CN112104212A - Three-phase two-switch rectifier filter inductor and hysteresis loop control switching frequency design method - Google Patents

Three-phase two-switch rectifier filter inductor and hysteresis loop control switching frequency design method Download PDF

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CN112104212A
CN112104212A CN202010965739.1A CN202010965739A CN112104212A CN 112104212 A CN112104212 A CN 112104212A CN 202010965739 A CN202010965739 A CN 202010965739A CN 112104212 A CN112104212 A CN 112104212A
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phase
current
pass filter
inductance
delay
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CN112104212B (en
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张镠钟
彭辉
王芳瑞
游江
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Harbin Engineering University
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/12—Arrangements for reducing harmonics from ac input or output

Abstract

The invention discloses a method for designing filter inductance and hysteresis control switching frequency of a three-phase two-switch rectifier, which comprises the following steps of 1, enabling each phase of inductance current at a converter side to be in a critical continuous state under a required load, and calculating inductance value L of each phase of inductance at the converter side; step 2, calculating the delay T of the low-pass filter for filtering the inductive currentd(ii) a Step 3, if the frequency is fs＝1/TsThe time delay of the signal passing through the second-order low-pass filter is TdCorresponding phase delay
Figure DDA0002682237610000011
Satisfies the following conditions:
Figure DDA0002682237610000012
according to a second-order low-pass filter, obtaining the angular frequency omegasOf the signal of
Figure DDA0002682237610000013
The natural resonant angular frequency ω of the second order low pass filter is calculatedn(ii) a And 4, step 4: switching frequency f of the actual converter with filter delaysNatural resonant angular frequency f using a low-pass filtern. The invention is beneficial to reducing the harmonic distortion of the current at the network side and improving the sine degree of the current at the network side; an engineering design method for switching frequency design under current hysteresis control is provided.

Description

Three-phase two-switch rectifier filter inductor and hysteresis loop control switching frequency design method
Technical Field
The invention relates to a method for designing the side filter inductance and hysteresis loop control switching frequency of a three-phase two-switch rectifier converter, belonging to the technical field of power electronics.
Background
Current hysteresis comparison control is typically used in systems where good current control is desired, and the actual switching frequency when hysteresis control is employed may not be easily controlled, subject to the hysteresis width, and too high a switching frequency may cause switching device damage. The invention provides a three-phase two-switch rectifier filter inductor and hysteresis loop control switching frequency design method. The three-phase two-switch rectifier has a good power factor correction function (namely, the current and the voltage on the power grid side are approximately sinusoidal) on the premise that the inductive current on the converter side needs to be interrupted or approximately in a critical continuous state (as the continuity of the inductive current on the converter side is enhanced, the current on the power grid side tends to be distorted).
Disclosure of Invention
In view of the above prior art, the technical problem to be solved by the present invention is to provide a method for ensuring that the current (approximate) critical continuity of the inverter side inductor is ensured, and a filter inductor and hysteresis loop control switching frequency design method of a three-phase two-switch rectifier based on the condition.
In order to solve the technical problem, the invention provides a method for designing the filter inductance and hysteresis loop control switching frequency of a three-phase two-switch rectifier, which comprises the following steps:
step 1: inverter-side individual-phase inductor current i12、i22And i32The critical continuous state is presented under the required load, and the inductance L of each phase at the converter side is calculated12、L22And L32Inductance value of, L12、L22And L32Have the same inductance value L, L satisfies:
Figure BDA0002682237590000011
wherein, TsFor a desired switching period, VoFor rating the voltage, V, on the DC sidecmIs the voltage peak of the AC side capacitor, ImFor each phase inductance L of the converter side12、L22And L32A peak value of a current fundamental component;
step 2: calculating for the inductor current i12、i22And i32Delay T of low-pass filter for filteringd；
And step 3: if the frequency is fs＝1/TsIs passed through a second order low pass filter
Figure BDA0002682237590000012
Has a time delay of TdCorresponding phase delay
Figure BDA0002682237590000013
Satisfies the following conditions:
Figure BDA0002682237590000014
ωsto correspond to angular frequency, omegas＝2πfsAccording to a second order low pass filter, the angular frequency is ωsOf the signal of
Figure BDA0002682237590000015
Simultaneously, the following requirements are met:
Figure BDA0002682237590000021
the natural resonant angular frequency ω of the second order low pass filter is calculatedn；
And 4, step 4: in at least one position of
Figure BDA0002682237590000022
Switching frequency f of the actual converter under filter delay conditionssUsing low-pass filters
Figure BDA0002682237590000023
Natural resonant angular frequency of
Figure BDA0002682237590000024
The invention also includes:
step 2TdSatisfies the following conditions:
Figure BDA0002682237590000025
the invention has the beneficial effects that: the method for calculating the inductance value of the converter side is used for realizing the interruption or critical continuity of the inductance current of the converter side of the three-phase two-switch rectifier, is favorable for reducing the harmonic distortion of the current of the network side and improving the sine degree of the current of the network side; an engineering design method for designing the switching frequency under the control of the current hysteresis is provided, so that the switching frequency under the control of the current hysteresis can be predicted and controlled, and the system loss analysis, the drive circuit design and the reliability improvement are facilitated.
Drawings
Fig. 1 is a three-phase two-switch rectifier topology circuit of the present invention.
FIG. 2 is a schematic diagram of a voltage-current dual closed-loop control structure according to the present invention.
FIG. 3 is a flow chart of the method of the present invention.
Detailed Description
The following further describes the embodiments of the present invention with reference to the drawings.
The invention aims to provide a method for designing the filter inductance and the hysteresis loop control switching frequency of a three-phase two-switch rectifier converter side. The three-phase two-switch rectifier has a good power factor correction function (i.e. the grid side current and voltage are approximately sinusoidal) on the premise that the inductor current of the converter side needs to be intermittent or approximately in a critical continuous state (as the continuity of the inductor current of the converter side is enhanced, the grid side current tends to be distorted). Current hysteresis comparison control is typically used in systems where good current control is desired, and the actual switching frequency when hysteresis control is employed may not be easily controlled, subject to the hysteresis width, and too high a switching frequency may result in high switching losses and cause switching device damage. In view of the above problems, the present invention provides a method for ensuring (approximate) critical continuity of the inductor current on the converter side, and a hysteresis loop control switching frequency design method based on the condition.
The invention comprises the following steps: the method comprises a converter side inductance value calculation method for ensuring the interruption or critical continuity of a converter side inductance current required by obtaining a low harmonic distortion network side current, and a converter side current hysteresis loop control switching frequency determination method based on the requirement.
With reference to fig. 1, the input side of the circuit is formed by L11, L21, L31, C1, C2, C3, L12, L22, and L32 to form a three-phase T-type filter network, and the midpoints of filter capacitors C1, C2, and C3 are connected to the midpoints of 2 series-connected switching tubes S1 and S2 and the midpoints of 2 series-connected capacitors Cd1 and Cd2 to form a three-level (positive, negative, and zero) structure. The circuit structure is shown in fig. 1. Because of the existence of the neutral line, the upper half bridge and the lower half bridge are mutually independent to form a partial decoupling foundation, and the voltage borne by the switching device is only 1/2 of the output voltage, so that the model selection requirement on the switching tube is reduced.
The three-phase two-switch rectifier and the voltage and current double closed-loop control structure thereof are mature technologies, in the invention, only the current loop controller inside the three-phase two-switch rectifier is replaced by a current hysteresis comparator, the change is very easy to understand for the relevant professional technicians, for comparison, a common system control block diagram is provided, and as shown in fig. 2, a double closed-loop control strategy is adopted, namely, a voltage outer loop and a current inner loop are combined. The task of the voltage outer ring is to sample output voltage and compare the output voltage with given value, the difference value is multiplied by the maximum (minimum) value of three-phase alternating voltage through PI regulation to be used as phase given value, then the maximum (minimum) value of actually input three-phase current is sampled, and the difference value of the two is compared with triangular carrier to generate a driving signal to drive the MOS tube. The MOS tubes of the upper bridge arm and the lower bridge arm are completely independent and do not influence each other. The benefits of such control are: each phase is optimally controlled to the maximum extent (within the interval of 2 pi/3), the control algorithm is simple, and the digital control method is adopted, so that the cost is low and the cost performance is high.
The applicant's studies have shown that for the three-phase two-switch shown in figure 1Turning off the rectifier when the current i of each phase of the converter side is12、i22And i32When the intermittent or critical continuous state is presented under the required load condition, the network side inductive current i11、i21And i31Has smaller harmonic distortion and can obtain approximate unit power factor performance.
The first embodiment is as follows:
the invention comprises the following steps:
(1) according to the inductive current i of each phase of the converter side12、i22And i32(theoretically steady state time i12、i22And i32Having approximately the same waveform except for 120 deg. phase difference) to exhibit a critical continuous state at the desired load, according to the requirements
Figure BDA0002682237590000031
The inductance value of each phase inverter side (inductance L of three inverter sides) was calculated12、L22And L32Have the same value L). Furthermore, TsFor a desired switching period, VoFor rating the voltage, V, on the DC sidecmIs an AC side capacitor (C)1、C2And C3) Voltage peak of (1)mFor each phase inductance L of the converter side12、L22And L32Peak value of the fundamental component of the current.
(2) According to
Figure BDA0002682237590000032
Calculating for the inductor current i12、i22And i32Delay of the filtered low pass filter (these three currents will be used for current hysteresis control).
(3) In a second order low-pass filter (e.g. taking
Figure BDA0002682237590000033
) Under the condition of the reaction, the reaction kettle is used for heating,
Figure BDA0002682237590000041
assuming a frequency fs＝1/Ts(corresponding to the converter switching frequency, corresponding to the angular frequency omegas＝2πfs) Has a time delay of T through a second-order low-pass filter as shown by (A1)d(corresponding phase delays of
Figure BDA0002682237590000042
And according to (A1), the angular frequency is ωsThe phase delay of the signal of (a2) is shown.
Figure BDA0002682237590000043
Then the natural resonant angular frequency ω of the corresponding second-order low-pass filter can be obtained from (a2)n(corresponding to the bandwidth of the second order low pass filter) to determine the filter parameters that meet the requirements.
(4) The designed inductance value according to the (1) is small, so that the converter side inductance current can be ensured to be approximately discontinuous under the required load condition, and the time required by the actual converter side inductance current to generate the current change quantity in the range limited by the loop width of the hysteresis comparator is relative to the filtering delay T of the (A1)dCan be ignored, and therefore, under the condition of having the (A1) filtering delay, the switching frequency of the practical converter and the natural resonant angular frequency of the low-pass filter (A1) can be enabled
Figure BDA0002682237590000044
Are similar, so that the switching frequency fs≈fn。
Example two:
with reference to fig. 1 to 3, the design method of the present invention is implemented as follows:
(1) according to the inductive current i of each phase of the converter side12、i22And i32(theoretically steady state time i12、i22And i32Having approximately the same waveform except for 120 ° phase difference) to exhibit a critical continuous state at the required load, voltage phase at unity power factor isInductor current at 90 °:
Figure BDA0002682237590000045
in the formula ImThe peak value of the average value of the phase currents.
The loop current equation in the inductor is:
Figure BDA0002682237590000046
in the formula, VcmIs the peak value of the capacitance voltage at the AC input side.
The peak value of the average value of the phase currents is:
Figure BDA0002682237590000047
according to fig. 1, the dc side can be regarded as two Boost converters connected in series, the two dc capacitors are equal in voltage,
Figure BDA0002682237590000051
obtaining:
Figure BDA0002682237590000052
the simultaneous expressions (1) to (4) give:
Figure BDA0002682237590000053
in the formula, TsFor a desired switching period, Po＝3VcmIm/2 is the output power of the rectifier, VoFor rating the voltage, V, on the DC sidecmIs an AC side capacitor (C)1、C2And C3，C1、C2AndC3equal capacitance) of the voltage peak, ImFor each phase inductance L of the converter side12、L22And L32Peak value of the fundamental component of the current.
(2) According to formula (6)
Figure BDA0002682237590000054
Calculating for the inductor current i12、i22And i32The filtered low pass filter delays (these three currents will be used for current hysteresis control).
(3) In a second order low-pass filter (e.g. with a desired damping ratio)
Figure BDA0002682237590000058
) Under the condition of the reaction, the reaction kettle is used for heating,
Figure BDA0002682237590000055
if the frequency is fs＝1/Ts(corresponding to the converter switching frequency, corresponding to the angular frequency omegas＝2πfs) The time delay of the signal passing through the second-order low-pass filter shown in the formula (7) is Td(corresponding phase delays of
Figure BDA0002682237590000056
And according to equation (7), the angular frequency is ωsThe phase delay of the signal of (2) is shown as (8).
Figure BDA0002682237590000057
Then the natural resonant angular frequency ω of the corresponding second-order low-pass filter can be calculated according to equation (8)n(corresponding to the bandwidth of the second order low pass filter) to determine the filter parameters that meet the requirements.
(4) The designed inductance value according to the step (1) is small, so that the inductance current at the converter side can be ensured under the condition of required loadApproximately intermittent, the time required for the actual converter side inductor current to generate a current variation in a range defined by the loop width of the hysteresis comparator is relative to the filter delay T given in (3)dIs negligible, and therefore, with the filter delay shown in equation (3), the switching frequency of the actual converter and the natural resonant angular frequency of the low-pass filter equation (3) will be made
Figure BDA0002682237590000061
Are similar and therefore have a switching frequency fs≈fn。

Claims (2)

1. A method for designing filter inductance and hysteresis loop control switching frequency of a three-phase two-switch rectifier is characterized by comprising the following steps:
step 1: inverter-side individual-phase inductor current i12、i22And i32The critical continuous state is presented under the required load, and the inductance L of each phase at the converter side is calculated12、L22And L32Inductance value of, L12、L22And L32Have the same inductance value L, L satisfies:
Figure FDA0002682237580000011
wherein, TsFor a desired switching period, VoFor rating the voltage, V, on the DC sidecmIs the voltage peak of the AC side capacitor, ImFor each phase inductance L of the converter side12、L22And L32A peak value of a current fundamental component;
step 2: calculating for the inductor current i12、i22And i32Delay T of low-pass filter for filteringd。
And step 3: if the frequency is fs＝1/TsIs passed through a second order low pass filter
Figure FDA0002682237580000012
Has a time delay of TdCorresponding phase delay
Figure FDA0002682237580000019
Satisfies the following conditions:
Figure FDA0002682237580000013
ωsto correspond to angular frequency, omegas＝2πfsAccording to a second order low pass filter, the angular frequency is ωsOf the signal of
Figure FDA0002682237580000014
Simultaneously, the following requirements are met:
Figure FDA0002682237580000015
the natural resonant angular frequency ω of the second order low pass filter is calculatedn。
And 4, step 4: in at least one position of
Figure FDA0002682237580000016
Switching frequency f of the actual converter under filter delay conditionssUsing low-pass filters
Figure FDA0002682237580000017
Natural resonant angular frequency of
Figure FDA0002682237580000018
2. The method of claim 1, wherein the method comprises the steps of: step 2 said TdSatisfies the following conditions:
Figure FDA0002682237580000021
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