CN111917677A - CPFSK demodulation baseband implementation method - Google Patents

CPFSK demodulation baseband implementation method Download PDF

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CN111917677A
CN111917677A CN202010838086.0A CN202010838086A CN111917677A CN 111917677 A CN111917677 A CN 111917677A CN 202010838086 A CN202010838086 A CN 202010838086A CN 111917677 A CN111917677 A CN 111917677A
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不公告发明人
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Beijing Rongwei Technology Co ltd
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    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/18—Phase-modulated carrier systems, i.e. using phase-shift keying
    • H04L27/22—Demodulator circuits; Receiver circuits
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/18—Phase-modulated carrier systems, i.e. using phase-shift keying
    • H04L27/20—Modulator circuits; Transmitter circuits
    • H04L27/2003—Modulator circuits; Transmitter circuits for continuous phase modulation

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Abstract

The invention discloses a CPFSK demodulation baseband realization method, which relates to the technical field of communication modulation and comprises the following steps: acquiring a CPFSK signal as an intermediate frequency signal IF, determining to perform operation on the length of N-2N +1 symbols, and judging the nth symbol in the middle; acquiring CPFSK signals on N symbol lengths, calibrating the polarity of a first symbol, and acquiring the initial phase of a carrier of an ith symbol on the current length of the first symbol; performing integral operation on the obtained transmission symbol sequence A to obtain a middle symbol an +1 as the decision of the middle symbol of the received waveform; the two paths of signals of the acquired normalized signal pass through a low-pass filter respectively to obtain an in-phase component and a positive-phase component. The invention has strong anti-interference capability on the distortion of amplitude and phase caused by multipath fading, simple method, low cost, high-speed telemetering demodulation, obvious reduction of bit error rate and demodulation threshold, and wide application range.

Description

CPFSK demodulation baseband implementation method
Technical Field
The invention relates to the technical field of communication modulation, in particular to a CPFSK demodulation baseband implementation method.
Background
Modulation plays a very important role in communication systems. The role of the modulation is to shift the spectrum of the modulated information from low frequencies to high frequencies to suit the channel transmission. Correspondingly, demodulation is needed at the receiving end to recover the modulated information, so that demodulation is an important technology affecting the performance of the communication system.
The demodulation method is divided into two types, non-coherent demodulation and coherent demodulation according to the standard of whether carrier recovery is needed or not. Coherent demodulation extracts and recovers carrier information through a phase-locked loop during demodulation, and demodulates modulation information through the recovered carrier signal and an input signal. Non-coherent demodulation, as the name implies, does not require carrier information extraction for demodulation. Coherent demodulation is widely used because coherent demodulation performance is better than non-coherent demodulation in terms of demodulation performance, in which a phase-locked loop is a key component in a coherent demodulation system. In terms of hardware implementation complexity, coherent demodulation requires extraction of coherent carriers, so that a circuit is complex and implementation difficulty is high; accordingly, the non-coherent demodulation circuit is easy to implement, and therefore, the non-coherent demodulation circuit is also practically applied to many occasions.
PCM/FM is a widely used target range telemetry scheme. Two new telemetry schemes, Multi-h CPM (Multi-modulation index continuous phase modulation) and FQPSK (Feher-based QPSK) are proposed. At present, in the PCM/FM telemetering system in China, the PCM code rate is low and can only reach 2Mbps at most. With the practical requirements of range telemetry, the 2Mbps code rate bottleneck has not been able to meet the transmission requirements of high rate PCM data.
Therefore, a CPFSK demodulation baseband implementation method is needed.
An effective solution to the problems in the related art has not been proposed yet.
Disclosure of Invention
Aiming at the problems in the related art, the invention provides a CPFSK demodulation baseband implementation method to overcome the technical problems in the prior related art.
The technical scheme of the invention is realized as follows:
a CPFSK demodulation baseband implementation method comprises the following steps:
step S1, acquiring the CPFSK signal as an intermediate frequency signal IF in advance, determining that the calculation is performed on N ═ 2N +1 symbol lengths, and determining an nth symbol in the middle;
step S2, collecting CPFSK signals in the length of N symbols, calibrating the polarity of the first symbol, and obtaining the initial phase of the carrier of the ith symbol in the current length;
step S3, performing an integration operation on the acquired transmission symbol sequence a, where the likelihood ratio is expressed as:
Figure BDA0002640410030000021
wherein m is 22n，θ1Is a random variable phase;
step S4, obtaining a middle symbol an +1 as the judgment of the middle symbol of the received waveform;
step S5, a normalized signal is obtained, expressed as:
r(t)＝cos(ωot+Δω∫g(τ)dτ+θ1)
and step S6, the two paths of signals pass through low-pass filters respectively to obtain in-phase and positive-phase components.
Further, the nth symbol in the middle of the decision is expressed as:
the signal scaled at the first symbol length is represented as:
Figure BDA0002640410030000022
wherein, (2P)1/2Is the amplitude of the signal; a1 is the value of the first symbol, and takes-1 or 1; theta 1 is the initial phase of the intermediate frequency signal s (t) and takes the values of phi and phi](ii) a h is the modulation index and T is the symbol period.
Further, the CPFSK is included as a phase continuous phase, and the signal represented by the ith symbol length is:
Figure BDA0002640410030000023
where ai is the value of the ith symbol.
Further, the calibrating the polarity of the first symbol includes the following steps:
signals over the acquisition length are acquired, expressed as: s (t, an +1, Ak, θ i);
wherein, Ak is a2 n-dimensional array { a1, a 2., an, an + 2. + a2n +1 }; an +1 is +1 or-1;
acquiring all possible transmission waveforms in the observation length, and calculating the waveforms and the received signals t (r);
the polarity of the (n +1) th symbol is determined, and the likelihood ratio l is calibrated and expressed as:
Figure BDA0002640410030000031
where, r (t) is s (t) + N (t), and t (N) is gaussian noise with bilateral power spectral density of 2N 0W/Hz.
Further, the method comprises the following steps:
calibrating the phase characteristic of the random variable, and expressing as:
Figure BDA0002640410030000032
where the signals are positive and in-phase components, the signals are represented as:
Figure BDA0002640410030000033
which is characterized in that the material is a mixture of,
Figure BDA0002640410030000034
is the phase of the continuous phase wave;
a scaled complex signal, expressed as:
z±1i2＝2P|∫r(t)exp j[ω0t+φi(t)]dt|2
further, the acquiring the normalized signal further includes the following steps:
a scaled complex signal, expressed as:
R(t)＝cos(Δω∫g(τ)dτ+θ1)-j sin(Δω∫g(τ)dτ+θ1)
the locally stored waveform over the observation interval is scaled, as:
S(t)＝cos(Δω∫g(τ)dτ+θ2)+j sin(Δω∫g(τ)dτ+θ2)
scaling the local waveform with the received signal operation, expressed as:
R(t)·S(t)
＝cos(θ2-θ1)+j sin(θ2-θ1)
＝IB+jQB。
further, the two signals respectively pass through a low-pass filter to obtain in-phase and positive-phase components, which are expressed as:
the number of samples within the length of acquiring the whole observation interval N-5 is NUM, and the integral value thereof is expressed as:
M＝NUM·(IB_sample+jQB_sample)
wherein t (n)
IB_sampleIs IBThe sampling value of (2); qB_sampleIs QBThe sampling value of (2);
the perfect square value, expressed as:
Figure BDA0002640410030000041
the invention has the beneficial effects that:
the invention provides a CPFSK demodulation baseband realization method, which comprises the steps of obtaining a CPFSK signal as an intermediate frequency signal IF in advance, determining to perform operation on the length of N-2N +1 symbols, and judging the nth symbol in the middle; acquiring CPFSK signals on N symbol lengths, calibrating the polarity of a first symbol, and acquiring the initial phase of a carrier of an ith symbol on the current length of the first symbol; performing integral operation on the obtained transmission symbol sequence A to obtain a middle symbol an +1 as the decision of the middle symbol of the received waveform; the two paths of signals of the acquired normalized signal respectively pass through the low-pass filter to obtain the in-phase component and the positive-phase component, so that the high-speed remote sensing demodulation has strong anti-interference capability on the amplitude and phase distortion caused by multipath fading, the method is simple, the cost is low, the high-speed remote sensing demodulation can obviously reduce the error rate and the demodulation threshold, and the application range is wide.
Drawings
In order to more clearly illustrate the embodiments of the present invention or the technical solutions in the prior art, the drawings needed in the embodiments will be briefly described below, and it is obvious that the drawings in the following description are only some embodiments of the present invention, and it is obvious for those skilled in the art to obtain other drawings without creative efforts.
Fig. 1 is a schematic structural diagram of a CPFSK demodulation baseband implementation method according to an embodiment of the present invention;
fig. 2 is a schematic block diagram of an MSD algorithm of a CPFSK demodulation baseband implementation method according to an embodiment of the present invention;
fig. 3 is a schematic diagram illustrating a Z value calculation flow of a CPFSK demodulation baseband implementation method according to an embodiment of the present invention.
Detailed Description
The technical solutions in the embodiments of the present invention will be clearly and completely described below with reference to the drawings in the embodiments of the present invention, and it is obvious that the described embodiments are only a part of the embodiments of the present invention, and not all of the embodiments. All other embodiments that can be derived by one of ordinary skill in the art from the embodiments given herein are intended to be within the scope of the present invention.
According to the embodiment of the invention, a CPFSK demodulation baseband realization method is provided.
As shown in fig. 1 to fig. 3, the CPFSK demodulation baseband implementation method according to the embodiment of the present invention includes the following steps:
step S1, acquiring the CPFSK signal as an intermediate frequency signal IF in advance, determining that the calculation is performed on N ═ 2N +1 symbol lengths, and determining an nth symbol in the middle;
step S2, collecting CPFSK signals in the length of N symbols, calibrating the polarity of the first symbol, and obtaining the initial phase of the carrier of the ith symbol in the current length;
step S3, performing an integration operation on the acquired transmission symbol sequence a, where the likelihood ratio is expressed as:
Figure BDA0002640410030000051
wherein m is 22n，θ1Is a random variable phase;
step S4, obtaining a middle symbol an +1 as the judgment of the middle symbol of the received waveform;
step S5, a normalized signal is obtained, expressed as:
r(t)＝cos(ωot+Δω∫g(τ)dτ+θ1)
and step S6, the two paths of signals pass through low-pass filters respectively to obtain in-phase and positive-phase components.
Further, the nth symbol in the middle of the decision is expressed as:
the signal scaled at the first symbol length is represented as:
Figure BDA0002640410030000061
wherein, (2P)1/2Is the amplitude of the signal; a1 is the value of the first symbol, and takes-1 or 1; theta 1 is the initial phase of the intermediate frequency signal s (t) and takes the values of phi and phi](ii) a h is the modulation index and T is the symbol period.
Further, the CPFSK is included as a phase continuous phase, and the signal represented by the ith symbol length is:
Figure BDA0002640410030000062
where ai is the value of the ith symbol.
Further, the calibrating the polarity of the first symbol includes the following steps:
signals over the acquisition length are acquired, expressed as: s (t, an +1, Ak, θ i);
wherein, Ak is a2 n-dimensional array { a1, a 2., an, an + 2. + a2n +1 }; an +1 is +1 or-1;
acquiring all possible transmission waveforms in the observation length, and calculating the waveforms and the received signals t (r);
the polarity of the (n +1) th symbol is determined, the likelihood ratio is calibrated to be 1, and the method is expressed as:
Figure BDA0002640410030000063
where, r (t) is s (t) + N (t), and t (N) is gaussian noise with bilateral power spectral density of 2N 0W/Hz.
Further, the method comprises the following steps:
calibrating the phase characteristic of the random variable, and expressing as:
Figure BDA0002640410030000064
where the signals are positive and in-phase components, the signals are represented as:
Figure BDA0002640410030000071
which is characterized in that the material is a mixture of,
Figure BDA0002640410030000072
is the phase of the continuous phase wave;
a scaled complex signal, expressed as:
z±1i2＝2P|∫r(t)exp j[ω0t+φi(t)]dt|2
further, the acquiring the normalized signal further includes the following steps:
a scaled complex signal, expressed as:
R(t)＝cos(Δω∫g(τ)dτ+θ1)-j sin(Δω∫g(τ)dτ+θ1)
the locally stored waveform over the observation interval is scaled, as:
S(t)＝cos(Δω∫g(τ)dτ+θ2)+j sin(Δω∫g(τ)dτ+θ2)
scaling the local waveform with the received signal operation, expressed as:
R(t)·S(t)
＝cos(θ2-θ1)+j sin(θ2-θ1)
＝IB+jQB。
further, the two signals respectively pass through a low-pass filter to obtain in-phase and positive-phase components, which are expressed as:
the number of samples within the length of acquiring the whole observation interval N-5 is NUM, and the integral value thereof is expressed as:
M＝NUM·(IB_sample+jQB_sample)
wherein t (n)
IB_sampleIs IBThe sampling value of (2); qB_sampleIs QBThe sampling value of (2);
the perfect square value, expressed as:
Figure BDA0002640410030000073
by means of the scheme, the CPFSK signal is acquired in advance as an intermediate frequency signal IF, operation is determined to be carried out on the length of N-2N +1 symbols, and the nth symbol in the middle is judged; acquiring CPFSK signals on N symbol lengths, calibrating the polarity of a first symbol, and acquiring the initial phase of a carrier of an ith symbol on the current length of the first symbol; performing integral operation on the obtained transmission symbol sequence A to obtain a middle symbol an +1 as the decision of the middle symbol of the received waveform; the two paths of signals of the acquired normalized signal respectively pass through the low-pass filter to obtain the in-phase component and the positive-phase component, so that the high anti-interference capability on the distortion in the amplitude and phase caused by multipath fading is realized, the method is simple, the cost is low, the high-speed and high-speed telemetering demodulation is realized, and the application range is wide.
In addition, specifically, the received signal t (r) is correlated with the positive and in-phase components of all possible transmitted signals t(s). And adding correlation values of all the intermediate symbols an + 1-1, comparing the correlation values and selecting the intermediate symbol an +1 with the largest added value as the decision of the intermediate symbol of the received waveform.
Since m 22N 2N-1 and an +1 equals either +1 or-1, there is a local need to generate both positive and same phase components of possibly all possible transmitted waveforms. So that 2N +1 waveforms need to be stored locally for correlation with the received signal t (r) over N symbol lengths.
In addition, the correlation value between the received baseband complex signal and the locally stored waveform containing the same PCM sequence over the length of the observation interval. Similarly, the baseband complex signal is operated on all other locally stored waveforms containing other PCM sequences, whose perfect squared values are smaller than the Z value, which makes a decision on the received signal possible.
In addition, the specific work flow of the baseband complex signal angle rotation MSD demodulator is as follows:
1) before demodulation, 8 kinds of local complex signal waveforms in one symbol period are stored in a ROM of an FPGA, and each kind of local complex signal waveform comprises two kinds of component waveforms of a same phase and a positive phase.
2) After A/D sampling is carried out on the received PCM/FM, digital down-conversion is carried out to obtain I, Q two paths of signals, and the two paths of signals are combined into a path of complex signal.
3) On the basis of symbol synchronization, in a symbol period, a path of complex signals obtained after digital down-conversion and 8 kinds of locally stored complex signals are subjected to correlation operation, and the obtained correlation values are still complex signals.
4) And performing angle rotation operation on the correlation value complex signal obtained in the last step by using a CORDIC algorithm, wherein the rotated angle value is determined according to the correlation condition among 128 possibly received symbol sequences.
5) The value after the previous angular rotation is fed into a 5-stage delay pipeline.
6) The data in the 5-stage delay pipeline is organically combined and added according to the correlation between the symbols of the possible transmission sequences according to the correlation condition between 128 symbol sequences, and then a complete square value is obtained.
7) The perfect square values obtained in the last step are averagely divided into two types according to the polarity (-1 or +1) of the intermediate symbols of 128 transmission sequences, 64 perfect square values belonging to the same type are added, and then the two added values are subtracted. The result is used as the basis for judging the polarity of the middle symbol of the received sequence.
Compared with the traditional frequency discrimination algorithm, the MSD algorithm has better performance, and can obviously reduce the error rate and the demodulation threshold because of utilizing the characteristic of the continuous phase of the PCM/FM signal. At a bit error rate of 10-4Under the condition, the Eb/N0 can be reduced by about 3d B.
In summary, according to the above technical solution of the present invention, the CPFSK signal is obtained in advance as the intermediate frequency signal IF, and it is determined that the calculation is performed on N ═ 2N +1 symbol lengths, and the nth symbol in the middle is determined; acquiring CPFSK signals on N symbol lengths, calibrating the polarity of a first symbol, and acquiring the initial phase of a carrier of an ith symbol on the current length of the first symbol; performing integral operation on the obtained transmission symbol sequence A to obtain a middle symbol an +1 as the decision of the middle symbol of the received waveform; the two paths of signals of the acquired normalized signal respectively pass through the low-pass filter to obtain the in-phase component and the positive-phase component, so that the high anti-interference capability on the distortion in the amplitude and phase caused by multipath fading is realized, the method is simple, the cost is low, the high-speed and high-speed telemetering demodulation is realized, and the application range is wide.
The above description is only for the purpose of illustrating the preferred embodiments of the present invention and is not to be construed as limiting the invention, and any modifications, equivalents, improvements and the like that fall within the spirit and principle of the present invention are intended to be included therein.

Claims (7)

1. A CPFSK demodulation baseband realization method is characterized by comprising the following steps:
acquiring a CPFSK signal as an intermediate frequency signal IF in advance, determining to perform operation on the length of N-2N +1 symbols, and judging an nth symbol in the middle;
acquiring CPFSK signals on N symbol lengths, calibrating the polarity of a first symbol, and acquiring the initial phase of a carrier of an ith symbol on the current length of the first symbol;
and performing integral operation on the acquired transmission symbol sequence A, wherein the likelihood ratio is expressed as:
Figure FDA0002640410020000011
wherein m is 22n，θ1Is a random variable phase;
obtaining an intermediate symbol an +1 as a decision of a received waveform intermediate symbol;
a normalized signal is obtained, expressed as:
r(t)＝cos(ωot+Δω∫g(τ)dτ+θ1)
the two paths of signals respectively pass through a low-pass filter to obtain in-phase and positive-phase components.
2. The CPFSK demodulation baseband implementation method of claim 1, wherein said decision of the middle nth symbol is expressed as:
the signal scaled at the first symbol length is represented as:
Figure FDA0002640410020000012
wherein, (2P)1/2Is the amplitude of the signal; a1 is the value of the first symbol, and takes-1 or 1; theta 1 is the initial phase of the intermediate frequency signal s (t) and takes the values of phi and phi](ii) a h is the modulation index and T is the symbol period.
3. The CPFSK demodulation baseband implementation method of claim 2, further comprising the CPFSK being phase continuous phase, wherein the ith symbol length represents the signal:
Figure FDA0002640410020000013
where ai is the value of the ith symbol.
4. The CPFSK demodulation baseband implementation method of claim 3, wherein said calibrating the polarity of the first symbol comprises the steps of:
signals over the acquisition length are acquired, expressed as: s (t, an +1, Ak, θ i);
wherein, Ak is a2 n-dimensional array { a1, a 2., an, an + 2. + a2n +1 }; an +1 is +1 or-1;
acquiring all possible transmission waveforms in the observation length, and calculating the waveforms and the received signals t (r);
the polarity of the (n +1) th symbol is determined, the likelihood ratio is calibrated to be 1, and the method is expressed as:
Figure FDA0002640410020000021
where, r (t) is s (t) + N (t), and t (N) is gaussian noise with bilateral power spectral density of 2N 0W/Hz.
5. The CPFSK demodulation baseband implementation method of claim 4, wherein the likelihood ratio further comprises the steps of:
calibrating the phase characteristic of the random variable, and expressing as:
Figure FDA0002640410020000022
where the signals are positive and in-phase components, the signals are represented as:
Figure FDA0002640410020000023
which is characterized in that the material is a mixture of,
Figure FDA0002640410020000024
is the phase of the continuous phase wave;
a scaled complex signal, expressed as:
z±1i2＝2P|∫r(t)exp j[ωot+φi(t)]dt|2
6. the CPFSK demodulation baseband implementation method according to claim 5, wherein said obtaining a normalized signal further comprises the steps of:
a scaled complex signal, expressed as:
R(t)＝cos(Δω∫g(τ)dτ+θ1)-j sinΔω∫g(τ)dτ+θ1)
the locally stored waveform over the observation interval is scaled, as:
S(t)＝cos(Δω∫g(τ)dτ+θ2)+j sin(Δω∫g(τ)dτ+θ2)
scaling the local waveform with the received signal operation, expressed as:
R(t)·S(t)
＝cos(θ2-θ1)+j sin(θ2-θ1)
＝IB+jQB。
7. the CPFSK demodulation baseband implementation method of claim 6, wherein the two signals respectively pass through a low-pass filter to obtain in-phase and positive-phase components, which are expressed as:
the number of samples within the length of acquiring the whole observation interval N-5 is NUM, and the integral value thereof is expressed as:
M＝NUM·(IB_sample+jQB_sample)
wherein t (n)
IB_sampleIs IBThe sampling value of (2); qB_sampleIs QBThe sampling value of (2);
the perfect square value, expressed as:
Z＝NUM2·(IB-sample 2+QB-sample 2)。
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Citations (3)

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Publication number Priority date Publication date Assignee Title
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* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JPH04362834A (en) * 1991-06-10 1992-12-15 Nec Corp Cpfsk modulation circuit and cpfsk demodulation circuit
CN105024962A (en) * 2015-05-25 2015-11-04 中国电子科技集团公司第十研究所 GMSK signal low-complexity coherent demodulation method
CN109039966A (en) * 2018-08-01 2018-12-18 上海华虹集成电路有限责任公司 A kind of demodulation method based on decision-feedback, the GFSK signal of low complex degree

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Title
S.P.MAJUMDER等: "Effect of Polarization Mode Dispersion on Optical Heterodyne CPFSK System", 《GLOBECOM "00 - IEEE. GLOBAL TELECOMMUNICATIONS CONFERENCE. CONFERENCE RECORD (CAT. NO.00CH37137)》 *
孔冬: "高速PCM/FM遥测解调技术研究", 《中国优秀硕士学位论文全文数据库信息科技辑(月刊)2013年第01期》 *
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Application publication date: 20201110

RJ01 Rejection of invention patent application after publication