CN111769615A

CN111769615A - Operational amplifier power supply device, power supply method and battery management system

Info

Publication number: CN111769615A
Application number: CN202010658178.0A
Authority: CN
Inventors: 周号
Original assignee: Zhuhai Maiju Microelectronics Co Ltd
Current assignee: Zhuhai Maiju Microelectronics Co Ltd
Priority date: 2020-07-09
Filing date: 2020-07-09
Publication date: 2020-10-13
Anticipated expiration: 2040-07-09
Also published as: CN111769615B

Abstract

The utility model provides an operational amplifier power supply unit for battery voltage monitoring devices, battery voltage monitoring devices are used for monitoring the voltage of each battery of group battery, and the group battery includes a plurality of batteries of positive negative pole series connection, and the negative pole of first battery is as the negative pole end of group battery and is connected to the earthing terminal, and the positive pole of the N battery is as the positive pole end of group battery, include: an operational amplifier that generates a control signal that samples a battery voltage of each of the plurality of batteries based on the battery voltage in the battery pack, the operational amplifier including a power supply positive terminal and a power supply negative terminal; and a level shifter generating a first voltage and a second voltage based on a voltage difference between both ends of the ith battery, wherein a power supply voltage of a power supply positive terminal of the operational amplifier is generated based on the first voltage, and a power supply voltage of a power supply negative terminal of the operational amplifier is generated based on the second voltage. The present disclosure also provides a battery management system.

Description

Operational amplifier power supply device, power supply method and battery management system

Technical Field

The present disclosure relates to the field of battery management, and in particular, to an operational amplifier power supply device, a power supply method, and a battery management system.

Background

In the battery system, overcharge and overdischarge of the battery not only reduce the lifespan of the battery, but also may cause safety accidents such as explosion and fire in severe cases.

Assuming that the voltage difference between the positive electrode and the negative electrode of a single battery is V (bat), the voltages to the ground of the positive electrode and the negative electrode of each battery of the high-voltage series battery pack are respectively n × V (bat), (n-1) × V (bat), … 2 × V (bat), because the battery pack is formed by connecting n batteries in series, usually V (bat) is about 3.7V, and n is 2-100, the absolute voltage of the positive electrode and the negative electrode of any battery to the ground (AGND) is very high, about 10-400V, so that a high-voltage resistant device is needed to collect voltage signals, and the high-voltage resistant device usually needs a special high-voltage semiconductor manufacturing process and a larger chip, which increases the cost of the chip, such as the cost of an operational amplifier.

In addition, because a high voltage easily causes parasitic diode leakage to affect measurement accuracy, such a high absolute voltage also causes limited sampling accuracy.

The high voltage can also cause reliability problems, because the acquisition switch is always exposed to the high voltage, gate hot carrier injection is easily generated, so that the threshold voltage fails, or the source and drain voltage-resistant drift regions are connected in series, and finally, the reliability problems such as system short circuit or complete failure of the sampling circuit are caused.

Disclosure of Invention

In order to solve at least one of the above technical problems, the present disclosure provides an operational amplifier power supply apparatus, a power supply method, and a battery management system.

According to an aspect of the present disclosure, an operational amplifier power supply device for a battery voltage monitoring device for monitoring voltages of respective batteries of a battery pack including a plurality of batteries whose positive and negative poles are connected in series, the plurality of batteries being first to nth batteries, a negative pole of the first battery serving as a negative terminal of the battery pack and being connected to a ground terminal, and a positive pole of the nth battery serving as a positive terminal of the battery pack, where N is an integer of 2 or more, includes:

an operational amplifier that generates a control signal that samples a battery voltage of each of a plurality of batteries based on a battery voltage in the battery pack, the operational amplifier including a power supply positive terminal and a power supply negative terminal; and

a level shifter generating a first voltage and a second voltage based on a voltage difference across an ith battery, where i is greater than or equal to 2 and less than or equal to N,

wherein a power supply voltage of a power supply positive terminal of the operational amplifier is generated based on the first voltage, and a power supply voltage of a power supply negative terminal of the operational amplifier is generated based on the second voltage.

According to one embodiment of the present disclosure, the level shifter generates the first voltage and the second voltage based on a difference between both ends voltage of an i-th battery and a difference between both ends voltage of an i + 1-th battery.

According to one embodiment of the present disclosure, the battery pack further comprises an N-type MOSFET, a drain of the N-type MOSFET is connected to a voltage of a positive terminal of the battery pack, a source of the N-type MOSFET is connected to a positive power supply terminal of the operational amplifier, and a gate of the N-type MOSFET is connected to the first voltage.

According to one embodiment of the present disclosure, the power supply system further comprises a P-type MOSFET, wherein a source of the P-type MOSFET is connected to a negative power supply terminal of the operational amplifier, a drain of the P-type MOSFET is connected to a ground voltage, and a gate of the P-type MOSFET is connected to the second voltage.

According to one embodiment of the present disclosure, the power supply device further comprises a second N-type MOSFET and a current source, wherein a source of the second N-type MOSFET is connected to a power supply negative terminal of the operational amplifier and one end of the current source, the other end of the current source is grounded, a drain of the second N-type MOSFET is connected to a voltage of a positive terminal of the battery pack, and a gate of the second N-type MOSFET is connected to the second voltage.

According to one embodiment of the present disclosure, the power supply further comprises a gain amplifier, wherein a positive input terminal of the gain amplifier is connected to the second voltage, and a negative input terminal of the gain amplifier is connected to an output terminal of the gain amplifier, and an output terminal of the gain amplifier is connected to a negative power supply terminal of the operational amplifier.

According to another aspect of the present disclosure, an operational amplifier power supply method for a battery voltage monitoring device for monitoring voltages of respective batteries of a battery pack including a plurality of batteries whose positive and negative poles are connected in series, the plurality of batteries being first to nth batteries, a negative pole of the first battery serving as a negative terminal of the battery pack and being connected to a ground terminal, and a positive pole of the nth battery serving as a positive terminal of the battery pack, where N is an integer equal to or greater than 2, the operational amplifier generating a control signal for sampling a battery voltage of each of the plurality of batteries based on the battery voltages in the battery pack, the operational amplifier including a positive power supply terminal and a negative power supply terminal, the method comprising:

generating a first voltage and a second voltage based on a voltage difference between two ends of the ith battery, wherein i is more than or equal to 2 and less than or equal to N; and

generating a supply voltage for a supply positive terminal of the operational amplifier based on the first voltage and generating a supply voltage for a supply negative terminal of the operational amplifier based on the second voltage.

According to one embodiment of the present disclosure, the first voltage and the second voltage are generated based on a voltage difference across an i-th battery and a voltage difference across an i + 1-th battery.

According to one embodiment of the present disclosure, a drain of the N-type MOSFET is connected to a voltage of a positive terminal of the battery pack, a source of the N-type MOSFET is connected to a positive power supply terminal of the operational amplifier, and a gate of the N-type MOSFET is connected to the first voltage, the power supply voltage of the positive power supply terminal of the operational amplifier is a difference between the first voltage and a gate-source voltage of the N-type MOSFET.

According to another aspect of the present disclosure, a battery management system includes: the battery voltage monitoring device is used for monitoring the voltage of each battery of the battery pack.

Drawings

The accompanying drawings, which are included to provide a further understanding of the disclosure and are incorporated in and constitute a part of this specification, illustrate exemplary embodiments of the disclosure and together with the description serve to explain the principles of the disclosure.

Fig. 1 shows a schematic diagram of a battery voltage monitoring device according to one embodiment of the present disclosure.

Fig. 2 shows a schematic diagram of a transconductance cell, according to one embodiment of the present disclosure.

Fig. 3 shows a schematic diagram of a voltage conversion unit according to an embodiment of the present disclosure.

Fig. 4 shows a control signal diagram according to an embodiment of the present disclosure.

Fig. 5 shows a schematic diagram of a battery voltage monitoring device according to one embodiment of the present disclosure.

Fig. 6 shows a circuit diagram of a current compensation approach according to another embodiment of the present disclosure.

Fig. 7 shows a partial enlarged view of fig. 6.

Fig. 8 shows a partial enlarged view of fig. 6.

Fig. 9 shows a voltage acquisition module schematic according to one embodiment of the present disclosure.

Fig. 10 shows a circuit diagram of a conventional operational amplifier.

Fig. 11 shows an operational amplifier circuit diagram according to one embodiment of the present disclosure.

Figure 12 shows a high output resistance current source powered operational amplifier architecture diagram according to one embodiment of the present disclosure.

Figure 13 illustrates a high output resistance current source powered operational amplifier circuit diagram according to one embodiment of the present disclosure.

FIG. 14 shows a P-N dual source follower powered operational amplifier architecture diagram according to one embodiment of the present disclosure.

Fig. 15 shows a circuit diagram of a VSS generator according to one embodiment of the present disclosure.

Fig. 16 shows a schematic diagram of an operational amplifier and a level shifter according to one embodiment of the present disclosure.

FIG. 17 illustrates a circuit diagram of a P-N dual source follower powered operational amplifier according to one embodiment of the present disclosure.

Fig. 18 shows a schematic diagram of a level shifter according to an embodiment of the present disclosure.

FIG. 19 shows an operational amplifier architecture diagram for an N-N dual source follower supply, according to one embodiment of the present disclosure.

Figure 20 shows an operational amplifier architecture diagram for an N-type source follower and gain amplifier supply according to one embodiment of the present disclosure.

Fig. 21 shows a schematic diagram of a voltage conversion unit according to an embodiment of the present disclosure.

Fig. 22 shows a circuit diagram of an operational amplifier according to one embodiment of the present disclosure.

Fig. 23 shows a circuit diagram of an operational amplifier according to one embodiment of the present disclosure.

Fig. 24 shows a battery management system schematic according to an embodiment of the present disclosure.

Detailed Description

The present disclosure will be described in further detail with reference to the drawings and embodiments. It is to be understood that the specific embodiments described herein are for purposes of illustration only and are not to be construed as limitations of the present disclosure. It should be further noted that, for the convenience of description, only the portions relevant to the present disclosure are shown in the drawings.

It should be noted that the embodiments and features of the embodiments in the present disclosure may be combined with each other without conflict. Technical solutions of the present disclosure will be described in detail below with reference to the accompanying drawings in conjunction with embodiments.

Fig. 1 provides a battery voltage monitoring device for monitoring the voltages of battery packs BAT 1-BATn. The battery pack includes a plurality of batteries of which positive and negative electrodes are connected in series, the plurality of batteries are a first battery BAT1 to an nth battery BAT, the negative electrode of the first battery BAT1 serves as the negative electrode terminal of the battery pack and is connected to a ground terminal (AGND), and the positive electrode of the nth battery BAT serves as the positive electrode terminal of the battery pack, where N is an integer greater than or equal to 2.

The battery monitoring apparatus may include a voltage conversion unit 100, a gating unit 200, and a current compensation unit 300.

The number of the voltage conversion units is M, wherein M is more than or equal to 2 and less than or equal to N, the M conversion units respectively receive the battery voltage between the positive end and the negative end of each battery from the second battery to the Nth battery, each conversion unit comprises a first operational amplifier, a first transistor, a first transimpedance unit and a second resistor, one end of each first transimpedance unit receives the positive end voltage of each battery, the other end of each first transimpedance unit is connected to the negative input end of the first operational amplifier, the positive input end of each first operational amplifier receives the negative end voltage of each battery, the output end of each first operational amplifier is connected with the grid of the first transistor, the other end of each first transimpedance unit and one end of each second transimpedance unit are respectively connected with the source and the drain of the first transistor, the other end of each second transimpedance unit is connected to the ground end, and when the first transistors are switched on, a first current flows through the first transimpedance units, A first transistor and a second transimpedance unit, and a battery voltage of each battery is measured by a voltage of the second transimpedance unit.

Wherein the first and second transimpedance units are shown as resistors in fig. 1, but the first and second transimpedance units may also be transistors, or a series unit of a resistor and a transistor. For example, as shown in fig. 2, the first transimpedance unit and the second transimpedance unit may be a series circuit of a P-type MOSFET and a resistor, a series circuit of an N-type MOSFET and a resistor, or a P-type MOSFET or an N-type MOSFET.

The following description will take a resistor as an example, but those skilled in the art will understand that the other methods shown above are also applicable to the technical solution of the present disclosure.

Fig. 3 shows a schematic diagram of a voltage converting unit 100 according to an embodiment of the present disclosure. Taking the voltage converting unit of BATn as an example, the voltage converting unit 100 may include a firstOperational amplifier OTA1, first transistor MPBn, first resistor R_naAnd a second resistor R_nb. The first transistor MPBn is a P-type MOSFET here.

Wherein, the positive pole of the battery BATn can be connected with the filtering and ESD protection resistor R_FOne terminal of (1), resistance R_FA filter capacitor C connected with the PINn terminal_FOne end of which is connected with a resistor R_FOne terminal of (C), a capacitor_FAnd the other end thereof is connected to the negative electrode (PIN (n-1) terminal) of the battery BATn.

Resistance R_naOne end of the resistor R is connected with the PINn end_naIs connected to the source of the first transistor MPBn, a resistor R_naThe connection node SFBn to the first transistor MPBn is connected to the negative input INN of the first operational amplifier OTA 1. The drain of the first transistor MPBn is connected with a second resistor R_nb(sampling resistor) and a resistor R_nbThe other end of the first switch is connected with the ground terminal. The positive input terminal INP of the first operational amplifier OTA1 is connected to the negative terminal of the battery bat, i.e., to the PIN (n-1) terminal, and the first operational amplifier OTA1 is powered by VCC.

A second resistor R_nbVoltage V of_RnbAnd outputs the voltage as the voltage of the battery BATn.

The voltage conversion unit 100 described above may be employed for all of the second to nth batteries. For the first battery BAT1 (i.e. the battery with its negative electrode grounded), the positive electrode of the battery BAT1 may be connected to the filter and ESD protection resistor R_FOne terminal of (1), resistance R_FConnected with PIN1 terminal, and filter capacitor C_FOne end of which is connected with a resistor R_FOne terminal of (C), a capacitor_FAnd the other end thereof is connected to the negative electrode (PIN0 terminal) of the battery BAT 1. And a first resistance R_1aAnd a second resistor R_1bThe two ends are connected to the PIN1 end and the PIN0 end after series connection. Resistance R_1bThe voltage across both ends is output as the voltage of battery BAT 1.

In the method, the voltage difference between the positive electrode and the negative electrode of any battery is converted and amplified into the absolute voltage of the reference ground, so that the absolute voltage can be converted into the low voltage, high-voltage resistant devices are avoided, and the absolute voltage can be acquired with high precision.

The first operational amplifier OTA1, with MPB2, …, MPB (n-1), MPBn, respectively, form a plurality of negative feedback loops, each of which converts the voltage difference between the two terminals of the corresponding single battery into a voltage to ground. Either negative feedback loop is shown in figure 3.

Resistance R_naIs connected to the chip pin PINn, which passes through a resistor R_FConnected to the positive electrode of the nth battery BATn, the voltage of the positive electrode of BATn to the reference ground is VBATn, and the nth R of the resistor is not flowed through by current_FWhen is, i.e. V_PINnVBATn. Resistance R_naIs connected to the negative input INN of the first operational amplifier OTA 1.

The positive input terminal INP of the first operational amplifier OTA1 is connected to the chip PIN PIN (n-1), and the PIN (n-1) terminal is connected to the chip PIN PIN (n-1) through a resistor R_FA resistor R connected to the positive electrode of the (n-1) th battery BAT (n-1) and passing through the (n-1) th battery in the absence of current_FWhen, V_PIN(n-1)VBAT (n-1) is the voltage of the positive electrode of BAT (n-1) to ground.

According to the negative feedback principle, V is assumed that the voltage gain of the first operational amplifier OTA1 is large enough and the input offset voltage is 0 (ideal case)_SFBn＝V_PIN(n-1). Then the resistance R_naVoltage difference V between two ends_Rna＝V_PINn-V_PIN(n-1)Through a resistance R_naCurrent of (I)_sn＝[V_PINn-V_PIN(n-1)]/R_na. Because of the resistance R_naIs connected to the PINn terminal, which is then connected to the resistor R_FIs connected to the positive electrode of BATn, and thus flows through the resistor R_FIs also equal to I_snI.e., [ V ]_PIN(n)-V_PIN(n-1)]/R_na。

Then the resistance R_FVoltage difference V between two ends_RFn＝R_F*[V_PINn-V_PIN(n-1)]/R_na。

In the circuit structure, when the voltage difference between two ends of the Nth battery is converted and collected, the voltage conversion units of other batteries are in an off state, that is, the first operational amplifiers OTA1 and MPB corresponding to other batteries are in an off state, so that the first operational amplifiers OTA1 and MPB are connected with PIN (N-1)Connected n-1 th resistor R_FThe current flowing is 0, then V_RF(n-1)＝0。

According to the above description, I_sn＝[V_PINn-V_PIN(n-1)]/R_na＝[V_BATn-V_BAT(n-1)]/(R_na+R_F)，[V_BATn-V_BAT(n-1)]Equal to the voltage difference V (BATn) between the positive and negative poles of the Nth battery. Resistance R_naCorresponding to the sampling resistance of voltage to current.

Current I_snFlows into a source terminal SFBn of the MPBn of the P-type source follower, flows out from a drain terminal of the MPBn, and flows into a resistor R_nbResistance R_nbIs connected to the drain of the MPBn, a resistor R_nbIs connected to the frame reference ground (AGND). Resistance R_nbThe voltage difference between the two ends (i.e. the resistance R)_nbHigh end to "ground" voltage) is: v_Rnb＝I_sn*R_nb＝R_nb*[V_BATn-V_BAT(n-1)]/(R_na+R_F)＝V(BATn)*R_nb/(R_na+R_F)。

From the above formula, the resistance R can be seen_nbHas the function of converting the current I_snAnd then the voltage difference between the two ends of the Nth battery is converted into the voltage relative to the reference ground.

The gating unit 200 of the present disclosure may include N switches sw1, sw2, … …, swn, wherein the N switches sw1, sw2, … …, swn input corresponding N voltages V_R1b、V_R2b、……、V_Rnb. When the battery voltage between the positive electrode end and the negative electrode end of the ith battery in the battery pack is measured, the gating unit measures the battery voltage of the ith battery by measuring the voltage of the second resistor corresponding to the ith battery according to the voltage conversion unit corresponding to the ith battery, and the voltage conversion units corresponding to other batteries are in a closed state, wherein i is more than or equal to 1 and less than or equal to N. Wherein the switches sw1, sw2, … …, swn are controlled by a control signal Φ sw. A schematic waveform diagram of the control signal Φ sw is shown in fig. 3. When high, the corresponding switch is opened.

Current compensation of the present disclosureThe unit 300 generates a compensation current of an ith battery based on a voltage of a second resistor corresponding to the ith battery in the battery pack, and the compensation current of the ith battery is supplied to the one end of a first resistor corresponding to the ith battery. The current compensation unit 300 receives the N voltages V selected from the gate unit 200_R1b、V_R2b、……、V_RnbAnd according to the voltage, the current compensation unit 300 compensates the current I of the battery corresponding to the voltage_snCompensation is performed.

From formula V_Rnb＝V(BATn)*R_nb/(R_na+R_F) It can be seen that the voltage difference between the two ends of the Nth battery is converted and amplified, and then is compared with the voltage V of the reference ground_RnbAnd a resistor R_FIn connection with, R_FThe value of the resistance is related to the actual application scene, and the value range is large, and is usually 100-10 k ohm. R_naTypically about 100 kohm, R_nb＝0.5*R_naWhen R is 50k ohm_na>>R_FThen V_RnbCan be represented as V_Rnb＝V(BATn)*R_nb/(R_na+R_F)＝V(BATn)*(1-R_F/R_na)*R_nb/R_naThen, because R_FIs R_F/R_naThe error is about 0.2% -20%. Therefore, it is necessary to pass through R_FSo that the current finally flows through R_FIs 0 so that R is_FThe voltage difference between both ends is 0, then V_Rnb＝V(BATn)*R_nb/R_naThe voltage to the reference ground and R after conversion and amplification_FIndependently, conversion and amplification errors are eliminated.

In addition, it can be known from any subunit of the voltage converting unit of fig. 3 that: when phi is_sw2At high level, the first operational amplifier OTA1 and transistor MPB2 of BAT2 are turned on, start to work, consume current, convert and amplify the voltage difference between the two ends of the 2 nd battery BAT2 with reference to ground, and output V_R2b(ii) a … … when phi_sw(n-1)First operation of the (n-1) th battery BAT (n-1) at high levelThe amplifier OTA1 and MPB (n-1) are turned on, start to work, consume current, convert and amplify the voltage difference between the two ends of the (n-1) th battery BAT (n-1) to the reference ground, and output V_R(n-1)b. When phi is_swnWhen the voltage is high, the first operational amplifiers OTA1 and MPBn of the Nth battery BATn are turned on, start to work, consume current, convert and amplify the voltage difference between the two ends of the Nth battery BATn to the reference ground, and output V_Rnb。

In one complete whole voltage conversion and amplification period, i.e. over n phi₁At high level, i.e. from phi_sw1High level to phi_swnDuring high level time of (phi)_C1BHigh level to phi_CnBThe electric charges consumed from each battery in the overall circuit of the voltage converting unit of the high-voltage series battery pack in fig. 1 during the high level time are respectively:

Q(BAT1)＝(I_s1+I_s2+…+I_sn)*t_H(Φ1),

Q(BAT2)＝(I_s2+…+I_sn)*t_H(Φ1),

Q(BAT(n-1))＝(I_s(n-1)+I_s(n))*t_H(Φ1),

Q(BATn)＝I_s(n)*t_H(Φ1)。

t_H(Φ1)the high duration of the voltage sampling clock. The charge consumed by the voltage converting unit in each cell during the voltage conversion and amplification period is unbalanced, and as can be seen from the above, the charge consumed by the 1 st cell is N times that of the nth cell, usually I_s1＝I_s2＝……＝I_sn. Because of the unbalance of the charge consumption of each battery, along with the increase of charge-discharge cycles, the voltage of each single battery is gradually differentiated, and a certain battery is discharged to an undervoltage state in advance compared with other batteries in the discharging process, or the certain battery is fully charged in advance compared with other batteries in the charging process, so that the service life of the whole battery pack is greatly shortened.

To solve the problem of R_FIntroduced switching and amplification errors, and problems of imbalance between cells, are therefore addressed hereinThe disclosure employs a current compensation unit 300 as described in fig. 1.

In the current compensation unit 300 shown in fig. 1, the current compensation unit generates a compensation current of an ith cell based on a voltage of a second resistor corresponding to the ith cell in the battery pack, and the compensation current of the ith cell is supplied to the one end of a first resistor corresponding to the ith cell. It is noted that the compensation current for each cell in the present disclosure is derived from the voltage of each cell relative to ground reference. Therefore, the compensation current through the present disclosure will be more accurate.

The current compensation unit comprises a second operational amplifier, a second N-type transistor and N-1 current mirror circuits, wherein the positive input end of the second operational amplifier is connected with the output of the gating unit so as to receive the battery voltage difference of the ith battery, i is more than or equal to 1 and less than or equal to N-1, the output end of the second operational amplifier is connected with the grid electrode of the second transistor, the source electrode of the second transistor is connected with one end of a third resistor, the other end of the third resistor is connected with the grounding end, the connection node of the second transistor and the third resistor is connected with the negative input end of the second operational amplifier, the drain electrode of the second transistor is connected with the current mirror circuits, and the N-1 current mirror circuits are enabled to output the compensation current of the ith battery according to the battery voltage difference of the ith battery.

The current compensation unit comprises a third N-type transistor, the grid electrode of the third N-type transistor is connected with the positive input end of the second operational amplifier, the source electrode of the third N-type transistor is connected with the grounding end through a fourth resistor, the drain electrode of the third N-type transistor receives the compensation current of the (N-1) th battery, and when the gating unit generates the compensation current of the (N-1) th battery based on the voltage difference of the (N) th battery, the compensation current of the (N) th battery is equal to the compensation current of the (N-1) th battery.

The current compensation unit comprises a third N-type transistor, the grid electrode of the third N-type transistor is connected with the positive input end of the second operational amplifier, the source electrode of the third N-type transistor is connected with the grounding end through a fourth resistor, the drain electrode of the third N-type transistor is connected with the booster circuit, the booster circuit is used for generating boosted voltage larger than the voltage of the positive electrode end of the Nth battery, when the switching-on unit generates the compensation current of the Nth battery based on the voltage difference of the Nth battery, the compensation current of the Nth battery is obtained based on the boosted voltage generated by the booster circuit, and the compensation current of the Nth battery is supplied to the first end of the first resistor corresponding to the Nth battery.

The voltages of the positive and negative electrodes of each battery in the series battery pack to the reference ground after conversion and amplification are respectively V_R1b、V_R2b、V_R3b、……、V_R(n-1)b、V_RnbThe N voltages are fed from phi as shown_swA one-out-of-multiple gating unit 200 controlled by the sampling control signal, the output voltage of the gating unit 200 is V_comp，Φ_swIs formed by phi_sw1～Φ_swnWhen the corresponding gating switches sw 1-swn are opened at high level, V is formed by a group of clock control signals_comp＝V_RnbN is 1 to N, and N is the total number of the series-connected batteries.

V_compThe input is input to the positive input end of a second operational amplifier OTA2, the second operational amplifier OTA2 and an N-type source follower MNBC0 form a negative feedback loop, the MNBC0, the MNBCn and the MNBCC are N-type MOSFETs with the same type and width-length ratio, the MNBC0, the MNBCn and the MNBCC are connected with the output end of the second operational amplifier OTA2 at the grids, and a resistor R_2c＝R_3c＝R_1c。

The second operational amplifier OTA2 and the N-type source follower MNBC0 form a negative feedback loop, and V is set to be 0 (ideal) assuming that the voltage gain of the second operational amplifier OTA2 is large enough and the input offset voltage is 0_s0＝V_comp，I_cp0＝V_s0/R_1c＝V_comp/R_1cIn circuit design, R is selected_1c＝R_1b＝R_2b＝……＝R_(n-1)b＝R_nbWhen is equal to phi_sw1At a high level, phi_sw2～Φ_swnWhen the voltage is low, the sw1 is on, the sw 2-swn are off, and V is_comp＝V_R1bAnd R is_1c＝R_1bThen, I_cp0＝V_comp/R_1c＝V_R1b/R_1b＝I_s1，I_cp0Flowing into P-type MOS transistors MPC0 and MPCB0 to form compensation referenceThe current is applied.

In the current compensation unit, MPC0& MPCB0, MPC1& MPCB1, MPC2& MPCB2, … … MPC (n-1) & MPCB (n-1) are connected through switches swcb1/swca 1-swcb (n-1)/swca (n-1) to form a current mirror, and sources of P-type MOS transistors MPC0, MPC1, … … and MPC (n-1) are directly connected to VCC, namely, the highest voltage end of the series battery pack, the positive electrode of BATn, VCC being VBATn, which is the voltage of the positive electrode of the highest battery against the reference ground, namely, VCC of the highest voltage end of the series battery pack against the reference ground.

Timing signal phi_C1&Φ_C1B、……、Φ_C(n-1)&Φ_C1B(n-1)Respectively control switch swca1&swcb1、……、swca(n-1)&The on and off of swcb (n-1) are switched on and off with the timing shown in FIG. 4.

When phi is_sw11 (high level), Φ_C10 (low level), Φ_C1B1 (high level), and Φ_C2～Φ_C(n-1)＝1，Φ_C2B～Φ_C(n-1)BWhen the signal value is equal to 0, the swca1 is turned off, the swcb1 is turned on, the swca2 to the swca (n-1) are turned on, the swcb2 to the swcb (n-1) are turned off, and the MPC0&MPCB0 and MPC1&MPCB1 forms a current mirror, then I_cp0＝I_cp1。I_cp1Flows into PMOS tube MPC1 from VCC end&MPCB1, then into PIN PIN1, i.e. resistor R_FLeft end of because of the signal phi_sw1＝1，V_comp＝V_R1bThen, I_cp1＝I_cp0＝V_comp/R_1c＝V_R1b/R_1cAnd due to R_1c＝R_1bThen, I_cp0＝I_s1。

That is, the current flows into the resistor R_1bCurrent of (I)_s1Can be directly generated by a mirror current I generated from the highest voltage point VCC of the series battery_cp1Compensated, i.e. without power supply from BAT1, while flowing through resistor R_FIs also 0 because of the resistance R_FA positive electrode connected in series with the battery BAT1 and a resistor R_1bThe high end of (c).

In the same way, it can be obtained that when phi_sw(n-1)1 (high level), Φ_C(n-1)0 (low level), Φ_C(n-1)B＝1(High level), phi_C1～Φ_C(n-2)＝1，Φ_C1B～Φ_C(n-2)BWhen equal to 0, because of the signal phi_sw(n-1)When 1, V_comp＝V_R(n-1)bThe MPC0 is formed by switching off swca (n-1), switching on swcb (n-1), switching on swca 1-swca (n-2) and switching off swcb 1-swcb (n-2)&MPCB0 and MPCC (n-1)&MPCB (n-1) forms a current mirror, then I_cp(n-1)＝I_cp0＝I_s(n-1)。

That is to say into the resistor R_(n-1)bCurrent of (I)_s(n-1)Can be directly generated by a mirror current I generated from the highest voltage point VCC of the series battery_cp(n-1)Compensated, i.e. without the need for power supply from BAT (n-1), while passing through the (n-1) th R_FIs also 0.

By passing through_sw1～Φ_sw(n-1)A group of clock-controlled gating switches sw 1-sw (n-1) to make the output voltage V of the sampling gating unit_compThe output V can be selected in one-to-one correspondence_R1b～V_R(n-1)b，V_comp＝V_R1b～V_R(n-1)bBecause of the compensation of the reference current I_cp0＝V_comp/R_1c。

And V_compThe selected unit dynamically selects and outputs V_R1b～V_R(n-1)bTherefore, when converting and amplifying the voltage difference across different cells to a reference ground, the reference current I is compensated_cp0Also follows the voltage difference of the corresponding battery, so that I_cp0Exactly equal to I_s1～I_s(n-1)Thereby realizing accurate dynamic compensation such that the voltage conversion unit does not absorb current from the BATs 1-BAT (n-1) when the 1 st to n-1 th batteries are converted, that is, no current flows through the 1 st to n-1 th resistors R_FThat is, the 1 st to n-1 st resistors R are eliminated_FThe compensation current corresponding to each battery flows out from VCC, and VCC makes the highest positive pole of the series battery pack completely pass through each battery from BAT1 to BATn, so the problem of unbalanced consumption current of BAT 1-BAT (n-1) is thoroughly solved.

In the aboveThrough phi_sw1＝1、Φ_C1＝1、Φ_C1B0 to phi_sw(n-1)＝1、Φ_C(n-1)＝1、Φ_C(n-1)B0, during the conversion and amplification of the voltage with respect to the reference, I_cp0～I_cp(n-1)All generated based on VCC, flows into PINs PIN 1-PIN (n-1), i.e. into resistor R_1b～R_(n-1)bThe current flows only from the high voltage to the low voltage, the voltages of the PINs PIN 1-PIN (n-1) are all smaller than VCC, and the voltage V of PIN (n-1)_PIN(n-1)The voltage difference between the two ends of the highest battery is lower than VCC, namely VCC-V_PIN(n-1)V (batn), around 3.7V.

This gives rise to the problem of compensating for the flow through the n-th resistor R_FCurrent of (3) to eliminate the nth resistor R_FResulting in conversion and amplification errors.

I.e. when phi_swnWhen 1 (high level), the nth first operational amplifier OTA1 and MPBn are turned on, the voltage difference across the nth battery BATn is converted and amplified with reference to ground, and V is output_Rnb。

Because of the nth resistor R in the figure_FThe left end is connected to the positive pole of the nth battery, i.e. VBATn, i.e. VCC, which is the highest voltage of the system with respect to the reference ground, and in the circuit configuration shown in fig. 1, a compensation current I cannot be generated_cpnFlows into the nth filter R_FThe right end of the resistor, i.e., the PINn end.

In order to eliminate the n-th resistor R_FThe error introduced, as shown in FIG. 1, is introduced by introducing MNBCn and R_2cThe channel width-length ratios of the MNBC0 and the MNBCn are the same, the gates of the MNBCn and the MNBC0 are connected to the output end of the second operational amplifier OTA2, and the resistor R_2c＝R_1cThen V_s0＝V_snAnd V is_s0＝V_compWhen phi is_swnWhen 1 (high level), the nth gate switch swn is turned on, V_comp＝V_RnbTherefore I is_cn＝V_sn/R_2c＝V_comp/R_2c＝V_Rnb/R_nb＝I_sn。

Further, when Φ_swn＝1At (high level), phi in the cell current compensation unit_BATnWhen the switch swt1 is turned on at1 (high), the compensation current I_cnInto the PIN (n-1) PIN, i.e. the (n-1) th resistor R_FTo the right end of the housing. I is_snInto the nth resistor R_FRight end of (1), I_cn＝I_sn，R_F(n)＝R_F(n-1)＝R_FTherefore I is_cn*R_F(n-1)＝I_sn*R_F(n) of (a). Thus the nth resistor R_FIs equal to the (n-1) th filter resistor R_FIs measured.

V_PINn-V_PIN(n-1)VBATn-VBAT (N-1) ═ V (batn), V (batn) is the voltage across the nth battery, thus V_Rnb＝V(BATn)*R_na/R_nbThereby eliminating the nth resistor R_FResulting in conversion and amplification errors.

From the above analysis, it can be seen that the resistances R of BAT1 to BAT are eliminated based on the dynamic current compensation unit shown in fig. 1_FThe resulting error in converting to ground and amplifying the voltage across the battery also eliminates the problem of current imbalance consumed from BATs 1-BAT (n-1). That is to say that the first and second electrodes,

Q(BAT1)＝(I_s1+I_s2+…+I_sn+I_cn)*t_H(Φ1),

Q(BAT2)＝(I_s1+I_s2+…+I_sn+I_cn)*t_H(Φ1),

Q(BAT(n-1))＝(I_s1+I_s2+…+I_sn+I_cn)*t_H(Φ1),

in the present circuit configuration, since VCC is the highest voltage of the battery pack, a current is generated to flow into the nth resistor R_FI.e. the chip pin PINn, so that I cannot be compensated for_snSo that BAT 1-BAT (n-1) consume more I than BATn in a complete voltage conversion and amplification cycle_cn*t_H(Φ1)But has been greatly reduced compared to a scheme without dynamic compensation.

To further eliminate the low consumption charge I of BATn_cn*t_H(Φ1)In the present disclosure, a boost circuit, such as a boost charge pump and an inductive boost converter, is used to generate a voltage higher than the maximum voltage VCC for outputting a compensation current I_cpnFlow into chip pin PIN (n), compensate I_snUnbalanced charge amount I of Nth battery_cn*t_H(Φ1)。

For example, as shown in FIG. 5, when phi_swnWhen 1 (high level), phi in the battery current compensation unit_BATnWhen the voltage is 1 (high level), the switch swt1 is turned on, the transistor MNBBn is turned on, the boosted voltage is generated by the voltage boosting circuit, and the compensation current I is generated_cpn，I_cpnFlowing into pin PINn, i.e. n-th resistor R_FTo the right end of the housing. I is_snInto the nth resistor R_FRight end of (1), I_cpn＝I_sn。

According to a further embodiment of the present disclosure, the battery voltage monitoring device of the present disclosure further comprises a voltage collecting circuit, wherein the voltage collecting circuit comprises an N-type transistor MNBCC and a resistor R_3CAn N-type transistor MNBB1, and a P-type transistor MPCBB. Wherein the MNBC0 is an N-type MOSFET with the same type and width-length ratio size as the MNBCn and MNBCC, the gates of the MNBC0 and the MNBCn and MNBCC are connected to the output end of the second operational amplifier OTA2, and the resistor R_2c＝R_3c＝R_1c. When the voltage difference of the ith battery is sampled, the MPCBB, MNBB1 and MNBCC are turned on, and the compensation current I flowing through the ith battery_cpBThrough the compensation current at the resistor R_3cThe voltage difference of the ith battery is obtained after the voltage formed by the voltage acquisition module is acquired, and an ADC converter and the like can be arranged subsequently, so that the voltage difference is output through an output signal D_outAnd finishing the acquisition of the voltage.

Fig. 6 shows a circuit diagram of a current compensation approach according to another embodiment of the present disclosure. Fig. 7 and 8 show a partial enlarged view of fig. 6.

The voltages to earth after the voltage difference of the positive electrode and the negative electrode of each battery in the battery pack is converted and amplified are respectively V_R1b、V_R2b、V_R3b、...、V_R(n-1)b、V_R(n)bThese voltages areIs fed into a channel phi as shown in the figure_SWA sampling gating unit with one out of multiple controlled by sampling clock signal and output voltage of the sampling gating unit is V_COMP，Φ_SWIs formed by phi_SW1～Φ_SW(n)When the corresponding gating switches sw 1-sw (n) are opened at high level, V is_COMP＝V_R(n)bAnd N is 1: N, and N is the total number of the series batteries.

V_COMPInput to the positive input terminal of an operational amplifier OTA2, OTA2 and N-type source follower MNBC0, R_C0Forming a negative feedback loop, the MNBC0 to MNBC (N) and the MNBCC1 are N-type MOSFETs with the same type and width-length ratio size, and the gates of the MNBC0 to MNBCn and the MNBCC1 are all connected to the output end of the OTA 2. Wherein, the resistance R_C1＝R_C2＝R_C3＝R_Cn. V assumes that the voltage gain of the operational amplifier OTA2 is large enough and the input offset voltage is 0 (ideal case)_s0＝V_COMPThus I_cp0＝V_s0/R_C1＝V_COMP/R_C1Selecting R_C1＝R_1b＝R_2b＝…＝R_(n-1)b＝R_nbWhen phi is_sw1At a high level, phi_sw2～Φ_swnWhen the voltage is low, the sw1 is on, the sw 2-swn are off, and V is_COMP＝V_R1bAnd because of R_C1＝R_1bThen Icp₀＝V_COMP/R_C1＝V_R1b/R_1b＝I_s1，I_CP0Flows into the PMOS tubes MPC0 and MPCB0 to form the compensation reference current.

In the current compensation unit of FIG. 8, the switches swcb1/swca1 are respectively formed by phi_C1&Φ_C1BControl when phi_C11 (high level), Φ_C1BWhen the signal level is 0 (low level), the switch swcb1 is turned on, the switch swca1 is turned off, and the MPC0&MPCB0 and MPC1&The MPCB1 is connected through a switch swcb1/swca1 to form a current mirror, then I_cp0＝I_cp1＝V_COMP/R_C1＝V_R1b/R_1b＝I_s1. The sources of the P-type MOS transistors MPC0 and MPC1 are both directly connected to VCC, i.e. the highest voltage end of the series battery, i.e. the anode of BAT (n), VCC-VBAT (n)Vbat (n) is the positive-to-ground voltage of the highest cell, i.e., VCC, which is the highest end-to-ground voltage of the series battery.

I_cp1Flows into a P-type MOS tube MPC1 from a VCC end&MPCB1, and then into PIN PIN1, i.e. filter resistor R_FInto the sampling resistor R, i.e. into the left end of the sampling resistor R_1bCurrent of (I)_s1Can be directly generated by a mirror current I generated from the highest voltage point VCC of the series battery_cp1Compensated, i.e. without power supply from BAT1, because R_FA positive electrode connected in series with BAT1 and a resistor R_1bSo as to flow through R_FIs also 0.

Timing signal phi_C2&Φ_C2B～Φ_C(n)&Φ_C1B(n)The gates of switches MNNBB2 to mnbb (n) are controlled, respectively, and the timing is shown in fig. 4. When phi is_sw21, the 2 nd gate switch is opened, V_COMP＝V_R2b，V_COMP/R_C2＝I_c2，V_R2b/R_2b＝I_s2Because of V_COMP＝V_R2b，R_C2＝R_2bThen, I_c2＝I_s2。

Because of phi_C21 (high level), Φ_C2B0 (low level), Φ_C3～Φ_C(n)＝0，Φ_C2B～Φ_C(n)BWhen the voltage is 1, MNNBB2 is on, MNNBB3 to mnbb (n) are off, and I is_c2Flows into the first filter resistor R through MNNBB2_F(Filter resistor R connected to negative terminal of BAT2_F) Then the voltage drop across the filter resistor RF is V (R)_F1)＝I_c2*R_FFlows into the 2 nd filter resistor R_F(Filter resistor R connected to the positive electrode of BAT2_F) Has a current of I at the right end_s2Then the voltage drop across the 2 nd filter resistor RF is V (R)_F2)＝I_s2*R_F。

Because of I_c2＝I_s2Then V (R)_F1)＝V(R_F2) At this time R_2aThe voltage difference between the two ends is V_BAT2-V(R_F2)-(V_BAT1-V(R_F1))＝(V_BAT2-V_BAT1)+(V(R_F1)-V(R_F2))

＝(V_BAT2-V_BAT1). Wherein (V)_BAT2-V_BAT1) Is the voltage across the 2 nd cell. Thus flowing into the sampling resistor R_2bCurrent of (I)_s2Can be directly formed by I_c2Is compensated for, I_s2At the 2 nd filter resistor R_FThe generated voltage drop is I_c2The voltage drops at the 1 st filter resistor RF cancel out, so that R_FDoes not have any effect on the battery voltage sampling.

The same principle, I_s3At the 3 rd filter resistor R_FThe generated voltage drop is I_c3At the 3 rd filter resistor R_FThe resulting voltage drops cancel out … …. Similarly, when the filter resistance R is matched with the nth filter resistance R_FWhen the generated voltage drop is cancelled, the booster circuit described above may be used. And will not be described in detail herein.

Fig. 9 shows a circuit schematic of a voltage acquisition module according to one example of the present disclosure. Wherein the control signal phi as shown in figure 4 is passed through the voltage acquisition module₁、Φ₂And phi₁Is delayed by a signal phi_1d、Φ_1dBThe transistors in the figure are controlled, so that the voltage acquisition is realized through the charging and discharging of the resistors Csp and Csn. By the voltage acquisition module shown in fig. 9, high-precision sample-and-hold can be realized.

In practical applications, the first operational amplifier OTA1 in the voltage converting unit corresponding to each battery also needs to consume current, so if the nth voltage converting unit is powered from the nth battery bat (n), similar I will be generated_snResulting in the problem of unequal charge consumption by each battery,

to address the battery imbalance problem due to the current consumed by the first operational amplifier OTA1, the first operational amplifier OTA1 in the present disclosure is all powered by the highest voltage VCC of the battery pack.

Fig. 10 shows a circuit schematic of a conventional operational amplifier OTA 1. In the circuit shown in fig. 10, the output terminal of the operational amplifier OTA1 is connected to ground via a series circuit of a resistor and a capacitor. This requires the use of high voltage resistant metal capacitors.

In one example of the present disclosure, the output terminal of the first operational amplifier corresponding to each of the second to nth batteries is connected to the one terminal of the first resistor corresponding to each of the batteries via a resistor and capacitor series circuit.

As shown in fig. 11, the output terminal of the operational amplifier OTA1 is connected to the PINn terminal via a resistor and capacitor series circuit. By connecting the compensation capacitor C1 and the compensation resistor R1 to the PINn terminal, not to the ground terminal, the use of a high-voltage-resistant metal capacitor can be avoided, and the cost can be reduced.

However, a resistor and capacitor network R for compensating the resistor R1 and filtering out the battery voltage fluctuation outside the chip pin_FAnd C_FIn series, wherein R_FAnd C_FThe low-pass filter network is used for preventing ESD (electrostatic discharge) from breaking down the internal circuit of the chip and filtering out the voltage fluctuation R of the battery_FAnd C_FThe magnitude of the value of (A) depends on the severity of the electrical environment of the application scenario, R_FTypically in the range of 100ohm to 10k ohm, C_FThe value range is typically 100nF to 10uF, since R_FA large variation of the value, resulting in zero point of compensation with R_FChange because the zero point of compensation is approximately equal to 1/[ C [)₁*(R_F+R₁)]。

Thus, in further examples of the disclosure, to address the cause R_FAnd C_FLow-pass filter network and compensation resistor R formed₁Compensation zero introduced in series with R_FThe problem of drift, a further solution is proposed.

Fig. 12 shows a high output resistance current source powered OTA1 architecture, while fig. 13 shows a circuit schematic of a high output resistance current source powered operational amplifier OTA 1.

In the schemes shown in fig. 12 and 13, the operational amplifier core circuit does not need to adopt a high-voltage tube design, the matching is better, the chip area is smaller, and the compensation capacitor does not need to adopt a high-voltage capacitor nor use a high-voltage capacitorExist due to R_FAnd C_FLow-pass filter network and compensation resistor R formed₁Compensation zero introduced in series with R_FThe problem of drift.

As shown in fig. 12, the supply terminal VPP of the first operational amplifier OTA1 is connected to VCC via a current source IBP, and the ground terminal VPN of the first operational amplifier OTA1 is connected to ground via a current source IBN. In I_BN＝I_BPIn this case, it can be realized that the first operational amplifier OTA1 does not directly absorb current from the battery sampling terminal pin (n), thereby eliminating the imbalance of the consumed current of each battery and ensuring that the battery voltage of the current battery collected by the first operational amplifier OTA1 is completely accurate.

According to a further embodiment of the present disclosure, there is also provided a first operational amplifier powered by a P-N source follower. The level shifter generates a first control signal and a second control signal based on a voltage difference between batteries of an ith battery, wherein a power supply terminal of a corresponding first operational amplifier of the ith battery is connected to a source terminal of an N-type MOSFET, a drain terminal of the N-type MOSFET is connected to a positive terminal voltage of the nth battery, a ground terminal of the corresponding first operational amplifier of the ith battery is connected to a source terminal of a P-type MOSFET, the drain terminal of the P-type MOSFET is connected to a ground voltage, the first control signal is connected to a gate terminal of the N-type MOSFET, and the second control signal is connected to the gate terminal of the P-type MOSFET.

According to a further embodiment of the present disclosure, there is provided an operational amplifier power supply device for a battery voltage monitoring device for monitoring voltages of respective batteries of a battery pack including a plurality of batteries whose positive and negative poles are connected in series, the plurality of batteries being first to nth batteries, a negative pole of the first battery serving as a negative pole terminal of the battery pack and being connected to a ground terminal, and a positive pole of the nth battery serving as a positive pole terminal of the battery pack, where N is an integer greater than or equal to 2, including: an operational amplifier that generates a control signal that samples a battery voltage of each of the plurality of batteries based on the battery voltage in the battery pack, the operational amplifier including a power supply positive terminal and a power supply negative terminal; and the level converter generates a first voltage and a second voltage based on the voltage difference between the two ends of the ith battery, wherein i is more than or equal to 2 and less than or equal to N, the power supply voltage of the power supply positive end of the operational amplifier is generated based on the first voltage, and the power supply voltage of the power supply negative end of the operational amplifier is generated based on the second voltage.

The level shifter generates a first voltage and a second voltage based on a voltage difference across an ith battery and a voltage difference across an (i + 1) th battery.

Further, the embodiment further comprises an N-type MOSFET, wherein the drain electrode of the N-type MOSFET is connected with the voltage of the positive electrode terminal of the battery pack, the source electrode of the N-type MOSFET is connected with the power supply positive terminal of the operational amplifier, and the gate electrode of the N-type MOSFET is connected with the first voltage.

The power supply circuit further comprises a P-type MOSFET, wherein the source electrode of the P-type MOSFET is connected with the power supply negative terminal of the operational amplifier, the drain electrode of the P-type MOSFET is connected with the grounding voltage, and the gate electrode of the P-type MOSFET is connected with the second voltage.

Fig. 14 shows an architecture diagram of this embodiment. As shown in fig. 14, the level shifter includes an input terminal for obtaining a battery voltage difference of an ith battery (i is any one of a2 nd battery to an nth battery). The level shifter generates a first control signal VNG and a second control signal VPG. The first control signal VNG is output to the N-type MOSFET MNhvBufH, and the second control signal VPG is output to the P-type MOSFET MPhvBufH. The source of the N-type MOSFET MNhvBufH is connected to the supply terminal VPP of the first operational amplifier OTA1, the drain of the N-type MOSFET MNhvBufH is connected to the voltage VCC, the source of the P-type MOSFET mphvbbufh is connected to the ground terminal VPN of the first operational amplifier OTA1, and the drain of the P-type MOSFET mphvbbufh is connected to the output terminal of the ground voltage generator (VSS generator).

Figure 15 shows a circuit diagram of a VSS generator according to one example of the present disclosure. Wherein the VSS generator generates the VSS voltage based on the battery voltage of the 1 st battery and by a clock signal. Further, one example of a specific circuit of the configuration pattern in fig. 14 is shown in fig. 15.

When the architecture shown in fig. 14 is adopted, the first operational amplifier in the above-described manner may take the form of a structure as shown in fig. 16, for example, a level shifter and an operational amplifier are provided in the same chip. In addition, the configuration shown in fig. 16 can be employed in the same manner as in the embodiments shown in fig. 19 and 20. FIG. 17 illustrates a circuit diagram of a P-N dual source follower powered operational amplifier according to one embodiment of the present disclosure.

Three implementations of a level shifter are shown in fig. 18. For the left side view of fig. 18, VNG-VPG ═ V (DZ1), where V (DZ1) is the reverse breakdown voltage of zener diode DZ1, typically around 6V. For the middle view of fig. 18, VNG-VPG ═ VH (or VM) -VL, where VH (or VM) and VL are external bias voltages. For the right side view of fig. 18, VNG-VPG ═ V (DZ1), where V (DZ1) is the reverse breakdown voltage of zener diode DZ1, typically around 6V.

For the supply voltages VPP and VPN of the first operational amplifier in the circuit shown in fig. 14, VPP-VPN ═ VNG-vgs (mnhvbufh) ] - [ VPG + vsg (mphvbufh) ]

＝(VNG-VPG)-[VGS(MNhvBufH)+VSG(MPhvBufH)]

Generally, vgs (MNhvBufH) is 1V, and vsg (MPhvBufH) is 1V, where vgs (MNhvBufH) is the voltage difference between the gate (G) voltage and the source (S) voltage of NMOS transistor MNhvBufH, and vsg (MPhvBufH) is the voltage difference between the gate (G) voltage and the source (S) voltage of PMOS transistor MPhvBufH. Thus VPP-VPN ≈ 2V ≈ 4V. Compared with the method of supplying power to the operational amplifier by VCC, the method shown in fig. 14 and 18 can effectively reduce the power supply voltage of the operational amplifier, thereby avoiding the use of a high-voltage resistant operational amplifier.

An architecture diagram of an N-N dual source follower powered first operational amplifier OTA1 is provided in FIG. 19. The operational amplifier power supply device comprises an N-type MOSFET, wherein the drain electrode of the N-type MOSFET is connected with the voltage of the positive electrode terminal of the battery pack, the source electrode of the N-type MOSFET is connected with the power supply positive terminal of the operational amplifier, and the grid electrode of the N-type MOSFET is connected with the first voltage. The power supply circuit further comprises a second N-type MOSFET and a current source, wherein the source electrode of the second N-type MOSFET is connected to the power supply negative terminal of the operational amplifier and one end of the current source, the other end of the current source is grounded, the drain electrode of the second N-type MOSFET is connected with the voltage of the positive terminal of the battery pack, and the grid electrode of the second N-type MOSFET is connected with a second voltage.

The level shifter generates a first control signal and a second control signal based on a voltage difference between batteries of an ith battery, wherein a power supply terminal of a first operational amplifier corresponding to the ith battery is connected to a source terminal of a first N-type MOSFET, a drain terminal of the first N-type MOSFET is connected to a positive terminal voltage of the nth battery, the first control signal is connected to a gate terminal of the N-type MOSFET, and a ground terminal of the first operational amplifier corresponding to the ith battery is connected to a source terminal of a second N-type MOSFET and then grounded via a current source, a drain terminal of the second N-type MOSFET is connected to the positive terminal voltage of the nth battery, and the second control signal is connected to the gate terminal of the second N-type MOSFET.

VPP-VPN＝[VNG-VGS(MNhvBufH)]-[VPG-VGS(MNhvBufL)]

＝(VNG-VPG)-[VGS(MNhvBufH)-VGS(MNhvBufL)]

Generally, vgs (MNhvBufH) ≈ vgs (mnhvbufl), where vgs (MNhvBufH) is the voltage difference between the gate (G) voltage and the source (S) voltage of NMOS transistor MNhvBufH, and vsg (MPhvBufH) is the voltage difference between the gate (G) voltage and the source (S) voltage of PMOS transistor MPhvBufH. Thus, VPP-VPN (VNG-VPG) -0V (VNG-VPG) is approximately 6V. In the mode shown in fig. 19, the supply voltage of the operational amplifier can be effectively reduced, and the use of a high-voltage resistant operational amplifier can be avoided.

An architecture diagram of a first operational amplifier OTA1 powered by an N-type source follower and unity gain amplifier is shown in fig. 20. The operational amplifier power supply device further comprises an N-type MOSFET, wherein the drain electrode of the N-type MOSFET is connected with the voltage of the positive electrode end of the battery pack, the source electrode of the N-type MOSFET is connected with the power supply positive end of the operational amplifier, and the grid electrode of the N-type MOSFET is connected with the first voltage. The positive input end of the gain amplifier is connected with the second voltage, the negative input end of the gain amplifier is connected with the output end of the gain amplifier, and the output end of the gain amplifier is connected with the power supply negative end of the operational amplifier.

The level shifter generates a first control signal and a second control signal based on the voltage difference of the batteries of the ith battery, wherein the power supply terminal of the first operational amplifier corresponding to the ith battery is connected with the source electrode of the first N-type MOSFET, the drain electrode of the first N-type MOSFET is connected with the positive terminal voltage of the nth battery, the first control signal is connected with the grid electrode of the N-type MOSFET, the grounding terminal of the first operational amplifier corresponding to the ith battery is connected with the output terminal of the gain amplifier, the positive input terminal of the gain amplifier is connected with the second control signal, the negative input terminal of the gain amplifier is connected with the output of the gain amplifier to form a negative feedback loop, and the power supply terminal of the gain amplifier is connected with the positive terminal voltage of the nth battery and the grounding terminal.

In the mode shown in fig. 20, VPP-VPN ═ VNG-VGS (MNhvBufH) ] -VPG, VGS (MNhvBufH) is the voltage difference between the gate (G) voltage and the source (S) voltage of the NMOS transistor MNhvBufH, and VGS (MNhvBufH) ≈ 1V. In this way, the VPP-VPN-VPG is VNG-1V-VPG, and VPN-VPG is the output of the negative feedback voltage buffer with unit gain. In the mode shown in fig. 19, the supply voltage of the operational amplifier can be effectively reduced, and the use of a high-voltage resistant operational amplifier can be avoided.

In the above example, the description was made with reference to the case where the first transistor connected to the output of the first operational amplifier OTA1 is a P-type transistor.

The output voltage of the first operational amplifier OTA1, i.e. the gate voltage of MPB, is a high voltage to ground (AGND), V_OUTn＝V_BATn-V_GS[MPBn]，V_BAT(n)The voltage of the positive electrode of the nth battery to earth is equivalent to the series voltage of the nth battery, and is usually n x 3.7V, V_GS[MPB(n)]Is the source-drain voltage of MPB. Therefore, the internal current mirror and the current source load of the first operational amplifier OTA1 need to adopt a high-voltage-resistant MOSFET for circuit design, the chip area occupied by the high-voltage-resistant MOSFET is large, and meanwhile, because the MOSFET is always exposed to high voltage for working, the long-term accumulation effect of gate hot carrier injection also easily causes the reliability problem of threshold voltage deviation or source-drain series connection.

To solve this problem, as shown in fig. 21, the P-type source follower MPB driven by the first operational amplifier OTA1 is replaced with an N-type source follower (N-type MOSFET), and the output voltage V of the first operational amplifier OTA1_OUTn＝V_Rnb+V_GS[MNBn]，V_R(n)bUsually 0.5 times as much asThe cell voltage is usually 0.5 × 3.7V, and the source-drain voltage V of MNB_GS[MNBn]Usually 1 to 1.5V, so V_OUTnThe current source load and the current mirror inside the first operational amplifier OTA1 are designed with high voltage tolerant MOSFETs for low voltage to ground, thus avoiding. For the design of other parts, details are not repeated, and reference may be made to the above description.

Fig. 22 and 23 are circuit diagrams of operational amplifiers that can be used in the mode shown in fig. 21. It should be noted that the operational amplifier described above is equally applicable to the manner shown in fig. 21.

According to the above form of operational amplifier, the present disclosure still further provides an operational amplifier power supply method, the operational amplifier being used for a battery voltage monitoring device for monitoring voltages of respective batteries of a battery pack, the battery pack including a plurality of batteries whose positive and negative poles are connected in series, the plurality of batteries being first to nth batteries, a negative pole of the first battery being a negative pole terminal of the battery pack and being connected to a ground terminal, and a positive pole of the nth battery being a positive pole terminal of the battery pack, where N is an integer equal to or greater than 2, the operational amplifier generating a control signal for sampling a battery voltage of each of the plurality of batteries based on the battery voltages in the battery pack, the operational amplifier including a power supply positive terminal and a power supply negative terminal, comprising: generating a first voltage and a second voltage based on a voltage difference between two ends of the ith battery, wherein i is more than or equal to 2 and less than or equal to N; and generating a supply voltage of the supply positive terminal of the operational amplifier based on the first voltage and generating a supply voltage of the supply negative terminal of the operational amplifier based on the second voltage.

The first voltage and the second voltage are generated based on a voltage difference across the ith battery and a voltage difference across the (i + 1) th battery.

The drain electrode of the N-type MOSFET is connected with the voltage of the positive electrode end of the battery pack, the source electrode of the N-type MOSFET is connected with the positive power supply end of the operational amplifier, the grid electrode of the N-type MOSFET is connected with a first voltage, and the power supply voltage of the positive power supply end of the operational amplifier is the difference between the first voltage and the grid source voltage of the N-type MOSFET.

For other features in the method, reference may be made to the above description of specific implementations, which are not described herein again.

As shown in fig. 24, according to another embodiment of the present disclosure, a battery management system is provided. The battery management circuit is externally connected with the charger and the load, and the battery voltage monitoring circuit independently monitors the voltage condition of each battery in the battery pack. The specific implementation of the battery voltage monitoring circuit can refer to the above description, and is not described in detail herein. The battery management system can perform channel selection by controlling the arithmetic unit to detect the condition of the battery voltage of a certain section, perform analog-to-digital conversion after obtaining the battery voltage and control the arithmetic unit to perform processing, and the communication unit can provide information for controlling the arithmetic unit to the outside or receive information from the outside.

According to the above technical contents of the present disclosure, at least the following technical problems can be solved.

The utility model provides a high accuracy battery voltage monitoring circuit can gather the real-time voltage of each section battery in the high-pressure series application accurately, carries out the control protection of overcharging or overdischarging to the voltage of multisection series connection battery, ensures in time to discover the abnormal conditions of excessive charge-discharge of battery and battery broken string.

The scheme for independently compensating the current of each voltage acquisition module is provided, the current compensation problem of the highest battery is innovatively solved, meanwhile, the condition that the current consumption of each battery is unbalanced due to a voltage acquisition and conversion circuit in a series battery pack is well eliminated, the phenomenon of voltage unbalance of each battery after the battery pack is used for a long time is avoided, and the service life of the battery pack is prolonged.

Because a high-precision dynamic current compensation scheme is adopted, the acquired battery voltage of each section is completely real battery voltage instead of being subjected to ESD resistance and battery filter resistance R_FThe voltage after voltage reduction flows through the resistor R because of the adoption of current compensation_FIs 0, so R_FIs 0, the input voltage of the voltage conversion unit is completely equal to the battery voltage.

In the description herein, reference to the description of the terms "one embodiment/mode," "some embodiments/modes," "example," "specific example," or "some examples," etc., means that a particular feature, structure, material, or characteristic described in connection with the embodiment/mode or example is included in at least one embodiment/mode or example of the application. In this specification, the schematic representations of the terms used above are not necessarily intended to be the same embodiment/mode or example. Furthermore, the particular features, structures, materials, or characteristics described may be combined in any suitable manner in any one or more embodiments/modes or examples. Furthermore, the various embodiments/aspects or examples and features of the various embodiments/aspects or examples described in this specification can be combined and combined by one skilled in the art without conflicting therewith.

Furthermore, the terms "first", "second" and "first" are used for descriptive purposes only and are not to be construed as indicating or implying relative importance or implicitly indicating the number of technical features indicated. Thus, a feature defined as "first" or "second" may explicitly or implicitly include at least one such feature. In the description of the present application, "plurality" means at least two, e.g., two, three, etc., unless specifically limited otherwise.

It will be understood by those skilled in the art that the foregoing embodiments are merely for clarity of illustration of the disclosure and are not intended to limit the scope of the disclosure. Other variations or modifications may occur to those skilled in the art, based on the foregoing disclosure, and are still within the scope of the present disclosure.

Claims

1. An operational amplifier power supply device for a battery voltage monitoring device for monitoring voltages of respective batteries of a battery pack including a plurality of batteries whose positive and negative poles are connected in series, the plurality of batteries being first to nth batteries, a negative pole of the first battery serving as a negative pole terminal of the battery pack and being connected to a ground terminal, and a positive pole of the nth battery serving as a positive pole terminal of the battery pack, where N is an integer equal to or greater than 2, comprising:

2. The operational amplifier power supply of claim 1 wherein the level shifter generates the first voltage and the second voltage based on a voltage difference across an i-th cell and a voltage difference across an i + 1-th cell.

3. Operational amplifier supply unit according to claim 1 or 2, characterized in that it further comprises an N-type MOSFET, the drain of which is connected to the voltage of the positive terminal of the battery, the source of which is connected to the positive supply terminal of the operational amplifier and the gate of which is connected to the first voltage.

4. The operational amplifier power supply of claim 3 further comprising a P-type MOSFET having a source connected to the negative supply terminal of the operational amplifier, a drain connected to a ground voltage, and a gate connected to the second voltage.

5. The operational amplifier power supply of claim 3 further comprising a second N-type MOSFET having a source connected to the negative power supply terminal of the operational amplifier and one end of the current source, the other end of the current source being connected to ground, a drain of the second N-type MOSFET being connected to the voltage of the positive terminal of the battery pack, and a gate of the second N-type MOSFET being connected to the second voltage.

6. The operational amplifier power supply of claim 3 further comprising a gain amplifier having a positive input connected to the second voltage and a negative input connected to an output of the gain amplifier, the output of the gain amplifier being connected to the negative power supply terminal of the operational amplifier.

7. A power supply method for an operational amplifier used for a battery voltage monitoring device for monitoring voltages of respective batteries of a battery pack including a plurality of batteries whose positive and negative poles are connected in series, the plurality of batteries being first to nth batteries, a negative pole of the first battery serving as a negative terminal of the battery pack and being connected to a ground terminal, and a positive pole of the nth battery serving as a positive terminal of the battery pack, where N is an integer equal to or greater than 2, the operational amplifier generating a control signal for sampling a battery voltage of each of the plurality of batteries based on the battery voltages in the battery pack, the operational amplifier including a positive power supply terminal and a negative power supply terminal, the method comprising:

8. The operational amplifier power supply method according to claim 7, wherein the first voltage and the second voltage are generated based on a difference between both ends voltage of an i-th battery and a difference between both ends voltage of an i + 1-th battery.

9. The operational amplifier power supply method according to claim 7 or 8, wherein a drain of an N-type MOSFET is connected to a voltage of a positive terminal of the battery pack, a source of the N-type MOSFET is connected to a positive power supply terminal of the operational amplifier, and a gate of the N-type MOSFET is connected to the first voltage, the positive power supply terminal of the operational amplifier being supplied with a power supply voltage which is a difference between the first voltage and a gate-source voltage of the N-type MOSFET.

10. A battery management system, comprising: a cell voltage monitoring device according to any one of claims 1 to 9 for monitoring the voltage of individual cells of a battery.