CN111464040B - DCDC framework applicable to different input power grids and control method thereof - Google Patents
DCDC framework applicable to different input power grids and control method thereof Download PDFInfo
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- CN111464040B CN111464040B CN202010409357.0A CN202010409357A CN111464040B CN 111464040 B CN111464040 B CN 111464040B CN 202010409357 A CN202010409357 A CN 202010409357A CN 111464040 B CN111464040 B CN 111464040B
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M3/00—Conversion of dc power input into dc power output
- H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
- H02M3/44—Conversion of dc power input into dc power output with intermediate conversion into ac by combination of static with dynamic converters; by combination of dynamo-electric with other dynamic or static converters
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- B—PERFORMING OPERATIONS; TRANSPORTING
- B60—VEHICLES IN GENERAL
- B60L—PROPULSION OF ELECTRICALLY-PROPELLED VEHICLES; SUPPLYING ELECTRIC POWER FOR AUXILIARY EQUIPMENT OF ELECTRICALLY-PROPELLED VEHICLES; ELECTRODYNAMIC BRAKE SYSTEMS FOR VEHICLES IN GENERAL; MAGNETIC SUSPENSION OR LEVITATION FOR VEHICLES; MONITORING OPERATING VARIABLES OF ELECTRICALLY-PROPELLED VEHICLES; ELECTRIC SAFETY DEVICES FOR ELECTRICALLY-PROPELLED VEHICLES
- B60L53/00—Methods of charging batteries, specially adapted for electric vehicles; Charging stations or on-board charging equipment therefor; Exchange of energy storage elements in electric vehicles
- B60L53/20—Methods of charging batteries, specially adapted for electric vehicles; Charging stations or on-board charging equipment therefor; Exchange of energy storage elements in electric vehicles characterised by converters located in the vehicle
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
- H02J7/00—Circuit arrangements for charging or depolarising batteries or for supplying loads from batteries
- H02J7/02—Circuit arrangements for charging or depolarising batteries or for supplying loads from batteries for charging batteries from ac mains by converters
- H02J7/04—Regulation of charging current or voltage
- H02J7/06—Regulation of charging current or voltage using discharge tubes or semiconductor devices
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M3/00—Conversion of dc power input into dc power output
- H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
- H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
- H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
- H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
- H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
- H02M3/3353—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having at least two simultaneously operating switches on the input side, e.g. "double forward" or "double (switched) flyback" converter
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- Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
- Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
- Y02T—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO TRANSPORTATION
- Y02T10/00—Road transport of goods or passengers
- Y02T10/60—Other road transportation technologies with climate change mitigation effect
- Y02T10/70—Energy storage systems for electromobility, e.g. batteries
-
- Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
- Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
- Y02T—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO TRANSPORTATION
- Y02T10/00—Road transport of goods or passengers
- Y02T10/60—Other road transportation technologies with climate change mitigation effect
- Y02T10/7072—Electromobility specific charging systems or methods for batteries, ultracapacitors, supercapacitors or double-layer capacitors
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- Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
- Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
- Y02T—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO TRANSPORTATION
- Y02T10/00—Road transport of goods or passengers
- Y02T10/60—Other road transportation technologies with climate change mitigation effect
- Y02T10/72—Electric energy management in electromobility
-
- Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
- Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
- Y02T—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO TRANSPORTATION
- Y02T10/00—Road transport of goods or passengers
- Y02T10/80—Technologies aiming to reduce greenhouse gasses emissions common to all road transportation technologies
- Y02T10/92—Energy efficient charging or discharging systems for batteries, ultracapacitors, supercapacitors or double-layer capacitors specially adapted for vehicles
-
- Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
- Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
- Y02T—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO TRANSPORTATION
- Y02T90/00—Enabling technologies or technologies with a potential or indirect contribution to GHG emissions mitigation
- Y02T90/10—Technologies relating to charging of electric vehicles
- Y02T90/14—Plug-in electric vehicles
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- Engineering & Computer Science (AREA)
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- Transportation (AREA)
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- Dc-Dc Converters (AREA)
Abstract
The invention discloses a DCDC framework suitable for different input power grids and a control method thereof, wherein the DCDC framework comprises a primary side conversion module, a first transformer T1, a secondary side conversion module, a controller, a second transformer T2 and a secondary side rectification module, a primary side winding W5 of the second transformer T2 is connected with a primary side winding W1 of the first transformer T1 in series and then is connected with an output end of the primary side conversion module, a secondary side winding W6 of the second transformer T2 is connected with an input end of the secondary side rectification module, an output end of the secondary side rectification module is connected with an output end of the secondary side conversion module in parallel, and a primary side or secondary side of the second transformer T2 is provided with a change-over switch K; the controller controls the change-over switch K to put the second transformer T2 and the secondary rectifying module into or out of operation according to the bus voltage of the primary converting module; the invention solves the problem of multi-path parallel current sharing of the high-power vehicle-mounted OBC rear-stage DCDC and the problem of compatibility of single-phase and three-phase input voltages; has the advantages of small number of devices, simplicity, easy realization and the like.
Description
Technical Field
The invention belongs to the technical field of power supplies, and particularly relates to a DCDC architecture suitable for different input power grids and a control method thereof.
Background
With the development of society, environmental pollution and energy shortage are getting more attention, and the development of new energy automobiles is an effective way to solve the two problems. With the development of new energy automobile technology, the continuous voyage mileage is higher and higher, the capacity requirement of a power battery is higher and higher, and the charging time of the battery is shorter and shorter, so that the power of a vehicle-mounted charger (OBC for short) is rapidly required to be improved. Currently, high-power OBCs are designed by adopting multiple paths in parallel, as shown in fig. 1. In the current design, because the parameters of the components have tolerance, the problem of uneven flow can occur, so that the design of each path needs to leave a comparison allowance and additional hardware processing is added to ensure the flow equalization, and over-design is caused, so that the cost is over-high. In addition, the design composition of the current OBC is shown in fig. 2, and the current OBC consists of two stages of PFC and DCDC in series, the high-power output is based on the scene that the input alternating voltage is three-phase, and the vehicle-mounted OBC is often required to be compatible with single-phase and three-phase input. At three-phase input, PFC output voltage is typically 800V, i.e., DCDC input voltage is 800V; in single-phase input, since the transformer turn ratio of DCDC is fixed, the PFC input voltage needs to be raised to 800V at the time of single-phase input, which increases loss of PFC and results in inefficiency.
Therefore, how to design a high-power multi-path parallel current sharing method without increasing extra hardware cost and solve the problem of compatibility of primary and secondary side turn ratios of DCDC transformers with different busbar voltages of single-phase input and three-phase input is a technical problem to be solved in the industry.
Disclosure of Invention
In order to solve the defects in the prior art, the invention provides a DCDC architecture suitable for different input power grids and a control method thereof.
The technical scheme adopted by the invention is to design a DCDC architecture suitable for different input power grids, which comprises a primary side conversion module, a first transformer T1, a secondary side conversion module, a controller, a second transformer T2 and a secondary side rectification module which are sequentially connected, wherein a primary side winding W5 of the second transformer T2 is connected with a primary side winding W1 of the first transformer T1 in series and then is connected with an output end of the primary side conversion module, a secondary side winding W6 of the second transformer T2 is connected with an input end of the secondary side rectification module, an output end of the secondary side rectification module is connected with an output end of the secondary side conversion module in parallel, and a primary side or secondary side of the second transformer T2 is provided with a change-over switch K; the controller controls the change-over switch K to put the second transformer T2 and the secondary rectifying module into operation or out of operation according to the bus voltage of the primary converting module.
The switch K may be connected in parallel to two ends of the primary winding W5 of the second transformer T2.
The switch K may be connected in parallel to two ends of the secondary winding W6 of the second transformer T2.
The DCDC architecture comprises a charging mode and an inversion mode; when the bus voltage is higher than a threshold value M in a charging mode, the controller controls the switch K to be opened; when the bus voltage is not higher than a threshold value M in the charging mode, the controller controls the switch K to be closed; in the inversion mode, the controller controls the switch K to be closed.
The threshold M is 600 volts.
The ratio of the primary winding W1 to the secondary winding W2 of the first transformer T1 is equal to the ratio of the primary winding W5 to the secondary winding W6 of the second transformer T2; the number of turns of the primary winding W1 of the first transformer T1 is equal to that of the primary winding W5 of the second transformer T2, and the wire diameters are equal; the number of turns of the secondary winding W2 of the first transformer T1 is equal to that of the secondary windings W6 of the second transformer T2, and the wire diameters are equal.
The power switch in the secondary side conversion module adopts an active device, and the power switch in the secondary side rectification module adopts a passive device.
The secondary side rectifying module adopts a bridge rectifying module and comprises a first diode D1, a second diode D2, a third diode D2 and a fourth diode D4.
The primary side conversion module adopts a full-bridge structure and comprises a first power switch Q1, a second power switch Q2, a third power switch Q3 and a fourth power switch Q4; the secondary side conversion module adopts a full-bridge structure and comprises a fifth power switch Q5, a sixth power switch Q6, a seventh power switch Q7 and an eighth power switch Q8.
The secondary winding W2 of the first transformer T1 is connected with the secondary conversion module after passing through the blocking capacitor C2; the secondary winding W6 of the second transformer T2 is directly connected with the secondary rectifying module.
A resonance capacitor C1 is connected in series between the primary winding of the first transformer T1 and the primary winding of the second transformer T2.
The primary winding of the first transformer T1 is connected in series with a first resonant inductor Lr1.
The primary winding of the second transformer T2 is connected in series with a second resonant inductor Lr2.
The first transformer T1 and the second transformer T2 are integrated on the same magnetic core.
The invention also designs a control method of the DCDC framework suitable for different input power grids, and the DCDC framework adopts the DCDC framework suitable for different input power grids. During control, only the primary side conversion module and the secondary side conversion module are controlled, and the secondary side rectification module is not controlled, so that the control mode is simplified, and the cost of parts is reduced. The specific control mode is as follows: a time sequence difference phi exists between a control signal for driving the power switch in the primary side conversion module and a control signal for driving the power switch in the secondary side conversion module, and the controller controls the advance or the retard of the time sequence difference phi; the timing difference phi advances to increase the gain of the DCDC framework and the output power of the secondary side conversion module; the time sequence difference phi lag can reduce the gain of the DCDC framework and reduce the output power of the secondary side conversion module.
In a charging mode, the controller comprises a collector for collecting the input voltage (Vin) of the primary side conversion module, and a collector for collecting the output voltage (VoHV) and the output current (IoHV) which are common to the secondary side conversion module and the secondary side rectification module; the controller respectively performs sampling calibration on the output current (IoHV) and the output voltage (VoHV), and obtains output power through power operation (Power Calculation); performing difference value operation on the sampled and scaled output current (IoHV) and an output current reference value (IrefHV), performing loop compensation on the difference value of the sampled and scaled output current (IoHV) and the output current reference value (IrefHV), performing small operation on the obtained compensation value and a preset voltage loop preset value (VsetHV), and taking the small value as a voltage loop reference value (VrefHV); and carrying out difference operation on the output voltage (VoHV) after sampling and scaling and the voltage loop reference value (VrefHV), carrying out loop compensation on the difference value of the output voltage (VoHV) and the voltage loop reference value (VrefHV), calculating the time sequence difference phi by using the obtained compensation value and the output power, and driving power switches in the primary side conversion module and the secondary side conversion module through PWM operation (PWM Generator).
In an inversion mode, the controller comprises a collector for collecting the input voltage (Vin) of the primary side conversion module, and a collector for collecting the output voltage (VoHV) and the output current (IoHV) which are common to the secondary side conversion module and the secondary side rectification module; the controller performs sampling calibration on an input voltage (Vin), performs difference operation with an input current reference value (Vrefvin), performs loop compensation on the difference value of the input voltage (Vin) and the input current reference value (Vrefvin) to obtain the time sequence difference phi, and drives the power switches in the primary side conversion module and the secondary side conversion module through PWM operation (PWM Generator).
The technical scheme provided by the invention has the beneficial effects that:
the invention solves the problem of multi-path parallel current sharing of the high-power vehicle-mounted OBC rear-stage DCDC and the problem of compatibility of single-phase and three-phase input voltages under the condition of not increasing the cost; has the advantages of small number of devices, simplicity, easy realization and the like.
Drawings
The invention is described in detail below with reference to examples and figures, wherein:
FIG. 1 is a schematic block diagram of a prior art multi-channel DCDC parallel operation;
fig. 2 is a functional block diagram of a charger;
FIG. 3 is a circuit diagram of the invention with a resonant inductor in series;
FIG. 4 is a circuit diagram of the invention with two resonant inductors in series;
FIG. 5 is a circuit diagram of a combination of first and second transformers of the present invention;
FIG. 6 is a control waveform diagram of the primary side conversion module;
FIG. 7 is a control waveform diagram of the secondary side conversion module;
FIG. 8 is a control wave timing difference map for the primary and secondary side conversion modules;
FIG. 9 is a graph of secondary side conversion module control waveforms versus bridge arm midpoint voltage waveforms;
FIG. 10 is a graph of the waveforms of the respective output currents and total output currents of the secondary side first and second conversion modules;
FIG. 11 is a graph showing the waveform of the output current of the primary side conversion module, the midpoint voltage of the bridge arm of the primary side conversion module, and the midpoint voltage of the bridge arms of the first and second conversion modules of the secondary side;
FIG. 12 is a schematic diagram of control wave timing differences waveforms for a primary side conversion module and a secondary side conversion module;
fig. 13 is a waveform comparison chart of the midpoint voltage v_ef of the second conversion module bridge arm, the midpoint voltage v_cd of the first conversion module bridge arm, the primary side conversion module output voltage v_ab, the primary side conversion module output current Ip, and the DCDC architecture output current IoHV when the single-phase power grid is connected;
fig. 14 is a schematic block diagram of the controller control in the charging mode;
fig. 15 is a schematic block diagram of the controller control in the inversion mode.
Detailed Description
The present invention will be described in further detail with reference to the drawings and examples, in order to make the objects, technical solutions and advantages of the present invention more apparent. It should be understood that the specific embodiments described herein are for purposes of illustration only and are not intended to limit the scope of the invention.
The invention discloses a DCDC architecture suitable for different input power grids, which comprises a primary side conversion module, a first transformer T1, a secondary side conversion module, a controller, a second transformer T2 and a secondary side rectification module which are sequentially connected, wherein a primary side winding W5 of the second transformer T2 is connected with a primary side winding W1 of the first transformer T1 in series and then is connected with an output end of the primary side conversion module, a secondary side winding W6 of the second transformer T2 is connected with an input end of the secondary side rectification module, an output end of the secondary side rectification module is connected with an output end of the secondary side conversion module in parallel, and a primary side or secondary side of the second transformer T2 is provided with a change-over switch K; the controller controls the change-over switch K to put the second transformer T2 and the secondary rectifying module into operation or out of operation according to the bus voltage of the primary converting module.
Referring to the schematic block diagram of the charger shown in fig. 2, in the preferred embodiment, the front end of the DCDC architecture is sequentially connected to an ac input end, an EMI filter, and a PFC circuit, so as to provide dc power for the DCDC circuit. The alternating current input end is connected with different external input power grids, and can be a three-phase power grid or a single-phase power grid. The method is applied to the vehicle-mounted charger, and the output end of the DCDC circuit is mainly connected with the high-voltage battery in the vehicle.
In a preferred embodiment (see fig. 3), the switch K is connected in parallel across the primary winding W5 of the second transformer T2. In other embodiments, the switch K is connected in parallel to two ends (circuit diagram is not shown) of the secondary winding W6 of the second transformer T2.
The DCDC architecture comprises a charging mode and an inversion mode; when the bus voltage is higher than a threshold value M in a charging mode, the controller controls the switch K to be opened; when the bus voltage is not higher than a threshold value M in the charging mode, the controller controls the switch K to be closed; in the inversion mode, the controller controls the switch K to be closed. The switch K adopts one of a bidirectional switch or a relay.
In a preferred embodiment the threshold M is 600 volts. When the invention is applied to the charger, the bus voltage is higher than the threshold value M, and the representative charger is connected with a three-phase power grid. And the bus voltage is not higher than the threshold value M, and the representative charger is connected with a single-phase power grid.
In a preferred embodiment, the ratio of the primary winding W1 to the secondary winding W2 of the first transformer T1 is equal to the ratio of the primary winding W5 to the secondary winding W6 of the second transformer T2; the number of turns of the primary winding W1 of the first transformer T1 is equal to that of the primary winding W5 of the second transformer T2, and the wire diameters are equal; the number of turns of the secondary winding W2 of the first transformer T1 is equal to that of the secondary windings W6 of the second transformer T2, and the wire diameters are equal.
The power switch in the secondary side conversion module adopts an active device, and the power switch in the secondary side rectification module adopts a passive device.
In a preferred embodiment, the secondary side rectifying module adopts a bridge rectifying module, and the bridge rectifying module includes a first diode D1, a second diode D2, a third diode D2, and a fourth diode D4.
Referring to the preferred embodiment shown in fig. 3, the primary side conversion module adopts a full-bridge structure, and includes a first power switch Q1, a second power switch Q2, a third power switch Q3, and a fourth power switch Q4; the secondary side conversion module adopts a full-bridge structure and comprises a fifth power switch Q5, a sixth power switch Q6, a seventh power switch Q7 and an eighth power switch Q8. The secondary side conversion module control signals and input voltages are shown in fig. 9. The first power switch Q1, the second power switch Q2, the third power switch Q3, the fourth power switch Q4, the fifth power switch Q5, the sixth power switch Q6, the seventh power switch Q7, and the eighth power switch Q8 adopt: one of a MOSFET, a SiC MOSFET, an IGBT parallel diode and a GAN HEMT.
In a preferred embodiment, the secondary winding W2 of the first transformer T1 is connected to the secondary conversion module after passing through the blocking capacitor C2; the secondary winding W6 of the second transformer T2 is directly connected with the secondary rectifying module. As shown in fig. 3, 4, 5.
In a preferred embodiment, a resonant capacitor C1 is connected in series between the primary winding of the first transformer T1 and the primary winding of the second transformer T2. As shown in fig. 3, 4, 5.
Referring to the embodiment shown in fig. 3, the primary winding of the first transformer T1 is connected in series with a first resonant inductor Lr1. I.e. only one vibrating inductor is connected in series with the whole primary side.
Referring to the embodiment shown in fig. 4, the primary winding of the second transformer T2 is connected in series with a second resonant inductor Lr2. Namely, the whole primary side is connected with two vibrating inductors in series.
Referring to the embodiment shown in fig. 5, the first transformer T1 and the second transformer T2 are integrated on the same core. That is, the first transformer T1 and the second transformer T2 may be provided separately or may be provided in combination.
It should be noted that the resonant inductor may be provided in different embodiments by different methods, and may be one resonant inductor, two separate resonant inductors, or integrated into the same magnetic core. The resonant inductance can be an independent element or leakage inductance of the transformer.
The present invention will be described below with reference to fig. 3 by taking the use of the present invention in a charger as an example.
The charging mode is connected with a three-phase power grid:
when three phases are input, the bus voltage is 800V, namely vin=800V in fig. 3, a switch K is turned off, and Q1 and Q3 in the primary side conversion module form a primary side first bridge arm, and a bridge arm midpoint a; q2 and Q4 form a primary side second bridge arm, and a bridge arm midpoint B; the resonant inductor Lr1, the winding W1 of the transformer T1, the resonant capacitor C1 and the winding W5 of the transformer T2 are connected in series, one end of the resonant capacitor Lr2 is connected to the midpoint A of the first bridge arm of the primary side, and the other end is connected to the midpoint B of the second bridge arm of the primary side, so that the primary side series structure of the transformers T1 and T2 is formed. Q5 and Q7 in the secondary side conversion module form a secondary side first bridge arm, a bridge arm midpoint C, Q6 and Q8 form a secondary side second bridge arm, a bridge arm midpoint D, a transformer T1 winding W2 is connected in series with a blocking capacitor C2, one end of the transformer T1 winding W2 is connected to the secondary side first bridge arm midpoint C, the other end is connected to the secondary side second bridge arm midpoint D, and an output capacitor C4 is connected in parallel to form a secondary side conversion module output HV1; the secondary side rectifying module comprises a third secondary side bridge arm and a bridge arm midpoint E, wherein the third secondary side bridge arm is formed by D1 and D3; d2 and D4 form a fourth leg of the secondary side, the midpoint F of the leg; one end of a winding W6 of the transformer T2 is connected with a midpoint E of a third bridge arm of the secondary side, the other end of the winding W6 is connected with a midpoint F of a fourth bridge arm of the secondary side, a parallel output capacitor C5 forms a secondary side rectifying module to output HV2, an output positive end of HV1 and an output positive end of HV2 are connected together, an output negative end of HV1 and an output negative end of HV2 are connected together, and HV1 and HV2 are connected in parallel to form high-voltage HV output. According to the connection, the winding W2 of the transformer T1 and the winding W6 of the transformer T2 form parallel connection. Wherein the transformer T1 winding W1: w2 transformer T2 winding W5: the turn ratio and the number of turns of W6 are the same.
The control mode is as follows: in the topology of fig. 3, the controller controls HV voltage and current by driving the power switches Q1-Q4 in the primary side conversion module and the power switches Q5-Q8 in the secondary side conversion module, specifically, the primary side conversion module: q1 and Q4 are driven in unison, both are 50% duty cycles; q2 and Q3 are driven in unison, with 50% duty cycle, and Q1, Q4 and Q2, Q3 are driven in diametrically opposite directions, as shown in FIG. 6; a secondary side conversion module: q5 and Q8 are driven in unison, both are 50% duty cycles; q6 and Q7 drive are identical, both are 50% duty cycle, Q5, Q8 and Q6, Q7 drive are diametrically opposed, as shown in FIG. 7; the above-mentioned 50% duty ratio is implemented to prevent the upper and lower switching tubes of the pair of bridge arms from being simultaneously turned on and short-circuited, and a dead time is subtracted from the upper and lower switching tubes, and the above-mentioned 50% duty ratio is collectively referred to as including the dead time.
Parallel current sharing principle: primary-secondary side turn ratio W1 for two transformers T1: primary-secondary ratio of W2 to T2W 5: w6 are equal, the primary sides of the transformers are connected in series, the voltage of the input voltage Vin is applied to the resonant cavity, the primary side W1 of the transformer T1 and the primary side W5 of the transformer T2, the currents flowing through the primary windings of the two transformers are identical, and even if the resonant parameter Lr and the resonant capacitor Cr parameter deviate, the currents flowing through the two transformers are identical. The primary side turn ratio and the secondary side turn ratio of the two transformers are the same, the two paths of output are in short circuit, the output voltages of the two paths are the same, the current coupling between the transformers T1 and T2 and the secondary side is the same, namely the output current Io1 of the secondary side conversion module and the output current Io2 of the secondary side rectification module are equal, the total output current is controlled by controlling the driving of the primary side output and the secondary side output of the secondary side conversion module, and the output current of the secondary side rectification module is automatically equal to the output current of the secondary side conversion module. Because the number of turns of primary and secondary windings of the transformers T1 and T2 is the same, the currents output by the secondary conversion module and the secondary rectification module are automatically balanced, and no additional current sharing treatment is needed. In addition, the secondary side rectifying module is diode rectifying, diode conduction is determined by primary side current zero crossing, the problem of transformer magnetic bias caused by slight difference of driving of the secondary side converting module and the active device is avoided, and the secondary side rectifying module can omit a blocking capacitor in the control mode, namely, a C2 capacitor is not needed in the secondary side rectifying module.
The charging mode is connected with a single-phase power grid:
at single-phase input, the bus voltage decreases by half, i.e. vin=400V, and switch K closes. In the primary side conversion module, a primary side winding W5 of a transformer T2 is short-circuited, input voltage is applied to a resonant cavity and the primary side winding W1 of the transformer T1, parameters of the resonant cavity are the same as resonant points, and the resonant points of single-phase and three-phase input are shown in the following formula 1;
equation 1
The input voltage is reduced by half, the transformer of the primary side conversion module is reduced by half (the primary side winding W5 is short-circuited T2), the resonance point is unchanged, and when the single-phase input is carried out, the gain of the single transformer T1 is the same, compared with the three-phase input, the power of the single phase is reduced by half, the primary side current is the same, the turn ratio of the transformer T1 and the design of the winding wire diameter are the same, and the over-design caused by the compatibility of the single phase and the three phases is avoided.
Inversion mode (inversion mode does not divide three-phase or single-phase grid):
in the topology of fig. 3, switch K is closed, i.e. transformer T2 winding W5 is shorted. HV is the input of energy and Vin is the output, opposite to the charging mode. The primary side is the energy output side, and the secondary side HV is the energy input side.
The control mode is as follows: the controller controls the energy output by driving the primary side conversion module power switches Q1-Q4 and the secondary side conversion module power switches Q5-Q8. Specifically, the primary conversion modules Q1 and Q4 are driven in unison, both being 50% duty cycle; q2 and Q3 are driven identically, both at 50% duty cycle, Q1, Q4 and Q2, Q3 are driven diametrically opposite, as shown in FIG. 6. Q5 and Q8 in the secondary side conversion module are consistent in drive and are 50% duty ratio; q6 and Q7 drive are identical, both 50% duty cycle, Q5, Q8 and Q6, Q7 drive are diametrically opposed, as shown in FIG. 7. The above-mentioned 50% duty ratio is implemented to prevent the upper and lower switching tubes of the pair of bridge arms from being simultaneously turned on and short-circuited, and a dead time is subtracted from the upper and lower switching tubes, and the above-mentioned 50% duty ratio is collectively referred to as including the dead time.
Specific examples are:
the hardware framework and the control mode of the invention are applied; when operating in three phases, vin=800V, high voltage output 300V, hv output 21KW, i.e.: the high voltage HV outputs the total current 70A, and the current sharing performance of the secondary side conversion module output and the secondary side rectification module output is compared.
Table 1: simulation parameters
The simulation results are shown in table 2, and the waveforms of the output current and the total output current of the secondary side first and second conversion modules are compared with each other as shown in fig. 10. As can be seen from table 2, the current output by the secondary side conversion module and the current output by the secondary side rectification module have almost no deviation, which proves the feasibility of the control mode. Fig. 11 shows a graph of primary side conversion module output current, primary side conversion module bridge arm midpoint voltage, secondary side first and second conversion module (i.e., secondary side conversion module and secondary side rectification module) bridge arm midpoint voltage waveforms.
Table 2: simulation results
Fig. 13 is a waveform comparison chart of the midpoint voltage v_ef of the second conversion module bridge arm, the midpoint voltage v_cd of the first conversion module bridge arm, the output voltage v_ab of the primary conversion module, the output current Ip of the primary conversion module and the output current IoHV of the DCDC architecture in a state that the relay K is closed when vin=400V is connected to the single-phase power grid and the primary winding W5 of the transformer T2 is short-circuited. From the figures it can be seen that: (1) The total current of the high-voltage output is reduced by half in single-phase input compared with three-phase input, namely: the output power is reduced by half; (2) The voltage V_EF at the midpoint E, F of the output bridge arm of the secondary side rectifying module is 0, which indicates that all output power is output from the secondary side converting module; (3) The primary current Ip is the same in both single-phase and three-phase inputs.
Table 3: single-phase and three-phase input current contrast
The invention also discloses a control method of the DCDC framework suitable for different input power grids, and the DCDC framework adopts the DCDC framework suitable for different input power grids; to adjust the output power of the DCDC architecture, there is a timing difference Φ between the control signals driving the power switches in the primary side conversion module and the control signals driving the power switches in the secondary side conversion module, as shown in fig. 8 and 12. Namely, a time sequence difference phi exists between Q1 and Q4 in the primary side conversion module and Q5 and Q8 in the secondary side conversion module, and a time sequence difference phi exists between Q2 and Q3 in the primary side conversion module and Q6 and Q7 in the side first conversion module. The controller controls the advance or the retard of the time sequence difference phi; the timing difference phi advances to increase the gain of the DCDC framework and the output power of the secondary side conversion module; the time sequence difference phi lag can reduce the gain of the DCDC framework and reduce the output power of the secondary side conversion module. The primary side conversion module is used as a reference, and the time sequence difference phi is shifted left and right, and is specifically as follows: the secondary side conversion module drives the leading primary side conversion module to move right, and the secondary side conversion module drives the lagging primary side conversion module to move left. The timing difference phi advances to increase the gain of the DCDC framework and the output power of the secondary side conversion module; the time sequence difference phi lag can reduce the gain of the DCDC framework and reduce the output power of the secondary side conversion module.
Referring to fig. 14, a schematic block diagram of the controller control in the charge mode is shown. The controller comprises a collector for collecting the input voltage (Vin) of the primary side conversion module, and a collector for collecting the output voltage (VoHV) and the output current (IoHV) which are common to the secondary side conversion module and the secondary side rectification module; in a charging mode, the controller performs sampling scaling on an output current (IoHV) and an output voltage (VoHV), respectively, and obtains output power through power operation (Power Calculation); performing difference value operation on the sampled and scaled output current (IoHV) and an output current reference value (IrefHV), performing loop compensation on the difference value of the sampled and scaled output current (IoHV) and the output current reference value (IrefHV), performing small operation on the obtained compensation value and a preset voltage loop preset value (VsetHV), and taking the small value as a voltage loop reference value (VrefHV); and carrying out difference operation on the output voltage (VoHV) after sampling and scaling and the voltage loop reference value (VrefHV), carrying out loop compensation on the difference value of the output voltage (VoHV) and the voltage loop reference value (VrefHV), calculating the time sequence difference phi by using the obtained compensation value and the output power, and driving power switches in the primary side conversion module and the secondary side conversion module through PWM operation (PWM Generator).
When the secondary side conversion module current IoHV sampled by the controller is smaller than the actually set IrefHV or the sampled secondary side conversion module side voltage VoHV is smaller than the set voltage loop reference value (VrefHV), the controller moves to the right to increase the time sequence difference phi and increase the gain. On the contrary, when the secondary side conversion module side current IoHV sampled by the controller is greater than the set current reference value refHV or the sampled secondary side conversion module side voltage VoHV is greater than the set voltage loop reference value VrefHV, the controller moves to the left to increase the timing difference Φ to reduce the gain.
Referring to fig. 15, a schematic block diagram of the controller control in the inversion mode is shown. The controller comprises a collector for collecting the input voltage (Vin) of the primary side conversion module, and a collector for collecting the output voltage (VoHV) and the output current (IoHV) which are common to the secondary side conversion module and the secondary side rectification module; in the inversion mode, the controller performs sampling calibration on the input voltage (Vin), performs difference operation with an input voltage reference value (VrefVin), performs loop compensation on the difference value of the input voltage (Vin) and the input voltage reference value (VrefVin) to obtain the time sequence difference phi, and drives the power switches in the primary side conversion module and the secondary side conversion module through PWM operation (PWM Generator).
There is a timing difference Φ between the primary and secondary side conversion modules, i.e., between the primary and secondary side conversion modules Q1, Q4, Q5, Q8, and between the primary and secondary side conversion modules Q2, Q3, Q6, Q7, as shown in fig. 8 and 12. The primary side conversion module is used as a reference, and the time sequence difference phi is shifted left and right, and is specifically as follows: the secondary side conversion module drives the leading primary side conversion module to move right, and the secondary side conversion module drives the lagging primary side conversion module to move left. Shifting the inversion mode Φ right may decrease the gain (i.e., decrease the primary side conversion module output power), and shifting Φ left may increase the gain (i.e., increase the primary side conversion module output power).
And controlling the power switch cycle fs of the primary side conversion module and the secondary side conversion module while controlling phi. The primary side conversion module has the significance that as the inductance Lr1, the inductance Lr2 and the capacitance C1 exist in the primary side conversion module, the inductance Lr1, the inductance Lr2 and the capacitance C1 form a network which can change in equivalent reactance along with the change of a switching period, and the mathematical expression is as follows:
-equation 2
Where Z (fs) is the change in equivalent reactance of Lr1, lr2 and C1 with fs. In practical application, in order to improve efficiency, the Z (fs) is changed by controlling fs to further obtain optimal matching characteristics.
The above examples are illustrative only and are not intended to be limiting. Any equivalent modifications or variations to the present application without departing from the spirit and scope of the present application are intended to be included within the scope of the claims of the present application.
Claims (16)
1. The utility model provides a DCDC framework suitable for different input electric wire netting, includes primary side conversion module, first transformer T1, vice limit conversion module who connects gradually to and controller, its characterized in that: the secondary side winding W6 of the second transformer T2 is connected with the input end of the secondary side rectifying module, the output end of the secondary side rectifying module is connected with the output end of the secondary side converting module in parallel, and a change-over switch K is arranged on the primary side or the secondary side of the second transformer T2; the controller controls the change-over switch K to put the second transformer T2 and the secondary rectifying module into operation or out of operation according to the bus voltage of the primary converting module;
a time sequence difference phi exists between a control signal for driving the power switch in the primary side conversion module and a control signal for driving the power switch in the secondary side conversion module, and the controller controls the advance or the retard of the time sequence difference phi;
the controller comprises a collector for collecting the input voltage (Vin) of the primary side conversion module, and a collector for collecting the output voltage (VoHV) and the output current (IoHV) which are common to the secondary side conversion module and the secondary side rectification module; in a charging mode, the controller performs sampling scaling on an output current (IoHV) and an output voltage (VoHV), respectively, and obtains output power through power operation (Power Calculation); performing difference value operation on the sampled and scaled output current (IoHV) and an output current reference value (IrefHV), performing loop compensation on the difference value of the sampled and scaled output current (IoHV) and the output current reference value (IrefHV), performing small operation on the obtained compensation value and a preset voltage loop preset value (VsetHV), and taking the small value as a voltage loop reference value (VrefHV); and carrying out difference operation on the output voltage (VoHV) after sampling and scaling and the voltage loop reference value (VrefHV), carrying out loop compensation on the difference value of the output voltage (VoHV) and the voltage loop reference value (VrefHV), calculating the time sequence difference phi by using the obtained compensation value and the output power, and driving power switches in the primary side conversion module and the secondary side conversion module through PWM operation (PWM Generator).
2. The DCDC architecture of claim 1 adapted for different input grids, wherein: the switch K is connected in parallel with two ends of the primary winding W5 of the second transformer T2.
3. The DCDC architecture of claim 1 adapted for different input grids, wherein: the switch K is connected in parallel with two ends of the secondary winding W6 of the second transformer T2.
4. A DCDC architecture for different input grids according to claim 2 or 3, characterized in that: the DCDC architecture comprises a charging mode and an inversion mode;
when the bus voltage is higher than a threshold value M in a charging mode, the controller controls the switch K to be opened;
when the bus voltage is not higher than a threshold value M in the charging mode, the controller controls the switch K to be closed;
in the inversion mode, the controller controls the switch K to be closed.
5. The DCDC architecture for different input power grids of claim 4, wherein: the threshold M is 600 volts.
6. The DCDC architecture for different input power grids of claim 4, wherein: the ratio of the primary winding W1 to the secondary winding W2 of the first transformer T1 is equal to the ratio of the primary winding W5 to the secondary winding W6 of the second transformer T2; the number of turns of the primary winding W1 of the first transformer T1 is equal to that of the primary winding W5 of the second transformer T2, and the wire diameters are equal; the number of turns of the secondary winding W2 of the first transformer T1 is equal to that of the secondary windings W6 of the second transformer T2, and the wire diameters are equal.
7. The DCDC architecture of claim 1 adapted for different input grids, wherein: the power switch in the secondary side conversion module adopts an active device, and the power switch in the secondary side rectification module adopts a passive device.
8. The DCDC architecture of claim 7, adapted for different input grids, wherein: the secondary side rectifying module adopts a bridge rectifying module and comprises a first diode D1, a second diode D2, a third diode D2 and a fourth diode D4.
9. The DCDC architecture of claim 7, adapted for different input grids, wherein: the primary side conversion module adopts a full-bridge structure and comprises a first power switch Q1, a second power switch Q2, a third power switch Q3 and a fourth power switch Q4; the secondary side conversion module adopts a full-bridge structure and comprises a fifth power switch Q5, a sixth power switch Q6, a seventh power switch Q7 and an eighth power switch Q8.
10. The DCDC architecture of claim 1 adapted for different input grids, wherein: the secondary winding W2 of the first transformer T1 is connected with the secondary conversion module after passing through the blocking capacitor C2; the secondary winding W6 of the second transformer T2 is directly connected with the secondary rectifying module.
11. The DCDC architecture of claim 1 adapted for different input grids, wherein: a resonance capacitor C1 is connected in series between the primary winding of the first transformer T1 and the primary winding of the second transformer T2.
12. The DCDC architecture of claim 1 adapted for different input grids, wherein: the primary winding of the first transformer T1 is connected in series with a first resonant inductor Lr1.
13. The DCDC architecture for different input power grids of claim 11, wherein: the primary winding of the second transformer T2 is connected in series with a second resonant inductor Lr2.
14. The DCDC architecture of claim 1 adapted for different input grids, wherein: the first transformer T1 and the second transformer T2 are integrated on the same magnetic core.
15. The control method of the DCDC framework suitable for different input power grids is characterized by comprising the following steps of: the DCDC architecture employing the DCDC architecture of any one of claims 1 to 14 adapted for different input grids; a time sequence difference phi exists between a control signal for driving the power switch in the primary side conversion module and a control signal for driving the power switch in the secondary side conversion module, and the controller controls the advance or the retard of the time sequence difference phi; the timing difference phi advances to increase the gain of the DCDC framework and the output power of the secondary side conversion module; the time sequence difference phi lag can reduce the gain of the DCDC framework and reduce the output power of the secondary side conversion module.
16. The method for controlling a DCDC architecture adapted to different input power grids according to claim 15, wherein: the controller comprises a collector for collecting the input voltage (Vin) of the primary side conversion module, and a collector for collecting the output voltage (VoHV) and the output current (IoHV) which are common to the secondary side conversion module and the secondary side rectification module;
in the inversion mode, the controller performs sampling calibration on the input voltage (Vin), performs difference operation with an input current reference value (VrefVin), performs loop compensation on the difference value of the input voltage (Vin) and the input current reference value (VrefVin) to obtain the time sequence difference phi, and drives the power switches in the primary side conversion module and the secondary side conversion module through PWM operation (PWM Generator).
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