CN110927680B - Broadband receiving digital beam forming method based on digital deskew and frequency domain equalization - Google Patents

Abstract

The invention discloses a broadband receiving digital beam forming method based on digital de-slope and frequency domain equalization. Limited by the FPGA clock frequency, the data rate conversion is performed on the broadband intermediate frequency signal sent to the FPGA after sampling, and the signal is divided into multi-phase Representation; according to the receiving beamforming pointing requirements, the delay requirements of the corresponding array elements are calculated, and the delay is divided into integer delay and fractional delay according to the sampling frequency of the signal; the sampling signal is filtered by polyphase interpolation after integer delay. The quadrature processing method is used to realize the conversion of the intermediate frequency signal to the baseband signal; the equalization filter is designed to realize the joint correction of the transmitting and receiving channels, and the frequency domain equalization method based on the inverse Fourier transform is used to obtain the filter coefficient; The fractional delay generated by the equalization filter and multiple accumulators is used to de-slope the local oscillator phase. The CORDIC algorithm is used to realize the digital de-slope processing and fractional delay compensation of the equalized echo signal. The CORDIC output is formed by Taylor weighting and Bayliss weighting. Low side lobes and difference beams.

Description

A Digital Beamforming Method for Broadband Reception Based on Digital De-slope and Frequency Domain Equalization

technical field

The invention relates to a broadband receiving digital beam forming method based on digital de-slope and frequency domain equalization, and belongs to the technical field of array signal processing.

Background technique

The existing digital array radar using wideband LFM signal generally adopts the analog de-skew processing method at the receiving end. The method is simple and efficient, and solves the problem of receiving and processing wideband LFM signals well. On the one hand, the de-slope reception of the echo signal at the receiving end can effectively reduce the bandwidth of the received signal and the system sampling rate, and reduce the difficulty of collecting echo data and the amount of data processing; on the other hand, since the de-slope output is a frequency and The echo delay is proportional to the single-frequency signal. Therefore, only one FFT is needed to realize the pulse compression of the signal. Compared with the traditional frequency-domain matched filtering method, which requires two FFT operations, the amount of calculation and output delay are greatly reduced. .

However, the digital array radar using the analog de-slope method has complex system, high cost, high hardware overhead, and easy to introduce nonlinear errors, which leads to difficulty in calibration. In addition, the analog de-slope method is only suitable for LFM signals, which limits the use of other broadband signals, reduces the flexibility of the digital array radar system, and is not conducive to the expansion or reconstruction of the digital array radar system. With the continuous development of digital integrated circuit technology and digital signal processing technology, digital de-slope methods are gaining more and more attention. Compared with the analog de-slope method, the digital de-slope method can realize the de-slope reception of the signal in the digital domain, which not only saves the complex analog equipment, reduces the system complexity and cost, but also avoids the frequency modulation of the analog de-slope local oscillator. Problems such as nonlinearity and inconsistency of amplitude and phase improve the flexibility of the radar system, which is beneficial to the expansion or reconstruction of the radar system.

Unlike transmit digital beamforming, which uses uniform weighting to obtain maximum transmit gain, receive digital beamforming usually uses Taylor weighting and Bayliss weighting to form low sidelobe and difference beams. , the low sidelobe of the receiving beam puts forward higher requirements on the consistency between the receiving channels. In addition, the mismatch within the receive channel will also affect the pulse compression performance of the received signal in the beam direction, so a joint correction of the transmit and receive channel mismatch is required.

SUMMARY OF THE INVENTION

The invention provides a broadband receiving digital beam forming method based on digital de-slope and frequency domain equalization. The frequency domain equalization algorithm realizes the channel mismatch correction; the accumulator is used to generate the fractional delay to de-skew the local oscillator phase, and the CORDIC algorithm is used to realize the compensation of the digital de-slope and the fractional delay.

The present invention adopts the following technical solutions for solving the above-mentioned technical problems:

The present invention provides a broadband receiving digital beam forming method based on digital de-slope and frequency domain equalization, comprising the following steps:

Step 1. The polyphase representation of the sampled echo signal and the polyphase realization of the integer delay, the specific steps are as follows:

(1a) According to the band-pass sampling theorem, the bandwidth of the echo signal is B, the intermediate frequency carrier frequency is 3B, and the sampling frequency of the echo signal is 4B; the sampled echo signal is divided into four phases, and the data rate of each phase is B;

(1b) Determine the delay amount of each receiving unit relative to the reference unit according to the geometric structure of the antenna array, and decompose it into an integer delay amount and a fractional delay amount according to the sampling frequency of the echo signal, specifically:

The integer delay component T _l and fractional delay component F _l of the lth receiving unit relative to the reference unit are expressed as:

T _l =floor(τ _l /T _s )

In the formula, l=0, 1, ..., L-1, L is the number of receiving units, τ _l is the delay requirement of the l-th receiving unit, θ _t is the transmit beam direction, f _c is the radio frequency carrier frequency, T _s = 1/f _s is the echo signal sampling interval, floor( ) is rounded down;

的整数延时处理，得到数字延时信号，其中q＝0，1，…，3，mod(·)为取余函数；(1c) Perform data rearrangement and integer delay processing on the sampled echo signal: the original qth phase of the sampled echo signal becomes the mod(q+T _l , 4)th phase after data rearrangement processing , and then the integer delay amount is

The integer delay processing of , obtains a digital delay signal, where q=0, 1, ..., 3, mod( ) is the remainder function;

Step 2. Orthogonal transformation and equalization processing of the integer delay signal, specific steps:

(2a) Using the digital orthogonal transform method based on polyphase interpolation filtering to realize the orthogonal processing of integer delay signals;

(2b) using a frequency domain equalization filter to perform equalization processing on the output of step (2a);

Step 3. Digital de-slope and sum-difference beamforming based on CORDIC algorithm, specific steps:

(3a) According to the requirement of fractional delay, generate fractional delay to de-slope the local oscillator phase;

(3b) based on the output of the frequency-domain equalization filter and the fractional delay de-slope local oscillator phase generated in step (3a), the CORDIC algorithm is used to realize digital de-slope processing and fractional delay compensation of the equalized echo signal;

(3c) The output of step (3b) is subjected to Taylor weighting and Bayliss weighting to form low sidelobe and difference beams.

As a further optimization scheme of the present invention, the digital orthogonal transform based on polyphase interpolation filtering in step (2a) is specifically:

In the formula, h(4r+q') is the coefficient of the q'th phase interpolation filter, x _in (4p+q-4r-q') is the input of the q'th phase interpolation filter, y _out (4p+q) is the q-th phase output of the polyphase interpolation filter, q=0, 1, 2, 3, and N _h is the number of filter coefficients.

As a further optimization scheme of the present invention, the design of the frequency domain equalization filter in step (2b) is specifically:

x＝0，1，…，2^K′-1，2^K′为测量频点的个数，等值插值后的频域均衡滤波器频率响应H_R，l(Ix+i)表示为：Set the measurement frequency of the channel frequency response as

x=0, 1, ..., 2 ^K' -1, 2 K ^' is the number of measurement frequency points, the frequency response of the frequency domain equalization filter after equal interpolation H _{R, l} (Ix+i) is expressed as:

In the formula, _i = 0, 1, . Response, k=0, 1, ..., 2 ^K-2 -1, 2 ^K is the number of sampling points of the signal during frequency response measurement;

Assuming that the number of coefficients of the equalization filter is M, the coefficient h _l (m) of the frequency domain equalization filter of the lth receiving channel is expressed as:

h _l (m)=h _{R, l} (m)

In the formula, m=0, 1,..., M-1, n=0, 1,..., I 2 ^K' -1, IFFT[ ] is the inverse Fourier transform, w(n) is the Hamming window weight function, h _{R, l} (n) is the impulse response of the frequency-domain equalization filter.

As a further optimization scheme of the present invention, in step (3a), the generation of fractional delay de-oblique local oscillator phase is specifically:

Assuming that the output of the orthogonal transform takes the qth phase output, the generation of the fractional delay de-slope local oscillator phase is expressed as:

In the formula, q=0, 1, 2, 3, p=0, 1, ..., BT _p -1.

As a further optimization scheme of the present invention, the polyphase interpolation filter adopts a half-band filter, and its design is realized by the Filter Designer tool in MATLAB software, the sampling frequency is 4B, and the passband width is 4B.

Compared with the prior art, the present invention adopts the above technical scheme, and has the following technical effects:

1) Utilize the orthogonal transformation processing technology based on polyphase interpolation filtering to realize the conversion of intermediate frequency signal to baseband signal;

2) Using Hamming weighted frequency domain equalization technology to achieve mismatch correction of transmit and receive channels;

3) The accumulator is used to generate the fractional delay de-slope local oscillator phase, and the CORDIC algorithm is used to realize digital de-slope processing and fractional delay compensation, which avoids the direct generation of digital de-slope local oscillator and the use of multipliers.

Description of drawings

Figure 1 is a polyphase implementation of the present invention for integer delays.

FIG. 2 is a schematic diagram of the orthogonal transformation based on polyphase filtering in the present invention.

Fig. 3 is the channel characteristic measurement scheme of the present invention.

FIG. 4 is a structural block diagram of the hardware implementation of the equalization filter of the present invention.

FIG. 5 is a structural block diagram of the hardware implementation of digital de-slope and sum-difference beamforming according to the present invention.

FIG. 6 is the frequency response of the polyphase filter designed by the present invention under different quantization bit widths, wherein (a) is the amplitude response, and (b) is the phase response.

FIG. 7 is the amplitude response of the output result of the orthogonal transformation of the present invention and the Hamming weighted pulse compression output, wherein (a) is the amplitude response, and (b) is the pulse compression output (Hamming window weighting).

Fig. 8 is the probability curve of the amplitude and phase root mean square error measured by the measuring method of the present invention, wherein (a) is the amplitude root mean square error probability curve, (b) is the phase root mean square error probability curve.

FIG. 9 is a sum-difference beam pattern of the digital de-skew output of the present invention, wherein (a) is the receiving sum beam, and (b) is the phase-receiving difference beam.

FIG. 10 is a pattern diagram of the sum and difference beams before and after filtering by the equalization filter designed by the present invention, wherein (a) is the sum beam, and (b) is the difference beam.

Fig. 11 is the PSL of the sum and difference beams before filtering by the equalization filter designed by the present invention, the zero point depth of the difference beam and the root mean square error probability curve of the angle measurement, wherein (a) is the probability curve of the sum beam PSL, (b) is the difference beam PSL Probability curve, (c) is the probability curve of the difference beam null depth, (d) is the probability curve of the root mean square error of the angle measurement.

Fig. 12 is the sum and difference beam PSL after filtering by the equalization filter designed by the present invention, the zero point depth of the difference beam and the root mean square error probability curve of angle measurement, wherein (a) is the sum beam PSL probability curve, (b) is the difference beam PSL Probability curve, (c) is the probability curve of the difference beam null depth, (d) is the probability curve of the root mean square error of the angle measurement.

Detailed ways

Embodiments of the present invention are described in detail below, examples of which are illustrated in the accompanying drawings, wherein the same or similar reference numerals refer to the same or similar elements or elements having the same or similar functions throughout. The embodiments described below with reference to the accompanying drawings are exemplary and are only used to explain the present invention, but not to be construed as a limitation of the present invention.

It will be understood by those skilled in the art that, unless otherwise defined, all terms (including technical and scientific terms) used herein have the same meaning as commonly understood by one of ordinary skill in the art to which this invention belongs. It should also be understood that terms such as those defined in general dictionaries should be understood to have meanings consistent with their meanings in the context of the prior art and, unless defined as herein, are not to be taken in an idealized or overly formal sense. explain.

Below in conjunction with accompanying drawing, the technical scheme of the present invention is described in further detail:

A broadband receiving digital beam forming method based on digital de-slope and frequency domain equalization proposed by the present invention, the specific implementation steps are as follows:

Step 1. The polyphase representation of the sampled signal and the polyphase realization of the integer delay, the specific steps are as follows:

(1a) According to the band-pass sampling theorem, the signal bandwidth is taken as B, the intermediate frequency carrier frequency is taken as 3B, and the sampling frequency of the echo signal is taken as 4B. Limited by the clock frequency of the FPGA, the sampled data is sent to the FPGA for four-phase representation. The data rate of one phase is B;

(1b) Determine the delay amount of each receiving unit relative to the reference unit according to the geometric structure of the antenna array, and decompose it into an integer delay amount and a fractional delay amount according to the sampling frequency of the signal.

Assuming that the number of receiving units is L, the integer delay and fractional delay of the lth receiving unit can be expressed as:

T _l =floor(τ _l /T _s )

In the formula, l=0,1,...,L-1, τ _l is the delay requirement of the lth receiving unit, θ _t is the transmit beam direction, f _c is the radio frequency carrier frequency, T _s =1/f _s is Signal sampling interval, T _l is the integer delay component, F _l is the fractional delay component, and floor( ) is rounded down;

其中q＝0,1,…,3，mod(·)为取余函数，整数延时通过寄存器实现，具体实现参考图1。(1c) Perform data rearrangement and integer delay processing on the four-phase data, the original qth phase data of the sampled signal becomes the rearranged mod(q+ _T1,4 )th phase data, and the integer delay The amount of time is

Among them, q=0,1,...,3, mod(·) is the remainder function, and the integer delay is realized by the register. Refer to Figure 1 for the specific realization.

Step 2. Orthogonal transformation and equalization processing of the integer delay signal, specific steps:

(2a) The digital orthogonal transformation method based on polyphase interpolation filtering is used to realize the quadrature processing of the integer delay signal. Since the bandwidth of the integer delay signal is B, the four-phase complex data only needs to take one of the phases to meet the requirements. , which can save hardware multiplier resources. For the specific implementation, refer to Figure 2;

Suppose the ADC input analog IF signal is:

是信号的相位调制，f_IF是信号中频载波频率。where a(t) is the envelope of the signal,

is the phase modulation of the signal and fIF is the _IF carrier frequency of the signal.

对上述信号进行采样，则采样后的数字信号序列可以表示为：Pick

Sampling the above signal, the sampled digital signal sequence can be expressed as:

为信号的同相分量，

为信号的正交分量，将s(k)分四相表示，可以进一步简写成：In the formula,

is the in-phase component of the signal,

is the quadrature component of the signal, and s(k) is divided into four phases, which can be further abbreviated as:

s(4p+0)= _sI (4p+0)

s(4p+1)=(-1) ^x+1 s _Q (4p+1)

s(4p+2)=- _sI (4p+2)

s(4p+3)=(-1) ^x s _Q (4p+3)

仔细观察上式，可以发现，中频信号可以通过基带信号无失真的表示，正交分量的符号与x的取值有关，一般取x＝1。但是同相分量与正交分量在时间上相差半个采样点，需要在时域上进行延时补偿，延时补偿可以通过内插因子为4的多相插值滤波器的分支滤波器实现。It should be noted that the sampling frequency of each phase signal is only the same as the original sampling frequency.

Carefully observe the above formula, it can be found that the intermediate frequency signal can be represented without distortion by the baseband signal, and the sign of the quadrature component is related to the value of x, generally taking x=1. However, the time difference between the in-phase component and the quadrature component is half a sampling point, and delay compensation needs to be performed in the time domain. The delay compensation can be realized by a branch filter of a polyphase interpolation filter with an interpolation factor of 4.

Assuming that the input signal of a polyphase interpolation filter with an interpolation factor of 4 is x _in (4p+q), the filter output y _out (4p+q) can be expressed as:

In the formula, h(4r+q') is the coefficient of the q'th phase interpolation filter, x _in (4p+q-4r-q') is the input of the q'th phase interpolation filter, y _out (4p+q) is the q-th phase output of the polyphase interpolation filter, q=0, 1, 2, 3, and N _h is the number of filter coefficients. The polyphase interpolation filter adopts a half-band filter, the filter design is realized by the Filter Designer tool in MATLAB software, the sampling frequency is 4B, and the passband width is

(2b) using a frequency domain equalization filter to perform equalization processing on the output of step (2a), so as to realize joint correction of the mismatch between transmitting and receiving channels;

The traditional channel characteristics are generally measured by injecting LFM signals, but this method requires a high signal-to-noise ratio of the input signal. In view of the fact that the amplitude and phase distortion change slowly with frequency within the operating bandwidth, it can be considered to improve the measurement accuracy by measuring the amplitude and phase errors of multiple single-frequency signals of different frequencies, and then fitting the frequency of the actual channel through equal interpolation. response.

Design the measurement scheme of the channel frequency response as shown in Figure 3. Assuming that the number of measurement frequency points is 2 ^K′ , set the measurement frequency points as:

In the formula, x=0, 1, ..., 2 ^K' -1.

If the single-frequency signal with frequency f _x is sent into the transmitting channel, the coupled output signals of the L transmitting channels can be expressed as:

In the formula, A _T (f _x ) and θ _T (f _x ) are the amplitude response and phase response at the frequency point f _x after the L transmit channels are synthesized, respectively.

s _T (t) is coupled into all receiving channels after the power divider, then the output signal of the lth receiving channel can be expressed as:

In the formula, n=0, 1, ..., 2 ^K -1, 2 ^K is the effective number of sampling data, _{AR, l} (f) and θR _{, l} (f) are the frequency points of the lth receiving channel respectively magnitude and phase responses at f _x ,

The 0th phase output after the orthogonal transformation of the output signal s _{R, l, x} (n) of the lth receiving channel can be expressed as:

In the formula, k=0, 1, ..., 2 ^K-2 -1.

Taking the Fourier transform of s' _{R, l, x} (k), the frequency response of the equalization filter after equal interpolation can be expressed as:

In the formula, i=0, 1, ..., I-1, I is the interpolation multiple, S' _{R, l, x} (k) is the baseband frequency response of the x-th single-frequency signal injected by the l-th receiving channel, k =0,1,..., ^2K- 2-1.

Assuming that the number of coefficients of the equalization filter is M, the coefficient h _l (m) of the frequency domain equalization filter of the lth receiving channel is expressed as:

h _l (m)=h _R,l (m)

In the formula, m=0,1,...,M-1, n=0,1,...,I·2 ^K' -1, IFFT[·] is the inverse Fourier transform, w(n) is the Hamming window weight function, h _R,l (n) is the impulse response of the equalization filter in the frequency domain, and the implementation of the equalization filter refers to Fig. 4 for details.

Step 3. Digital de-slope and sum-difference beamforming based on CORDIC algorithm. Refer to Figure 5 for the specific implementation. The specific steps are as follows:

(3a) According to the requirement of fractional delay, the fractional delay de-slope local oscillator phase is generated by multiple accumulators.

Considering that the receiving processing of each array element is the same, the processing of the received echo signal of the lth array element is taken as an example. Considering ideal reception, assuming that there is a stationary target with a distance R in the direction of θ _t (array element 0 is the reference), the ideal echo signal received by the lth array element can be expressed as:

C＝3×10⁸m/s，f_c为射频载波频率，T_p为信号脉冲宽度。where _AR is the target scattering coefficient,

C=3×10 ⁸ m/s, f _c is the radio frequency carrier frequency, and T _p is the signal pulse width.

Considering the characteristics of the ideal receiving channel and the channel gain is 1, the wideband IF signal amplified by the mixing filter can be expressed as:

和分数延时

则采样后的信号可以表示为：Sampling s_r _IF,l (t) at sampling frequency f _s =4B, dividing τ _l into integer delays

and fractional delay

Then the sampled signal can be expressed as:

Perform T _l integer delays on s_r _{IF, l} (k), and the delay output can be expressed as:

Perform ideal orthogonal transform processing on s_r′ _IF,l (k), and the output baseband signal can be expressed as:

In the formula,

Assuming that the output of the orthogonal transform takes the qth phase output, the fractional delay de-slope local oscillator phase that needs to be generated can be expressed as:

的产生可以通过2个累加器实现，具体实现参考图5。In the formula, q=0, 1, 2, 3, p=0, 1, ..., BT _p -1. It can be seen from the above formula that the generation of the phase (4a _l +8qb+16b)p can be realized by a single accumulator, and the phase

The generation of can be realized by 2 accumulators, the specific realization refers to Figure 5.

(3b) The output of the equalization filter and the fractional delay de-slope local oscillator phase generated in step (3a) adopt the CORDIC algorithm to realize digital de-slope processing and fractional delay compensation of the echo signal.

The CORDIC algorithm is an algorithm that can solve operations such as trigonometric functions and exponential functions by simply shifting and adding operations. It can perform operations in three rotating coordinate systems, namely circular coordinate system, linear coordinate system and hyperbolic coordinate. Tie. In each coordinate system, the CORDIC algorithm has two working modes, namely the rotation mode and the vector mode. The equation of each iteration of the CORDIC algorithm in the circular rotation mode is expressed as:

x ^(i'+1) = x ^(iv) -d _i' (2 ^-i' y ^(i') )

y ^(i′+1) = y ^(i′) +d _i′ (2 ^-i′ x ^(i′) )

z ^(i′+1) = z ^(i′) -d _i ′θ ^(i′)

In the formula, x ^(i′) , y ^(i′) , z ^(i′) represent the data before the i′+1 iteration; x ^(i′+1) , y ^(i′+1) , z ^{( i'+1)} represents the data after the i'+1th iteration; θ ^(i') = arctan(2 ^-i' ), arctan() represents the arc tangent function; the symbol d _i' is a decision operator, using to determine the direction of rotation, and

After X iterations we get:

x ^(M) = K _X (x ⁽⁰⁾ cosz ⁽⁰⁾ -y ⁽⁰⁾ sinz ⁽⁰⁾ )

y ^(M) = K _X (y ⁽⁰⁾ cosz ⁽⁰⁾ +x ⁽⁰⁾ sinz ⁽⁰⁾ )

where K _X is the scaling factor,

(3c) The output of step (3b) is subjected to Taylor weighting and Bayliss weighting to form low sidelobe and difference beams.

The algorithm and processing method of the present invention have been verified, and satisfactory application effects have been achieved:

1. Experimental conditions: signal form is LFM signal, ADC quantization bit width is 14bits, signal bandwidth B=400MHz, time width is _Tp = _20.48us , sampling rate is fs=1600MHz, RF carrier frequency fc= _15GHz , intermediate frequency The frequency f _IF =1200MHz; the channel characteristic measurement is K=16, K'=6, the interpolation multiple I=128 when the channel characteristic is fitted, and the number of equalization filter coefficients M=13. Assuming that the target direction is 30°, the number of array elements L=32.

2. Simulation content:

Simulation 1: Design an interpolation filter, the filter adopts a half-band filter, based on the following parameters: pass-band cutoff frequency 200MHz, sampling frequency 1600MHz, pass-band fluctuation 0.01dB; (a) and (b) in Figure 6 give The frequency response of the designed filter under different quantization bit widths.

Simulation 2: The quantization bit width of the filter is 16 bits. Based on the filter designed above, the orthogonal transformation is performed, and the ideal baseband signal is taken as the reference object. (a) and (b) in Figure 7 show the orthogonal transformation of this patent. The magnitude response of the output results and the orthogonal transform output Hamming weighted pulse compression output. In the figure, LPG represents the pulse compression signal-to-noise ratio loss, and PSL represents the pulse compression output peak sidelobe ratio.

Simulation 3: Assuming that the signal-to-noise ratio of the input signal is 20dB when the channel characteristics are measured, and the traditional measurement method is used as the reference object, (a) and (b) in Figure 8 show the amplitude measured by the measurement method proposed in this patent. and phase rms error probability curves.

Simulation 3: Taking the ideal received sum-difference beam as the reference object, (a) and (b) in Figure 9 show the sum-difference beam pattern of the digital de-oblique output when the reference direction is 300. In the figure, PSL represents the beam direction The peak-to-side lobe ratio of the graph.

Simulation 4: The amplitude RMS error between receiving channels is 1dB, the phase RMS error is 20°, the in-band amplitude fluctuation is 1dB, and the in-band phase fluctuation is 10°. (a) and (b) in Figure 10 give Before and after equalization and difference beam patterns.

Simulation 5: In order to avoid chance, 100 simulations were performed for statistical analysis. (a) to (d) in Figure 11 show the PSL of the pre-equalization and the difference beam, the zero depth of the difference beam and the RMS error probability curve of the angle measurement. (a) to (d) in 12 give the equalized sum and difference beam PSL, difference beam null depth and angle measurement rms error probability curves.

3. Analysis of simulation results:

It can be seen from (a) and (b) in FIG. 7 that the performance of the orthogonal transform output of the present patent is basically consistent with that of an ideal baseband signal.

It can be seen from (a) and (b) in Figure 8 that, in the case of low signal-to-noise ratio, compared with the traditional measurement method, the method proposed in this patent can significantly reduce the RMS error of amplitude and phase measurement.

From (a) and (b) in Figure 9, it can be seen that compared with the ideal case, the PSL change of the sum and difference beams can be ignored, the peak loss of the sum beam is about -0.38dB, and the zero point depth of the difference beam is about -56.42dB. , which is basically consistent with the ideal and poor beam performance.

From (a) and (b) in Figure 10, (a) to (d) in 11 and (a) to (d) in Figure 12, it can be seen that the beam performance after the equalization filter designed in this patent is used. It has been greatly improved, and the performance is basically the same as that of Figure 8, and the loss of beam performance can be ignored.

The simulation results show the effectiveness of the patented broadband receiving digital beamforming method based on digital de-slope and frequency domain equalization.

The above is only a specific embodiment of the present invention, but the protection scope of the present invention is not limited to this, any person familiar with the technology can understand the transformation or replacement that comes to mind within the technical scope disclosed by the present invention, All should be included within the scope of the present invention, therefore, the protection scope of the present invention should be subject to the protection scope of the claims.

Claims (5)

1. the broadband receiving digital beamforming method based on digital de-slope and frequency domain equalization, is characterized in that, comprises the following steps: Step 1. The polyphase representation of the sampled echo signal and the polyphase realization of the integer delay, the specific steps are as follows: (1a) According to the band-pass sampling theorem, the bandwidth of the echo signal is B, the intermediate frequency carrier frequency is 3B, and the sampling frequency of the echo signal is 4B; the sampled echo signal is divided into four phases, and the data rate of each phase is B; (1b) Determine the delay amount of each receiving unit relative to the reference unit according to the geometric structure of the antenna array, and decompose it into an integer delay amount and a fractional delay amount according to the sampling frequency of the echo signal, specifically: The integer delay component T _l and fractional delay component F _l of the lth receiving unit relative to the reference unit are expressed as:

T _l =floor(τ _l /T _s )

In the formula, l=0,1,...,L-1, L is the number of receiving units, τ _l is the delay requirement of the lth receiving unit, θ _t is the transmit beam direction, f _c is the radio frequency carrier frequency, T _s = 1/f _s is the echo signal sampling interval, floor( ) is rounded down; (1c) Perform data rearrangement and integer delay processing on the sampled echo signal: the original qth phase of the sampled echo signal becomes the mod(q+ _T1,4 )th phase after data rearrangement processing , and then the integer delay amount is

The integer delay processing of , obtains a digital delay signal, where q=0,1,...,3, mod( ) is the remainder function; Step 2. Orthogonal transformation and equalization processing of the integer delay signal, specific steps: (2a) Using the digital orthogonal transform method based on polyphase interpolation filtering to realize the orthogonal processing of integer delay signals; (2b) using a frequency domain equalization filter to perform equalization processing on the output of step (2a); Step 3. Digital de-slope and sum-difference beamforming based on CORDIC algorithm, specific steps: (3a) According to the requirement of fractional delay, generate fractional delay to de-slope the local oscillator phase; (3b) based on the output of the frequency-domain equalization filter and the fractional delay de-slope local oscillator phase generated in step (3a), the CORDIC algorithm is used to realize digital de-slope processing and fractional delay compensation of the equalized echo signal; (3c) The output of step (3b) is subjected to Taylor weighting and Bayliss weighting to form low sidelobe and difference beams. 2. the broadband receiving digital beamforming method based on digital de-slope and frequency domain equalization according to claim 1, is characterized in that, in step (2a), based on the digital orthogonal transformation of polyphase interpolation filtering, be specially:

In the formula, h(4r+q') is the coefficient of the q'th phase interpolation filter, x _in (4p+q-4r-q') is the input of the q'th phase interpolation filter, y _out (4p+q) is the q-th phase output of the polyphase interpolation filter, q=0, 1, 2, 3, and N _h is the number of filter coefficients. 3. the broadband receiving digital beamforming method based on digital de-slope and frequency domain equalization according to claim 1, is characterized in that, the design of frequency domain equalization filter in step (2b), is specially: Set the measurement frequency of the channel frequency response as

x=0,1,…,2 ^K′ -1, 2 K ^′ is the number of measurement frequency points, the frequency response of the frequency domain equalization filter after equal interpolation H _R,l (Ix+i) is expressed as:

In the formula, i=0,1,...,I-1, I is the interpolation multiple, S' _R,l,x (k) is the baseband frequency of the xth single-frequency signal injected by the lth receiving channel measured Response, k=0,1,…,2 ^K-2 -1, 2 ^K is the number of sampling points of the signal during frequency response measurement; Assuming that the number of coefficients of the equalization filter is M, the coefficient h _l (m) of the frequency domain equalization filter of the lth receiving channel is expressed as:

h _l (m)=h _R,l (m) In the formula, m=0,1,...,M-1, n=0,1,...,I·2 ^K' -1, IFFT[·] is the inverse Fourier transform, w(n) is the Hamming window weight function, h _R,l (n) is the impulse response of the frequency-domain equalization filter. 4. the broadband reception digital beamforming method based on digital de-slope and frequency domain equalization according to claim 1, is characterized in that, in step (3a), fractional delay de-slopes the generation of local oscillator phase, is specially: Assuming that the output of the orthogonal transform takes the qth phase output, the generation of the fractional delay de-slope local oscillator phase is expressed as:

In the formula, q = 0, 1, 2, 3, p = 0, 1, ..., BT _p -1, and f _IF is the signal intermediate frequency carrier frequency. 5. the broadband reception digital beamforming method based on digital de-slope and frequency domain equalization according to claim 2, is characterized in that, polyphase interpolation filter adopts half-band filter, and its design adopts the Filter Designer tool in MATLAB software Implementation, sampling frequency 4B, passband width

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