CN110854483B - Broadband miniaturized 90-degree phase shifter and design method thereof - Google Patents

Broadband miniaturized 90-degree phase shifter and design method thereof Download PDF

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CN110854483B
CN110854483B CN201911186873.5A CN201911186873A CN110854483B CN 110854483 B CN110854483 B CN 110854483B CN 201911186873 A CN201911186873 A CN 201911186873A CN 110854483 B CN110854483 B CN 110854483B
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phase shifter
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CN110854483A (en
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曹媛
韩占岭
闫孝姮
谢国民
万晨
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Liaoning Technical University
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    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01P—WAVEGUIDES; RESONATORS, LINES, OR OTHER DEVICES OF THE WAVEGUIDE TYPE
    • H01P1/00—Auxiliary devices
    • H01P1/18—Phase-shifters
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01P—WAVEGUIDES; RESONATORS, LINES, OR OTHER DEVICES OF THE WAVEGUIDE TYPE
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Abstract

The invention provides a broadband miniaturized 90-degree phase shifter and a design method thereof, wherein the phase shifter comprises two independent circuit units, namely a main line unit and a reference line unit, and a lambda/4 TRD coupling line with two open ends is adopted to replace a3 lambda/4 reference line in the traditional Schiffman phase shifter. The design method comprises the following steps: firstly, a calculation formula of return loss and insertion loss of a TRD coupling line with two open ends is deduced by adopting S parameter matrix analysis of a TRD coupler, and the change rule of the TRD coupling line along with a coupling coefficient is determined; then, analyzing the performance influence of the TRD coupling line coupling coefficient and the main line odd-even mode characteristic impedance ratio on the improved miniaturized Schiffman phase shifter by adopting ADS simulation; designing the size of the phase shifter, and carrying out simulation analysis on the performance of the phase shifter; finally, the correctness of the design method is verified by adopting HFSS simulation, physical processing and test methods. Compared with the existing Schiffman phase shifter, the length of the reference line of the phase shifter provided by the invention is reduced by two thirds, and the miniaturization is fully realized.

Description

Broadband miniaturized 90-degree phase shifter and design method thereof
Technical Field
The invention relates to a phase shifter, in particular to a broadband miniaturized 90-degree phase shifter and a design method thereof.
Background
The main line unit of the traditional 90 DEG Schiffman phase shifter mainly comprises a section of lambda/4 parallel coupling line with a straight-through end and an isolation end connected together, microwave signals can generate 180 DEG phase shift through the parallel coupling line, and the reference line unit comprises a section of 3 lambda/4 uniform transmission line and can generate 270 DEG phase shift. When the input signals of the main line and the reference line are in the same phase, the transmission of the main line and the reference line can enable a phase difference of 90 degrees to exist between two paths of output signals and the output signals have broadband phase shifting characteristics, and the broadband phase shifting characteristics are widely applied to a feed network of a bandwidth circularly polarized antenna.
Disclosure of Invention
The technical problem to be solved by the invention is to provide a broadband miniaturized 90-degree phase shifter and a design method thereof aiming at the defects of the prior art, the miniaturized phase shifter is reasonable in structural design, scientific and rigorous in design and manufacture, can effectively realize the miniaturization of the phase shifter, meets more diversified use requirements, and is simple and convenient to operate and strong in practicability.
In order to solve the technical problems, the invention adopts the technical scheme that: a broadband miniaturization 90-degree phase shifter is characterized by comprising two independent circuit units, namely a main line unit and a reference line unit, wherein an input port P3 of the main line unit is connected with an output port P4 through a parallel coupling line, the parallel coupling line is composed of parallel and communicated transmission lines, two ports are arranged on each of the two transmission lines, the two ports on one transmission line are respectively an input end a1 and a through end a2, the ports on the other transmission line are a coupling end a3 and an isolation end a4, the through end a2 is connected with the isolation end a4, the input port P3 is connected with an input end a1, the output port P4 is connected with a coupling end 3, the input port P1 of the reference line unit is connected with an output port P2 through a TRD coupling line, the TRD coupling line is made by loading capacitance elements on the parallel coupling lines at equal intervals, an input end b1 and an isolation end b2 are arranged on one transmission line of the TRD coupling lines, a coupling end b3 and a through end b4 are arranged on the other transmission line, the through end b4 and the coupling end b3 are both open-circuited, the input port P1 is connected with the input end b1, the relative positions of the isolation end b2 and the through end b4 are opposite to the relative positions of the isolation end a4 and the through end a2, the output port P2 is connected with the isolation end b2, the odd-mode characteristic impedance and the even-mode characteristic impedance of the parallel coupling line in the main line unit are respectively Ze2And Zo2Electrical length of theta2The characteristic impedances of odd and even modes of TRD coupling lines in the reference line unit are respectively Ze1、Zo1Electrical length of theta1And C is the parameter of loading capacitance on the TRD coupling line, and the length of the TRD coupling line is lambda/4.
In addition, the invention also provides a design method for preparing the broadband miniaturized 90-degree phase shifter, which is characterized by comprising the following steps of:
step one, structural design and theoretical analysis of the phase shifter: analyzing and deducing a calculation formula of return loss and insertion loss of the TRD coupling line by adopting an S parameter matrix of the TRD coupler so as to determine a change rule of the TRD coupling line along with a coupling coefficient;
step two, parametric analysis of the phase shifter: analyzing the performance influence of the TRD coupling line coupling coefficient and the main line unit odd-even mode characteristic impedance ratio on the phase shifter by adopting ADS simulation;
step three, designing and manufacturing the phase shifter and analyzing the performance of the phase shifter: and calculating to obtain the physical size of the phase shifter by using transmission line comprehensive software in ADS, and modeling, simulating and analyzing by using HFSS electromagnetic simulation software.
Preferably, the structural design and theoretical analysis of the phase shifter in the first step specifically include the following operation methods:
s101, obtaining an S parameter matrix of the TRD coupler as shown in the specification
Figure GDA0003123455560000031
Where k is the coupling coefficient of the TRD coupled line. When the TRD coupling line only has an input signal at the port b1 and the normalized voltage of the input signal is 1, no signal is output at the port b2 which is an isolation terminal, and the normalized voltages of the output signals at the port b3 and the port b4 are k and k respectively
Figure GDA0003123455560000032
Two paths of output signals are totally reflected by an open load, enter a TRD coupling line, are combined in the TRD coupling line and are output by a port P1 and a port P2, and the normalized voltages of the port P1 and the port P2 are respectively 2k2-1 and
Figure GDA0003123455560000033
where-j indicates that the phase of the output signal at port P2 lags 90 deg. with respect to the input signal at port P1 when in the TRD coupled line
Figure GDA0003123455560000034
When the output signal of the port P1 is zero, ideal input and output matching characteristics can be realized;
s102, the central working frequency of the reference line unit is f0The calculation formulas of the return loss RL and the insertion loss IL are respectively as follows:
Figure GDA0003123455560000041
Figure GDA0003123455560000042
wherein S is11Voltage reflection coefficient, S, representing port P121Representing the voltage transfer coefficient from port P1 to port P2.
S103, drawing a change curve of the return loss and the insertion loss of the reference line unit along with the coupling coefficient k of the TRD coupling line, and determining the coupling coefficient k according to the change curve;
s104, calculating the odd and even mode characteristic impedance Z of the TRD coupling linee1And Zo1Calculating the parameter C of the loading capacitor: firstly, a full-pass network is realized for a main line unit under the condition that
Figure GDA0003123455560000043
Z0For the load impedance matched with the P3 and P4 ports, the S parameter of the main line unit at this time is:
S33＝S44＝0；
Figure GDA0003123455560000044
Figure GDA0003123455560000045
Figure GDA0003123455560000046
Δψ＝phase(S21)-phase(S43)＝-90°-ψ；
wherein S is33、S44Respectively representing the voltage reflection coefficients, S, of the ports P3 and P421、S43、S34Represents the voltage transfer coefficients, θ, of port P1 to port P2, port P3 to port P4, and port P4 to port P32Is the coupling line electrical length of the main line cell, Ze2And Zo2The characteristic impedance of odd and even modes of the parallel coupling lines in the main line unit is shown, Δ ψ shows the phase difference of the output signal of the phase shifter at the center operating frequency, Δ ψ is determined only by the main line unit, and when the length of the parallel coupling lines in the main line unit is also λ/4, Δ ψ is 90 °.
Loading and paralleling capacitors C between coupled lines in reference line uniteCapacitor C and parallel capacitor CeThe calculation formula of (a) is as follows:
Figure GDA0003123455560000051
Figure GDA0003123455560000052
Figure GDA0003123455560000053
Figure GDA0003123455560000054
Figure GDA0003123455560000055
Figure GDA0003123455560000056
wherein N is the loading number of the capacitor, k is the coupling coefficient, and theta1Is the coupling line electrical length.
Preferably, the parametric analysis of the phase shifter in the second step specifically includes the following steps:
s201, obtaining a change curve of an S parameter of the phase shifter along with a TRD coupling line coupling coefficient k, a change curve of an impedance bandwidth of the phase shifter along with the TRD coupling line coupling coefficient k and a change curve of a phase difference delta psi of the phase shifter along with the TRD coupling line coupling coefficient k by using ADS circuit simulation software;
s202, obtaining a change curve of a phase difference delta psi of the phase shifter along with a ratio g of even mode characteristic impedance and odd mode characteristic impedance of the main line unit by using ADS circuit simulation software;
and S203, performing parameter analysis based on the graphs obtained in S201 and S202.
Preferably, in step three, a variation curve of the S parameter of the phase shifter with the frequency and a variation curve of the phase difference of the output signal of the phase shifter with the frequency, which are simulated according to the physical model, are obtained through HFSS electromagnetic simulation software.
Compared with the prior art, the invention has the following advantages:
1. compared with the existing Schiffman phase shifter, the phase shifter designed by the invention has the advantages that the length of the reference line is reduced by two thirds, the miniaturization is fully realized, the structural design is scientific and reasonable, more various use requirements can be met, and the phase shifter can be popularized and applied.
2. The phase shifter designed by the invention can control the bandwidth of the phase shifter by adjusting the coupling coefficient of the TRD coupling line, and can effectively inhibit adjacent channel interference when in use.
3. The phase shifter designed by the invention can control the input-output matching characteristic and the phase shift ripple characteristic of the phase shifter by adjusting the coupling coefficient of the TRD coupling line, namely, the introduction of the TRD coupling line can not only reduce the circuit size, but also improve the design flexibility of the phase shifter.
The present invention will be described in further detail with reference to the accompanying drawings and examples.
Drawings
Fig. 1 is a schematic diagram of a phase shifter designed according to the present invention.
Fig. 2 is a graph showing the variation of return loss and insertion loss of a reference line unit according to the TRD coupling line coupling coefficient k in the present invention.
Fig. 3 is a graph showing the variation of the S-parameter of the phase shifter according to the present invention with the coupling coefficient k of the TRD coupling line.
Fig. 4 is a graph showing the variation of the impedance bandwidth of the phase shifter according to the present invention with the coupling coefficient k of the TRD coupling line.
Fig. 5 is a graph showing the variation of the phase difference Δ ψ of the phase shifter in accordance with the coupling coefficient k of the TRD coupling line in the present invention.
Fig. 6 is a graph showing the variation of the phase difference Δ ψ of the phase shifter of the present invention with the ratio g of the even mode characteristic impedance to the odd mode characteristic impedance of the main line unit.
FIG. 7 is a schematic diagram of a phase shifter according to the present invention.
FIG. 8 is a graph showing the variation of S-parameter with frequency of a phase shifter actually fabricated in the present invention.
Fig. 9 is a graph showing the variation of the phase difference of the output signal with frequency in the phase shifter actually fabricated in the present invention.
Fig. 10 is a schematic diagram of the unit structure of the capacitively-loaded coupled line of the present invention.
Detailed Description
As shown in fig. 1, the present invention provides a wideband miniaturized 90 ° phase shifter, which includes two independent circuit units, a main line unit and a reference line unit, the input port P3 of the main line unit is connected with the output port P4 through a parallel coupling line, the parallel coupling line is composed of parallel transmission lines which are arranged and communicated, two ports are arranged on the two transmission lines, the two ports on one transmission line are respectively an input end a1 and a straight-through end a2, the ports on the other transmission line are a coupling end a3 and an isolation end a4, the through end a2 is connected with the isolation end a4, the input port P3 is connected with the input end a1, the output port P4 is connected with the coupling end a3, the input port P1 of the reference line unit is connected to the output port P2 through a TRD coupled line, the TRD coupling line is made by loading capacitance elements on parallel coupling lines at equal intervals, and one transmission line in the TRD coupling line.An input end b1 and an isolation end b2 are arranged on the main line unit, a coupling end b3 and a through end b4 are arranged on the other transmission line, the through end b4 and the coupling end b3 are both open-circuited, an input port P1 is connected with the input end b1, the relative positions of the isolation end b2 and the through end b4 are opposite to the relative positions of the isolation end a4 and the through end a2, an output port P2 is connected with the isolation end b2, and the odd-mode characteristic impedance and the even-mode characteristic impedance of the parallel coupling line in the main line unit are respectively Ze2And Zo2Electrical length of theta2The characteristic impedances of odd and even modes of TRD coupling lines in the reference line unit are respectively Ze1、Zo1Electrical length of theta1And C is the parameter of loading capacitance on the TRD coupling line, and the length of the TRD coupling line is lambda/4.
A design method for making said broadband miniaturized 90 ° phase shifter, characterized by the steps of:
step one, structural design and theoretical analysis of the phase shifter: and analyzing and deducing a calculation formula of the return loss and the insertion loss of the TRD coupling line by adopting an S parameter matrix of the TRD coupler so as to determine the change rule of the TRD coupling line along with the coupling coefficient.
Step two, parametric analysis of the phase shifter: and analyzing the influence of the TRD coupling line coupling coefficient and the main line unit odd-even mode characteristic impedance ratio on the performance of the phase shifter by adopting ADS simulation.
Step three, designing and manufacturing the phase shifter and analyzing the performance of the phase shifter: and calculating to obtain the physical size of the phase shifter by using transmission line comprehensive software in ADS, and modeling, simulating and analyzing by using HFSS electromagnetic simulation software.
In this embodiment, the structural design and theoretical analysis of the phase shifter in the first step specifically include the following operation methods:
s101, obtaining ideal matching (S)110) and isolation characteristics (S)210), the TRD coupler is implemented with the condition that the odd-even mode electrical lengths of the TRD coupler differ by pi. According to the microwave network theory, the S parameter matrix of the TRD coupler can be deduced as follows
Figure GDA0003123455560000091
Where k is the coupling coefficient of the TRD coupled line. As shown in FIG. 1, when the TRD coupling line has only the input signal at the port b1 and the normalized voltage of the input signal is 1, the port b2 is an isolated terminal and has no signal output, and the normalized voltages of the output signals at the port b3 and the port b4 are k and k respectively
Figure GDA0003123455560000092
Two paths of output signals are totally reflected by an open load, enter a TRD coupling line, are combined in the TRD coupling line and are output by a port P1 and a port P2, and the normalized voltages of the port P1 and the port P2 are respectively 2k2-1 and
Figure GDA0003123455560000093
where-j indicates that the phase of the output signal at port P2 is 90 degrees behind the input signal at port P1 when in the TRD coupled line
Figure GDA0003123455560000094
When the output signal of the port P1 is zero, the ideal input/output matching characteristic can be realized.
S102, the central working frequency of the reference line unit is f0The calculation formulas of the return loss RL and the insertion loss IL are respectively as follows:
Figure GDA0003123455560000095
Figure GDA0003123455560000096
wherein S is11Voltage reflection coefficient, S, representing port P121Representing the voltage transfer coefficient from port P1 to port P2.
S103, drawing a graph 2 of the change of the return loss and the insertion loss of the reference line unit with the coupling coefficient k of the TRD coupling line, and as can be seen from fig. 2, the smaller the coupling coefficient of the TRD coupling line is, the larger the return loss of the reference line unit is, i.e., the better the matching performance of the input/output port is, and the smaller the insertion loss is. When the design index requires that the input return loss of the phase shifter in the whole working frequency band is more than 10dB, the coupling coefficient k of the TRD coupling line is less than 0.81, and the insertion loss of the reference line unit is less than 0.45 dB. However, reducing the coupling coefficient of the TRD coupled line will reduce the bandwidth of the phase shifter, so the phase shifter must be designed with a compromise between matching performance and bandwidth, and an appropriate coupling coefficient must be selected.
S104, after the coupling coefficient is determined, the odd-even mode characteristic impedance Z of the TRD coupling line is calculated according to the odd-even mode analysis methode1And Zo1And a parameter C of the loading capacitance. Fig. 10 is a schematic diagram of a unit structure of the capacitively-loaded coupled line. In the figure Ze1And Zo1The characteristic impedances of the even and odd modes of the parallel coupled lines, respectively. At the central position, a capacitor C is loaded between the coupled lines and connected in parallele。
The calculation formula is as follows:
Figure GDA0003123455560000101
Figure GDA0003123455560000102
Figure GDA0003123455560000103
Figure GDA0003123455560000104
Figure GDA0003123455560000105
Figure GDA0003123455560000111
wherein N is the loading number of the capacitor, k is the coupling coefficient, and theta1Is the coupling line electrical length.
Based on the steps, design parameters of all parts of the TRD coupler based on the periodic capacitive loading can be calculated.
Since the more the number of the nodes of the capacitive loading coupling line unit, the wider the operating bandwidth of the TRD coupling line, but when the number of the nodes is changed from 3 to 6, the increased bandwidth is small, and the insertion loss of the TRD coupling line will increase with the increase of the number of the nodes, so that the 3-unit loading is most suitable to be selected. In addition, when the electrical length θ of the parallel coupled lines is1When it is equal to pi/2, the loading capacitance C shown in FIG. 10eWill become zero. To eliminate loading capacitance CeIn order to reduce the complexity of the phase shifter, the lambda 4 parallel coupling line is adopted to realize the TRD coupling line in the improved reference line unit structure.
S105, the main line unit can realize a full-pass network, and the realization condition is that
Figure GDA0003123455560000112
Z0The load impedance is matched with the ports P3 and P4, and the S parameter of the main line unit is
S33＝S44＝0
Figure GDA0003123455560000113
Figure GDA0003123455560000114
Figure GDA0003123455560000115
Δψ＝phase(S21)-phase(S43)＝-90°-ψ
Wherein S is33、S44Respectively representing the voltage reflection coefficients, S, of the ports P3 and P421、S43、S34Is a voltage transmission coefficient, θ2Is the main coupling line electrical length. Δ ψ represents an output signal phase difference of the phase shifter at the center operating frequency, Δ ψ is determined only by the main line cell, and Δ ψ is 90 ° when the parallel coupling line length in the main line cell is also λ/4.
Preferably, the parametric analysis of the phase shifter in the second step specifically includes the following steps:
s201, obtaining a change curve of an S parameter of the phase shifter along with a TRD coupling line coupling coefficient k by using ADS circuit simulation software as shown in fig. 3, a change curve of an impedance bandwidth of the phase shifter along with the TRD coupling line coupling coefficient k as shown in fig. 4, and a change curve of a phase difference delta psi of the phase shifter along with the TRD coupling line coupling coefficient k as shown in fig. 5, wherein the invariant parameters are g 1.0 and theta is1＝π/2，θ2As can be seen from fig. 3, the coupling coefficient k of the TRD coupling line is related to the input matching characteristic (i.e., S) at the central operating frequency of the phase shifter11) The influence is great. As can be seen from FIG. 4, | S increases as k increases from 0.71 to 0.8011The impedance bandwidth of | < -10dB is improved from 33.6% to 42.1%, but excessive coupling (k > 0.81) will deteriorate the input matching characteristic | S at the central operating frequency of 1GHz11I will be greater than-10 dB; as can be seen from fig. 5, when k increases from 0.71 to 0.80, the bandwidth of 90 ° ± 10 ° phase difference increases from 32.6% to 39.6%, and the bandwidth of 90 ° ± 5 ° phase difference increases from 28.0% to 37.2%. Furthermore, as can be seen from fig. 3 and 5, two transmission zeros exist near 0.73 GHz and 1.23GHz, which can be used to suppress adjacent channel interference, but the transmission zero positions also correspond to phase jumps, which makes the operating bandwidth of the phase shifter designed by the present invention less than 50%.
S202, obtaining a variation curve of the phase difference delta phi of the phase shifter along with the ratio g of the even-mode characteristic impedance and the odd-mode characteristic impedance of the main line unit by using ADS circuit simulation software, wherein k is 0.8 and theta is a constant parameter, and the variation curve is shown in FIG. 61＝π/2，θ2As can be seen from fig. 6, as g increases, the ripple of the phase difference curve around the center operating frequency increases. The variation law is consistent with the characteristics of the Schiffman phase shifter with the traditional structure. When g increases from 1.0 to 1.6, the phase difference isThe bandwidth of 90 +/-10 degrees is improved from 39.6 percent to 42.2 percent;
s203, based on the comprehensive analysis of fig. 3 to fig. 6, the maximum operation bandwidth of the phase shifter designed by the present invention is about 42%. Although this is less than the operating bandwidth of the existing Schiffman phase shifter, the length of the reference line of the phase shifter designed by the invention is reduced by two thirds, and the 90 ° phase shifter with medium bandwidth is also needed in practical application. For example, the frequency range covered by a Global Positioning System (GPS) circularly polarized broadband antenna which simultaneously works in L1, L2 and L5 bands is 1.164 to 1.587GHz (the relative bandwidth is 30.8%), the working frequency range of an ultrahigh frequency radio frequency identification reader antenna is 840 to 960MHz (the relative bandwidth is 13.3%), and the feeding network can be designed by adopting the miniaturized Schiffman phase shifter provided herein.
In this embodiment, the third step specifically includes the following implementation processes: determining the central working frequency of the phase shifter to be 1.375GHz and the characteristic impedance Z of the port0Equal to 50 omega. The dielectric substrate used to carry the phase shifters was 1.5mm thick and had a relative dielectric constant of 2.65. In order to make the return loss at the central operating frequency larger than 24dB, the coupling coefficient k of the TRD coupling line is selected to be 0.728, and the electrical length theta of the coupling microstrip line in the TRD coupling line is selected1Chosen to be pi/2 to eliminate its loading capacitance, thereby reducing the complexity of the circuit. When k and theta1After the determination, according to the design step of the TRD coupling line, calculating the electrical parameter of the TRD coupling line to be Ze1＝126.0Ω、Zo174.0 Ω and 2.7 pF. To achieve the 90 ° phase shift function and to reduce the size, θ2Also chosen to be pi/2. At this time, the ratio g of the main line even-mode characteristic impedance to the main line odd-mode characteristic impedance can be adjusted to obtain a proper operation bandwidth and a small phase ripple. After the electric parameters of the TRD coupling line and the main coupling line are modeled, simulated and adjusted in the ADS, g is determined to be 1.55 so as to meet the working frequency coverage of 1.164-1.587 GHz required by the application of a GPS system. In use mode
Figure GDA0003123455560000141
And
Figure GDA0003123455560000142
calculating the characteristic impedance of odd-even mode of the main coupling line as Zo2＝40.2Ω、Ze262.2 Ω. And then calculating the physical size of each transmission line by using transmission line comprehensive software in the ADS according to the transmission line characteristic parameters. To take into account the effects of transmission line discontinuities, simulations and analyses were modeled in the electromagnetic simulation software HFSS. Finally, the physical dimensions of the circuit shown in FIG. 7 are: w0＝4.07mm，W1＝1.04mm，W2＝3.73mm，W3＝0.50mm，S1＝0.76mm，S2＝0.60mm，L1＝37.5mm，L237.1 mm. According to the structural parameters, a TRD coupling line-based miniaturized phase shifter is manufactured, and the circuit size is 58mm multiplied by 42 mm.
The change curve of the S parameter curve of the phase shifter along with the frequency is obtained by HFSS simulation software for the designed circuit and is shown in figure 8, and the change curve of the phase difference curve of the phase shifter along with the frequency is shown in figure 9, and as can be seen from figure 8, the frequency range of the phase shifter with the return loss larger than 10dB is 1.133-1.587 GHz (the relative bandwidth is 33.3%), and the insertion loss in the frequency range is less than 1.4 dB; the frequency range with the insertion loss less than 3dB is 1.077-1.606 GHz (the relative bandwidth is 39.4%). As can be seen from fig. 9, the phase difference generated when the two paths of the phase shifter transmit signals is stabilized near 90 °, the frequency range of the phase difference fluctuation between ± 5 ° is 1.136 to 1.584GHz (relative bandwidth is 32.9%), the frequency range of the phase difference fluctuation between ± 10 ° is 1.114 to 1.598GHz (relative bandwidth is 35.7%), the operating frequency band of the GPS system (L1, L2, and L5 bands) is completely covered, and the requirements of practical applications can be fully satisfied.
The above description is only for the preferred embodiment of the present invention, and is not intended to limit the present invention in any way. Any simple modification, change and equivalent changes of the above embodiments according to the technical essence of the invention are still within the protection scope of the technical solution of the invention.

Claims (4)

1. A broadband miniaturization 90-degree phase shifter is characterized by comprising a main line unit and a reference line unitAn input port P3 of the main line unit is connected with an output port P4 through parallel coupling lines, the parallel coupling lines are composed of parallel transmission lines which are arranged in parallel and communicated, two ports are arranged on the two transmission lines, the two ports on one transmission line are respectively an input end a1 and a through end a2, the ports on the other transmission line are a coupling end a3 and an isolation end a4, the through end a2 is connected with the isolation end a4, the input port P3 is connected with an input end a1, the output port P4 is connected with a coupling end a3, the input port P1 of the reference line unit is connected with an output port P2 through a TRD coupling line, the TRD coupling line is made by loading capacitance elements on the parallel coupling lines at equal intervals, one transmission line in the TRD coupling line is provided with an input end b1 and an isolation end b2, the other transmission line is provided with a coupling end b3 and a through end b4, the through end b4 and the coupling end b3 are both open-circuited, the input port P1 is connected with the input end b1, the relative positions of the isolation end b2 and the through end b4 are opposite to the relative positions of the isolation end a4 and the through end a2, the output port P2 is connected with the isolation end b2, the odd-mode characteristic impedance and the even-mode characteristic impedance of the parallel coupling line in the main line unit are respectively Ze2And Zo2Electrical length of theta2The characteristic impedances of odd and even modes of TRD coupling lines in the reference line unit are respectively Ze1、Zo1Electrical length of theta1And C is the parameter of loading capacitance on the TRD coupling line, and the length of the TRD coupling line is lambda/4.
2. A design method for preparing a broadband miniaturized 90 ° phase shifter according to claim 1, comprising the steps of:
step one, structural design and theoretical analysis of the phase shifter: analyzing and deducing a calculation formula of return loss and insertion loss of the TRD coupling line by adopting an S parameter matrix of the TRD coupler so as to determine a change rule of the TRD coupling line along with a coupling coefficient; the specific operation is as follows:
s101, obtaining an S parameter matrix of the TRD coupler as shown in the specification
Figure FDA0003123455550000021
In the formula, k is the coupling coefficient of the TRD coupling line; when the TRD coupling line only has an input signal at the port b1 and the normalized voltage of the input signal is 1, no signal is output at the port b2 which is an isolation terminal, and the normalized voltages of the output signals at the port b3 and the port b4 are k and k respectively
Figure FDA0003123455550000022
Two paths of output signals are totally reflected by an open load, enter a TRD coupling line, are combined in the TRD coupling line and are output by a port P1 and a port P2, and the normalized voltages of the port P1 and the port P2 are respectively 2k2-1 and
Figure FDA0003123455550000023
where-j indicates that the phase of the output signal at port P2 is 90 degrees behind the input signal at port P1 when in the TRD coupled line
Figure FDA0003123455550000024
When the output signal of the port P1 is zero;
s102, the central working frequency of the reference line unit is f0The calculation formulas of the return loss RL and the insertion loss IL are respectively as follows:
Figure FDA0003123455550000027
Figure FDA0003123455550000025
s103, drawing a change curve of the return loss and the insertion loss of the reference line unit along with the coupling coefficient k of the TRD coupling line, and determining the coupling coefficient k according to the change curve;
s104, calculating the odd and even mode characteristic impedance Z of the TRD coupling linee1And Zo1Calculating the parameter C of the loading capacitor: firstly, the main line unit is realized with a full-through networkUnder the conditions of
Figure FDA0003123455550000026
Z0For the load impedance matched with the P3 and P4 ports, the S parameter of the main line unit at this time is:
S33＝S44＝0；
Figure FDA0003123455550000031
Figure FDA0003123455550000032
Figure FDA0003123455550000033
Δψ＝phase(S21)-phase(S43)＝-90°-ψ；
wherein S is33、S44、S11Respectively representing the voltage reflection coefficients, S, of the ports P3, P4 and P121、S43、S34Representing the voltage transmission coefficient, theta2Is the main coupling line electrical length, Ze2And Zo2The characteristic impedance of odd and even modes of the parallel coupling lines in the main line unit is shown, delta phi represents the phase difference of output signals of the phase shifter at the central working frequency, the delta phi is determined by the main line unit only, and when the length of the parallel coupling lines in the main line unit is also lambda/4, the delta phi is 90 degrees;
loading and paralleling capacitors C between coupled lines in reference line uniteCapacitor C and parallel capacitor CeThe calculation formula of (a) is as follows:
Figure FDA0003123455550000034
Figure FDA0003123455550000035
Figure FDA0003123455550000036
Figure FDA0003123455550000037
Figure FDA0003123455550000041
Figure FDA0003123455550000042
wherein N is the loading number of the capacitor, k is the coupling coefficient, and theta1Is the coupling line electrical length;
step two, parametric analysis of the phase shifter: analyzing the performance influence of the TRD coupling line coupling coefficient and the main line unit odd-even mode characteristic impedance ratio on the phase shifter by adopting ADS simulation;
step three, designing and manufacturing the phase shifter and analyzing the performance of the phase shifter: and calculating to obtain the physical size of the phase shifter by using transmission line comprehensive software in ADS, and modeling, simulating and analyzing by using HFSS electromagnetic simulation software.
3. The method as claimed in claim 2, wherein the parametric analysis of the phase shifter in step two comprises the following steps:
s201, obtaining a change curve of an S parameter of the phase shifter along with a TRD coupling line coupling coefficient k, a change curve of an impedance bandwidth of the phase shifter along with the TRD coupling line coupling coefficient k and a change curve of a phase difference delta psi of the phase shifter along with the TRD coupling line coupling coefficient k by using ADS circuit simulation software;
s202, obtaining a change curve of a phase difference delta psi of the phase shifter along with a ratio g of even mode characteristic impedance and odd mode characteristic impedance of the main line unit by using ADS circuit simulation software;
and S203, performing parameter analysis based on the graphs obtained in S201 and S202.
4. The design method of a broadband miniaturized 90 ° phase shifter according to claim 2, wherein the variation curve of the S parameter of the phase shifter with frequency and the variation curve of the phase difference of the output signal of the phase shifter with frequency are obtained by HFSS electromagnetic simulation software according to the simulation of a physical model in step three.
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