CN110692185A - DC voltage conversion device - Google Patents
DC voltage conversion device Download PDFInfo
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- CN110692185A CN110692185A CN201880012565.8A CN201880012565A CN110692185A CN 110692185 A CN110692185 A CN 110692185A CN 201880012565 A CN201880012565 A CN 201880012565A CN 110692185 A CN110692185 A CN 110692185A
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M3/00—Conversion of dc power input into dc power output
- H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
- H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
- H02M3/10—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
- H02M3/145—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
- H02M3/155—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
- H02M3/156—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
- H02M3/158—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
- H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
- H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
- H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
- H02M7/4811—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode having auxiliary actively switched resonant commutation circuits connected to intermediate DC voltage or between two push-pull branches
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M1/00—Details of apparatus for conversion
- H02M1/0048—Circuits or arrangements for reducing losses
- H02M1/0054—Transistor switching losses
- H02M1/0058—Transistor switching losses by employing soft switching techniques, i.e. commutation of transistors when applied voltage is zero or when current flow is zero
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- Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
- Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
- Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
- Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
- Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes
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- Engineering & Computer Science (AREA)
- Power Engineering (AREA)
- Dc-Dc Converters (AREA)
Abstract
A DC voltage conversion device (10) is provided with: a first switching element (SW1) connected to the internal power supply line Ls; a low-pass filter (13); an auxiliary resonant circuit (20) including an auxiliary switching element (SA) and an auxiliary reactor (LA) connected in series, and having one end (La1) of the auxiliary reactor (LA) connected to a first connection point (N1) which is a connection point between the first switching element (SW1) and the low-pass filter (13); and a switch control unit (11) that performs zero-voltage switching on the first switching element (SW1) and zero-current switching on the auxiliary switching element (SA). The auxiliary resonant circuit (20) includes an auxiliary Capacitor (CA) connected between the second terminal (D) of the auxiliary switching element (SA) and the ground line (Lg) or the internal power supply line (Ls).
Description
Technical Field
The technology disclosed in the present specification relates to a dc voltage conversion device, and more particularly, to a dc voltage conversion device including an auxiliary resonant circuit.
Background
Conventionally, in order to suppress switching loss and high-frequency noise generated by switching by causing a switching element to perform so-called soft switching, a method of providing an auxiliary resonant circuit in a dc voltage conversion device has been widely performed. As a DC voltage conversion device including such an auxiliary resonant circuit, for example, a DC-DC converter (DC voltage conversion device) disclosed in patent document 1 is known. In the DC-DC converter of this document, the period during which the second main switch (low-side switch) and the auxiliary switch included in the auxiliary resonant circuit are simultaneously turned on is optimized based on the current flowing to the smoothing reactor, thereby preventing the occurrence of an excessive power loss.
Prior art documents
Patent document
Patent document 1: japanese patent laid-open publication No. 2004-129393
Disclosure of Invention
Problems to be solved by the invention
However, the DC-DC converter of the above document becomes an energy source of the auxiliary resonant circuit, that is, the power supply becomes the output voltage Vout. Since the output voltage Vout is a voltage obtained by converting the input voltage Vin, the loss in the auxiliary resonant circuit is added as a conversion loss. This is not necessarily advantageous for the high efficiency of the converter. In the configuration in which the output voltage Vout is applied to the auxiliary resonant circuit as described above, when the input voltage Vin is converted into the very low output voltage Vout, the voltage at the point M in fig. 1 of document 1 does not reach the input voltage Vin even if the output voltage Vout is applied to the auxiliary resonant circuit, and there is a possibility that soft switching (zero-voltage switching) of the first main switch S1 cannot be performed. In this case, the switching loss in the first main switch S1 increases.
In recent years, when a dc voltage conversion device is used as a power circuit such as an electric vehicle that requires a large current, in other words, when used for a large capacitive load, the dc voltage conversion device causes a small conversion loss with a large reduction in conversion efficiency. Therefore, a dc voltage conversion device capable of maintaining high conversion efficiency even when applied to a large capacitive load is desired.
The technology disclosed in the present specification has been made in view of the above-described circumstances, and provides a dc voltage conversion device capable of maintaining efficient conversion efficiency when applied to a large capacitive load.
Means for solving the problems
A dc voltage conversion device disclosed in the present specification is a dc voltage conversion device that converts a dc input voltage applied from a main power supply into an output voltage having a predetermined voltage value, and includes: an internal power cord connected to the main power supply; a first switching element connected to the internal power line; a low-pass filter having one end connected to one end of the first switching element; an auxiliary resonant circuit including an auxiliary switching element and an auxiliary reactor connected in series, one end of the auxiliary reactor being connected to a first connection point which is a connection point of the first switching element and the low-pass filter; a reflow part connected between the first connection point and a ground line; and a switching control unit that performs zero-voltage switching on the first switching element and performs zero-current switching on the auxiliary switching element, wherein the auxiliary switching element of the auxiliary resonant circuit includes: a first terminal connected to the other end of the auxiliary reactor; and a second terminal that turns on and off a connection with respect to the first terminal, the auxiliary resonance circuit including an auxiliary capacitor connected between the second terminal of the auxiliary switching element and a ground line or the internal power supply line.
According to this configuration, the auxiliary capacitor is connected to the auxiliary switching element of the auxiliary resonant circuit. Therefore, energy related to the auxiliary resonance can be supplied from the auxiliary capacitor to the auxiliary resonance circuit. Then, the first switching element is zero-voltage-switched, and the auxiliary switching element is zero-current-switched. Therefore, according to the dc voltage conversion device of the present configuration, even when applied to a large capacitive load, efficient conversion efficiency can be maintained.
In the above dc voltage converter, the switching control unit may generate a control signal for turning on and off the auxiliary switching element, and an on period of the control signal may include a charging period in which the auxiliary capacitor is charged with a first current flowing to the first switching element that exceeds at least an output current.
According to this configuration, the auxiliary capacitor can be charged with the surplus energy generated at the time of voltage conversion, and the auxiliary resonant circuit can be operated by the energy stored in the auxiliary capacitor. This makes it possible to effectively utilize surplus energy associated with voltage conversion, and to improve conversion efficiency.
In the above dc voltage converter, the switching control unit may generate the control signal in a discharge period in which the auxiliary capacitor is discharged and in the on period in which the control signal is generated in the charge period.
According to this configuration, the period of the current flowing to the auxiliary switching element can be shortened by dividing the discharge period in which the auxiliary capacitor is discharged and the charge period in which the gate control signal of the auxiliary switching element is generated into the on period. This can reduce the conduction loss of the auxiliary switching element.
In the dc voltage converter, the auxiliary capacitor may be connected between the auxiliary switching element and the ground line.
According to this configuration, the charge and discharge control of the auxiliary capacitor can be performed more easily than in the case where the auxiliary capacitor is connected between the auxiliary switching element and the internal power supply line.
In the above dc voltage conversion device, the dc voltage conversion device may further include a voltage stabilizing diode connected between a second connection point, which is a connection point between the auxiliary reactor and the auxiliary switching element, and the ground line.
According to this configuration, the potential of the second connection point can be stabilized while the auxiliary switching element is off by the voltage stabilizing diode. That is, when the MOSFET is used as the auxiliary switching element while the auxiliary switching element is off and the second switching element is on, it is conceivable that the potential of the second connection point varies via the parasitic capacitance of the auxiliary switching element. In this case, the voltage stabilizing diode can suppress variation in the potential of the second connection point.
In the dc voltage converter, the auxiliary switching element may be two auxiliary switching elements connected in series and controlled by the switching control unit with the same control signal.
According to this configuration, the auxiliary switching element is configured by two auxiliary switching elements connected in series, whereby the on-resistance as the auxiliary switching element increases. Thus, by reducing the on current flowing when the auxiliary switching element is turned on, the on resistance loss of the auxiliary switching element can be reduced as compared with the case where there is one auxiliary switching element. Since the loss (electric power) is proportional to the square of the current, in this case, the amount of decrease in the on-resistance loss due to a decrease in the on-current is larger than the amount of increase in the on-resistance loss due to an increase in the on-resistance. Therefore, the on-resistance loss can be reduced.
In the dc voltage converter, the return unit may be configured by a second switching element that is zero-voltage-switched by the switching control unit.
According to this configuration, the on/off control of the second switching element can appropriately set the reflux period, and the zero-voltage switching can suppress the switching loss of the second switching element.
Further, the dc voltage converter may further include: a first parallel capacitor connected in parallel to the first switching element; and a second parallel capacitor connected in parallel with the reflow part.
According to this configuration, the speed of the potential change of the first connection point in the so-called dead time can be adjusted when the first switching element and the second switching element are turned off.
Effects of the invention
According to the dc voltage conversion device disclosed in the present specification, when applied to a large capacitive load, high conversion efficiency can be maintained.
Drawings
Fig. 1 is a schematic circuit diagram showing a dc voltage conversion device according to embodiment 1.
Fig. 2 is a schematic timing chart showing the operation of the dc voltage conversion device.
Fig. 3 is a schematic partial circuit diagram showing the flow of current in the dc voltage conversion device.
Fig. 4 is a schematic partial circuit diagram showing the flow of current in the dc voltage conversion device.
Fig. 5 is a schematic partial circuit diagram showing the flow of current in the dc voltage conversion device.
Fig. 6 is a schematic partial circuit diagram showing the flow of current in the dc voltage conversion device.
Fig. 7 is a schematic partial circuit diagram showing the flow of current in the dc voltage conversion device.
Fig. 8 is a schematic partial circuit diagram showing the flow of current in the dc voltage conversion device.
Fig. 9 is a schematic partial circuit diagram showing the flow of current in the dc voltage conversion device.
Fig. 10 is a schematic time chart showing the operation of the dc voltage conversion device according to embodiment 2.
Fig. 11 is a partial circuit diagram showing another embodiment of the auxiliary capacitor.
Fig. 12 is a partial circuit diagram showing another embodiment of the auxiliary capacitor.
Detailed Description
< embodiment 1>
A dc voltage conversion device 10 according to embodiment 1 will be described with reference to fig. 1 to 9.
1. Structure of DC voltage conversion device
In the present embodiment, the DC voltage conversion device 10 is a so-called chopper-type step-down DC-DC converter, and steps down an input voltage Vin of a DC power applied from a battery serving as the main power supply 40 to convert the input voltage into an output voltage Vout of the DC power having a predetermined voltage value. The input voltage Vin is, for example, 48V, and the output voltage Vout is, for example, 24V.
In the present embodiment, the dc voltage conversion device 10 is disposed for use in, for example, an HV vehicle in which a gasoline engine and a traveling motor are mounted, and is applied to a power circuit that supplies electric power to a large-capacity load 50 such as the traveling motor. The application of the dc voltage conversion device 10 is not limited to the HV vehicle, and is not limited to the vehicle. Further, the load to which the dc voltage conversion device 10 is applied is not necessarily limited to a large capacitance load. The present invention is not limited to the step-down DC-DC converter, and may be applied to a step-up DC-DC converter as described later.
As shown in fig. 1, the dc voltage conversion device 10 includes an internal power supply line Ls, a first switching element SW1, a first parallel capacitor C1, a switch control unit 11, a low-pass filter 13, a reflux unit 14, and an auxiliary resonant circuit 20.
The internal power supply line Ls is connected to the battery 40, and supplies electric power from the battery 40 to each unit of the dc voltage conversion device 10.
The first switching element SW1 is formed of an N-channel MOSFET including a body diode D1 in the present embodiment. The drain terminal D of the first switching element SW1 is connected to the internal power supply line Ls.
The first parallel capacitor C1 is connected in parallel with the first switching element SW 1. The first parallel capacitor C1 is not limited to a single element, and may be a parasitic capacitor of the first switching element SW 1.
The low-pass filter 13 has a known structure, and is configured of, for example, a smoothing reactor Lo and a smoothing capacitor Co as shown in fig. 1. An input terminal 13a (corresponding to one end of the low-pass filter) of the low-pass filter 13 is connected to a source terminal S (corresponding to one end of the first switching element) of the first switching element SW1, and an output terminal 13b of the low-pass filter 13 is connected to an output terminal of the dc voltage conversion device 10. The low-pass filter 13 receives the potential Vn1 of the first node N1, i.e., the first node voltage Vn1, and outputs the output voltage Vout obtained by smoothing the first node voltage Vn 1.
The return portion 14 is connected between the first connection point N1 and the ground line Lg, and has a known configuration. In the present embodiment, as shown in fig. 1, the return unit 14 is constituted by the second switching element SW 2. The second switching element SW2 is formed of an N-channel MOSFET including a body diode D2, and is switched at zero voltage by the switching controller 11.
The second parallel capacitor C2 is connected in parallel with the second switching element SW 2. The second parallel capacitor C2 can adjust the speed of change of the first connection point voltage Vn1 in which the first switching element SW1 and the second switching element SW2 are in an off state, a so-called dead time. Like the first parallel capacitor C1, the second parallel capacitor C2 is not limited to a single element, and may be a parasitic capacitor of the first switching element SW 1.
In this way, since the return unit 14 is configured by the second switching element SW2, the return period can be appropriately set by on/off control of the second switching element SW2, and the switching loss of the second switching element SW2 can be suppressed by zero-voltage switching. The configuration of the reflow unit 14 is not limited to this, and may be configured by, for example, one reflow diode.
The auxiliary resonant circuit 20 includes an auxiliary switching element SA, an auxiliary reactor LA, an auxiliary capacitor CA, and a voltage stabilizing diode D21 connected in series. The auxiliary resonant circuit 20 is connected to a first connection point N1 which is a connection point between the first switching element SW1 and the low-pass filter 13. Specifically, one end LA1 of the auxiliary reactor LA not connected to the auxiliary switching element SA is connected to the first connection point N1. Here, the reactance of the auxiliary reactor LA is set to be sufficiently smaller than the smoothing reactor Lo.
The auxiliary switching element SA is formed of an N-channel MOSFET in the present embodiment. The auxiliary switching element SA includes a parasitic capacitor Cp and a body diode D3. The auxiliary switching element SA includes a drain terminal D, a gate terminal G, and a source terminal S. The source terminal S is connected to the other end La2 of the auxiliary reactor, and the drain terminal D is connected to and disconnected from the source terminal S in accordance with a gate control signal GA applied to the gate terminal G. Here, the source terminal S of the auxiliary switching element SA is an example of a "first terminal", and the drain terminal D of the auxiliary switching element SA is an example of a "second terminal".
The voltage stabilizing diode D21 is connected between the second connection point N2, which is the connection point between the auxiliary reactor LA and the auxiliary switching element SA, and the ground line Lg. Specifically, the cathode of the voltage stabilizing diode D21 is connected to the second connection point N2, and the anode of the voltage stabilizing diode D21 is connected to the ground line Lg. The voltage stabilizing diode D21 stabilizes the voltage Vsa between the drain and the source of the auxiliary switching element SA when the auxiliary switching element SA is turned off.
The auxiliary capacitor CA is connected between the auxiliary switching element SA and the ground line Lg. Specifically, one end of the auxiliary capacitor CA is connected to the drain terminal D of the auxiliary switching element SA, and the other end of the auxiliary capacitor CA is connected to the ground line Lg. Here, the drain terminal D of the auxiliary switching element SA corresponds to "one end on the side not connected to the auxiliary reactor of the auxiliary switching element". The auxiliary capacitor CA supplies energy at the time of resonance of the auxiliary resonance circuit 20 by charging and discharging. The auxiliary capacitor CA is charged with energy when the current flowing through the first switching element SW1, i.e., the first current Isw1, and the current flowing through the second switching element SW2, i.e., the second current Isw2, are higher than the output current (load current) Io.
The switch controller 11 is connected to the switching elements (SW1, SW2, SA) and generates gate control signals (G1, G2, GA) for controlling the on/off switches of the switching elements. Specifically, the switch controller 11 switches the first and second switching elements (SW1 and SW2) by so-called Zero Voltage Switching (ZVS) in accordance with the gate control signals (G1 and G2). The switching control unit 11 switches the auxiliary switching element SA by so-called Zero Current Switching (ZCS) in accordance with the gate control signal GA. The switching elements (SW1, SW2, SA) are not limited to N-channel MOSFETs. For example, an IGBT or the like may be used.
2. Operation of DC voltage conversion device
Next, the operation of the dc voltage conversion device 10 will be described with reference to fig. 2 to 9.
As shown in fig. 2, when the first switching element SW1 is in the off state and the second switching element SW2 is in the on state, that is, the return state, that is, at time t0 in the synchronous rectification state, the auxiliary switching element SA starts the resonance operation by the auxiliary resonance circuit 20 when turned on in response to the gate control signal GA, that is, when switched by the Zero Current (ZCS).
Then, in a first period K1 which is a period from time t0 to time t1, a current flows as shown in fig. 3. That is, the resonance current Irs, which is a current flowing through the auxiliary switching element SA and the auxiliary reactor LA, increases, and the second current Isw2, which is a current flowing through the second switching element SW2, decreases accordingly. The speed of increase of the resonance current Irs depends on the magnitude of the reactance of the auxiliary reactor LA. In the first period K1, the output current (load current) Io, which is the current flowing through the smoothing reactor Lo, is constant. The output current Io is not limited to the first period K1, and is substantially constant.
During the first period K1, the second switching element SW2 is turned off. The second current Isw2 further decreases and becomes 0 in association with the turning-off of the second switching element SW 2. On the other hand, the resonant current Irs increases and reaches the magnitude of the output current Io at time t 1.
Fig. 4 shows the flow of current during a period from time t1 to time t2 when the first node voltage Vn1 is greater than the input voltage Vin (second period K2). In the second period K2, since the first and second switching elements (SW1, SW2) are in the off state (so-called dead time), the resonant current Irs mainly flows through the first parallel capacitor C1 and the second parallel capacitor C2 (see the currents Ic1 and Ic2 in fig. 4). At this time, the first parallel capacitor C1 is discharged, while the second parallel capacitor C2 is charged. Therefore, the first junction voltage Vn1 rises. Here, the first junction voltage Vn1 is equal to the second voltage Vsw2, which is the drain-source voltage of the second switching element SW 2. Accordingly, as shown in fig. 2, the second voltage Vsw2 rises in the second period K2.
At time t2, when the first junction voltage Vn1 (second voltage Vsw2) reaches the input voltage Vin, the body diode D1 of the first switching element SW1 is turned on, and the first voltage Vsw1, which is the drain-source voltage of the first switching element SW1, becomes 0. Fig. 5 shows a state of the on period of the body diode D1, that is, from time t2 to time t3 (third period K3). At this time, a first current Isw1 in the reverse direction flows via the body diode D1.
During the on period of the body diode D1, the first switching element SW1 is turned on (time t 3). That is, the first switching element SW1 is a Zero Voltage Switch (ZVS). At this time, a voltage in the reverse direction (input voltage Vin — auxiliary capacitor voltage (charging voltage) Vca) is applied to auxiliary reactor LA. Then, as shown in fig. 2, after time t3, the current flowing to the first switching element SW1, i.e., the first current Isw1, increases, and the resonance current Irs decreases. After time t4 when the value of the first current Isw1 reaches the output current Io, the resonant current Irs becomes 0. Fig. 6 (a) shows the flow of current immediately after the first switching element SW1 is turned on at time t3, and fig. 6 (b) shows the flow of current in a period from time t3 to time t4 (fourth period K4).
Then, at time t5 after a predetermined time from time t4 at which resonant current Irs becomes 0, first switching element SW1 is turned off. Fig. 6 (c) shows the flow of current in the period from time t4 to time t5 (fifth period K5).
When the first switching element SW1 is turned off at time t5, the first current Isw1 commutates to the first parallel capacitor C1 and the second parallel capacitor C2 (see currents Ic1 and Ic2 in fig. 7). At this time, the first junction voltage Vn1 (second voltage Vsw2) drops sharply. After time t6 when the first junction voltage Vn1 reaches 0V, the body diode D2 is turned on (see fig. 8), and the second voltage Vsw2 is maintained at 0V. At this time, the second current Isw2 flows via the body diode D2.
At a time t7 after the time t6 at which the body diode D2 is turned on, the second switching element SW2 is turned on, that is, is zero-voltage switched, and so-called synchronous rectification is started. At this time, the charging of the auxiliary capacitor CA based on the second current Isw2 is continued (refer to fig. 9). Then, the charging current (reverse resonance current) Irs decreases as the charging voltage Vca of the auxiliary capacitor CA increases.
At a time t8 when the resonant current Irs reaches 0, the auxiliary switching element SA is turned off, i.e., Zero Current Switching (ZCS) is performed. The synchronous rectification continues from time t8 to time t9 when the second switching element SW2 is turned off next. After the time t9, the operation from the time t0 is repeated. The 1 cycle from the time t0 to the time t9 is, for example, 10 μ s (microseconds).
Here, the period from time t5 to time t7 is a so-called dead time. Fig. 7 shows the flow of current in the first half of the dead time, that is, in the period from time t5 to time t6 (sixth period K6), and fig. 8 shows the flow of current in the second half of the dead time, that is, in the period from time t6 to time t7 (seventh period K7).
Note that the time t0 to the time t4 correspond to the discharge period KH of the auxiliary capacitor CA, and the time t4 to the time t8 correspond to the charge period KJ of the auxiliary capacitor CA. The voltage Vca at the time of discharging the auxiliary capacitor CA is determined according to the capacitance of the auxiliary capacitor CA.
As shown in fig. 2, the charge period KJ includes a period during which the first current Isw1 exceeds the output current Io (a period from time t4 to time t 5) and a period during which the second current Isw2 exceeds the output current Io (a period from time t6 to time t 8). The on period (from time t0 to time t8) of the gate control signal (an example of a control signal) GA of the auxiliary capacitor CA includes a charging period KJ. That is, the on period of the gate control signal (an example of the control signal) GA of the auxiliary capacitor CA includes a charging period KJ in which the auxiliary capacitor CA is charged with the first current Isw1 exceeding the output current Io and the second current Isw2 exceeding the output current Io.
It is assumed that the oscillation is caused by the current flowing through the parasitic capacitance Cp of the auxiliary switching element SA after the time t8 when the auxiliary switching element SA is turned off and the current flowing through the auxiliary switching element SA and the body diode D3 becomes 0. Specifically, in the circuit formed by the auxiliary reactor LA, the parasitic capacitance Cp, the auxiliary switching element SA, and the second switching element SW2, it is conceivable that the voltage Vsa between the drain and the source of the auxiliary switching element SA largely oscillates due to oscillation. However, the generation of such a vibration of the voltage Vsa can be suppressed by the voltage stabilizing diode D21.
Here, the on/off timing of each gate control signal (G1, G2, GA) is determined by a known method. That is, the on/off timing is determined by the switch control unit 11 based on a comparison between a detection signal from a detection circuit (not shown) that detects an electrical quantity such as the resonance current Irs and a reference value. Alternatively, it is determined in advance by calculation based on circuit constants such as the reactance value of the auxiliary reactor LA and the capacitance of the auxiliary capacitor CA. In this case, the determined timing data is stored in a memory or the like of the switch control unit 11, and the switch control unit 11 determines the on/off timing based on the stored data. Alternatively, the on/off timing is determined based on both the detection signal and the stored data.
3. Effect of embodiment 1
In embodiment 1, an auxiliary capacitor CA is connected to an auxiliary switching element SA of an auxiliary resonant circuit 20. Therefore, energy related to the auxiliary resonance can be supplied from the auxiliary capacitor CA to the auxiliary resonance circuit 20. The first switching element SW1 and the second switching element SW2 are zero-voltage-switched, and the auxiliary switching element SA is zero-current-switched. Therefore, according to the dc voltage conversion device 10 of embodiment 1, even when applied to a large capacitive load, efficient conversion efficiency can be maintained.
The on period (from time t0 to time t8 in fig. 2) of the gate control signal GA of the auxiliary switching element SA generated by the switch control unit 11 includes a charging period (from time t4 to time t5 and from time t6 to time t8 in fig. 2) in which the auxiliary capacitor CA is charged with the first current Isw1 exceeding the output current Io and the second current Isw2 exceeding the output current Io. Therefore, the auxiliary capacitor CA can be charged with the surplus energy generated at the time of voltage conversion, and the auxiliary resonant circuit 20 can be operated by the energy stored in the auxiliary capacitor CA. This makes it possible to effectively utilize surplus energy associated with voltage conversion, and to improve conversion efficiency.
Note that, the on period of the gate control signal GA may not include a charging period (from time t6 to time t8 in fig. 2) in which the auxiliary capacitor CA is charged with the second current Isw2 that exceeds the output current Io. In short, the on period of the gate control signal GA may include a charging period in which the auxiliary capacitor CA is charged with the first current Isw1 that exceeds at least the output current Io.
In embodiment 1, a voltage stabilizing diode D21 connected between the second connection point N2 and the ground line Lg is provided. The potential of the second connection point N2 can be stabilized while the auxiliary switching element SA is off by the voltage stabilizing diode D21. That is, in the period (eighth period K8) in which the auxiliary switching element SA21 is off and the second switching element SW2 is on, when an N-channel MOSFET is used as the auxiliary switching element SA as in embodiment 1, it is conceivable that the potential of the second connection point N2 fluctuates due to oscillation of the parasitic capacitance Cp via the auxiliary switching element SA, and the potential of the second connection point N2 rises more than the input voltage Vin. However, in this case, the voltage stabilizing diode D21 can suppress the variation in the potential of the second connection point N2. The voltage stabilizing diode D21 may be omitted.
The voltage applied to each of the switching elements (SW1, SW2, SA) may be substantially the same level as the input voltage Vin. Therefore, each switching element can be made to be a small rated component, and thus conduction loss can be reduced.
< embodiment 2>
Embodiment 2 is explained with reference to fig. 10. The on/off timings of the gate control signals (G1, G2, GA) of the respective switching elements (SW1, SW2, SA) generated by the switch controller 11 are different from those of embodiment 1. Therefore, only the differences between the on and off timings of the gate control signals (G1, G2, GA) will be described below.
In embodiment 2, the switching control unit is particularly configured to generate an on period of the gate control signal GA of the auxiliary switching element SA for a discharge period KH and a charge period KJ during which the auxiliary capacitor CA is discharged.
That is, in embodiment 1, as shown in fig. 2, the on period of the gate control signal GA is a period from time t0 to a period immediately after time t8 in fig. 2 so that the discharge period KH and the charge period KJ of the auxiliary capacitor CA are continuous. However, in embodiment 2, as shown in fig. 10, the on period of the gate control signal GA is divided into two periods, i.e., a period from time t0 to time t2 in fig. 10 corresponding to the discharge period KH of the auxiliary capacitor CA, and a period from time t3 to time t5 in fig. 10 corresponding to the charge period KJ.
Then, the first switching element SW1 is turned on at time t1 in the discharge period KH in fig. 10, and the first switching element SW1 is turned off at time t4 in the charge period KJ in fig. 10.
Further, the second switching element SW2 is turned off at a time point when the auxiliary switching element SA is turned on in the discharge period KH, i.e., at an approximate time point t0 in fig. 10, and the second switching element SW2 is turned on at a time point when the auxiliary switching element SA is turned off in the charge period KJ, i.e., at an approximate time point t5 in fig. 10.
In this way, by dividing the discharge period KH and the charge period KJ during which the auxiliary capacitor CA is discharged into the on period during which the gate control signal GA for the auxiliary switching element SA is generated, the period during which the current flows into the auxiliary switching element SA can be shortened as compared with the case of embodiment 1. This can reduce the conduction loss of the auxiliary switching element SA.
< other embodiment >
The present invention is not limited to the embodiments described above with reference to the drawings, and for example, the following embodiments are also included in the technical scope of the present invention.
(1) In the above-described embodiment, the example in which the auxiliary capacitor is configured by the auxiliary capacitor CA connected between the auxiliary switching element and the ground line has been described, but the present invention is not limited to this. For example, as shown in fig. 11, the auxiliary capacitor may be configured by a first auxiliary capacitor CA1 connected between the auxiliary switching element SA and the ground line Lg, and a second auxiliary capacitor CA2 connected between a connection point N3 between the first auxiliary capacitor CA1 and the auxiliary switching element SA and the internal power supply line Ls. In this case, the capacitance of each capacitor is preferably approximately CA2 × CA1 — 2 × CA 2.
Alternatively, as shown in fig. 12, the auxiliary capacitor CA may be configured by an auxiliary capacitor connected between the auxiliary switching element and the internal power supply line. Even in these cases, an auxiliary resonant circuit using the energy stored in the auxiliary capacitor CA can be configured by the same control as in the above-described embodiment.
(2) In the above embodiment, the example in which the auxiliary switching element SA is configured by one auxiliary switching element SA is shown, but the present invention is not limited to this. For example, the auxiliary switching element SA may be configured by two auxiliary switching elements connected in series and controlled simultaneously by the switching control unit.
It may also be formed by an auxiliary switching element SA 21.
(3) In the above embodiment, the example in which the return unit 14 is configured by the second switching element SW2 and the second parallel capacitor C2 is shown, but the present invention is not limited to this. For example, the reflow unit 14 may be formed of one reflow diode.
(4) In the above-described embodiment, the example in which the DC voltage conversion device 10 is applied to a chopper-type step-down DC-DC converter has been described, but the present invention is not limited to this, and for example, the DC voltage conversion device 10 may be applied to a chopper-type step-up DC-DC converter. Fig. 11 shows an example of control and operation when the DC voltage conversion device 10 is applied to a chopper-type step-up DC-DC converter.
Description of the reference symbols
10 … DC voltage conversion device
11 … switch control part
13 … low-pass filter
14 … reflux unit (second switch element)
20 … auxiliary resonant circuit
40 … Main Power supply
C1 … first parallel capacitor
C2 … second shunt capacitor
CA … auxiliary capacitor
D … drain terminal (second terminal) of second switching element
D21 … voltage stabilizing diode
LA … auxiliary reactor
Lg … grounding wire
Ls … internal power cord
N1 … first connection point
N2 … second connection Point
S … Source terminal (first terminal) of second switching element
SA … auxiliary switch element
SW1 … first switch element (N-channel MOSFET) SW2 … second switch element (N-channel MOSFET).
Claims (8)
1. A DC voltage conversion device for converting a DC input voltage applied from a main power supply into an output voltage having a predetermined voltage value, the DC voltage conversion device comprising:
an internal power cord connected to the main power supply;
a first switching element connected to the internal power line;
a low-pass filter having one end connected to one end of the first switching element;
an auxiliary resonant circuit including an auxiliary switching element and an auxiliary reactor connected in series, one end of the auxiliary reactor being connected to a first connection point which is a connection point of the first switching element and the low-pass filter;
a reflow part connected between the first connection point and a ground line; and
a switch control unit for performing zero-voltage switching on the first switching element and performing zero-current switching on the auxiliary switching element,
the auxiliary switching element of the auxiliary resonant circuit comprises: a first terminal connected to the other end of the auxiliary reactor; and a second terminal for turning on/off the connection with respect to the first terminal,
the auxiliary resonant circuit includes an auxiliary capacitor connected between the second terminal of the auxiliary switching element and a ground line or the internal power supply line.
2. The direct current voltage conversion apparatus according to claim 1,
the switch control unit generates a control signal for turning on and off the auxiliary switching element,
the on period of the control signal includes a charging period in which the auxiliary capacitor is charged with a first current flowing to the first switching element that exceeds at least an output current.
3. The direct current voltage conversion apparatus according to claim 2,
the switch control section generates an on period of the control signal for a discharge period during which the auxiliary capacitor is discharged and for the charge period.
4. The direct current voltage conversion device according to any one of claims 1 to 3,
the auxiliary capacitor is connected between the auxiliary switching element and the ground line.
5. The direct current voltage conversion device according to any one of claims 1 to 4,
the dc voltage conversion device further includes a voltage stabilizing diode connected between a second connection point, which is a connection point between the auxiliary reactor and the auxiliary switching element, and the ground line.
6. The direct current voltage conversion device according to any one of claims 1 to 5,
the auxiliary switching element is composed of two auxiliary switching elements connected in series and controlled by the switching control unit with the same control signal.
7. The direct current voltage conversion device according to any one of claims 1 to 6,
the return unit is configured by a second switching element that is zero-voltage switched by the switching control unit.
8. The direct current voltage conversion device according to any one of claims 1 to 7,
the dc voltage conversion device further includes:
a first parallel capacitor connected in parallel to the first switching element; and
and a second parallel capacitor connected in parallel with the return portion.
Applications Claiming Priority (3)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| JP2017042872A JP2018148725A (en) | 2017-03-07 | 2017-03-07 | DC voltage converter |
| JP2017-042872 | 2017-03-07 | ||
| PCT/JP2018/005840 WO2018163794A1 (en) | 2017-03-07 | 2018-02-20 | Direct-current voltage conversion device |
Publications (1)
| Publication Number | Publication Date |
|---|---|
| CN110692185A true CN110692185A (en) | 2020-01-14 |
Family
ID=63448896
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CN201880012565.8A Pending CN110692185A (en) | 2017-03-07 | 2018-02-20 | DC voltage conversion device |
Country Status (4)
| Country | Link |
|---|---|
| JP (1) | JP2018148725A (en) |
| CN (1) | CN110692185A (en) |
| DE (1) | DE112018001219T5 (en) |
| WO (1) | WO2018163794A1 (en) |
Cited By (1)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN112054689A (en) * | 2020-08-30 | 2020-12-08 | 哈尔滨工程大学 | Isolated three-level direct current converter |
Families Citing this family (1)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN116134717A (en) * | 2020-09-01 | 2023-05-16 | 华为数字能源技术有限公司 | DC/DC converter, voltage gain switching method and system |
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| JP2002252965A (en) * | 2001-02-22 | 2002-09-06 | Denso Corp | Power conversion apparatus using auxiliary resonance commutation circuit |
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| GB9206020D0 (en) * | 1992-03-19 | 1992-04-29 | Astec Int Ltd | Transition resonant convertor |
| JPH06269165A (en) * | 1993-03-11 | 1994-09-22 | Sanken Electric Co Ltd | Pwm dc-dc converter |
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| JPH11178319A (en) * | 1997-12-12 | 1999-07-02 | Toyo Electric Mfg Co Ltd | Gate signal control of power conversion device with auxiliary resonance communication circuit |
| JP4304862B2 (en) * | 2000-12-28 | 2009-07-29 | 富士電機システムズ株式会社 | Power converter |
| JP4017490B2 (en) * | 2002-10-02 | 2007-12-05 | 株式会社デンソー | DC / DC converter |
| JP4193606B2 (en) * | 2003-06-26 | 2008-12-10 | 株式会社デンソー | DC / DC converter |
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2017
- 2017-03-07 JP JP2017042872A patent/JP2018148725A/en active Pending
-
2018
- 2018-02-20 CN CN201880012565.8A patent/CN110692185A/en active Pending
- 2018-02-20 DE DE112018001219.0T patent/DE112018001219T5/en not_active Withdrawn
- 2018-02-20 WO PCT/JP2018/005840 patent/WO2018163794A1/en active Application Filing
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| JP2002171759A (en) * | 2000-05-10 | 2002-06-14 | Hitachi Medical Corp | Dc-dc converter and x-ray high-voltage generator using the same |
| JP2002252965A (en) * | 2001-02-22 | 2002-09-06 | Denso Corp | Power conversion apparatus using auxiliary resonance commutation circuit |
| CN101047335A (en) * | 2006-03-31 | 2007-10-03 | 雅达电子国际有限公司 | Zero-voltage-switching DC-DC converters with synchronous rectifiers |
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| CN112054689A (en) * | 2020-08-30 | 2020-12-08 | 哈尔滨工程大学 | Isolated three-level direct current converter |
Also Published As
| Publication number | Publication date |
|---|---|
| DE112018001219T5 (en) | 2019-11-21 |
| JP2018148725A (en) | 2018-09-20 |
| WO2018163794A1 (en) | 2018-09-13 |
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