CN110082602B - Full-impedance measuring circuit and measuring device - Google Patents
Full-impedance measuring circuit and measuring device Download PDFInfo
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Abstract
The invention discloses a full impedance measuring circuit and a measuring device, wherein the measuring circuit comprises: a first transimpedance amplifier, a second transimpedance amplifier, a first differential amplifier, a second differential amplifier, a reference resistor, a sensor, and four programmable multiplexers; the current injection end of the sensor is used for inputting a current signal, and the voltage output end of the sensor is used for detecting a response voltage signal; the positive input terminal of the first transimpedance amplifier and the positive input terminal of the second transimpedance amplifier are both grounded; four programmable multiplexers can switch the circuit between the two forms. According to the invention, through an alternative measurement mode, the waveforms of the measured sensor response voltage and the reference resistance response voltage both contain noise information of two channels, so that the noise contained in the waveforms of the sensor response voltage and the reference resistance response voltage is ensured to be consistent, then the noise is synchronously reduced through a three-parameter sine wave fitting algorithm of each channel, and accurate amplitude and phase estimation values of each channel are obtained, so that the measurement precision of the circuit can be further improved.
Description
Technical Field
The invention belongs to the technical field of impedance measurement, and particularly relates to a full-impedance measurement circuit and a measurement device.
Background
Conductivity (Conductivity), Temperature (Temperature), and Depth (Depth) sensors (CTD or thermohaline Depth sensors for short) are the most basic and important sensors for monitoring the environment of a body of water. It not only provides the conductivity, temperature and pressure parameters directly, but also can be used for calculating salinity parameter and depth parameter. These parameters are essential for the development of various water area studies. Taking the marine research as an example, the parameters can be used for monitoring the flowing, circulating and climate change processes of seawater, can also provide background physical parameters for the research of biogeochemistry and marine ecosystems, and have great significance in the aspects of researching global climate problems, monitoring marine ecological environment and the like. Meanwhile, the temperature and salinity parameters also provide necessary background compensation parameters for other various ocean sensors.
Commonly used sensors for warm salt depth include: electrode type conductivity sensor, PRT (platinum resistance thermometer) bridge type temperature sensor, piezoresistor bridge type pressure sensor, their measurement to CTD parameter can be converted into the measurement to corresponding impedance finally. Therefore, the design method of the impedance measurement circuit determines the quality of the final CTD parameter measurement result.
At present, the impedance measurement circuit which is common at home and abroad comprises the following three types: the first is a frequency-based impedance measurement circuit, the second is a square wave excitation-based impedance measurement circuit, and the third is a sine wave excitation-based impedance measurement circuit. Wherein the first and second impedance measurement circuits do not support full impedance measurement and only the third impedance measurement circuit supports full impedance measurement. The impedance measuring circuit based on sine wave excitation adopts a sine wave signal as an excitation source, so that the response voltage of the sensor is also in a sine wave form, and the response voltage in the sine wave form is transmitted to a digital circuit part through an ADC (analog-to-digital converter). And then the digital circuit operates a corresponding sine wave fitting algorithm (when the frequency of the excitation signal is accurately known, a three-parameter fitting algorithm is operated without iterative operation, and when the frequency of the excitation signal is not accurately known, a four-parameter fitting algorithm is operated without iterative operation), so that the amplitude and the phase of the impedance can be calculated. Therefore, the impedance measurement circuit can support full impedance measurement, and higher measurement accuracy can be obtained through a sine wave fitting algorithm. However, the impedance measuring circuit has the following problems: 1) whether real-time measurements can be supported depends on whether the excitation signal frequency is precisely known; 2) the impedance measurement accuracy is affected by the drift of the performance parameters of the components in the circuit with temperature or time, and therefore drift compensation is required.
At present, common drift compensation methods at home and abroad are divided into the following methods.
The first solution is to build a circuit using components with very low temperature coefficients. On one hand, however, components with extremely low temperature coefficients are expensive, which greatly increases the cost of circuit design; on the other hand, some components (such as differential amplifiers) cannot achieve extremely low temperature coefficients under the current production process. Therefore, this method cannot solve the component drift problem well.
The second solution is to add an extra reference resistor with extremely low temperature coefficient to provide a reference response voltage, switch the measurement object to be a sensor or the reference resistor through a multiplexer, and finally obtain an accurate measurement resistance value according to the relationship between the sensor response voltage and the reference resistor response voltage, thereby reducing the drift influence. Compared with the first method, the method not only effectively solves the drift problem, but also reduces the cost, but also doubles the overall measurement time by alternately measuring the response voltages of the sensor and the reference resistor through a single channel.
The third solution is to use a two-channel parallel design, where two channels simultaneously measure the response voltages from the sensor and the reference resistor, and then use a seven-parameter sine wave fitting algorithm to obtain accurate response voltage values of the sensor and the reference resistor. Although the analog signal acquisition of the method adopts a double-channel parallel design, the seven-parameter sine wave fitting algorithm is essentially based on the improvement of four parameters on a double-channel measurement scene (only the noise of two channels is comprehensively considered for carrying out overall fitting calculation), iterative calculation is still needed, real-time signal processing cannot be realized, and the overall measurement time is not shortened.
The three drift compensation methods cannot ensure the real-time performance of the impedance measurement circuit and cannot process signals in real time.
Disclosure of Invention
In order to solve the technical problems in the prior art, the invention provides a full-impedance measuring circuit and a measuring device.
In order to achieve the purpose, the invention provides the following technical scheme:
a full impedance measurement circuit comprising:
a first transimpedance amplifier, a second transimpedance amplifier, a first differential amplifier, a second differential amplifier, a reference resistor, a sensor, and a plurality of programmable multiplexers;
the reference resistor is used for providing a reference voltage value; the current injection end of the sensor is used for inputting an excitation current signal, and the voltage output end of the sensor is used for providing a response voltage signal; a positive input terminal of the first transimpedance amplifier and a positive input terminal of the second transimpedance amplifier are both grounded;
a plurality of said programmable multiplexers couple the first transimpedance amplifier, the second transimpedance amplifier, the first differential amplifier, the second differential amplifier, and the reference resistor to switch the circuit between:
in a first form:
a current injection end of the sensor is coupled between a negative input terminal of the first transimpedance amplifier and an output terminal of the first transimpedance amplifier;
a voltage output of the sensor is coupled between a positive input terminal of the first differential amplifier and a negative input terminal of the first differential amplifier;
the reference resistance is coupled between the negative input terminal of the second transimpedance amplifier and the output terminal of the second transimpedance amplifier;
the reference resistance is further coupled between a positive input terminal of the second differential amplifier and a negative input terminal of the second differential amplifier;
in a second form:
a current injection end of the sensor is coupled between a negative input terminal of the second transimpedance amplifier and an output terminal of the second transimpedance amplifier;
a voltage output of the sensor is coupled between a positive input terminal of the second differential amplifier and a negative input terminal of the second differential amplifier;
the reference resistance is coupled between a negative input terminal of the first transimpedance amplifier and an output terminal of the first transimpedance amplifier;
the reference resistor is also coupled between a positive input terminal of the first differential amplifier and a negative input terminal of the first differential amplifier.
In the above technical solution, the full impedance measurement circuit further includes:
a first input resistance coupled to a negative input terminal of the first transimpedance amplifier;
a second input resistance coupled to a negative input terminal of the second transimpedance amplifier.
A full impedance measurement device comprising: two measurement channels, wherein,
the first measurement channel sequentially comprises: the first I/V conversion unit, the switching unit and the first differential amplification unit;
the second measurement channel sequentially comprises: the second I/V conversion unit, the switching unit and the second differential amplification unit;
the switching unit is shared by two measuring channels and is internally provided with a programmable multiplexer; the programmable multiplexer can switch the measuring objects of the first I/V conversion unit and the second I/V conversion unit between the sensor and the reference resistor;
the first differential amplifying unit and the second differential amplifying unit may amplify analog response voltage signals of the first measurement channel and the second measurement channel, respectively.
In the above technical solution, before two measuring channels, there are also provided: the device comprises a waveform generating unit, a digital-to-analog conversion unit and a filtering buffer unit; the output end of the filtering buffer unit is respectively connected with the input ends of the two measurement channels;
the waveform generating unit can generate digital sine wave signals with controllable amplitude, frequency and phase;
the digital-to-analog conversion unit can convert the digital sine wave signal into an analog sine wave signal;
the filtering buffer unit may perform low-pass filtering on the analog sine wave signal.
In the above technical solution, the first measurement channel is further connected to a second differential amplifier unit, and the second differential amplifier unit is connected to a third differential amplifier unit, and the third differential amplifier unit is connected to a fourth differential amplifier unit: the first low-pass filtering unit and the first analog-to-digital conversion unit;
on the second measurement channel, the second differential amplification unit is connected with: the second low-pass filtering unit and the second analog-to-digital conversion unit;
the first low-pass filtering unit and the second low-pass filtering unit can respectively perform low-pass filtering on the analog response voltage signals amplified by the first measuring channel and the second measuring channel;
the first analog-to-digital conversion unit and the second analog-to-digital conversion unit can respectively convert the analog response voltage signals filtered by the first measurement channel and the second measurement channel into corresponding digital response voltage signals.
In the above technical solution, the full impedance measuring apparatus is further provided with a digital signal processing unit;
the input end of the digital signal processing unit is respectively connected with the output ends of the first analog-to-digital conversion unit and the second analog-to-digital conversion unit;
the digital signal processing unit can integrate the digital response voltage signals of the first measuring channel and the second measuring channel, and run a three-parameter sine wave fitting algorithm to obtain the response voltage values of the sensor and the reference resistor.
In the above technical solution, the digital-to-analog conversion unit, the first analog-to-digital conversion unit, and the second analog-to-digital conversion unit have the same clock source control and operate at the same sampling frequency.
In the above technical solution, the digital signal processing unit is provided with an FPGA and an MCU.
In the above technical solution, the model of the FPGA is MAX10, and the model of the MCU is STM32F 1.
In the technical scheme, the reference resistance is a metal foil resistance with a temperature coefficient within 20 PPM/DEG C.
In another aspect, the invention provides a method for performing CTD measurement by using the measuring circuit or the measuring device.
Preferably, the method comprises:
(1) generating an excitation signal in the form of a digital sine wave;
(2) converting the digital sine wave signal into an analog sine wave signal;
(3) low-pass filtering the analog sine wave signal;
(4) the measuring objects (sensors or reference resistors with extremely low temperature coefficients) of the two I/V conversion units are switched by switching the programmable multiplexer, so that the aim of alternately measuring analog response voltage signals from the sensors or the reference resistors in a dual-channel parallel mode is fulfilled;
(5) amplifying the analog response voltage signals of the two channels respectively;
(6) respectively carrying out low-pass filtering on the analog response voltage signals amplified by the two channels;
(7) respectively converting the analog response voltage signals filtered by the two channels into corresponding digital response voltage signals;
(8) and the digital response voltage signals of the two channels are reintegrated in a software programming mode, a real-time three-parameter sine wave fitting algorithm is operated to obtain the respective response voltage values of the sensor and the reference resistor, and the accurate full impedance value corresponding to the parameter measured by the sensor is further obtained according to the proportional relation between the two.
The integration referred to herein means the addition of the response voltage signals of the same device under test (reference resistance or sensor) measured at different channels during an alternating period.
The invention has the advantages that:
the invention supports full impedance measurement and ensures the real-time performance of measurement by a circuit design mode based on a three-parameter sine wave fitting algorithm.
The invention can obviously reduce or eliminate the error of the drift of the components in the full-impedance measuring circuit to the measuring result of the sensor by introducing the reference resistor, namely, the aim of temperature compensation is achieved.
According to the invention, the response voltage of the sensor and the response voltage of the reference resistor can be synchronously measured in a parallel measurement mode, and the integral measurement time is not influenced.
According to the invention, through an alternative measurement mode, the waveforms of the measured sensor response voltage and the reference resistance response voltage both contain noise information of two channels, so that the noise contained in the waveforms of the sensor response voltage and the reference resistance response voltage is ensured to be consistent, then the noise is synchronously reduced through a three-parameter sine wave fitting algorithm of each channel, and accurate amplitude and phase estimation values of each channel are obtained, so that the measurement precision of the circuit can be further improved.
Drawings
Fig. 1 is a schematic structural diagram of an embodiment of a full impedance measurement apparatus according to the present invention.
Fig. 2 is a circuit diagram of a conventional single-channel full-impedance measurement circuit.
Fig. 3 is a circuit diagram of a full impedance measurement circuit of the present invention.
Fig. 4 is a circuit diagram of a first form of the full impedance measurement circuit shown in fig. 3.
Fig. 5 is a circuit diagram of a second form of the full impedance measurement circuit shown in fig. 3.
Fig. 6 is a circuit diagram of another embodiment of the full impedance measurement apparatus of the present invention.
Detailed Description
The present invention will be described in further detail with reference to examples and drawings, but the present invention is not limited thereto.
The full-impedance measuring device provided by the invention can be used for measuring the response voltage from the sensor and the reference resistor alternately through the parallel double channels and fitting the three-parameter sine waves corresponding to the respective channels, and finally, under the condition that the overall measuring time of the system is not increased, the error caused by the performance parameter drift of components in the circuit to the full-impedance measurement can be remarkably reduced or even eliminated, namely, the drift compensation purpose is achieved.
One embodiment of the full impedance measuring apparatus of the present invention is shown in fig. 1, and includes:
a waveform generation unit: generating an excitation signal in the form of a digital sine wave;
a digital-to-analog conversion unit: converting the digital sine wave signal into an analog sine wave signal;
a filtering buffer unit: the analog sine wave signal is subjected to low-pass filtering, and a buffering effect is achieved, so that the load capacity of the circuit is improved;
an impedance measurement unit: the device can be subdivided into two I/V conversion units (I/V conversion unit 1 and I/V conversion unit 2) and a switching unit, and has the specific function that the measuring objects (sensors or reference resistors with extremely low temperature coefficients) of the two I/V conversion units are switched through a programmable multiplexer in the switching unit so as to achieve the purpose of alternately measuring analog response voltage signals from the sensors or the reference resistors in a dual-channel parallel mode;
low-pass filtering unit 1 and low-pass filtering unit 2: respectively carrying out low-pass filtering on the analog response voltage signals amplified by the two channels;
analog-to-digital conversion unit 1 and analog-to-digital conversion unit 2: respectively converting the analog response voltage signals filtered by the two channels into corresponding digital response voltage signals;
a digital signal processing unit: and the digital response voltage signals of the two channels are reintegrated in a software programming mode, a real-time three-parameter sine wave fitting algorithm is operated to obtain the respective response voltage values of the sensor and the reference resistor, and the accurate full impedance value corresponding to the parameter measured by the sensor is further obtained according to the proportional relation between the two.
The digital-to-analog conversion unit, the analog-to-digital conversion unit 1 and the analog-to-digital conversion unit 2 are controlled by the same clock source and work at the same sampling frequency, so that the frequencies of the excitation signal and the response signal are accurately known, and a three-parameter sine wave fitting algorithm can be perfectly supported.
As shown in fig. 3, a full impedance measuring circuit 1 of the present invention includes: input resistance Rin1And an input resistance Rin2A transimpedance amplifier I/V1 and I/V2, a differential amplifier IA1 and IA2, a reference resistor RcalSensor, and four programmable multiplexers: programmable multiplexer MUX1, programmable multiplexer MUX2, programmable multiplexer MUX3, programmable multiplexer MUX 4; wherein,
reference resistance RcalProviding a reference voltage value; the current injection ends (I + and I-) of the Sensor are used for inputting current signals, and the voltage output ends (V + and V-) are used for detecting response voltage signals;
input deviceResistance Rin1A negative input terminal coupled to transimpedance amplifier I/V1; input resistance Rin2Coupled to the negative input terminal of the transimpedance amplifier I/V2, the positive input terminal of the transimpedance amplifier I/V1 and the positive input terminal of the transimpedance amplifier I/V2 are both grounded.
The circuit switches between two forms, as shown in fig. 4, the first form being:
the current injection end of the Sensor is coupled between the negative input terminal of the transimpedance amplifier I/V1 and the output terminal of the transimpedance amplifier I/V1;
the voltage output terminal of the Sensor is coupled between the positive input terminal of the differential amplifier IA1 and the negative input terminal of the differential amplifier IA 1;
reference resistance RcalCoupled between the negative input terminal of the transimpedance amplifier I/V2 and the output terminal of the transimpedance amplifier I/V2;
reference resistance RcalAlso coupled between the positive input terminal of differential amplifier IA2 and the negative input terminal of differential amplifier IA 2;
as shown in fig. 5, the second form is:
the current injection end of the Sensor is coupled between the negative input terminal of the transimpedance amplifier I/V2 and the output terminal of the transimpedance amplifier I/V2;
the voltage output terminal of the Sensor is coupled between the positive input terminal of the differential amplifier IA2 and the negative input terminal of the differential amplifier IA 2;
reference resistance RcalCoupled between the negative input terminal of the transimpedance amplifier I/V1 and the output terminal of the transimpedance amplifier I/V1;
reference resistance RcalAnd is also coupled between the positive input terminal of differential amplifier IA1 and the negative input terminal of differential amplifier IA 1.
The basic principle of the invention is explained in detail below:
fig. 2 shows a circuit diagram of a conventional single-channel full-impedance measuring circuit, which includes an impedance measuring unit and a differential amplifying unit. Wherein I/V is a trans-impedance amplifier, and C/T/D Sensor is a four-terminal Sensor (which may be conductivity)One of the sensors temperature, depth, etc.), IA is a differential amplifier. Suppose VinIs a sine wave excitation signal R after digital-to-analog conversion and filtering bufferinginIs the input resistance of the transimpedance amplifier I/V, the input current I can be calculated as:
assuming that the differential amplification factor of the differential amplifier is G, and the amplitude of the sensor response voltage obtained after differential amplification and digital signal processing is U, the impedance value | Z corresponding to the parameter measured by the sensor is finally obtainedSThe size of |:
can be obtained by the arrangement of the formula [1] and the formula [2 ]:
from formula [3]Analyzable, impedance value | ZSI size and differential amplification factor G, trans-impedance amplifier input resistance RinIn this regard, the accuracy of the impedance measurement is affected by the drift in time or temperature of the differential amplifier, the transimpedance amplifier, and the input resistor.
The drift compensation method of the invention is thus introduced. Fig. 3 to 5 are circuit diagrams of the full impedance measurement of the present invention, which include circuit diagrams corresponding to the impedance measurement unit and the differential amplification unit.
As shown in FIG. 3, the dual channels are designed to be completely symmetrical, i.e., the input resistor Rin1And Rin2The transimpedance amplifiers I/V1 and I/V2 and the differential amplifiers IA1 and IA2 comprise a subsequent low-pass filter unit 1 and a subsequent low-pass filter unit 2, and an analog-to-digital conversion unit 1 and an analog-to-digital conversion unit 2 which are all formed by components with the same parameters. The part within the dashed box of fig. 3 corresponds to the switching unit shown in fig. 1, which consists of the following components:
(1) sensor of four-terminal type: the current injection ends (I + and I-) of the sensor are used for inputting current signals, and the voltage output ends (V + and V-) are used for detecting response voltage signals;
(2) reference resistor R with extremely low temperature coefficient and ultra-high precisioncal: providing a reference voltage value;
(3) four programmable multiplexers: the device is divided into two groups of MUX1+ MUX3 and MUX2+ MUX4 and is used for switching measurement objects. Wherein, MUX1 and MUX2 are used to select the negative feedback end loads (sensor or reference resistor) of transimpedance amplifiers I/V1 and I/V2, and MUX3 and MUX4 are used to select the input signals of the differential input ends of differential amplifiers IA1 and IA 2.
Based on the circuit diagrams of fig. 2 to 5, the following describes a specific implementation principle of the "two-channel parallel alternate measurement" of the full-impedance measurement circuit of the present invention:
first, we make the following two assumptions:
(1) the waveform stored by the waveform generating unit is a digital sine waveform with N points;
(2)Vinthe analog sine wave excitation signal is obtained by sampling digital sine wave with m Hz sampling frequency and filtering and buffering by the D/A conversion unit, and the period (i.e. measurement period) of the analog sine wave excitation signal is
The programming controls the switching unit to switch in k measurement periods, and the analog-to-digital conversion unit 1 and the analog-to-digital conversion unit 2 simultaneously convert the analog response voltage signals detected by the respective channels into digital response voltage signals at a sampling rate of m Hz (assuming that the digital response voltage signals obtained by the channels 1 and 2 are x and y, respectively). The sampling time is controlled to be 2k measurement periods, and then x and y both contain the sensor response voltage vsAnd a reference resistance response voltage vcAssuming that:
x＝x1+x2[4]
y＝y1+y2[5]
formula [4][5]In, x1、x2Respectively representing the 1 st to k th periods of the signal xWaveform information of the k +1 th to 2k th periods, y1、y2Respectively represent the waveform information of the 1 st to k th periods and the k +1 th to 2k th periods of the signal y. Assuming that during the 1 st to k th cycles, channel 1 measures the response voltage of the sensor and channel 2 measures the response voltage of the reference resistor (see fig. 4), then there are:
x1＝vs1，y1＝vc1[6]
formula [6]]In, vs1Representing the sensor response voltage, v, of cycles 1 to kc1Representing the reference resistance response voltage for periods 1 through k. Then, during the (k + 1) th to (2) th cycles, channel 1 measures the response voltage of the reference resistor, and channel 2 measures the response voltage of the sensor (as shown in fig. 5), i.e.:
x2＝vc2，y2＝vs2[7]
by substituting formula [6] [7] for formula [4] [5 ]:
x＝vs1+vc2[8]
y＝vc1+vs2[9]
from formula [8][9]It can be seen intuitively that channel 1 and channel 2 achieve an alternative measurement of the sensor response voltage vsAnd a reference resistance response voltage vcThe purpose of (1).
Inputting the two paths of digital response voltage signals x and y into a digital signal processing unit, and performing the following digital signal processing processes in the digital signal processing unit: by means of software programming, the formula [8 ]][9]The described signals x and y are re-integrated in real time (for example, the response voltage of the sensor measured by the channel 1 in the 1-k period is integrated and added with the response voltage of the sensor measured by the channel 2 in the k + 1-2 k period, and the response voltage of the reference resistor measured by the channel 2 in the 1-k period is integrated and added with the response voltage of the reference resistor measured by the channel 1 in the k + 1-2 k period), and finally, the response voltages v and v of the sensor measured in the 2k periods can be obtained respectivelysAnd a reference resistance response voltage vcDigital sine waveform of (a):
vs＝vs1+vs2＝x1+y2[10]
vc＝vc1+vc2＝y1+x2[11]
from formula [10][11]It can be seen that the sensor response voltage vsAnd a reference resistance response voltage vcBoth contain the noise of channel 1 and channel 2 (i.e., v can be considered to besAnd vcThe contained noise is consistent), and then the above v is adjustedsAnd vcRunning a real-time three-parameter sine wave fitting algorithm respectively to obtain respective response voltage amplitudes (respectively marked as U)sAnd Uc) And a phase.
As shown in fig. 3, since the dual channels adopt a completely symmetrical design, it is noted that:
Rin1＝Rin2r (i.e. I)1＝I2＝I) [12]
G1＝G2＝G [13]
And the impedance value | Z corresponding to the parameter measured by the sensorsensor| and reference resistance value RcalAccording to the formula [12][13]Can be respectively calculated as:
the formula [14] [15] can be collated:
from formula [16]It can be seen that the drift compensation design method avoids removing the reference resistor RcalTemperature drift of other components vs. impedance ZsensorInfluence of | measurement accuracy, and reference resistance RcalThe resistance components with extremely low temperature coefficient and extremely high precision can be selected, so that the influence of the drift of the components in the circuit on the measurement precision of the sensor can be basically eliminated. In addition, the method adopts a double-channel parallel alternate measurement mode, and comprehensively considers double channels from the hardware levelThe noise of the invention achieves the same fitting effect as that of a seven-parameter fitting algorithm, but the invention only needs to run a three-parameter fitting algorithm in real time in the aspect of signal processing without iterative computation.
Another embodiment of the present invention will be described below by taking the circuit configuration diagram shown in fig. 6 as an example.
The present embodiment may be considered as a sensor measurement system comprising an analog circuit for full impedance measurement and a complementary digital circuit. The functions and the selection of the analog and digital circuits will be described in detail below.
The analog circuit is mainly used for realizing the functions of specific double-channel mutual correction and full impedance measurement, and comprises the following parts:
(1) DAC: the digital-to-analog converter is a digital-to-analog conversion unit and is used for converting the digital sine wave signals from the waveform generation unit into analog sine wave signals, and a DAC module with the resolution of 12bits to 18bits and the sampling rate of more than 1MHz can be selected;
(2) BUF _ LPF: the active low-pass filter is a filtering buffer unit and is used for low-pass filtering the analog sine wave signal and buffering the analog sine wave signal to improve the load capacity of a circuit, and the active low-pass filter can be composed of a low-noise precise operational amplifier, a high-precision resistor and a high-precision capacitor;
(3) I/V1 and I/V2: the transimpedance amplifier is an I/V conversion unit and is used for converting input current signals of respective channels into voltage signals of a negative feedback end, namely response voltage signals from a sensor or a reference resistor, and can also be formed by a low-noise precise operational amplifier;
(4) MUX1 to MUX 4: the multiplexer is used for selecting the feedback end loads (sensors or reference resistors) of the trans-impedance amplifiers (I/V1 and I/V2) and the input signals of the differential amplifiers (IA1 and IA2) so as to achieve the purpose of dual-channel parallel alternate measurement, and four programmable double SPDT analog switches can be selected;
(5) C/T/D Sensor: the four-terminal sensor comprises a four-electrode conductivity sensor, a PRT bridge type temperature sensor, a piezoresistor bridge type pressure sensor and the like;
(6)Rcal: the reference resistor is used for providing reference voltage for the response voltage of the sensor, and a metal foil resistor with the precision of 0.01-0.1% and the temperature coefficient within 20 PPM/DEG C can be selected;
(7) IA1 and IA 2: the differential amplifier corresponds to the differential amplification unit 1 and the differential amplification unit 2 in the invention content, is used for differentially amplifying analog response voltage signals of respective channels, and can select a digital programmable instrument amplifier or a differential amplifier with fixed amplification factor;
(8) LPF1 and LPF 2: the passive low-pass filter is a low-pass filtering unit and is used for carrying out low-pass filtering on the analog response voltage signal subjected to differential amplification so as to eliminate high-frequency noise in the signal, and the passive low-pass filter can be formed by selecting a high-precision resistor and a high-precision capacitor;
(9) ADC1 and ADC 2: the analog-to-digital converter is an analog-to-digital conversion unit, and is configured to convert the analog response voltage signal subjected to the low-pass filtering into a digital response voltage signal for subsequent digital signal processing, and may select an ADC module with a resolution of 12bits to 18bits and a sampling rate of 1MHz or higher.
In the above, the three parts (4), (5) and (6) constitute a switching unit; (3) the two paths of components mentioned in the five parts of (4), (7), (8) and (9) need to adopt components with the same parameter so as to achieve the completely symmetrical circuit design effect.
The digital circuit in this embodiment is mainly used for the function realization of the waveform generation unit and the digital signal processing unit, and includes the following parts:
FPGA: the FPGA is mainly used for realizing high-speed signal processing, comprises the steps of generating a sine excitation waveform, operating a three-parameter sine wave fitting algorithm and realizing necessary logic control of a measuring process, and can select an FPGA chip of a mainstream manufacturer, such as an MAX10 series of Altera;
MCU: the micro control unit is used for assisting the FPGA to realize the overall logic control, communication control and power control of the sensor measuring system, and a low-power-consumption micro control chip can be selected, such as a low-power-consumption series of STM32F 1;
FLASH Memory: the flash memory is used for storing online measurement data, is convenient for a later-stage upper computer to read and use, and can select a NOR flash memory of a mainstream manufacturer;
USB Port: and the USB interface is used for communicating with an upper computer at the PC end, and the upper computer is responsible for reading, analyzing, displaying and other functions of the measured data of the sensor.
The invention supports full impedance measurement and ensures the real-time performance of measurement by a circuit design mode based on a three-parameter sine wave fitting algorithm.
The invention can obviously reduce or eliminate the error of the drift of the components in the full-impedance measuring circuit to the measuring result of the sensor by introducing the reference resistor, namely, the aim of temperature compensation is achieved.
According to the invention, the response voltage of the sensor and the response voltage of the reference resistor can be synchronously measured in a parallel measurement mode, and the integral measurement time is not influenced.
According to the invention, through an alternative measurement mode, the waveforms of the measured sensor response voltage and the reference resistance response voltage both contain noise information of two channels, so that the noise contained in the waveforms of the sensor response voltage and the reference resistance response voltage is ensured to be consistent, then the noise is synchronously reduced through a three-parameter sine wave fitting algorithm of each channel, and accurate amplitude and phase estimation values of each channel are obtained, so that the measurement precision of the circuit can be further improved.
Having described embodiments of the present invention, the foregoing description is intended to be exemplary, not exhaustive, and not limited to the embodiments disclosed. Many modifications and variations will be apparent to those of ordinary skill in the art without departing from the scope and spirit of the described embodiments. The terminology used herein is chosen in order to best explain the principles of the embodiments, the practical application, or improvements made to the technology in the marketplace, or to enable others of ordinary skill in the art to understand the embodiments disclosed herein.
Claims (8)
1. A full impedance measurement circuit, comprising:
the circuit comprises a first transimpedance amplifier, a second transimpedance amplifier, a first differential amplifier, a second differential amplifier, a reference resistor, a sensor and a plurality of programmable multiplexers, wherein the parameters of the same components are the same, and symmetrical double channels are formed;
the reference resistor is used for providing a reference voltage value; the current injection end of the sensor is used for inputting an excitation current signal, and the voltage output end of the sensor is used for providing a response voltage signal; a positive input terminal of the first transimpedance amplifier and a positive input terminal of the second transimpedance amplifier are both grounded;
a plurality of said programmable multiplexers couple the first transimpedance amplifier, the second transimpedance amplifier, the first differential amplifier, the second differential amplifier, and the reference resistor to switch the circuit between the following two modes for alternate measurement:
in a first form:
a current injection end of the sensor is coupled between a negative input terminal of the first transimpedance amplifier and an output terminal of the first transimpedance amplifier;
a voltage output of the sensor is coupled between a positive input terminal of the first differential amplifier and a negative input terminal of the first differential amplifier;
the reference resistance is coupled between the negative input terminal of the second transimpedance amplifier and the output terminal of the second transimpedance amplifier;
the reference resistance is further coupled between a positive input terminal of the second differential amplifier and a negative input terminal of the second differential amplifier;
in a second form:
a current injection end of the sensor is coupled between a negative input terminal of the second transimpedance amplifier and an output terminal of the second transimpedance amplifier;
a voltage output of the sensor is coupled between a positive input terminal of the second differential amplifier and a negative input terminal of the second differential amplifier;
the reference resistance is coupled between a negative input terminal of the first transimpedance amplifier and an output terminal of the first transimpedance amplifier;
the reference resistor is further coupled between a positive input terminal of the first differential amplifier and a negative input terminal of the first differential amplifier,
wherein, the sensor response voltage signal and the reference resistance response voltage measured by the first channel are integrated and added with the sensor response voltage and the reference resistance response voltage measured by the second channel to respectively obtain the sensor response voltage signal v measured in each periodsAnd a reference resistance response voltage signal vcOf the sensor response voltage signal vsAnd a reference resistance response voltage vcBoth containing noise of both channels, based on the sensor response voltage signal vsAnd a reference resistance response voltage signal vcAnd calculating the impedance to be measured by referring to the resistance value.
2. The full impedance measurement circuit of claim 1, further comprising:
a first input resistance coupled to a negative input terminal of the first transimpedance amplifier;
a second input resistance coupled to a negative input terminal of the second transimpedance amplifier.
3. A full impedance measurement device, comprising: two measuring channels, each having the same component parameters, form a symmetrical double channel, wherein,
the first measurement channel sequentially comprises: the first I/V conversion unit, the switching unit and the first differential amplification unit;
the second measurement channel sequentially comprises: the second I/V conversion unit, the switching unit and the second differential amplification unit;
the switching unit is shared by two measuring channels and is internally provided with a programmable multiplexer; the programmable multiplexer can switch the measuring objects of the first I/V conversion unit and the second I/V conversion unit between the sensor and the reference resistor for alternate measurement;
the first and second differential amplifying units may amplify analog response voltage signals of the first and second measurement channels, respectively,
wherein, the sensor response voltage signal and the reference resistance response voltage measured by the first channel are integrated and added with the sensor response voltage and the reference resistance response voltage measured by the second channel to respectively obtain the sensor response voltage signal v measured in each periodsAnd a reference resistance response voltage signal vcOf the sensor response voltage signal vsAnd a reference resistance response voltage vcBoth containing noise of both channels, based on the sensor response voltage signal vsAnd a reference resistance response voltage signal vcAnd calculating the impedance to be measured by referring to the resistance value.
4. The full impedance measurement device of claim 3, wherein before the two measurement channels, there are further provided: the device comprises a waveform generating unit, a digital-to-analog conversion unit and a filtering buffer unit; the output end of the filtering buffer unit is respectively connected with the input ends of the two measurement channels;
the waveform generating unit can generate digital sine wave signals with controllable amplitude, frequency and phase;
the digital-to-analog conversion unit can convert the digital sine wave signal into an analog sine wave signal;
the filtering buffer unit may perform low-pass filtering on the analog sine wave signal.
5. The full impedance measurement device of claim 4,
on the first measurement channel, the first differential amplification unit is connected with: the first low-pass filtering unit and the first analog-to-digital conversion unit;
on the second measurement channel, the second differential amplification unit is connected with: the second low-pass filtering unit and the second analog-to-digital conversion unit;
the first low-pass filtering unit and the second low-pass filtering unit can respectively perform low-pass filtering on the analog response voltage signals amplified by the first measuring channel and the second measuring channel;
the first analog-to-digital conversion unit and the second analog-to-digital conversion unit can respectively convert the analog response voltage signals filtered by the first measurement channel and the second measurement channel into corresponding digital response voltage signals.
6. The full impedance measurement device of claim 5, further comprising a digital signal processing unit;
the input end of the digital signal processing unit is respectively connected with the output ends of the first analog-to-digital conversion unit and the second analog-to-digital conversion unit;
the digital signal processing unit can integrate the digital response voltage signals of the first measuring channel and the second measuring channel, and run a three-parameter sine wave fitting algorithm to obtain the response voltage values of the sensor and the reference resistor.
7. The full impedance measurement device of claim 5, wherein the digital-to-analog conversion unit, the first analog-to-digital conversion unit, and the second analog-to-digital conversion unit have the same clock source control and operate at the same sampling frequency.
8. The full impedance measurement device of claim 6, wherein the reference resistance is a metal foil resistance with a temperature coefficient within 20PPM/° C.
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| CN114184856A (en) * | 2021-12-06 | 2022-03-15 | 中国计量科学研究院 | Thermal noise detection device based on fully differential structure and noise thermometer |
| CN114812915B (en) * | 2022-06-24 | 2022-10-18 | 中国空气动力研究与发展中心低速空气动力研究所 | Pressure scanning valve circuit |
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