CN110082602B - A full impedance measurement circuit and measurement device - Google Patents

Abstract

The invention discloses a full impedance measurement circuit and a measurement device. The measurement circuit comprises: a first transimpedance amplifier, a second transimpedance amplifier, a first differential amplifier, a second differential amplifier, a reference resistor, a sensor, and four programming multiplexer; the current injection terminal of the sensor is used for inputting the current signal, and the voltage output terminal is used for detecting the response voltage signal; the positive input terminal of the first transimpedance amplifier and the positive input terminal of the second transimpedance amplifier are both grounded; Four programmable multiplexers allow the circuit to switch between the two forms. The present invention makes the measured waveforms of the sensor response voltage and the reference resistance response voltage contain the noise information of the two channels by means of alternate measurement, thereby ensuring that the noises contained in the two waveforms tend to be consistent, and then pass through the respective channels. The three-parameter sine wave fitting algorithm simultaneously reduces noise and obtains their respective accurate amplitude and phase estimates, which can further improve the measurement accuracy of the circuit.

Description

A full impedance measurement circuit and measurement device

technical field

The invention belongs to the technical field of impedance measurement, and in particular relates to a full impedance measurement circuit and a measurement device.

Background technique

Conductivity (Conductivity), temperature (Temperature), and depth (Depth) sensors (CTD or temperature, salinity and depth sensor for short) are the most basic and important sensors for monitoring the water environment. It not only directly provides conductivity, temperature, pressure parameters, but also can be used to calculate salinity parameters and depth parameters. These parameters are essential to carry out various water studies. Taking marine research as an example, these parameters can not only be used to monitor the flow, circulation, and climate change process of seawater, but also provide background physical parameters for the study of biogeochemistry and marine ecosystems. etc. are of great significance. At the same time, the temperature and salinity parameters also provide essential background compensation parameters for various other marine sensors.

Commonly used temperature and salt depth sensors include: electrode conductivity sensor, PRT (platinum resistance thermometer, platinum resistance thermometer) bridge temperature sensor, piezoresistor bridge pressure sensor, their measurement of CTD parameters can finally be converted into pairs Corresponding impedance measurements. Therefore, the design method of the impedance measurement circuit determines the quality of the final CTD parameter measurement.

At present, the common impedance measurement circuits at home and abroad include the following three types: the first is a frequency-based impedance measurement circuit, the second is an impedance measurement circuit based on square wave excitation, and the third is an impedance measurement circuit based on sine wave excitation. Among them, the first and second types do not support full impedance measurement, and only the third impedance measurement circuit supports full impedance measurement. The impedance measurement circuit based on sine wave excitation uses a sine wave signal as the excitation source, so the response voltage of the sensor is also in the form of a sine wave, and the response voltage in the form of a sine wave is transmitted to the digital circuit part through an ADC (analog-to-digital converter). Then the digital circuit runs the corresponding sine wave fitting algorithm (when the frequency of the excitation signal is accurately known, the three-parameter fitting algorithm is run, and no iterative operation is required; when the excitation signal is not accurately known, the four-parameter fitting algorithm is run, which requires iteration. operation), so that the magnitude and phase of the impedance can be calculated. Therefore, the impedance measurement circuit can support full impedance measurement, and a higher measurement accuracy can be obtained through the sine wave fitting algorithm. However, this impedance measurement circuit has the following problems: 1) Whether it can support real-time measurement depends on whether the frequency of the excitation signal is accurately known; 2) The impedance measurement accuracy will be affected by the drift of the performance parameters of the components in the circuit with temperature or time, so Drift compensation is required.

At present, the common drift compensation methods at home and abroad are divided into the following categories.

The first solution is to build circuits with components with very low temperature coefficients. However, on the one hand, components with extremely low temperature coefficients are expensive, which greatly increases the cost of circuit design; on the other hand, some components (such as differential amplifiers) cannot achieve extremely low temperature coefficients under the current production process. Therefore, this method cannot solve the problem of component drift very well.

The second solution is to add an extra reference resistance with a very low temperature coefficient to provide a reference response voltage, switch the measurement object to the sensor or the reference resistance through the multiplexer, and finally according to the difference between the sensor response voltage and the reference resistance response voltage. The relationship between the two obtains an accurate measured resistance value, thereby reducing the effect of drift. Compared with the first method, this method can effectively solve the drift problem and reduce the cost, but the overall measurement time is doubled by alternately measuring the response voltage of the sensor and the reference resistor through a single channel.

The third solution is to use a dual-channel parallel design. The two channels measure the response voltage from the sensor and the reference resistor at the same time, and then use a seven-parameter sine wave fitting algorithm to obtain accurate sensor and reference resistor response voltage values. Although the analog signal acquisition of this method adopts two-channel parallel design, the essence of the seven-parameter sine wave fitting algorithm is based on the improvement of four parameters in the two-channel measurement scene (only the overall fitting calculation is performed by comprehensively considering the noise of the two channels) , iterative calculation is still required, real-time signal processing cannot be achieved, and the overall measurement time is not shortened.

None of the above three drift compensation methods can guarantee the real-time performance of impedance measurement circuit measurement, and cannot achieve real-time signal processing.

SUMMARY OF THE INVENTION

In order to solve the technical problems existing in the prior art, the present invention provides a full impedance measurement circuit and a measurement device.

To achieve the above object, the present invention provides the following technical solutions:

A full impedance measurement circuit, comprising:

a first transimpedance amplifier, a second transimpedance amplifier, a first differential amplifier, a second differential amplifier, a reference resistor, a sensor, and a plurality of programmable multiplexers;

The reference resistor is used to provide a reference voltage value; the current injection terminal of the sensor is used to input an excitation current signal, and the voltage output terminal is used to provide a response voltage signal; the positive input terminal of the first transimpedance amplifier and the The positive input terminals of the second transimpedance amplifier are all grounded;

A plurality of the programmable multiplexers are connected to the first transimpedance amplifier, the second transimpedance amplifier, the first differential amplifier, the second differential amplifier, and the reference resistor, so that the circuit can be switched in the following two forms :

First form:

The current injection end of the sensor is coupled between the negative input terminal of the first transimpedance amplifier and the output terminal of the first transimpedance amplifier;

The voltage output of the sensor is coupled between the positive input terminal of the first differential amplifier and the negative input terminal of the first differential amplifier;

the reference resistor is coupled between the negative input terminal of the second transimpedance amplifier and the output terminal of the second transimpedance amplifier;

the reference resistor is also coupled between the positive input terminal of the second differential amplifier and the negative input terminal of the second differential amplifier;

The second form:

The current injection end of the sensor is coupled between the negative input terminal of the second transimpedance amplifier and the output terminal of the second transimpedance amplifier;

The voltage output of the sensor is coupled between the positive input terminal of the second differential amplifier and the negative input terminal of the second differential amplifier;

the reference resistor is coupled between the negative input terminal of the first transimpedance amplifier and the output terminal of the first transimpedance amplifier;

The reference resistor is also coupled between the positive input terminal of the first differential amplifier and the negative input terminal of the first differential amplifier.

In the above technical solution, the full impedance measurement circuit further includes:

a first input resistor coupled to the negative input terminal of the first transimpedance amplifier;

A second input resistance coupled to the negative input terminal of the second transimpedance amplifier.

A full impedance measurement device, comprising: two measurement channels, wherein,

The first measurement channel sequentially includes: a first I/V conversion unit, a switching unit, and a first differential amplifying unit;

The second measurement channel sequentially includes: a second I/V conversion unit, a switching unit, and a second differential amplifying unit;

The switching unit is shared by two measurement channels, and a programmable multiplexer is arranged in it; the programmable multiplexer can make the first I/V conversion unit and the second I/V The measurement object of the conversion unit, switching between sensor and reference resistance;

The first differential amplifying unit and the second differential amplifying unit may amplify the analog response voltage signals of the first measurement channel and the second measurement channel, respectively.

In the above technical solution, before the two measurement channels, a waveform generation unit, a digital-to-analog conversion unit, and a filter buffer unit are connected in sequence; the output ends of the filter buffer unit are respectively connected with the input terminals of the two measurement channels. end connection;

the waveform generating unit, which can generate a digital sine wave signal whose amplitude, frequency and phase are controllable;

the digital-to-analog conversion unit, which can convert the digital sine wave signal into an analog sine wave signal;

The filtering buffer unit can perform low-pass filtering on the analog sine wave signal.

In the above technical solution, the first measurement channel is further connected after the first differential amplifying unit: a first low-pass filtering unit and a first analog-to-digital conversion unit;

On the second measurement channel, after the second differential amplifying unit is further connected: a second low-pass filtering unit and a second analog-to-digital conversion unit;

The first low-pass filtering unit and the second low-pass filtering unit may perform low-pass filtering on the amplified analog response voltage signals of the first measurement channel and the second measurement channel, respectively;

The first analog-to-digital conversion unit and the second analog-to-digital conversion unit may respectively convert analog response voltage signals filtered by the first measurement channel and the second measurement channel into corresponding digital response voltage signals.

In the above technical solution, the full impedance measurement device is further provided with a digital signal processing unit;

The input ends of the digital signal processing unit are respectively connected with the output ends of the first analog-to-digital conversion unit and the second analog-to-digital conversion unit;

The digital signal processing unit can integrate the digital response voltage signals of the first measurement channel and the second measurement channel, run a three-parameter sine wave fitting algorithm, and obtain the response voltage values of the sensor and the reference resistor.

In the above technical solution, the digital-to-analog conversion unit, the first analog-to-digital conversion unit, and the second analog-to-digital conversion unit are controlled by the same clock source and work at the same sampling frequency.

In the above technical solution, an FPGA and an MCU are provided in the digital signal processing unit.

In the above technical solution, the model of the FPGA is MAX10, and the model of the MCU is STM32F1.

In the above technical solution, the reference resistance is a metal foil resistance with a temperature coefficient within 20PPM/°C.

Another aspect of the invention provides a method for CTD measurement using the above-mentioned measurement circuit or measurement device.

Preferably, the method includes:

(1) Generate an excitation signal in the form of a digital sine wave;

(2) converting the above-mentioned digital sine wave signal into an analog sine wave signal;

(3) low-pass filtering the above-mentioned analog sine wave signal;

(4) Switch the measurement object (sensor or reference resistance with extremely low temperature coefficient) of the two I/V conversion units by switching the programmable multiplexer, so as to achieve dual-channel parallel alternate measurement from the sensor or reference resistance The purpose of the analog response to the voltage signal;

(5) respectively amplifying the analog response voltage signals of the above two channels;

(6) respectively low-pass filtering the amplified analog response voltage signals of the above two channels;

(7) respectively converting the analog response voltage signals filtered by the above two channels into corresponding digital response voltage signals;

(8) Reintegrate the digital response voltage signals of the above two channels through software programming, and run a real-time three-parameter sine wave fitting algorithm to obtain the respective response voltage values of the sensor and the reference resistor. The proportional relationship between the two, and then obtain the accurate full impedance value corresponding to the parameters measured by the sensor.

Integration, as mentioned here, refers to summing the response voltage signals of the same device under test (reference resistor or sensor) measured on different channels during an alternating period.

The advantages of the present invention are:

Through the circuit design method based on the three-parameter sine wave fitting algorithm, the present invention not only supports the full impedance measurement, but also ensures the real-time performance of the measurement.

The present invention can significantly reduce or eliminate the error brought by component drift in the full impedance measurement circuit to the measurement result of the sensor by introducing a reference resistance, that is, to achieve the purpose of temperature compensation.

In the present invention, the measurement of the response voltage of the sensor and the response voltage of the reference resistance can be performed synchronously by means of parallel measurement, without affecting the overall measurement time.

The present invention makes the measured waveforms of the sensor response voltage and the reference resistance response voltage contain the noise information of the two channels by means of alternate measurement, thereby ensuring that the noises contained in the two waveforms tend to be consistent, and then pass through the respective channels. The three-parameter sine wave fitting algorithm simultaneously reduces noise and obtains their respective accurate amplitude and phase estimates, which can further improve the measurement accuracy of the circuit.

Description of drawings

FIG. 1 is a schematic structural diagram of a specific embodiment of the full impedance measurement device of the present invention.

FIG. 2 is a circuit diagram of a conventional single-channel full impedance measurement circuit.

FIG. 3 is a circuit diagram of the full impedance measurement circuit of the present invention.

FIG. 4 is a circuit diagram of a first form of the full impedance measurement circuit shown in FIG. 3 .

FIG. 5 is a circuit diagram of a second form of the full impedance measurement circuit shown in FIG. 3 .

FIG. 6 is a circuit diagram of another specific embodiment of the full impedance measurement device of the present invention.

Detailed ways

The present invention will be described in further detail below with reference to the embodiments and the accompanying drawings, but the embodiments of the present invention are not limited thereto.

The full impedance measurement device of the present invention measures the response voltage from the sensor and the reference resistor alternately through parallel dual channels, and the three-parameter sine wave fitting corresponding to the respective channels, finally without increasing the overall measurement time of the system. Significantly reduce or even eliminate the error caused by the drift of the performance parameters of the components in the circuit to the full impedance measurement, that is, to achieve the purpose of drift compensation.

A specific embodiment of the full impedance measurement device of the present invention is shown in FIG. 1 , including:

Waveform generation unit: generate excitation signal in the form of digital sine wave;

Digital-to-analog conversion unit: converts the above-mentioned digital sine wave signal into an analog sine wave signal;

Filtering buffer unit: perform low-pass filtering on the above-mentioned analog sine wave signal, and play a buffering role to improve the load capacity of the circuit;

Impedance measurement unit: can be subdivided into two I/V conversion units (I/V conversion unit 1 and I/V conversion unit 2) and switching unit, the specific function is to use the programmable multiplexer in the switching unit To switch the measurement objects (sensors or reference resistors with extremely low temperature coefficient) of the two I/V conversion units, in order to achieve the purpose of alternately measuring the analog response voltage signals from the sensors or reference resistors in parallel with two channels;

Differential amplifying unit 1 and differential amplifying unit 2: respectively amplify the analog response voltage signals of the above two channels;

Low-pass filtering unit 1 and low-pass filtering unit 2: respectively perform low-pass filtering on the analog response voltage signals amplified by the above two channels;

The analog-to-digital conversion unit 1 and the analog-to-digital conversion unit 2: respectively convert the analog response voltage signals filtered by the above two channels into corresponding digital response voltage signals;

Digital signal processing unit: Reintegrate the digital response voltage signals of the above two channels through software programming, and run a real-time three-parameter sine wave fitting algorithm to obtain the respective response voltage values of the sensor and reference resistor. The proportional relationship between them can be obtained, and then the accurate full impedance value corresponding to the parameters measured by the sensor can be obtained.

Among them, the digital-to-analog conversion unit, the analog-to-digital conversion unit 1, and the analog-to-digital conversion unit 2 are controlled by the same clock source and work at the same sampling frequency, so the frequencies of the excitation signal and the response signal are accurately known, which can perfectly support the three-parameter sine wave fitting algorithm.

The full impedance measurement circuit of the present invention, as shown in FIG. 3 , the full impedance measurement circuit 1 includes: an input resistance R _in1 and an input resistance R _in2 , a transimpedance amplifier I/V1 and a transimpedance amplifier I/V2, a differential amplifier IA1 and a differential Amplifier IA2, reference resistor R _cal , sensor Sensor, and four programmable multiplexers: programmable multiplexer MUX1, programmable multiplexer MUX2, programmable multiplexer MUX3, Programmable multiplexer MUX4; where,

The reference resistor R _cal provides the reference voltage value; the current injection terminals (I+ and I-) of the sensor Sensor are used to input the current signal, and the voltage output terminals (V+ and V-) are used to detect the response voltage signal;

The input resistor R _in1 is coupled to the negative input terminal of the transimpedance amplifier I/V1; the input resistor R _in2 is coupled to the negative input terminal of the transimpedance amplifier I/V2, the positive input terminal of the transimpedance amplifier I/V1 and the transimpedance amplifier I/ The positive input terminals of V2 are both grounded.

The circuit switches in two forms, as shown in Figure 4, the first form is:

The current injection end of the sensor Sensor is coupled between the negative input terminal of the transimpedance amplifier I/V1 and the output terminal of the transimpedance amplifier I/V1;

The voltage output end of the sensor Sensor is coupled between the positive input terminal of the differential amplifier IA1 and the negative input terminal of the differential amplifier IA1;

The reference resistor R _cal is coupled between the negative input terminal of the transimpedance amplifier I/V2 and the output terminal of the transimpedance amplifier I/V2;

The reference resistor R _cal is also coupled between the positive input terminal of the differential amplifier IA2 and the negative input terminal of the differential amplifier IA2;

As shown in Figure 5, the second form is:

The current injection end of the sensor Sensor is coupled between the negative input terminal of the transimpedance amplifier I/V2 and the output terminal of the transimpedance amplifier I/V2;

The voltage output terminal of the sensor Sensor is coupled between the positive input terminal of the differential amplifier IA2 and the negative input terminal of the differential amplifier IA2;

The reference resistance R _cal is coupled between the negative input terminal of the transimpedance amplifier I/V1 and the output terminal of the transimpedance amplifier I/V1;

The reference resistor R _cal is also coupled between the positive input terminal of the differential amplifier IA1 and the negative input terminal of the differential amplifier IA1 .

The basic principle of the present invention is described in detail below:

Figure 2 shows the circuit diagram of a traditional single-channel full impedance measurement circuit, including the impedance measurement unit and the differential amplifier unit. Among them, I/V is a transimpedance amplifier, C/T/D Sensor is a four-terminal sensor (which can be one of conductivity, temperature, depth and other sensors), and IA is a differential amplifier. Assuming that V _in is the sine wave excitation signal after digital-to-analog conversion and filtering and buffering, and R _in is the input resistance of the transimpedance amplifier I/V, the input current I can be calculated as:

Assuming that the differential amplification factor of the differential amplifier is G, and the sensor response voltage amplitude obtained after differential amplification and digital signal processing is U, the impedance value |Z _S | corresponding to the parameters measured by the final sensor is:

According to formula [1] and formula [2], we can get:

It can be analyzed from formula [3] that the magnitude of the impedance value |Z _S | is related to the differential amplification factor G and the input resistance R _in of the transimpedance amplifier, that is, the accuracy of the impedance measurement value is affected by the differential amplifier, the transimpedance amplifier and the input resistance time drift or The influence of temperature drift.

Thus, the drift compensation method of the present invention is derived. 3 to 5 are full impedance measurement circuit diagrams of the present invention, including schematic circuit diagrams corresponding to the impedance measurement unit and the differential amplifying unit.

As shown in Figure 3, the dual channel adopts a fully symmetrical design, namely input resistors R _in1 and R _in2 , transimpedance amplifiers I/V1 and I/V2, differential amplifiers IA1 and IA2, including subsequent low-pass filter units 1 and 1 The low-pass filter unit 2, the analog-to-digital conversion unit 1 and the analog-to-digital conversion unit 2 are all constituted by components with the same parameters. The part in the dashed box in Figure 3 corresponds to the switching unit shown in Figure 1, which consists of the following components:

(1) Four-terminal sensor Sensor: the current injection terminals (I+ and I-) of the sensor are used to input the current signal, and the voltage output terminals (V+ and V-) are used to detect the response voltage signal;

(2) Very low temperature coefficient, ultra-high precision reference resistance R _cal : provide reference voltage value;

(3) Four programmable multiplexers: divided into two groups of MUX1+MUX3 and MUX2+MUX4, which are used for measurement object switching. Among them, MUX1 and MUX2 are used to select the negative feedback terminal load (sensor or reference resistor) of the transimpedance amplifiers I/V1 and I/V2, and MUX3 and MUX4 are used to select the input signals of the differential input terminals of the differential amplifiers IA1 and IA2.

Based on the circuit diagrams of FIGS. 2 to 5 , the specific implementation principle of the “dual-channel parallel alternate measurement” of the full impedance measurement circuit of the present invention is described below:

First, we make the following two assumptions:

(1) The waveform stored by the waveform generation unit is a digital sinusoidal waveform of N points;

(2) V _in is the analog sine wave excitation signal that the digital-to-analog conversion unit samples the digital sine waveform at the sampling frequency of m Hz and is filtered and buffered, then the period of the analog sine wave excitation signal (that is, the measurement period) is

The programming controls the switching unit to switch in k measurement cycles, and the analog-to-digital conversion unit 1 and the analog-to-digital conversion unit 2 simultaneously convert the analog response voltage signal detected by the respective channel into a digital response voltage signal at a sampling rate of m Hz (assuming channel 1). The digital response voltage signals obtained from channel 2 and channel 2 are respectively x and y). If the sampling time is controlled to be 2k measurement cycles, then both _x and y contain the waveform information of the sensor response voltage vs and the reference resistance response voltage _vc , assuming:

x=x ₁ +x ₂ [4]

y=y ₁ +y ₂ [5]

In formula [4][5], x ₁ and x ₂ respectively represent the waveform information of the 1st to kth cycles and k+1 to 2kth cycles of the signal x, and y ₁ and y ₂ respectively represent the 1st to kth cycles of the signal y. waveform information of the k+1 to 2kth cycles. Assuming that in the first to k cycles, channel 1 measures the response voltage of the sensor, and channel 2 measures the response voltage of the reference resistance (as shown in Figure 4), there are:

x ₁ =v _s1 , y ₁ =v _c1 [6]

In formula [6], v _s1 represents the sensor response voltage in the first to k cycles, and v _c1 represents the reference resistance response voltage in the first to k cycles. Then, in the k+1 to 2kth cycle, channel 1 measures the response voltage of the reference resistor, and channel 2 measures the response voltage of the sensor (as shown in Figure 5), namely:

x ₂ =v _c2 , y ₂ =v _s2 [7]

Substitute formula [6][7] into formula [4][5] to get:

x=v _s1 +v _c2 [8]

y=v _c1 +v _s2 [9]

It can be seen intuitively from equations [8] and [9] that channel 1 and channel 2 achieve the purpose of alternately measuring the sensor response voltage _vs and the reference resistance response voltage _vc .

Input the above-mentioned two-channel digital response voltage signals x and y into the digital signal processing unit, and the following digital signal processing process is performed in the digital signal processing unit: by means of software programming, the signals described in formula [8][9] are x and y are reintegrated in real time (for example, the sensor response voltage measured on 1 to k cycles channel 1 is integrated and summed with the sensor response voltage measured on k+1 to 2k cycles channel 2, the 1 to k cycle channel 2 The measured response voltage of the reference resistance is integrated and added with the response voltage of the reference resistance measured in channel 1 of _k +1 to 2k cycles), and finally the sensor response voltage vs and the reference resistance response of 2k cycles can be obtained respectively. Digital sinusoidal waveform of voltage v _c :

v _s =v _s1 +v _s2 =x ₁ +y ₂ [10]

v _c =v _c1 +v _c2 =y ₁ +x ₂ [11]

It can be seen from formula [10][11] that both the sensor response voltage v _s and the reference resistance response voltage v _c contain the noise of channel 1 and channel 2 (that is, it can be considered that the noise contained in v _s and v _c are consistent ), and then run a real-time three-parameter sine wave fitting algorithm on the above v _s and v _c respectively to obtain the respective response voltage amplitudes (respectively denoted as U _s and U _c ) and phases.

As shown in Figure 3, since the dual channel adopts a completely symmetrical design, then write:

R _in1 =R _in2 =R (ie I ₁ =I ₂ =I) [12]

G ₁ =G ₂ =G [13]

The impedance value |Z _sensor | and the reference resistance value R _cal corresponding to the parameters measured by the sensor can be calculated as:

Arrangement [14][15] can be obtained:

It can be seen from equation [16] that this drift compensation design method avoids the influence of the temperature drift of other components except the reference resistor R _cal on the measurement accuracy of the impedance |Z _{sensor |} Resistive components with low temperature coefficient and extremely high precision, so the influence of the drift of components in the circuit on the measurement accuracy of the sensor can be basically eliminated. In addition, the method adopts the dual-channel parallel alternate measurement method, comprehensively considers the noise of the dual-channel from the hardware level, and achieves the same fitting effect as the seven-parameter fitting algorithm, but the present invention only needs to run in real time in terms of signal processing. Three-parameter fitting algorithm without iterative computation.

Another specific embodiment of the present invention is described below by taking the circuit structure diagram shown in FIG. 6 as an example.

What this embodiment describes can be considered as a sensor measurement system, which includes an analog circuit for full impedance measurement and a digital circuit matched with it. The function and selection of the analog circuit and the digital circuit are described in detail below.

The analog circuit is mainly used for the realization of the specific dual-channel mutual correction and full impedance measurement, including the following parts:

(1) DAC: digital-to-analog converter, which is a digital-to-analog conversion unit, which is used to convert the digital sine wave signal from the waveform generation unit into an analog sine wave signal. The resolution is 12bits to 18bits, and the sampling rate is DAC module above 1MHz;

(2) BUF_LPF: Active low-pass filter, which is a filter buffer unit, used to low-pass filter the above-mentioned analog sine wave signal, and as a buffer to improve the load capacity of the circuit, you can choose low-noise precision It is composed of operational amplifier and high-precision resistors and capacitors;

(3) I/V1 and I/V2: transimpedance amplifiers, which are I/V conversion units, used to convert the input current signal of the respective channel into a voltage signal at the negative feedback terminal, that is, the response from the sensor or reference resistor The voltage signal can also be composed of a low-noise precision operational amplifier;

(4) MUX1 to MUX4: Multiplexers for selecting the feedback load (sensor or reference resistor) of the transimpedance amplifier (I/V1 and I/V2) and the input signal of the differential amplifier (IA1 and IA2) , in order to achieve the purpose of dual-channel parallel alternate measurement, four programmable dual SPDT analog switches can be selected;

(5) C/T/D Sensor: Four-terminal sensor, including four-electrode conductivity sensor, PRT bridge temperature sensor, piezoresistor bridge pressure sensor, etc.;

(6) R _cal : reference resistance, used to provide a reference voltage for the response voltage of the sensor, a metal foil resistance with an accuracy of 0.01% to 0.1% and a temperature coefficient within 20PPM/°C can be selected;

(7) IA1 and IA2: differential amplifiers, corresponding to the "differential amplifying unit 1" and "differential amplifying unit 2" in the summary of the invention, used to differentially amplify the analog response voltage signals of the respective channels, digital programmable instrumentation amplifiers can be selected Or a differential amplifier with a fixed magnification;

(8) LPF1 and LPF2: passive low-pass filters, which are low-pass filtering units used to low-pass filter the differentially amplified analog response voltage signal to eliminate high-frequency noise in the signal, which can Use high-precision resistors and capacitors to form;

(9) ADC1 and ADC2: an analog-to-digital converter, which is an analog-to-digital conversion unit, used to convert the low-pass filtered analog response voltage signal into a digital response voltage signal for subsequent digital signal processing , ADC modules with a resolution of 12bits to 18bits and a sampling rate of more than 1MHz can be selected.

In the above content, (4), (5), (6) three parts constitute the switching unit; (3), (4), (7), (8), (9) mentioned in the five parts The two-way components need to use components with the same parameters to achieve a completely symmetrical circuit design effect.

The digital circuit in this embodiment is mainly used for the function realization of the waveform generation unit and the digital signal processing unit, and includes the following parts:

FPGA: Field Programmable Gate Array, mainly used for high-speed signal processing, including generating sinusoidal excitation waveforms, running three-parameter sinusoidal wave fitting algorithms, and implementing necessary logic control of the measurement process. You can choose FPGA chips from mainstream manufacturers, such as Altera’s MAX10 series;

MCU: Micro-control unit, which is used to assist FPGA to realize the overall logic control, communication control and power control of the sensor measurement system. Low-power microcontroller chips can be selected, such as the low-power series of STM32F1;

FLASH Memory: Flash memory, used to store online measurement data, which is easy to read and use by the host computer in the later stage, and NOR flash memory of mainstream manufacturers can be selected;

USB Port: The USB interface is used to communicate with the host computer on the PC side. The host computer is responsible for reading, parsing, and displaying sensor measurement data.

Through the circuit design method based on the three-parameter sine wave fitting algorithm, the present invention not only supports the full impedance measurement, but also ensures the real-time performance of the measurement.

The present invention can significantly reduce or eliminate the error caused by component drift in the full impedance measurement circuit to the measurement result of the sensor by introducing a reference resistance, that is, to achieve the purpose of temperature compensation.

In the present invention, the measurement of the response voltage of the sensor and the response voltage of the reference resistance can be performed synchronously by means of parallel measurement, without affecting the overall measurement time.

The present invention makes the measured waveforms of the sensor response voltage and the reference resistance response voltage contain the noise information of the two channels by means of alternate measurement, thereby ensuring that the noise contained in the two waveforms tends to be consistent, and then pass through the respective channels. The three-parameter sine wave fitting algorithm simultaneously reduces noise and obtains their respective accurate amplitude and phase estimates, which can further improve the measurement accuracy of the circuit.

Various embodiments of the present invention have been described above, and the foregoing descriptions are exemplary, not exhaustive, and not limiting of the disclosed embodiments. Numerous modifications and variations will be apparent to those of ordinary skill in the art without departing from the scope and spirit of the described embodiments. The terminology used herein was chosen to best explain the principles of the various embodiments, the practical application or technical improvement in the marketplace, or to enable others of ordinary skill in the art to understand the various embodiments disclosed herein.

Claims (8)

1. a full impedance measurement circuit, is characterized in that, comprises: The first transimpedance amplifier, the second transimpedance amplifier, the first differential amplifier, the second differential amplifier, the reference resistor, the sensor, and a plurality of programmable multiplexers, all of the same components have the same parameters, forming a symmetrical dual channel ; The reference resistor is used to provide a reference voltage value; the current injection terminal of the sensor is used to input an excitation current signal, and the voltage output terminal is used to provide a response voltage signal; the positive input terminal of the first transimpedance amplifier and the The positive input terminals of the second transimpedance amplifier are all grounded; A plurality of the programmable multiplexers are connected to the first transimpedance amplifier, the second transimpedance amplifier, the first differential amplifier, the second differential amplifier, and the reference resistor, so that the circuit can be switched in the following two forms , take alternate measurements: First form: The current injection end of the sensor is coupled between the negative input terminal of the first transimpedance amplifier and the output terminal of the first transimpedance amplifier; The voltage output of the sensor is coupled between the positive input terminal of the first differential amplifier and the negative input terminal of the first differential amplifier; the reference resistor is coupled between the negative input terminal of the second transimpedance amplifier and the output terminal of the second transimpedance amplifier; the reference resistor is also coupled between the positive input terminal of the second differential amplifier and the negative input terminal of the second differential amplifier; The second form: The current injection end of the sensor is coupled between the negative input terminal of the second transimpedance amplifier and the output terminal of the second transimpedance amplifier; The voltage output of the sensor is coupled between the positive input terminal of the second differential amplifier and the negative input terminal of the second differential amplifier; the reference resistor is coupled between the negative input terminal of the first transimpedance amplifier and the output terminal of the first transimpedance amplifier; The reference resistor is also coupled between the positive input terminal of the first differential amplifier and the negative input terminal of the first differential amplifier, Among them, the sensor response voltage signal and reference resistance response voltage measured by the first channel are integrated and added together with the sensor response voltage and reference resistance response voltage measured by the second channel to obtain the sensor response voltage signal measured in each cycle. The waveforms of the response voltage signal _{vc of v s and the reference resistance, the response voltage signal of the sensor v s and the response voltage v c of the reference} resistance both contain the noise of two channels, based on the response voltage signal of the sensor v _s and the response voltage of the reference resistance The signal _vc and the reference resistance value calculate the impedance to be measured. 2. full impedance measurement circuit according to claim 1, is characterized in that, also comprises: a first input resistor coupled to the negative input terminal of the first transimpedance amplifier; A second input resistance coupled to the negative input terminal of the second transimpedance amplifier. 3. A full-impedance measurement device, characterized in that it comprises: two measurement channels, each of which has the same components with the same parameters, forming a symmetrical dual channel, wherein, The first measurement channel sequentially includes: a first I/V conversion unit, a switching unit, and a first differential amplifying unit; The second measurement channel sequentially includes: a second I/V conversion unit, a switching unit, and a second differential amplifying unit; The switching unit is shared by two measurement channels, and a programmable multiplexer is arranged in it; the programmable multiplexer can make the first I/V conversion unit and the second I/V The measurement object of the conversion unit is switched between the sensor and the reference resistance, and the measurement is alternated; The first differential amplifying unit and the second differential amplifying unit can respectively amplify the analog response voltage signals of the first measurement channel and the second measurement channel, Among them, the sensor response voltage signal and reference resistance response voltage measured by the first channel are integrated and added together with the sensor response voltage and reference resistance response voltage measured by the second channel to obtain the sensor response voltage signal measured in each cycle. The waveforms of the response voltage signal _{vc of v s and the reference resistance, the response voltage signal of the sensor v s and the response voltage v c of the reference} resistance both contain the noise of two channels, based on the response voltage signal of the sensor v _s and the response voltage of the reference resistance The signal _vc and the reference resistance value calculate the impedance to be measured. 4. The full impedance measurement device according to claim 3, characterized in that, before the two measurement channels, there are also connected in sequence: a waveform generation unit, a digital-to-analog conversion unit, and a filter buffer unit; the filter buffer The output terminals of the unit are respectively connected with the input terminals of the two measurement channels; the waveform generating unit, which can generate a digital sine wave signal whose amplitude, frequency and phase are controllable; the digital-to-analog conversion unit, which can convert the digital sine wave signal into an analog sine wave signal; The filtering buffer unit can perform low-pass filtering on the analog sine wave signal. 5. The full impedance measurement device according to claim 4, characterized in that, On the first measurement channel, after the first differential amplifying unit is further connected: a first low-pass filtering unit and a first analog-to-digital conversion unit; On the second measurement channel, after the second differential amplifying unit is further connected: a second low-pass filtering unit and a second analog-to-digital conversion unit; The first low-pass filtering unit and the second low-pass filtering unit may perform low-pass filtering on the amplified analog response voltage signals of the first measurement channel and the second measurement channel, respectively; The first analog-to-digital conversion unit and the second analog-to-digital conversion unit may respectively convert analog response voltage signals filtered by the first measurement channel and the second measurement channel into corresponding digital response voltage signals. 6. The full impedance measurement device according to claim 5, characterized in that, further provided with a digital signal processing unit; The input ends of the digital signal processing unit are respectively connected with the output ends of the first analog-to-digital conversion unit and the second analog-to-digital conversion unit; The digital signal processing unit can integrate the digital response voltage signals of the first measurement channel and the second measurement channel, run a three-parameter sine wave fitting algorithm, and obtain the response voltage values of the sensor and the reference resistor. 7 . The full impedance measurement device according to claim 5 , wherein the digital-to-analog conversion unit, the first analog-to-digital conversion unit, and the second analog-to-digital conversion unit are controlled and operated by the same clock source. 8 . at the same sampling frequency. 8 . The full impedance measurement device according to claim 6 , wherein the reference resistance is a metal foil resistance with a temperature coefficient within 20PPM/°C. 9 .

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