CN108832305B - Cassegrain vortex field antenna based on super surface - Google Patents
Cassegrain vortex field antenna based on super surface Download PDFInfo
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- H—ELECTRICITY
- H01—ELECTRIC ELEMENTS
- H01Q—ANTENNAS, i.e. RADIO AERIALS
- H01Q19/00—Combinations of primary active antenna elements and units with secondary devices, e.g. with quasi-optical devices, for giving the antenna a desired directional characteristic
- H01Q19/10—Combinations of primary active antenna elements and units with secondary devices, e.g. with quasi-optical devices, for giving the antenna a desired directional characteristic using reflecting surfaces
- H01Q19/18—Combinations of primary active antenna elements and units with secondary devices, e.g. with quasi-optical devices, for giving the antenna a desired directional characteristic using reflecting surfaces having two or more spaced reflecting surfaces
- H01Q19/19—Combinations of primary active antenna elements and units with secondary devices, e.g. with quasi-optical devices, for giving the antenna a desired directional characteristic using reflecting surfaces having two or more spaced reflecting surfaces comprising one main concave reflecting surface associated with an auxiliary reflecting surface
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- H—ELECTRICITY
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- H01Q—ANTENNAS, i.e. RADIO AERIALS
- H01Q15/00—Devices for reflection, refraction, diffraction or polarisation of waves radiated from an antenna, e.g. quasi-optical devices
- H01Q15/14—Reflecting surfaces; Equivalent structures
- H01Q15/16—Reflecting surfaces; Equivalent structures curved in two dimensions, e.g. paraboloidal
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Abstract
The invention discloses a Cassegrain vortex field antenna based on a super surface, which mainly solves the problems that the conventional vortex field antenna is large in focal length, large in phase compensation error and large in occupied space of the whole antenna. It includes carrier (1), primary reflector (2), secondary reflector (3), feed (4) and bearing structure (5), the carrier adopts concave surface structure, primary reflector is conformal with the carrier, it is main, secondary reflector all adopts the super surface structure of phase place sudden change, the focus of primary reflector is less than the geometric focus of carrier, be used for realizing the short burnt effect, primary reflector is parabolic cylinder structure, this primary reflector includes main dielectric layer, main reflection stratum and main phase control layer, wherein main phase control layer is by a plurality of align to grid, and constitute according to the main becket micro-structure of heliciform global distribution, be used for producing the vortex electromagnetic wave. The invention can shorten the focal length of the Cassegrain vortex field antenna, reduce phase compensation errors, efficiently excite vortex electromagnetic waves and can be used for communication and radar.
Description
Technical Field
The invention belongs to the technical field of antennas, and particularly relates to a Cassegrain vortex field antenna which can be used for communication and imaging.
Technical Field
In recent years, the demand for communication capacity is rapidly increased, and vortex electromagnetic wave communication has good orthogonality due to different modes, so that a large number of same-frequency multiplexing channels can be formed, and the frequency spectrum utilization rate and the communication capacity are greatly improved, so that the research is focused on people. In vortex electromagnetic wave communication, vortex electromagnetic waves are effectively excited as a key link, and a vortex field antenna with good directionality and high-quality spiral phase distribution can realize remote transmission, identification and multiplexing of the vortex electromagnetic waves. The Cassegrain antenna is characterized in that a hyperboloid auxiliary reflecting surface is added on the basis of a parabolic antenna, and electromagnetic waves are reflected by the auxiliary reflecting surface and a main reflecting surface to obtain a highly directional radiation pattern. Compared with a common parabolic antenna, the radiation performance of the antenna can be optimized by the aid of the auxiliary reflecting surface added to the Cassegrain antenna, the feed source is placed close to the vertex of the main reflecting surface, the length of the feed line is obviously shortened, loss and system noise coefficient are reduced, a super-surface structure with small phase gradient change is introduced to the main reflecting surface and the auxiliary reflecting surface, accurate adjustment and control of vortex field phases can be achieved, and vortex electromagnetic waves can be efficiently excited.
However, after the geometric structure of the parabolic main reflecting surface of the cassegrain antenna is determined, the focal length of the antenna is also determined, flexible adjustment of the focal length cannot be realized, if the focal length is shortened, the curvature of the parabolic main reflecting surface is increased, the height of the main reflecting surface under the same aperture is increased, and higher requirements are put forward for antenna processing.
In the existing research, a microwave reflecting surface is mostly adopted to construct a vortex field antenna to excite vortex electromagnetic waves, for example, in the invention of Chinese patent with application publication number CN 105322285A and name of "an orbital angular momentum antenna", the invention discloses an orbital angular momentum antenna, which comprises a paraboloid reflecting surface and a spiral antenna feed source, wherein the center corresponding to the spiral with the minimum radius of the spiral antenna feed source is positioned at the focus of the paraboloid reflecting surface, the spiral antenna feed source generates vortex electromagnetic waves, and emergent waves are obtained after the vortex electromagnetic waves are reflected by the paraboloid. Although the antenna can realize the excitation of the vortex electromagnetic field to a certain extent, the vortex electromagnetic wave is directly generated as the feed source, the feed source configuration is complex, the feed source must be placed at the position of the geometric focal length of the paraboloid, the focal length cannot be shortened, and the occupied space of the antenna is larger.
Disclosure of Invention
The invention aims to provide a cassegrain vortex field antenna based on a super surface aiming at the defects of the prior art, so that the structure of the antenna is simplified, the phase compensation error is reduced, the focal length of the cassegrain vortex field antenna is reduced, and the occupied space of the antenna is saved.
In order to achieve the above purpose, the Cassegrain vortex field antenna based on the super surface comprises
The reflector comprises a carrier 1, a main reflector 2, an auxiliary reflector 3, a feed source 4 and a support structure 5, wherein the main reflector 2 is conformal with the carrier 1, the conformal structure is a central hollow structure, the feed source 4 adopts a pyramid horn antenna, the support structure 5 consists of four hard plastic rods, and each plastic rod is respectively connected with the same-side end points of the main reflecting surface 2 and the auxiliary reflecting surface 3; the method is characterized in that:
the carrier 1 adopts a concave structure; the main reflector 2 adopts a phase mutation concave super-surface structure constructed based on the generalized Snell's law, and the focal length of the main reflector 2 is smaller than the geometric focal length of the carrier 1, so that the overall height of the antenna is reduced; the secondary reflector 3 adopts a hyperbolic characteristic phase mutation super-surface structure constructed based on the generalized Snell's law;
the main reflector 2 comprises a main dielectric layer 21, a main reflecting layer 22 and a main phase regulating layer 23, wherein the main phase regulating layer 23 is composed of m × n main metal ring microstructures 231 which are uniformly distributed, the scattering parameter phase of each auxiliary metal ring microstructure is different, all the main metal ring microstructures 231 are integrally distributed in a spiral shape and used for generating vortex electromagnetic waves, m is larger than or equal to 12, and n is larger than or equal to 12.
Preferably, the concave structure adopted by the carrier 1 is a concave paraboloid cylindrical structure formed by translating an upward opening parabola, and the concave paraboloid cylindrical structure is bent upwards from the center to two side edges along the vertical direction of a generatrix of the cylindrical surface, the bending degree follows the equation of the upward opening parabola, and the center thickness is smaller than the edge thickness.
Preferably, the size of the hollow cross section of the conformal center of the main reflector 2 and the carrier 1 is the same as that of the cross section of the waveguide part of the pyramidal horn antenna, and the feed source 4 is installed at the hollow position.
Preferably, the main medium layer 21 is a concave structure, the upper surface of which is printed with a main phase control layer 23, and the lower surface of which is printed with a main reflection layer 22;
preferably, each primary metal ring microstructure 231 has a size corresponding to an incident angle θ of an incident electromagnetic wave at a position thereof with respect to the primary mirror 2i1And the phase compensation value phi (x, y, z). All primary metal ring microstructures 231 have a gradually decreasing phase gradient from the center to the edge.
Preferably, the secondary reflector 3 is of a square structure and comprises a secondary dielectric layer 31, a secondary reflection layer 32 and a secondary phase regulation layer 33, the secondary reflection layer 32 is printed on the upper surface of the secondary dielectric layer 31, the secondary phase regulation layer 33 is printed on the lower surface of the secondary dielectric layer 31, the phase regulation layer 33 is composed of i × j secondary metal ring microstructures 331 which are evenly etched on the upper surface of the secondary dielectric layer 31, i is more than or equal to 4, j is more than or equal to 4, and the size of each secondary metal ring microstructure 331 is determined by the incident angle theta of electromagnetic waves at the position of the secondary metal ring microstructure 331 relative to the secondary reflector (3)i2And the phase compensation value phi (x, y).
Preferably, the feed source 4 is a pyramidal horn antenna.
Compared with the prior art, the invention has the following advantages:
1. the main reflecting surface adopts the concave mirror, and the phase mutation super-surface structure constructed based on the generalized Snell's law is introduced on the concave main reflecting mirror, so that the focal length of the main reflecting mirror is shortened, and the space occupied by the antenna is saved.
2. The main reflector and the secondary reflector are both composed of the dielectric layer, the reflecting layer printed on one side surface of the dielectric layer and the phase regulating layer printed on the other side surface of the dielectric layer, so that the invention has the characteristics of simple structure, easy processing and low cost.
3. According to the invention, as the phase control layers of the main reflector and the secondary reflector are arranged according to the change of the incident angle of the electromagnetic wave, the precision of phase compensation is improved, and vortex electromagnetic wave can be efficiently excited.
Drawings
FIG. 1 is a schematic view of the overall structure of the present invention;
FIG. 2 is a schematic view of the main mirror structure of the present invention;
FIG. 3 is a schematic view of the construction of the secondary mirror of the present invention;
FIG. 4 is a schematic diagram of the electromagnetic wave propagation path and feed source design principle of the present invention;
FIG. 5 is a two-dimensional radiation pattern at a frequency of 20GHz according to an embodiment of the invention;
FIG. 6 is a simulation diagram of S11 at 19.0 GHz-21.0 GHz according to an embodiment of the invention;
FIG. 7 is a cross-sectional view of the xoy plane at 20GHz frequencies with electric fields of 375mm, 750mm, 1500mm, and 3000mm, respectively, for an embodiment of the invention.
Detailed Description
The invention is further described below with reference to the following figures and specific examples.
Referring to fig. 1, the present invention comprises a carrier 1, a primary mirror 2, a secondary mirror 3, a feed 4 and a support structure 5. Carrier 1 is located the holistic below of antenna, and the conformal inlaying of main mirror 2 is at the upper surface of carrier 1, and both centers fretwork for installation feed 4, feed 4 adopt pyramid horn antenna, divide into waveguide part and flare angle part, and this waveguide part is standard WR51 waveguide. A secondary mirror 3 is located directly above the primary mirror 2 and the feed 4, which secondary mirror 3 is connected to the primary mirror 2 by a support structure 5.
The numerical values of the hollow positions are quantized as follows:
a Cartesian coordinate system is established by taking the center of the upper surface of the main reflector 2 as a coordinate origin, the x axis is along the bending direction of the cylindrical surface, the y axis is along the generatrix direction of the cylindrical surface, and the z axis is perpendicular to the x axis and the y axis. Because the standard WR51 waveguide section size of the waveguide part of the horn antenna is the same as the hollow section size, the variation interval of the hollow position of the carrier 1 along the coordinate x is [ -7.495mm,7.495mm ], the variation interval along the coordinate y is [ -4.255mm,4.255mm ], and the variation interval along the coordinate z is [ -10mm, -0.5mm ] according to the specific size of the standard WR51 waveguide. The variation range of the hollow position of the main reflector 2 along the coordinate x is [ -7.495mm,7.495mm ], the variation range along the coordinate y is [ -4.255mm,4.255mm ], and the variation range along the coordinate z is [ -0.5mm, 0mm ].
The carrier 1 is curved upward along the x-axis from the center to both side edges, the degree of curvature following the parabolic equation with the opening upward: z (1/600) x, the center thickness being less than the edge thickness.
The main reflector 2, the secondary reflector 3 and the feed source 4 are arranged in a positive feed mode, namely, the central points of the main reflector 2, the secondary reflector 3 and the feed source 4 are on the same straight line. The supporting structure 5 is composed of four rigid plastic rods, each plastic rod is connected with the same side end point of the main reflecting surface 2 and the auxiliary reflecting surface 3, and the length of each plastic rod is 139.18mm in the present embodiment but not limited thereto.
Referring to fig. 2, the main reflector 2 has a concave structure, and includes a main dielectric layer 21, a main reflective layer 22, and a main phase control layer 23, where the main reflective layer 22 is printed on the lower surface of the main dielectric layer 21, and the main phase control layer 23 is printed on the upper surface of the main dielectric layer 21.
The main dielectric layer 21 is a concave paraboloid cylindrical structure, the thickness of the medium is 0.5mm, the relative dielectric constant is 4.4, and the relative magnetic permeability is 1, the length of the main dielectric layer 21 along the x axis is 222.40mm, the length of the main dielectric layer 21 along the y axis is 225mm, the size is set mainly by considering the precondition that the integral main mirror 2 can obtain better wave front calibration effect under the design frequency of 20GHz when the integral main mirror has enough electrical size, the variation interval of the main dielectric layer 21 along the coordinate x is [ -111.2mm, 111.2mm ], the variation interval along the coordinate y is [ -112.5mm, 112.5mm ], and the variation interval along the coordinate z is [ -0.5mm,27.00 mm.
The main reflective layer 22 is composed of a concave paraboloid cylindrical metal plate, and is embedded on the lower surface of the main dielectric layer 21, and since the size value of the main reflective layer 22 cannot be larger than the size of the main dielectric layer 21, according to the variation interval of the coordinate value of the main dielectric layer 21, the present embodiment is not limited to the case where the central coordinate of the main reflective layer 22 is (0, 0, -0.5mm), the variation interval along the coordinate x is [ -111.2mm, 111.2mm ], the variation interval along the coordinate y is [ -112.5mm, 112.5mm ], and the variation interval along the coordinate z is [ -0.5mm,27.00mm ].
The main phase control layer 23 is composed of 3576 main metal ring microstructures 231 uniformly arranged on the upper surface of the main dielectric layer 21 for generating vortex electromagnetic waves. The primary metal ring microstructure 231 is a square metal ring, since the primary metal ring microstructure 231The range of the coordinate values of (a) is not larger than the size of the main dielectric layer 21, so that the variation range of the main metal ring microstructure 231 along the coordinate x is [ -109.83mm, 109.83mm, according to the variation range of the coordinate values of the main dielectric layer 21]The variation range along the coordinate y is [ -112.5mm, 112.5mm]The variation interval along the coordinate z is [0mm, 27.00mm ]]The centers of adjacent primary metal ring microstructures 231 were spaced 3.75mm apart in the x-direction and 3.75mm apart in the y-direction. Side length L of each main metal ring microstructure 2311And line width w1Angle of incidence theta of the incident electromagnetic wave with respect to the main mirror 2 from the position thereofi1And the phase compensation value Φ (x, y, z), the phase compensation value Φ (x, y, z) at the position of each main metal ring microstructure 231 is firstly required to satisfy the requirement of reflecting the plane wave, and is calculated as follows:
where d Φ ═ k (sin θ)i1-sinθr1) dr denotes the derivative of phi (x, y, z) with respect to r, where
θi1Is the angle of incidence, θ, of the incident electromagnetic wave with respect to the primary mirror 2r1In order to reflect the electromagnetic wave at a reflection angle with respect to the main mirror 2, k is 24 °/mm, which is a 20GHz electromagnetic wave propagation constant, f is 101.58mm, which is a focal length of the main mirror 2, Φ0Is an arbitrary constant phase value. In order to make the main metal ring microstructure 231 satisfy the phase compensation value of the reflected vortex electromagnetic wave on the basis, a vortex factor M θ can be added to the original formula, where M represents the modal value of the electromagnetic vortex, θ is the vortex angle, and finally the phase compensation value Φ (x, y, z) of the main metal ring microstructure 231 is calculated as follows:
since the phase compensation value phi (x, y, z) required by the primary metal ring microstructure 231 is increased by the phase of the vortex angle theta relative to the plane wave antenna, the structural parameters of the primary metal ring microstructure need to be realized within a wider rangeAlternatively, the structural parameters that each primary metal ring microstructure 231 satisfies are determined from the phase compensation values Φ (x, y, z) that the primary metal ring microstructures 231 need to satisfy at different position coordinates, and these parameters include: incident angle thetai1Is [0 DEG ], 57.63 DEG]The variation interval of phase compensation value phi (x, y, z) is [ -180 DEG, 180 DEG]Length of side L1The variation interval is [1.12mm, 3.5mm]Line width w1The variation interval is [0.1mm,0.55mm]All the primary metal ring microstructures 231 are distributed in a spiral shape as a whole, and the phase gradient from the center to the edge becomes gradually smaller.
Referring to fig. 3, the secondary reflector 3 includes a secondary dielectric layer 31, a secondary reflective layer 32, and a secondary phase adjusting layer 33, where the secondary dielectric layer 31 is square, the secondary reflective layer 32 is printed on the upper surface of the secondary dielectric layer, and the secondary phase adjusting layer 33 is printed on the lower surface of the secondary dielectric layer.
In this example, it is assumed, but not limited to, that the sub-medium layer 31 has a thickness of 0.5mm, a relative dielectric constant of 4.4, and a relative magnetic permeability of 1, and that the sub-medium layer 31 has a variation range of [ -22.5mm,22.5mm ] along the coordinate x, a variation range of [ -22.5mm,22.5mm ] along the coordinate y, and a variation range of [86.1mm, 86.6mm ] along the coordinate z.
The secondary reflection layer 32 is composed of a square plane metal plate and is embedded on the upper surface of the secondary dielectric layer 31, and since the size value of the secondary reflection layer 32 cannot be larger than that of the secondary dielectric layer 31, the secondary reflection layer 32 has a central coordinate of (0, 0, 86.6mm), a variation range of [ -22.5mm,22.5mm ] along the coordinate x, a variation range of [ -22.5mm,22.5mm ] along the coordinate y, and a fixed coordinate value z of 86.6mm along the coordinate z.
The secondary phase adjusting and controlling layer 33 is composed of a plurality of secondary metal ring microstructures 331 uniformly distributed on the lower surface of the secondary dielectric layer 31, the number of the secondary metal ring microstructures 331 is determined by the size of the secondary phase adjusting and controlling layer 33, in this example, but not limited to, 324 secondary metal ring microstructures 331 are taken, the secondary metal ring microstructures 331 are square metal rings, the distance between the centers of the adjacent secondary metal ring microstructures 331 in the x direction is 2.5mm, the distance in the y direction is 2.5mm, and the change interval of the secondary metal ring microstructures 331 along the coordinate x is [ -21.25mm,21.25mm]The variation range along the coordinate y is [ -21.25mm,21.25mm]Having fixed coordinates along the coordinate zThe value z is 86.1 mm. Side length L of each secondary metal ring microstructure 3312And line width w2Angle of incidence theta of electromagnetic wave with respect to the sub-mirror 3 from its locationi2And a phase compensation value phi (x, y), the phase compensation value phi (x, y) of the position of each secondary metal ring microstructure 331 is calculated as follows:
where d Φ ═ k (sin θ)i2-sinθr2) dr denotes the derivative of phi (x, y) with respect to r, where
θi2Is the angle of incidence, θ, of the incident electromagnetic wave with respect to the secondary mirror 3r2In order to reflect the electromagnetic wave at a reflection angle with respect to the sub-mirror 3, k is 24 °/mm and 20GHz propagation constant, L is 48mm, and L is the distance between the phase center of the feed source 4 and the sub-phase adjustment layer 33, Lh38.1mm is the distance between the phase center of the feed source 4 and the main phase regulation layer 23; l + Lh86.1mm is the distance between the secondary phase adjusting layer 33 and the primary phase adjusting layer 23, which is equal to the z-axis coordinate value of each secondary metal ring microstructure 331, i.e. the fixed coordinate value z ═ L + Lh86.1mm and satisfies f>l+Lh，Φ0Is an arbitrary constant phase value.
Determining the structural parameters of each secondary metal ring microstructure 331 according to the phase compensation values phi (x, y) required to be satisfied by calculating the secondary metal ring microstructures 331 at different position coordinates, wherein the parameters comprise the incidence angle thetai2Phase compensation value phi (x, y) and side length L2Line width w2I.e. angle of incidence thetai2Is [0 DEG ], 31.68 DEG]The variation range of the phase compensation value phi (x, y) is [ -171.35 DEG, 179.28 DEG]Length of side L2The variation interval is [1.12mm, 2.3mm ]]Line width w2The variation interval is [0.1mm,0.55mm ]]。
Referring to fig. 4, the phase center F1 of the feed source 4 is located at the center of the foremost opening face of the angular aperture portion in the z direction, and has coordinates of (0, 0, 38.1mm), and the sub-mirrorThe virtual focus F2 of the sub-mirror 3 coincides with the focus of the main mirror 2 with coordinates (0, 0, 101.58mm), and the real focus of the sub-mirror 3 coincides with the phase center F1 of the feed 4. The virtual focal length of the secondary reflector 3 is f-L-Lh15.48mm, and the real focal length is L48 mm, and f-L-L is satisfiedh<l。
The pyramid horn antenna used by the feed source 4 comprises a waveguide part and an opening angle part, wherein the waveguide part is a standard WR51 waveguide, the single-mode transmission frequency range is 14.5 GHz-22.0 GHz, the change interval of the waveguide part along a coordinate x is [ -7.495mm,7.495mm ], the change interval along the coordinate y is [ -4.255mm,4.255mm ], and the change interval along the coordinate z is [ -10mm,0mm ] according to the size value of the standard WR51 waveguide. The angular sector has a variation range of [ -11.43mm,11.43mm ] along the coordinate x, a variation range of [ -8.89mm,8.89mm ] along the coordinate y, and a variation range of [0mm, 38.1mm ] along the coordinate z. The length A of the opening long edge of the forefront end of the opening angle part along the x axis is 22.86mm, the side length d of the auxiliary reflector 3 can be obtained from the change interval [ -22.5mm,22.5mm ] of the auxiliary reflector 3 along the coordinate x is 45mm, and A and d satisfy the following relational expression:
where f is 101.58mm, L is the focal length of the main mirror 2h38.1mm is the distance between the phase center of the feed 4 and the center of the main phase adjusting layer 23 of the main mirror 2.
The present example, but not limited to, shows that the focal length of the primary mirror 2 is 101.58mm, and the geometric focal length of the carrier 1 is 150mm, and the focal length of the primary mirror 2 is 32.28% shorter than the focal length of the carrier 1, which illustrates the effect of achieving a shorter focal length.
The technical effects of the present invention will be further described in detail with reference to the results of simulation experiments.
1. Simulation conditions and contents:
and electromagnetic simulation software CST 2017.
As can be seen from fig. 5(a), the angles of the radiation directions of the two main beams in the E plane of the embodiment of the present invention are-4 ° and 4 °, wherein the gain of the-4 ° main beam is 22.02dBi, and the gain of the 4 ° main beam is 22.97dBi, which indicates that the present invention can obtain a larger gain in the E plane.
As can be seen from fig. 5(b), the angles of the radiation directions of the two main beams in the H plane are-3 ° and 3 °, wherein the gain of the-3 ° main beam is 22.99dBi, and the gain of the 3 ° main beam is 21.74dBi, which indicates that the present invention can obtain a larger gain in the H plane.
As can be seen from FIG. 6, S11 is all lower than-10 dB in the frequency band of 19.0GHz to 21.0GHz, which shows that the embodiment of the invention has good matching characteristics.
Fig. 7 shows the electric field distribution in the square observation plane with the side length of 375mm when the distance from the antenna is 375mm, 750mm, 1500mm, 3000mm respectively, and as can be seen from fig. 7, when the distance from the antenna is 375mm and 750mm, the observation plane is located in the near field region of the antenna, the electric field distribution is relatively disordered, when the distance from the antenna is 100 wavelengths and 200 wavelengths, the observation plane is located in the far field region of the antenna, the electric field distribution is spiral, it is in accordance with the conclusion that the phase value of the electric field distribution rotating for a circle changes 360 °, and the phase value in the opposite angular direction is opposite.
In conclusion, the vortex electromagnetic wave transmitting antenna is used for transmitting vortex electromagnetic waves, can shorten the focal length of a vortex field antenna, reduces phase compensation errors, simplifies the antenna structure, and is suitable for the fields of communication, imaging and the like.
Claims (9)
1. A Cassegrain vortex field antenna based on a super surface comprises a carrier (1), a main reflector (2), an auxiliary reflector (3), a feed source (4) and a supporting structure (5), wherein the main reflector (2) is conformal to the carrier (1), the conformal center is of a hollow structure, the feed source (4) adopts a pyramid horn antenna, the supporting structure (5) consists of four rigid plastic rods, and each plastic rod is respectively connected with the same-side end points of the main reflector (2) and the auxiliary reflector (3); the method is characterized in that:
the carrier (1) adopts a concave surface structure; the main reflector (2) adopts a phase mutation concave super-surface structure constructed based on the generalized Snell's law, and the focal length of the main reflector is smaller than the geometric focal length of the carrier, so that the overall height of the antenna is reduced; the secondary reflector (3) adopts a hyperbolic characteristic phase mutation super-surface structure constructed based on the generalized Snell's law;
the main reflector (2) comprises a main dielectric layer (21), a main reflecting layer (22) and a main phase regulating layer (23), wherein the main phase regulating layer (23) is composed of m × n main metal ring microstructures (231) which are uniformly distributed, the phase compensation value of each main metal ring microstructure (231) is different, all the main metal ring microstructures (231) are integrally distributed in a spiral shape and used for generating vortex electromagnetic waves, m is more than or equal to 12, n is more than or equal to 12, and the size of each main metal ring microstructure (231) is determined by the incident angle theta of the incident electromagnetic waves relative to the main reflector (2) at the position of the main metal ring microstructure (231)i1And phase compensation value phi (x, y, z);
all primary metal ring microstructures (231) have a decreasing phase gradient from center to edge.
2. The antenna of claim 1, wherein: the concave surface structure adopted by the carrier (1) is a concave paraboloid cylindrical structure formed by translating a parabola with an upward opening, the concave paraboloid cylindrical structure is upwards bent from the center to two side edges along the vertical direction of a cylindrical surface generatrix, the bending degree follows the parabola equation with the upward opening, and the center thickness is smaller than the edge thickness.
3. The antenna of claim 1, wherein: the size of the hollow cross section of the conformal center of the main reflector (2) and the carrier (1) is the same as that of the section of the waveguide part of the pyramidal horn antenna, and the feed source (4) is installed at the hollow position.
4. The antenna of claim 1, wherein: the main medium layer (21) is of a concave structure, the main phase regulating layer (23) is printed on the upper surface of the main medium layer (21), and the main reflecting layer (22) is printed on the lower surface of the main medium layer (21).
5. The antenna of claim 1, wherein: the auxiliary reflector (3) is of a square structure and comprises an auxiliary dielectric layer (31), an auxiliary reflecting layer (32) and an auxiliary phase adjusting layer (33); the secondary reflection layer (32) is printed on the upper surface of the secondary dielectric layer (31), and the secondary phase adjustment layer (33) is printed on the lower surface of the secondary dielectric layer (31).
6. The antenna of claim 5, wherein the secondary phase control layer (33) is composed of i × j secondary metal ring microstructures (331) uniformly etched on a dielectric substrate, i is larger than or equal to 4, j is larger than or equal to 4, and the size of each secondary metal ring microstructure (331) is determined by the incident angle theta of incident electromagnetic waves at the position of each secondary metal ring microstructure relative to the secondary reflector (3)i2And the phase compensation value Φ (x, y) determines:
the phase compensation value phi (x, y) of the position of each secondary metal ring microstructure (331) is calculated as follows:
7. The antenna of claim 1, wherein: and the virtual focus of the secondary reflector (3) is positioned above the secondary reflector (3), the real focus is positioned below the secondary reflector (3), the virtual focus is coincided with the focus of the main reflector (2), and the real focus is coincided with the phase center of the feed source (4).
8. The antenna of claim 1, wherein: the virtual focal length of the secondary reflector (3) is f-L-LhThe real focal length is L and satisfies f-L-Lh<L, where f is the focal length of the main mirror (2), LhIs the distance between the phase center of the feed source (4) and the main phase regulating layer (23) of the main reflector (2).
9. The antenna of claim 1, wherein: the length A of the long edge of the opening at the forefront end of the flare angle part of the pyramidal horn antenna adopted by the feed source (4) and the length d of the side of the secondary reflector (3) satisfy the following relational expression:
wherein f is the main mirror: (2) Focal length of (L)hIs the distance between the phase center of the feed source (4) and the main phase regulation layer (23).
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| CN108832305A (en) | 2018-11-16 |
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