CN108011600B - Reconfigurable radio frequency and microwave power amplifier - Google Patents
Reconfigurable radio frequency and microwave power amplifier Download PDFInfo
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- CN108011600B CN108011600B CN201711463812.XA CN201711463812A CN108011600B CN 108011600 B CN108011600 B CN 108011600B CN 201711463812 A CN201711463812 A CN 201711463812A CN 108011600 B CN108011600 B CN 108011600B
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Abstract
The invention relates to a reconfigurable radio frequency and microwave power amplifier, which comprises an input radio frequency switch, an adjustable input matching network, a power amplification stage, an adjustable output matching network, an output radio frequency switch, a tunable wave impedance control network and a bias and control circuit which are connected in sequence; the adjustable input matching network and the adjustable output matching network respectively comprise at least one stage of LC network with a fundamental wave impedance conversion function, the LC network comprises at least one reconfigurable device, and the reconfigurable device is connected with a passive device in the LC network in parallel. The invention realizes the reconfigurable harmonic impedance control network and fundamental impedance matching network through the adjustable input matching network, the tunable wave impedance control network and the adjustable output matching network, and can reconstruct the load impedance network in real time when the power amplifier works in different bands of the linear power amplification mode, thereby meeting the fundamental impedance and harmonic load required by each frequency band.
Description
Technical Field
The invention relates to the field of mobile communication, in particular to a reconfigurable radio frequency and microwave power amplifier.
Background
In the field of mobile communication represented by smart phones, a radio frequency and microwave power amplifier (power amplifier Module, PA Module, or PAM, hereinafter referred to as a radio frequency power amplifier or a power amplifier) is a very important element in a communication terminal, and its performance directly affects the quality of mobile communication and also directly determines the single charging service life of the mobile communication terminal device. Radio frequency engineers and chip design engineers must continually improve power amplifier performance while controlling or reducing the complexity and cost of the radio frequency front-end.
At present, a saturated power amplifier is needed by a 2G wireless communication terminal, a linear power amplifier is needed by a 3G/4G/5G terminal, and the linear power amplifier needed by the 3G/4G/5G terminal is needed to meet fundamental wave impedance needed by the frequency band and harmonic wave load impedance needed by an F-type power amplifier on different bands in different paths, so that the linearity and the power addition efficiency are low when the power amplifier module is used as the linear power amplifier. In addition, although the 3G/4G/5G frequency band covers the 2G frequency band, two current power amplifiers still have to coexist in the radio frequency front end of the smart phone, and work in the 2G mode and the 3G/4G mode respectively, so that the complexity and the cost of the radio frequency front end further increase with the arrival of the 5G communication mode. Meanwhile, the multimode multi-frequency power amplifier modules required by the 4G and 5G require higher linearity and efficiency, and the bandwidth covered by the amplifier modules is increased, which is difficult to achieve in the prior art.
Disclosure of Invention
The invention aims to provide a reconfigurable radio frequency and microwave power amplifier, aiming at a 3G/4G/5G linear power amplifier module, fundamental wave impedance and harmonic wave load impedance required by F-type power amplifier on different bands in the same amplifying path are met in real time, and linearity and power addition efficiency of the module as linear power amplifier are improved. In addition, the linear power amplifier is reconstructed to enable the linear power amplifier to be compatible with a working mode required by the saturated power amplifier, a 2G GSM amplifier module in the radio frequency front end is omitted, and complexity and cost of the radio frequency front end are reduced.
The technical scheme for solving the technical problems is as follows: the reconfigurable radio frequency and microwave power amplifier comprises an input radio frequency switch, an adjustable input matching network, a power amplification stage, an adjustable output matching network, an output radio frequency switch, a tunable wave impedance control network and a bias and control circuit which are sequentially connected, wherein the tunable wave impedance control network is connected between a common end of the power amplification stage and the adjustable output matching network and the ground;
The adjustable input matching network and the adjustable output matching network respectively comprise at least one stage of LC network with a fundamental wave impedance conversion function, the LC network comprises at least one reconfigurable device, and the reconfigurable device is connected with a passive device in the LC network in parallel;
The tunable wave impedance control network comprises a fixed harmonic impedance control network and at least one reconfigurable device, the reconfigurable device being connected in parallel with a passive device in the fixed harmonic impedance control network;
the reconfigurable device is a reconfigurable capacitor or a reconfigurable inductor, the reconfigurable capacitor is formed by connecting a radio frequency switch and a capacitor in series, and the reconfigurable inductor is formed by connecting the radio frequency switch and the inductor in series;
The bias and control circuit is respectively connected with the input radio frequency switch, the power amplification stage, the output radio frequency switch, the radio frequency switch in the adjustable input matching network, the radio frequency switch in the adjustable output matching network and the radio frequency switch in the tunable wave impedance control network.
The beneficial effects of the invention are as follows: the reconfigurable radio frequency and microwave power amplifier realizes the reconfigurable harmonic impedance control network and the fundamental impedance matching network through the adjustable input matching network, the tunable wave impedance control network and the adjustable output matching network, and can reconstruct the load impedance network in real time when the power amplifier works in different bands of a linear power amplification mode, thereby meeting the fundamental impedance and harmonic load required by each frequency band.
On the basis of the technical scheme, the invention can be improved as follows.
Further, at least one backup impedance conversion network for providing an increased or decreased fundamental load impedance to the power amplifier stage in cooperation with the adjustable output matching network, the backup impedance conversion network being connected between the adjustable output matching network and the output radio frequency switch;
The standby impedance conversion network comprises a direct current blocking capacitor connected between the adjustable output matching network and the output radio frequency switch, and a reconfigurable impedance conversion network with one end connected between the direct current blocking capacitor and the output radio frequency switch and the other end grounded.
Further, the reconfigurable impedance conversion network is specifically a reconfigurable inductor, or a reconfigurable capacitor, or a series inductor and a reconfigurable capacitor, or a series capacitor and a reconfigurable inductor;
The bias and control circuit is also connected with a radio frequency switch in the reconfigurable impedance transformation network.
The beneficial effects of adopting the further scheme are as follows: the reconfigurable radio frequency and microwave power amplifier introduces a standby impedance conversion control network on the basis of realizing a reconfigurable harmonic impedance control network and a fundamental impedance matching network, so as to realize obviously reduced fundamental load impedance for the power amplifier, enable the 3G/4G/5G linear power amplifier to be used as a linear power amplifier and also to work in a saturated state, be compatible with a 2G communication mode, omit a GSM saturated power amplifier in the current communication terminal and reduce the complexity and cost of a radio frequency front end in mobile communication equipment.
Further, the power amplification stage comprises at least one stage of amplification stage of transistor, and when the power amplification stage comprises more than two stages of amplification stages, an interstage matching network is arranged between two adjacent stages of amplification stages.
Further, the inter-stage matching network is a fixed inter-stage matching network or an adjustable inter-stage matching network, and the fixed inter-stage matching network and the adjustable inter-stage matching network respectively comprise at least one stage of LC network with a fundamental wave impedance conversion function;
At least one reconfigurable device is also included in the tunable inter-stage matching network, and the reconfigurable device is connected in parallel with a passive device in an LC network of the tunable inter-stage matching network;
The bias and control circuit is also connected with a radio frequency switch in the adjustable inter-stage matching network.
Further, an equivalent circuit of the fixed harmonic impedance control network in the tunable wave impedance control network is that one capacitor is connected with one inductor in series, or one capacitor is connected with one inductor in parallel, or one capacitor is alone, or one inductor is alone.
Further, at least two radio frequency signal channels are arranged in the input radio frequency switch; and a plurality of radio frequency signal channels are arranged in the output radio frequency switch.
Further, a parasitic inductance suppression network for suppressing parasitic inductance is further provided in the tunable wave impedance control network, the parasitic inductance suppression network is arranged between a reconfigurable device in the tunable wave impedance control network and ground, and the parasitic inductance suppression network includes at least one reconfigurable device.
The beneficial effects of adopting the further scheme are as follows: parasitic inductance can be generated when a reconfigurable device in the tunable wave impedance control network works, so that the resonance point of a new resonance network after reconstruction is likely to deviate from expectations and deviate from a reconstruction target, and the parasitic inductance inhibits the setting of the network, so that the influence of the parasitic inductance can be avoided, and ideal harmonic impedance is provided.
Drawings
FIG. 1 is a block diagram of a radio frequency front end system in a typical 2G/3G/4G system capable smart phone;
FIG. 2 is a block diagram of a conventional 3G/4G power amplifier module;
FIG. 3 is a schematic diagram of a load matching structure of a first exemplary power amplifier according to the prior art;
FIG. 4 is a schematic diagram of a load matching structure of a second exemplary power amplifier according to the prior art;
FIG. 5 is a schematic diagram of a third exemplary load matching structure of a conventional power amplifier;
FIG. 6 is a schematic diagram of a load matching structure of a fourth exemplary power amplifier according to the prior art;
FIG. 7 is a schematic diagram of a fifth exemplary prior art power amplifier load matching architecture;
FIG. 8 is a schematic diagram of a typical prior art network structure for controlling the second harmonic impedance to be low;
FIG. 9 is a schematic diagram of an output matching network of a conventional power amplifier;
FIG. 10 is a schematic diagram of a first configuration of an adjustable output matching network in a reconfigurable RF and microwave power amplifier according to the present invention;
FIG. 11 is a schematic diagram of a second reconfigurable architecture of an adjustable output matching network in a reconfigurable RF and microwave power amplifier according to the present invention;
FIG. 12 is a schematic diagram of a third configuration of an adjustable output matching network in a reconfigurable RF and microwave power amplifier according to the present invention;
FIG. 13 is an equivalent structural schematic diagram of the entire matching network of FIG. 12;
FIG. 14 is a schematic diagram of a reconfigurable harmonic impedance control network in a reconfigurable RF and microwave power amplifier according to the present invention;
FIG. 15 is an equivalent structural schematic diagram of the entire network of FIG. 14;
FIG. 16 is a schematic diagram of another reconfigurable structure of a reconfigurable harmonic impedance control network in a reconfigurable RF and microwave power amplifier according to the present invention
FIG. 17 is a schematic diagram of a reconfigurable RF and microwave power amplifier compatible 2G/3G/4G three modes of construction;
Fig. 18 is a schematic diagram of the overall structure of a reconfigurable rf and microwave power amplifier according to the present invention.
Detailed Description
The principles and features of the present invention are described below with reference to the drawings, the examples are illustrated for the purpose of illustrating the invention and are not to be construed as limiting the scope of the invention.
First, the state of the art is analyzed:
in the 2G era, GSM is the main part, mobile communication only pursues good voice communication effect, the radio frequency power amplifier of a functional mobile phone only needs to cover 4 frequency bands of GSM850/EDGE900/DCS1800/PCS1900, and correspondingly, the power amplifier required by the 2G mobile communication terminal belongs to a saturated power amplifier, only has requirements on saturated output power, power addition efficiency and harmonic suppression in a saturated state, and has no requirement on linearity in power backspacing. After twenty-first century, the communication mode rapidly goes through EDGE/WCDMA/TDS-CDMA/CDMA2000 3G age, and goes into 4G age containing TD-LTE/LTE-FDD two communication systems, the data transmission rate of mobile communication is obviously improved, the theoretical peak rate of 20MHz 100RB64QAM FDD-LTE is 100Mbps, and compared with 14.4Mbps of 3GWCDMA, the data transmission rate is improved by one order of magnitude. And the data transmission rate of 100Mbps is only a theoretical value of a single antenna, if the MIMO technology (Multiple-Input Multiple-Output transmission technology) is adopted, namely a plurality of transmitting antennas and receiving antennas are respectively used at a transmitting end and a receiving end, so that signals are transmitted and received simultaneously through the transmitting end and the receiving end, the data transmission rate is improved, and in a4 x 4MIMO system, even if only 3/4 channel coding is adopted, the peak rate can be up to 300Mbps, the corresponding user experience is completely different, and a user who frequently uses a smart phone to surf the internet every day must have deep experience. The 3G/4G and 5G power amplifiers belong to linear power amplifiers, and important indexes include linearity, power additional efficiency and higher harmonics of certain specific frequency bands.
Fig. 1 is a block diagram of a typical radio frequency Front End (RFE) system in a smart phone supporting a 2G/3G/4G system, and the HB power amplifier path is omitted in fig. 1. As can be seen from fig. 1, the LB comprises two amplifiers, GSM and 3G/4G, and although the 3G/4G band has completely covered the GSM band, the GSM amplifier cannot be omitted since the saturated power amplifier and the linear power amplifier are not compatible with each other. Likewise MB also includes both GSM and 3G/4G amplifier paths.
With the proposal of phase2, phase3 and phase5 smart phone platform, a multimode multi-frequency mobile phone needs to support different operating frequency bands of multiple countries and regions, and for this purpose, many different devices are added from the output end of the linear power amplifier to the antenna, as shown in fig. 1, including two radio frequency switches (RF switches), a diplexer (duplex), and a diplexer (Diplexer). One of the RF switches is inside the power amplifier module and the other RF switch is between the diplexer and the diplexer. Each device passing through after the radio frequency power output of the power amplifier will necessarily increase some insertion loss to the power, taking the current most typical MTK phase2 mobile phone platform band8 (880 MHz-915 MHz) as an example, the most conventional single-pole-throw (SPST) radio frequency switch in industry will have a typical insertion loss value of 0.3-0.4 dB, whereas the radio frequency switch commonly used in the power amplifier module is a single-pole multi-throw radio frequency switch, the insertion loss of the conventional SP7T radio frequency switch in band8 is about 0.5dB, the insertion loss typical value of the diplexer is about 2.5dB, and the insertion loss typical value of the diplexer is about 0.5dB, which means that the radio frequency signal insertion loss is about 4dB from the output end of the power amplifier to the antenna end, in order to cancel these insertion losses, the output of the radio frequency power amplifier must have a margin of 4dB, that is the actual output value of the power amplifier is about 2.5 times the actual required value of the antenna end, which requires that the linear power amplifier can output higher power.
According to different frequency bands, the power amplification channels in the power amplifier module of the 4G smart phone are divided into three power amplification channels of low band/middle band/high band, and fig. 2 is a block diagram of the current 3G/4G power amplifier module in the industry, which can be seen to comprise three signal amplification channels, and the three signal amplification channels consist of three power amplifiers and radio frequency band selection switches corresponding to the three power amplifiers. The low band signal amplifying channel needs to cover all frequency bands including band5/band8/band12/band13/band14/band17/band20/band26/band27/band28, the frequency range is 699 MHz-915MHz,middle band, the frequency range is 1710 MHz-2025MHz,high band, and the frequency range is 2300 MHz-2690 MHz, the ratio of the bandwidth to the center frequency point of the frequency band is greater than 20% from the aspect of bandwidth analysis, that is, three power amplifiers belong to the broadband power amplifier, but the class F power amplifier capable of improving the linear output power and the power additional efficiency cannot realize broadband coverage, and in order to understand the problem, the design knowledge of the class F and class IF radio frequency power amplifiers is introduced below.
The DC power consumption expression of the power amplifier circuit is as follows:
Where P out,fund represents the fundamental power, which is the power that the power amplifier design engineer wants to get from the load, strictly speaking, the engineer always wants to get the rf power from the load as high as possible and maintain good linearity without sacrificing other important indexes of the power amplifier (including gain and power added efficiency PAE). And higher harmonic power And heat dissipation P dis caused by thermal parasitics can increase direct current power consumption P DC and reduce the efficiency of the power amplifier. However, whether the existence of the higher harmonic signals is a certain disadvantageous factor or not means that the design thought of the harmonic control type power amplifier is to realize specific harmonic impedance through a harmonic impedance control network, so that harmonic voltage signals exist at the collector of the amplifying unit, and certain specific signal waveforms, such as square waves and half sine waves, are formed by superposition with fundamental wave signals, thereby protecting the fundamental wave, enabling the fundamental wave signals to still not generate saturation distortion and cut-off distortion under a larger swing state, and enabling the power amplifier to output higher linear power under the premise of not sacrificing PAE.
The F-type power amplifier is a typical representative of a harmonic suppression type power amplifier, the design principle is that on the premise of not influencing an output matching network, the impedance of harmonic waves is controlled through a harmonic control network, harmonic control is carried out on a load end of the power amplifier, the harmonic control comprises even harmonic short circuit and odd harmonic open circuit (usually only second harmonic and third harmonic are processed), a voltage signal similar to a square wave and a current signal similar to half sine wave are obtained at an output end, and researches prove that the square wave voltage signal can obviously improve the linear output power of fundamental waves and improve the linearity index of the power amplifier, and corresponding knowledge can refer to RF power amplifier of wireless communication by Steve C.crips, chapter 6 of the second edition. Meanwhile, although the higher harmonic signal of the F-class power amplifier does not ensure that the voltage and the current signals are 0 at the same time, the even harmonic voltage component is 0, the odd harmonic current component is 0, and the power calculation formula is adopted
Pout,nf＝V*I*cosθ
Wherein no power is generated as long as one of the voltage and the current is 0. Therefore, the higher harmonic waves of the ideal F-type power amplifier do not generate power, the direct current power consumption is not increased, and the circuit efficiency of the power amplifier can be obviously improved.
The IF (inverse F) type power amplifier is a type of power amplifier with waveforms which are quite opposite to those of the F type, and by controlling an even harmonic impedance open circuit and an odd harmonic impedance short circuit (only second harmonic and third harmonic are usually processed), the voltage waveform presents a half sine wave, the current waveform is a rectangular wave, the performance of the achieved power amplifier is quite different from that of the F type, the IF type power amplifier can output higher saturated power, the power amplifier achieves higher power additional efficiency in a saturated state, but linearity in a power back-off state is sacrificed, so the IF type power amplifier concept is often used for designing 2G GSM power amplifiers.
According to the analysis, the linear power amplifier needs to adopt the F-type design principle, and the harmonic impedance control network is reasonably designed, so that the output power of the power amplifier is improved, and the power additional efficiency of the amplifier is improved. Unfortunately, the current industry design method cannot meet the harmonic impedance state required by each band in the broadband while meeting the class F, but only the band in the center of the band can be balanced to work in the class F, but the two ends of the band cannot be balanced, so that the performance of the side band is sacrificed.
Taking the amplifier of the MB amplifying path in fig. 2 as an example, according to the design principle of the class F power amplifier, the output matching network necessarily includes a harmonic impedance control network, and fig. 3 to 7 list five typical load matching structures of the power amplifier, where all the output matching networks include such a harmonic impedance control network. Fig. 8 is a typical network controlling the second harmonic impedance to be low, and for f 0 =1860 MHz, two passive device parameters in the second harmonic control network are required to meet the following resonance conditions
The design difficulty of the broadband RF and microwave power amplifier is that the second harmonic control network cannot meet the following conditions at the same time
However, only if the three are really satisfied at the same time, the MB power amplifier can be ensured to meet the design principle of the F-type power amplifier.
Taking an LB power amplifier as an example, f 1 represents the lowest frequency point 699MHz of the coverage frequency band required by the LB power amplifier, and f 2 represents the highest frequency point 915MHz. According to the design principle of the class F power amplifier, the load impedance of the second harmonic 2F 1～2f2 of the F 1～f2 needs to be designed as a short circuit to realize low impedance, and the load impedance of the third harmonic 3F 1～3f2 needs to be designed as an open circuit to realize high impedance. Therefore, 2f 2＜＜3f1 must be ensured. However, 2f 2＝1830MHz,3f1 =2097 MHz, which are very close to each other, and the current smart phone power amplifier design method, which uses a harmonic impedance control network with a fixed component value, cannot achieve such bandwidth requirements at all, which is the difficulty of actually achieving the broadband class F amplifier state of the radio frequency and microwave power amplifier.
Besides the design obstacle of broadband indexes faced by the harmonic impedance control network introduced above, the design of fundamental impedance is also faced by the design obstacle, the design difficulty of fundamental impedance is introduced below, the design obstacle formed by broadband indexes of the 4G radio frequency power amplifier module is analyzed, and the reason that the 3G/4G power amplifier and the 2G power amplifier still coexist in the scheme of the radio frequency front end of the mobile phone is analyzed.
As shown in fig. 2, in the rf power amplifier module, three amplification channels LB, MB, HB include, in addition to respective power amplifiers, corresponding input matching networks, inter-stage matching networks, and output matching networks, where the design targets of the three networks are similar, and a specific fundamental load Z opt,fund is implemented for a pre-amplification unit or transducer. The fundamental wave load Z opt,fund which is needed to be realized by the output matching network refers to the characteristic impedance of a fundamental wave frequency point which is presented backwards from the power stage S2 amplifying unit in the amplifying path, and is realized by the output matching network. At present, the fundamental wave load Z opt,fund is realized by adopting impedance conversion networks with different stages (usually one stage or two stages) and different structures through fixed passive devices in the industry, and the relationship between the fundamental wave output power Z opt,fund and the fundamental wave output power Z opt,fund of the power amplifier is as follows:
Where V DD is the dc bias voltage of the collector of the amplifying cell in fig. 2 and V knee is determined by the characteristics of the transistors used in the amplifying cell. According to the calculation formula, once the design index of the power amplifier is determined, if the power amplifier has only one optimal fundamental load Z opt,fund, the performance index of the power amplifier is directly deviated from the optimal value. It has also been found that 2G saturated power amplifiers require lower fundamental loading Z opt,fund than 3G/4G linear power amplifiers, as saturated power amplifiers seek greater saturated output power and power added efficiency in saturated conditions.
Taking the matching structure shown in fig. 4 as an example and taking the conventional fundamental wave load value Z opt,fund =3 ohm of the 4G linear power amplifier as an example, the parameters of the load matching network of the MB are calculated, so as to explain the design difficulty of the broadband radio frequency power amplifier.
The frequency band range that the internal MB channel of the power amplifier module needs to cover is 1710 MHz-2025 MHz, including common frequency bands such as band1(1920～1980MHz),band2(1850～1910MHz),band3(1717～1785MHz),band4(1710～1755MHz),band33(1900～1920MHz),band34(2010～2025MHz),band39(1880～1920MHz), and unusual frequency bands such as band 9/10/35/36/37. Because the different frequency bands need to share one MB power amplifying path, in order to maximize the performance of all the frequency bands, the design idea in the current industry is to design a load near a center frequency point f 0 =1860 MHz, calculate the value of each passive device in the matching network by the following formula,
ω*L1＝Zopt*Q
ω＝2*π*f0
L 1＝0.45nH,L2＝1.84nH,C1＝12.3pF,C2 = 3.01pF is calculated. The problem is that the component values in the matching network are all calculated according to f 0 =1860 MHz, so that Z opt,fund_1860 =3 ohm can be satisfied, the minimum frequency point f 1 =1710 MHz of the MB frequency band can obtain the load Z out,fund_1710 =2.69-j0.75ohm, the maximum frequency point f 2 =2025 MHz can obtain the load Z opt,fund_2025 =3.46+j0.46 ohm, the frequency band far from f 0 =1860 MHz can obtain the load gradually deviating from the 3ohm target impedance within the frequency band covered by the MB amplifying path, which means that the power amplifier can show linear degree degradation, efficiency reduction, large in-band gain fluctuation and the like on the side bands, and these are the most important performance indexes of the power amplifier module.
From the above analysis, on the one hand, the output matching network determines the fundamental load Z opt,fund of the power amplifier, which directly determines the saturated output power of the power amplifier; on the other hand, the harmonic impedance control network determines the harmonic impedance obtained by the amplifying unit, and the harmonic impedance directly affects the working state and type of the power amplifier, and determines the linear power, saturated power and power added efficiency of the power amplifier.
Aiming at the linear power amplifier, the current design method faces two problems, firstly, the output matching network cannot be designed to realize broadband fundamental wave impedance in a limited space inside a module; second, all bands cannot be satisfied simultaneously with the harmonic impedance required by class F power amplifiers.
For the radio frequency front end scheme in the mobile phone, the 3G/4G linear power amplifier and the 2G saturated power amplifier need the diametrically opposite harmonic impedance state and the fundamental wave load impedance with larger difference, the linear power amplifier needs the second harmonic impedance short circuit and the third harmonic impedance open circuit, the harmonic impedance state needed by the saturated power amplifier is opposite, and meanwhile, the saturated power amplifier needs a lower fundamental wave impedance value.
The present invention is explained in conjunction with the above summary of the prior art:
As shown in fig. 18, a reconfigurable radio frequency and microwave power amplifier comprises an input radio frequency switch, an adjustable input matching network, a power amplification stage, an adjustable output matching network, an output radio frequency switch, a tunable wave impedance control network and a bias and control circuit which are sequentially connected, wherein the tunable wave impedance control network is connected between a common end of the power amplification stage and the adjustable output matching network and the ground; the adjustable input matching network and the adjustable output matching network respectively comprise at least one stage of LC network with a fundamental wave impedance conversion function, the LC network comprises at least one reconfigurable device, and the reconfigurable device is connected with a passive device in the LC network in parallel; the tunable wave impedance control network comprises a fixed harmonic impedance control network and at least one reconfigurable device, the reconfigurable device being connected in parallel with a passive device in the fixed harmonic impedance control network; the reconfigurable device is a reconfigurable capacitor or a reconfigurable inductor, the reconfigurable capacitor is formed by connecting a radio frequency switch and a capacitor in series, and the reconfigurable inductor is formed by connecting the radio frequency switch and the inductor in series; the bias and control circuit is respectively connected with the input radio frequency switch, the power amplification stage, the output radio frequency switch, the radio frequency switch in the adjustable input matching network, the radio frequency switch in the adjustable output matching network and the radio frequency switch in the tunable wave impedance control network.
The invention further comprises at least one standby impedance transformation network for providing an increased or decreased fundamental load impedance to the power amplifier stage in cooperation with the adjustable output matching network, the standby impedance transformation network being connected between the adjustable output matching network and the output radio frequency switch; the standby impedance conversion network comprises a direct current blocking capacitor connected between the adjustable output matching network and the output radio frequency switch, and a reconfigurable impedance conversion network, one end of which is connected between the direct current blocking capacitor and the output radio frequency switch, and the other end of which is grounded;
The reconfigurable impedance conversion network is specifically a reconfigurable inductor, a reconfigurable capacitor, a series inductor and a reconfigurable capacitor, or a series capacitor and a reconfigurable inductor;
The bias and control circuit is also connected with a radio frequency switch in the reconfigurable impedance transformation network.
The power amplification stage comprises at least one stage of amplifying stage of transistor, and when the power amplification stage comprises more than two stages of amplifying stages, an interstage matching network is arranged between two adjacent stages of amplifying stages.
The inter-stage matching network is a fixed inter-stage matching network or an adjustable inter-stage matching network, and the fixed inter-stage matching network and the adjustable inter-stage matching network respectively comprise at least one stage of LC network with a fundamental wave impedance conversion function; at least one reconfigurable device is also included in the tunable inter-stage matching network, and the reconfigurable device is connected in parallel with a passive device in an LC network of the tunable inter-stage matching network; the bias and control circuit is also connected with a radio frequency switch in the adjustable inter-stage matching network.
The equivalent circuit of the fixed harmonic impedance control network in the tunable wave impedance control network is formed by connecting a capacitor in series with an inductor, or connecting a capacitor in parallel with an inductor, or connecting a capacitor alone, or connecting an inductor alone.
At least two radio frequency signal channels are arranged in the input radio frequency switch; and a plurality of radio frequency signal channels are arranged in the output radio frequency switch.
The tunable wave impedance control network is also provided with a parasitic inductance suppression network for suppressing parasitic inductance, the parasitic inductance suppression network is arranged between a reconfigurable device in the tunable wave impedance control network and the ground, and the parasitic inductance suppression network comprises at least one reconfigurable device.
The following description of the invention will be made with reference to the accompanying drawings:
as in the analysis of the background section, the output matching network of the MB 3G/4G amplifier is typically designed at 1860MHz frequency, and L 1＝0.45nH,L2＝1.84nH,C1＝12.3pF,C2＝3.1pF,Zopt,fung_1860 = 3ohm is calculated. If two passive devices in the secondary resonance network take the value of L f = 0.6nH, then C f = 2.89pF and resonate at the second harmonic of 1860MHz, so that the second harmonic load of 1860MHz can be obtained as low impedance, and Z 2nd_1860 approximately 0.44ohm, if the Q values of the two elements are improved, the second harmonic impedance value can be further reduced. Whereas for the side bands of MB (1710-2025 MHz), including band4 (1710-1755 MHz) and band34 (2010-2025 MHz), on the one hand the fundamental impedance has deviated from the design value of 3ohm, z opt,fund_1710＝2.69-j0.75ohm,Zopt,fund_2025 =3.46+j0.46 ohm; on the other hand, the harmonic loads of band4 and band34 have also deviated from the optimal values required for class F amplifiers, Z 2nd_1710 =1.15-j3.37ohm, and the second harmonic impedance of band34 has also deviated from the required low impedance point, Z 2nd_2025 =0.41+j2.6ohm.
The invention adopts a method of reconstructing the load matching network and the harmonic impedance control network of the power amplifier in real time, so as to reconstruct an amplifier module within the frequency band of the radio frequency signal which is required to be amplified at the time according to the real-time requirement, ensure that the amplifier can work in the F-type power amplifier state when working in any frequency band, and improve the performance. For example, the matching network value of the MB amplifying path in the original module is calculated and designed according to f 0 =1860 MHz, at a certain moment, the control signal from the baseband chip received by the module is known to be about to transmit the band4 (1710-1755 MHz) signal at the next moment to amplify the power, the module adjusts the component value of the matching network in the module, reconstructs the load network, and changes the component value in the matching network from the default value (L1＝0.45nH,L2＝1.84nH,C1＝12.3pF,C2＝3.1pF,Lf＝0.6nH,Cf＝2.89pF) to the optimum component value (L1＝0.49nH,L2＝2nH,C1＝13.35pF,C2＝3.28pF,Lf＝0.6nH,Cf＝3.61pF), required by the band4, so that the load of the MB power amplifier can be reconstructed into the optimum impedance state required by the band4, and Z opt,fund_1710＝3ohm,Z2nd_1710 is about 0.4ohm. Similarly, if the control signal from the baseband chip informs the module that the radio frequency signal from the band34 of the transceiver is about to be received, the module adjusts the component values of the matching network in the module, reconstructs the load network, and changes the load network to the optimal component value (L1＝0.41nH,L2＝1.67nH,C1＝11.3pF,C2＝2.78pF,Lf＝0.6nH,Cf＝2.58pF), required by the band34, so that the load of the MB power amplifier can be reconstructed into the optimal impedance state required by the band34, and Z opt,fund_2025＝3ohm,Z2nd_2025 approximately equal to 0.48ohm.
Fig. 9 is a schematic diagram of an output matching network of a current power amplifier, and after a power amplification stage, a harmonic impedance network, an output matching network, and a radio frequency switch chip are sequentially implemented, typically using a silicon-on-insulator (SOI) process or a pseudo high electron mobility transistor (Pseudomorphic HEMT, pHEMT) process. (the present invention is mainly exemplified by the reconstruction of the load matching structure shown in FIG. 9. Of course, the present invention can also be used in the reconstruction of other load matching structures, such as the structures shown in FIGS. 3, 5, 6 and 7)
Example 1:
Fig. 10 is a schematic diagram of a structure for reconstructing an inductance in the tunable output matching network based on the load matching structure in fig. 9, for the whole bandwidth of MB from 1710MHz to 2025MHz, the variation range of the required capacitance C1 is 11.3pF to 13.3pF, and the minimum value is 11.3pF, then the capacitance C1 in fig. 9 can be reconstructed, that is, as shown in fig. 10, a capacitance C 12 series radio frequency switch S 1 parallel to the capacitance C 11 is reserved and designed on the radio frequency switch chip, And a capacitor C 13 in parallel with the capacitor C 11, the radio frequency switch S 2 being connected in series, the capacitor C 12 and the capacitor C 13 being reconfigured, the capacitor C 11 is implemented by using an SMD capacitor with a fixed value. The connection or disconnection of the two capacitors of the capacitor C 12 and the capacitor C 13 is realized by controlling the connection or disconnection of the switch S 1、S2 in real time through a control signal from the bias and control circuit. Capacitor C 11 =11.3 pF, implemented with SMD capacitors, capacitor C 12 =1 pF and capacitor C 13 =1 pF can be designed on a radio frequency switch chip, implemented with MIM capacitors. Thus, the capacitance value of the capacitor C 1 in fig. 9 is switched between 11.3pF, 12.3pF and 13.3pF in real time, and the capacitor C 1 in the reconfigurable adjustable output matching network is realized.
In other embodiments, according to different frequency and impedance requirements, other capacitances in the output matching network are reconstructed by adopting different numbers and different values of capacitances, and the method is also suitable for reconstructing capacitances in other matching networks, which need to change capacitance values, including capacitances in the input matching network and the interstage matching network.
Example 2;
Fig. 11 is a schematic structural diagram of the reconstruction of the inductance in the tunable output matching network based on the load matching structure in fig. 9, in which the required inductance L 2 varies from 1.67nH to 2.0nH for the entire bandwidth 1710MHz to 2025MHz of MB in fig. 9, the inductance L 2 in fig. 9 can be reconstructed, that is as shown in fig. 11, Inductance L 21 =2nh, using substrate wound inductance or SMD inductance, inductance L 22 =23 nH and inductance L 23 =18 nH using wound or SMD inductance on the radio frequency switch chip: inductance L 22 and inductance L 23 are reconstructed. Wherein the inductance L 22 = 23nH series radio frequency switch S 3, the reconstruction inductance L 23 = 18nH series radio frequency switch S 4, When S 3、S4 is all disconnected, only the inductor L 21 =2nh is connected to the matching network; When the switch is turned on 3 and the switch is turned off 4, a parallel structure of the inductor L 21 and the inductor L 22 is obtained, the equivalent inductance is 1.84nH; When both S 3、S4 are on, inductor L 21, inductor L 22 and inductor L 23 are simultaneously connected to the matching network, equivalent to an inductance of 1.67 nH.
In other embodiments, according to different frequency and impedance requirements, other inductors in the output matching network can be reconstructed by using different numbers and different values of inductors, and the method is also suitable for reconstructing the inductors needing to change the inductance value in other matching networks, including the input matching network and the interstage matching network inductors.
Example 3:
fig. 12 is a schematic diagram of another structure for reconstructing the inductance in the tunable output matching network based on the load matching structure in fig. 9, in which the required inductance L 2 varies from 1.67nH to 2.0nH for the entire bandwidth 1710MHz to 2025MHz of MB in fig. 9, the inductance L 2 in fig. 9 can be reconstructed, that is as shown in fig. 12, inductance L 21 = 1.67nH, capacitance CL 22 = 0.4pF series radio frequency switch S 3, capacitance CL 23 = 0.4pF series radio frequency switch S 4, The capacitor CL 22 and the capacitor CL 23 are reconstructed, the inductor L 21 adopts a substrate winding inductor or an SMD inductor, the capacitor CL 22, capacitor CL 23 employs a MIM capacitor on the rf switch chip. When S 3、S4 is disconnected, only L 21 =1.67 nH is connected to the matching network, and the inductance value is most suitable for the amplifier to work in band34 (2010-2025 MHz); When one of the switches is turned on S 3、S4, the inductor L 21 is then either in parallel with the capacitor CL 22 or the inductor L 21 is in parallel with the capacitor CL 23, The equivalent structure of the whole matching network is shown in fig. 13, wherein R on3 is the on equivalent resistance of the switch S 3, the resistance value of which varies with the size and shape of the radio frequency switch S 3, but the magnitude of the resistance value of which is small, Typical values in conventional SOI processes are P on3 < 0.2ohm, which is negligible. At 1860MHz, the network is equivalent to an inductor with an inductance value of 1.84nH, and in band2 (1850-1910 MHz), the equivalent inductance value of the network changes in a range of 1.836 nH-1.848 nH, and the state is suitable for the amplifier module to work in band2 frequency band; When both S3 and S4 are connected, the inductor L 21, the capacitor CL 22 and the capacitor CL 23 are simultaneously connected into a matching network, the equivalent variation range of the network is 1.975 nH-2.01 nH in the band3/band4 (1710-1785 MHz) frequency band, This condition is known to be suitable for operation of the amplifier module in the band3/band4 band. This scheme achieves the required addition of two MIM capacitors CL 22 and CL 23, with a capacitance of only 0.4pF, using MIM capacitors on the rf switch chip, with little need to increase the area of the chip.
In other embodiments, according to different frequency and impedance requirements, other inductors in the output matching network can be reconstructed by using different numbers and different values of capacitance, and the method is also suitable for reconstructing the inductors in other matching networks, including the input matching network and the interstage matching network, wherein the inductance value of the inductors needs to be changed.
In combination with the ideas of the reconstruction inductance and the reconstruction capacitance, in other embodiments, the aim of reconstructing the values of components in the matching network can be achieved by using only any one of the reconstruction inductance and the reconstruction capacitance, or by comprehensively using a plurality of reconstruction modes.
Example 4:
in order to realize that more frequency bands meet the design principle of the F-class power amplifier, an embodiment of a reconstruction scheme for a harmonic control network is still introduced by taking an MB amplifier as an example. As shown by calculation, in fig. 9, taking the inductance L f =0.6 nH in the secondary resonant network, at three frequency points of 1710MHz, 1860MHz and 2025MHz, the corresponding resonant capacitance C f should be 3.6pF, 3.1pF and 2.6pF respectively, so that the harmonic impedance control network can be correspondingly reconfigured, and the reconstructed structure is shown in fig. 14, The tunable wave impedance control network comprises a fixed harmonic impedance control network and is composed of a capacitor C f2 and a capacitor C f3 (capacitor C f2 and capacitor C f3 are reconstructed), The fixed harmonic impedance control network consists of a fixed capacitor C f1 connected in series with a fixed inductor L f, C f1 is realized by adopting an SMD capacitor or an HBT on-chip capacitor with a fixed capacitance value, L f is formed by adopting a bonding wire or a metal wire on a substrate or an SMD device, or at least two of the three. Capacitor C f2 is connected in series with radio frequency switch S 5, capacitor C f3 is connected in series with radio frequency switch S 6,Cf2、S5、Cf3、S6 implemented on a radio frequency switch die, Or may be implemented on the die on which the amplifier stage is located. In this embodiment, the three capacitors respectively take a value of C f1＝2.6pF,Cf2＝0.5pF,Cf3 =0.5pf. When S 5、S6 is disconnected, C f1 and L f resonate at the second harmonic of 2025MHz, providing more ideal second harmonic low impedance for band 34; when one of S 5、S6 is on and the other is off, the resonant network resonates at the second harmonic of 1860MHz, providing more ideal second harmonic low impedance for band 2; when S 5、S6 is conducted, the harmonic network resonates at the second harmonic of 1710MHz, and provides ideal second harmonic low impedance for band3 and band 4.
With the above scheme for reconstructing the harmonic impedance control network, it is found that there may be a disadvantage when designing the power amplifier module, because parasitic inductance exists in the branches of S 5 and S 6, which are connected in series with C f2 and C f3, on the rf switch chip. The inductance of the inductance L f in the secondary resonant network is small, and when S 5 or S 6 is on, if the parasitic inductance in the C f2 series S 5 network is in a magnitude comparable to the value of L f, The resonance point of the new resonant network after reconstruction tends to deviate from the expected one, and whether or not such deviation occurs, depends on the design of the trace in the specific embodiment. As shown in fig. 15, for the equivalent circuit of the entire network when both S 5 and S 6 are on, it is easy to understand that if the parasitic inductance L X satisfies L x. Gtoreq.lf, The resonance point formed by the secondary resonance network after reconstruction is located at 3.3GHz, and the second harmonic frequency bands (3.42 GHz-3.57 GHz) of band3 and band4 are deviated. For the case where the parasitic inductance L X is relatively large, another embodiment of a reconfigurable harmonic impedance control network is proposed, as shown in fig. 16, C f1 is implemented by using an SMD capacitor with a fixed capacitance value, capacitor C f2 is connected in series with a radio frequency switch S 5, The capacitor C f3 is connected in series with the radio frequency switch S 6.Lf＝0.6nH,Cf1 =3.6pF, and the radio frequency switch S 6.Lf＝0.6nH,Cf1 =3.6pF and the capacitor C f3 are resonant at a second harmonic frequency point of 1710MHz to provide second harmonic low impedance for band3 and band 4; the values of C f2 and C f3 are designed according to the following relationship:
Where f a＝2025MHz,fb =1086 MHz.
When S 5、S6 is disconnected, C f1 and L f resonate at the second harmonic of 1710MHz, and ideal second harmonic low impedance is provided for band3 and band 4; when S 5 is on and S 6 is off, the system adds a new resonant network which resonates at the second harmonic of 2025MHz and provides ideal second harmonic low impedance for band 34; when both S 5、S6 are on, the harmonic network resonates at the second harmonic of 1860MHz, providing a relatively ideal second harmonic low impedance for band 2.
Similarly, in other embodiments, according to different frequencies and impedance requirements, different numbers and different values of reconstruction capacitors can be used to achieve other harmonic impedance control purposes in cooperation with different fixed harmonic impedance networks, and the method is suitable for reconstructing other harmonic impedance control networks.
Since harmonic impedance control is not limited to providing low impedance for a harmonic, the impedance of a harmonic can be far from the low impedance or present high impedance through different structures and different capacitance and inductance values. For example, if the inductor in the fixed harmonic impedance network is omitted, the purpose of presenting high impedance at the third harmonic of a certain fundamental wave or far away from low impedance can be achieved by directly adopting a proper capacitor to be grounded, so that the impedance state required by the class F power amplifier is better achieved. In connection with the method disclosed herein, the tunable wave impedance control network may be comprised of another fixed harmonic impedance control network in combination with a different number of reconfigurable capacitors. The further fixed harmonic impedance control network may comprise only at least one fixed capacitance or only one fixed inductance.
According to the reconstruction concept disclosed above, the MB radio frequency power amplifier works in each band, and the harmonic impedance control network can be reconstructed in real time, so that the performance index of the approximate ideal F-class power amplifier is obtained.
Example 5:
According to the reconstruction concept disclosed above, the tunable wave impedance control network can be reconstructed and resonated in other states, so that the state of third harmonic high impedance is achieved, and the harmonic impedance state required by the IF power amplifier is realized. An embodiment of the relevant reconstruction as a class IF power amplifier, i.e. a saturated power amplifier required for the 2G communication mode, is described below.
The saturated power amplifier required for the 2G communication mode has the following three differences compared to the 3G/4G linear power amplifier: 1. the power amplifier stage is biased to a greater current state; 2. designing a lower Z opt,fund; 3. the third harmonic is short circuited. In order to achieve that a 3G/4G linear power amplifier can operate in a 2G saturated power amplifier mode, a lower Z opt,fund is required, typically up to around 2 ohms. Such as the analysis of the "background art" section,
The reduction of Z opt,fund can obtain a higher P opt,fund, the reduction of Z opt,fund can be achieved by adding a first-stage impedance transformation network, fig. 17 is an embodiment of a power amplifier compatible with three modes of 2G/3G/4G, and fig. 17 can be used simultaneously with fig. 11, 12 and 14. As shown in fig. 17, the device includes an MB input rf switch, an MB input matching network, a first amplifying stage, an MB inter-stage matching network, a second amplifying stage, an output matching network (which is a network of the tunable output matching networks after S 1、S2、S3 and S 4 are disconnected), a standby impedance conversion network, a tunable wave impedance control network, a bias and control circuit, and an MB SP6T output rf switch.
The tunable wave impedance control network comprises a fixed harmonic impedance control network, a reconfigurable inductor L f2 and two reconfigurable capacitors C f2 and C f3, wherein the fixed harmonic impedance control network is formed by connecting a capacitor C f1 in series with an inductor L f1, a capacitor C f2 is connected in series with a radio frequency switch S 1, a capacitor C f3 is connected in series with a radio frequency switch S 2, and an inductor L f2 is connected in series with a radio frequency switch S 3. In other embodiments, the adjustable impedance control network may be composed of a different number of fixed harmonic impedance control networks and a different number of reconfigurable capacitors and a different number of reconfigurable inductors, where the fixed harmonic impedance control networks may also be composed of harmonic impedance control networks with different structures.
The standby impedance transformation network is composed of a DC blocking capacitor C 3 and a reconfigurable inductor connected in parallel to the ground. The reconfigurable inductor is a radio frequency switch S 7 connected in series with an inductor L 3.
In the embodiment of fig. 17, the device values in the output matching network are set to L 1＝0.45nH,L2＝1.84nH,C1＝12.3pF,C2 = 3.1pF, the setting of the device values is designed according to the working mode and the frequency band of the power amplifier, the values can also be realized by adopting fixed passive devices in parallel connection with reconfigurable devices, and the reconfiguration structures in some matching networks, including adjustable output matching networks, are omitted here. When the power amplifier needs to operate in the 3G/4G mode band2 band, the fundamental wave (1850-1910 MHz) load impedance Z opt,fund needs to be designed to be about 3 ohms, and the bias and control circuit controls the rf switch S 2、S3、S4 to be turned off (see fig. 11, 12 and 14), and the fundamental wave load impedance network can provide the band2 fundamental wave impedance for the second amplifying stageThe tunable wave impedance control network resonates at the second harmonic of band 2. When working in other 3G/4G mode MB frequency bands, the aim of complete coverage can be achieved through the structure shown in FIG. 14.
When the power amplifier needs to operate in 2G GSM HB (GSM 1800:1710-1785 MHz, GSM 1900:1850-1910 MHz), its fundamental (1710-1910 MHz) load impedance Z opt,fund needs to be reduced to 2 ohms, while the tunable wave impedance control network resonance is needed at the third harmonic. The radio frequency switch S 1、S2 is turned off and S 3、S4 is turned on (see fig. 11, 12 and 14) by the bias and control circuit, and the fundamental load impedance network provides the fundamental impedance for the second amplifying stageThe tunable wave impedance control network resonates at the third harmonic. Correspondingly, a radio frequency channel is added on the output radio frequency switch chip and is connected to a 2G GSM HB radio frequency output pin.
It is clear to those skilled in the art that, according to the design method disclosed in the present invention, in other embodiments, the number of reconfigurable capacitors and reconfigurable inductors in the tunable wave impedance control network, the capacitance value and the inductance value can be further increased or decreased, so as to achieve the purpose of covering other target frequency bands.
In other embodiments, the value of L 3 in the standby impedance transformation network may be changed to achieve the goal of obtaining other optimal fundamental loads in this or other frequency bands after the standby impedance transformation network is enabled.
In other embodiments, the purpose of covering other frequency bands or matching to other target impedance values is achieved by increasing or adjusting the number of standby impedance conversion networks.
In other real-time examples, other components may be used to form the standby impedance transformation network, for example, the reconfigurable inductor connected in parallel to the ground may be replaced by a reconfigurable capacitor, or a network in which the reconfigurable capacitor is connected in series with the inductor, or a network in which the reconfigurable inductor is connected in series with the capacitor.
According to the design concept disclosed by the invention, in other embodiments, the 3G/4G LB power amplifier path can be designed as a reconfigurable power amplifier so as to achieve the purpose of meeting the design principle of the F-class power amplifier on each different 3G/4G LB frequency band in real time, and the 3G/4G LB power amplifier can also work in a saturated power amplifier state and is compatible with a 2G LB GSM mode, so that one LB GSM power amplifier is omitted for the radio frequency front end proposal, and the complexity and cost of the radio frequency front end are reduced.
According to the design concept disclosed in the present invention, in other embodiments, the 3G/4G/5G power amplifier may be designed as a reconfigurable power amplifier, so as to be compatible with other radio frequency power amplifiers such as WLAN power amplifier, bluetooth power amplifier, zigbee power amplifier, NB-IoT power amplifier, etc., so as to omit the corresponding power amplifier channel, save cost and save complexity of the radio frequency front end, as shown in fig. 18, RFin is an input of the power amplifier in any operation mode and frequency band, RFout is an output of the power amplifier in any operation mode and frequency band, and has versatility.
The foregoing description of the preferred embodiments of the invention is not intended to limit the invention to the precise form disclosed, and any such modifications, equivalents, and alternatives falling within the spirit and scope of the invention are intended to be included within the scope of the invention.
Claims (5)
1. A reconfigurable radio frequency and microwave power amplifier, characterized by: the adjustable output matching network comprises an input radio frequency switch, an adjustable input matching network, a power amplification stage, an adjustable output matching network, an output radio frequency switch, a tunable wave impedance control network and a bias and control circuit which are sequentially connected, wherein the tunable wave impedance control network is connected between the common end of the power amplification stage and the adjustable output matching network and the ground;
The adjustable input matching network and the adjustable output matching network respectively comprise at least one stage of LC network with a fundamental wave impedance conversion function, the LC network comprises at least one reconfigurable device, and the reconfigurable device is connected with a passive device in the LC network in parallel;
The tunable wave impedance control network comprises a fixed harmonic impedance control network and at least one reconfigurable device, the reconfigurable device being connected in parallel with a passive device in the fixed harmonic impedance control network;
the reconfigurable device is a reconfigurable capacitor or a reconfigurable inductor, the reconfigurable capacitor is formed by connecting a radio frequency switch and a capacitor in series, and the reconfigurable inductor is formed by connecting the radio frequency switch and the inductor in series;
The bias and control circuit is respectively connected with the input radio frequency switch, the power amplification stage, the output radio frequency switch, the radio frequency switch in the adjustable input matching network, the radio frequency switch in the adjustable output matching network and the radio frequency switch in the tunable wave impedance control network;
The power amplifier further comprises at least one standby impedance conversion network for providing increased or decreased fundamental load impedance to the power amplifier stage in cooperation with the adjustable output matching network, the standby impedance conversion network being connected between the adjustable output matching network and an output radio frequency switch;
The standby impedance conversion network comprises a direct current blocking capacitor connected between the adjustable output matching network and the output radio frequency switch, and a reconfigurable impedance conversion network, one end of which is connected between the direct current blocking capacitor and the output radio frequency switch, and the other end of which is grounded;
The power amplification stage comprises at least one stage of amplifying stage of transistor, and when the power amplification stage comprises more than two stages of amplifying stages, an interstage matching network is arranged between two adjacent stages of amplifying stages;
The tunable wave impedance control network is also provided with a parasitic inductance suppression network for suppressing parasitic inductance, the parasitic inductance suppression network is connected between a reconfigurable device in the tunable wave impedance control network and ground, and the parasitic inductance suppression network comprises at least one reconfigurable device.
2. A reconfigurable radio frequency and microwave power amplifier according to claim 1, wherein: the reconfigurable impedance conversion network is specifically a reconfigurable inductor, a reconfigurable capacitor, a series inductor and a reconfigurable capacitor, or a series capacitor and a reconfigurable inductor;
The bias and control circuit is also connected with a radio frequency switch in the reconfigurable impedance transformation network.
3. A reconfigurable radio frequency and microwave power amplifier according to claim 1, wherein: the inter-stage matching network is a fixed inter-stage matching network or an adjustable inter-stage matching network, and the fixed inter-stage matching network and the adjustable inter-stage matching network respectively comprise at least one stage of LC network with a fundamental wave impedance conversion function;
At least one reconfigurable device is also included in the tunable inter-stage matching network, and the reconfigurable device is connected in parallel with a passive device in an LC network of the tunable inter-stage matching network;
The bias and control circuit is also connected with a radio frequency switch in the adjustable inter-stage matching network.
4. A reconfigurable radio frequency and microwave power amplifier according to claim 1 or 2, characterized in that: the equivalent circuit of the fixed harmonic impedance control network in the tunable wave impedance control network is formed by connecting a capacitor in series with an inductor, or connecting a capacitor in parallel with an inductor, or connecting a capacitor alone, or connecting an inductor alone.
5. A reconfigurable radio frequency and microwave power amplifier according to claim 1 or 2, characterized in that: at least two radio frequency signal channels are arranged in the input radio frequency switch; and a plurality of radio frequency signal channels are arranged in the output radio frequency switch.
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| CN218183313U (en) * | 2022-09-05 | 2022-12-30 | 深圳飞骧科技股份有限公司 | Power amplifier with reconfigurable matching network and communication equipment |
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| US9673766B1 (en) * | 2016-05-18 | 2017-06-06 | Nxp Usa, Inc. | Class F amplifiers using resonant circuits in an output matching network |
| CN207588812U (en) * | 2017-09-08 | 2018-07-06 | 牛旭 | A kind of restructural radio frequency and microwave power amplifier |
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