CN103401456A

CN103401456A - Dual-modulating wave dual-carrier modulation method for voltage type three-level neutral point clamped converter

Info

Publication number: CN103401456A
Application number: CN2013102873313A
Authority: CN
Inventors: 李宁; 王跃; 王兆安
Original assignee: Xian Jiaotong University
Current assignee: Xian Jiaotong University
Priority date: 2013-07-09
Filing date: 2013-07-09
Publication date: 2013-11-20
Anticipated expiration: 2033-07-09
Also published as: CN103401456B

Abstract

The invention discloses a dual-modulation wave dual-carrier modulation method for a voltage-type three-level neutral point clamp converter, which comprises the following steps: (1) obtaining a three-phase modulation wave according to the carrier PWM modulation method; (2) converting each phase The modulation wave U _x is decomposed into the upper modulation wave U _xp and the lower modulation wave U _xn ; (3) After comparing the upper modulation wave U _xp and the lower modulation wave U _xn with the carrier, output the final PWM pulse to the voltage type three-level medium point clamp converter. According to the principle that the average midpoint current in each carrier period is 0, the present invention solves the up-modulation wave U _xp and down-modulation wave U _xn without affecting the DC capacitor voltage balance.

Description

Double modulation wave and double carrier modulation method for voltage-type three-level neutral-point clamped converter

【技术领域】【Technical field】

本发明涉及电力电子技术领域，特别涉及一种电压型三电平中点钳位变流器调制方法。The invention relates to the technical field of power electronics, in particular to a modulation method for a voltage-type three-level midpoint clamp converter.

【背景技术】【Background technique】

近年来，三电平中点箝位逆变器在中高压领域的应用越来越广泛，技术也越来越成熟，但是，在其固有的中点电位平衡问题方面仍然有待完善。In recent years, the application of three-level neutral-point clamped inverters in the medium and high voltage fields has become more and more extensive, and the technology has become more and more mature. However, its inherent neutral-point potential balance problem still needs to be improved.

三电平逆变器的调制方法有载波PWM法和空间矢量PWM(space vectorPWM，SVPWM)法。The modulation methods of the three-level inverter include the carrier PWM method and the space vector PWM (space vector PWM, SVPWM) method.

SVPWM由于电压利用率高、矢量选择灵活，受到了广泛的欢迎，但这种方法算法复杂、计算量大。相反，基本的载波PWM却具有算法简单、易于实现的优点，然而，由于三电平NPC逆变器需要控制中点电位平衡，且普遍采用零序电压注入的方法，这又给载波PWM带来了以下不足：1）所需注入零序电压的准确值计算非常复杂，这抵消了基本载波PWM算法简单的优点；2）在部分调制度和功率因数下，中点电位低频波动仍然无法完全消除，这不仅需要增大直流节点电容，还会增加逆变输出的低频谐波。SVPWM has been widely welcomed because of its high voltage utilization rate and flexible vector selection, but this method has complex algorithms and a large amount of calculation. On the contrary, the basic carrier PWM has the advantages of simple algorithm and easy implementation. However, since the three-level NPC inverter needs to control the neutral point potential balance, and the method of zero-sequence voltage injection is generally used, this brings the carrier PWM The following disadvantages are overcome: 1) The calculation of the exact value of the required injected zero-sequence voltage is very complicated, which offsets the simple advantages of the basic carrier PWM algorithm; 2) Under partial modulation and power factor, the low-frequency fluctuation of the midpoint potential still cannot be completely eliminated , which not only needs to increase the DC node capacitance, but also increases the low-frequency harmonics of the inverter output.

实际上，通过在调制波中注入适当的零序电压，载波PWM也可以获得和SVPWM同样高的电压利用率。如果能够进一步克服载波PWM在上述中点电位平衡控制方面的不足，必将促进其在三电平逆变器中的应用，并进一步提高逆变器的性能。In fact, by injecting an appropriate zero-sequence voltage into the modulated wave, carrier PWM can also obtain the same high voltage utilization rate as SVPWM. If the shortcomings of carrier PWM in the above-mentioned midpoint potential balance control can be further overcome, its application in three-level inverters will be promoted, and the performance of the inverter will be further improved.

【发明内容】【Content of invention】

本发明的目的在于提供一种全调制度、全数字化、不对直流电容电压平衡造成影响的电压型三电平中点钳位变流器双调制波双载波调制方法。The purpose of the present invention is to provide a dual-modulation-wave and dual-carrier modulation method for a voltage-type three-level neutral-point clamp converter with full modulation degree, full digitization, and no influence on the balance of DC capacitor voltage.

为了实现上述目的，本发明采用的技术方案是：In order to achieve the above object, the technical scheme adopted in the present invention is:

电压型三电平中点钳位变流器双调制波双载波调制方法，包括以下步骤：The dual modulation wave dual carrier modulation method for a voltage-type three-level neutral point clamping converter comprises the following steps:

（1）根据载波PWM调制方法，得到三相调制波：(1) According to the carrier PWM modulation method, the three-phase modulation wave is obtained:

$\{\begin{matrix} {U u}_{a a} = = M m sin sin ((ωt ωt)) \\ {U u}_{b b} = = M m sin sin ((ωt ωt - - 22 π π / / 33)) \\ {U u}_{c c} = = M m sin sin ((ωt ωt + + 22 π π / / 33)) \end{matrix} - - - - - - ((11))$

其中M为调制度，ω为基波角频率；Where M is the degree of modulation, ω is the fundamental angular frequency;

（2）将各相调制波U_x分解为上调制波U_xp和下调制波U_xn；上调制波U_xp和下调制波U_xn的公式如下：(2) Decompose the modulation wave U _x of each phase into the upper modulation wave U _xp and the lower modulation wave U _xn ; the formulas of the upper modulation wave U _xp and the lower modulation wave U _xn are as follows:

$\{\begin{matrix} {U u}_{xp xp} = = \frac{{U u}_{x x} + + {U u}_{z z} + + 11 - - k k}{22} ((00 \leq \leq {U u}_{xp xp} \leq \leq 11)) \\ {U u}_{xn xn} = = \frac{{U u}_{x x} + + {U u}_{z z} - - 11 + + k k}{22} ((- - 11 \leq \leq {U u}_{xn xn} \leq \leq 00)) \end{matrix} ((x x = = a a,, b b,, c c)) - - - - - - ((22))$

公式（2）中k=k₁或k₂；当k=k₁时，U_Z=U_Z1;当k=k₂时，U_Z=U_Z2;In formula (2), k=k ₁ or k ₂ ; when k=k ₁ , U _Z =U _Z1 ; when k=k ₂ , U _Z =U _Z2 ;

（3）将k带入式（2）中得到每相的上调制波U_xp和下调制波U_xn，将上调制波U_xp、下调制波U_xn与载波进行比较后输出最终的PWM脉冲给电压型三电平中点钳位变流器。(3) Put k into formula (2) to get the up-modulation wave U _xp and down-modulation wave U _xn of each phase, compare the up-modulation wave U _xp , down-modulation wave U _xn with the carrier and output the final PWM pulse It is a voltage-type three-level neutral point clamping converter.

相对于现有技术，本发明的有益效果是：本发明提供了一种全调制度、全数字化、不对直流电容电压平衡造成影响的电压型三电平载波调制策略，该方法实现简单，应用效果好。Compared with the prior art, the beneficial effects of the present invention are: the present invention provides a voltage-type three-level carrier modulation strategy with full modulation degree, full digitalization, and no influence on the DC capacitor voltage balance. The method is simple to implement and the application effect is good.

【附图说明】【Description of drawings】

图1是三电平二极管钳位型变流器主电路拓扑图；Figure 1 is a topological diagram of the main circuit of a three-level diode-clamped converter;

图2是传统三电平二极管钳位型变流器载波调制策略的三相调制波（M=1）；Figure 2 is the three-phase modulation wave (M=1) of the traditional three-level diode-clamped converter carrier modulation strategy;

图3是k=k₁，U_z=U_z1时本发明三电平二极管钳位型变流器双调制波双载波调制方法的三相调制波（M=1）；Fig. 3 is the three-phase modulation wave (M=1) of the dual modulation wave dual carrier modulation method of the three-level diode-clamped converter of the present invention when k=k ₁ and U _z =U _z1 ;

图4是k=k₂，U_z=U_z2时本发明三电平二极管钳位型变流器双调制波双载波调制方法的三相调制波（M=1）。Fig. 4 is the three-phase modulation wave (M=1) of the dual modulation wave dual carrier modulation method of the three-level diode-clamped converter of the present invention when k=k ₂ and U _z =U _z2 .

【具体实施方式】【Detailed ways】

下面结合附图对本发明作进一步说明。The present invention will be further described below in conjunction with accompanying drawing.

如图1所示给出了本发明中二极管钳位型三电平变流器的原理图，包括三相交流部分（若为三电平逆变器结构，则三相交流部分为负载；若为三电平整流器、静止无功发生器等装置，则三相交流部分为交流源加三相交流平波电抗器）、三电平直流侧外接部分（若为三电平逆变器结构，则直流侧外接部分为直流电压源，该直流源可为实际电源，也可为通过交流电源整流得到的直流源；若为三电平整流器结构，则直流侧外接部分为负载；若为三电平静止无功发生器，则直流侧无外接部分）、三电平NPC变流器主电路部分、电压传感器、电流传感器、AD转换芯片和数字处理器，其中，电压传感器检测三相交流部分电压和直流侧各电容电压、电流传感器检测交流侧各相电流，电压传感器和电流传感器通过AD转换芯片与数字处理器连接，数字处理器与五通过相应的驱动电路控制三电平变流器中各功率器件的开关。As shown in Figure 1, the principle diagram of the diode-clamped three-level converter in the present invention is provided, including the three-phase AC part (if it is a three-level inverter structure, the three-phase AC part is the load; if For devices such as three-level rectifiers and static var generators, the three-phase AC part is an AC source plus a three-phase AC smoothing reactor), and the external part of the three-level DC side (if it is a three-level inverter structure, Then the external part of the DC side is a DC voltage source, which can be an actual power supply or a DC source obtained by rectifying an AC power supply; if it is a three-level rectifier structure, the external part of the DC side is a load; if it is a three-voltage static var generator, there is no external part on the DC side), three-level NPC converter main circuit part, voltage sensor, current sensor, AD conversion chip and digital processor, in which the voltage sensor detects the voltage of the three-phase AC part The capacitor voltage and current sensors on the DC side detect the phase currents on the AC side. The voltage sensors and current sensors are connected to the digital processor through the AD conversion chip. Switches for power devices.

本发明电压型三电平中点钳位变流器双调制波双载波调制方法的具体步骤如下：The specific steps of the dual modulation wave dual carrier modulation method of the voltage-type three-level neutral point clamp converter of the present invention are as follows:

(1)根据载波PWM调制方法，得到相应的三相调制波,其中M为调制度，ω为基波角频率；(1) According to the carrier PWM modulation method, the corresponding three-phase modulation wave is obtained, where M is the degree of modulation, and ω is the angular frequency of the fundamental wave;

（2）将各相调制波U_x（x=a,b,c）分解为上调制波U_xp和下调制波U_xn，根据每个载波周期中的平均中点电流为0的原则，求解上调制波U_xp和下调制波U_xn（其中k为M和ωt的函数，U_z为三相调制波中叠加的零序分量）；(2) Decompose the modulation wave U _x (x=a,b,c) of each phase into the upper modulation wave U _xp and the lower modulation wave U _xn , and according to the principle that the average midpoint current in each carrier cycle is 0, solve Up modulation wave U _xp and down modulation wave U _xn (where k is a function of M and ωt, U _z is the superimposed zero-sequence component in the three-phase modulation wave);

本发明中给出其具体求解过程：Provide its specific solution process in the present invention:

在传统的三相调制波中叠加零序分量U_z可以提高整个三电平系统的电压利用率，新的调制波为：Superimposing the zero-sequence component U _z in the traditional three-phase modulation wave can improve the voltage utilization of the whole three-level system, and the new modulation wave is:

U_x'＝U_x+U_z(x＝a,b,c) （3）U _x '＝U _x +U _z (x＝a,b,c) (3)

根据上调制波和下调制波的定义：According to the definition of up-modulation wave and down-modulation wave:

U_xp+U_xn＝U_x+U_z(x＝a,b,c) （4）U _xp +U _xn ＝U _x +U _z (x＝a,b,c) (4)

定义d_xo为一个载波周期中x相连接到直流侧电容中点O的占空比，则有：Define d _xo as the duty cycle of phase x connected to the midpoint O of the DC side capacitor in a carrier cycle, then:

d_xo＝1+U_xn-U_xp （5）d _xo ＝1+U _xn -U _xp (5)

欲使载波调制策略在每个开关周期中对直流电容电压均无影响，应保证：To make the carrier modulation strategy have no effect on the DC capacitor voltage in each switching cycle, it should be ensured that:

Σ(1+U_xn-U_xp)·i_x≡0 （6）Σ(1+U _xn -U _xp )·i _x ≡0 (6)

由于i_x与U_x之间的相位差不固定，一个比较实用的解为：Since the phase difference between i _x and U _x is not fixed, a more practical solution is:

1+U_an-U_ap＝1+U_bn-U_bp＝1+U_cn-U_cp （7）1+U _an -U _ap ＝1+U _bn -U _bp ＝1+U _cn -U _cp (7)

且需满足：And need to meet:

$\{\begin{matrix} {U u}_{xp xp max max} = = {- - U u}_{xn xn min min} \\ {U u}_{xp xp min min} = = {U u}_{xn xn max max} = = 00 \end{matrix} ((x x = = a a,, b b,, c c)) - - - - - - ((88))$

令：make:

1+U_xn-U_xp＝k （9）1+U _xn -U _xp = k (9)

有：have:

$\{\begin{matrix} {U u}_{xp xp} = = \frac{{U u}_{x x} + + {U u}_{z z} + + 11 - - k k}{22} \\ {U u}_{xn xn} = = \frac{{U u}_{x x} + + {U u}_{z z} - - 11 + + k k}{22} \end{matrix} - - - - - - ((1010))$

根据减小调制算法开关损耗的原则计算k值和U_z Calculate the k value and U _z according to the principle of reducing the switching loss of the modulation algorithm

式（10）中有k和U_z两个变量,为了减小系统的开关损耗,可以使一段时间内U_xp为最值（0或1），U_xn为最值（0或-1）,解得:There are two variables k and U _z in formula (10). In order to reduce the switching loss of the system, it is possible to make U _xp the maximum value (0 or 1) and U _xn the maximum value (0 or -1) for a period of time. Solutions have to:

$\{\begin{matrix} {k k}_{xp xp ((11))} = = {U u}_{x x} + + {U u}_{z z} - - 11 \\ {k k}_{xp xp ((00))} = = {U u}_{x x} + + {U u}_{z z} + + 11 \\ {k k}_{xn xn ((00))} = = {- - U u}_{x x} - - {U u}_{z z} + + 11 \\ {k k}_{xn xn ((- - 11))} = = {- - U u}_{x x} - - {U u}_{z z} - - 11 \end{matrix} ((x x = = a a,, b b,, c c)) - - - - - - ((1111))$

上式中的k_xy(j),x=a,b,c,y=p,n,j=1,0,-1,表示使得x（a,b,c）相的y(p,n)调制波为j（1,0,-1）值时对应的k解。k _xy(j) in the above formula, x=a,b,c,y=p,n,j=1,0,-1, means that x (a,b,c) phase y(p,n ) is the corresponding k solution when the modulating wave is j (1,0,-1).

上式中存在如下约束条件：There are the following constraints in the above formula:

$\{\begin{matrix} {k k}_{xp xp ((11))} = = {- - k k}_{xn xn ((00))},, {k k}_{xp xp ((00))} = = {- - k k}_{xn xn ((- - 11))},, ((x x = = a a,, b b,, c c)) \\ {k k}_{xz xz ((j j))} &NotEqual; &NotEqual; {k k}_{yz yz ((j j))},, ((x x,, y the y = = a a,, b b,, c c,, x x &NotEqual; &NotEqual; y the y,, z z = = p p,, n no,, j j = = 1,0 1,0,, - - 11)) \\ {k k}_{xp xp ((11))} &NotEqual; &NotEqual; {k k}_{xp xp ((00))},, {k k}_{xn xn ((00))} &NotEqual; &NotEqual; {k k}_{xn xn ((- - 11))},, ((x x = = a a,, b b,, c c)) \end{matrix} - - - - - - ((1212))$

由于以上的式子中存在U_z这个未知数，可以联立两个k_xy(j)求解U_z：Since the unknown U _z exists in the above formula, two k _xy(j) can be combined to solve U _z :

${U u}_{zl zl} = = \{\begin{matrix} 11 - - {U u}_{x x},, {k k}_{xp xp ((11))} = = {k k}_{xn xn ((00))} \\ {- - U u}_{x x},, {k k}_{xp xp ((11))} = = {k k}_{xn xn ((- - 11))} {ork ork}_{xp xp ((00))} = = {k k}_{xn xn ((00))} \\ - - 11 - - {U u}_{x x},, {k k}_{xp xp ((00))} = = {k k}_{xn xn ((- - 11))} \end{matrix} ((x x = = a a,, b b,, c c)) - - - - - - ((1313))$

${U u}_{z z 22} = = \{\begin{matrix} 11 - - \frac{{U u}_{x x} + + {U u}_{y the y}}{22},, (({k k}_{xp xp} = = {k k}_{yn yn ((00))} {ork ork}_{xn xn ((00))} = = {k k}_{yp yp ((11))})) \\ - - \frac{{U u}_{x x} + + {U u}_{y the y}}{22},, (\begin{matrix} {k k}_{xp xp ((11))} = = {k k}_{yn yn ((- - 11))} {ork ork}_{xp xp ((00))} = = {k k}_{yn yn ((00))} \\ {ork ork}_{xn xn ((00))} = = {k k}_{yp yp ((00))} {ork ork}_{xn xn ((- - 11))} = = {k k}_{yp yp ((11))} \end{matrix}) \\ - - 11 - - \frac{{U u}_{x x} + + {U u}_{y the y}}{22},, (({k k}_{xp xp ((00))} = = {k k}_{yn yn ((- - 11))} {ork ork}_{xn xn ((- - 11))} = = {k k}_{yp yp ((00))})) \end{matrix} ((x x,, y the y = = a a,, b b,, c c,, x x &NotEqual; &NotEqual; y the y)) - - - - - - ((1414))$

U_z和k值需满足：U _z and k values need to meet:

$\{\begin{matrix} 00 \leq \leq {U u}_{xp xp} = = \frac{{U u}_{x x} + + {U u}_{z z} + + 11 - - k k}{22} \leq \leq 11 \\ - - 11 \leq \leq {U u}_{xn xn} = = \frac{{U u}_{x x} + + {U u}_{z z} - - 11 + + k k}{22} \leq \leq 00 \end{matrix} - - - - - - ((1515))$

满足上式的U_z和k的解及其对应的ωt的范围为：The solutions of U _z and k satisfying the above formula and their corresponding ranges of ωt are:

上式中的k₁/U_z1和k₂/U_z2这两组解会使得d_xo恒为0，这个结果最终会使得三电平调制退化为两电平调制，需舍弃，故实际中只存在如下的解：The two sets of k ₁ /U _z1 and k ₂ /U _z2 solutions in the above formula will make d _xo always 0, and this result will eventually degenerate the three-level modulation into two-level modulation, which needs to be discarded, so in practice only The following solutions exist:

$\{\begin{matrix} k k = = &PlusMinus; &PlusMinus; [[11 + + (({U u}_{b b} - - {U u}_{c c})) / / 22]],, {U u}_{z z} = = - - (({U u}_{b b} + + {U u}_{c c})) / / 22,, ωt ωt &Element; &Element; [[00,, π π / / 66]] \cup \cup [[1111 π π / / 6,2 6,2 π π]] \\ k k = = &PlusMinus; &PlusMinus; [[11 + + (({U u}_{b b} - - {U u}_{a a})) / / 22]],, {U u}_{z z} = = - - (({U u}_{a a} + + {U u}_{b b})) / / 22,, ωt ωt &Element; &Element; [[π π / / 66,, π π / / 22]] \\ k k = = &PlusMinus; &PlusMinus; [[11 + + (({U u}_{c c} - - {U u}_{a a})) / / 22]],, {U u}_{z z} = = - - (({U u}_{c c} + + {U u}_{a a})) / / 22,, ωt ωt &Element; &Element; [[π π / / 2,5 2,5 π π / / 66]] \\ k k = = &PlusMinus; &PlusMinus; [[11 + + (({U u}_{c c} - - {U u}_{b b})) / / 22]],, {U u}_{z z} = = - - (({U u}_{b b} + + {U u}_{c c})) / / 22,, ωt ωt &Element; &Element; [[55 π π / / 66,, 77 π π / / 66,,]] \\ k k = = &PlusMinus; &PlusMinus; [[11 + + (({U u}_{a a} - - {U u}_{b b})) / / 22]],, {U u}_{z z} = = - - (({U u}_{a a} + + {U u}_{b b})) / / 22,, ωt ωt &Element; &Element; [[77 π π / / 6,3 6,3 π π / / 22]] \\ k k = = &PlusMinus; &PlusMinus; [[11 + + (({U u}_{a a} - - {U u}_{c c})) / / 22]],, {U u}_{z z} = = - - (({U u}_{c c} + + {U u}_{a a})) / / 22,, ωt ωt &Element; &Element; [[33 π π / / 2,11 2,11 π π / / 66]] \end{matrix} - - - - - - ((1717))$

上式中的k和U_z为分段函数，k在每个π/3区域中有两个解，U_z在每个π/3区域中有一个解，共有2⁶=64个解，其中的绝大多数解都使得U_xp和U_xn不连续，当U_xp和U_xn连续时有：The k and U _z in the above formula are piecewise functions, k has two solutions in each π/3 area, U _z has one solution in each π/3 area, and there are 2 ⁶ =64 solutions in total, where Most of the solutions make U _xp and U _xn discontinuous, when U _xp and U _xn are continuous:

$\{\begin{matrix} {U u}_{xy xy ((\frac{π π}{66} - -))} = = {U u}_{xy xy ((\frac{π π}{66} + +))} \\ {U u}_{xy xy ((\frac{π π}{22} - -))} = = {U u}_{xy xy ((\frac{π π}{22} + +))} \\ {U u}_{xy xy ((\frac{55 π π}{66} - -))} = = {U u}_{xy xy ((\frac{55 π π}{66} + +))} \\ {U u}_{xy xy ((\frac{77 π π}{66} - -))} = = {U u}_{xy xy ((\frac{77 π π}{66} + +))} \\ {U u}_{xy xy ((\frac{33 π π}{22} - -))} = = {U u}_{xy xy ((\frac{33 π π}{22} + +))} \\ {U u}_{xy xy ((\frac{1111 π π}{66} - -))} = = {U u}_{xy xy ((\frac{1111 π π}{66} + +))} \end{matrix} ((x x = = a a,, b b,, c c,, y the y = = p p,, n no)) - - - - - - ((1818))$

根据上式，解得使x相的上调制波U_xp（下调制波U_xn）连续的解为：According to the above formula, the solution to make the up-modulation wave U _xp (down-modulation wave U _xn ) of phase x continuous is:

（3）将k带入式（2）中得到新型调制策略每相的上下调制波U_xp和U_xn，将上下调制波U_xp和U_xn与载波进行比较后输出最终的PWM脉冲给电压型三电平中点钳位变流器。(3) Bring k into formula (2) to get the up and down modulation waves U _xp and U _xn of each phase of the new modulation strategy, compare the up and down modulation waves U _xp and U _xn with the carrier and output the final PWM pulse to the voltage type Three-level neutral point clamped converter.

图3和图4分别为k和Uz分别为k₁、U_z1和k₂、U_z2时的三相上下调制波U_xp和U_xn，相较于传统的单相单调波策略，本发明提出的新型调制策略增加自由度k和U_z，通过k和U_z的合理取值就可以使得调制策略对中点电压的平衡无影响，k和U_z的推导过程较为复杂，但结果十分简单，实现简便易行，同时不受调制度和负载的影响。但需要说明的是，本发明提出的调策略会增加功率器件的开关损耗，与传统的载波SPWM策略相比，增加了1/3倍的器件开关损耗。Figure 3 and Figure 4 show the three-phase up-down modulation waves U _xp and U _xn when k and Uz are k ₁ , U _z1 and k ₂ , and U _z2 respectively. Compared with the traditional single-phase monotonic wave strategy, the present invention proposes The new modulation strategy of the new modulation strategy increases the degrees of freedom k and U _z , and the modulation strategy has no effect on the balance of the midpoint voltage through reasonable values of k and U _z . The derivation process of k and U _z is relatively complicated, but the result is very simple. The implementation is simple and easy, and it is not affected by the degree of modulation and load. However, it should be noted that the modulation strategy proposed by the present invention will increase the switching loss of the power device, and compared with the traditional carrier SPWM strategy, it will increase the switching loss of the device by 1/3 times.

Claims

1. the two carrier modulating methods of voltage-type three level neutral-point-clamped current transformer double modulation ripples, is characterized in that, comprises the following steps:

(1), according to the Carrier-based PWM modulator approach, obtain the three-phase modulations ripple:

\{\begin{matrix} U_{a} = M \sin (ωt) \\ U_{b} = M \sin (ωt - 2 π / 3) \\ U_{c} = M \sin (ωt + 2 π / 3) \end{matrix} - - - (1)

Wherein M is modulation degree, and ω is the first-harmonic angular frequency;

(2) with each phase modulating wave U _xBe decomposed into modulating wave U _xpWith modulated ripple U _xnUpper modulating wave U _xpWith modulated ripple U _xnFormula as follows:

\{\begin{matrix} U_{xp} = \frac{U_{x} + U_{z} + 1 - k}{2} (0 \leq U_{xp} \leq 1) \\ U_{xn} = \frac{U_{x} + U_{z} - 1 + k}{2} (- 1 \leq U_{xn} \leq 0) \end{matrix} (x = a, b, c) - - - (2)

K=k in formula (2) ₁Or k ₂Work as k=k ₁The time, U _Z=U _Z1; Work as k=k ₂The time, U _Z=U _Z2;

(3) k is brought in formula (2) into the upper modulating wave U that obtains every phase _xpWith modulated ripple U _xn, with upper modulating wave U _xp, modulated ripple U _xnCompare the final pwm pulse of rear output to voltage-type three level neutral-point-clamped current transformers with carrier wave.