CN102664859A - Synchronization and channel estimation scheme for multi-band orthogonal frequency division multiplexing (OFDM) ultra wideband receiver - Google Patents

Abstract

The invention discloses an integral synchronization and channel estimation scheme for a multi-band orthogonal frequency division multiplexing (OFDM) ultra wideband receiver, and is applied to a high-speed short-distance ultra wideband wireless communication background. A baseband hopping multi-band implementation scheme is adopted; a self-correlation and energy detection-based symbol timing method is disclosed; the influence of single-carrier frequency offset under a baseband hopping mechanism on a system is equivalently consistent with that under radio frequency hopping, so that an improved maximum likelihood estimation method is disclosed; frequency-domain phase deviation caused by sampling frequency offset is compensated on the basis of phase equalization; and a channel is compensated on the basis of overlap-add and frequency-domain equalization, and residue symbol timing deviation, carrier phase deviation, amplitude attenuation caused by carrier frequency offset, and amplitude attenuation and integer sample deviation, which are caused by the sampling frequency offset, are also compensated. By the scheme, the tolerance of a guard interval is fully utilized, the system has high error performance, and the overall complexity of an algorithm is lowered.

Description

Synchronous and the channel estimation scheme of multi-band OFDM ultra wideband receiver

Technical field

Synchronous and channel estimation method is the key component of decision systems performance in the wireless communication receiver; The present invention propose in the multi-band OFDM ultra wideband receiver complete synchronously and channel estimation scheme, be specifically related to frame detection, symbol timing, carrier frequency synchronization, sample frequency and reach channel estimation and equalization synchronously.This scheme can be widely used in the high speed short distance super broad band radio communication system based on the multi-band OFDM mode.

Background technology

Ultra broadband (UWB) system has transmission rate height, low in energy consumption, advantage such as antijamming capability is strong, is the developing direction of following wireless communication field.Mb-ofdm (MB-OFDM) is one of UWB physical-layer techniques scheme; Be proposed physical layer standard, and adopted by European ECMA-368, international ISO/IEC26907 and Chinese GB/T26229-2010 standard as wireless personal local area network IEEE802.15.3a.MB-OFDM-UWB is described as the high speed short-distance wireless communication technology of future generation that replaces bluetooth, the highest support 480Mbps transmission rate, and " Wireless USB 2.0 " is otherwise known as.The ECMA-368 standard code following system parameters: the 3.1-10.6GHz frequency range is divided into 14 widely is the sub-band of 528MHz, adopt the OFDM technology in the band; Every band comprises 128 number of sub-carrier, and it is made up of 100 data subcarriers, 12 pilot sub-carriers, 10 protection subcarriers and 6 gap carrier waves, and 37 zero suffix as protection at interval; The OFDM symbol is modulated on the carrier wave of different center frequency by time-frequency code control timesharing, realizes the multiband transmission; Comprise 24 synchronous training sequences and 6 channel estimation sequence in the standard preamble sequences.

OFDM wireless communication system band efficiency is high, and ability of anti-multipath is strong, but responsive to the Time and Frequency Synchronization deviation.The symbol timing offset causes FFT calculation window positional fault, in the CP-OFDM system if timing advance but be no more than protection and then only cause at interval phase place deflection, if definite time delay or timing advance and exceed the protection interval then can cause intersymbol interference (ISI); Carrier frequency offset causes each sub-carrier orthogonality to be destroyed, and brings and disturbs (ICI) between subcarrier; Sampling frequency deviation can bring ICI and ISI simultaneously.ICI and ISI can cause receiving terminal planisphere generation disperse, and systematic function seriously descends, therefore must be correct carry out Time and Frequency Synchronization.Ofdm system generally include 3 aspect contents synchronously: frame detect and symbol regularly; Carrier frequency offset is estimated and compensation; Sampling frequency deviation is estimated and compensation.Frame detects to be generally according to frame head information and has judged whether that the valid data frame arrives, and is the first synchronous link.Frame detects and requires very accurately the frame of leakage or empty frame can not be arranged, otherwise can produce very big error code.Timing synchronization act as the original position that finds the OFDM symbol, the start bit of window when promptly confirming to calculate FFT is so that correctly carry out the quadrature demodulation of multicarrier.Symbol timing position mistake can cause serious ISI, greatly influences systematic function.Carrier synchronization mainly is that carrier frequency is carried out subcarrier spacing being carried out normalized carrier wave frequency deviation can being divided into integer frequency offset and fractional part of frequency offset again synchronously.Carrier wave frequency deviation comes the asynchronous of self-channel Doppler frequency shift and transceiver local oscillator, destroys sub-carrier orthogonality, causes ICI.Sampling clock mainly is that sampling clock frequency is carried out synchronously synchronously, and the frequency deviation of sampling clock is caused by the asynchronous of control clock of transceiver digital-to-analogue and analog to digital conversion.The sampling clock frequency deviation can cause ICI, when the frequency deviation cumulative effect reaches an integral multiple sampled value, is equivalent to introduce a symbol timing offset, then also can bring ISI, so can get up to consider timing synchronization and sampling clock synchronization combining.

Wireless channel is divided into frequency selective fading and flat fading channel according to the big I of multidiameter delay, can be divided into rapid fading and slow fading channel according to communication terminal translational speed speed.Different channel circumstances is different to the systematic function influence, and is also different to the requirement of channel estimation ability.Channel estimation method generally realizes based on training sequence and pilot tone, correct design channel estimating supplementary and satisfy performance requirement and channel estimation method that complexity is lower is the key of ofdm system.Under the time invariant channel, channel remains unchanged in the frame, can estimate based on leading training sequence; In time varying channel, can use variation continuous or the tracking channel that scattered pilot is real-time, the pattern of pilot tone is by channel coherence time and coherence bandwidth decision, and the time domain interval of pilot tone should be less than the coherence time of channel, and frequency domain interval should be less than the coherence bandwidth of channel.

The impulse response of UWB channel can be expressed as

$h\left(t\right)=X\underset{n=1}{\overset{N}{\mathrm{\Σ}}}\underset{k=1}{\overset{K\left(n\right)}{\mathrm{\Σ}}}{\mathrm{\α}}_{\mathrm{nk}}\mathrm{\δ}(t-T_n-{\mathrm{\τ}}_{\mathrm{nk}})$

Wherein X is the lognormal stochastic variable, represents the amplitude gain of channel; N be observe bunch number, K (n) is the multipath number of receiving in n bunch, α=p _Nkβ _Nk, p _NkBe the discrete random variable of equiprobability+1 and-1, β _NkIt is the channel coefficients of the obeys logarithm normal distribution of k paths in n bunch.T _nBe the n bunch of time of advent, τ _NkBe the time delay of k paths in n bunch, the two all obeys Poisson distribution.In addition, the multipath average power becomes two exponential decay models

$E\left[{\left|h_{k,l}\right|}^2\right]={\mathrm{\Ω}}_0{e}^{-\frac{T_n}{\mathrm{\Γ}}}{e}^{-\frac{{\mathrm{\τ}}_{\mathrm{nk}}}{\mathrm{\γ}}}$

Wherein E [.] expression statistical average, Ω ₀Headed by reach the average power in footpath, Γ be a bunch arrival rate, γ is an arrival rate directly.IEEE 802.15.3a working group has confirmed 4 kinds of standard UWB channel model CM1-CM4 according to the difference of above-mentioned several kinds of parameters, and wherein the CM4 channel is the most abominable.

Table 1UWB channel model parameter

Model parameter	CM1	CM2	CM3	CM4
					Bunch arrival rate Λ (1/ns)	0.0233	0.4	0.0667	0.0667
Multipath arrival rate λ (1/ns) in bunch	2.5	0.5	2.1	2.1
					Bunch damped expoential Γ	7.1	5.5	14	24
Multipath fading exponent gamma in bunch	4.3	6.7	7.9	12
					Root mean square time delay expansion (ns)	5	8	15	26
NP(85％)	4.2	6.5	9.7	15.6
					The footpath number is got in emulation	8	12	16	26

Summary of the invention

The objective of the invention is to propose complete synchronous and channel estimation scheme, comprise that frame detection, thick timing, thin timing, carrier frequency synchronization, sample frequency reach channel equalization synchronously based on the MB-OFDM-UWB system of ECMA-368 standard code.The aim that this scheme is confirmed is to correct the synchronism deviation that exists in all reality and the influence of channel, strive making system to have good error performance, and algorithm complex is tried one's best low so that the follow-up hardware that carries out algorithm is realized.

Technical scheme of the present invention:

The present invention proposes to be applicable to the synchronous and channel estimating complete scheme of MB-OFDM-UWB system: the multiband implementation method that adopts BB-FH; Frame detects and slightly regularly adopts improved S-C method, to the correction that lags behind of thick timing position; The thin characteristics of system zero prefix of regularly utilizing synchronously adopt the least energy ratio method, and confirm the hunting zone of minimum ratio according to thick timing offset scope, equally to the correction that lags behind of thin timing position; The influence of the single carrier frequency deviation of RF part under the BB-FH mechanism is equal to the influence of RF part multicarrier frequency deviation under the RF frequency hopping dehop mechanism, and carrier wave frequency deviation adopts the ML method synchronously, carries out frequency offset estimating with the equal method of making even, and the symbol of 3 frequency bands of feedforward compensation; System need not thin Frequency Synchronization; Time domain OLA operation is converted into the CP-OFDM system of equivalence with the ZP-OFDM system, the amplitude fading that frequency domain channel equalization compensates simultaneously that multipath disturbs, residue symbol timing offset, carrier wave skew and CFO, SFO cause; Sample frequency is used the method for frequency domain phase equalization synchronously, and the time-domain symbol deviation that it causes absorbs with CP equally.

Beneficial effect of the present invention:

The present invention proposes the complete synchronous and channel estimation optimizing scheme that is applicable to the MB-OFDM-UWB system, system ISI and ICI that the Time and Frequency Synchronization deviation is caused carry out the high accuracy elimination, and the influence of ultra broadband multipath channel is estimated and compensation efficiently.Simultaneously, this scheme makes full use of system protection tolerance power at interval, realizes the maximized collaborative work of each several part algorithm when guaranteeing systematic function, and to reduce the system receiver overall complexity, the hardware of being convenient to system is realized.

Description of drawings

Fig. 1 is a base band dehop theory diagram

Fig. 2 is preamble synchronization and channel estimation sequence

Fig. 3 is system synchronization and channel estimating global design scheme

Fig. 4 is the frame related operation result in the timing synchronously

Fig. 5 (a) is a thick timing offset histogram under the CM1 channel

Fig. 5 (b) is a thick timing offset histogram under the CM2 channel

Fig. 5 (c) is a thick timing offset histogram under the CM3 channel

Fig. 5 (d) is a thick timing offset histogram under the CM4 channel

Fig. 6 (a) is a thin timing offset histogram under the CM1 channel

Fig. 6 (b) is a thin timing offset histogram under the CM2 channel

Fig. 6 (c) is a thin timing offset histogram under the CM3 channel

Fig. 6 (d) is a thin timing offset histogram under the CM4 channel

Fig. 7 is a revised thickness timing offset probability under the CM4 channel

Fig. 8 is the root-mean-square error curve of the residual carrier frequency deviation under the different frequency method for synchronous

Fig. 9 is the QPSK planisphere before and after the carrier frequency synchronization

Figure 10 is the system BER performance under different carrier frequency deviation and the channel model

Figure 11 is a system BER performance of removing thin timing front and back under the CM4 channel

Embodiment

Be described further below in conjunction with accompanying drawing with through the embodiment specific embodiments of the invention:

The present invention has designed the complete synchronous and channel estimation scheme of multi-band OFDM radio ultra wide band system, and it is characterized in that: this programme may further comprise the steps:

A. system adopts baseband digital domain frequency hopping and the multiband implementation of separating jumping, behind the baseband transmitter Waveform shaping, carries out frequency hopping, before the receiver matched filtering, implements dehop;

B. the carrier wave in the receiver is initiated on base band first mid-band frequency, thick regularly synchronously to system implementation, to corrections that lag behind of thick timing position, and according to revised position startup dehop;

C. implement thin timing synchronously based on the information of 3 frequency bands after starting dehop, thin timing position is carried out the second order lag correction;

D. estimate three frequency band frequency deviations respectively based on the preamble synchronization sequence, it is all back as the second frequency band frequency offset estimating value to make even, and releases other two frequency bands frequency deviation;

E. utilize the overlap-add operation that the ZP-OFDM system is converted into the CP-OFDM system;

F. carry out frequency domain zeroth order channel equalization based on leading channel estimation sequence;

G. carry out the phase deviation that frequency domain phase equalization causes with compensating sampling frequency offset based on pilot tone;

H. the symbol timing position is carried out leading revising in advance the integral multiple sample value deviation that causes in order to compensating sampling frequency offset.

Under the baseband hopping mechanism described in the step a, only by the local oscillator control of single-frequency, this carrier frequency is the centre frequency of second frequency band in 3 frequency bands to the frequency mixer of system's transceiver.On-F, 0, three frequencies of F, 3 band spectrums that base band generates are modulated to radio frequency through second carrier wave integral body to the single band frequency spectrum in base band timesharing frequency hopping, and wherein F is 2 times of single band baseband bandwidth.

Thick timing described in the step b synchronously institute based on normalized autocorrelation adjudicate estimate for

$C\left(i\right)=\frac{\left|\underset{k=i}{\overset{i+3M-1}{\mathrm{\Σ}}}{r}^{*}\left(k\right)r(k+21M)\right|}{\underset{k=i}{\overset{i+3M-1}{\mathrm{\Σ}}}{\left|r(k+21M)\right|}^2}$

Wherein M is the OFDM symbol lengths, k the data of r (k) for receiving.

Thin timing described in the step c synchronously institute based on energy window ratio adjudicate estimate for

$z\left(i\right)=\frac{\underset{q^{\′}=0}{\overset{3R-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{H-1}{\mathrm{\Σ}}}{\left|r(i-m+q^{\′}M)\right|}^2}{\underset{q^{\′}=0}{\overset{3R-1}{\mathrm{\Σ}}}\underset{n=1}{\overset{H}{\mathrm{\Σ}}}{\left|r(i+n+q^{\′}M)\right|}^2}$

Wherein H is a length of window, and R is a number of symbols.

The formula based on synchronizing sequence estimation frequency deviation described in the steps d does

${\mathrm{\&epsiv;}}_q=\frac{\mathrm{<figure-callout\; id="2"\; label="N"\; filenames="HSA00000720770000031.png,HSA00000720770000032.png"\; state="\{\{state\}\}">N</figure-callout>}}{2\mathrm{\&pi;}3M{D}_q}\mathrm{angle}\left(\underset{k=1}{\overset{R-D_q}{\mathrm{\&Sigma;}}}\underset{n=1}{\overset{M}{\mathrm{\&Sigma;}}}{r_{q,k,n}}^{*}{r}_{q,k+D_q,n}\right)$

D wherein _qSatisfy

ε _QmaxBe the greatest normalized frequency deviation of q frequency band, N is a number of sub carrier wave, and k is the symbol sequence number, and n is a data sequence number.

Overlap-add operation described in the step e can be described as

r _i′＝r _i+λ _i

Wherein, i time domain OFDM symbol receiving of system is 1 * (N+N _g) vector

${\mathrm{\&lambda;}}_i=[r_{i,N},r_{i,N+1},\&CenterDot;\&CenterDot;\&CenterDot;,r_{i,N+N_g-1},0],$ Wherein 0 is the null vector of 1 * N.

Frequency domain zeroth order channel equalization process described in the step f can be described as

${R_{i,k}}^{\&prime;}=R_{i,k}\frac{S_{\mathrm{CE},k}}{R_{\mathrm{CE},k}}$

S wherein _{CE, k}, R _{CE, k}Be respectively transmitting terminal and receiving terminal channel estimation sequence value, R at k number of sub-carrier place _{I, k}Be the data on i symbol k of the receiving terminal number of sub-carrier.

Frequency domain phase equalization process described in the step g can be described as

${\hat{s}}_i=\underset{k_j\&Element;K}{\mathrm{\&Sigma;}}{k}_j{\mathrm{\&theta;}}_{i,j}/\underset{k_j\&Element;\mathrm{<figure-callout\; id="2"\; label="K\; k\; j"\; filenames="HSA00000720770000031.png,HSA00000720770000032.png"\; state="\{\{state\}\}">K</figure-callout>}}{\mathrm{\&Sigma;}}{k_j}^{}{{}_{}}^2$

K wherein _jBe pilot sub-carrier sequence number, θ _{I, j}Be i the phase deviation that symbol j number of sub-carrier place estimates.

The preparatory correction value of symbol timing advance described in the step h equals issuable maximum integer times sample value deviation number in the frame that sampling frequency offset causes.

Embodiment

Make up 200bps pattern UWB system according to the ECMA-368 standard, the synchronous and channel estimation scheme that the present invention proposes is carried out emulation.

Shown in Figure 4 is the related operation result in one frame when signal to noise ratio is 0dB under the CM4 channel, can see that C (i) changes between 0 to 1, a relevant peaks is arranged in the frame, and the peak is more sharp-pointed.A threshold value G is confirmed in emulation, when continuous 2 C (i) when becoming less than G greater than G, confirm as and detected frame.

Fig. 5 (a) (b) (c) (d) is respectively thick timing offset histogram when SNR is 5dB under CM1, CM2, CM3, the CM4 channel, and wherein the UWB channel setting is that a frame changes once, and the number of frame is about 1500 in the emulation.Can find out that timing offset presents Gaussian distribution substantially under the various channel models, under CM1～CM4 channel, thick timing offset scope is respectively [1; 4], [2,7], [1; 11], [4,20], timing offset minimum and distribution are the narrowest under the CM1 channel; Under the most serious CM4 channel of multipath, timing offset maximum and distribution are the wideest, and the visible S-C legal time is responsive to multipath synchronously.Can know by Fig. 5, under the 5dB signal to noise ratio maximum hysteresis deviation for the CM4 channel down corresponding 20, so correction value is decided to be 20, revise then back four kinds of channels under in advance slightly that the timing offset scope is [21 ,-16], [22 ,-13], [21 ,-9], [16,0].The channel estimation methods of native system adopts OLA and channel equalization, and it allows less remaining timing offset, and deviation range is [(N _g-L), and 0], N wherein _g=37, L is the channel multi-path number, the ISI that can avoid multipath disperse and symbol timing offset and effect to bring like this.The emulation value of L is seen table 1 under four kinds of channels, can calculate the maximum leading timing offset that allows under four kinds of channels through this table and be respectively 29,25,21,11.It is thus clear that the maximum leading deviation 16 under the CM4 channel has slightly exceeded allowed band, can bring certain ISI, causes the decline of system signal noise ratio.The thick maximal correlation method that regularly adopts is that multipath is limited, and timing position is that main footpath is the position of most powerful path, and is not necessarily first footpath, so timing error is bigger.

Take all factors into consideration implementation complexity, confirm that thin energy window length regularly is 5, adopt leading continuous 8 groups of synchronizing symbols.Fig. 6 is the remaining deviation histogram after the thin timing synchronously when SNR is 5dB under CM1～CM4 channel.From figure, can obviously find out, relatively remain timing offset with thick timing results and be greatly improved.Under CM1～CM4 channel, thin timing offset scope is respectively [0,1], [0; 3], [1,2], [0; 4], thin maximum hysteresis deviation regularly is 4 under the CM4 channel, so the hysteresis modifying factor confirms as 4; The maximum down leading timing offset of revised CM4 channel is-4, satisfies the condition [11,0] that does not produce ISI under the CM4 channel.

Fig. 7 is under signal to noise ratio 5dB, normalization frequency deviation 0.0384, the CM4 channel, the residue timing error probability graph through lagging behind and revising after the thickness timing.From figure, can obviously find out thin timing, and time synchronization method does not receive the influence of frequency deviation synchronously for the effect that reduces timing offset.

Fig. 8 is corresponding residual frequency deviation root-mean-square error (MSRE) curve of the method for average, MMSE method and direct method.As can be seen from the figure, be better than pair of symbols based on all symbol performances, the MMSE method is very close with the method performance of making even equal, and the two all is better than directly utilizes 3 band information estimation approach.During the 20dB signal to noise ratio, estimate that based on two group codes the frequency deviation equal estimated accuracy of making even again is 10 ^-5The order of magnitude can reach 10 based on the estimation of all symbols ^-6The order of magnitude, but on implementation complexity utilizes many group codes to estimate to be higher than two group codes, intends here only adopting based on the estimation frequency deviation method of two groups of synchronizing symbols and comes the planisphere behind the emulation compensate of frequency deviation, sees whether residual frequency deviation produces considerable influence.Fig. 9 (a) (b) is depicted as under signal to noise ratio 5dB, the maximum frequency deviation, adds the QPSK planisphere of frequency deviation synchronized algorithm front and back.Set the planisphere parameter in the emulation, make it can show whole 522 * 100 constellation point in the frame simultaneously.As can be seen from the figure, this frequency synchronization method has carried out effective correction to planisphere, and constellation point is distributed in four quadrants, and the situation of not crossing the border occurs.This shows, estimate that based on two group codes the frequency deviation equal method of making even again can reach performance requirement, and it implements more easyly, complexity is lower.

The back that estimates is according to the frequency deviation proportionate relationship and then can draw the frequency deviation of 3 frequency bands.Utilize the feedforward compensation method to compensate the signal of three frequency bands respectively during carrier frequency compensation; Begin all sample values are carried out the rotation of linear increment phase place in time domain from the first sampled point of present frame; The sequence number that multiply by phase place is the sequence number of the sampled point in the frame; And every at a distance from a symbol, phase hit once, the phase hit cycle is 3 symbols.Compensation is during next frame, compensates according to the frequency offset estimating value of next frame, and the sequence number that multiply by phase place is again since 0.All sample values also have a common excess phase deviation in one frame of compensation back like this, and it can be corrected by the frequency domain channel estimation with carrier wave initial phase deviation.

Carrier wave frequency deviation makes the phase place rotation that there be in each symbol; And with the increase of symbol sequence number, the anglec of rotation constantly increases.Coarse frequency synchronously after, symbol amplitude decay and ICI that remaining very little frequency deviation causes all can ignore, but can cause the rotation of planisphere with the cumulative effect of the increase phase deviation of symbol sequence number, possibly cause the symbol judgement mistake that sequence number is bigger in the frame.Therefore, need need to determine whether further thin frequency synchronization algorithm through observing the planisphere of receiving terminal under the said frequencies method for synchronous.This emulation based on system frame structure be 30 leading training symbols and add 522 useful information symbols, 522 the maximum information numbers of symbols that can get that information symbol is standard code wherein.As can beappreciated from fig. 9, all constellation point all do not have big phase place deflection, explain that residual frequency deviation does not produce significantly influence to the symbol that sequence number is bigger in the frame, so need not further thin Frequency Synchronization.

During the simulated channel equalization performance; Add timing and carrier synchronization effects in the system; Receiving terminal add that incoming frame detects and thick regularly, thin timing, carrier frequency synchronization, OLA and channel equalization algorithm, Figure 10 is zero system's bit error rate curve when reaching maximum for carrier wave frequency deviation under CM1 and the CM4 channel.Can find out that when systematic function was better than maximum frequency deviation when not having frequency deviation, systematic function was better than the CM4 channel under the same frequency deviation CM1 channel.System's bit error rate when signal to noise ratio is 3dB under the maximum frequency deviation CM4 channel is 10 ^-6The order of magnitude, the channel equalization method that visible the present invention proposes has been given good correction to the UWB channel.

The influence that adds all synchronism deviation factors in the analogue system, carrier wave and sampling clock relative frequency deviation 40ppm.Figure 11 is the system's bit error rate curve chart under the CM4 channel.The error performance indexs of being with alliance to stipulate system: under the 480Mbps pattern, when SNR was 8dB, system's bit error rate should reach 10 more ^-5The order of magnitude.This analogue system is based on the 200Mbps pattern, from figure, can leave increase BER along with signal to noise ratio and have and reduce, and system's bit error rate can reach 10 when SNR was 3dB ^-5The order of magnitude, the compliance with system performance demands is not having under the thin situation regularly, and along with the increase of signal to noise ratio, the trend that BER reduces slows down slowly gradually, and this is owing to a certain amount of ISI causes, and system's bit error rate can reach 10 after the 4dB ^-5The order of magnitude satisfies performance requirement equally.This shows that ISI is less to the influence of systematic function, all less because thick timing advance deviation falls into the power in several footpaths of length and multipath channel end of last symbol time delay expansion, thus only produced less ISI in the system.

Claims (9)

1. The multi-band OFDM ultra wideband receiver synchronously and channel estimation scheme, it is characterized in that: this programme may further comprise the steps:

   A. system adopts baseband digital domain frequency hopping and the multiband implementation of separating jumping, behind the baseband transmitter Waveform shaping, carries out frequency hopping, before the receiver matched filtering, implements dehop;

   B. the carrier wave in the receiver is initiated on base band first mid-band frequency, thick regularly synchronously to system implementation, to corrections that lag behind of thick timing position, and according to revised position startup dehop;

   C. implement thin timing synchronously based on the information of 3 frequency bands after starting dehop, thin timing position is carried out the second order lag correction;

   D. estimate three frequency band frequency deviations respectively based on the preamble synchronization sequence, it is all back as the second frequency band frequency offset estimating value to make even, and releases other two frequency bands frequency deviation;

   E. utilize the overlap-add operation that the ZP-OFDM system is converted into the CP-OFDM system;

   F. carry out frequency domain zeroth order channel equalization based on leading channel estimation sequence;

   G. carry out the phase deviation that frequency domain phase equalization causes with compensating sampling frequency offset based on pilot tone;

   H. the symbol timing position is carried out leading revising in advance the integral multiple sample value deviation that causes in order to compensating sampling frequency offset.
2. 2. the synchronous and channel estimation scheme of multi-band OFDM ultra wideband receiver according to claim 1; It is characterized in that: under the baseband hopping mechanism described in the step a; Only by the local oscillator control of single-frequency, this carrier frequency is the centre frequency of second frequency band in 3 frequency bands to the frequency mixer of system's transceiver.On-F, 0, three frequencies of F, 3 band spectrums that base band generates are modulated to radio frequency through second carrier wave integral body to the single band frequency spectrum in base band timesharing frequency hopping, and wherein F is 2 times of single band baseband bandwidth.
3. 3. multi-band OFDM ultra wideband receiver according to claim 1 is characterized in that synchronously and channel estimation scheme: the thick timing described in the step b synchronously institute based on normalized autocorrelation adjudicate estimate for

   $C\left(i\right)=\frac{\left|\underset{k=i}{\overset{i+3M-1}{\mathrm{\&Sigma;}}}{r}^{*}\left(k\right)r(k+21M)\right|}{\underset{k=i}{\overset{i+3M-1}{\mathrm{\&Sigma;}}}{\left|r(k+21M)\right|}^2}$

   Wherein M is the OFDM symbol lengths, k the data of r (k) for receiving.
4. 4. multi-band OFDM ultra wideband receiver according to claim 1 is characterized in that synchronously and channel estimation scheme: the thin timing described in the step c synchronously institute based on energy window ratio adjudicate estimate for

   $z\left(i\right)=\frac{\underset{q^{\&prime;}=0}{\overset{3R-1}{\mathrm{\&Sigma;}}}\underset{m=0}{\overset{H-1}{\mathrm{\&Sigma;}}}{\left|r(i-m+q^{\&prime;}M)\right|}^2}{\underset{q^{\&prime;}=0}{\overset{3R-1}{\mathrm{\&Sigma;}}}\underset{n=1}{\overset{H}{\mathrm{\&Sigma;}}}{\left|r(i+n+q^{\&prime;}M)\right|}^2}$

   Wherein H is a length of window, and R is a number of symbols.
5. 5. the synchronous and channel estimation scheme of multi-band OFDM ultra wideband receiver according to claim 1 is characterized in that: the formula based on synchronizing sequence estimation frequency deviation described in the steps d does

   ${\mathrm{\&epsiv;}}_q=\frac{N}{2\mathrm{\&pi;}3M{D}_q}\mathrm{angle}\left(\underset{k=1}{\overset{R-D_q}{\mathrm{\&Sigma;}}}\underset{n=1}{\overset{M}{\mathrm{\&Sigma;}}}{r_{q,k,n}}^{*}{r}_{q,k+D_q,n}\right)$

   D wherein _qSatisfy

   

   ε _QmaxBe the greatest normalized frequency deviation of q frequency band, N is a number of sub carrier wave, and k is the symbol sequence number, and n is a data sequence number.
6. 6. the synchronous and channel estimation scheme of multi-band OFDM ultra wideband receiver according to claim 1 is characterized in that: the overlap-add operation described in the step e can be described as

   r _i′＝r _i+λ _i

   Wherein, i time domain OFDM symbol receiving of system is 1 * (N+N _g) vector

   

   ${\mathrm{\&lambda;}}_i=[r_{i,N},r_{i,N+1},\&CenterDot;\&CenterDot;\&CenterDot;,r_{i,N+N_g-1},0],$ Wherein 0 is the null vector of 1 * N.
7. 7. the synchronous and channel estimation scheme of multi-band OFDM ultra wideband receiver according to claim 1, it is characterized in that: the frequency domain zeroth order channel equalization process described in the step f can be described as

   ${R_{i,k}}^{\&prime;}=R_{i,k}\frac{S_{\mathrm{CE},k}}{R_{\mathrm{CE},k}}$

   S wherein _{CE, k}, R _{CE, k}Be respectively transmitting terminal and receiving terminal channel estimation sequence value, R at k number of sub-carrier place _{I, k}Be the data on i symbol k of the receiving terminal number of sub-carrier.
8. 8. the synchronous and channel estimation scheme of multi-band OFDM ultra wideband receiver according to claim 1, it is characterized in that: the frequency domain phase equalization process described in the step g can be described as

   ${\hat{s}}_i=\underset{k_j\&Element;K}{\mathrm{\&Sigma;}}{k}_j{\mathrm{\&theta;}}_{i,j}/\underset{k_j\&Element;K}{\mathrm{\&Sigma;}}{k_j}^2$

   K wherein _jBe pilot sub-carrier sequence number, θ _{I, j}Be i the phase deviation that symbol j number of sub-carrier place estimates.
9. 9. multi-band OFDM ultra wideband receiver according to claim 1 is characterized in that synchronously and channel estimation scheme: the preparatory correction value of symbol timing advance described in the step h equals issuable maximum integer times sample value deviation number in the frame that sampling frequency offset causes.

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