CN102347695A - Series resonant converter - Google Patents
Series resonant converter Download PDFInfo
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- CN102347695A CN102347695A CN2011100309990A CN201110030999A CN102347695A CN 102347695 A CN102347695 A CN 102347695A CN 2011100309990 A CN2011100309990 A CN 2011100309990A CN 201110030999 A CN201110030999 A CN 201110030999A CN 102347695 A CN102347695 A CN 102347695A
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M3/00—Conversion of dc power input into dc power output
- H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
- H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
- H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
- H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
- H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M1/00—Details of apparatus for conversion
- H02M1/08—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M1/00—Details of apparatus for conversion
- H02M1/0048—Circuits or arrangements for reducing losses
- H02M1/0054—Transistor switching losses
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- Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
- Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
- Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
- Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
- Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes
Abstract
The invention provides an efficient series resonant converter capable of reducing switch loss and improving efficiency by arranging a separate capacitor in a secondary winding of a transformer for increasing resonance current of a primary winding of the transformer rapidly when a switch unit is switched on. The series resonant converter is capable of reducing the switch loss by satisfying the condition of zero voltage and zero current switch when the switching unit is switched on and the condition of zero voltage switch when the switching unit is switched off. Moreover, the efficiency of the series resonant coverter can be improved through rapid increase of the resonance current during charging of the secondary capacitor on the secondary winding.
Description
Technical field
The present invention relates to series resonant converter, relate in particular to through improving the resonance current waveform the efficient series resonant converter of the efficient of the switching loss that reduces and increase is provided.
Background technology
DC-to-DC converter is to be used for the electronic circuit of direct current (DC) source from a voltage level conversion to another voltage level.Usually, DC-to-DC converter converts direct voltage to interchange (AC) voltage, raises through transformer and perhaps reduces this alternating voltage, and will convert direct voltage to through the alternating voltage that raises or reduce.
Series resonant converter (SRC) is an example of DC-to-DC converter.
Fig. 1 is the circuit diagram of conventional SRC.This SRC uses the resonance that is generated by inductor Lr and capacitor Cr, and presents good conversion efficiency.
With reference to figure 1, this SRC comprises switch element 20, LC resonant circuit 30, transformer TX, bridge rectifier 40 and gate driver 51.Switch element 20 comprises that a plurality of switch S 1-S4 will be will change over alternating voltage from the direct voltage of input voltage source 10 through the alternation direct voltage.This LC resonant circuit 30 is connected to switch element 20, and comprises the resonant inductor Lr resonant capacitor Cr that is one another in series and connects.LC resonant circuit 30 uses the resonance that is generated by resonant inductor Lr resonant capacitor Cr to change the frequency characteristic from the alternating voltage of switch element 20.Transformer TX is with primary voltage, and promptly the alternating voltage from LC resonant circuit 30 converts secondary voltage to.Bridge rectifier 40 converts secondary alternating voltage to direct voltage.Gate driver 51 control switch unit 20 are with the amplitude and the shape of control load electric current.
SRC also comprises the direct voltage from bridge rectifier 40 is carried out filtering and will be applied to the capacitor C of load 60 through the direct voltage of filtering
0
Full-bridge pulse-width modulation (PWM) transducer that SRC is to use four the semiconductor switch S1-S4 such as insulated gate bipolar transistor (IGBT) or mos field effect transistor (MOSFET) with full bridge structure interconnection to realize.Switch S 1-S4 is connected in parallel to inverse parallel diode D1-D4, and parallelly connected with buffer condenser CS1-CS4.
Switch element 20 through under the control of gate driver 51, connect synchronously or turn-off pair of switches S1 and S4 or another to switch S2 and S3 with converting direct-current voltage into alternating-current voltage.Alternating voltage is transferred to the secondary winding TX2 of transformer TX through LC resonant circuit 30.
This transformer TX is through the energy of secondary winding TX2 output from LC resonant circuit 30.Induced potential among the secondary winding TX2 is confirmed by the ratio of the quantity of the quantity of the number of turn among the secondary winding TX2 and the number of turn among the elementary winding TX1.
The bridge rectifier 40 that comprises four rectifier diode RD1-RD4 will convert direct voltage to from the alternating voltage of inducting of secondary winding TX2 output.This direct voltage is through capacitor C
0Carry out filtering, and output to load 60 then.
This gate driver 51 turns on and off switch S 1-S4 during the driving of SRC and power conversion process.Gate driver 51 received pulse voltage signals are as input and generate drive signal, promptly gating signal so that switch S 1-S4 turn on and off.
During the switching manipulation of semiconductor switch S1-S4, change voltage and current with predetermined delay in each switch and gradient.Therefore, when switch S 1-S4 turns on and off, can exist voltage and current can be applied to the section of switch, the i.e. section of voltage and current part crossover synchronously.In this section, the switching loss corresponding to the product V * I of voltage and current can take place.
For example, when IGBT by the time, apply fully at the IGBT two ends that tail current can continue to flow after the voltage, thereby cause serious switching loss.
Switching loss reduces the efficient of transducer and causes the switch heating.In addition, the switching frequency of switching loss and switch increases pro rata, thus the maximum switching frequency of limit switch.
In order to reduce switching loss, the various switching mechanisms such as zero voltage switch (ZVS), Zero Current Switch (ZCS) and zero-voltage zero-current (ZVZCS) have been proposed.
In order to realize that ZVS, ZCS and ZVZCS to reduce switching loss, as shown in Figure 1, provide the load resonant transducer, it uses LC resonance through the elementary winding TX1 that inductor Lr and capacitor Cr is connected to transformer TX.LC resonance can allow transducer to generate the voltage and current waveform that satisfies no-voltage and zero current condition.In addition, the LC resonant circuit can allow load voltage and load current vibration, realizes ZVS, ZVC or ZVZCS thus.
Fig. 2 is the resonance current i that describes to depend on the switching frequency fs in the load resonant transducer
LThe curve chart of characteristic.The switching manipulation of load resonant transducer depends on that switching frequency can be divided into two patterns, promptly discontinuous conduction mode (DCM) and continuous conduction mode (CCM).In DCM, in the switching frequency fs section lower, carry out switching manipulation than the resonance frequency fr of inductor and capacitor.In CCM, in the switching frequency fs section higher, carry out switching manipulation than resonance frequency fr.
Fig. 3 a and 3b are the exemplary voltages describing to be caused by the LC resonance in the conventional load resonance converter and the curve chart (for convenience's sake, half-bridge structure being shown) of current waveform.Fig. 3 a illustrates the harmonic wave among the DCM, and Fig. 3 b illustrates the harmonic wave among the CCM.Current i
LIndication inductor current and voltage v
CThe indication condenser voltage.
With reference to figure 3a, the switching frequency fs in DCM than the low one-period of resonance frequency fr during, connect when allowing electric current to flow when the pair of switches of full bridge structure, electric current gathers in inductor Lr.The electric energy that gathers is passed to capacitor Cr, increases condenser voltage v thus
CAccumulate in after electric energy among the inductor Lr dissipates fully, the polarity upset of capacitor, thus cause current i
LFlow in the other direction.Pair of switches is turn-offed then.Therefore, appearance does not wherein have the discontinuous segment that electric current flows.
In this discontinuous segment, when another applies voltage v during to switch connection on opposite direction
CAnd current i
LCorrespondingly, because there is the wherein discontinuous mobile discontinuous segment of electric current, connect switch so can carry out zero current.
With reference to figure 3b, the switching frequency fs in CCM than the high one-period of resonance frequency fr during, inductor current i when pair of switches is connected
LIncrease, and condenser voltage v then
CInductor current i during increase
LReduce.Electric current is no longer mobile when pair of switches is turn-offed.When another during to switch connection (, zero current is connected switch), to apply voltage in the other direction.As a result,, voltage on opposite direction, applies electric current when beginning to descend.Like this, accomplished a switch periods.That is electric current continuous flow during one-period.
In these two patterns, (PFM) controls output current through pulse frequency modulated.In DCM, higher frequency causes the increase of output current.In CCM, lower frequency causes the increase of resonance current, increases the load current of exporting after the full-wave rectification thus.
In CCM, when under the same frequency operating condition, connecting switch resonance current i when will give resonance current near the waveform of square wave
LWhen sharply increasing, this resonance current can have the effective value of increase.As a result, transducer can present the efficient of improvement.
Therefore, need be used for increasing resonance current i fast at CCM
LThe technology that has the transducer of improved efficiency with acquisition.
Summary of the invention
The present invention is contemplated to and solves above-described prior art problems, and one side of the present invention the resonance current waveform in the switching manipulation provides the efficient series resonant converter that can reduce switching loss and improve efficient in the continuous conduction mode through improving.
According to an aspect of the present invention, efficient series resonant converter comprises: comprise the switch element of a plurality of switches, be used for through the alternation direct voltage direct current (DC) voltage transitions being become to exchange (AC) voltage; The LC resonant circuit, it comprises the resonant inductor resonant capacitor that is connected in series, and is connected to switch element, and uses the resonance that is generated by resonant inductor resonant capacitor to change the frequency characteristic of transmission from the alternating voltage of switch element; The transformer that comprises elementary winding and secondary winding, this elementary winding is connected to the LC resonant circuit, and in induced potential in the secondary winding and the secondary winding in quantity and the elementary winding of the number of turn ratio of the quantity of the number of turn proportional; Secondary capacitor, its be connected to Secondary winding of transformer and and transformers connected in parallel; The full-bridge rectifier that comprises a plurality of rectifier diodes, the alternating voltage that is used for inducting at secondary winding converts direct voltage to; And gate driver, its detection is connected to the conducting of the inverse parallel diode of switch, and when the inverse parallel diode current flow, exports the conducting gating signal to connect switch.
Secondary capacitor can have the electric capacity littler than resonant capacitor.
Secondary capacitor can have the impedance lower than the circuit of being made up of rectifier diode and load, and the secondary capacitor that is used in the load current charging of inducting in the secondary winding can cause the resonance current of LC resonant circuit to increase fast.
Gate driver can comprise: be connected to first resistor to the input of its input pulse voltage signal; Be parallel-connected to first resistor of input end and use the capacitor of the conducting pulse voltage charging that applies through input; The semiconductor switch that comprises source electrode, grid and drain electrode, source electrode, grid and drain electrode are connected respectively to the output node of input, capacitor and are connected to the gate node of output of the switch of switch element; The 3rd resistor that between the gate node of the drain electrode of semiconductor switch and output, connects; And be connected to input to form the 4th resistor of conducting path through first resistor and capacitor.
This second resistor can have the resistance bigger than the 4th resistor.
The accompanying drawing summary
According to the following description of the exemplary embodiment that combines appended accompanying drawing to provide, of the present invention abovely will become obviously with others, characteristic and advantage, wherein:
Fig. 1 is the circuit diagram of conventional series resonant converter;
Fig. 2 is a curve chart of describing to depend on the resonance current characteristic of the switching frequency in the load resonant transducer;
Fig. 3 a and 3b are the exemplary voltages describing to be caused by the LC resonance in the conventional load resonance converter and the curve chart of current waveform;
Fig. 4 is the circuit diagram of the series resonant converter of one exemplary embodiment according to the present invention;
Fig. 5 is a curve chart of describing the voltage and current waveform of the continuous conduction mode in the series resonant converter of one exemplary embodiment according to the present invention;
Fig. 6 is the comparative curve chart of the voltage and current waveform of the continuous conduction mode in depicted example property series resonant converter and the conventional series resonant converter;
Fig. 7-the 14th illustrates the circuit diagram of the operation of each pattern in the series resonant converter of one exemplary embodiment according to the present invention;
Figure 15 is the circuit diagram of the gate driver of the series resonant converter of one exemplary embodiment according to the present invention;
Figure 16 illustrates the pulse voltage signal of conduct to the input signal of gate driver shown in Figure 15;
Figure 17 illustrates the gating signal of conduct from the output signal of gate driver shown in Figure 15;
Figure 18 is resonance current and the gating signal of the depicted example property series resonant converter curve chart with respect to the time; And
Figure 19-the 22nd illustrates the circuit diagram of the operation of each pattern in the gate driver of one exemplary embodiment according to the present invention.
Embodiment
Specifically describe referring now to the exemplary embodiment of accompanying drawing invention.
The present invention provides and can reduce the series resonant converter that switching loss is improved efficient simultaneously through improving the resonance current waveform.More specifically, the resonance current waveform that the series resonant converter of one exemplary embodiment can improve through the capacitor that is added into Secondary winding of transformer in the switching manipulation of continuous conduction mode according to the present invention reduces switching loss and improves efficient.
Fig. 4 is the circuit diagram of the series resonant converter (SRC) of one exemplary embodiment according to the present invention.
With reference to figure 4, this SRC comprises switch element 20, LC resonant circuit 30, transformer TX, capacitor C2, bridge rectifier 40 and gate driver 51.Switch element 20 comprises a plurality of switch S 1-S4, and these a plurality of switch S 1-S4 are used for will becoming to exchange (AC) voltage from direct current (DC) voltage transitions of input voltage source 10 through the alternation direct voltage.LC resonant circuit 30 uses LC resonance with the frequency characteristic of conversion from the alternating voltage of switch element 20.Transformer TX is with primary voltage, and promptly the alternating voltage from LC resonant circuit 30 converts secondary voltage to.Capacitor C2 is connected to the secondary winding TX2 of transformer TX, and parallelly connected with transformer TX.The alternating voltage that bridge rectifier 40 will be inducted in the secondary winding TX2 of transformer TX converts direct voltage to.Gate driver 51 control switch unit 20 are with the amplitude and the shape of control load electric current.
The similar that this SRC has is in the structure of conventional SRC.That is, switch element 20 comprises four semiconductor switch S1-S4 such as IGBT or MOSFET, and it connects with full bridge structure; Inverse parallel diode D1-D4 is parallel-connected to switch S 1-S4; And buffer condenser CS1-CS4 is parallel-connected to diode D1-D4.
In addition, realize this SRC with the structure that is similar to conventional SRC structure.Promptly; Pair of switches S1 in the switch element 20 and S4 or another are connected synchronously or are turn-offed down in the drive signal (gating signal) of gate driver 51 switch S2 and S3; With with converting direct-current voltage into alternating-current voltage, and this alternating voltage is transferred to the elementary winding TX1 of transformer TX through LC resonant circuit 30; LC resonant circuit 30 comprises resonant inductor Lr resonant capacitor Cr, and the elementary winding TX1 that they are connected in series to transformer TX between another contact node of the contact node of switch S 1 and S2 and switch S 3 and S4 is to reduce switching loss; Transformer TX converts primary voltage to the secondary voltage at the secondary winding TX2 two ends of transformer TX; The bridge rectifier 40 that comprises rectifier diode RD1-RD4 converts the alternating voltage of inducting at secondary winding TX2 two ends to direct voltage; And should be through the direct voltage of rectification by capacitor C
0Carry out filtering, output to load 60 then.
But in this embodiment, SRC also comprises another capacitor C2, and this capacitor C2 is parallel-connected to transformer TX on secondary winding TX2.This secondary capacitor C2 allows SRC to produce the resonance current waveform that improves, and reduces the efficient that switching loss is improved SRC simultaneously thus.
For the operation of each pattern of the following stated, secondary capacitor C2 has the electric capacity littler than resonant capacitor Cr.For example, secondary capacitor C2 can have 1/20～1/5 electric capacity (for example 0.3 μ F) of the electric capacity (for example 3 μ F) that is resonant capacitor Cr.
The secondary capacitor C2 that is added can make SCR have the low impedance of combination than rectifier diode RD1-RD4 and load 60.
Operation in each pattern of the SRC that comprises secondary capacitor C2 will be described with reference to the drawings.
Fig. 5 is voltage Vc and an electric current I of describing the continuous conduction mode (CCM) among the SRC of one exemplary embodiment according to the present invention
LThe curve chart of waveform.Fig. 6 is voltage Vc and the electric current I of the CCM among depicted example property SRC and the conventional SRC
LThe comparative curve chart of waveform.
With reference to figure 5, I
LThe expression resonant inductor current, V
CExpression resonant capacitor voltage, the moment that pair of switches S1 among the initial point indication full-bridge switch S1-S4 of pattern 1 and S4 connect, t
2The moment that indication pair of switches S1 and S4 turn-off, t
4Indicate another moment to switch S2 and S3 connection, and t
7The moment that indication pair of switches S2 and S3 turn-off.
Fig. 7-the 14th illustrates the circuit diagram of the operation of SRC in each pattern.
Pattern 1: connect switch S1 and charge (with reference to the pattern 1 of figure 5 with S4 and to secondary capacitor
And Fig. 7)
When switch S 1 is connected with S4, flow through elementary winding TX1, resonant capacitor Cr and the switch S 4 of switch S 1, resonant inductor Lr, transformer TX from the resonance current of input voltage source 10.Resonance current is inductor current I
LFlows shown in pattern 1, condenser voltage V simultaneously
cRaise.Load current on the secondary winding TX2 that inducts from elementary winding TX1 is to having the secondary capacitor C2 charging that is parallel-connected to transformer TX of low electric capacity.The interpolation of capacitor C2 makes SRC have the low impedance of combination than rectifier diode RD1-RD4 and load 60.In addition, to make resonance current be induced current I to secondary capacitor C2
LIncrease fast, like Fig. 5 and shown in Figure 6.
Pattern 2: apply electric current (with reference to pattern 2 and Fig. 8 of figure 5) to load
When secondary capacitor C2 charges fully with the load current on the secondary winding TX2 in pattern 1 and becomes when equaling the voltage at secondary winding TX2 two ends, load current flows to load 60 through rectifier diode RD1 and RD4.
Mode 3: switch S 1 is turn-offed and electric current afterflow (with reference to mode 3 and Fig. 9 of figure 5) with S4
When switch S 1 and S4 in pattern 2 when removing gating signal and turn-off; Owing to buffer condenser CS1 and CS4 switch S 1 and S4 separately the voltage at two ends slowly rise, and the buffer condenser CS2 that has charged in the pattern formerly that is connected in parallel with switch S 2 and S3 discharges with CS3.Buffer condenser CS1 and CS4 are with the inductor current I that flows through resonant inductor Lr
LCharging, and the inductor current of afterflow simultaneously I
LIn mode 3, buffer condenser CS2 and CS3 discharge, and buffer condenser CS1 and CS4 charging simultaneously.When the voltage at buffer condenser CS1 and CS4 two ends becomes when equaling voltage source V dc, buffer condenser CS2 and CS3 discharge fully, thereby cause switch S 1 and S4 to turn-off.The energy that in resonant inductor Lr resonant capacitor Cr, gathers allows load current to continue to flow to load 60.
Pattern 4: the conducting of inverse parallel diode D2 and D3 (with reference to pattern 4 and Figure 10 of figure 5)
Because switch S 1 is turn-offed with S4 in mode 3, flow through inverse parallel diode D2 and the D3 that is connected with S3 with switch S 2 so flow through the freewheel current of resonant inductor Lr, and therefore sharply reduce through voltage source.The voltage at switch S 2 and S3 two ends then approaches zero.Correspondingly, wherein in the no-voltage section of electric current through inverse parallel diode D2 and D3 afterflow, promptly wherein inverse parallel diode D2 and D3 connect or the no-voltage section of positive bias in the conducting gating signal apply the zero voltage switch that allows switch S 2 and S3.
Pattern 5: the no-voltage of switch S 2 and S3 and the connection switch of zero current are (with reference to the mould of figure 5
Formula 5 and Figure 11)
The freewheel current that in mode 3, flows through inverse parallel diode D2 and D3 sharply is decreased to zero through voltage source.Then, voltage source allows electric current on opposite direction, to flow, and flows to the resonance current I of resonant inductor Lr through switch S 2 and S3
LOn opposite direction, flow.The secondary capacitor C2 repid discharge of charging in pattern 1 and pattern 2.By with pattern 1 in the direction of opposite current on flow through on the secondary winding TX2 that the primary current of elementary winding TX1 inducts load current with pattern 1 in the direction of opposite current on flow through secondary capacitor C2.Correspondingly, secondary capacitor C2 is charged by load current again, is inductor current I thereby cause resonance current
LIncrease fast, like Fig. 5 and shown in Figure 6.That is, (, the voltage of switch ends be zero) applies the conducting gating signal when inverse parallel diode D2 and D3 conducting, and when the reversing of electric current, with zero current condition switch switch S 2 and S3.Its result is that realization no-voltage and Zero Current Switch condition minimize switching loss thus.
Pattern 6: apply electric current (with reference to pattern 6 and Figure 12 of figure 5) to load
Except when switch S 2 inductor current I when connecting with S3
LFlowing through on the opposite direction outside the resonant inductor Lr resonant capacitor Cr, pattern 6 is identical with pattern 2.When secondary capacitor C2 equaled the secondary voltage on the secondary winding TX2 by charging of the load current on the secondary winding TX2 and voltage fully, load current flowed to load 60 through rectifier diode RD2 and RD3.
Mode 7: switch S 2 is turn-offed and electric current afterflow (with reference to mode 7 and Figure 13 of figure 5) with S3
Except inductor current I
LFlowing through on the opposite direction outside resonant inductor Lr resonant capacitor Cr and switch S 2 and the S3 shutoff, mode 7 is identical with mode 3.Particularly, in mode 7, the voltage at switch S 2 and S3 two ends slowly increases through buffer condenser CS2 and CS3, and the buffer condenser CS1 and the CS4 that formerly charge in the pattern discharge.At this moment, the inductor current that flows through resonant inductor Lr is to buffer condenser CS2 and CS3 charging and by afterflow.When the voltage at buffer condenser CS2 and CS3 two ends becomes when equaling voltage source V dc, buffer condenser CS1 and CS4 discharge fully, and switch S 2 is turn-offed with S3.The energy that in resonant inductor Lr resonant capacitor Cr, gathers allows load current to continue to flow to load 60.
Pattern 8: the conducting of inverse parallel diode D1 and D4 (with reference to pattern 8 and Figure 14 of figure 5)
Except inductor current I
LFlowing through resonant inductor Lr resonant capacitor Cr on the opposite direction and be connected to outside the inverse parallel diode D1 and D4 conducting of switch S 1 and S4, pattern 8 is identical with pattern 4.
Eight patterns have been described during a switch periods.After pattern 8, inductor current I
LReversing, and repeat that switch S wherein 1 is connected with S4 and the pattern 1 (with reference to figure 7) of secondary capacitor C2 charging.After the pattern 1, also repeat pattern 2-pattern 8.
After pattern 8 during recovery pattern 1, when being zero (, when the voltage at switch S 1 and S4 two ends) applies the conducting gating signal when inverse parallel diode D1 and D4 conducting.When the reversing of electric current, with zero current condition diverter switch S1 and S4.Therefore, satisfy no-voltage and Zero Current Switch condition, minimize switching loss thus.
Thus; Because added the secondary capacitor C2 littler than the electric capacity of the resonant capacitor Cr in the LC resonant circuit unit 30 to the secondary winding TX2 of transformer TX, thus in initial level electric current flow through secondary capacitor C2 and unsupported 60 and elementary winding TX1 on resonance current I
LIncrease fast, thereby produce trapezoidal current waveform, as shown in Figure 5.Correspondingly, as shown in Figure 6, under same frequency, this resonance current I
LCan have than having the bigger effective value of sine-shaped conventional resonance current.
In other words, as shown in Figure 6,,, improve efficient thus so the effective value of resonance current increases the amount (' A+C-B ') of shaded area under the same switch frequency because SRC allows resonance current to increase quickly than conventional SRC.
It is 10 times electric capacity of conventional condenser capacitance that the inductor energy of increase area ' C ' allows buffer condenser CS1-CS4 to have, and realizes the no-voltage turn-off criterion thus.
Correspondingly, owing under same frequency, can obtain bigger effectively current value, therefore might improve the efficient of this SRC.In addition, might be through being reduced in the resonance current I under the same load current conditions
LMaximum reduce conduction loss.
In addition, the inductor current I that increases fast through secondary capacitor C2
LThe energy that allows to be stored among the inductor Lr increases.Therefore, might enlarge markedly the electric capacity of the buffer condenser CS1-CS4 that is connected to switch S 1-S4 two ends, the switching loss when reducing switch S 1-S4 shutoff thus.
Correspondingly, carry out be used for when switch S 1-S4 connects with switch S 1-S4 separately the voltage at two ends remain on zero zero voltage switch, reduce switching loss thus.
In conventional SRC, because under no-voltage and zero current condition, connect switch S1-S4, and switch S 1-S4 does not turn-off under no-voltage and zero current condition, so switching loss takes place.But in exemplary SRC, switch S 1-S4 turn-offs under zero voltage condition and connects.
On the other hand, when inverse parallel diode D1-D4 conducting, be not easy to connect switch S1-S4 exactly so that can satisfy the zero current of switch S 1-S4 and the condition of no-voltage.
In application with wide loading range because switch S 1-S4 shutoff and the cycle between the diode D1-D4 conducting separately separately depend on loading condition and change, so be difficult to estimate when to apply the conducting gating signal.
In one embodiment, when the voltage approaches zero at switch S 1-S4 two ends, gate driver automatically applies the conducting gating signal when diode D1-D4 conducting.
Gate driver allows in wide opereating specification, to carry out no-voltage and zero current is connected switch, and also is provided for the dead time compensation of stable operation.
Figure 15 is the circuit diagram of the gate driver among the SRC of one exemplary embodiment according to the present invention.
With reference to Figure 15, gate driver 51 comprises first resistor R 11, capacitor C11, semiconductor switch, second resistor R 12, the 3rd resistor R 13 and the 4th resistor R 14.First resistor R 11 is connected to input 52, and pulse voltage signal inputs to this input 52.Capacitor C11 is parallel-connected to first resistor R 11 at input 52 places, and uses the conducting pulse voltage that applies through input 52 and 53 to charge.This semiconductor switch comprises source electrode, grid and drain electrode, the gate node 55 of output that this source electrode, grid and drain electrode are connected respectively to the output node of input 51, capacitor C11 and are connected to the switch S 1-S4 of switch element 20.Second resistor R 12 is connected between the collector node 54 of first resistor R 11 and output.The 3rd resistor R 13 is connected between the gate node 55 of drain electrode and output of semiconductor switch.The 4th resistor R 14 is connected to input 52 to form conducting path through first resistor R 11 with capacitor C11.
In the circuit of configuration thus, apply to input 52 and 53 as input signal have (+) and (-) polarity pulse voltage signal with the generation gating signal.Figure 16 illustrates the example of the pulse voltage signal that applies through input.
Semiconductor switch can use mos field effect transistor (MOSFET) to realize.When semiconductor applied positive conducting pulse voltage through the input 52 and 53 to gate driver 51 and connects through capacitor C11 is charged, this conducting gating signal provided to connect the switch S 1-S4 (hereinafter being called main switch) of switch element 20 through the gate node 55 of the 3rd resistor R 13 and output.
On the other hand, MOSFET turn-offs when the input 52 and 53 to gate driver 51 applies negative conducting pulse voltage, and wherein electric current flows through the The built-in body diode, thereby main switch S1-S4 is turn-offed.
The conducting (that is, the zero-voltage state of main switch) that gate driver 51 detects the inverse parallel diode D1-D4 that is connected to main switch S1-S4, and when inverse parallel diode D1-D4 conducting, provide the conducting gating signal to connect main switch S1-S4.Therefore, the no-voltage that realizes main switch S1-S4 is connected switch.
In addition, second resistor R 12 and the 4th resistor R 14 are connected in series to first resistor R 11, can flow through in second resistor R 12 and the 4th resistor R 14 so that flow through the electric current of first resistor R 11.This second resistor R 12 can have the resistance bigger than the 4th resistor R 14.
Described in hereinafter; When before the maximum dead time of setting by capacitor C11, realizing the zero voltage condition of main switch S1-S4 (; When being connected to the inverse parallel diode D1-D4 conducting of main switch S1-S4); The electric current that flows through the 4th resistor R 14 can flow through resistor R 12, thereby connects MOSFET.
Operation with gate driver in each pattern of description.
With reference to Figure 15, the input of gate driver 15 connects so that apply pulse voltage through transformer TX11.When the elementary winding to transformer TX11 applied as the pulse voltage signal of input signal with control main switch S1-S4, the secondary pulses voltage signal of inducting from elementary winding on the secondary winding caused gate driver 51 to produce signals so that main switch S1-S4 turns on and off.
In each operator scheme, the input 52 and 53 that applies (+) and (-) voltage will be called pin _ 1 (pin _ 1) and pin _ 2 (pin _ 2) under the situation of not mentioning transformer.
Figure 17 illustrates the gating signal of when pulse voltage signal shown in Figure 16 is used as input signal, exporting from gate driver shown in Figure 15.Figure 18 describes the curve chart of the gating signal of resonance current and SRC with respect to the time.Figure 19-the 22nd illustrates the circuit diagram of the operation of gate driver in each pattern of one exemplary embodiment according to the present invention.
(+) gate voltage applies pattern 1: positive pulse voltage application (with reference to Figure 19)
When the input to gate driver 51 applied positive gate voltage, as shown in figure 16, pin _ 1 52 were positive terminals, and pin _ 2 53 are negative pole ends.Therefore, electric current flows through capacitor C11, first resistor R 11 and the 4th resistor R 14 (high resistance) from pin _ 1 52, and capacitor C11 is recharged.When capacitor C11 continued to be charged to the conducting voltage of P channel mosfet, MOSFET conducting and electric current flowed to the 6th resistor R 16 through MOSFET and the 3rd resistor R 13, thereby connected main switch S1-S4 (with reference to Figure 21).This operator scheme is called maximum dead time pattern.Gate driver 51 is configured to when before the maximum dead time of the following stated, being connected to the inverse parallel diode D1-D4 conducting of main switch S1-S4 (, when realizing the zero voltage condition of main switch), and main switch S1-S4 is connected in the automatic conducting of MOSFET thus.
(+) gate voltage applies pattern 2: the no-voltage on of no-voltage detecting pattern and main switch
Close (with reference to Figure 20 and 21)
Inverse parallel diode D1-D4 conducting before the maximum dead time when capacitor C11 is charged to conducting MOSFET in pattern 1; And during the voltage approaches zero between the collector and emitter of main switch S1-S4; Electric current flows through capacitor C11, resistor R 11 and resistor R 12 (low resistance), as shown in figure 20.At this constantly, be different from maximum dead time pattern, because current flows through resistor R12 and capacitor C11 are connected MOSFET thus by quick charge.As a result, with reference to Figure 21, along with the MOSFET conducting, electric current flows through the 3rd resistor R 13 and the 6th resistor R 16, thereby connects main switch S1-S4.In other words, when the voltage of inverse parallel diode D1-D4 at conducting before of predetermined maximum dead time and main switch S1-S4 two ends equalled zero, main switch S1-S4 had nothing to do and connects automatically in predetermined maximum dead time.Correspondingly, because gate driver 51 switch element 20 to transducer when inverse parallel diode D1-D4 conducting provides the conducting gating signal, connect switch (with reference to the pattern 5 of transducer) so in switch element 20, realize the no-voltage of main switch S1-S4.The on-delay that the step indication that when pulse is risen, generates gating signal shown in Figure 17 is caused by the capacitor C11 through charging.
(-) gate voltage applies pattern 1: shutdown mode
The shutoff of this pattern indication main switch S1-S4.When negative grid voltage shown in Figure 16 was applied to the input of gate driver 51, pin _ 1 52 and pin _ 2 53 were respectively negative pole end and positive terminal.Correspondingly, under the situation that does not have delay, flow through the body diode of the 6th resistor R 16 and MOSFET, thereby turn-off main switch S1-S4 from the electric current of pin _ 2 53.
Therefore, when the voltage approaches zero at main switch S1-S4 two ends and diode D1-D4 conducting, gate driver 51 detects the conducting of diode D1-D4 and automatically applies the grid Continuity signal.As a result, might realize the zero voltage switch of the main switch S1-S4 among the SRC shown in Figure 4.
Like this; Because added the little secondary capacitor of electric capacity than the resonant capacitor in the LC resonant circuit unit to Secondary winding of transformer; So electric current flows through secondary capacitor and resonance current on the unsupported and elementary winding increases fast in initial level, thereby generates trapezoidal current waveform.Therefore, under same frequency, this resonance current can have than have the bigger effective value of conventional resonance current of sinusoidal waveforms.
Therefore, owing to the effective current that under the same switch frequency, increases, this SRC can present the efficient higher than conventional SRC.In addition, it is 10 times electric capacity of conventional condenser capacitance that the inductor energy of increase allows buffer condenser to have, and realizes the no-voltage turn-off criterion thus.
In addition, the effective current value of the increase under same frequency allows SRC to present the efficient of increase.Under same load current conditions, the maximum resonance electric current can be lowered, and reduces conduction loss thus.
In addition, the electric current that increases fast through secondary capacitor increases the energy of in inductor, storing.Therefore, might enlarge markedly the electric capacity of the buffer condenser that is connected to switch, thus the switching loss when reducing switch and turn-offing.
Correspondingly, when switch turn-offs, might realize allowing the voltage of each switch ends to remain on zero zero voltage switch.As a result, might reduce switching loss.
In other words, in conventional SRC, because switch S 1-S4 connects under no-voltage and zero current bar line, and switch S 1-S4 does not turn-off under no-voltage and zero current condition, and therefore switching loss takes place when switch turn-offs.But in exemplary SRC, switch S 1-S4 carries out under zero voltage condition and turn-offs and connect.
In addition, gate driver is simple to operate.That is, for zero current and the no-voltage on-condition of realizing switch, gate driver detects the conducting of inverse parallel diode when the switch ends voltage approaches zero, and when diode current flow, automatically applies the grid Continuity signal.
In addition, this gate driver allows in wide opereating specification, to carry out no-voltage and zero current is connected switch, and also is provided for the dead time compensation of stable operation.
Though in the disclosure, described some embodiment, these embodiment provide as just example for those of ordinary skills, and can to make various modifications and change and not deviate from the spirit and scope of the present invention be conspicuous.Correspondingly, scope of the present invention should only be limited accompanying claims and its equivalents.
Claims (7)
1. efficient series resonant converter comprises:
The switch element that comprises a plurality of switches is used for through the alternation direct voltage direct current (DC) voltage transitions being become to exchange (AC) voltage;
The LC resonant circuit that comprises the resonant inductor resonant capacitor that is connected in series; Said LC resonant circuit is connected to said switch element, and uses the resonance that is generated by said resonant inductor and said resonant capacitor to change the frequency characteristic of transmission from the said alternating voltage of said switch element;
The transformer that comprises elementary winding and secondary winding; Said elementary winding is connected to said LC resonant circuit, and the ratio of the quantity of the number of turn in the quantity of the number of turn in induced potential in the said secondary winding and the said secondary winding and the said elementary winding is proportional;
Be connected to said transformer said secondary winding and with the secondary capacitor of said transformers connected in parallel;
The bridge rectifier that comprises a plurality of rectifier diodes, the alternating voltage that is used for inducting at said secondary winding converts direct voltage to; And
Gate driver, said gate driver detect the conducting of the inverse parallel diode that is connected to said switch and when said inverse parallel diode current flow, export the conducting gating signal to connect said switch.
2. efficient series resonant converter as claimed in claim 1 is characterized in that, said secondary capacitor has the electric capacity littler than said resonant capacitor.
3. efficient series resonant converter as claimed in claim 1 or 2 is characterized in that said secondary capacitor has the electric capacity of the 1/20-1/5 that is said resonant capacitor electric capacity.
4. efficient series resonant converter as claimed in claim 1 or 2; It is characterized in that; The low impedance of circuit that said secondary capacitor can have than be made up of said rectifier diode and load, and the said secondary capacitor that is used in the load current charging of inducting in the said secondary winding causes the resonance current of said LC resonant circuit to increase fast.
5. efficient series resonant converter as claimed in claim 1 is characterized in that, said gate driver comprises:
Be connected to first resistor of input, pulse voltage signal inputs to said input;
Be parallel-connected to said first resistor of said input end and use the capacitor of the conducting pulse voltage charging that applies through said input;
The gate node of output that the semiconductor switch that comprises source electrode, grid and drain electrode, said source electrode, grid and drain electrode are connected respectively to the said output node of said input, said capacitor and are connected to the said switch of said switch element;
The 3rd resistor that between the said gate node of the said drain electrode of said semiconductor switch and said output, connects; And
Be connected to said input to form the 4th resistor of conducting path through said first resistor and said capacitor.
6. efficient series resonant converter as claimed in claim 5 is characterized in that, second resistor and first diode are one another in series and are connected between the collector node of said first resistor and said output.
7. efficient series resonant converter as claimed in claim 6 is characterized in that, said second resistor has than the big resistance of said the 4th resistor.
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| Application Number | Priority Date | Filing Date | Title |
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| KR1020100070290A KR101000561B1 (en) | 2010-07-21 | 2010-07-21 | Series resonant converter |
| KR10-2010-0070290 | 2010-07-21 |
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Cited By (2)
| Publication number | Priority date | Publication date | Assignee | Title |
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| CN105322812A (en) * | 2014-07-04 | 2016-02-10 | 现代自动车株式会社 | Pulse width modulation resonance converter and charger for vehicle using the same |
| CN110582930A (en) * | 2017-03-31 | 2019-12-17 | 森特姆阿德泰尔运输公司 | Hybrid power battery |
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| KR101385979B1 (en) | 2012-11-26 | 2014-04-16 | 한국전기연구원 | Series resonant converter |
| KR101721321B1 (en) * | 2015-01-28 | 2017-03-31 | 단국대학교 산학협력단 | Hybride type LED Power Supply |
| KR102009200B1 (en) | 2016-07-04 | 2019-08-09 | 숭실대학교산학협력단 | Bidirectional Converter with Auxiliary LC Resonant Circuit and Operating Method thereof |
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| CN107528477A (en) * | 2017-08-08 | 2017-12-29 | 西南交通大学 | A kind of quasi-resonance soft switch double-transistor flyback DC/DC converters |
| TWI745800B (en) * | 2018-12-21 | 2021-11-11 | 美商米沃奇電子工具公司 | Battery pack chargers, methods of controlling an operation of a battery pack charger, and charging circuits for a battery pack charger |
| KR102393664B1 (en) | 2020-11-17 | 2022-05-02 | 한국전기연구원 | Inverter switch gate driver and inverter |
| KR20240047770A (en) | 2022-10-05 | 2024-04-12 | 한국전기연구원 | Resonant converter and inverter with soft switching function |
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| KR101000561B1 (en) | 2010-12-14 |
| CN102347695B (en) | 2014-09-17 |
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