CN100553090C - Reduce the method and the circuit of the switching concussion in the switched power supply - Google Patents
Reduce the method and the circuit of the switching concussion in the switched power supply Download PDFInfo
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- CN100553090C CN100553090C CNB2007100878035A CN200710087803A CN100553090C CN 100553090 C CN100553090 C CN 100553090C CN B2007100878035 A CNB2007100878035 A CN B2007100878035A CN 200710087803 A CN200710087803 A CN 200710087803A CN 100553090 C CN100553090 C CN 100553090C
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Abstract
The present invention proposes a kind of method and circuit that reduces the switching concussion in the switched power supply, comprise two transistors in this switched power supply, and this two transistor can not enter closed condition simultaneously.Among the present invention the phase-locked loop can be set further, so that the output frequency of pulse width modulating control circuit equals a setpoint frequency.The present invention and synchronous switched power supplier are compared, and have the advantage of saving energy consumption, and can significantly shorten during the concussion.
Description
Technical field
The present invention relates to control method and the circuit of a kind of switched power supply (switching regulator), be meant a kind of method and circuit that reduces the switching concussion in the switched power supply especially.
Background technology
Switched power supply commonly used comprises voltage-dropping type (Buck), booster type (Booster) and three kinds on back-pressure type (Inverter).At first be illustrated with regard to the voltage-dropping type switched power supply, its circuit structure roughly as shown in Figure 1, voltage-dropping type switched power supply 1 includes two transistor switch Q1, Q2, see through pulse width modulating control circuit 10 and control the Push And Release of this two transistor Q1, Q2, use the magnitude of current and direction on the control inductance L, electric energy is sent to output OUT.Pulse width modulating control circuit 10 receives the feedback voltage that extracts from output, compares with a reference voltage Vref, how to control with decision and switches two transistor Q1, Q2.
In the prior art, early stage switched power supply, be complementary fully the switching time of its two transistor Q1, Q2, be called synchronous switched power supplier again, that is as shown in Figure 2, when transistor Q1 opened, transistor Q2 closed promptly; When transistor Q2 opened, transistor Q1 promptly closed.(in this specification, " unlatching " is meant complete conducting; " close " and be meant and be not conducting fully under the situation of not considering leakage current.) under this kind arrangement, the inductive current amount I of its correspondence
LWith direction shown in the 3rd waveform among the figure, when transistor Q1 open, when transistor Q2 closes, because of the voltage of the input IN voltage greater than output OUT, electric current flows in (among the figure with+the past output direction of expression) toward output OUT, and flow constantly increases; And when transistor Q2 unlatching, when transistor Q1 closes, because of the current potential of inductance left node Phase drops near 0, the voltage of output OUT is greater than the voltage of this node, so electric current trend changes, flow reduces before this, then changes toward flowing (among the figure with-expression opposite direction) in the other direction.
Fig. 3 and Fig. 4 illustrate booster type switched power supply 2 and back-pressure type switched power supply 3 respectively, its mode of operation and aforementioned similar, be by the comparative result of pulse width modulating control circuit 10 equally according to feedback voltage and reference voltage Vref, how decision switches two transistor Q1, Q2, comes the voltage of control output end OUT.Its detailed circuit mode of operation is known by present technique field person, does not repeat them here.
Please return and read Fig. 1 and Fig. 2, this kind switches the arrangement of two transistor Q1, Q2 synchronously, and its shortcoming is, when the inductive current direction when just changeing negative, the expression electric current is by output OUT, and ground connection is run off by the path of inductance L and transistor Q2, that is can lose output OUT energy.
Therefore, at prior art United States Patent (USP) the 6th, in 580, No. 258 cases, a kind of practice is proposed, its main concept as shown in Figure 5, be by suitable oxide-semiconductor control transistors Q1, Q2, make and promptly to close transistor Q2 by just changeing when bearing when the inductive current direction, so promptly do not have energy and run off, can reduce unnecessary consume from output OUT.As shown in FIG., transistor Q1, Q2 have one section time T of closing simultaneously, are called " sleep pattern " (sleep mode).The circuit concept of this case roughly as shown in Figure 6, wherein the signal of inductive current is represented in detecting, and in current comparator ICP, compared, the output of its comparative result and pulse width modulating control circuit 10 is after logical operation, and transistor Q2 is opened or closed to decision whether.
Yet the practice of this kind prior art has its shortcoming.When transistor Q1, Q2 closed simultaneously and enter sleep pattern, the electric current of its reality on inductance L and the voltage at node Phase place were not very desirable waveform, but as shown in Figure 7, when transistor Q1, Q2 close simultaneously, the inductance L electric current I
LNear concussion a little null value, and this moment node Phase place voltage V
PHBe simple harmonic quantity concussion (the damped simple harmonic motion) waveform that is obstructed.During this high frequency oscillation, circuit fails to enter stable state, and will produce the EMI noise of not desired, and is unsatisfactory, so should shorten it.
In view of this, the present invention promptly at the deficiency of above-mentioned prior art, proposes a kind of switched power supply that can address the above problem, with and control circuit and method.
Summary of the invention
The present invention's first purpose is to provide a kind of switched power supply, and itself and synchronous switched power supplier are compared, and has the advantage of saving energy consumption, but compares to the prior art practice shown in Figure 7 with Fig. 5, then can significantly shorten during the concussion.
The present invention's second purpose is to provide a kind of control method in order to the control switched power supply.
For reaching above-mentioned purpose, in one of them embodiment of the present invention, provide a kind of switched power supply, comprise: the first transistor, transistor seconds and inductor are electrically connected on a common node mutually; Pulse width modulating control circuit is in order to produce pulse width modulation signal; Operational amplifier with this common node voltage, is compared with a reference voltage; And multiplex circuit, its first output that is input as this operational amplifier, second is input as the output of this pulse width modulating control circuit, and this multiplex circuit is according to the electric current on this inductor, and one of its input is selected in decision; The grid of the output control transistor seconds of multiplex circuit.
In addition, according to another embodiment of the invention, also provide a kind of switched power supply, comprised: the first transistor, transistor seconds and inductor are electrically connected on a common node mutually; Pulse width modulating control circuit is in order to produce pulse width modulation signal; Operational amplifier will flow through the electric current of this inductor, compare with a reference current; And multiplex circuit, its first output that is input as this operational amplifier, second is input as the output of this pulse width modulating control circuit, and this multiplex circuit is according to the electric current that flows through this inductor, and one of its input is selected in decision; The grid of the output control transistor seconds of multiplex circuit.
In addition, according to another embodiment of the invention, a kind of method that reduces the switching concussion in the switched power supply also is provided, comprise following steps: provide a switched power supply, this switched power supply comprises the first transistor, transistor seconds and inductor, is electrically connected on a common node mutually; Close in the part of period at the first transistor,, compare with a reference voltage with this common node voltage; And according to comparative result, the grid of control transistor seconds, making transistor seconds is the low current circulation status.
Again, according to another embodiment of the invention, a kind of method that reduces the switching concussion in the switched power supply also is provided, comprise following steps: provide a switched power supply, this switched power supply comprises the first transistor, transistor seconds and inductor, is electrically connected on a common node mutually; Close in the part of period at the first transistor,, compare with a reference current with the electric current on this inductor; And according to comparative result, the grid of control transistor seconds, making transistor seconds is the low current circulation status.
In the various embodiments described above, can further provide the phase-locked loop, so that the output frequency of pulse width modulating control circuit equals a setpoint frequency.
Below will illustrate in detail, when the purpose that is easier to understand the present invention, technology contents, characteristics and the effect reached thereof by specific embodiment; Wherein, similar element indicates with identical symbol.
Description of drawings
Fig. 1 is the schematic circuit diagram of the voltage-dropping type switched power supply of prior art.
Fig. 2 is the exemplary waveforms figure of the synchronous switched power supplier of prior art.
Fig. 3 is the schematic circuit diagram of the booster type switched power supply of prior art.
Fig. 4 is the schematic circuit diagram of the back-pressure type switched power supply of prior art.
Fig. 5 is the ideal waveform schematic diagram of the 6th, 580, No. 258 cases of prior art United States Patent (USP).
Fig. 6 is the circuit generalized schematic of the 6th, 580, No. 258 cases of prior art United States Patent (USP).
Fig. 7 is the actual waveform schematic diagram of the 6th, 580, No. 258 cases of prior art United States Patent (USP).
Fig. 8 is the present invention's waveform schematic diagram.
Fig. 9 is the schematic circuit diagram of first embodiment of the invention.
Figure 10 is the waveform schematic diagram of Fig. 9 embodiment.
Figure 11 is the schematic circuit diagram of second embodiment of the invention.
Figure 12 is the waveform schematic diagram of Figure 11 embodiment.
Figure 13 is the schematic circuit diagram of third embodiment of the invention.
Figure 14 is the waveform schematic diagram of Figure 13 embodiment.
Figure 15 illustrates the circuit structure of phase-locked loop.
Figure 16, Figure 17, Figure 18 are the schematic circuit diagram of other three embodiment of the present invention.
Symbol description among the figure
1 voltage-dropping type switched power supply
2 booster type switched power supplies
3 back-pressure type switched power supplies
10 pulse width modulating control circuits 10
11 voltage-dropping type switched power supplies
30,31,32 multiplex circuits
41,42 logical circuits
50 phase-locked loops
51,52 fixed pulse width produce circuit
53,54 low pass filters
55 subtracters
The CP comparator
The ICP current comparator
The Iofs reference current
The Iref reference current
The IN input
L inductance (inductance value)
The OP operational amplifier
The OUT output
Q1, the Q2 transistor
The Vofs voltage source
Embodiment
Main concept of the present invention is not make transistor Q1, Q2 to close simultaneously; Electric current I on inductance L
LBe about to when just changeing negative, and not exclusively close transistor Q2, but it is in " weak conducting " state, allow that the electric current of low discharge passes through.So, compare with the prior art practice shown in Figure 2, the present invention still has the high efficiency advantage of saving energy consumption, but compares to the prior art practice shown in Figure 7 with Fig. 5, and then the present invention can significantly shorten the concussion time.
Above notion please refer to Fig. 8, and contrast Fig. 5 and Fig. 7, when being easier to understand.In the prior art, the role of transistor Q2 only is a switch, and therefore standard-sized sheet and full cut-off two states are only arranged.When in order to save energy consumption, it is aforementioned that transistor Q1, Q2 are entered " sleep pattern " time, transistor Q1, Q2 close simultaneously.But according to the present invention, then also it doesn't matter " sleep pattern "; In Fig. 8, the electric current I on inductance L
LIs about to when just changeing negative, and not exclusively closes transistor Q2, but among period T, Q2 converts to transistor " weak conducting " state, allow that the electric current of low discharge passes through.To this, as shown in the figure, three kinds of practices can be arranged, first kind of practice is to make transistor Q2 except conducting, all is in low current condition, as shown in first kind of waveform, transistor Q2 only comprises standard-sized sheet, low current two states; Or, as shown in second kind of waveform, make transistor Q2 when transistor Q1 conducting, still close fully, and only among period T, convert transistor Q2 to low current condition, so then transistor Q2 comprises standard-sized sheet, full cut-off, three kinds of states of low current; The third practice is that the low current condition that makes transistor Q2 is allowed when initial and reduced subsequently than high-amperage, even also the magnitude of current at switching initial stage be a height than the later stage.First kind circuit complexity is lower, and second kind is better on energy-saving effect, the third quicker elimination concussion, and the three respectively has quality, belongs to category of the present invention.
Be familiar with present technique person when finding immediately that more than transistor Q1, the Q2 in the explanation is to be example with NMOS.Certainly, transistor Q1, Q2 also can change with PMOS individually and make, and the oscillogram of its correspondence is from also different, but do not break away from notion of the present invention.
Please contrast Fig. 8 and Fig. 7 again, under above-mentioned arrangement of the present invention, when transistor Q1 closes and transistor Q2 in low current condition the time, that is in the drawings among the period T, the voltage V at node Phase place
PHThough be the simple harmonic quantity concussion waveform that is obstructed equally, its concussion decay fast more promptly arrives plateau far beyond prior art.It should be noted that, clear and definite for asking figure, be to illustrate the concussion waveform among the figure with the figure of comparatively exaggerating, and not fully not proportionally.In real conditions, see through the appropriate design of aftermentioned embodiment, can make the concussion shared period of waveform lower than icon.
Described " low current ", according to the present invention, be meant to be 1 μ A (micromicroampere) or more than it, but below the magnitude of current (not containing) of the complete conducting of transistor Q2, the magnitude of current in this scope.It should be noted that in addition though in the period T in Fig. 8, the grid-control voltage of transistor Q2 is to illustrate to be straight line, the present invention is not limited thereto; In period T, the grid-control voltage of transistor Q2 can be for changing waveform arbitrarily, only need its corresponding magnitude of current that produces, meet above-mentioned condition and get final product.
The mode of reaching of the waveform among above-mentioned Fig. 8 according to the present invention, has the multiple practice, and its first physical circuit embodiment please refer to Fig. 9.Present embodiment is to be example with the voltage-dropping type switched power supply, as shown in the figure, in voltage-dropping type switched power supply 11 of the present invention, except up and down bridge transistor switch Q1, Q2, inductance L, current comparator ICP, pulse width modulating control circuit 10, other includes an operational amplifier OP.The voltage at one of input of operational amplifier OP recipient node Phase place, another input is accepted the voltage a little less than output node voltage Vout.Represent that with voltage source V ofs the voltage that operational amplifier OP is received equals Vout-Vofs among the figure, but it should be noted that icon only is signal, do not represent to be provided with the voltage source V ofs of an entity; For example, can reach identical functions by the internal bias voltage value of operational amplifier OP two inputs.As the voltage source V ofs of entity is set, then this entity voltage source can be resistance, various diode, or the like.After operational amplifier OP compares the voltage of two inputs, according to comparative result, output signal L1.
The grid of transistor Q2 is subjected to multiplex circuit 30 control, decides transistor Q2 to be controlled by the output L2 of pulse width modulating control circuit 10, the output L1 of exclusive disjunction amplifier OP by the output of this multiplex circuit 30.When transistor Q2 was controlled by signal L2, its role was a switch, and when transistor Q2 is controlled by signal L1, then may be the low current condition of weak conducting.
The operation of foregoing circuit sees also Figure 10, understands when easier.The waveform of signal L1 at first is described, when transistor Q1 conducting and transistor Q2 when closing, the voltage at node Phase place equals the voltage of input; When transistor Q1 closes and during transistor Q2 conducting, the voltage at node Phase place equals 0; When a little less than the transistor Q2 during conducting, to there be micro-current after output OUT flows through inductance L and transistor Q2, to lead ground, right-hand voltage of expression inductance L is slightly larger than left, so the voltage at node Phase place is lower than output node voltage Vout slightly, through the feedback control mechanism of circuit, the magnitude of voltage that equals Vout-Vofs will be equilibrated at.By the circuit design of operational amplifier OP, in the time of can equaling Vout-Vofs at the voltage at node Phase place, make the weak conducting of transistor Q2.Therefore signal L1 is the syllogic waveform, as figure.Signal L2 is the output of pulse width modulating control circuit 10, is simple impulse waveform.Signal ZD is the output signal of current comparator ICP, contrast inductive current I
LWaveform, as inductive current I
LDuring greater than reference level Iref (reference level can be 0 or a little less than 0, decides on the reference input design of current comparator ICP), signal ZD is a high levels, as inductive current I
LWhen being equal to or less than reference level, signal ZD is a low level.(current comparator ICP can be a hysteresis comparator, shakes noise a little to filter.) when signal ZD was high levels, multiplex circuit 30 was selected signal L2; When signal ZD was low level, multiplex circuit 30 was selected signal L1; Therefore, the grid signal of transistor Q2 is as among the figure shown in the below.
The embodiment of Fig. 9 is not unique execution mode of the present invention.Please refer to Figure 11, another kind of multi-channel approach is shown, wherein it should be noted that, in the present embodiment, pulse width modulating control circuit 10 only need produce the required switch pulse signal of transistor Q1 grid, do not need to produce separately different pulse signals and come oxide-semiconductor control transistors Q2, only need utilize the anti-phase signal of this pulse signal to get final product, so the internal circuit configuration of pulse width modulating control circuit 10 is comparatively simplified.
In detail, please contrast and consult Figure 11 and Figure 12, effect because of NOR gate 41, only have when the output signal ZD of the output signal (hereinafter to be referred as the PWM signal) of 10 couples of transistor Q1 of pulse width modulating control circuit grid and current comparator ICP is all low level, the output SWO of NOR gate 41 just is a high levels (1), corresponding to the period T among Figure 12; See through multiplex circuit 31 and 32 effect this moment, the positive-negative input end of operational amplifier OP is selected the voltage and the Vout-Vofs at node Phase place respectively, so make conducting a little less than the transistor Q2.On the other hand, in the time of except that above-mentioned other, the positive-negative input end of operational amplifier OP is respectively anti-phase signal of PWM and PWM signal, that is the outputting level of operational amplifier OP is followed the anti-phase signal of PWM.So the grid signal waveform of transistor Q2 as shown in the figure.
Except that the 9th, 11 figure illustrated embodiment, Figure 13 marks another embodiment of the present invention, and the characteristics of present embodiment are to control the output pulse frequency of pulse width modulating control circuit 10, to reach effects such as for example avoiding audio zone, scramble.
In detail, present embodiment and Fig. 9 embodiment different are in being provided with a phase-locked loop (PLL) 50, and the voltage source V ofs of pressure reduction is provided is variable voltage source, controlled by the output signal of phase-locked loop 50.Please contrast and consult Figure 13 and Figure 14 (for simplifying drawing, omitting the concussion noise among the figure), suppose that desire is first waveform with the frequency setting of PWM signal, therefore be the frequency setting input of phase-locked loop 50 with this waveform.When the frequency of PWM signal is lower than setpoint frequency, if the required voltage Vout of output do not change, can increase by node Phase by the electric current that transistor Q2 leads ground, so that the voltage at node Phase place descends, so can force the rising of PWM signal frequency.In other words, when the frequency of PWM signal is lower than setpoint frequency, can make the output of phase-locked loop 50 heighten the voltage of variable voltage source Vofs, Vout-Vofs is descended, see through the effect of operational amplifier OP, will make the more conductings of transistor Q2, the voltage at the node Phase place of leaving behind, finally reach balance, make the frequency of PWM signal equal setpoint frequency.
Using identical notion, in the embodiment shown in fig. 11, add phase-locked loop 50, also is feasible, is familiar with present technique person when analogizing, and its circuit structure and signal waveform will not be given unnecessary details.
The concrete practice as for phase-locked loop 50, numerous embodiments can be arranged, take a single example at this, see also Figure 15, can make setpoint frequency signal and PWM signal produce circuit 51 and 52 by fixed pulse width respectively, produce in the circuit in fixed pulse width, triggering edge according to input signal, produce the pulse signal T1 and the T2 of fixed pulse width, make pulse signal T1 and T2 again, pulse signal T1 is become analog signal S1 and S2 with the frequency inverted of T2 respectively by low pass filter 53 and 54.Subtracter 55 subtracts each other analog signal S1 and S2, and its output can be in order to adjust variable voltage source Vofs.
See also Figure 16, this is an another embodiment of the present invention, except that the voltage that uses node Phase place compares, and also can be according to inductive current I
LCome the state of oxide-semiconductor control transistors Q2, as shown in the figure, present embodiment is with inductive current I
LCompare with reference current Iofs (in the side circuit, is with representative inductive current I
LVoltage signal compare with the voltage signal of representing reference current Iofs, so difference is a voltage signal), and see through operational amplifier OP, according to this comparative result, and produce the analog signal of the weak conducting state of oxide-semiconductor control transistors Q2.In detail, when signal ZD was high levels, multiplex circuit 30 was selected L2, and transistor Q2's is condition controlled in pulse width modulating control circuit 10 at this moment, is standard-sized sheet or complete shut-down two states; When signal ZD was low level, multiplex circuit 30 was selected L1, and transistor Q2's is condition controlled in the output of operational amplifier OP at this moment, is weak conducting state.Under the latter instance, see through feedback control mechanism, can make inductive current I
LIofs equates with reference current, in other words can when transistor Q1 closes transistor Q2 be controlled at weak conducting state, and the magnitude of current of controlling on it equals Iofs.
With Figure 13 similarly, phase-locked loop 50 can be set in the above-described embodiments, output pulse frequency with control pulse width modulating control circuit 10, its circuit for example can be with reference to Figure 17, the output of mat phase-locked loop 50, control variable reference current Iofs, that is adjust the weak conducting state of transistor Q2 and the magnitude of current on it.
The present invention also can do following variation under same concept except that the above.In above each embodiment, all be to use operational amplifier OP, directly use the grid of analog form oxide-semiconductor control transistors Q2, but the also available comparator C P of operational amplifier OP replaces.For example, Fig. 9 embodiment can be changed to Figure 18, and wherein when the voltage at node Phase place was lower than Vout-Vofs, the output of comparator C P was charged to capacitor C.When signal ZD selects L1, voltage when node Phase place equals Vout-Vofs, this moment, comparator C P was output as low level, but the grid of transistor Q2 system is controlled by the magnitude of voltage of capacitor C, and this magnitude of voltage can be set for and allows the weak conducting of transistor Q2, so, can reach purpose of the present invention equally.In like manner, the operational amplifier OP among the above all embodiment all can use the similar manner replacement, does not give unnecessary details in addition.
More than multiplex circuit 30,31,32 among each embodiment, and need not be a grid circuit, and can only be a node, as long as chosen input has the ability to cover (override) another input, get final product.
More than be example with the voltage-dropping type switched power supply, narrated notion of the present invention.Same notion also can be applicable to booster type switched power supply and back-pressure type switched power supply, but needs the reference input slightly modified with operational amplifier OP; Be familiar with present technique person and learn, seldom give at this and giving unnecessary details when analogizing.
Below at preferred embodiment the present invention is described, the above person only only is to make to be familiar with present technique person and to be easy to understand content of the present invention, is not the interest field that is used for limiting the present invention.For being familiar with present technique person, when can in spirit of the present invention, thinking immediately and various equivalence variation; For example, shown among each embodiment, be for example in the dividing potential drop mode, from output extraction feedback voltage signal, compare for pulse width modulating control circuit 10 and reference voltage Vref, but the mode of extraction feedback signal is not limited thereto.And for example, shown between each element, add the circuit element that does not influence the signal meaning, as delay circuit, drive grid or the like, do not influence spirit of the present invention.For another example, shown in transistor Q1 and Q2, can be incorporated within the integrated circuit, also can be arranged on outside the integrated circuit.And for example, the embodiment of Figure 16 also can be with reference to the embodiment of Figure 11, and does corresponding variation.So all a notion and spirit impartial for it a variation or modification according to the present invention all should be included in the present invention's the claim.
Claims (8)
1. method that reduces the switching concussion in the switched power supply comprises following steps:
A switched power supply is provided, and this switched power supply comprises the first transistor, transistor seconds, inductor, is electrically connected on a common node mutually;
Close in the part of period at the first transistor,, compare with a reference voltage with this common node voltage; And
According to comparative result, the grid of control transistor seconds, making transistor seconds is the low current circulation status.
2. the method for claim 1, wherein this low current circulation status is high at the magnitude of current at switching initial stage than the later stage.
3. the method for claim 1 more comprises: adjust this common node voltage, so that the switching frequency of the first transistor equals a setpoint frequency.
4. method as claimed in claim 3, wherein the step of this adjustment common node voltage comprises:
A phase-locked loop is provided;
In the phase-locked loop, this setpoint frequency is compared with the switching frequency of the first transistor;
According to comparative result, control above-mentioned reference voltage; And
The low current circulation status of control transistor seconds is to adjust this common node voltage.
5. method that reduces the switching concussion in the switched power supply comprises following steps:
A switched power supply is provided, and this switched power supply comprises the first transistor, transistor seconds, inductor, is electrically connected on a common node mutually;
Close in the part of period at the first transistor,, compare with a reference current with the electric current on this inductor; And
According to comparative result, the grid of control transistor seconds, making transistor seconds is the low current circulation status.
6. method as claimed in claim 5, wherein this low current circulation status is high at the magnitude of current at switching initial stage than the later stage.
7. method as claimed in claim 5 more comprises: adjust the electric current on this inductor, so that the switching frequency of the first transistor equals a setpoint frequency.
8. method as claimed in claim 7, wherein the step of the electric current on this adjustment inductor comprises:
A phase-locked loop is provided;
In the phase-locked loop, this setpoint frequency is compared with the switching frequency of the first transistor;
According to comparative result, control above-mentioned reference electric current; And
The low current circulation status of control transistor seconds is to adjust the electric current on this inductor.
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