CH708498B1 - Device and method for galvanic isolation and voltage balancing in a bidirectional DC-DC converter with high-frequency AC-AC transformer. - Google Patents

Abstract

The inventive DC-DC converter for electrical power transmission from a DC primary side to a DC secondary side comprises: • a 4-pole AC-AC transformer (4) for electrically isolated power transmission, • a primary-side bridge circuit, which is a primary side of the 4-pole AC-AC transformer (4) to the DC primary side connects, • an upper primary-side DC link capacitor (7) and a lower primary-side DC link capacitor (8), which are connected in series and together a voltage (U 1 (t)) of the DC voltage Primary side, • a secondary-side bridge circuit, which connects a secondary side of the 4-pole AC-AC transformer (4) to the DC secondary side, • an upper secondary side DC link capacitor (9) and a lower secondary side DC link capacitor (10) connected in series are switched and together a voltage (U 2 (t)) of the Support DC secondary side, • a device for balancing the voltage of the two primary-side DC link capacitors (7, 8) or the two secondary-side DC link capacitors (9, 10).

Description

Description: The invention relates to the field of power electronic circuits and relates to a bidirectional DC-DC converter with high-frequency AC-AC transformer, and to a method for its control.

In a bidirectional, voltage-fed 3- to 3/4-phase AC-AC converter (AC-AC Voltage Source Converter [VSC]), which is supplied by a network with low-impedance connection to a protective conductor and which a load fed with low-resistance protective conductor connection, circulating currents can form that close over the protective conductor. In order to prevent these circulating currents, galvanic isolation (also called potential separation) between the input and output sides of the AC-AC converter is typically provided. This galvanic isolation is achieved by a transformer. In addition, the use of the transformer can also prevent circulating currents between parallel-connected voltage-controlled AC-AC converters whose phases are connected to each other at the input and at the output of the converters. To save weight and to allow a compact design of the converter, the galvanic isolation, instead of using a mains frequency transformer at the line side input of the AC-AC converter, by installing a DC-DC converter with high-frequency transformer between two separate voltage intermediate circuits can be achieved become. From one DC link voltage, an AC voltage is generated by a half bridge or a full bridge; the other DC link voltage is generated by a rectifying half bridge or full bridge of an AC voltage. This results in an AC-DC-DC-AC converter structure as schematically shown in Fig. 4 (a).

The DC-DC converter can be realized as a resonance converter or as a dual active bridge converter (DAB converter). The half bridges and the full bridges can be implemented as 2-level, 3-level or multi-level switching structures. By increasing the number of voltage levels, it is possible to use power semiconductors with a lower blocking voltage resistance, which have lower switching losses and thus permit a higher switching frequency of the bridges. By choosing a higher switching frequency, the construction volume of the inductive components, in this case the transformer can be reduced. Active bridges are required at the input and output of the DC-DC converter if bi-directional power flow is to be possible. Typically, the active bridges of the resonant converter and the DAB converter are driven with a duty cycle close to 50%. In A. Esser and T. Kalker, "Contactless power supply of decentralized drives in handling equipment", etzArchiv Bd. 12 H. 8, 1990, the realization of a DC-DC converter by a DAB and a resonant converter is described. The DC-DC converter has a 2-level full bridge at the input and at the output. As a result, the contactless energy transfer for a drive is achieved. Such a resonance converter, each with two 2-level full bridges, is shown by way of example in FIG. 4 (b).

In contrast, EP 2 495 858 A1 shows possible embodiments of bidirectional DC-DC converters with galvanic isolation. Among other things, a realization of the isolated DC-DC converter is shown as a DAB converter, wherein each input and output side each a star point-clamped (English: Neutral Point Clamped [NPC]) 3-level half-bridge is used. Furthermore, CH 705 817 A2 discloses a DAB and a resonance converter which additionally have a saturable coil in order to achieve zero-current switching. The converters are each implemented with two neutral-clamped 3-level half-bridges.

In multi-level switching structures with three or more voltage levels and several DC link capacitors are used, both on the input and on the output side of the DC-DC converter. When supplying any load which leads to different loads of the output phases, the different load of the output phases is reflected in an unequal load of the DC link capacitors. Therefore, the voltages across the DC link capacitors with an additional circuit, which will be referred to as a voltage equalizer hereinafter, must be maintained at the same value. For a 3-level switching structure with DC link center tap, this voltage balancer, as shown in Fig. 4 (c), consist of a 2-level half-bridge whose output is connected via a balancing coil 51 to the center of the DC link capacitors is. In steady state and with slight differences in the DC link voltages, the half-bridge is driven with a duty cycle of close to 50%. H. Ertl, J.W. Kolar and F.C. Zach, "Analysis of an Extended DC-DC Flyback Converter with Inherent Input Voltage Symmetrization", in Proc. of the 35th International Conference on Power Conversion, Nuremberg, Germany, May 26-28, 1998, shows a voltage balancer as described to achieve equal voltage sharing across two series-connected DC link capacitors.

The DC-DC converter for galvanic isolation and the voltage balancers require a variety of power semiconductors, a high-frequency transformer and a balancing coil. The increased number of power semiconductors increases the cost of the converter and causes additional power dissipation. In addition, inductive components such as transformer and inductance make up a large part of the total volume of the converter and thus make the compact design of the converter more difficult.

The object of the invention is to provide a DC-DC converter, which allows an asymmetric load connected in series secondary side DC link capacitors. The converter should ensure a potential separation. Optionally, it should also allow bidirectional power flow.

The object is achieved by a DC-DC converter according to claim 1.

Thus, the DC-DC converter is used for electrical power transmission from a DC primary side to a DC secondary side. It indicates: • a 4-pole AC-AC transformer for potential-free power transmission, • a primary-side bridge circuit connecting one primary side of the 4-pole AC-AC transformer to the DC primary side, • an upper primary-side DC link capacitor and a lower primary-side DC link capacitor a secondary-side bridge circuit connecting a secondary side of the 4-pole AC-AC transformer to the DC secondary side, a lower secondary-side link capacitor, which are connected in series and together support a voltage U2 (t) of the DC secondary side, wherein the DC-DC converter means for balancing the voltage of the two primary-side DC link capacitors or the two se Having secondary circuit capacitors on the customer side.

In this case, the device may have an inductance for balancing the voltage, which builds up in the presence of an asymmetry of the voltage of the two primary-side DC link capacitors or the two secondary-side DC link capacitors, and counteracts this current asymmetry.

The inductance for balancing can • be formed by the magnetizing inductance of a transformer of the 4-pole AC-AC transformer. In this case, the magnetizing inductance act as energy storage of the transformer, wherein the magnetic relative permeability of the core material of the transformer is chosen so low that a current that occurs in the presence of asymmetry of the voltages across the primary-side or secondary-side DC link capacitors and acts as a magnetizing current of the transformer, not leads to saturation of the transformer. • be formed by an air gap in the magnetic core of the transformer or by using a powder core material. • be a balancing coil that is connected in parallel with a primary-side or a secondary-side connection pair of the 4-pole AC-AC transformer. • be distributed to the magnetizing inductance of the transformer and such a balancing coil.

A first aspect of the invention is thus: The primary-side bridge of the DC-DC converter is also used for the balancing of the primary side DC link voltages. As a result, less power semiconductors must be used. Furthermore, the functionality of the balun can be partially or completely built into the transformer, e.g. by adding an air gap in the transformer core or by using a powder core material for the construction of the transformer. As a result, a more compact design of the balancing coil is achieved (it does not have to take over the entire compensation current).

This is possible because typically the resonance converter resp. the DAB converter and the voltage symmetrizer are driven with a duty cycle close to 50%.

A primary-side full-bridge circuit and a secondary-side half-bridge circuit allow the same functionality, but use twice the number of power semiconductors on the primary side (assuming the same bridge topologies).

Another aspect of the invention is thus: A realization of the circuit according to the first aspect leads to unequal primary side and secondary side DC link voltages. The voltage difference can be influenced by the choice of system components. By means of additional control, the two primary-side or the two secondary-side DC link voltages can be regulated to the same value.

The dynamics of the AC-AC transformer should be much larger than those of the DC link.

In summary: About the magnetization inductance of the transformer, a mean current is constructed so that the primary current of the AC-AC transformer is averaging. That the transformer assumes the function of

Voltage balanced and galvanic isolation. The balancing coil ensures that the difference between the two primary-side voltages does not increase over time.

In the following, the subject invention based on preferred embodiments, which are illustrated in the accompanying drawings, explained in more detail. Each show schematically:

Fig. 1 (a) A first circuit embodiment of the invention, wherein the primary-side and secondary-side active bridges are realized with a star point-clamped 3-level topology and wherein the energy transfer is based on the concept of a resonance converter. In addition, a balancing coil is used to balance the DC link voltages.

Fig. 1 (b) to Fig. 1 (a) belonging primary side voltage, primary side current, secondary side voltage, secondary side current and voltage across the primary side resonant capacitor for a lower than upper load current.

Fig. 1 (c) A further circuit embodiment of the invention, based on Fig. 1 (a), wherein parallel to the balancing coil, a damping coil and a damping resistor are present.

Fig. 1 (d) Upper and lower primary side intermediate circuit voltages and upper and lower secondary side intermediate circuit voltage for the circuit of FIG. 1 (c) and for a larger middle lower than upper load current. The effects on the intermediate circuit voltages of a sudden increase in the mean value of the lower load current are also shown.

Fig. 1 (e) A further circuit embodiment of the invention, wherein the primary-side and secondary-side active bridges are realized with a 2-level topology and wherein the energy transfer is based on the concept of a resonance converter.

FIG. 2 (a) shows a control method or a controller structure with which the upper and lower primary-side intermediate circuit voltages can be regulated to the same values; FIG. or with which the upper and lower secondary-side intermediate circuit voltage can be regulated to the same values. The control method is applicable to the circuits shown in Figs. 1 (a), 1 (c) and 1 (e).

Fig. 2 (b) Upper and lower primary side intermediate circuit voltages and upper and lower secondary side intermediate circuit voltage for the circuit according to FIG. 1 (c) and for a larger middle lower than upper load current. The upper and lower secondary-side intermediate circuit voltages were regulated to the same values by the control method described in FIG. 2 (a). The effects on the intermediate circuit voltages of a sudden increase in the mean value of the lower load current are also shown.

3 (a) shows a further circuit embodiment of the invention, the primary-side and secondary-side active bridges being realized with a star-point clamped 3-level topology, and the energy transmission being based on the concept of a dual active bridge converter (DAB concept). verter).

FIG. 3 (b) Primary-side voltage, primary-side current, secondary-side voltage and secondary-side current belonging to FIG. 3 (a) for a greater middle lower than upper load current.

Fig. 4 (a) - (c) Structures known in the art.

Basically, the same or similar parts are provided with the same reference numerals in the figures.

A circuit engineering embodiment of the invention is shown in Fig. 1 (a): A DC-DC converter is shown, which connects a voltage U ^ t) with a voltage U2 (t). The converter has a primary-side intermediate circuit 1, a primary-side active bridge 2, a balancing coil 3, a 4-pole AC-AC transformer 4, a secondary-side active bridge 5 and a secondary-side intermediate circuit 6. The primary-side intermediate circuit 1 has an upper primary side DC link capacitor 7 and a lower primary side DC link capacitor 8, which are connected in series and together support the voltage U ^ t). The secondary-side intermediate circuit 6 has an upper secondary-side intermediate circuit capacitor 9 and a lower secondary-side intermediate circuit capacitor 10, which are connected in series and together support the voltage U2 (t). The balancing coil 3 and the 4-pole AC-AC transformer 4 are connected at a first pole to the output of the primary-side active bridge 2 and at a second pole to the center of the primary-side DC link 1. The 4-pole AC-AC transformer 4 is connected to a third pole to the output of the secondary-side active bridge 5, with a fourth pole to the center of the secondary-side intermediate circuit. The 4-pole AC-AC transformer represents a resonant converter and the active bridges 2, 5 are realized with a 2-level or 3-level topology. Representing the various known topologies of this type, a star point clamped 3-level topology is shown in Figure 1 (a).

The 4-pole AC-AC transformer 4 has a series connection of a resonance capacitor 11, an additional coil 12 and a transformer 13. The auxiliary coil 12 is disposed in Fig. 1 (a) on the primary side of the transformer 13, and it could equally be on the secondary side. The "primary side" is that side of the DC-DC converter on which the resonance capacitor 11 is located. Thus, on the primary side, an upper primary-side intermediate circuit capacitor 7 is loaded identically on average as a lower primary-side intermediate circuit capacitor 8. Load refers to the reference of energy. On the secondary side, a load of an upper secondary-side intermediate circuit capacitor 9, which may be different from the load of a lower secondary-side intermediate circuit capacitor 10.

In order to reduce switching losses in the primary-side active bridge 2 and the secondary-side active bridge 5, for a power flow from the primary side to the secondary side of a DC-DC converter, as shown in Fig. 1 (a), the primary-side bridge 2 is active while the secondary-side bridge 5 is used as a passive diode rectifier. For a power flow from the secondary side to the primary side of the DC-DC converter, the secondary-side bridge 5 is activated accordingly and the primary-side bridge 2 is used as a passive diode rectifier. It should be noted that for the second case of the power flow from the secondary side to the primary side, a balancing coil 3 is not absolutely necessary for the desired functionality of the DC-DC converter and therefore can also be omitted.

In the following, the operation of the circuit in Fig. 1 (a) is explained by means of a positive power flow from the primary side to the secondary side of a DC-DC converter, i. the primary-side bridge 2 is actively operated and the secondary-side bridge 5 is used as a passive diode rectifier. From this, the behavior of the DC-DC converter for other operating cases can be derived.

From the primary-side intermediate circuit 1, a rectangular alternating voltage up (t) is generated by means of the primary-side bridge 2, which rests against the balancing coil 3 and on the primary side of the 4-pole AC-AC converter 4 (solid line). From the secondary-side intermediate circuit 6, a rectangular AC voltage us (t) is generated by means of the secondary-side bridge 5, which is applied to the secondary side of the 4-pole AC-AC converter 4 (dotted line). The two voltages up (t) and us (t) are for a stationary state and for a larger mean value of a lower load current iL, 2 (t) compared to the mean value of an upper load current iL, i (t) in FIG. 2 (b). shown. An upper secondary-side intermediate circuit capacitor 9 is loaded with the upper load current iu1 (t), while a lower secondary-side intermediate circuit capacitor 10 is loaded with the lower load current iU2 (t).

By applying a positive voltage UPi1 (t) at the output of the primary-side bridge 2, a resonant circuit is positively excited and therefore forms a positive current pulse of the secondary side current is (t). The resonant circuit is formed by the primary-side resonant capacitor 11, the auxiliary coil 12 and the stray inductance of a transformer 13. Due to the rectifying action of the secondary-side bridge 5, this positive current pulse supplies the upper secondary-side intermediate circuit capacitor 9. The passive diode rectification of the secondary-side bridge 5 prevents the sign of the positive current pulse of the secondary-side current from changing. Thus, the secondary side current remains zero after reaching its zero crossing. The secondary-side current is (t) thus performs half a vibration with a slightly different from the resonant circuit period of the resonant circuit period. So that the positive current pulse reaches zero before a negative voltage -Up, 2 (t) is applied to the output of the primary-side bridge 2, the resonant frequency of the resonant circuit is slightly higher than the clock frequency at which the primary-side bridge 2 is a positive ( or negative) voltage to its output. Now, if the secondary-side current is zero, then by applying a negative voltage -Up, 2 (t) to the output of the primary-side bridge 2, a negative current pulse of the primary side current is (t). This negative current pulse of the secondary-side current is (t) reached before the next positive current pulse is fired, due to the diode rectification of the secondary-side bridge 5 is zero.

In the steady state, an upper voltage difference between the upper primary side voltage Up, i (t) and an upper secondary side voltage Us, i (t). The mean value of the positive current pulse of the secondary side current is (t) is proportional to an upper excitation voltage ures, e, P (t), with which the positive current pulse is excited. This upper excitation voltage is at the time in which a positive voltage is switched to the output of a primary-side bridge 2 equal to ures, e, p (t) = UPi1 (t) - uc, res (t) - Up, 2 (t ). Therefore, the upper excitation voltage ures, e, P (t) is directly proportional to the upper voltage difference. Consequently, in the stationary state, the upper excitation voltage ures, e, P (t) and the upper voltage difference at the time in which a positive voltage is applied to the output of the primary-side bridge 2 have such values that the positive current pulse has an average value, which is equal to the average value of an upper load current iL, i (t). The same applies to a lower voltage difference between a lower primary-side voltage Up, 2 (t) and a lower secondary-side voltage Us, 2 (t).

Since typically for a compact design of the transformer 13, the clock frequency high (eg> 10 kHz) is selected and the resonant frequency of the resonant circuit formed by the primary-side resonant capacitor 11, the additional coil 12 and the leakage inductance of the transformer 13, is above the clock frequency , the resonance capacitor 11 generally has a capacitance value smaller by factors than those of the upper primary side intermediate circuit capacitor 7, the lower primary side intermediate circuit capacitor 8, the upper secondary side intermediate circuit capacitor 9 and the lower secondary side intermediate circuit capacitor 10. Thus, for the following considerations, the dynamics of the 4-pole AC-AC transformer 4 can be considered to be faster than those of the primary DC link 1 and those of the secondary DC link 6 by factors. Therefore, for the time during which, for example, a voltage uc, res (t) across the primary-side resonant capacitor 11 changes, the upper primary-side voltage Up> 1 (t) above the upper primary-side intermediate circuit capacitor 7, the lower primary-side voltage Up, 2 (FIG. t) over the lower primary side DC link capacitor 8, an upper secondary side voltage Us, i (t) above the upper secondary side DC link capacitor 9 and a lower secondary side voltage Us, 2 (t) above the lower secondary side DC link capacitor 10 are considered to be constant.

The activation of the primary-side bridge 2 takes place with a duty cycle of substantially 50%, i. the power semiconductors of the primary-side bridge are switched in such a way that the time duration during which the positive voltage Up, i (t) is present at the output of the primary-side bridge 2, with the time duration during which the negative voltage -Up, 2 (t) at the output the primary side bridge 2 is applied, is substantially identical. The period during which the positive and the negative voltage applied corresponds to half the clock period minus the time, while a zero voltage at the output of the primary-side bridge 2 is applied. For star point-clamped 3-level topologies, such as those in Fig. 1 (a), the switching from positive or negative to negative or positive voltage takes place at the output of the primary-side bridge 2 via the application of the zero voltage. The switching of the power semiconductors is controlled by control signals Si for the primary-side bridge 2. Control signals S2 control the secondary-side bridge 5 analogously to this.

By way of example and to simplify the description, the transformer 13 has a ratio of 1: 1, which is why in the present description of a distinction between the secondary-side voltage and that which is transformed to the primary side is disregarded. However, the transmission ratio of the transformer 13 may also be different from 1: 1, without restricting the desired circuit functionalities.

So that the upper secondary-side voltage Us, i (t) and the lower secondary-side voltage Us, 2 (t) do not change on average over time, the mean value of the positive current pulse of a secondary-side current is (t) must be equal to the mean value an upper load current iL, i (t) be. Analogously, the negative mean value of the negative current pulse of the secondary-side current is (t) must correspond to the mean value of a lower load current i | _, 2 (t). A difference between the mean value of the upper and lower load currents iL, i (t), iL, 2 (t) accordingly also leads to a difference in the mean values of the positive and negative current pulses of the secondary side current is (t). Therefore, the secondary-side current is (t) has a mean value not equal to zero. The mean value of the secondary-side current is negative for a larger mean value of the lower load current iL, 2 (t) than the mean value of the upper load current iL, i (t). If this mean value in the secondary-side current is (t) also be present in a primary-side current ip (t), then the resonance capacitor 11 would be discharged and a short-term negative average would result in a voltage uc, res (t) across the resonance capacitor 11 triggering the primary-side bridge 2 with a fixed duty cycle, this short-term negative mean value of the voltage uc, res (t) would result in a positive mean value being formed in a voltage up '(t) across the primary side of the transformer 13 (in FIG Sum, the mean values of the voltage uc, res (t) of the voltage up '(t) and that on the auxiliary coil 12 must cancel each other out). For this purpose, it was assumed that the upper primary side voltage Up> 1 (t) and the lower primary side voltage Up, 2 (t) are identical. The positive mean value of the voltage up '(t) causes the magnetizing inductance of the transformer 13 to build up a positive mean value of a magnetizing current. This average magnetization current is superimposed with the resonant current pulses and counteracts the discharge of the resonance capacitor 11. The mean value of the magnetizing current is increased until the mean value of the voltage uc, res (t) reaches zero again. In the steady state, it follows that for identical primary-side voltages UPi1 (t), Up, 2 (t), the primary-side current ip (t) becomes averaged and that the mean value of the voltage uc, res (t) also settles to zero. This also means that the magnetizing current corresponds on average to the negative mean difference of an upper load current iL, i (t) and a lower load current iL, 2 (t). In order to ensure the functionality of the circuit in the manner described, the transformer 13 may not saturate by the constructed mean magnetizing current, resp. the transformer 13 must be able to store energy. Advantageously, therefore, the core of the transformer 13 will have a low magnetic relative permeability. Accordingly, by incorporating an air gap in the magnetic core of the transformer 13 or by the choice of a powder core material, a compact design can be achieved. The primary-side current ip (t), the secondary-side current is (t) and the voltage uc, res (t) are shown for the stationary state in Fig. 1 (b).

The resonant capacitor 11 has, as explained, the task of exciting a resonant current pulse. In addition, in the case that an upper primary-side voltage UPi1 (t) is different from a lower-side primary voltage Up, 2 (t), the resonance capacitor 11 takes an average value of the voltage uc, res (t), so that the voltage up '(t ) located on the primary side of the transformer 13 is averaged (after the above explained behavior of the circuit).

As a result, the magnetizing current does not change on average by the magnetizing inductance of the transformer 13. Typically, the two primary side voltages Up, i (t), Up, 2 (t) are very close to each other, whereby the magnetizing current largely corresponds to the negative difference of the average values of the upper load current iL, i (t) to the lower load current iL2 (t).

Furthermore, the primary-side voltage Ui (t) of a DC-DC converter, as shown in FIG. 1 (a), is typically impressed. Due to the activation of the primary-side active bridge 2 with a fixed duty cycle, a defined division of the total voltage Ui (t) on the upper primary side DC link capacitor 7 and the lower primary side DC link capacitor 8 is achieved by adding the balancing coil 3. With a positive difference between the upper primary-side voltage Up, i (t) across the upper primary-side intermediate circuit capacitor 7 and the lower primary-side voltage Up, 2 (t) above the lower primary-side intermediate circuit capacitor, a positive mean value of the voltage up (t) is present the balancing coil 3 at. As a result, a positive mean value of a balancing current is established in the balancing coil 3. This leads to a discharge of the upper primary side DC link capacitor 7 and to a charging of the lower primary side DC link capacitor 8. The greater the difference between the two primary side voltages Up, i (t), Up, 2 (t), the greater the mean value of the balancing current , Therefore, the two primary-side voltages Up, i (t), Up, 2 (t) are the same until the voltage up (t) applied to the balancing coil 3 is averaged. Taking into account the ohmic resistance of the balancing coil 3 or the ohmic resistances of the terminals of the balancing coil 3, there may be a small difference between the voltages Up1 (t), Up, 2 (t) via both primary-side DC link capacitors 7, 8 come. For oscillations in the balancing current and thus in the primary-side voltages UP> 1 (t), Up, 2 (t) in the event of a sudden change in an upper load current iL, i (t) and / or a lower load current iL, 2 (t) 1 (a) can be supplemented with damping elements as shown in FIG. 1 (c): In this further circuit embodiment of the invention, an attenuation coil 15 is formed parallel to the balancing coil 3 and a damping resistor 14, wherein the damping coil 15 and the damping resistor are connected in series. In Fig. 1 (c) a possible realization of a parallel RL attenuation is shown. A serial RL damping, a parallel RC damping or a pure ohmic damping would also be conceivable. The time courses of the primary-side intermediate circuit voltages Up, i (t), Up, 2 (t) are shown for the circuit of Fig. 1 (c) in Fig. (D).

If the primary-side active bridge 2 and / or the secondary-side bridge 5 is realized by a 3-level topology that does not allow a direct transition from positive to negative (and vice versa) output voltages of the primary side bridge 2 (like those shown in FIG 1 (a)), a balancing coil 3 is compulsory so that the difference between the upper primary-side voltage UPi1 (t) and the lower-side primary voltage Up, 2 (t) does not constantly increase over time. The reason for this is the application of zero voltage to the output of the primary-side bridge 2. Between the transitions from a positive to a negative and from a negative to a positive output voltage of the primary-side bridge 2 is an interval, while the zero voltage at the output of primary-side bridge 2 is placed. During this interval, the voltage u0, res (t) across a resonance capacitor 11 is directly proportional negative to a transformer 13 at. The mean values of the voltage ifc, res (t) over each such interval are different for both intervals. For the assumed case that the mean value of a lower load current ii_, 2 (t) is greater than the mean value of an upper load current ii_, i (t), the result is a negative voltage mean value of the voltage uc, res (t) over the two Intervals for equal primary-side voltages UPi1 (t), Up, 2 (t). This results in an increased mean value of the voltage up '(t) across the transformer 13. As a result, the mean value of the magnetizing current flowing through the magnetizing inductance of a transformer 13 is increased. This represents a steady-state disturbance and may cause the difference between the upper primary-side voltage UPi1 (t) and the lower-side primary voltage Up, 2 (t) to steadily increase over time. For the sake of brevity, this problem will not be discussed further.

For a realization of the primary-side and secondary-side bridges 2, 5 with a 2-level topology, the problem just described does not exist, and the balancing coil 3 can be omitted. Fig. 1 (e) thus represents another possible circuit implementation of the invention. Alternatively, the primary-side active half-bridge 2 can also be replaced by a primary-side full bridge. For these two alternatives of the circuit implementation of the invention based on 2-level topologies, however, it is necessary that the voltage distribution between the upper and lower primary side DC link capacitor 7, 8 is influenced by means of another converter. This converter can be upstream of the DC-DC converter.

For a realization of a primary-side bridge 2 with a 3-level topology, however, the switching losses are lower at elevated clock frequencies than in a realization with a 2-level topology. To a compact resp. Achieving low-volume construction of a transformer 13 will typically increase the clock frequency at which the power semiconductors switch. Since a balancing coil 3 typically has to absorb only a small average current in comparison to the mean difference between an upper load current iL, i (t) and a lower load current iLi2 (t), it can be made compact.

A disadvantage of the circuit embodiment of the invention according to FIG. 1 (a), the existing differences between the upper primary-side voltage Up, i (t) over the upper primary side DC link capacitor 7 and the lower primary side voltage Up, 2 (t) above the lower primary-side intermediate circuit capacitor 8 and between the upper secondary-side voltage USi1 (t) above the upper secondary-side intermediate circuit capacitor 9 and the lower secondary-side voltage Us, 2 (t) above the lower secondary side intermediate circuit capacitor 10. These differences in the primary-side and secondary-side voltages depend on the circuit parameters and the load on the secondary side of a DC-DC converter. However, this disadvantage can be solved for either the primary side or the secondary side of the DC-DC converter. To achieve this, the activation of the primary-side active bridge 2 can not be done with a fixed duty cycle, but with a variable. A control method 16 or the structure of a controller for realizing the desired function is shown in FIG. 2 (a). From a voltage Ux, i (t), a voltage Ux, 2 (t), a primary-side current ip (t) and a secondary-side current is (t), drive signals Sx are generated. The two voltages Ux, i (t) and Ux, 2 (t) can be either the upper primary-side voltage Up, i (t) and the lower primary-side voltage Up, 2 (t) or the upper secondary-side voltage Us, i (t ) and the lower secondary side voltage Us, 2 (t). The drive signals Sx are amplified by a driver and control the power semiconductors of a secondary-side active bridge 5 either in the form of S-, the power semiconductors of a primary-side active bridge 2 or in the form of S2. A possible control method 16 is also shown in FIG. 2 (a): A positive difference between, for example, the upper secondary-side voltage Us, i (t) and the lower secondary-side voltage Us, 2 (t) leads via an external proportional-integral-differential (PID) controller 17 to an increased differential setpoint iPs, SOii (t) for a differential current ips (t) between the primary current ip (t) and the secondary current is (t). An internal PID controller 18 requires a mean bridge branch output voltage U | (t) at the output of the primary-side bridge 2 (or the secondary-side bridge 5) due to the current error between the differential setpoint ipSjSOii (t) and the differential current ips (t) the error between the differential setpoint ips, soii (t) and the differential current ips (t) becomes zero. A pulse width modulation unit (PWM unit) 19 generates from the middle bridge branch output voltage Ui (t) the drive signals Sx for the primary side active bridge 2 (or the secondary side active bridge 5). Finally, the desired mean bridge branch output voltage ui (t) in the form of up (t) [or us (t)] is applied to the output of the primary-side bridge 2 (or the secondary-side bridge 5). The time profiles of the two voltages across the upper and lower primary side DC link capacitors 7, 8 and the voltages across the upper and lower secondary side DC link capacitors 9, 10 are for the circuit shown in Fig. 1 (a) and with the in Fig. 2 (a ) shown in Fig. 2 (b) reproduced. As explained for the activation of the primary-side active bridge 2 with a fixed duty cycle, a balancing coil 3 is not absolutely necessary for a realization of the primary-side bridge 2 and the secondary-side bridge 5 as a 2-level topology; for a 3-level topology, however.

In a further circuit embodiment of the invention, the 4-pole AC-AC transformer 4 is not designed as a resonant converter but as a dual active bridge converter (DAB converter). This circuit is shown in FIG. 3 (a) and has no resonance capacitor 11. In Fig. 3 (a) a realization of a primary-side active bridge 2 and a secondary-side active bridge 5 with a star point-clamped 3-level topology is shown. However, the two bridges may also have a 2-level or other 3-level topology. The primary-side active bridge 2 and the secondary-side active bridge 5 are each provided with a duty cycle of 50% (optionally including the period for a 3-level topology, during the respective zero voltage at the output of the primary-side and secondary-side active bridges 2, 5 applied). The primary-side bridge 2 thus applies a rectangular voltage up (t) to its output, an upper primary-side voltage Up, i (t) and the negated value of a lower primary-side voltage Up, 2 (t) at the output of the primary-side during the same first time duration Abut bridge 2. The secondary-side bridge 5 accordingly applies a rectangular voltage us (t) to its output, wherein an upper secondary-side voltage US | 1 (t) and the negated value of a lower secondary-side voltage Us, 2 (t) during the same second time period at the output of the secondary side Bridge 5 abut. The first and the second time periods are identical and correspond to almost half the clock period. A clock period starts at the moment in which the output voltage of the primary-side bridge 2 (or secondary-side bridge 5) from a negative voltage for a 2-level topology or zero voltage for a 3-level topology of the primary-side bridge 2 (resp. the secondary-side bridge 5) changes to a positive voltage. The clock period immediately ends before a change in the output voltage of the primary-side bridge 2 (or the secondary-side bridge 5) from a negative voltage for a 2-level topology or zero voltage for a 3-level topology of the primary-side bridge 2 (FIG. or the secondary-side bridge 5) takes place to a positive voltage. For a positive power flow from the primary side to the secondary side of a DC-DC converter is a phase-shifted clocking of the primary-side bridge 2 and the secondary-side bridge 5. Therefore, the drive signals for the power semiconductors of the secondary-side bridge 5 by Tvz (t) is delayed to the drive signals for the Power semiconductor of the primary-side bridge 2, as shown in Fig. 3 (b). For a positive power flow from the secondary side to the primary side of the DC-DC converter, the timing of the primary side bridge 2 is delayed relative to that of the secondary side bridge 5 accordingly. During the time Tvz (t), a primary-side current ip (t), and therefore also a secondary-side current is (t), can be significantly increased or decreased via an additional coil 12 and / or the leakage inductance of the transformer 13. Thus, the transmitted power is increased by an increase in the delay Tvz (t) (or phase shift) and reduced in an analogous manner by reducing the delay T ^ t). As a result, a secondary-side voltage U2 (t) can be regulated to a reference value (phase-shift control).

Analogous to the explanations to the preceding embodiments, here too the secondary-side current is (t) has an average which corresponds to the difference of the average values of the upper load current iL, i (t) and the lower load current iL, 2 (t) , Thus, the upper secondary side voltage Us, i (t) across the upper secondary link capacitor 9 and the lower secondary side voltage Us, 2 (t) across the lower secondary side link capacitor 10 is maintained at a constant voltage level on average. If a primary-side current ip (t) were not mean-free, but had the same mean value as the secondary-side current is (t), then, for example, for a larger average of the lower load current ii_, 2 (t) would be higher than the mean value of the upper one Load current ii_, i (t) charge the upper primary side DC link capacitor 7 and the lower primary side DC link capacitor 8 discharged. The upper primary-side voltage UPi1 (t) across the upper primary-side intermediate circuit capacitor 7 thus increases, and the lower primary-side voltage Up, 2 (t) across the lower primary-side capacitor 8 decreases. This, in turn, causes a positive average of a voltage up (t) across the transformer 13, which is above the magnetizer

Claims (16)

is present, since the upper primary-side voltage Up, i (t) and the negated value of the lower primary-side voltage Up, 2 (t) for a fixed duty cycle in equal shares at the output of the primary-side bridge 2. By this positive mean value of the voltage up (t) across the transformer 13, an average value of a magnetizing current is increased by the magnetizing inductance of the transformer 13. This positive mean value of the magnetizing current acts as a discharge current for the upper primary-side capacitor 7 and thus counteracts the charging of the upper primary-side capacitor. In the same way, the mean value of the magnetizing current corresponds to a charging current of the lower primary side intermediate circuit capacitor, which counteracts the discharging of the lower primary side intermediate circuit capacitor. The magnetizing current is built up in the middle until the voltage up- (t) is again averaged. This is the case when the upper primary-side voltage Up, i (t) and the lower primary-side voltage are the same on average. Analogous considerations apply to the secondary side of the 4-pole AC-AC transformer 4. Thus, the circuit shown in FIG. 3 (a) has, on average, equal upper and lower primary side voltages Up, i (t), Up, 2 (t) and same upper and lower secondary voltages Us, i (t), Us, 2 (t). Accordingly, the primary-side current ip (t) is also free of medium-value and the magnetizing current circulating on the primary side of a DC-DC converter average of the average difference of the upper load current iL, i (t) and the lower load current iL, 2 (t) corresponds, cf. Fig. 3 (b). In order to ensure the functionality of the circuit in the manner described, the transformer 13 may not saturate by the constructed mean magnetizing current, resp. the transformer 13 must be able to store energy. Advantageously, therefore, the core of the transformer 13 will have a low magnetic relative permeability. Accordingly, by incorporating an air gap in the magnetic core of the transformer 13 or by the choice of a powder core material, a compact design can be achieved. List of Reference Numerals 1 Primary-side intermediate circuit 2 Primary-side active bridge 3 Balancing coil 1. DC-DC converter for electrical power transmission from a DC primary side to a DC secondary side, comprising • a 4-pole AC-AC transformer (4) for isolated power transmission,. a primary-side bridge circuit, which connects a primary side of the 4-pole AC-AC transformer (4) to the DC primary side, • an upper primary side DC link capacitor (7) and a lower primary side DC link capacitor (8), which are connected in series and together a secondary-side bridge circuit, which connects a secondary side of the 4-pole AC-AC transformer (4) to the DC secondary side, • an upper secondary side DC link capacitor (9) and a lower secondary-side intermediate circuit capacitor (10), which are connected in series and together support a voltage (U2 (t)) of the DC secondary side, wherein the DC-DC converter, a device for balancing the voltage of the two primary-side DC link capacitors (7, 8 ) or the two secondary side DC link capacitors (9, 10). 2. DC-DC converter according to claim 1, wherein the device for balancing the voltage of the two primary-side DC link capacitors or the two secondary-side DC link capacitors has an inductance for balancing the voltage, which in the presence of an asymmetry of the voltage of the two primary-side DC link capacitors (7, 8). or the two secondary side DC link capacitors (9, 10) builds up a current, and this current counteracts this asymmetry. 3. DC-DC converter according to claim 2, wherein the inductance for balancing by the magnetizing inductance of a transformer (13) of the 4-pole AC-AC transformer (4) is formed. 4. DC-DC converter according to claim 3, wherein the magnetizing inductance acts as an energy storage of the transformer (13) and the magnetic relative permeability of the core material of the transformer (13) is so low that a current which is in the presence of an asymmetry of the voltages across the primary side or secondary side DC link capacitors (7, 8, 9, 10) occurs and acts as a magnetizing current of the transformer (13), does not lead to a saturation of the transformer (13). 4-pole AC-AC transformer 5 Secondary active bridge 6 Secondary DC link 7 Upper primary DC link capacitor 8 Lower DC link capacitor 9 Upper secondary DC link capacitor 10 Lower secondary DC link capacitor 11 Primary resonant capacitor 12 Primary auxiliary coil 13 Differential (PID) Controller 18 Inner Proportional-Integral-Derivative (PID) Controller 19 Pulse Width Modulation Unit (PWM Unit) Claims 5. DC-DC converter according to claim 4, wherein the magnetizing inductance is formed by an air gap in the magnetic core of the transformer (13) or by using a powder core material. 6. DC-DC converter according to claim 2, wherein the inductance for balancing a balancing coil (3), which is connected in parallel to a primary-side or a secondary-side terminal pair of the 4-pole AC-AC transformer (4). 7. DC-DC converter according to claim 6, wherein the inductance for balancing on the magnetizing inductance of the transformer (13) and the balancing coil (3) is distributed, wherein the balancing coil (3) parallel to a primary-side or a secondary-side terminal pair of the 4-pole AC-AC transformer (4) is switched. 8. DC-DC converter according to claim 7, comprising a damping circuit for damping oscillations of the voltage across the upper and lower primary side DC link capacitor (7, 8) and above the upper and lower secondary side DC link capacitor (9, 10). 9. DC-DC converter according to claim 8, wherein the damping circuit is realized by a series circuit of a damping coil (14) and a damping resistor (15) which are mounted parallel to the balancing coil (3). 10. DC-DC converter according to one of claims 1 to 9, comprising a resonant circuit (11, 12) connected in series with one of the terminals of the 4-pole AC-AC transformer (4), wherein a primary-side bridge circuit (2 ) and a secondary-side bridge circuit (5) have a 3-level topology. 11. DC-DC converter according to one of claims 1 to 9, comprising a resonant circuit (11, 12) connected in series with one of the terminals of the 4-pole AC-AC transformer (4), wherein the primary-side bridge circuit (2 ) and the secondary-side bridge circuit (5) have a 2-level topology. 12. DC-DC converter according to one of claims 1 to 9, comprising a resonant circuit (11, 12) connected in series with one of the terminals of the 4-pole AC-AC transformer (4), wherein the primary-side bridge circuit (2 ) has a 3-level topology and the secondary-side bridge circuit (5) has a 2-level topology. 13. DC-DC converter according to one of claims 1 to 9, comprising a resonant circuit (11, 12) connected in series with one of the terminals of the 4-pole AC-AC transformer (4), wherein the primary-side bridge circuit (2 ) has a 2-level topology and the secondary-side bridge circuit (5) has a 3-level topology. 14. DC-DC converter according to one of claims 1 to 13, comprising a controller, with which the two voltages across the upper and lower primary side DC link capacitor (7, 8) or the two voltages across the upper and lower secondary side DC link capacitor (9, 10) can be actively regulated with a control method (16) to the same values. 15. DC-DC converter according to claim 14, wherein the controller has a structure with the following elements: • an external proportional-integral-derivative controller (17) for generating a differential setpoint ips, soii (t) for a differential current ips (t) between a primary current ip (t) and a secondary current is (t) based on a difference between the respective two voltages (Ux, i, Ux, 2); • an inner PID controller (18) which generates from the difference between the differential setpoint ips, s0ii (t) and the difference current ips (t) a mean bridge branch output voltage ui (t); • a pulse width modulation unit (19) which generates from the middle bridge branch output voltage U | (t) control signals Sx for the primary-side bridge circuit (2) or the secondary-side bridge circuit (5), so that the average bridge branch output voltage U | (t) in the time average at the outputs of the primary-side or secondary-side bridge circuit (2, 5) is applied. 16. DC-DC converter according to one of claims 1 to 5, which is designed as a dual active bridge converter.