CA2528799A1 - System and method for power line communications - Google Patents

Abstract

A communication system for a power line is described. A transmission system of the communication system divides the time axis into a number of time slots synchronized such that one time slot starts at the zero crossing of the power line signal.
These time slots are referred to as channels and are numbered from 1 to n. A modulation method is described to is narrow band continuous phase FSK, where a number m of modulating frequencies are used, arranged such that an integral number of full cycles fit into each channel time slot for all m frequencies. The system transmits during only a subset of the available time slots (channels) concentrated near the zero crossing of the power line waveform where the noise is typically minimal.

Description

SYSTEM AND METHOD FOR POWER LINE COMMUNICATIONS
FIELD OF THE INVENTION

[0001] This invention relates to a system and method of communications for power line media, particularly transmission in the presence of high amplitude, non stationary noise sources connected to the line.

SUMMARY OF THE INVENTION

[0002] A method for transmitting data over a power line in a time period is provided.
The method comprises: dividing the time period into a number of time slots synchronized such that one time slot starts at a zero crossing of a power line signal for transmitting the data, each time slot being relating to a channel and being numbered from 1 to n;
modulating a narrow band continuous phase FSK in which a number m of modulating frequencies are used, and arranged such that an integral number of full cycles fit into each time slot for each channel for all m frequencies; and transmitting data during only a subset of the available time slots concentrated near the zero crossing of the power line signal.

BRIEF DESCRIPTION OF THE DRAWINGS

[0003] Aspects of the invention will become more apparent from the following description of specific embodiments thereof and the accompanying drawings which illustrate, by way of example only, the principles of the invention. In the drawings, where like elements feature like reference numerals (and wherein individual elements bear unique alphabetical suffixes):

[0004] Figure 1 is a chart of the FSK burst slots in one half power line cycle for a system of an embodiment;

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[0005] Figure 2 is a diagram of Correlation receiver for a two frequency FSK
of the system of Figure 1;

[0006] Figure 3 is block diagram of an analog front end (AFE) of the system of Figure 1;

[0007] Figure 4 is a schematic diagram of the AFE of Figure 3;

[0008] Figure 5 is a schematic diagram of a transmitter circuit of the AFE of Figure 4;

[0009] Figure 6 is a schematic diagram of a low pass filter of the AFE of Figure 4;

[0010] Figure 7 is a schematic diagram of a high pass filter of the AFE of Figure 4;

[0011] Figure 8 is a schematic diagram of a protection circuit of the AFE of Figure 4;

[0012] Figure 9 is a schematic diagram of a band pass filter and amplifier of the AFE
of Figure 4;

[0013] Figure 10 is a schematic diagram of a high pass filter of the AFE of Figure 4;

[0014] Figure 11 is a schematic diagram of a protection diode circuit of the AFE of Figure 4;

[0015] Figure 12 is a schematic diagram of a band pass filter and amplifier of the AFE of Figure 4;

[0016] Figure 13 is a schematic diagram of another low pass filter of the AFE
of Figure 4;

[0017] Figure 14 is a schematic diagram of another band pass filter and amplifier of the AFE of Figure 4;

[0018] Figure 15 is a schematic diagram of a limiter of the AFE of Figure 4;

[0019] Figures 16 and 16a are schematic diagrams of an automatic gain control amplifier of the AFE of Figure 4; and [0020] Figure 17 is chart showing an optimized reception and transmission of multiple frequencies using a sine wave for the system of Figure 1.

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DETAILED DESCRIPTION OF EMBODIMENTS

[0021] The description which follows, and the embodiments described therein, are provided by way of illustration of an example, or examples, of particular embodiments of the principles of the present invention. These examples are provided for the purposes of explanation, and not limitation, of those principles and of the invention. In the description, which follows, like parts are marked throughout the specification and the drawings with the same respective reference numerals.

[0022] Current high speed communication on power line media (e.g. standard in house wiring) uses a variety of modulation techniques to overcome the highly noisy environment. Two types of systems have been commonly used. Firstly, wideband systems that uses spread spectrum to combat the interference may be used, see for example:

5,574,748 and 5,090,024 - Spread spectrum communications system for network;
5,263,046 - Spread-spectrum chirp communication with sharply defined bandwidth;
6,243,413 - Modular home-networking communication system and method using disparate communication channels;
6,616,254 - Code shift keying transmitter for use in a spread spectrum communications system;
5,579,335 - Split band processing for spread spectrum communications; and 5,748,671 - Adaptive reference pattern for spread spectrum detection, the contents of which are hereby incorporated by reference.

[0023] Secondly, narrow band systems that use one or more frequencies modulated in frequency or phase may also be used. See for example:

5,504,454 - Demodulator for powerline carrier communications; and 4,475,217 - Receiver for phase-shift modulated carrier signals, the contents of which are hereby incorporated by reference.

TDO-RED #8299833 v. 1 [0024] But Type of modulation techniques may also use various kind of synchronization, see for example:

6,734,784 - Zero crossing based powerline pulse position modulated communication system;
6,577,231 - Clock synchronization over a powerline modem network for multiple devices;
6,784,790 - Synchronization/reference pulse-based powerline pulse position modulated communication system;
6,907,472 - Distributed synchronization mechanism for shared communications media based networks; and 5,553,081 - Apparatus and method for detecting a signal in a communications system, the contents of which are hereby incorporated by reference.

[0025] The type of system used is also a function of the frequency spectrum allowed in the country of use. Most countries do not allocate enough spectrum for the wideband spread spectrum systems, so narrowband systems have been favoured, see for example:

- USA: FCC, PART 15 47 CFR CH.1 A, RADIO FREQUENCY DEVICES
(PART 15);
- EUROPE: EN50065-1 - SIGNALING ON LOW-VOLTAGE ELECTRICAL
INSTALLATIONS IN THE FREQUENCY RANGE 3 kHz TO 148.5 kHz; and - CANADA: ICES-006, Issue 1, August 25, 2001, AC Wire Carrier Current Devices (Unintentional Radiators), the contents of which are hereby incorporated by reference.

[0026] One feature of all these systems is that they use continuous transmission for each message, where a message typically consists of 100's of bits. However it has been observed from a large sample of data from the field that the noise on typical power lines where a number of disturbing devices are connected is not constant in either time or frequency, but exhibits quiet periods in both dimensions. The transmission system of the TDO-RED #8299833 v. I

present invention thus uses both time and frequency diversity to improve the robustness of the system in the presence of large amounts of non stationary power line noise, thereby achieving significantly improved performance in very adverse conditions.

[0027] In an aspect of the present invention, the transmission system divides the time axis into a number of time slots synchronized such that one time slot starts at the zero crossing of the power line signal (50 or 60 Hz depending on the region). These time slots are called channels and numbered from 1 to n. The modulation method used is narrow band continuous phase FSK, where a number m of modulating frequencies are used, arranged such that an integral number of full cycles fit into each channel time slot for all m frequencies. The system transmits during only a subset of the available time slots (channels) concentrated near the zero crossing of the power line waveform where the noise is typically minimal.

[0028] The system uses diversity by transmitting the same bit over one or more channels (time slots) and one or more frequencies. It uses a positive acknowledgment protocol with a reverse channel to tell the transmitter which redundancy method to use at any given time. The transmitter and receiver are both synchronized to the power line signal zero crossing and the default transmission method is the lowest bit rate using the maximum diversity. The system additionally uses a cyclic redundancy check (CRC) polynomial to detect the correct reception of messages - if the CRC is not received correctly, no acknowledgment is sent and the transmitter will revert to its default high redundancy state after some programmable delay.

[0029] In the paragraphs that follow, an embodiment of the system is described using a particular example of 4 channels and 2 frequencies on a 60 Hz power line.
However it should be clear to anyone versed in the art that this can be readily expanded to a n channels and m frequencies as well as the use on other power line frequencies (e.g. 50 Hz), in other embodiments.

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Burst mode FSK

[0030] Referring to Figure 1, tor the embodiment, the transmission method chosen is traditional FSK with two frequencies. The period of the power line waveform is divided into a number of segments, and transmission occurs during some but not all these segments. Thus consider segments of 600 sec - in a 60 Hz power line the period is 16.67 msec and the half period is 8.33 msec, giving 14 time slots of 595 sec in one half period. Of these the system transmits in 4 timeslots of 600 sec , leaving the rest of the period empty. The 4 timeslots are arranged asymmetrically with one before the zero crossing and 3 after - this is illustrated in Figure 2 and are numbered as channel 1 to 4.

[0031] The system of the embodiment uses continuous phase FSK with the transmitted signals:

sm (t) = T cos(27rf.t + 27c m Of t) m=1,2 and Af chosen such that:

_k ~f T

which assures seamless switching at the end of the burst T. Choosing T 600 sec and:
70 =116.7 kHz T
4f=~=16.7kHz f2 =f,+Of=133.3kHz completes the definition of the bursts. The receiver uses a traditional bank of correlators as shown in Figure 3. The correlators are synchronized to the zero crossing of the power TDO-RED #8299833 v. 1 line waveform and the output is sampled at the nearest peak at the end of the period T, minimizing the effect of any jitter in the zero crossing detection.

[0032] The cross correlation of these signals sampled at T is given by:
Pmk 1 f sm (t)sk (t) dt 2~

T
2~ f F32 cos(27r f~.t + 27c mOf t) =~ cos(2~ t + 2~ kOf t) dt [0033] This equation has two parts, one at DC and the other at 2 times the carrier frequency f, The result of the integration of the two parts is:

pmk = T f cos(2;t( m- k)Of t) dt T
+T f cos(47c ft + 2;t (m + k)Af t) dt - sin(27c( m- k)Af T) + sin(47c fT + 27c (m + k)Af T) 2;r( m- k)Of T 41r fT + 2;r (m + k)Of T
=1 if m=k = 0 otherwise since we have chosen f, = T= 70 and Af = T = 10. Note that this correlation is normalized to 1 because we divided it by the signal power. This also gives a clue about the effect of jitter in the zero crossing mentioned above. In this case the correlation is over a shorter period, resulting in less energy at the output. If we synchronize the correlator by taking the largest output sample near the end of the original burst, the effect can be approximated as the ratio the reduced burst length due to jitter Tred to the original burst length T as shown below:

TDO-RED #8299833 v. 1 Pmk 2s e S. (t)Sk (t) dt Ted cos(2;z f t+ 2~ kAf t) dt 1 FLT cos(2~ f t+ 2~ mOf t) FLT
=2s f ' -TYed sin(2;r( m- k)OfT,ed )+ sin(4;T fT + 27c (m + k)Of Tred ) T 27C( m- k)Af TPed 47c f T+ 2;z (m + k)Of Tred =T~ed ifm=k T
= 0 otherwise where we have assumed that TYed = fe and Tred = Af are still integers (this just says that we correlate over an integral number of cycles of fl and f2).

Transmission methods using time and frequency diversity [0034] The four time slots may be viewed as four independent channels. Thus diversity techniques are used to improve robustness in the presence of noise.
In particular we use both time and frequency diversity by transmitting multiple copies on different channels and using one or two frequencies as further explained below. Various combining techniques are then used to improve the robustness of the detection.

[0035] For the embodiment, the following transmission methods A - E may be used in this case, although this list is not exhaustive, and more possibilities exist particularly in the case of more channels and more frequencies:

A. (480bps).2-frequencies. 1-bit per channel. 4-bits per burst.

B. (240bps).2-frequencies. 2-bits per burst, Channel 0 and 1 merged. Channel 2 and 3 merged C. (120bps).2-frequencies. 1-bit per burst, Channel 0 and 1 merged. Channel 2 and 3 merged. a '0' is a transition from F 1 to F2, a '1' is a transition from F2 to Fl D. (120bps).2-frequencies. 1-bit per burst, All Channel Merged TDO-RED #8299833 v. I

E. (60bps).2-frequencies. 1/2-bit per burst, All Channel Merged, a'0' is a transition from Fl to F2, a' 1' is a transition from F2 to Fl [0036] These are illustrated graphically in the following tables 1 to 5:
Burst 1 bO bl b3 b4 Table 1 Method A

Burst 1 bO bO bl bl Table 2 Method B

Burst 1 Fl Fl F2 F2 F2 F2 Fl Fl Table 3 Method C
Burst 1 bO bO b0 bO
Table 4 Method D

Burst 1 Burst 2 Fl F1 Fl Fl F2 F2 F2 F2 F2 F2 F2 F2 Fl Fl Fl Fl Table 5 Method E
Detection algorithms using time and frequency diversity [0037] For the embodiment, the detector uses time and frequency diversity methods to improve the robustness of the transmission. The 4 channels and the 2 frequencies are used to make a combined decision depending on the transmitted sequence. The receiver TDO-RED #8299833 v. 1 monitors the channel and makes a decision on which transmission method is likely to yield the best result. A reverse channel protocol is used to communicate this decision to the transmitter.

[0038] Outlined below are some of the detection methods that can be used used, although it will be appreciated that this list is not exhaustive and other additional methods may also be used as is evident to anyone versed in this art. The basic idea is to use time diversity first by either repeating the same information in a number of time slots or reducing the number of time slots used, ignoring the ones that are too noisy.
This is then combined with frequency diversity by using only one of the two frequencies to make the decision, ignoring the other one judged to be too noisy. In the general case we use k out of n time slots and 1 out of m frequencies.

[0039] Method A:

1. Use maximum likelihood decision from correlator [0040] Method B

1. Use maximum likelihood decision from correlator on merged channels 2. Use maximum likelihood decision from correlator only on merged channels 1,2 [0041] Method C

1. Use maximum likelihood decision from correlator on merged channels 1,2 and 3,4 then apply differential decoding 2. Use maximum likelihood decision from correlator only on merged channels 1,2 then apply differential decoding 3. Same as 1 or 2 but monitor only change in F1 4. Same as 1 or 2 but monitor only change in F2 TDO-RED 48299833 v. I

[0042] Method D

1. Use maximum likelihood decision from correlator on merged channels 1,2,3,4 2. Use maximum likelihood decision from correlator only on merged channels 1,2 3. Use maximum likelihood decision from correlator only on channel 1 4. Use maximum likelihood decision from correlator only on channel 2 [0043] Method E

1. Use maximum likelihood decision from correlator on merged channels 1,2,3,4 in burst 1 and burst 2 and apply differential decoding 2. Use maximum likelihood decision from correlator only on merged channels 1,2 in burst 1 and burst 2 and apply differential decoding 3. Use maximum likelihood decision from correlator only on channel 1 in burst 1 and burst 2 and apply differential decoding 4. Use maximum likelihood decision from correlator only on channel 2 in burst 1 and burst 2 and apply differential decoding 5. Same as 1 to 4 but monitor only change in F 1 6. Same as 1 to 4 but monitor only change in F2 Synchronization, startup and tracking [0044] For the embodiment, the system uses a simple link layer protocol for startup and tracking. Synchronization is achieved by detecting the zero crossing of the power line signal and then looking for the maximum of the larger correlator output to determine the end of the burst near T micro seconds after the zero crossing (note that the correlator will contain part of the second burst if the zero crossing is detected late or noise only if it is detected early due to jitter , but the effect of this is small as shown above).

[0045] The link layer protocol transmits messages bounded by a start of message sync pattern at the beginning and a CRC at the end of the message. The receiver uses this TDO-RED #8299833 v. 1 CRC to determine if correct operation has been achieved and sends a positive acknowledgement to the transmitter to that effect.

[0046] It should be noted that other link layer protocols may also be used in conjunction with the transmission system in other embodiments, as will be evident to anyone versed in this art.

[0047] Startup is achieved by transmitting at the lowest bit rate, ('/2 bit per burst in this case). Once successful transmission at this bit rate is achieved (correct CRC
received), the receiver monitors all channels and all frequencies to determine if a higher bit rate could be sustained. It then communicates to the transmitter via a control message to use one of the other transmission patterns and switches its detection algorithm accordingly. It should be noted that another implementation can start with the highest bit rate and reduce it in case of bad CRC. Improved robustness is achieved by positively acknowledging each message. This allows the transmitter to revert to the lowest bit rate in case the channel deteriorates to the point where the receiver is not receiving correct data and does not send an acknowledgement.

Analog Front End Requirements [0048] The analog front end (AFE) is an analog circuit composed of a transmission and a reception circuit. This circuit provides the connection from the digital signal processing portions of the system to the analog portion of the power line. The receiver circuit is always on, whereas the transmitter circuit must be enabled with a logical high (1) on the TX_Enb pin. Both circuits have a protection diode circuit to limit spikes and signals present on the powerline and pass through the coupling.

[0049] Referring to Figure 3, a simplify block diagram of the AFE is shown.
Referring to Figure 4, a circuit schematic showing additional details of the AFE is shown.

[0050] For the embodiment, the following provides a summary of different filters that may be used in the AFE:

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D 1 1 ' Filter 1 Characteristics Filter type Butterworth Low-Pass Order 4 Cut off Frequency 180 KHz -3 dB
Filter 2 Characteristics Butterworth High-Pass Filter type Order 2 Cut off Frequency 190 KHz -3 dB
Filter 3 Characteristics Filter type (Type) Band-Pass Order for the high pass 10 Order for the low pass 4 Cut off Frequency (Low) 106 KHz -6 dB
Cut off Frequency (High) 160 KHz -6 dB
Frequency Center 125 KHz Gain 20dB
AGC Characteristics Gain >30dB
Response delay 30 S
Power Amplifier Characteristics Input impedance 60052 Output impedance <1 S2 Gain l 1 1 dB
Bandwidth 80-150KHz Power 2.25Wpeak (3Vpeak in 40) Protection Short circuit and over Protected by coupler voltage impedance Distortion -60dB (3' harmonic) Output impedance: 5052 Table 6 - AFE Specifications [0051] Details regarding different aspects of the AFE are now described in turn.
Transmitter Circuit [0052] Referring to Figure 5, the amplifier section of the AFE is made up of two stages:

= The transmission filter (TX filter).

TDO-RED #8299833 v. 1 = A voltage/current stage amplifies the input signal with low-distortion to meet FCC, ICES and CENELEC requirements. The output stage haslow output impedance. The amplifier is controlled by the Tx_Enb signal. When Tx_Enb is low, the current stage is high impedance to allow power line signal to be received by the RX section. When Tx_Enb is high, the current stage amplifies the signal from the voltage amplifier and transmits it to the coupler.

[0053] The amplification is 11 1 dB. So the range is 1.7 volts peak-to-peak.
The output impedance is less than 1 Q when transmitting and more than 250 S2 in idle state.
The transmitter uses an integrated circuit to simplify the amplification. The integrated circuit amplifier supports low impedance on the power line without distorting the signal transmission.

[0054] For the embodiment, a 2 amplifiers that work with a bridge configuration to be able to transmit a 6Vpp signal on the line from a single 5V supply is used.
The output of the transmission amplifier is not protected against shorts between ground and output.
The output signal is transmitted at 6 Vpp for a load greater than 6 Q. For a load smaller than 6 S2, the output signal decreases but the distortion stays at a low level to avoid transmitting harmonics on the power line.

[0055] The band-pass filter of the amplifier ranges from 80 kHz to 150 kHz. As the RHINO IC delivers a pulse witdh modulation signal, we need to filter it by using a passif low pass filter to reshape the signal to amplify. The transmitting filter is used to filter the signal taken from the RHINO IC and to feed it to the voltage/current amplifier. This is accomplished by eliminating the high frequencies of the TX signal at the input of the amplifier. This may be done by a low pass filter as shown in Figure 6.

Receiver Circuit [00561 For the embodiment, the receiving circuit is always enabled. It receives the signal from the power line and filters it for the RHINO IC. The receiver have several functions to achieve for the RHINO IC:

TDO-RED #8299833 v. I

- Extract the signal of the noise present at the output of the coupling circuit by a efficiency filtering - Compress the signal without clipping it to preserve the shape when the maximum amplitude is reached - Amplify the signal when it is necessary; depending of the attenuation present on the powerline - Warm the RHINO IC as it compress the signal and when the line is noisy [0057] To do these functions, the receiver is divided in several sections which are independent of each other. Depending of performances required for different commercial applications, this division provides an easy ability to remove sections. For the embodiment, the following sections are implemented:

- A high pass filter - A protection diode circuit - A band-pass filter and amplifier - A high pass filter - A protection diode circuit - A band-pass filter and amplifier - A low pass filter - A band-pass filter and amplifier - A limiter that react as a protection for the RHINO IC
- an automatic Gain Control amplifier which control gain of 2 of the three amplifiers [0058] The different sections clean the signal but do not saturate it to preserve the shape of the signal. The minimum signal to be detected by the AFE is 30 V if the noise floor is lower than -97 dB (Vpp). This gives a sensitivity of -97 dB.

[0059] Each of these sections are now described in turn. Referring first to Figure 7, a high pass filter is shown.

[0060] Referring to Figure 8, a protection diode circuit is shown.
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[0061] Referring to Figure 9, a band-pass filter and amplifier is shown.
[0062] Referring to Figure 10, a high pass filter is shown.

[0063] Referring to Figure 11, a protection diode circuit is shown.
[0064] Referring to Figure 12, a band-pass filter and amplifier is shown.
[0065] Referring to Figure 13, a low pass filter is shown.

[0066] Referring to Figure 14, a band-pass filter and amplifier is shown.

[0067] Referring to Figure 15, a limiter that react as a protection for the RHINO IC is shown.

[0068] Referring to Figures 16 and 16a, an automatic gain control (AGC) amplifier that controls gain of 2 of the three amplifiers is shown.

100691 The AGC is designed to have fast response within 80 S. This is enough fast to control amplification of the different section of the receiver and gives feedback to the RHINO IC.

[0070] Referring now to Figure 17, an optimized reception and transmission of multiple frequencies using a single sine wave is now described and shown. As shown, there is a 356pts 2.5khz sine wave stored in RAM memory (for a sampling rate of 888888.8 samples per seconds). This single sine wave is used by the DSP to perform DTFT on any frequency that is a multiple of 2.5khz.

[0071] The following function performs the DTFT at 110khz using this table.
Notice that only N register must be modified in order to select any frequency.:

moveu.w #356,LC // 2c moveu.w #(32768 + 356-1),M01 // RO and Rl are configured as MOD(356) Addressing move.w #44,N // 44 x 2.5khz = 110khz parameter moveu.w #DFTTable + 89,R0 // Imaginary Part (Cos) offset of 90degrees moveu.w #DFTTable,Rl Real Part (Sin) no offset moveu.w #TestBuffer,R3 ADC Data Ptr clr a x:(rl)+N,yl // real part+ add(N) to rl nop clr b x:(r0)+N,yO x:(r3)+,x0 // imaginary part+ add(N) to rO
DOSLC _ENDOFLOOP

TDO-RED #8299833 v. 1 mac yl,xO,a x:(rl)+N,yl // real part + add(N) to rl mac yO,xO,b x:(rO)+N,yO x:(r3)+,xO // imaginary part + add(N) to rO
ENDOFLOOP:

[0072] It will be appreciated that the same principles may be used in transmission in order to generate a pulse width modulated wave.

[0073] Although the invention has been described with reference to certain specific embodiments, various modifications thereof will be apparent to those skilled in the art without department from the scope of the invention.

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Claims

WE CLAIM:

1. A method for transmitting data over a power line in a time period, the comprising:
dividing the time period into a number of time slots synchronized such that one time slot starts at a zero crossing of a power line signal for transmitting the data, each time slot being relating to a channel and being numbered from 1 to n;

modulating a narrow band continuous phase FSK in which a number m of modulating frequencies are used, and arranged such that an integral number of full cycles fit into each said time slot for each said channel for all m frequencies; and transmitting data during only a subset of the available time slots concentrated near the zero crossing of the power line signal.