BR112012022574B1 - APPARATUS AND METHOD FOR PROCESSING AN INPUT AUDIO SIGNAL USING CASCADED FILTER BANKS - Google Patents

Abstract

apparatus and method for processing an input audio signal using cascading filter banks. an apparatus for processing an input audio signal (2300) depends on a cascade of filter banks, the cascade comprising a synthesis filter bank (2304) for synthesizing an intermediate audio signal (2306) from the input audio signal (2300), the input audio signal being represented by a plurality of first sub-track signals (2303) generated by an analysis filter bank (2302), wherein a number of filter bank channels of the filter bank of synthesis (2304) is less than a number of channels in the analysis filter bank (2302). the apparatus further comprises another analysis filter bank (2307) for generating a plurality of second sub-track signals (2308) of the intermediate audio signal (2306), wherein the other analysis filter bank has a number of channels being different of the number of channels of the synthesis filter bank (2304), such that a sampling rate of a sub-range signal of the plurality of second sub-range signals (2308) is different from a sampling rate of a first sub-range signal of the plurality of first subrange signals (2303).

Description

TECHNICAL FIELD

The present invention relates to audio source coding systems that make use of a harmonic transposition method for high frequency reconstruction (HFR), and to digital effect processors, for example, the so-called exciters, where the generation of harmonic distortion adds luminosity to the processed signal, and to time extenders, where the duration of a signal is extended while maintaining the spectral content of the original.

HISTORY OF THE INVENTION

In PCT WO 98/57436 the concept of transposition was established as a method for recreating a high frequency range from a lower frequency range of an audio signal. Substantial bitrate savings can be achieved using this concept in audio coding. In an HFR-based audio coding system, a low-bandwidth signal is processed by a center waveform encoder and higher frequencies are regenerated using transposition and additional very low bitrate side information that describe the target spectral shape on the encoder side. For low bit rates, where the bandwidth of the encoded central signal is narrow, it becomes increasingly important to recreate a high bandwidth with perceptually pleasing characteristics. Harmonic transposition defined in PCT WO 98/57436 performs very well for complex musical material in a situation with low cross frequency. The principle of harmonic transposition is that a sinusoid with frequency co is mapped to a sinusoid with frequency Tco where T > 1 is an integer that defines the order of transposition. In contrast, a single sideband (SSB) modulation-based HER method maps a sinusoid with frequency co to a sinusoid with frequency co + Δco where Δco is a fixed frequency shift. Given a low bandwidth center signal, a jarring ring artifact can result from SSB transposition.

To achieve the best possible audio quality, state-of-the-art high-quality harmonic HFR methods employ complex modulated filter banks, e.g. a Short Duration Fourier Transform (STFT), with high frequency resolution and a high degree of oversampling to achieve the required audio quality. Fine resolution is required to avoid unwanted intermodulation distortion that arises from nonlinear processing of sinusoid sums. With sufficiently high frequency resolution, i.e. narrow subranges, high quality methods aim to have a maximum of one sinusoid in each subrange. A high degree of time oversampling is necessary to avoid the alternate type of distortion, and a certain degree of frequency oversampling is necessary to avoid pre-echoes for transient signals. The obvious disadvantage is that the computational complexity can become high.

Block-based subband harmonic transposition is another HFR method to suppress intermodulation products, in which case a filter bank with coarse frequency resolution and a lower degree of oversampling is employed, for example, a multichannel QMF bank. In this method, a time block of complex subrange samples is processed by a common phase modifier while the superposition of several modified samples forms an output subrange sample. This has the net effect of suppressing the intermodulation products that would otherwise occur when the subband input signal consists of multiple sinusoids. Transposition based on block-based subband processing has much lower computational complexity than high-quality transpositions and achieves nearly the same quality for many signals. However, the complexity is still much higher than for HFR methods based on trivial SSB, since a plurality of analysis filter banks, each processing signal of different transposition orders T, is needed in a typical application of HFR to synthesize the required bandwidth. Additionally, a common approach is to adapt the sampling rate of the input signals to fit the analysis filterbanks of a constant size, although the filterbanks process signals of different transposition orders. It is also common to apply low-pass filters to input signals to obtain output signals, processed in different transposition orders, with non-overlapping spectral energy densities.

The storage or transmission of audio signals is generally subject to strict bit rate restrictions. In the past, encoders were forced to drastically reduce the bandwidth of transmitted audio only when a very low bit rate was available. Currently, modern audio codecs can encode ultra-wideband signals using Bandwidth Extension (BWE) methods [1-12]. These algorithms rely on a parametric representation of the high-frequency (HF) content that is generated from the low-frequency (LF) portion of the decoded signal by means of transposition to the HF spectral region ("repair") and application of a post-triggered parameter. -processing. The LF part is encoded with any audio or speech encoder. For example, the bandwidth extension methods described in [1-4] rely on a single sideband modulation (SSB), often also called the "copy" method, to generate multiple HF chunks.

Recently, a new algorithm, which employs a bank of phase vocoders [15-17] to generate the different chunks, was presented [13] (see figure 20) . This method was developed to avoid the auditory roughness that is commonly seen in signals subject to the SSB bandwidth extension. However, since the BWE algorithm runs on the decoder side of a codec stream, computational complexity is a serious problem. Prior art methods, especially HBE based on phase vocoder, value a very high computational complexity compared to SSB based methods.

As described above, existing bandwidth extension schemes apply only one repair method to a given signal block at a time, be it SSB-based repair [1-4] or HBE vocoder-based repair [15]. — 17], Additionally, modern audio encoders [19-20] offer the possibility to switch the repair method globally on a time block basis between alternative repair schemes.

SSB copy repair introduces unwanted harshness to the audio signal, but it is computationally simple and preserves the transient envelope time. Furthermore, the computational complexity is significantly high over the very simple SSB copy method.

SUMMARY OF THE INVENTION

When it comes to a reduction in complexity, sampling rates are of particular importance. This is due to the fact that a high sample rate means high complexity and a low sample rate usually means low complexity due to the reduced number of operations required. On the other hand, however, the situation in bandwidth-extending applications is particularly such that the sample rate of the central encoder output signal will typically be so low that this sample rate is too low for a signal of bandwidth. full band. Indicated differently, when the sampling rate of the decoder output signal is, for example, 2 or 2.5 times the maximum frequency of the central encoder output signal, then an extension of the bandwidth, for example, by a factor of 2 means that a sample rate up operation is required so that the sample rate of the extended bandwidth signal is too high that the sample can "cover" the additionally generated high frequency components.

Additionally, filter banks such as analysis filter banks and synthesis filter banks are responsible for a considerable amount of processing operations. Thus, the size of the filter banks, i.e. if the filter bank is a filter bank with 32 channels, a filter bank with 64 channels or even a filter bank with a higher number of channels will significantly influence the complexity. of the audio processing algorithm. Generally, it can be said that a high number of filter bank channels requires more processing operations and thus higher complexity than a small number of filter bank channels. Given this situation, in bandwidth extension applications and also in other audio processing applications where different sample rates are an issue, such as in applications like vocoder or any other audio effect applications, there is an independence specifies between complexity and audio sample rate or bandwidth, which means that operations for upsampling or subband filtering can dramatically improve complexity without specifically influencing audio quality in a good way when the wrong tools or algorithms are chosen for specific operations.

The present invention aims to provide an improved audio processing concept, which allows low processing complexity on the one hand and good audio quality on the other.

This object is achieved by an apparatus for processing an input audio signal according to claim 1 or 18, a method for processing an input audio signal according to claim 20 or 21, or a computer program , according to claim 22.

Embodiments of the present invention rely on a specific cascading of analysis filter banks and/or synthesis filter banks to achieve low resampling complexity without sacrificing audio quality. In one embodiment, an apparatus for processing an input audio signal comprises a synthesis filter bank for synthesizing an intermediate audio signal from the input audio signal, wherein the input audio signal is represented by a plurality of first subrange signals generated by an analysis filter bank placed in the processing direction before the synthesis filter bank, where a number of filter bank channels of the synthesis filter bank is less than a number of channels of the bank analysis filter. The intermediate signal is further processed by another analysis filter bank to generate a plurality of second sub-band signals of the intermediate audio signal, wherein the other analysis filter bank has a number of channels being different from the number of channels in the bank. synthesis filter such that a sampling rate of a sub-range signal of the plurality of sub-range signals is different from a sampling rate of a first sub-range signal of the plurality of first sub-range signals generated by the analysis filter bank.

The cascade of a synthesis filter bank and another subsequently connected analysis filter bank provides a sample rate conversion and additionally a modulation of the portion of the bandwidth of the original input audio signal that was fed to the filter bank. synthesis in a baseband.

This time-intermediate signal, which has now been extracted from the original input audio signal which may, for example, be an output signal from a central decoder of a bandwidth extension scheme, is now preferably represented as a sampled signal. critically modulated to the baseband, and it has been seen that this representation, i.e. the output signal resampled, when processed by another analysis filter bank to obtain a representation of the subband allows for low-complexity processing of other processing operations that may or may not occur and which may, for example, be bandwidth-extending processing operations such as non-linear subrange operations followed by high frequency reconstruction processing and a blending of the subranges in the final synthesis filter bank.

The present application provides different aspects of apparatus, methods or computer programs to process audio signals in the context of bandwidth extension and in the context of other audio applications, which are not related to bandwidth extension. The characteristics of the individual aspects subsequently described and claimed can be partially or completely combined, but they can also be used separately from one another, as the individual aspects already provide advantages with respect to perceptual quality, computational complexity and processor/memory resources when implemented. in a computer system or microprocessor.

The embodiments provide a method for reducing the computational complexity of a subband block-based harmonic HFR method by efficiently filtering and converting the sample rate of input signals to the analysis stages of the HFR filter bank. Also, the low pass filters applied to the input signals can be shown as obsolete in a transpose based on the subband block.

The present embodiments help to reduce the computational complexity of block-based subrange harmonic transposition by efficiently implementing multiple orders of subrange block-based transposition in the structure of a single pair of analysis and synthesis filter banks. Depending on perceptual quality versus computational complexity compromise, only a suitable subset of orders or all transposition orders can be performed collectively within a filter bank pair. Furthermore, a combined transposition scheme where only certain transpose orders are calculated directly where the remaining bandwidth is filled by playing available transposition orders, i.e. previously calculated, (e.g. 2nd order) and/or the central coded bandwidth. In this case, repair can be performed using every conceivable combination of source tracks available for playback.

Additionally, the findings provide a method for improving high quality harmonic HFR methods as well as subband block-based harmonic HFR methods by means of spectral alignment of HFR tools. In particular, high performance is achieved by aligning the spectral edges of the signals generated by the HFR to the spectral edges of the envelope fit frequency table. Furthermore, the spectral edges of the limiting tool are, by the same principle, aligned with the spectral edges of the signals generated by the HFR.

Other realizations are configured to improve the perceptual quality of transients while reducing computational complexity, for example by applying a repair scheme that applies a mixed repair consisting of harmonic repair and copy repair.

In specific embodiments, the individual filter banks of the cascading filter bank structure are quadrature mirror (QMF) filter banks, which depend on a prototype low-pass filter or modulated window using a set of modulation frequencies that define the center frequencies of the filter bank channels. Preferably, all window functions and prototype filters depend on each other in such a way that filters from filter banks of different sizes (filter bank channels) depend on each other as well. Preferably, the largest filter bank in a cascading structure of filter banks comprising, in embodiments, a first analysis filter bank, a subsequently connected filter bank, another analysis filter bank, and at some later stage of processing a final synthesis filter bank, has a response from the window function or prototype filter having a certain number of coefficients from the window function or prototype filter. The smaller sized filter banks are subsampled versions of this window function, which means that the window functions for the other filter banks are subsampled versions of the "large" window function. For example, if a filter bank is half the size of the large filter bank, then the window function has half the number of coefficients, and the coefficients of the smaller filter banks are derived by subsampling. In this situation, subsampling means that, for example, every second the filter coefficient is considered for the smallest filter bank having half the size. However, when there are other relationships between the filter bank sizes that are not integer values, a certain type of interpolation of the window coefficients is performed so that at the end of the window of the smallest filter bank is again a subsampled version of the window. biggest window.

Embodiments of the present invention are particularly useful in situations where only a part of the input audio signal is needed for further processing, and this situation particularly occurs in the context of extending harmonic bandwidth. In this context, vocoder-like processing operations are particularly preferred.

It is an advantage of the realizations that the realizations provide lower complexity for a QMF transposition by efficient time and frequency domain operations and an improved audio quality for QMF and DFT based harmonic spectral band reproduction using spectral alignment.

The realizations relate to audio source encoding systems employing, for example, a block-based subband harmonic transposition method for high frequency reconstruction (HFR), and for digital effect processors, for example, so-called exciters, where the generation of harmonic distortion adds luminosity to the processed signal, and to time extenders, where the duration of a signal is extended while maintaining the spectral content of the original. The embodiments provide a method for reducing the computational complexity of a subband block-based harmonic HFR method by means of efficient filtering and sample rate conversion of input signals prior to the HFR filter bank analysis stages. Further, the findings show that conventional low-pass filters applied to input signals are obsolete in a subband block HFR system. Additionally, the findings provide a method for improving high quality harmonic HFR methods as well as subband block-based harmonic HFR methods by means of spectral alignment of HFR tools. In particular, the embodiments teach how high performance is achieved by aligning the spectral edges of the signals generated by the HFR to the spectral edges of the frequency table of the enveloping fit. Furthermore, the spectral edges of the limiting tool are by the same principle aligned with the spectral edges of the signals generated by the HFR.

BRIEF DESCRIPTION OF THE DRAWINGS

The present invention will now be described in the form of illustrative examples, not limiting the scope or spirit of the invention, with reference to the accompanying drawings, in which:

Figure 1 illustrates the operation of a block-based transposition using transposition orders of 2, 3, and 4 in an HFR-enhanced decoder structure;

Figure 2 illustrates the operation of the non-linear sub-band extension units in Figure 1;

Figure 3 illustrates an efficient implementation of the transposition based on the block of Figure 1, where the resamplers and low-pass filters preceding the HFR analysis filter bank are implemented using multirate time-domain resamplers and low-pass filters based on in QMF;

Fig. 4 illustrates an example of building blocks for an efficient implementation of the multirate time domain resampler of Fig. 3;

Figure 5 illustrates the effect on an exemplary processed signal by the different blocks of Figure 4 for a transposition order of 2;

Figure 6 illustrates an efficient implementation of transposition based on the block in Figure 1, where the preceding resamplers and low-pass filters from the HFR analysis filter bank are replaced by the small subsampled synthesis filter bank operating on selected subranges of a 32-track analysis filter bank;

Figure 7 illustrates the effect on an exemplary processed signal by a subsampled synthesis filter bank of Figure 6 for a transposition order of 2;

Figure 8 illustrates the blocks in implementation of an efficient 2-factor multi-rate time-domain sample rate reducer;

Figure 9 illustrates the blocks in implementation of an efficient 3/2 factor multi-rate time domain sampling rate reducer;

Figure 10 illustrates the alignment of the spectral edges of the HFR transposition signals to the edges of the envelope-adjustable frequency bands in an improved HFR encoder;

Figure 11 illustrates a scenario where artifacts emerge due to misaligned spectral edges of the HFR transposition signals;

Figure 12 illustrates a scenario where the artifacts of Figure 11 are avoided as a result of aligned spectral edges of the HFR transposition signals;

Figure 13 illustrates the adaptation of the spectral edges in the limiting tool to the spectral edges of the HFR transposition signals;

Figure 14 illustrates the principle of subband harmonic transposition based on the block;

Fig. 15 illustrates an example scenario for applying subtrack block-based transposition using various transposition orders in an HFR enhanced audio codec;

Fig. 16 illustrates an exemplary prior art scenario for operating a multi-order transposition based on the sub-range block that applies a separate analysis filter bank by transposition order;

Fig. 17 illustrates an inventive example scenario for efficiently operating a subrange block-based multi-order transposition by applying a single 64-range QMF parsing filter bank;

Fig. 18 illustrates another example for forming a broad processing with the subband signal;

Fig. 19 illustrates a single sideband (SSB) modulation repair;

Figure 20 illustrates a harmonic bandwidth extension (HBE) repair;

Fig. 21 illustrates a mixed repair, where the first repair is generated by frequency propagation and the second recovery is generated by an SSB copy of a low frequency part;

Fig. 22 illustrates a mixed alternative repair using the first HBE repair for an SSB copy operation to generate a second recovery;

Figure 23 illustrates a preferred cascade structure of analysis and synthesis filterbanks;

Fig. 24a illustrates a preferred implementation of the small synthesis filter bank of Fig. 23;

Fig. 24b illustrates a preferred implementation of the other analysis filter bank of Fig. 23;

Figure 25a illustrates overviews of certain analysis and synthesis filter banks of ISO/IEC 14496-3: 2005(E), and particularly an implementation of an analysis filter bank that can be used for the analysis filter bank. analysis of Fig. 23 and an implementation of a synthesis filter bank that can be used for the final synthesis filter bank of Fig. 23;

Fig. 25b illustrates an implementation as a flowchart of the analysis filter bank of Fig. 25a;

Fig. 25c illustrates a preferred implementation of the synthesis filter bank of Fig. 25a;

Figure 26 illustrates an overview of the structure in the context of bandwidth extension processing; and

Fig. 27 illustrates a preferred implementation of processing the output of the subband signals by the other analysis filter bank of Fig. 23.

BRIEF DESCRIPTION OF THE DRAWINGS

The embodiments described below are merely illustrative and may provide a lower complexity of a QMF transposition by efficient time and frequency domain operations, and improved audio quality of harmonic SBR based on QMF and DFT by spectral alignment. It is understood that modifications and variations of the provisions and details described herein will be apparent to those skilled in the art. It is intended, therefore, to be limited only by the scope of the forthcoming patent claims and not by the specific details presented in the form of description and explanation of the embodiments herein.

Fig. 23 illustrates the preferred implementation of the apparatus for processing an incoming audio signal, where the incoming audio signal may be an in-line time domain input signal 2300 output by, for example, a central audio decoder 2301. The input audio signal is fed into a first analysis filter bank 2302 which is, for example, an analysis filter bank having M channels. Particularly, the parsing filter bank 2302 thus outputs M subband signals 2303, which have a sampling rate fs = fs/M. This means that the analysis filter bank is a critically sampled analysis filter bank. This means that the analysis filter bank 2302 provides, for each block of M samples input in line 2300, a single sample for each subband channel. Preferably, the analysis filterbank 2302 is a complex modulated filterbank which means that each subrange sample has a magnitude and a phase or equivalently a real part and an imaginary part. Thus, the line-in audio signal 2300 is represented by a plurality of first sub-range signals 2303 that are generated by the analysis filter bank 2302.

A subset of all first subrange signals is fed into a synthesis filter bank 2304. Synthesis filter bank 2304 has channels Ms, where Ms is less than M. Thus, not all subrange signals generated by the bank of filter 2302 are added to the synthesis filter bank 2304, but only a subset, that is, a certain smaller number of channels as indicated by 2305. In figure 23 of the embodiment, the subset 2305 protects a certain intermediate bandwidth, but from alternatively, the subpool can also protect a bandwidth starting with the channel of filter bank 1 of filter bank 2302 up to a channel having a channel number less than M, or alternatively, the subpool 2305 can also protect a group of sub-band signals aligned with the highest channel M and extended to a lower channel having a channel number higher than channel number 1. Alternatively, channel indexing can be started with zero depending on the notification currently used. Preferably, however, for bandwidth-extending operations a given intermediate bandwidth represented by the group of sub-band signals indicated at 2305 is fed into the synthesis filter bank 2304.

The other channels that do not belong to the 2305 group are not fed to the 2304 synthesis filter bank. The 2304 synthesis filter bank generates an intermediate audio signal 2306, which has a sampling rate of fs • Ms/M. Since Ms is less than M, the sampling rate of the intermediate signal 2306 will be less than the sampling rate of the line-in audio signal 2300. In this way, the intermediate signal 2306 represents an undersampled and demodulated signal corresponding to the signal of the bandwidth represented by sub-bands 2305, where the signal is demodulated to baseband, as the lowest channel of group 2305 is inserted into channel 1 of the Ms synthesis filter bank and the highest channel of block 2305 is inserted into the input 2304 block, away from some zero-fill operations for the lowest or highest channel to avoid aliasing problems at the edges of the 2305 subset. The apparatus for processing an input audio signal further comprises another filter bank of analysis 2307 to analyze the intermediate signal 2306, and the other analysis filter bank has MA channels, where MA is different from Ms and preferably is greater than Ms. When MA is greater than Ms, then the sampling rate of subrange signals emitted by the other analysis filter bank 2307 and indicated in 2308 will be lower than the sampling rate of a subrange signal 2303. However, when MA is smaller than Ms, then the sampling rate of a subband signal 2308 will be higher than a sampling rate of a subband signal of the plurality of first subband signals 2303.

In this way, the cascade of filter banks 2304 and 2307 (and preferably 2302) provide very efficient sample rate uplift and high quality downsampling or upsampling operations or generally a very efficient resampling processing tool. The plurality of second subband signals 2308 are preferably further processed in a processor 2309 which performs processing with the data resampled by the cascade of filter banks 2304, 2307 (and preferably 2302). Additionally, it is preferred that block 2309 also performs a sample rate up operation for bandwidth extension processing operations so that in the end the sub-bands emitted by block 2309 are at the same sampling rate as the emitted sub-bands. by block 2302. Then, in an application of bandwidth extension processing, these subranges are inserted together with the additional subranges indicated at 2310, which are preferably the low range subranges as, for example, generated by the filter bank of analysis 2302 into a synthesis filter bank 2311, which ultimately provides a processed time domain signal, e.g., an extended bandwidth signal having a sampling rate of 2fs. This sampling rate emitted by block 2311 is in this embodiment 2 times the sampling rate of the on-line signal 2300, and this sampling rate emitted by block 2311 is large enough that the additional bandwidth generated by processing in block 2309 can be represented in the processed time domain signal with high audio quality.

Depending on the particular application of the present invention of cascading filter banks, the filter bank 2302 may be in a separate device and an apparatus for processing an input audio signal may only comprise the synthesis filter bank 2304 and the other bank. analysis filter bank 2307. Stated differently, the analysis filter bank 2302 may be distributed separately from a "post"-processor comprising blocks 2304, 2307 and, depending on the implementation, blocks 2309 and 2311, as well.

In other embodiments, the application of the present invention which implements the cascading filter banks may be different in that a given device comprises the analysis filter bank 2302 and the smallest synthesis filter bank 2304, and the intermediate signal is provided to a different processor distributed by a different distributor or through a different distribution channel. So, the combination of the analysis filter bank 2302 and the smaller synthesis filter bank 2304 represents a very efficient way of reducing the sample rate and at the same time demodulating the signal from the bandwidth represented by the subset 2305 to the baseband. . This sample rate reduction and baseband demodulation was performed without any loss in audio quality, and particularly without any loss in audio information and thus is high quality processing.

The table in Figure 23 illustrates certain exemplar numbers for different devices. Preferably, the analysis filter bank 2302 has 32 channels, the synthesis filter bank has 12 channels, the other analysis filter bank has 2 times the channels of the synthesis filter bank, such as 24 channels, and the final synthesis filter 2311 has 64 channels. Generally stated, the number of channels in the analysis filter bank 2302 is large, the number of channels in the synthesis filter bank 2304 is small, the number of channels in the other analysis filter bank 2307 is medium, and the number of channels in the 2311 synthesis filter bank is too large. The sampling rates of the subband signals emitted by the analysis filter bank 2302 are fs/M. The intermediate signal has a sampling rate of fs • Ms/M. The subband channels of the other analysis filter bank indicated in 2308 have a sampling rate of fs • Ms/(M • MA) , and the synthesis filter bank 2311 provides an output signal having a sampling rate of 2fs , when processing at block 2309 doubles the sample rate. However, when the processing at block 2309 does not double the sample rate, then the sample rate output emitted by the synthesis filter bank will be correspondingly lower. Subsequently, other preferred embodiments relating to the present invention are discussed.

Figure 14 illustrates the principle of transposition based on the subtrack block. The input time domain signal is fed to an analysis filter bank 1401 which provides a wide variety of complex value subrange signals. These are fed to the sub-range processing unit 1402. The large variety of complex-valued output sub-ranges are fed to the synthesis filter bank 1403, which in turn outputs the modified time domain signal. The sub-track processing unit 1402 performs non-linear sub-track processing operations on a block basis so that the modified time domain signal is a transposed version of the input signal corresponding to a transposition order T>1. The concept of a comprising non-linear operations on blocks of more than one sub-band sample at a time, where subsequent blocks are windowed and overlapped added to generate the sub-band output signals.

Filter banks 1401 and 1403 can be of any complex exponential modulated type such as QMF or windowed DFT. They can be stacked the same or differently in modulation and can be defined from a wide range of prototype or window filters. It is important to know the quotient Δ/s / of the two parameters of the filter bank, measured in physical units. • ΔfA : the frequency spacing of the subband frequency of the analysis filter bank 1401; • ΔfS : the frequency spacing of the subband of the synthesis filter bank 1403.

. Uma sinusoide de saida da frequência fisica desejada transposta T•Ω. resultará da inserção da subfaixa da sintese com indice

. Assim, os valores do indice de subfaixa fonte apropriados do processamento da subfaixa para um determinado indice de subfaixa alvo m devem obedecer

In order to configure the processing of the subrange 1402 it is necessary to find the correspondence between the indices of the source and target subrange. It is observed that an input sinusoid of the physical frequency Ω will result in a main contribution occurring in the input subbands with index

. An output sinusoid of the desired physical frequency transposed T•Ω. will result from the insertion of the synthesis subrange with index

. Thus, the appropriate source subrange index values from subrange processing for a given target subrange index m must conform to

Fig. 15 illustrates an exemplary scenario for applying subband block-based transposition using various transposition orders in an HFR enhanced audio codec. A transmitted data stream is received at the central decoder 1501, which provides a low-bandwidth decoded central signal at a sampling frequency fs. The low frequency is resampled at the output sampling frequency 2fs by means of a complex modulated 32-band QMF analysis bank 1502 followed by a 64-band QMF synthesis bank (inverse QMF) 1505. The two filter banks 1502 and 1505 have the same physical resolution parameters Δfs=ΔfA and the HFR processing unit 1504 simply leaves through the unmodified lower sub-bands corresponding to the low signal of the center bandwidth. The high frequency content of the output signal is obtained by inserting the higher subbands of the 64-band QMF synthesis bank 1505 with the output bands of the multiple transposition unit 1503, subject to spectral shape and modification performed by the processing unit. of HFR 1504. Multiple transposition 1503 takes the decoded center signal as input and outputs a wide variety of subband signals representing the analysis of 64 QMF tracks of a superposition or combination of several transposed components. The objective is that if the HFR processing is bypassed, each component corresponds to a physical transposition of the integer number of the central signal, ( Z' = 2,3,... ).

Fig. 16 illustrates an exemplary prior art scenario for operating a multi-order transposition based on subrange block 1603 by applying a separate analysis filter bank by transposition order. Here three transposition orders T = 2,3,4 will be produced and managed in the domain of a 64-band QMF operating at the output sampling rate 2fs. The 1604 splice unit simply selects and combines the relevant subranges from each branch of the transpose factor into a single wide variety of QMF subranges to be fed into the HFR processing unit.

Consider first the case T = 2 . The objective is specifically that the processing stream of a 64-lane QMF parse 1602-2, a sub-band processing unit 1603-2, and a 64-lane QMF synthesis 1505 results in a physical transposition of T = 2 . The identification of these three blocks with 1401, 1402 and 1403 of figure 14, one considers that and Δfs/ΔfA=2 so that (1) results in the specification for 1603-2 that the correspondence between the source n and target subranges m is determined by n = m.

For the T = 3 case, the exemplary system includes a 1601-3 sample rate converter that converts the low input sample rate by a factor of 3/2 from fs to 2fs/3. The objective is specifically that the processing stream of the 64-lane QMF parsing 1602-3, the sub-band processing unit 1603-3, and the 64-lane QMF synthesis 1505 result in a physical transposition of T = 3 . The identification of these three blocks with 1401, 1402 and 1403 of figure 14, one considers due to resampling that Δfs/ΔfA=3 so that (1) provides the specification for 1603-3 that the correspondence between source subranges n and target subranges m is again determined by n=m.

For the T=4 case, the exemplary system includes a 1601-4 sample rate converter that converts the low input sample rate by a factor of two from fs to fs/2. The objective is specifically that the processing stream of the 64-lane QMF parsing 1602-4, the sub-band processing unit 1603-4, and the 64-lane QMF synthesis 1505 results in a physical transposition of 71 = 4. identification of these three blocks with 1401, 1402 and 1403 of figure 14, one considers due to resampling that Δfs/ΔfA=4 so that (1) provides the specification for 1603-4 that the correspondence between source subranges n and target subranges m also is given by n = m.

Figure 17 illustrates an inventive example scenario for efficiently operating a subrange block-based multi-order transposition by applying a single 64-range QMF analysis filter bank. In fact, the use of three separate QMF parse banks and two sample rate converters in Figure 16 results in relatively high computational complexity as well as some implementation disadvantages for frame-based processing due to rate conversion. sampling 1601-3. Current realizations teach to replace the two branches 1601-3 -> 1602-3 -> 1603-3 and 1601-4 -> 1602-4 -> 1603-4 by processing the subtrack 1703-3 and 1703-4, respectively, in that branch 1602-2 -> 1603-2 is kept unchanged compared to figure 16. All three transposition orders will now have to be performed in a filter bank domain with reference to figure 14, where Δfs/ΔfA=2 . For the case T = 3, the specification for 1703-3 given by (1) is that the correspondence between the source subranges n and the target subranges m is given by n≈2m/3 . For the case 7 = 4, the specification for 1703-4 given by (1) is that the correspondence between the source subranges n and the target subranges m is given by n≈2m . To further reduce the complexity, some transposition orders can be generated by copying the already calculated transposition orders or the central decoder output.

Figure 1 illustrates the operation of a subband block-based transposition using transposition orders of 2, 3, and 4 in an HFR-enhanced decoder structure such as SBR [ISO/IEC 14496-3:2009, "Information technology - Coding of audio-visual objects - Part 3: Audio]". The data stream is time domain decoded by the central decoder 101 and passed to the HFR module 103, which generates a high frequency signal from the central baseband signal. After generation, the signal generated by HFR is dynamically adjusted to match the original signal as closely as possible by means of additional transmitted information. This adjustment is performed by the HFR processor 105 on the subrange signals obtained from one or several QMF analysis banks. A typical scenario is where the core decoder operates on a time domain signal sampled at half the frequency of the input and output signals, i.e. the HFR decoder module will effectively resampling the core signal at twice the sampling frequency. This sample rate conversion is generally achieved by first filtering the signal from the central encoder by means of a 32-band QMF analysis bank 102. The sub-bands below are called cross-frequency, i.e., the lowest subset of the 32 sub-bands that contain all the energy of the central encoder signal is combined with the set of sub-bands that carry the signal generated by HFR. Generally, the number of sub-bands combined is 64, which, after filtering through the QMF synthesis bank 106, results in a signal converted from the sample rate of the central encoder combined with the output of the HFR module.

In transposition based on the subband block of the HFR module 103, three transposition orders T = 2, 3 and 4 must be produced and managed in the domain of a 64-band QMF operating at the output sampling rate 2fs. The input time domain signal is filtered with the low pass filter in blocks 103-12, 103-13 and 103-14. This is done so that the output signals, processed by the different transposition orders, have non-overlapping spectral contents. The signals are further downsampled (103-23, 103-24) to adapt the sampling rate of the input signals to the analysis filter bank of a constant size (in this case 64). It can be seen that the increase in sample rate from fs to 2fs can be explained by the fact that sample rate converters use sample rate reduction factors of T/2 instead of T, in which the latter would result in the transposed subband signals having the same sampling rate as the input signal. The reduced sample rate signals are fed to separate HFR analysis filter banks (103-32, 103-33, and 103-34), one for each transposition order, which provide a wide variety of subrange signals with complex value. These are entered at the non-linear subrange extension units (103-42, 103-43, and 103-44). The large variety of complex-valued output subranges are fed into the Merge/Combine Module 104 with the output of the subsampled analysis bank 102. The Merge/Merge unit simply merges the subranges of the central analysis filter bank 102 and each extension factor is subdivided into a single wide variety of QMF sub-ranges to be fed into the HFR 105 processing unit.

When the spectral signal of different transposition orders is set to not overlap, i.e., the 1st order spectrum of the transposition signal should start where the spectrum of the ends of the T-1 order signal, the transposed signals must be low-pass. . Thus, the traditional low-pass filters 103-12-103-14 in figure 1. However, through a simple exclusive selection among the sub-ranges available by the Join/Combine unit 104, the separate low-pass filters are redundant and can be avoided. . Also, the inherent lowpass feature provided by the QMF bank is explained by inserting different contributions from the transpose branches independently of the different subtrack channels in 104. It is also sufficient to apply the time span only to the tracks that are combined in 104 .

Figure 2 illustrates the operation of a non-linear sub-range extension unit. Block extractor 201 tests a finite structure of complex value input signal samples. The structure is defined by an indicator position of the input. This structure undergoes nonlinear processing at 202 and is subsequently windowed by a finite length window at 203. The resulting samples are added to the samples previously emitted in the superposition and addition unit 204 where the position of the output structure is defined by an exit indicator position. The entry indicator is increased by a fixed amount and the exit indicator is increased by the subrange extension factor times the same amount. An iteration of this chain of operations will produce an output signal with duration being the subrange extension factor times the duration of the input subrange signal, up to the length of the synthesis window.

While the SSB transposition employed by SBR [ISO/IEC 14496-3:2009, "Information technology - Coding of audiovisual objects - Part 3: Audio]" typically explains the entire baseband, excluding the first subband, to generate the high band, a harmonic transposition generally uses a smaller part of the spectrum of the center encoder. The amount used, the so-called source range, depends on the transposition order, the bandwidth extension factor, and the rules applied to the combined result, e.g. whether signals generated from different transposition orders are allowed to overlap in a way spectral or not. As a consequence, only a limited part of the harmonic transposition output spectrum for a given transposition order will currently be used by the HFR 105 module processing.

Fig. 18 illustrates another embodiment of an exemplary processing implementation for processing a single subband signal. The single subband signal was subjected to any kind of decimation before or after being filtered by an analysis filter bank not shown in figure 18. In this way, the time duration of the single subband signal is shorter than. the length of time before forming the decimation. The single subrange signal is fed to a block extractor 1800, which may be identical to the block extractor 201, but which may also be implemented differently. Block extractor 1800 in Fig. 18 operates using an exemplary block advance value/sample called e. The sample/block advance value can be variable or can be fixedly defined and is illustrated in Fig. 18 as an arrow on the block extractor box 1800. At the block extractor 1800 output, there are a plurality of extracted blocks. These blocks are highly overlapping, as the sample/feed value of the block e is significantly less than the block length of the block extractor. An example is that the block extractor extracts blocks from 12 samples. The first block comprises samples 0 to 11, the second block comprises samples 1 to 12, the third block comprises samples 2 to 13, and so on. In this embodiment, the sample/block feed value e is equal to 1, and there is an overlap of 11 folds.

The individual blocks are fed into a windower 1802 for windowing the blocks using a window function for each block. Additionally, an 1804 phase calculator is provided, which calculates a phase for each block. The 1804 phase calculator can use either the individual block before windowing or after windowing. Then, a p x k phase trim value is calculated and input to a 1806 phase regulator. The phase regulator applies the trim value to each sample in the block. Also, the k factor is equal to the bandwidth extension factor. When, for example, bandwidth extension by a factor of 2 is obtained, then the calculated p-phase for a block extracted by block extractor 1800 is multiplied by factor 2 and the adjustment value applied to each block sample at the regulator of phase 1806 is p multiplied by 2. This is an exemplary value/rule. Alternatively, the phase corrected for the synthesis is k * p, p + (k-l)*p.

So in this example the correction factor is 2 if multiplied or l*p if summed. Other values/rules can be applied to calculate the phase correction value.

In one embodiment, the single subrange signal is a complex subrange signal, and the phase of a block can be calculated in a plurality of different ways. One way is to consider the sample in the middle or around the middle of the block and calculate the phase of this complex sample. It is also possible to calculate the phase for each sample.

Although it is illustrated in figure 18 in the way that a phase regulator operates after the windower these two blocks can also be swapped, so that the phase adjustment is performed to the blocks extracted by the block extractor and a subsequent windowing operation is performed. . Since both operations, i.e., windowing and phase adjustment, are real-valued or complex-valued multiplications, these two operations can be summarized in a single operation using a complex multiplication factor, which is the product of a phase adjustment multiplication factor and a windowing factor.

Phase-adjusted blocks are inserted into an overlay/sum and amplitude correction block 1808, where phase-adjusted or windowed blocks are overlap-summed. Importantly, however, the sample/block advance value in block 1808 is different from the value used in block extractor 1800. Particularly, the sample/block advance value in block 1808 is greater than the value e used in the block. 1800, so that a time span of the signal emitted by block 1808 is obtained.

Thus, the processed subrange signal output by block 1808 has a length that is longer than the subrange signal input to block 1800. When the bandwidth span of two is obtained, then the sample/block advance value is used, which is twice the value in the corresponding block 1800. This results in a time extension by a factor of two. When, however, other time extension factors are required, then another sample/block advance value can be used so that the output of block 1808 has a required time duration.

To address the issue of overlap, an amplitude correction is preferably performed to address the issue of different overlaps in block 1800 and 1808. This amplitude correction could, however, also be introduced to the windower/phase regulator multiplication fact, but amplitude correction can also be performed after overlaying/processing.

In the example above with a block length of 12 and a sample/block advance value in the block extractor of one, the sample/block advance value for the overlay block/sum 1808 would be equal to two when an extension bandwidth by a factor of two is performed. This would still result in a five-block overlap. When a bandwidth extension by a factor of three is to be performed, then the sample/block advance value used by block 1808 would equal three, and the overlap would fall to an overlap of three. When a four-fold bandwidth extension is to be performed, then the 1808 overlap/sum block would have to use a sample/block advance value of four, which would still result in an overlap of more than two blocks.

Great computational savings can be obtained by restricting the input signals to the transposition branches to contain only the source track, and this at a sampling rate adapted to each transposition order. The basic block scheme of such a system for an HFR generator based on the subband block is illustrated in Figure 3. The input signal from the central encoder is processed by the dedicated sample rate reducers before the HFR analysis filter bank. .

The essential effect of each sample rate reducer is to filter the signal from the source track and bring it to the analysis filter bank at the lowest possible sample rate. Here, lowest possible refers to the lowest sampling rate that is still suitable for downstream processing, not necessarily the lowest sampling rate that avoids aliasing after decimation. Sample rate conversion can be achieved in several ways. Without limiting the scope of the invention, two examples will be given: the first shows the resampling performed by multirate time domain processing, and the second illustrates the resampling achieved by means of QMF subband processing.

Figure 4 shows an example of the blocks in a multirate time domain sample rate reducer for a transposition order of 2. The input signal, having a bandwidth B Hz, and a sampling frequency fs, is modulated. by a complex exponential (401) for frequency switching to start the source band to the DC frequency as

onde a fração foi reduzida pelo fator comum 8. Assim, o fator de interpolação é 3 (como visto da figura 5(c)) e o fator de dizimação é 8. Pela utilização de Identidades Nobres ["Multirate Systems And Filter Banks," P.P. Vaidyanathan, 1993, Prentice Hall, Englewood Cliffs], o decimador pode ser movido por todo o caminho para a esquerda, e o interpolador todo o caminho para a direita na figura 4. Desta forma, a modulação e a filtragem são feitas sobre a taxa de amostragem mais baixa possivel e a complexidade computacional é ainda reduzida.Examples of an input signal and the spectrum after modulation is described in figures 5(a) and (b). The modulated signal is interpolated (402) and filtered by a complex value low-pass filter with 0 and S/2 Hz passband limits (403). The spectra after the respective steps are shown in figures 5(c) and (d). The filtered signal is subsequently decimated (404) and the real part of the signal is calculated (405). The results after these steps are shown in figures 5(e) and (f). In this particular example, when T=2, B=0.6 (on a normalized scale, ie fs=2) , P2 is chosen as 24, to safely encompass the source range. The sampling rate reduction factor obtains

where the fraction was reduced by the common factor 8. Thus, the interpolation factor is 3 (as seen from Figure 5(c)) and the decimation factor is 8. By using Noble Identities ["Multirate Systems And Filter Banks," PP Vaidyanathan, 1993, Prentice Hall, Englewood Cliffs], the decimator can be moved all the way to the left, and the interpolator all the way to the right in figure 4. In this way, the modulation and filtering are done over the lowest possible sampling rate and the computational complexity is further reduced.

Another approach is to use the subrange outputs of the subsampled 32-range QMF analysis bank 102 already present in the HRF SBR method. The subranges that span the source ranges for the different transposition branches are synthesized in the time domain by the small QMF banks subsampled before the HFR analysis filterbank. This type of HFR system is illustrated in Figure 6. Small QMF banks are obtained by subsampling the original 64-band QMF bank, where the prototype filter coefficients are found by linear interpolation of the original prototype filter. Following the notation in Figure 6, the QMF synthesis bank before the 2nd order transposition branch has Q2=12 tracks (the subranges with zero-based indices from 8 to 19 in the 32-track QMF). To prevent aliasing in the synthesis process, the first (index 8) and last (index 19) ranges are set to zero. The resulting spectral output is shown in figure 7. Note that the transposition based on the block analysis filter bank has 202=24 bands, that is, the same number of bands as in the example based on the sample rate reducer of multirate time domain (figure 3).

When figure 6 and figure 23 are compared, it is clear that element 601 of figure 6 corresponds to analysis filter bank 2302 of figure 23. Furthermore, synthesis filter bank 2304 of figure 23 corresponds to element 602- 2, and the other analysis filter bank 2307 of Fig. 23 corresponds to element 603-2. Block 604-2 corresponds to block 2309 and combiner 605 may correspond to synthesis filter bank 2311, but in other embodiments, the combiner may be configured to output subrange signals and then the other synthesis filter bank connected to the combiner can be used. However, depending on the implementation, a given high frequency reconstruction as discussed in the context of Fig. 26 later may be performed before synthesis filtering by the synthesis filter bank 2311 or combiner 205, or it may be performed after synthesis filtering in the bank. of synthesis filter 2311 of figure 23 or after the combiner in block 605 of figure 6.

The other branches that span from 602-3 to 604-3 or that span from 602-T to 604-T are not illustrated in Figure 23, but can be implemented similarly, but with different filter bank sizes where T in figure 6 corresponds to a transposition factor. However, as discussed in the context of Figure 27, transposition by a transposition factor of 3 and transposition by a transposition factor of 4 can be introduced to the processing branch consisting of element 602-2 to 604-2 so that block 604-2 not only provides a transposition by a factor of 2, but also a transposition by a factor of 3 and a factor of 4, with a given synthesis filter bank is used as discussed in the context of figures 26 and 27 .

In figure 6 of the embodiment, Q2 corresponds to Ms and Ms is equal to, for example, 12. Furthermore, the size of the other analysis filter bank 603-2 corresponding to element 2307 is equal to 2MS as 24 in the embodiment.

Also, as described above, the lowest subrange channel and highest subrange channel of the synthesis filter bank 2304 can be entered with zeros to avoid aliasing problems.

The system described in Figure 1 can be viewed as a simplified special case of the resampling described in Figures 3 and 4. To simplify the arrangement, the modulators are omitted. Furthermore, all HFR analysis filtering is achieved using a 64-band analysis filter bank. Thus, P2 = P3 = P4 = 64 of figure 3, and the sampling rate reduction factors are 1, 1.5, and 2 for the 2nd, 3rd, and 4th transposition order branches respectively.

It is an advantage of the present invention that in the context of inventive critical sampling processing, subband signals from the 32-track QMF analysis bank corresponding to block 2302 of Fig. 23 or 601 of Fig. 6 as defined in MPEG4 (ISO/IEC 14496 -3) can be used. The definition of this analysis filter bank in the MPEG-4 Standard is illustrated at the top of figure 25a and is illustrated as a flowchart in figure 25b, which is also considered from the MPEG-4 Standard. The SBR (spectral bandwidth reproduction) of this standard is incorporated herein by reference. Particularly, the parsing filter bank 2302 of Fig. 23 or the 32-lane QMF 601 of Fig. 6 can be implemented as illustrated in Fig. 25a, the top and flowchart in Fig. 25b.

Furthermore, the synthesis filter bank illustrated in block 2311 of figure 23 can also be implemented as indicated at the bottom of figure 25a and as illustrated in the flow chart of figure 25c. However, any other filter bank definitions can be applied, but at least for the analysis filter bank 2302, the implementation illustrated in Figures 25a and 25b is preferred due to the robustness, stability and high quality provided by this filter bank of analysis having 32 MPEG-4 channels at least in the context of bandwidth extension applications such as spectral bandwidth reproduction, or generally indicated, the applications of high frequency reconstruction processing.

The 2304 synthesis filter bank is configured to synthesize a subset of the sub-tracks that span the source track for a transposition. This synthesis is done to synthesize the intermediate signal 2306 in the time domain. Preferably, the synthesis filter bank 2304 is a small QMF bank with subsampled actual value.

The time domain 2306 output of this filter bank is then fed into a QMF parsing bank with a complex value of twice the size of the filter bank. This QMF bank is illustrated by block 2307 of Fig. 23. This procedure allows for substantial savings in computational complexity, as only the relevant source track is transformed into the QMF sub-band domain having the frequency resolution doubled. The small QMF banks are obtained by subsampling the original 64-band QMF bank, where the prototype filter coefficients are obtained by linear interpolation of the original prototype filter. Preferably, the prototype filter associated with the MPEG-4 synthesis filter bank having 640 samples is used, where the MPEG-4 parsing filter bank has a window of 320 window samples.

onde Ms é o tamanho do banco de filtro de sintese subamostrado e kL representa o indice da subfaixa do primeiro canal do banco de QMF de 32 faixas para inserir o banco de filtro de sintese subamostrado. A matriz startSubband2kI> é listada na tabela 1. A função floor{x} circula o argumento x ao número inteiro mais próximo em direção a infinidade negativa. Tabela 1 - y = startSubband2kL(x)

The processing of the subsampled filterbanks is described in Figures 24a and 24b, which illustrate the flowcharts. The following variables are determined first:

where Ms is the size of the subsampled synthesis filter bank and kL represents the subrange index of the first channel of the 32-range QMF bank to enter the subsampled synthesis filter bank. The startSubband2kI> array is listed in Table 1. The floor{x} function loops the argument x to the nearest integer towards negative infinity. Table 1 - y = startSubband2kL(x)

Thus, the value Ms defines the size of the synthesis filter bank 2304 of figure 23 and KL is the first channel of the subset 2305 indicated in figure 23. Specifically, the value in the ftatieLow equation is defined in ISO/IEC 14496-3, section 4.6.18.3.2 which is also incorporated herein by reference. It should be noted that the Ms value is incremented by 4, which means that the size of the 2304 synthesis filter bank can be 4, 8, 12, 16, 20, 24, 28, or 32.

Preferably, synthesis filterbank 2304 is a real-valued synthesis filterbank. For this purpose, a set of subrange samples with real value Ms is calculated from subrange samples with new complex value Ms according to the first step of Fig. 24a. For this purpose, the equation is used

In the equation, exp() denotes the exponential complex function, i is the imaginary unit, and kL was defined earlier. • Alternate samples in matrix v through 2MS positions. Older 2MS samples are discarded. • The subrange samples with value in real M$ multiplied by the matrix N, that is, the matrix-vector product NV is calculated, where

onde

e

são definidos como o número inteiro e as partes fracionais de

, respectivamente. Os coeficientes da janela de c podem ser encontrados na tabela 4.A.87 do ISO/IEC 14496-3:2009.The output of this operation is stored in positions 0 to 2MS-1 of the matrix v. • Extract samples from v, according to the flowchart in figure 24a, to create the matrix g of element 10Ws. • Multiply the samples of matrix g by window Ci to produce matrix w. The window coefficients c± are obtained by the linear interpolation of the coefficients c, that is, through the equation

Where

and

are defined as the whole number and the fractional parts of

, respectively. The c window coefficients can be found in Table 4.A.87 of ISO/IEC 14496-3:2009.

Thus, the synthesis filter bank has a working prototype window calculator to calculate a prototype window function by subsampling or interpolation using a window function stored for a filter bank having a different size.

Compute the new output samples Ms by summing samples from the matrix w according to the last step in the flowchart of figure 24a.

onde

e

são definidos como o número inteiro e partes fracionais de

, respectivamente. Os coeficientes da janela de c podem ser encontrados na tabela 4.A.87 da ISO/IEC 14496-3:2009.Subsequently, the preferred implementation of the other analysis filter bank 2307 in Fig. 23 is illustrated with the flow chart in Fig. 24b. • Alternate the samples in the matrix x through the 2MS positions according to the first step of figure 24b. Older 2MS samples are discarded and new 2MS samples are stored in positions 0 to 2MS-1. • Multiply the x matrix samples by the c2i window coefficients. The c2i window coefficients are obtained by the linear interpolation of the c coefficients, that is, through the equation

Where

and

are defined as the whole number and fractional parts of

, respectively. The c window coefficients can be found in table 4.A.87 of ISO/IEC 14496-3:2009.

Thus, the other analysis filter bank 2307 has a working prototype window calculator to calculate a prototype window function by subsampling or interpolation using a stored window function for a filter bank having a different size. • Sum the samples according to the formula in the flowchart in figure 24b to create the matrix u of the 4MS element. • Compute the new subrange samples with new complex value 2MS by matrix-vector multiplication Mu, where

In the equation, exp() denotes the exponential complex function, and i is the imaginary unit.

A block diagram of a factor 2 sample rate reducer is shown in figure 8(a). The real-valued low-pass filter can be written H(z) = B(z)/A(z) , where B(z) is the non-recursive part (FIR) and A(z) is the recursive part (IIR ).

onde

However, for an efficient implementation, using Noble Identities to reduce computational complexity, it is useful to design a filter where all poles have multiplicity 2 (double poles) as A(z2). Thus, the filter can be factored as shown in figure 8(b). Using Noble Identity 1, the recursive part can be moved after the decimator as in figure 8(c). The non-recursive filter B(z) can be implemented using standard 2-component polyphase decomposition as

Where

Thus, the sample rate reducer can be structured as in figure 8(d). After using Noble Identity 1, the FIR part is calculated at the lowest possible sample rate as shown in Figure 8(e) . From Figure 8(e) it is easy to see that the FIR operation (delay, decimators and polyphase components) can be seen as a window sum operation using a two-sample input advance. For two input samples, a new output sample will be produced, effectively resulting in a reduction of the sampling rate by a factor of 2.

A block diagram of the sample rate reducer with the factor 1.5=3/2 is shown in figure 9(a). The real-valued low-pass filter can again be written H(z) - B(z)/A(z) where B(z) is the non-recursive part (FIR) and A(z) is the recursive part (IIR) ) As before, for an efficient implementation, using Noble Identity to reduce computational complexity, it is useful to design a filter where all poles have multiplicity 2 (double poles) or multiplicity 3 (triple poles) like A(z2) or A( z3) respectively. Here, dual poles are chosen as the design algorithm for the low pass filter is more efficient, although the recursive part is actually 1.5 times more complex to implement compared to the triple pole approach. Thus, the filter can be factored as shown in figure 9(b) . Using Noble Identity 2, the recursive part can be moved in front of the interpolator as in figure 9(c) . The non-recursive filter B(z) can be implemented using the polyphase decomposition of the standard component 2-3 = 6 as

enquanto as amostras com indice impar sao calculadas do grupo mais alto

. A operação de cada grupo (corrente de atraso, decimadores e componentes polifase) pode ser vista como a operação de soma pela janela utilizando um avanço de entrada de três amostras. Os coeficientes da janela utilizados no grupo superior são coeficientes com indice impar, enquanto o grupo inferior utiliza os coeficientes com indice par do filtro original B(z). Assim, para um grupo de três amostras de entrada, duas novas amostras de saida serão produzidas, efetivamente resultando em uma redução da taxa de amostragem de um fator 1,5.Thus, the sample rate reducer can be structured as in figure 9(d). After using Noble Identity 1 and 2, the FIR part is calculated at the lowest possible sample rate as shown in Figure 9(e). From figure 9(e) it is easy to see that the even-indexed output samples are calculated using the lower group of three polyphase filters

while odd-indexed samples are calculated from the highest group

. The operation of each group (delay current, decimators, and polyphase components) can be seen as the window sum operation using a three-sample input advance. The window coefficients used in the upper group are coefficients with odd index, while the lower group uses the coefficients with even index from the original filter B(z). Thus, for a group of three input samples, two new output samples will be produced, effectively resulting in a reduction of the sampling rate by a factor of 1.5.

The time domain signal from the central decoder (101 in Fig. 1) can also be subsampled using a smaller subsampled synthesis transform at the central decoder. The use of a smaller synthesis transformation also offers reduced computational complexity. Depending on the cross frequency, i.e. the bandwidth of the center encoder signal, the index of the synthesis transform size and the nominal size Q (Q < 1), results in a center encoder output signal having a rate of sampling Qfs. In order to process the subsampled central encoder signal in the examples described in the current application, all the analysis filter banks in Fig. of the sample rate (301-2, 301-3 and 301-T) of figure 3, the decimator 404 of figure 4 and the analysis filter bank 601 of figure 6. Apparently, Q has to be chosen so that all the filter bank sizes are integers.

Figure 10 illustrates the alignment of the spectral edges of the HFR transposition signals to the spectral edges of the envelope fit frequency table in an improved HFR encoder such as SBR [ISO/IEC 14496-3:2009, "Information technology - Coding of audio-visual objects - Part 3: Audio]". Figure 10(a) shows a stylistic graph of the low frequencies comprising the envelopment fit table, the so-called scaled factor ranges, which span the frequency range from the crossover frequency kx to the stop frequency ks. The scaled factor bands constitute the frequency grid used in an improved HFR encoder when adjusting the energy level of the high regenerated band frequency, i.e. the frequency envelope. To adjust the envelope, the signal energy is averaged over a block of time/frequency constrained by the edges of the scaled factor range and the edges of the selected time. If the signals generated by the different transposition orders are misaligned to the 15 scale factor bands, as illustrated in Figure 10(b), artifacts can arise if the spectral energy changes drastically near an edge of the transposition band, as the The wraparound adjustment process will keep the spectral structure within a range of the scaled factor. Thus, the proposed solution is to adapt the frequency edges of the transposed signals to the edges of the scaled factor bands as shown in figure 10(c). In the figure, the top edge of the signals generated by transposition orders 2 and 3 (T=2, 3) is reduced by a small amount, compared to figure 10(b), to align the frequency edges of the transposition bands. to the edges of the existing scale factor range.

A realistic scenario that shows the potential artifacts when using misaligned edges is described in the figure.

Figure 11(a) again shows the edges of the scaled factor range. Fig. 11(b) shows the generated non-HFR-adjusted signals of transposition orders T=2, 3 and 4 with the center decoded baseband signal. Figure 11(c) shows the envelope-adjusted signal when an envelope target plane is assumed. Blocks with squared areas represent scaled factor ranges with high intra-band energy variations, which can cause output signal anomalies.

Figure 12 illustrates the scenario of Figure 11, but this time using aligned edges. Figure 12(a) shows the edges of the scaled factor range, Figure 12(b) depicts the generated unadjusted signals by the HFR of transposition orders T=2, 3 and 4 with the baseband signal decoded center and , in line with figure 11(c), figure 12(c) shows the envelope-adjusted signal when an envelope target plane is assumed. As seen from the figure, there are no scaled factor bands with high-energy intraband variations due to misalignment of the transposed signal bands and the scaled factor bands, and thus potential artifacts are reduced.

Figure 13 illustrates the adaptation of the edges of the HFR limiting range, as described in, for example, SBR [ISO/IEC 14496-3:2009, "Information technology - Coding of audiovisual objects - Part 3: Audio]" to harmonic repairs in an improved HFR encoder. The limiter operates in frequency bands having a coarser resolution than the scale factor bands, but the principle of operation is the same. In the limiter, an average gain value for each of the limiting bands is calculated. The individual gain values, that is, the envelope gain values calculated for each of the scaled factor ranges, are not allowed to exceed the limiting average gain value by more than a certain multiplicative factor. The purpose of the limiter is to supply large variations of the scale factor range gains within each of the limiting ranges. While adapting the ranges generated by transposition to the ranges of the scale factor ensures small variations of intraband energy within a range of the scale factor, the adaptation of the edges of the limiting range to the edges of the transposition range, in accordance with the present invention, handles the largest differences in energy at scale between tracks processed by transposition. Fig. 13(a) shows the frequency limits of the signals generated by the HFR of transposition orders T=2, 3 and 4. The energy levels of the different transposed signals can be substantially different. Figure 13(b) shows the frequency bands of the limiter which are typically of constant width on a logarithmic frequency scale. The edges of the transposition frequency range are summed as constant bounding edges and the remaining bounding edges are recalculated to keep the logarithmic relationships as close as possible, as, for example, illustrated in Figure 13(c). Although some aspects have been described in the context of an apparatus, it is clear that these aspects also represent a description of the corresponding method, where a block or device corresponds to a method step or a characteristic of a method step. Analogously, the aspects described in the context of a method step also represent a description of a corresponding block or item or characteristic of a corresponding apparatus.

Other embodiments employ a mixed repair scheme which is shown in Fig. 21, where the mixed repair method within a time block is performed. For full coverage of the different regions of the HF spectrum, BWE comprises several fixes. In HBE, higher repairs require high transposition factors within the phase vocoders, which particularly deteriorate the perceptual quality of transients.

Thus, the realizations generate the higher-order repairs that occupy the upper spectral regions, preferably by calculating the efficient repair of the SSB copy, and the lower-order repairs that cover the middle spectral regions, for which harmonic structure preservation is desired. , preferably by HBE repair. The individual mix of repair methods can be static over time or, preferably, be signaled in the data stream.

For copying operation, low frequency information can be used as shown in figure 21. Alternatively, repair data that was generated using HBE methods can be used as illustrated in figure 21. The latter leads to a less dense tonal structure for the highest repairs. In addition to these two examples, every combination of copy and HBE is conceivable. The advantages of the proposed concepts are • Improved perceptual quality of transients Reduced computational complexity

Fig. 26 illustrates a preferred processing stream for bandwidth extension purposes, where different processing operations can be performed within the non-linear processing of the indicated sub-band in blocks 1020a, 1020b. The cascade of filter banks 2302, 2304, 2307 is represented in Fig. 26 by block 1010. Furthermore, block 2309 may correspond to elements 1020a, 1020b and the wraparound regulator 1030 may be placed between block 2309 and block 2311 of the Fig. 23 or can be placed after processing in block 2311. In this implementation, band-selective processing of the time domain signal processed as the bandwidth-extended signal is performed in the time domain instead of the sub-band domain, that exists before the synthesis filter bank 2311.

Fig. 26 illustrates an apparatus for generating an extended audio bandwidth signal from a low band input signal 1000 according to another embodiment. The apparatus comprises an analysis filter bank 1010, a subrange wide nonlinear subrange processor 1020a, 1020b, a subsequently connected envelope regulator 1030 or, generally indicated, a high frequency reconstruction processor operating on the reconstruction of the high frequency parameters. as, for example, input on the line of parameter 1040. The wraparound regulator, or as generally indicated, the high frequency reconstruction processor processes individual subband signals for each subband channel and inputs the processed subband signals for each subband channel in a synthesis filter bank 1050. The synthesis filter bank 1050 receives, in its lower channel input signals, a sub-band representation of the center signal from the low-band decoder. Depending on the implementation, the lowband can also be derived from the outputs of the analysis filter bank 1010 in figure 26. The transposed subband signals are fed to the highest filter bank channels of the synthesis filter bank to perform the reconstruction of high frequency.

Filterbank 1050 finally outputs a transposition output signal comprising the bandwidth spans by transposition factors 2, 3, and 4, and the signal output by block 1050 is not bandwidth-limited cross-frequency, that is, the highest frequency of the central encoder signal corresponding to the lowest frequency of SBR or the signal generated by HFR components.

In the embodiment of Fig. 26, the analysis filterbank performs twice oversampling and has a certain analysis subrange spacing 1060. The synthesis filterbank 1050 has a synthesis subrange spacing 1070 which is, in this embodiment, the twice the size of the sub-analysis spacing which results in a transposition contribution as will be discussed later in the context of Figure 27.

Fig. 27 illustrates a detailed implementation of a preferred embodiment of a non-linear sub-range processor 1020a in Fig. 26. The circuit illustrated in Fig. 27 receives as an input a single sub-range signal 108, which is processed in three "branches": A upper branch 110a is for a transposition by a transposition factor of 2. The middle branch of figure 27 indicated at 110b is for a transposition by a transposition factor of 3, and the lower branch in figure 27 is for a transposition by a transposition factor of 4 and is indicated by the reference numeral 110c. However, the actual transposition obtained by each processing element in Fig. 27 is only 1 (ie, no transposition) for branch 110a. The current transposition obtained by the processing element illustrated in Figure 27 for the middle branch 110b is equal to 1.5 and the current transposition for the lower branch 110c is equal to 2. This is indicated by the numbers in parentheses to the left of Figure 27, where the transposition factors T are indicated. The transpositions of 1, 5 and 2 represent a contribution of the first transposition obtained having a decimation operation on branches 110b, 110c and a time extension by the overlap-add processor. The second contribution, i.e. double the transposition, is obtained by the synthesis filter bank 105, which has a synthesis subrange spacing 107 that is twice the analysis subrange filter bank spacing. In this way, since the synthesis filter bank has twice the parsing subrange spacing, any decimation functionality does not occur at branch 110a.

Branch 110b, however, has a decimation functionality to get a transposition by 1.5. Due to the fact that the synthesis filter bank has twice the physical subrange spacing of the analysis filter bank, a transposition factor of 3 is obtained as indicated in figure 27 to the left of the block extractor for the second branch 110b .

Analogously, the third branch has a decimation functionality corresponding to a transposition factor of 2, and the final contribution of different subrange spacing in the analysis filter bank and the synthesis filter bank finally corresponds to a transposition factor of 4. of the third branch 110c.

Particularly, each branch has a block extractor 120a, 120b, 120c and each of these block extractors can be similar to the block extractor 1800 of Fig. 18. In addition, each branch has a phase calculator 122a, 122b and 122c, and the phase calculator can look like the 1804 phase calculator of figure 18. Also, each branch has a phase regulator 124a, 124b, 124c and the phase regulator can look like the 1806 phase regulator of figure 18. Also , each branch has a windower 126a, 126b, 126c, where each of these windowers can be similar to windower 1802 in figure 18. However, windowers 126a, 126b, 126c can also be configured to apply a rectangular window with a certain "zero padding" ". The transposition or repair signals from each branch 110a, 110b, 110c, in the embodiment of Fig. 27, are fed to the adder 128, which sums the contribution of each branch to the current subband signal to finally obtain the so-called transposition blocks in the output of adder 128. Then, an overlay-add procedure on overlay adder 130 is performed, and overlay adder 130 may be similar to overlay/sum block 1808 of Fig. 18. Overlay adder applies a lead value 2-e overlap-add value, where e is the overlay advance value or "lead value" of the block extractors 120a, 120b, 120c, and the overlap adder 130 outputs the transposed signal that is, in performing the Fig. 27, a single sub-band output for the k channel, that is, for the currently observed sub-band channel. The processing illustrated in Figure 27 is performed for each analysis sub-range or for a given group of analysis sub-ranges and, as illustrated in Figure 26, transposed sub-range signals are fed into the synthesis filter bank 1050 after being processed by the block 1030 to finally obtain the transposition output signal illustrated in Fig. 26 at the output of block 1050.

In one embodiment, the block extractor 120a of the first transposition branch 110a extracts 10 subtrack samples and subsequently a conversion of these 10 samples from QMF to polar coordinates is performed. This output, generated by the phase regulator 124a, is then directed to the windower 126a, which extends the output by zeros to the first and last value of the block, where this operation is equivalent to a windowing (synthesis) with a rectangular window of extension 10. Block extractor 120a at branch 110a does not perform a decimation. In this way, samples extracted by the block extractor are mapped to an extracted block at the same sample spacing as they were extracted.

However, this is different for branches 110b and 110c. Block extractor 120b preferably extracts a block of 8 subrange samples and distributes these 8 subrange samples into the extracted block at a different subrange sample spacing. The non-integer subrange sample inputs for the extracted block are obtained by an interpolation, and the QMF samples obtained with the interpolated samples are converted to polar coordinates and processed by the phase regulator. Then again, windowing on windower 126b is performed to extend the block emitted by phase regulator 124b by zeros for the first two samples and the last two samples, which operation is equivalent to a windowing (synthesis) with the rectangular window of extension 8.

Block extractor 120c is configured to extract a block with a time span of 6 subrange samples and performs a decimation of a decimation factor of 2, performs a conversion of the QMF samples to polar coordinates, and again performs an operation on the phase 124b, and the output is again extended by zeros, however now for the first three subrange samples and for the last three subrange samples. This operation is equivalent to a (synthesis) windowing with a rectangular window of extent 6.

The transpose outputs of each branch are then summed to form the combined QMF output by adder 128, and the combined QMF outputs are finally superimposed using overlap-add at block 130, where the advance value of overlap-add or value of advance is twice the advance value of block extractors 120a, 120b, 120c as discussed earlier.

One embodiment comprises a method for decoding an audio signal using block-based sub-band harmonic transposition, comprising filtering a center decoded signal through an M-band analysis filter bank to obtain a set of sub-band signals; synthesizing a subset of said subband signals by means of subsampled synthesis filterbanks having a reduced number of subbands, to obtain the subsampled source band signals.

One embodiment relates to a method for aligning the spectral edges of the range of signals generated by the HFR to the spectral edges used in a parametric process.

One embodiment relates to a method for aligning the spectral edges of the signals generated by the HFR to the spectral edges of the envelope fit frequency table comprising: searching for the highest edge in the envelope fit frequency table that does not exceed the fundamental limits of the envelope fit. bandwidth of the signal generated by HFR of the transposition factor T; and using the highest edge found as the frequency limit of the signal generated by HFR of the transposition factor T.

One embodiment relates to a method for aligning the spectral edges of the limiting tool to the spectral edges of the signals generated by the HFR comprising: adding the frequency edges of the signals generated by the HFR to the edge table used when creating the edges of the frequency range used by the limiting tool; and force the limiter to use the added frequency edges as constant edges and adjust the remaining edges correctly.

One embodiment relates to the combined transposition of an audio signal comprising several integer transposition orders into a domain of the low resolution filter bank where the transposition operation is performed on the time blocks of the subband signals.

Another realization concerns combined transposition, where transposition orders greater than 2 are embedded in an order 2 transposition environment.

Another realization refers to combined transposition, where transposition orders greater than 3 are embedded in an order 3 transposition environment, in which transposition orders less than 4 are performed separately.

Another embodiment relates to combined transposition, where transposition orders (e.g., transposition orders greater than 2) are created by reproducing previously calculated transposition orders (i.e., especially smaller orders) including the central coded bandwidth. Every conceivable combination of available transposition orders and bandwidth is possible without restrictions.

One realization refers to the reduction of computational complexity due to the reduced number of analysis filter banks that are needed for the transposition.

One embodiment relates to an apparatus for generating an extended bandwidth signal from an input audio signal, comprising: a repairman for repairing an incoming audio signal to obtain a repaired first signal and a recovered second signal the second recovered signal having a different repair frequency compared to the first repaired signal, wherein the first repaired signal is generated using a first repair algorithm, and the second recovered signal is generated using a second repair algorithm; and a combiner to combine the first repaired signal and the second recovered signal to obtain the extended bandwidth signal, and Another realization refers to this apparatus indeed, in which the algorithm of the first repair is a harmonic repair algorithm, and the 5 second repair algorithm is a non-harmonic repair algorithm.

Another embodiment relates to an earlier apparatus, in which the frequency of the first repair is less than the frequency of the second repair or vice versa.

Another embodiment concerns an earlier apparatus 10, in which the input signal comprises repair information; and wherein the repairman is configured to be controlled by the repair information extracted from the input signal to vary the first repair algorithm or the second repair algorithm in accordance with the repair information.

Another embodiment relates to a prior apparatus, in which the repairman is operative to repair subsequent blocks of audio signal samples, and in which the repairman is configured to apply the first repair algorithm and the second repair algorithm to the same block of audio. 20 audio samples.

Another embodiment relates to a prior apparatus, in which a repairman comprises, in arbitrary orders, a decimator controlled by a bandwidth extension factor, a filter bank, and an extender for a subband signal 25 of the filter bank. .

Another embodiment relates to a prior apparatus, in which the extender comprises a block extractor for extracting a number of overlapping blocks according to an extraction advance value; a phase regulator or windower to adjust subrange sample values in each block based on a window function or a phase correction; and an overlay/adder for performing an addition/overlay process of the phase- or window-fitted blocks using an overlay advance value greater than the extraction advance value.

Another embodiment relates to an apparatus for bandwidth spanning an audio signal comprising: a filter bank for filtering the audio signal to obtain signals with the sub-band reduced sampling rate; a plurality of different subband processors for processing different subband signals in different ways, the subband processors performing different time extension operations of the subband signal using different extension factors; and a union for joining the processed sub-tracks output from the plurality of different sub-track processors to obtain an extended audio bandwidth signal.

Another embodiment relates to an apparatus for reducing the sampling rate of an audio signal, comprising: a modulator; an interpolator using an interpolation factor; a complex low pass filter; and a decimator using a decimation factor, where the decimation factor is higher than the interpolation factor.

One embodiment relates to an apparatus for downsampling the sampling rate of an audio signal, comprising: a first filter bank for generating a plurality of sub-range signals of the audio signal, wherein a sampling rate of the sub-range signal is less than an audio signal sampling rate; at least one synthesis filter bank followed by an analysis filter bank to perform a sample rate conversion, the synthesis filter bank having a number of channels other than a number of channels of the analysis filter bank; a time extension processor for processing the sample rate converted signal; and a combiner for combining the time-extended signal and a low-band signal or a different time-extended signal.

Another embodiment relates to an apparatus for reducing the sampling rate of an audio signal by a non-integer sampling rate reduction factor, comprising: a digital filter; an interpolator having an interpolation factor; a polyphase element having odd and even offsets; and a decimator having a decimation factor being greater than the interpolation factor, the decimation factor and the interpolation factor being selected so that an index of the interpolation factor and the decimation factor is not an integer.

One embodiment relates to an apparatus for processing an audio signal, comprising: a core decoder having a synthesis transform size being less than a nominal transform size by a factor, such that an output signal is generated by the core decoder having a sampling rate less than a nominal sampling rate corresponding to the nominal transformation size; and a postprocessor having one or more filter banks, one or more time extenders, and a join, wherein a number of channels in the filter bank of one or more filter banks is reduced compared to a number as determined by the size. of nominal transformation.

Another embodiment relates to an apparatus for processing a lowband signal, comprising: a repair generator for generating various repairs using the audio lowband signal; an envelope regulator for adjusting an envelope signal using given scale factors for adjacent scaled factor ranges having edges of the scaled factor range, wherein the repair generator is configured to perform the various repairs such that an edge between the adjacent repairs coincides with an edge between the bands of the adjacent scaled factor on the frequency scale.

One embodiment relates to an apparatus for processing a lowband audio signal, comprising: a repair generator for generating various repairs using the lowband audio signal; and a wraparound trim limiter for limiting the wraparound trim values for a signal by limiting adjacent limiting ranges having edges of the limiting range, wherein the repair generator is configured to perform the various repairs such that an edge between adjacent repairs coincides with an edge between adjacent limiting bands on a frequency scale.

Inventive processing is useful for improving audio codecs that rely on a bandwidth extension scheme. Especially if optimal perceptual quality at a given bitrate is highly important and at the same time processing power is a limited resource.

Most prominent applications are audio decoders, which are generally implemented in portable i-devices and thus operate on a battery power supply.

The inventive encoded audio signal may be stored on a digital storage medium or may be transmitted on a transmission medium such as a wireless transmission medium or a wired transmission medium such as the Internet.

Depending on certain implementation requirements, embodiments of the invention can be implemented in hardware or in software. The implementation may be carried out using a digital storage medium, for example a floppy disk, a DVD, a CD, a ROM, a PROM, an EPROM, an EEPROM or a FLASH memory, having electronically readable control signals stored therein, that have cooperated (or may cooperate) with a programmable computer system so that the respective method is carried out.

Some embodiments according to the invention comprise a data carrier having electronically readable control signals, which can cooperate with a programmable computer system, so that one of the methods described herein is carried out.

Generally, embodiments of the present invention may be implemented as a computer program product with a program code, the program code being operative to perform one of the methods when the computer program product operates on a computer. Program code can, for example, be stored on a machine-readable carrier.

Other embodiments comprise computer program for performing one of the methods described herein, stored on a machine readable conveyor.

In other words, an embodiment of the inventive method is thus a computer program having a program code for performing one of the methods described herein when the computer program operates on a computer.

Another embodiment of the inventive methods is thus a data carrier (or a digital storage medium, or a computer readable medium) comprising, registered therein, the computer program for carrying out one of the methods described herein.

Another embodiment of the inventive method is thus a data stream or a sequence of signals representing the computer program to carry out one of the methods described herein. The data stream or signal sequence can, for example, be configured to be transferred over a data communication connection, for example via the Internet.

Another embodiment comprises a processing means, for example a computer, or a programmable logic device, configured or adapted to carry out one of the methods described herein.

Another embodiment comprises a computer having the computer program installed on it for performing one of the methods described herein.

In some embodiments, a programmable logic device (e.g., an array of field programmable gates) may be used to perform some or all of the functionality of the methods described herein. In some embodiments, an array of field programmable gates may cooperate with a microprocessor to perform one of the methods described herein. Generally, the methods are preferably performed by hardware.

The embodiments described above are merely illustrative for the principles of the present invention. It is understood that modifications and variations of the provisions and details described herein will be apparent to those skilled in the art. It is intended, therefore, to be limited only by the scope of the forthcoming patent claims and not by the specific details presented in the form of description and explanation of the embodiments herein. LITERATURE [1] M. Dietz, L. Liljeryd, K. Kjörling and O. Kunz, "Spectral Band Replication, a novel approach in audio coding," at the 112th AES Convention, Munich, May 2002. [2] S. Meltzer , R. Bohm and F. Henn, "SBR enhanced audio codecs for digital broadcasting such as "Digital Radio Mondiale" (DRM)," at the 112th AES Convention, Munich, May 2002. [3] T. Ziegler, A. Ehret , P. Ekstrand and M. Lutzky, "Enhancing mp3 with SBR: Features and Capabilities of the new mp3PRO Algorithm," at the 112th AES Convention, Munich, May 2002. [4] International Standard ISO/IEC 14496-3:2001/ FPDAM 1, "Bandwidth Extension," ISO/IEC, 2002. Speech bandwidth extension method and apparatus Vasu Iyengar et al. [5] E. Larsen, R.M. Aarts, and M. Danessis. Efficient high-frequency bandwidth extension of music and speech. At the 112th AES Convention, Munich, Germany, May 2002. [6] R. M. Aarts, E. Larsen, and O. Ouweltjes. A unified approach to low- and high-frequency bandwidth extension. At AES 115th AES Convention, New York, USA, October 2003. [7] K. Kãyhkõ. A Robust Wideband Enhancement for Narrowband Speech Signal. Research Report, Helsinki University of Technology, Acoustics and Audio Signal Processing Laboratory, 2001. [8] E. Larsen and R. M. Aarts. Audio Bandwidth Extension - Application to psychoacoustics, Signal Processing and Loudspeaker Design. John Wiley & Sons, Ltd, 2004. [9] E. Larsen, R.M. Aarts, and M. Danessis. Efficient high-frequency bandwidth extension of music and speech. At the 112th AES Convention, Munich, Germany, May 2002. [10] J. Makhoul. Spectral Analysis of Speech by Linear Prediction. IEEE Transactions on Audio and Electroacoustics, AU-21(3), June 1973. [11] US Patent Application 08/951,029, Ohmori, et al. Audio band width extending system and method. [12] US Patent Application 6895375, Malah, D & Cox, R.V. : System for bandwidth extension of Narrowband speech. [13] Frederik Nagel, Sascha Disch, "A harmonic bandwidth extension method for audio codecs," ICASSP International Conference on Acoustics, Speech and Signal Processing, IEEE CNF, Taipei, Taiwan, April 2009. [14] Frederik Nagel, Sascha Disch , Nikolaus Rettelbach, "A phase vocoder driven bandwidth extension method with novel transient handling for audio codecs," At the 126th AES Convention, Munich, Germany, May 2009. [15] M. Puckette. Phase-locked Vocoder. "IEEE ASSP Conference on Applications of Signal Processing to Audio and Acoustics, Mohonk 1995.", Rebel, A.: Transient detection and 5 preservation in the phase vocoder; citeseer.ist.psu.edu/679246.html. [16] Laroche L., Dolson M. : "Improved phase vocoder timescale modification of audio", IEEE Trans. Speech and Audio Processing, vol. 7, no. 3, pp. 323--332,. [17] US Patent Application 6549884 Laroche, J. & Dolson, M.: Phase-vocoder pitch-shifting. [18]Herre, J.; Faller, C.; Ertel, C.; Hilpert, J.; Holzer, A.; Spenger, C, "MP3 Surround: Efficient and Compatible Coding of Multi-Channel Audio," 116th Aud Convention. 15 Eng. Soc., May 2004. [19] Neuendorf, Max; Gournay, Philippe; Multrus, Markus; Lecomte, Jérémie; Bessette, Bruno; Geiger, Ralph; Bayer, Stefan; Fuchs, Guillaume; Hilpert, Johannes; Rettelbach, Nikolaus; Salami, Redwan; Schuller, Gerald; Lefebvre, Roch; Grill, Bernhard: Unified Speech and Audio Coding Scheme for High Quality at Lowbitrates, ICASSP 2009, April 19-24, 2009, Taipei, Taiwan. [20] Bayer, Stefan; Bessette, Bruno; Fuchs, Guillaume; Geiger, Ralph; Gournay, Philippe; Grill, Bernhard; Hilpert, Johannes; Lecomte, Jérémie; Lefebvre, Roch; Multrus, 25 Markus; Nagel, Frederick; Neuendorf, Max; Rettelbach, Nikolaus; Robliard, Julien; Salami, Redwan; Schuller, Gerald: A Novel Scheme for Low Bitrate Unified Speech and Audio Coding, 126th AES Convention, May 7, 2009, München.

Claims (17)

1. APPARATUS FOR PROCESSING AN INPUT AUDIO SIGNAL (2300), comprising: a synthesis filter bank (2304) for synthesizing an intermediate audio signal (2306) from the input audio signal (2300), the audio signal input (2300) being represented by a plurality of first subrange signals (2303) generated by an analysis filter bank (2302), wherein a number of filter bank (MS) channels of the synthesis filter bank ( 2304) is less than a number of channels (M) of the analysis filter bank (2302); and another analysis filter bank (2307) for generating a plurality of second subtrack signals (2308) of the intermediate audio signal (2306), wherein the other analysis filter bank (2307) has a number of channels (MA ) being different from the number of channels of the synthesis filter bank (2304), such that a sampling rate of a sub-band signal of the plurality of second sub-band signals (2308) is different from a sampling rate of a first signal subrange of the plurality of first subrange signals (2303); characterized in that the synthesis filter bank (2304) is configured to only process a subgroup (2305) of all first subrange signals (2303) of the plurality of first subrange signals representing the full bandwidth input audio signal (2300), and in which the synthesis filter bank (2304) is configured to generate the intermediate audio signal (2306) as a band segment of the full bandwidth input audio signal (2300) modulated to the band base. 2. APPLIANCE according to claim 1, characterized in that the synthesis filter bank (2304) is a filter bank with real value. 3. APPARATUS according to claim 1, characterized in that the number of first sub-range signals of the plurality of first sub-range signals (2303) is greater than or equal to 24, and in which the number of filter bank channels of the bank synthesis filter (2304) is less than or equal to 22. 4. APPARATUS according to any one of claims 1 to 3, in which the analysis filter bank (2302) is a complex value filter bank, in which the synthesis filter bank (2304) is characterized by comprising a real-valued calculator for calculating the real-valued sub-range signals from the first sub-range signals, wherein the real-valued sub-range signals calculated by the real-valued calculator are further processed by the synthesis filter bank (2304) to obtain the intermediate audio signal (2306). 5. APPARATUS according to any one of claims 1 to 4, characterized in that the other analysis filter bank (2307) is a complex value filter bank and is configured to generate the plurality of second sub-band signals (2308). ) as complex subrange signals. 6. APPLIANCE according to any one of claims 1 to 5, characterized in that the synthesis filter bank (2304), the other analysis filter bank (2307) or the analysis filter bank (2302) are configured to use subtested versions of the same filter bank window. 7. APPARATUS according to any one of claims 1 to 6, further comprising: a sub-track signal processor (2309) for processing the plurality of second sub-tracks (2308); and another synthesis filter bank (2311) for filtering a plurality of processed subtracks, wherein the other synthesis filter bank (2311), the synthesis filter bank (2304), the analysis filter bank (2302) or the other analysis filter bank (2307) are configured to use subtested versions of the same filter bank window, or where the other synthesis filter bank (2311) is configured to apply a synthesis window, and where the other analysis filter bank (2307), the synthesis filter bank (2304), or the analysis filter bank (2302) are configured to apply a subtested version of the synthesis window used by the other synthesis filter bank ( 2311). 8. APPARATUS as claimed in any one of claims 1 to 7, further comprising a subtrack processor (2309) for performing a subtrack non-linear processing operation to obtain a plurality of processed subtracks; a high frequency reconstruction processor (1030) for adjusting an input signal based on transmitted parameters (1040); and another synthesis filter bank (2311, 1050) for combining the input audio signal (2300) and the plurality of processed subtrack signals, wherein the high frequency reconstruction processor (1030) is configured to process an output. from the other synthesis filter bank (1050, 2311) or to process the plurality of processed sub-tracks, before the plurality of processed sub-tracks are fed to the other synthesis filter bank (2311, 1050). 9. APPARATUS according to any one of claims 1 to 8, characterized in that the synthesis filter bank (2304) is configured to set to zero an input on a channel of the lowest filter bank and on a channel of the bank. filter bank of the synthesis filter bank (2304). APPARATUS according to any one of claims 1 to 9, being configured to perform a block-based harmonic transposition, characterized in that the synthesis filter bank (2304) is an undertested filter bank. 11. APPARATUS according to any one of claims 1 to 10, further characterized in that it comprises a sub-track processor (2309), wherein the sub-track processor (2309, 1020a, 1020b) comprises: a plurality of different branches of processing (110a, 110b, 110c) for different transposition factors to obtain a transposition signal, wherein each processing branch is configured to extract (120a, 120b, 120c) blocks of subtrack samples; an adder (128) for adding the transposition signals to obtain the transposition blocks; and an overlay adder (130) for adding consecutive time transposition blocks using a block advance value being greater than a block advance value used to extract (120a, 120b, 120c) the blocks in the plurality of different branches processing (110a, 110b, 110c). 12. APPARATUS, according to any one of claims 1 to 11, further characterized in that it comprises: the analysis filter bank (2302), wherein the synthesis filter bank (2304) and the other filter bank of analyzer (2307) are configured to perform a sample rate conversion, a time extension processor (100a, 100b, 100c) to process the sample rate converted signal; and a combiner (2311, 605) for combining the processed subband signals generated by the time span processor to obtain a processed time domain signal. 13. APPARATUS according to any one of claims 1 to 12, characterized in that the number of channels of the other analysis filter bank (2307) is greater than the number of channels of the synthesis filter bank (2304). 14. APPARATUS FOR PROCESSING AN INPUT AUDIO SIGNAL (2300), comprising: an analysis filter bank (2302) having a number (M) of channels of the analysis filter bank, wherein the analysis filter bank ( 2302) is configured to filter the incoming audio signal (2300) to obtain a plurality of first sub-band signals (2303); and a synthesis filter bank (2304) for synthesizing an intermediate audio signal (2306) using a group (2305) of first subrange signals (2303), wherein the group comprises a number of subrange signals less than the number of filter bank channels of the analysis filter bank (2302), wherein the intermediate audio signal (2306) is the subtested representation of a portion of the bandwidth of the input audio signal (2300); characterized in that the synthesis filter bank (2304) is configured to only process a subgroup (2305) of all first subrange signals (2303) of the plurality of first subrange signals representing the full bandwidth input audio signal (2300), and in which the synthesis filter bank (2304) is configured to generate the intermediate audio signal (2306) as a band segment of the full bandwidth input audio signal (2300) modulated to the band base. 15. APPARATUS according to claim 14, characterized in that the analysis filter bank (2302) is critically the tested complex QMF filter bank, and in which the synthesis filter bank (2304) is a QMF filter bank of critically tested real value. 16. A METHOD FOR PROCESSING AN INPUT AUDIO SIGNAL (2300), comprising: synthesis filtering using a synthesis filter bank (2304) to synthesize an intermediate audio signal (2306) from the input audio signal (2300), the input audio signal (2300) being represented by a plurality of first sub-track signals (2303) generated by an analysis filter bank (2302), wherein a number of filter bank (MS) channels of the bank of synthesis filter (2304) is less than a number of channels (M) of the analysis filter bank (2302); and analysis filtering using another analysis filter bank (2307) to generate a plurality of second subtrack signals (2308) of the intermediate audio signal (2306), wherein the other analysis filter bank (2307) has a number of channels (MA) being different from the number of channels of the synthesis filter bank (2304), so that a sampling rate of a sub-band signal of the plurality of second sub-band signals (2308) is different from a sampling rate a first subrange signal of the plurality of first subrange signals (2303); characterized in that the synthesis filter bank (2304) is configured to only process a subgroup (2305) of all first subrange signals (2303) of the plurality of first subrange signals representing the full bandwidth input audio signal (2300), and in which the synthesis filter bank (2304) is configured to generate the intermediate audio signal (2306) as a band segment of the full bandwidth input audio signal (2300) modulated to the band base. 17. A METHOD FOR PROCESSING AN INPUT AUDIO SIGNAL (2300), comprising: analysis filtering using an analysis filter bank (2302) having a number (M) of channels of the analysis filter bank, wherein the analysis filter bank analysis filter (2302) is configured to filter the input audio signal (2300) to obtain a plurality of first sub-range signals (2303); and synthesis filtering using a synthesis filter bank (2304) to synthesize an intermediate audio signal (2306) using a group (2305) of first subrange signals (2303), where the group comprises a number of minor subrange signals than the number of filter bank channels of the analysis filter bank (2302), where the intermediate audio signal (2306) is the subtested representation of a portion of the input audio signal bandwidth (2300) ; characterized in that the synthesis filter bank (2304) is configured to only process a subgroup (2305) of all first subrange signals (2303) of the plurality of first subrange signals representing the full bandwidth input audio signal (2300), and in which the synthesis filter bank (2304) is configured to generate the intermediate audio signal (2306) as a band segment of the full bandwidth input audio signal (2300) modulated to the band base.

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