RU2541504C1 - Apparatus for selecting moving targets for pulse-to-pulse frequency tuning mode - Google Patents

Abstract

FIELD: radio engineering, communication.

SUBSTANCE: selection apparatus comprises a quartz-crystal oscillator, a pulse modulator, a stable driving generator, a first mixer, a high frequency generator, an antenna switch, an antenna, a second mixer, a filter, a first broadband intermediate frequency amplifier, a display, as well as a digital computer, a frequency synthesiser, an antenna angular position sensor, a high frequency amplifier, third and fourth mixers, a second broadband intermediate frequency amplifier, N frequency channels, each comprising an n-th frequency filter, quadrature phase detectors and an analogue-to-digital converter. The listed components are interconnected in a certain manner.

EFFECT: high efficiency of selecting moving targets in pulse-to-pulse carrier probing frequency tuning mode.

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Description

The invention relates to the category of radar devices and is intended for the selection of moving airborne objects (aircraft) moving against the background of passive interference (PP) during pulse-frequency tuning of the carrier frequency based on the method described in [1-4].

At present, the regimes of carrier frequency tuning from pulse to pulse are of increasing interest for domestic and foreign developers of promising radar stations. This is due to the fact that pulse-frequency tuning is one of the effective methods for protecting the radar from active interference. In addition, the use of frequency tunable signals (HF) also allows you to expand the information capabilities of radar stations in the following areas: increase the resolution in range; identification of air targets (CC) and selection of false objects; measurement of the radial extent of the center; determination of the composition of the group goal [5-22].

A device for moving targets selection (SAC), implemented in coherent-pulse radar, which includes a modulator, transmitter, antenna switch, antenna, auxiliary mixer, stable local oscillator, mixer, coherent local oscillator, phase detector, intermediate frequency amplifier (IF), additional frequency generator, filter, video amplifier and output device [23, p.348]. The output of the modulator is connected to the input of the transmitter, the first output of which is connected to the first input of the auxiliary mixer, and the second output is connected to the input of the antenna switch, the input-output of which is connected to the input-output of the antenna, and the output is connected to the first input of the mixer, the second input of which connected to the second output of the stable local oscillator, and the output is connected to the input of the amplifier, the output of which is connected to the second input of the phase detector, the first input of which is connected to the output of the coherent local oscillator, and the output is connected to the course of the filter, the output of which is connected to the input of the video amplifier, the output of which is connected to the input of the output device. In addition, the second output of the transmitter is connected to the first input of the auxiliary mixer, the second input of which is connected to the first output of the stable local oscillator, and the output is connected to the first input of the coherent local oscillator, the second input of which is connected to the output of the additional frequency generator.

The selection of moving targets in the described device [23, p.348] is based on the determination of the Doppler frequency, which depends on the radial velocity of the target and the magnitude of the carrier frequency of the probing signal. However, when using carrier-by-frequency tuning of the carrier frequency, the Doppler frequency from pulse to pulse will change according to an unknown law determined by the configuration of the target and the law of frequency tuning, which will not allow detecting air targets moving against the background of passive interference. Selection of moving targets in the frequency tuning mode should be based on new principles and approaches [1-4, 24], which necessitates the development of new SDC devices, i.e. devices with an updated structure.

A device is known SDS, implemented in a pulse-Doppler radar station [23, p. 367], including a synchronizer, a pulse modulator, a crystal oscillator, a stable master oscillator, a first mixer, a high-frequency generator, an antenna switch, an antenna, a filter, a second mixer, a first broadband UPCH, M range selection channels, local oscillator and indicator. Moreover, each range selection channel contains a range selector, a filter of one sideband, a main beam rejection filter, a Doppler frequency extraction mixer, N frequency channels, each of which contains a Doppler filter, an amplitude detector, a post-detector filter, and a threshold device. The first synchronizer output is connected to the first inputs of the range selectors, the second synchronizer output is connected to the input of the pulse modulator, the output of which is connected to the first input of the high-frequency generator, the second input of which is connected to the second output of the stable master oscillator, the first output of which is connected to the first input the first mixer, the second input of which is connected to the output of the crystal oscillator, the first output of the high-frequency generator is connected to the input of the antenna switch, the input-output of which -th output is connected to the antenna input, and an output connected to the second input of the second mixer having a first input connected to an output filter having an input connected to the output of the first mixer; the output of the second mixer is connected to the input of the first broadband amplifier, the output of which is connected to the second inputs of the range selectors. In each range selection channel, the range selector output is connected to the input of a filter of one sideband, the output of which is connected to the input of the main beam rejection filter, the output of which is connected to the first input of the Doppler frequency isolation mixer, the second input of which is connected to the local oscillator output, and the output is connected to the inputs of Doppler filters. In each frequency channel, the output of the Doppler filter is connected to the input of the amplitude detector, the output of which is connected to the input of the post-detector filter, the output of which is connected to the input of the threshold device. The outputs of the threshold devices of each frequency channel are connected to the indicator input.

The disadvantage of the described device is the inability to select moving targets during pulse-by-wave tuning of the carrier frequency, associated with the difficulty of determining the Doppler frequency, which from pulse to pulse has an unknown value, determined by the geometric design or structure of the air target and the law of tuning of the carrier frequency [2-4]. The specified disadvantage is advisable to eliminate. For this, it is necessary to make changes to the design of the SDS device described in [23, p. 367], which will allow to implement the method [1].

The objective of the invention is to develop a new improved design of the device SDS, capable of conducting effective selection of moving targets in the frequency tuning mode from pulse to pulse.

, Δf - шаг перестройки частоты от импульса к импульсу, квадратурный фазовый детектор (ФД) и аналого-цифровой преобразователь (АЦП). При этом первый выход ЦВМ соединяют со входом индикатора, второй выход ЦВМ соединяют со входом импульсного модулятора, третий выход ЦВМ соединяют со входом синтезатора частот, каждый n-й выход которого соединяют с первым входом квадратурного ФД соответствующего n-го частотного канала, а (N+1)-й выход синтезатора частот соединяют со вторым входом первого смесителя, выход которого соединяют со вторым входом генератора высокой частоты, выход антенного переключателя соединяют со входом УВЧ, выход которого соединяют со вторым входом второго смесителя, вход фильтра связывают с выходом четвертого смесителя, первый вход которого соединяют с выходом стабильного задающего генератора, выход кварцевого генератора соединяют со вторым входом четвертого смесителя и со вторым входом третьего смесителя, первый вход которого соединяют с выходом первого широкополосного УПЧ, а выход соединяют со входом второго широкополосного УПЧ, выход которого соединяют одновременно со входом каждого фильтра n-й частоты соответствующего n-го частотного канала, в каждом n-м частотном канале выход фильтра n-й частоты соединяют со вторым входом квадратурного ФД, первый и второй выходы которого соединяют с соответствующими первым и вторым входами АЦП, выход АЦП n-го частотного канала подключают к соответствующему n-му входу ЦВМ, (N+1)-й вход которой соединяют с выходом датчика углового положения антенны, механический вход которого механически связывают с механическим выходом антенны.The solution to this problem is achieved by the fact that the composition of the known device described above [23, p.367], including a crystal oscillator, a pulse modulator, a stable master oscillator, first and second mixers, a high-frequency generator, an antenna switch, an antenna, a filter, the first broadband IF, an indicator in which the output of the stable master oscillator is connected to the first input of the first mixer, the output of the pulse modulator is connected to the first input of the high-frequency generator, the output of which is connected to the input of the antennas nth switch, the input-output of which is connected to the input-output of the antenna, the output of the filter is connected to the first input of the second mixer, the output of which is connected to the input of the first broadband amplifier, additionally include a digital computer (digital computer), a frequency synthesizer, an antenna angle sensor, an amplifier high frequency (UHF), the third and fourth mixers, the second wideband IF, as well as N frequency channels, each of which contains a filter of the nth frequency f _n , where f _n = f _pr2 + nΔf, f _pr2 is the second intermediate frequency, $$n=\overline{\mathrm{one},N}$$

, Δf - frequency tuning step from pulse to pulse, quadrature phase detector (PD) and analog-to-digital converter (ADC). In this case, the first output of the digital computer is connected to the input of the indicator, the second output of the digital computer is connected to the input of the pulse modulator, the third output of the digital computer is connected to the input of the frequency synthesizer, each n-th output of which is connected to the first input of the quadrature PD of the corresponding n-th frequency channel, and (N The +1) -th output of the frequency synthesizer is connected to the second input of the first mixer, the output of which is connected to the second input of the high-frequency generator, the output of the antenna switch is connected to the input of the UHF, the output of which is connected to the second input of the second mixer sensor, the input of the filter is connected to the output of the fourth mixer, the first input of which is connected to the output of the stable master oscillator, the output of the quartz generator is connected to the second input of the fourth mixer and to the second input of the third mixer, the first input of which is connected to the output of the first broadband amplifier, and the output is connected to the input of the second broadband IF amplifier, the output of which is connected simultaneously with the input of each filter of the nth frequency of the corresponding n-th frequency channel, in each n-th frequency channel the output of the filter n -th frequency is connected to the second input of the quadrature PD, the first and second outputs of which are connected to the corresponding first and second inputs of the ADC, the ADC output of the n-th frequency channel is connected to the corresponding n-th input of the digital computer, the (N + 1) -th input of which is connected with the output of the antenna angular position sensor, the mechanical input of which is mechanically connected to the mechanical output of the antenna.

This construction of the structural diagram of the SDS device for the frequency-from-pulse to pulse-frequency tuning mode gives it the ability to select air targets (objects) moving against the background of passive interference in the carrier-frequency tuning mode.

The structural diagram of the proposed device SDS for the regime of tuning the frequency from pulse to pulse is shown in figure 1.

According to this scheme, an SDC device for a frequency from pulse to pulse adjustment mode comprises a crystal oscillator 1, a pulse modulator 2, a stable master oscillator 3, a first mixer 4, a high frequency generator 5, an antenna switch 6, an antenna 7, a second mixer 8, a filter 9, the first broadband amplifier 10, an indicator 11, a digital computer 12, a frequency synthesizer 13, the angular position sensor of the antenna 14, a high frequency amplifier 15, a third mixer 16, a fourth mixer 17, a second broadband amplifier 18, and also contains N cha -frequency channels, each of which contains n-th filter 19 frequency, quadrature phase detector 20 and the analog-to-digital converter 21 (Figure 1).

The SDS device for the frequency tuning mode from pulse to pulse works as follows.

излучения импульса на n-й частоте. В интересах последующей обработки данные вектора F сохраняются.Stable master oscillator 3 generates highly stable high-frequency electromagnetic waves at the main carrier frequency f ₀ , which are fed to the inputs of the first and fourth mixer. As a stable master oscillator 3, it is advisable to use a reflective klystron. The structure and principle of operation of stable master oscillators based on reflective klystrons are known and described in [25, p.207]. To emit signals with a random law of frequency change in the digital computer 12, a sequence of numbers of radiation frequencies is formed, which is distributed according to a random law and is used in an FH bundle. In this case, a vector F will be formed, the nth element of which determines the serial number $$\tilde{n}$$

pulse radiation at the nth frequency. In the interest of further processing, the data of the vector F is stored.

To ensure the regime of pulse-wise tuning of the carrier frequency in the frequency synthesizer 13, high-frequency oscillations are constantly generated at all n-th frequencies f _n = f _pr2 + nΔf. These high-frequency oscillations from the output of the frequency synthesizer 13 are sequentially fed to the second input of the first mixer 4 in accordance with the law of frequency tuning, which is determined by the signals entering the frequency synthesizer 13 from the output of the digital computer 12 in accordance with the stored vector F. The principle of operation of frequency synthesizers is described in [ 26, p. 29; 27, p. 247]. The constancy of oscillation generation eliminates the need to take into account transients during fast switching of frequencies. In addition, this makes it possible to use the signals generated by the frequency synthesizer as reference signals for conducting quadrature phase detection of reflected signals in quadrature PD 20.

, предлагается выбирать в качестве опорной дальности d-го дальностного канала.To solve this problem, it is proposed to divide the radar detection zone depicted in FIG. 2 (the area of space around the radar, limited by the near and far boundaries of the detection zone, in which, in the absence of deliberate interference, air targets are detected with a probability not lower than the given) by B = 2π / Θ _{β of the} azimuthal sectors, where Θ _β is the width of the directivity of the antenna in the azimuthal plane, and D = 2 (R _D -R _B ) / (c _i ) of the distance channels, where R _B and R _D are the distances to the near and far boundaries detection zones , C - electromagnetic wave propagation velocity, τ _i - pulse duration, range of _{R d = (R B + dcτ} i / 2) where d - number of range-channel $$(d=\overline{0,D-\mathrm{one}})$$

, it is proposed to choose the d-th range channel as the reference range.

To implement the method [1] in the proposed SDS device, it is necessary to form and periodically radiate into space using an antenna 7 rotating in azimuth three pairs of bursts of pulsed signals with tuning of the carrier frequency according to a random law. The radiation time of a triple of bursts of pulses should correspond to the time of azimuth change during antenna rotation by величину _β . The process of rotation of the antenna and its emission of triples of pairs of HF packets into space is constant. This provides a complete overview of the radar detection area. The combination of pairs of packets in triples is necessary for processing the reflected signals according to the method [1]. In the interest of increasing noise immunity, the law of variation of the carrier frequency should be different for each pair of packs. Within a pair of packs of the HRC, this law should be the same in both packs used.

To review the entire detection zone (figure 2), the antenna 7 must change the angular position in azimuth in the range from 0 to 360 degrees. Method [1] involves the absence of amplitude modulation of signals from one pair of HF packets by the directivity of the antenna during its rotation. To implement the method [1], the antenna rotation speed ω _A and the duration of the FH packs T _p must be chosen from conditions that allow us to neglect the specified amplitude modulation. It is known from [28, 29] that the directivity characteristic of the antenna in the azimuthal plane f (Θ) can be described by the arch sine function f (Θ) = sinc [π L sin (Θ) / λ], where Θ is the angle between the direction to the far zone the observation point and the normal of the antenna, L is the linear size of the antenna, λ is the wavelength. The study of the dependence of the modulation of the reflected signals on the rotation speed of the antenna is advisable to carry out within the beam width at half power (the width of the antenna pattern). Obviously, with this approach, the maximum changes in the function f (Θ) will be observed at values of the angle Θ close to ± Θ _β / 2. Figure 3 shows the main lobe of the antenna pattern and its fragment at values of Θ close to Θ _β / 2. Figure 3 shows that when the variable Θ changes by Θ _β / 40 (turning the antenna through the angle Θ _β / 40), the value of the function f (Θ) changes by no more than 0.026, when turning through the angle Θ _β / 20 it does not more than 0.05, when turning the angle Θ _β / 10 - no more than 0.1. To neglect changes in a function, its modulation should not exceed a few percent (up to 5). On this basis, a requirement has been developed for the emission of one pair of HF packets during the rotation of the antenna through an angle Θ _β / 20:

A variant of the sequence of sounding signals at ω _A = Θ _β / (40T _p ) is presented in Fig.4.

A sensor for the angular position of the antenna 14 is mechanically connected to the antenna 7, which is designed to generate signals that are further converted into a special code that carries information about the angular position of the antenna 7 at the moment of emission of the next (for example, second) pair of HF packets from the triple. The specified code goes to the digital computer.

To generate probe pulses with tuning of the carrier frequency, the signal of the stable master oscillator 3 is mixed in the first mixer 4 with one of the signals generated by the frequency synthesizer 13. The pulse duration τ _i , their pulse repetition period T _i and the frequency change order are determined by a digital computer 12. Principle of operation digital computers is known and described in [30, p.638; 31, p.77]. The signals from the digital computer output control the operation of pulse modulator 2. The pulse modulator generates video pulses of duration τ _i that control the operation of the high-frequency generator 5. The structure and principle of operation of pulse modulators are known and described in [30, p. 312].

The passband of the output filter of the first mixer 4 (not shown in FIG. 1) is selected in the range from f ₀ + f _{CR 2} to f ₀ + f _{CR 2} + NΔf. In this case, at the output of the first mixer 4, only a narrow-band signal will be present at one of the sensing frequencies, and multiple harmonics will be suppressed. It should be noted that the first, second, third and fourth mixers are known nonlinear elements designed for frequency conversion, the basic principles of which are described in [32, p.222].

The signal from the output of the first mixer 4 at a frequency f ₀ + f _pr2 + nΔf is fed to the second input of the high-frequency generator 5, which is advisable to use a flying klystron. The structure and principle of operation of flying klystrons are known and described in [25, p.199]. The high-frequency generator 5, using the signals from the output of the pulse modulator 2, generates microwave pulses with frequency tuning of a given duration τ _i , which are transmitted through the antenna switch 6 to the antenna 7, which emits them into space. Antenna switch 6 is a known device, the principles of its operation are described in [30, p.166]. It is advisable to use a reflector antenna for radiating the HF into space, the structure and principle of operation of which are described in [30, p. It should be noted that the described process of the formation and emission of HFs into space refers to only one of the azimuthal positions of the antenna during the review. The process of generating and emitting sounding signals as the antenna 7 rotates in azimuth is constant, which provides a complete overview of the radar detection zone.

Reflecting from the target, the radiated radio signals with a changed structure return to the antenna 7, are caught by it and pass into the receiving path, where the probing radio signals also “leak out”. The fact that probing radio signals entered the receiver was experimentally confirmed in [33–36]. From the output of the antenna 7, the received signals through the antenna switch 6 are fed to the input of the UHF 15, which carries out their primary amplification. The structure and principles of operation of high-frequency amplifiers are described in [27, p.353].

In the proposed device, a double frequency conversion is carried out, which is necessary to eliminate the influence of interference on the mirror channel. Frequency conversion is carried out using a quartz oscillator 1, a fourth mixer 17, a filter 9, a second mixer 8, a third mixer 16. The quartz generator 1 generates continuous stable electromagnetic waves at the first intermediate frequency f _pr1 , which are supplied to the third 16 and fourth 17 mixers. In the fourth mixer 17, these oscillations are mixed with the oscillations of the stable master oscillator 3, as a result of which a combination of high-frequency oscillations is generated at its frequencies that are multiples of the sum and difference of the main carrier frequency f ₀ and the first intermediate frequency f _pr1 , which are fed to filter 9. Band filter 9 is selected so that only a narrow-band radio signal at a difference frequency equal to f ₀ -f _{pr1 passes through its output} .

After the initial amplification, the reflected radio signals from the output of the UHF 15 are fed to the second input of the second mixer 8, the first input of which receives the signal from the output of the filter 9. The passband of the output filter of the second mixer 8 (not shown in Fig. 1) is selected in the range from f _pr1 + f _CR2 to f _CR1 + f _CR2 + NΔf.

The main amplification of the received signals is carried out in two broadband amplifiers of intermediate frequency 10 and 18. The principle of operation and structure of the amplifier are known and described in [27, p.376]. The signals from the output of the second mixer 8 enter the first broadband amplifier 10. The bandwidth of the first broadband amplifier 10 is selected in the range from f _pr1 + f _pr2 to f _pr1 + f _pr2 + NΔf, and the passband of the second broadband amplifier 18 is in the range from f _pr2 up to f _pr2 + NΔf. The amplified reflected signals at a frequency f _pr1 + f _pr2 + nΔf from the output of the first broadband amplifier 10 are fed to the first input of the third mixer 16 (at the second input of which there are signals at the first intermediate frequency f _pr1 ), which transfers the signal spectrum to the second intermediate frequency f _pr2 . For this, the passband of the output filter of the third mixer 16 (not shown in FIG. 1) is selected in the range from f _CR2 to f _CR2 + NΔf. The reflected signals at a frequency f _pr2 + nΔf from the output of the third mixer 16 are fed to the input of the second broadband amplifier 18, in which they are amplified.

The output of the second broadband amplifier 18 is connected simultaneously with the input of each filter of the nth frequency 19 of the corresponding n-th frequency channel, with which the signals are distributed over the frequency channels. The number of channels is equal to the number of frequencies N. The passband of the filter of the nth frequency 19 is selected in the range f _pr2 + nΔf ± Δf / 2. In the nth frequency channel, filter 19 is tuned to its nth frequency. That is, the filter 19 passes signals into its nth channel only to the frequency channel number. Moreover, for the correct filtering of the signals, it is necessary to fulfill the condition τ _i > 1 / (2Δf). It is assumed that the filters 19 and the filter 9 are band-pass filters, the structure and principle of operation of which are described in [37, p.202].

In the interest of further processing, the received signals from the outputs of the filter of the nth frequency 19 are fed to the input of the quadrature phase detector 20 of the corresponding nth channel, where they are divided into quadrature components (sine Im and cosine Re). The principles of quadrature processing of received signals using a quadrature phase detector 20 are known and described in [38–40]. To conduct quadrature processing, the quadrature phase detector 20 of each n-th frequency channel is supplied with reference oscillations at a frequency f _pr2 + nΔf, which are constantly generated by the frequency synthesizer 13 to generate sounding signals (ZS).

In each frequency channel, the quadrature components of the received signals Re and Im obtained using quadrature PD 20 are fed to the ADC 21, where they are converted to digital form. The analog-to-digital Converter 21 is a known device, the basic principles of which are described in [41, p.291; 42, p. 11]. The sampling frequency of the ADC 21 is selected according to the Kotelnikov theorem, according to which, for an unambiguous perception of the function by discrete values, the sampling frequency F _d should be chosen at least twice the frequency of the sampled function (analog signal at the output of the amplifier). For example, the maximum frequency of the input signal of the ADC 21 with f _pr2 = 30 MHz and ΔF = 150 MHz is 180 MHz. Therefore, at the indicated signal parameters, the sampling frequency F _d should be selected at least 360 MHz (F _d ≥360 MHz).

At the ADC output of 21 n-frequency channels, samples of the corresponding quadrature components (sine and cosine) of the reflected signal at a frequency f _pr2 + nΔf are generated, which in a complex form (as a complex number) are fed to the corresponding n-th inputs of digital computer 12, where their multichannel digital processing according to the method [1]. It includes: coordinated filtering of received signals, the formation of frequency characteristics (HF) in each range channel, the compensation of the radial movement of passive interference (rephasing of the frequency response), the formation of difference frequency characteristics, the compensation of the radial motion of the CC (rephasing of the differential frequency response), the construction of long-range portraits of the CC , obtaining three estimates of the radial speed of the CC and comparing them with each other, on the basis of which a decision is made on the presence of the CC in the processed range channel at the selected az mute. It should be noted that the described process of receiving and processing reflected signals refers to only one of the azimuthal positions of the antenna during the review. This process, as the antenna 7 rotates in azimuth, is continuous, which provides a complete overview of the detection zone of the radar station.

To carry out a coordinated filtering of the reflected signals of the nth frequency in the random access memory of the digital computer 12, “storing” the probe signal of the nth frequency that has leaked into the receiving path is performed. The fact that probing signals entered the receive path was experimentally confirmed in [33–36]. Information on the probe pulse of the nth frequency begins to be recorded at the moment the signal is transmitted to its radiation, which arrives from the output of the digital computer 12 to the input of the frequency synthesizer 13. Leaked into the receive path due to the imperfect antenna switch 6, the probe signal is digitized and converted into a complex form and is remembered in a complex conjugate form. This is necessary for digitally matched filtering according to the method described in [34-36]. The response of the digital matched filter is calculated as the convolution (the sum of the products with a shift) of the discrete values of the received signals and the complex conjugate ES at the reference times t _{d of} each range channel over an interval equal to the duration of the packet T _p .

Then, in the random access memory of digital computer 12, range selection is performed by dividing the responses of the matched filters into long-distance channels of length ct _i / 2, where c is the propagation velocity of electromagnetic waves. In each d-th range channel of the azimuth channel being processed, a vector G1 _d (ЧХ) is formed of N elements, and the complex value of the response of the matched receiver at the nth frequency at the reference range point of the d-th range channel is written to the nth element of this vector t _d (or nth term of the frequency factor). As a result, D frequency characteristics will be generated in the random access memory 12. This operation is illustrated in figure 5. It is similar to the operation of creating a two-dimensional matrix of D * N elements.

, где v_rПП=-V_{rПП max}, -V_{rПП max}+ΔV_rПП, -V_{rПП max}+2ΔV_rПП, -V_{rПП max}+3ΔV_rПП, …, V_{rПП max}; где V_{rПП max} - максимально возможная скорость пассивной помехи.The frequency characteristics obtained in each d-th range channel as a result of processing the first packet of the FH (vectors G1 _d ) are stored in the random access memory of the digital computer 12 for their subsequent subtraction from the rephased frequency characteristics obtained as a result of processing the second packet of the FH. To rephase, each element of the vectors G2 _d is multiplied by a complex phase factor $${\dot{\chi }}_n,v_{rPP}=\exp (-j\mathrm{four}{\pi }_n{v}_{rPP}{T}_p/c)$$

where v _rPP = -V _{rPP max} , -V _{rPP max} + ΔV _rPP , -V _{rPP max} + 2ΔV _rPP , -V _{rPP max} + 3ΔV _rPP , ..., V _{rPP max} ; where V _{rPP max} - the maximum possible speed of passive interference.

As a result, a two-dimensional matrix G2 _{d of} data from V1 = (2V _{rPP max} / ΔV _rPP +1) rows and N columns will be generated in each range channel of the processed azimuth channel. Next, the n-th elements of the vector G1 _d are subtracted from the rows of the matrix G2 _d , as a result of which a two-dimensional data matrix S _d consisting of V1 rows and N columns will be generated in each range channel. The above operations are schematically explained in Fig.6.

. Значение $$\tilde{n}$$

берется из сохраненного ранее вектора F. В связи с тем что значение радиальной скорости неизвестно, производится перебор возможных значений скорости v_r в диапазоне ±V_{r max} с шагом ΔV. В результате этого в каждом дальностном канале будет сформирована трехмерная матрица W_d данных, состоящая из N*V2 строк и V1*V2 столбцов, где V2=2V_{r max}/ΔV+1 [1].The next step in digital processing of received signals in digital computer 12 is the rephasing of the differential frequency response in each range channel of the processed azimuth channel by multiplying each n-th element of the differential frequency response by the phase factor $${\dot{\xi }}_{n,v_r}=\exp [-j\mathrm{four}\pi f_n{v}_r({\mathrm{<figure-callout\; id="0"\; label="t"\; filenames="00000020.png,00000021.png"\; state="\{\{state\}\}">t</figure-callout>}}_0+\tilde{n}{T}_i)/c]$$

. Value $$\tilde{n}$$

is taken from the previously saved vector F. Due to the fact that the radial velocity value is unknown, the possible values of the velocity v _{r are sorted out} in the range ± V _{r max} with a step ΔV. As a result of this, a three-dimensional data matrix W _d will be formed in each range channel, consisting of N * V2 rows and V1 * V2 columns, where V2 = 2V _{r max} / ΔV + 1 [1].

Then, in each range channel of the azimuth channel being processed, by performing the inverse fast Fourier transform with complex data vectors of each row of the three-dimensional matrix W _d , a three-dimensional matrix W1 _{d is formed} , which is equal in size to the matrix W _d . After that, the maximum value of the modulus of the complex signal is found in the matrix W1 _d , and the complex values of all elements of this matrix are divided by the found value (normalization of the elements of the matrix W1 _d ). After that, the entropy value of the data H is calculated for each row of the matrix W1 _d . As a result, a two-dimensional matrix H _d of V1 * V2 elements is formed in each range channel. These operations are illustrated in Fig.7. The first indices of the matrix elements shown in Fig. 7 indicate the frequency number, the second - the selected value of the radial velocity of the PP, the third - the selected value of the radial velocity of the target.

и $${\widehat{v}}_{rППd\min }$$

, соответствующих наименьшему значению энтропии $$H_d({\widehat{v}}_{rd\min },{\widehat{v}}_{rППd\min })$$

, с помощью которых определяются оценки радиальной скорости воздушной цели $${\widehat{V}}_{rd}$$

и пассивной помехи $${\widehat{V}}_{rППd}$$

в каждом дальностном канале:At the final stage, in each range channel of the azimuth channel being processed, there is a column number $${\widehat{v}}_{rd\min }$$

and $${\widehat{v}}_{rPPd\min }$$

corresponding to the lowest value of entropy $$H_d({\widehat{v}}_{rd\min },{\widehat{v}}_{rPPd\min })$$

by which the estimates of the radial speed of an air target are determined $${\widehat{V}}_{rd}$$

and passive interference $${\widehat{V}}_{rPPd}$$

in each range channel:

, где p - порядковый номер обрабатываемой пары пачек из состава тройки, сохраняются в специальной двумерной матрице размером 3*D, обновляемой после обработки каждой тройки пар пачек СПЧ.Radial velocity estimates obtained in each range channel of the azimuth channel being processed $${\widehat{V}}_{rd}^{(p)}$$

, where p is the serial number of the processed pair of packs from the triple, are stored in a special two-dimensional matrix of size 3 * D, updated after processing each triple of pairs of packs of HFs.

и $${\widehat{V}}_{rd}^{(p+2)}$$

в каждом дальностном канале. При выполнении условийOn this, the processing of the first pair of packs of the HRC is terminated. To eliminate false alarm and make the right decision about the presence of a target in the processed range channel, radiation and processing of at least two more pairs of HF packets with other laws of frequency change are necessary. For this, the above process of radiation and signal processing is repeated twice. Processing each pair of HF bundles according to the algorithm described above allows one to obtain two more estimates of the radial velocity of the target $${\widehat{V}}_{rd}^{(p+\mathrm{one})}$$

and $${\widehat{V}}_{rd}^{(p+2)}$$

in each range channel. Under the conditions

.where ΔV is the threshold determined by the accuracy of the estimation (measurement) of speed, a decision is made on the presence in the d-th range channel of a target moving with radial speed $${\widehat{V}}_{rd}=({\widehat{V}}_{rd}^{(p)}+{\widehat{V}}_{rd}^{(p+\mathrm{one})}+{\widehat{V}}_{rd}^{(p+2)})/3$$

.

The decision on whether the target belongs to the b-th azimuth channel is generated by the digital computer 12 based on the signal generated by the angular position sensor of the antenna 14 and containing information about the normal position of the antenna 7 at the time of the emission of the second pair of processed HF packets from the triple.

The processing results from the first output of the digital computer 12 are fed to the input of the indicator 11. As an indicator 11, it is advisable to use the circular viewing indicator, which is a known device for visual display of information, the basic principles of the device and its operation are described in [30, p.397; 42, p. 264]. The results of signal processing are displayed in the form of information about the azimuth, range and speed of the target on the indicator 11.

The proposed new construction of a moving target selection device circuit has an advantage over the prototype device, which is expressed in giving the noise-free sounding mode with tuning of the carrier frequency the ability to select moving CCs against the background of passive interference and measure their radial velocities. The new device structure allows for multi-channel processing of reflected HF signals, which provides effective selection of moving targets in the airspace survey locator when tuning the carrier frequency of the radar station from pulse to pulse.

The proposed device is recommended for use in promising radars of a circular or sector review of the air traffic control service, as well as in landing radars of aerodromes.

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Claims (1)

A device for moving targets for pulse-to-frequency tuning, including a crystal oscillator, pulse modulator, stable master oscillator, first mixer, high-frequency generator, antenna switch, antenna, second mixer, filter, first intermediate-frequency broadband amplifier, indicator, wherein the output of the stable master oscillator is connected to the first input of the first mixer, the output of the pulse modulator is connected to the first input of the high-frequency generator, the output of which oedinen to the input of the antenna switch, whose input-output connected to the input-output of the antenna filter output is connected to a first input of a second mixer whose output is connected to the input of a first broadband intermediate frequency amplifier,
characterized in that the device also includes a digital computer, a frequency synthesizer, an antenna angle sensor, a high-frequency amplifier, a third and fourth mixer, a second intermediate-frequency broadband amplifier, and N frequency channels, where N is the number of frequencies used in the device tunings, each of which contains a filter of the nth frequency f _n , a quadrature phase detector and an analog-to-digital converter, the first output of a digital computer connected to the input ohm of the indicator, the second output of the digital computer is connected to the input of the pulse modulator, the third output of the digital computer is connected to the input of the frequency synthesizer, each n-th output of which is connected to the first input of the quadrature phase detector of the corresponding n-th frequency channel, a (N + 1 ) -th output is connected to the second input of the first mixer, the output of which is connected to the second input of the high-frequency generator, the output of the antenna switch is connected to the input of the high-frequency amplifier, the output of which is connected they are removed from the second input of the second mixer, the input of the filter is connected to the output of the fourth mixer, the first input of which is connected to the output of the stable master oscillator, the output of the quartz generator is connected to the second input of the fourth mixer and to the second input of the third mixer, the first input of which is connected to the output of the first broadband amplifier intermediate frequency, and the output is connected to the input of the second broadband amplifier of the intermediate frequency, the output of which is connected simultaneously with the input of each filter of the nth the frequency of the corresponding nth frequency channel, in each nth frequency channel the output of the filter of the nth frequency is connected to the second input of the quadrature phase detector, the first and second outputs of which are connected to the corresponding first and second inputs of the analog-to-digital converter, the output of the analog-to-digital the converter of each n-th frequency channel is connected to the corresponding n-th input of a digital computer, the (N-1) -th input of which is connected to the output of the antenna angle sensor, the mechanical input of which is mechanically ki associated with the mechanical output of the antenna.

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