JP6510360B2 - Wireless communication system and wireless communication method - Google Patents

Description

  The present invention relates to a wireless communication system and a wireless communication method.

[Foreword]
At present, with the explosive spread of smartphones, microwave resources with high convenience are depleted in frequency resources. As measures, the transition from the 3rd generation mobile phone to the 4th generation mobile phone and the assignment of new frequency bands are being performed. However, since there are many operators who want to provide services, the frequency resources allocated to each operator are limited.

  In the service of mobile phones, studies are underway to improve the frequency utilization efficiency by a multi-antenna system using a plurality of antenna elements. The already widespread wireless standard IEEE (The Institute of Electrical and Electronics Engineers, Inc.) 802.11n uses Multiple Input Multiple Output (MIMO) transmission technology that uses multiple antenna elements for both transmission and reception. Space multiplexing transmission. Thus, in IEEE 802.11n, transmission capacity is increased to improve frequency utilization efficiency. The term "MIMO" is generally used on the assumption that a transmitting station and a receiving station have multiple antenna elements. When the receiving side is a single antenna element, the term MISO (Multiple Input Single Output) is used instead of MIMO. However, in the following, the term MIMO is used in the sense of including all of them.

  In addition, as a recent communication technology, it is divided into a plurality of frequency components (subcarriers) such as orthogonal frequency division multiplexing (OFDM) modulation scheme and SC-FDE (single carrier frequency domain equalization) scheme, and the frequency axis is divided. A system that performs signal processing is common. The following description will be made using the term “subcarrier” on the premise of a general scheme common to OFDM and SC-FDE without particular distinction.

In the MIMO transmission technology, more efficient transmission can be performed by knowing transmission path information between the transmitting station and the receiving station. As the simplest example, the case of multiple antennas on the receiving side has been shown, but in the case of including N antenna elements on the transmitting side and only one antenna element on the receiving side, N antenna elements The directivity control is performed on the transmission side such that the signals to be transmitted are combined in phase by the antenna element on the reception side. This can increase the line gain. Specifically, when channel information from the j-th antenna element of the transmitting station to the antenna element of the receiving station in the k-th subcarrier is h _j ^(k) , The transmission weight w _j ^(k) of ¹⁾ is calculated, and the transmission signal multiplied by this is transmitted from each antenna element (equal gain combining). Note that the channel information in the present specification takes into account the practical circuit configuration and the procedure of signal processing and control, and strictly speaking, an amplifier, a filter, etc. in an RF (Radio Frequency) circuit of a transmitting system and a receiving system. It is assumed that complex phase rotation and amplitude fluctuation information and the like are included.

A vector (h ₁ ^(k) ,..., H _j ^(k) ,..., H _N ^(k) ) having channel information corresponding to each of the first antenna element on the transmitting side to the Nth antenna element as a channel vector h ^It is called ^(k) . Also, a vector (w ₁ ^(k) ,..., W _j ^(k) ,..., W _N ^(k) ) ^T (T ₁ ^{) in} which the transmission weight corresponding to the first antenna element on the transmission side to the Nth antenna element is a component Represents transposition ⁾ is referred to as a transmission weight vector w ^(k) . Strictly speaking, the channel vector in the downlink → h ^(k) (the symbol “→” in front of “h ^(k) ” is a symbol given on h to represent a vector) is a line The vector, transmit weight vector → w ^(k), should be written as a column vector. However, in the following, for the sake of simplicity, the symbol “→” is omitted and the row vector and the column vector are described without distinction. Also, in the following description, the notation relating to the reception signal Rx, the transmission signal Tx and the noise n should be similarly given “→” to clearly indicate that it is a vector, but since there is no misleading notation “→” I omit and explain. The reception signal Rx is given by the following equation (2) for the transmission signal Tx and the noise n.

Substituting equation (1) into equation (2), the value obtained by adding the absolute value of each component h _j ^(k) of channel vector h ^(k) over all antenna components is obtained as a channel gain. If the transmission power from each antenna element is kept the same as in the case of transmission by one antenna, the amplitude of the received signal is N times that in the case of transmission by one antenna element in the case of N antenna elements Expected to be. The received signal power is improved to N ² times since it is proportional to the square of the amplitude. This value is a gain when a plurality of antenna elements are used as an array antenna. Strictly speaking, the gain of the array antenna itself is generally interpreted as N times the received power (as a result of evaluation under a constant total transmission power). Assuming an environment, the description will be made based on the case where the transmission power per element is constant.

  In general, it is known that the increase in transmission capacity with respect to the improvement amount of the signal-noise ratio (SNR) is larger in the low SNR region and smaller in the high SNR region according to Shannon's theorem. Therefore, rather than aiming to improve the transmission capacity by improving the channel gain, the reception side is often provided with a plurality of antenna elements, and the transmission capacity is often aimed to be improved by spatial multiplexing. MIMO transmission technology aims to increase the transmission capacity by spatial multiplexing.

FIG. 1 shows an outline of MIMO transmission. Here, as an explanation focusing on a certain frequency, a suffix “k” representing a subcarrier or a frequency is omitted. In FIG. 1, reference numeral 11 denotes a transmitting station, and reference numeral 12 denotes a receiving station. In this example, both the transmitting station 11 and the receiving station 12 have two antenna elements, and channel information (amplitude, complex phase) between the transmitting antenna # 1 of the transmitting station 11 and the receiving antenna # 1 of the receiving station 12 Information representing the amount of rotation h ₁₁ , channel information between the transmitting antenna # 1 of the transmitting station 11 and the receiving antenna # 2 of the receiving station 12 (information representing the amplitude and amount of rotation of complex phase) h ₂₁ The channel information (information representing the amplitude and the amount of rotation of the complex phase) between the transmitting antenna # 2 of the station 11 and the receiving antenna # 1 of the receiving station 12 is h ₁₂ , the transmitting antenna # 2 of the transmitting station 11 and the receiving station 12 The channel information (amplitude, information representing the amount of rotation of the complex phase) with respect to the receiving antenna # 2 of h is expressed as h _{22 and} signals t ₁ and t ₂ transmitted from two transmitting antennas of the transmitting station 11 , 2 reception of the receiving station 12 Between the signals r ₁ and r ₂ received by the antenna, noise signals n ₁ and n ₂ are used and expressed by the following equation (3).

  Basically, in MIMO transmission, the signal on the transmission side is estimated based on the reception signal on the reception side and the channel matrix. If the noise term in equation (3) is sufficiently small, the transmission signal can be estimated from the reception signal by multiplying both sides by the inverse of the channel matrix. The transmission characteristics can be improved by multiplying the predetermined transmission weight matrix on the transmission side and further multiplying the predetermined reception weight on the reception side, and more efficient transmission becomes possible. For example, when channel information between a plurality of transmitting antenna elements and a receiving antenna element is known, the channel matrix is subjected to singular value decomposition (SVD) and transmission in the eigenmode is performed. Maximize transmission capacity.

  Specifically, the channel matrix H is decomposed into a diagonal matrix D having a unitary matrix U and V and singular values λ as diagonal components as shown in the following equation (4).

  At this time, if a unitary matrix V is used as the transmission weight matrix, the reception signal vector Rx is given by the following equation (5) for the transmission signal vector Tx and the noise vector n.

On the receiving side, the following equation (6) is obtained by multiplying the matrix U ^H of Hermitian conjugates of the unitary matrix U.

In the equation (6), since the non-diagonal components of the diagonal matrix D are zero, the cross terms of the transmission signal are already canceled and the signals are separated. Also, although the noise vector is multiplied by the reception weight matrix U ^H and converted to the noise vector n ′, which is expressed by rotating the coordinate axis and represented, the statistical features of the vector remain equivalent to the original noise vector. FIG. 2 shows a conceptual diagram of eigenmode transmission. 2 (a) shows a basic MIMO channel, FIG. 2 (b) shows a situation in which transmission and reception weight matrices are multiplied, and FIG. 2 (c) shows a virtual transmission path formed by eigenmode transmission. There is. The channel matrix H between the transmitting and receiving antennas of the transmitting station 11 and the receiving station 12 is, as shown in equation (4), a matrix D having singular values λ in each diagonal component by singular value decomposition, and two unitary It is expressed by the product of the matrix U and V ^H. Here, as shown in FIG. 2 (b) and equation (5), when using the transmission weight matrix V on the transmission side and the reception weight matrix U ^H on the reception side, the unitary matrix part of equation (4) is multiplied and united. It becomes a matrix, and as a result, as shown in equation (6), it becomes possible to represent the matrix D with zero non-diagonal components and non-zero diagonal components. This corresponds to a situation in which transmission paths of virtual channels represented by singular values λ ₁ , λ ₂ ... Are parallelly connected, as shown in FIG. 2 (c). At this time, the square value of the absolute value of each singular value λ corresponds to the channel gain of the individual signal sequence. Each singular value λ is a different value for each signal system. The transmission capacity can be maximized by optimizing the transmission mode in accordance with the singular value of the eigenmode. The transmission mode is a specific mode of signal transmission determined by a combination of modulation multi-level number and error correction code rate.

  Here, if each component of the channel matrix of the MIMO channel is independent and uncorrelated, the absolute value of each singular value becomes a relatively large value. For example, when there are many reflected waves and the received power of the sight wave is relatively low, each singular value will have a relatively large value as described above. On the other hand, when the transmitting / receiving antenna is in the line-of-sight environment and there is not much reflected wave, only the absolute value of the first singular value is extremely large and the absolute value of the singular value after the second singular value is extremely It tends to be smaller. For this reason, it is generally said that MIMO transmission is suitable for a multipath environment, and not so suitable when the line of sight is dominant.

  The above is a description of a single-user MIMO transmission technology assuming one base station apparatus and one terminal station apparatus. The same description can be extended to multi-user MIMO in which communication is simultaneously performed on the same frequency axis between one base station apparatus and a plurality of terminal stations. In multi-user MIMO, in general, each terminal station apparatus communicates with a smaller number of antenna elements than the total number of signal systems to be spatially multiplexed. Therefore, in the downlink, directivity control is performed in advance for suppressing inter-user interference on the transmission side. Although the specific equation is slightly different, basically, as with the above-mentioned eigenmode transmission, after grasping the channel matrix, a transmission weight in accordance with it is used.

  Further, in the above description, the explanation was made focusing on the downlink, but also in the uplink, it is possible to perform communication using the channel information after grasping the channel information in advance similarly. For example, in the processing as the array antenna described first, in addition to using the in-phase combining weight given by equation (1) as the receiving weight, it is given by the following equation (7) as the maximum ratio combining weight It is also possible to use one.

The constant C in the equation (7) is a coefficient which is appropriately determined. Among the components of the vector, those with large absolute values of h _j ^(k) are added with a large weight, and C is determined such that a small signal is added with a small weight. In this way, a signal with a large SNR is emphasized, and adjustments are made so that the noise of the signal with a low SNR does not excessively affect.

  In addition, although downlink channel information is required to calculate transmission weights, this can be realized by various forms of channel feedback. In the simplest example, the terminal station apparatus can receive a training signal transmitted by the base station apparatus on the downlink, and the reception result can be accommodated in uplink control information and notified. In general, a channel estimation method that performs such loopback is called explicit feedback. Besides this, for example, calibration coefficients necessary for converting uplink channel information and downlink channel information in the base station apparatus are acquired in advance, and channel estimation is performed on the uplink. After that, it is also possible to estimate downlink channel information by multiplying this calibration coefficient. This method is called implicit feedback. Implicit feedback is generally favored for large scale antennas where the number of antennas is huge. However, in the present invention, the method of channel feedback is not particularly limited, and it is assumed that a general channel estimation method can be used.

Future Mobile Network Directions
As described above, with the explosive spread of smartphones, a further increase in transmission capacity is required. In current wireless communication research, technology is being studied for the 5th generation mobile phone following the 4th generation mobile phone, and in this case, 10 times more transmission capacity than the 4th generation is realized. Is required. Here, not only the increase in transmission capacity of one base station apparatus and the wireless system under the control thereof, but also the increase in transmission capacity per unit area are required together. Specifically, in areas where people gather, such as Shinjuku, Shibuya, Ginza and Otemachi, simply increasing the transmission capacity of the wireless system can not cope with it, and reduce the area area covered by one base station device (hereinafter referred to as The same transmission capacity is realized in a smaller area (referred to as “small cell”), and a large number of small cells are set to realize a transmission capacity proportional to the number of small cells. However, since this small cell is required to be installed at a place where people gather and need further transmission capacity, it is not possible to sufficiently design a station as a macro cell having a large area area. Originally, if multiple frequency channels are available to introduce a small cell while the frequency resources are exhausted, it will not utilize that resource as frequency reuse (frequency reuse) but multiple at the same place It is preferable to increase the total transmission capacity by utilizing the frequency channel of Therefore, there is a need for a technology that allows repeated installation of small cells at relatively short distances without station design even with the same frequency channel.

  Furthermore, while the frequency resources of the microwave band are exhausted, it is necessary to secure a certain frequency bandwidth in order to realize a transmission capacity of 10 Gbit / s or more, and for that purpose, utilization of higher frequency bands is expected. Ru. However, since the free space propagation loss increases by 20 dB when the frequency becomes 10 times in line design, the reach of propagation is reduced to 1/10 in the line-of-sight environment with the same transmission power. Furthermore, with regard to increasing the output of the high-power amplifier on the transmission side, the higher the frequency, the more difficult it becomes, and while limiting the transmission power per one antenna, a transmission capacity of 10 Gbit / s or more sufficient for line design Technologies that can be realized are required.

  From such a point of view, large-scale MIMO (Massive MIMO) transmission technology is currently attracting attention. In Massive MIMO transmission technology, the number of antennas on the base station apparatus side is increased by at least one digit or more compared to the conventional MIMO transmission that is several at most and several tens to several hundreds of antenna elements are used Then, the channel gain improvement to the destination terminal station apparatus and the interference to the terminal station apparatus other than the destination are reduced. There are various variations in the implementation method and application method of Massive MIMO, and in the case of a so-called small cell, the reduction of interference to terminal stations other than the destination is utilized for “suppression of inter-cell interference”. Also, as Massive MIMO transmission technology, as in the prior art, of course, simultaneous communication with a plurality of terminal stations is possible in addition to use as single-user MIMO where a large-scale MIMO matrix is simply used in a single terminal station. There is also a use as multi-user MIMO that performs Here, in the case of use as multi-user MIMO, it is possible to utilize a very redundant number of antenna elements for “suppression of inter-user interference” between terminal stations within the same area unlike in the case of small cells. It is. Below, a large scale antenna system (for example, refer nonpatent literature 1 to nonpatent literature 4) is briefly demonstrated as an example of art about these large scale antennas.

[Overview of large scale antenna system]
FIG. 3 is a schematic view of a large scale antenna system. In FIG. 3, a base station apparatus 1, a radio station apparatus 2, a line of sight wave 3, a stable reflection wave 4 by a structure, multiple reflection waves 5 to 6 near the ground, and a structure 7 are shown. In the large-scale antenna system of FIG. 3, the base station device 1 includes a large number (for example, 100 or more) of antenna elements, and is installed at a high place such as a roof of a building or a high steel tower. Similarly, the radio station apparatus 2 is installed at high places such as the roof of a building, on the roof of a house, on a telegraph pole or a steel tower. Therefore, the base station device 1 and the radio station device 2 are generally in a line-of-sight environment, and in addition to the path of the line of sight wave 3 and the path of the stable reflected wave 4 by the large stable structure 7, etc. There are mixed paths of multiple reflection waves 5 and 6 due to moving objects such as cars and people in the vicinity. In addition, especially when using a directional antenna, the reception level of the multiple reflected waves 5 and 6 near the ground becomes lower than that of the line of sight wave 3 and the stable reflected wave 4 or the like.

  FIG. 4 is a diagram showing an impulse response in the line-of-sight environment and the non-line-of-sight environment. FIG. 4 (a) shows an impulse response in a non-line-of-sight environment, and FIG. 4 (b) shows an impulse response in a line-of-sight environment. In FIGS. 4A and 4B, the horizontal axis represents delay time, and the vertical axis represents the reception level of each delay wave. In the case of the non-line-of-sight environment shown in FIG. 4 (a), there are no direct wave components in the line-of-sight section, many multiple reflected waves of various paths exist as components, and each amplitude and complex phase are randomly and strongly with time. fluctuate.

  On the other hand, when assuming a line-of-sight environment like the large-scale antenna system shown in FIG. 3, the stable path of the stable reflected wave 4 by the line-of-sight 3 and the structure 7 has a high level. As shown in FIG. 4 (b), multiple reflected waves of a time-varying path generally have a larger delay amount than the stable reflected wave 4 due to the line-of-sight 3 and the structure 7 due to multiple reflection and attenuation along the path length The level gets smaller. When such channel information is acquired a plurality of times and averaged, the stable path component has stable values each time in both amplitude and complex phase. However, the components of the time-varying path are randomly combined and averaged on the complex space to approach an average value of zero. Therefore, averaging enables effective extraction of only the stable component. The absolute channel information depends on the symbol timing, and if the symbol timing is different, averaging of channel information can not be appropriately performed. In order to solve such a problem, in Non-Patent Document 2, relative channel information (or each channel information divided by the channel information of the reference antenna based on the complex phase of the reference antenna element) is considered. The technology to utilize good) is introduced. If such averaging does not accompany, it is possible to discuss using absolute channel information without using relative channel information, but even in such a case, relative channel information is There is no problem with using it. In the following description, channel information is basically treated as relative channel information based on the complex phase of the reference antenna in order to be comprehensively handled including the averaging process.

  The base station apparatus 1 (see FIG. 3) calculates transmission / reception weights based on the time-independent channel information of stable paths obtained in this manner. The base station apparatus 1 performs directivity control for performing in-phase combination with a large number of antenna elements using the calculated transmission / reception weights. By using the transmission and reception weights described above, the base station apparatus 1 can increase the directivity gain to the radio station apparatus of the communication partner to be the target of directivity control in proportion to the square of the number N of antennas.

Further, since the directivity gain of the interference to the radio station apparatus other than the destination remains N times, a gap of N times in a simple calculation is relatively generated between the desired signal and the interference signal. As a result, the expected value of Signal to Interference Ratio (SIR) is ₁₀ Log ₁₀ (N) [dB]. This expected value is 20 dB when N is 100. Further, if the radio station apparatus with small correlation is selectively spatially multiplexed, further improvement of the SIR characteristic is expected, and higher spatial multiplexing can be realized.

In Non-Patent Document 3 and Non-Patent Document 4, a combination of radio station apparatuses with lower correlation of channel information (channel correlation) and techniques for further suppressing interference which can not be suppressed by the above-mentioned transmission and reception weights is selected. Technology is introduced. In order to realize ultra-high-order spatial multiplexing, it is important to combine radio station apparatuses with small correlation of channel information. A channel vector h _j ^(k) ("h _j ^(k) " is a vector with the channel information on the kth subcarrier between the multiple antenna elements of the base station apparatus and the antenna element of the j-th radio station apparatus being a vector Yes, the symbol “→” should originally be added on h to clearly indicate that it is a vector, but it will be omitted. Hereinafter, it will be omitted similarly in the description below) and another i-th wireless station The channel correlation with the channel vector h _i ^(k) in the device is given by equation (8) below.

The system assumed a sight environment, assuming a virtual channel model consisting only of sight waves, when the correlation of the channel vector h _i between the antenna and the base station apparatus of the radio station apparatus ^_(k) is small In the case where the correlation is large, the situation is not suitable for spatial multiplexing.

[About large-scale MIMO in small cells]
In the description of Equation (8) above, it is assumed that the wireless station apparatus side is a single antenna, and if different wireless station apparatuses, the correlation of the channel vector is generally lowered due to the spatial spread. It was assumed. On the other hand, in the fifth generation mobile phone, a large number of antennas are mounted on the wireless station apparatus side and a large number of antennas are also mounted on the base station apparatus side in order to improve the throughput per user. In the recent research report, with the number of antenna elements on the base station side being 256 elements, and the number of antenna elements on the wireless station side being 16 elements, a study was made to increase the capacity by large scale MIMO of 256 × 16 size. There is. Here, it is assumed that the radio station apparatus carried by the user is small in size and portable. Furthermore, for example, in consideration of reducing mutual interference between adjacent small cells, and installation in a place where people concentrate, for example, base stations are installed on the walls of existing buildings (for example, about 20 m above ground level) It is expected that radiation is given to each antenna element to irradiate a limited area in the form of looking down from above. In this case, installing a large antenna on the wall of a building or the like is not preferable from the viewpoints of safety and ease of installation. In the case of using a high frequency band such as a millimeter wave or a quasi-millimeter wave, as the wavelength becomes shorter, miniaturization of the antenna element and formation of directivity of the antenna become easy, and a very narrow area even in the base station device It is expected to use a small antenna set packed with a large number of antennas.

  In this case, if, for example, the channel correlation between the j-th and i-th antennas of the plurality of antennas in the single radio station apparatus is determined by the above equation (8), the antenna spacing of the radio station apparatus on the user side is Under the condition that it is very short and the line of sight between the base station apparatus and the radio station apparatus can be secured, it can be regarded as a MIMO channel in which channel correlation between antenna elements is very large. As described above, when the base station apparatus is installed in a form such as looking down from below at a high place such as a wall of a building and the user holds a smartphone etc. in hand, the antenna elements of the base station apparatus and the radio station apparatus Is generally expected to be a prospect environment, and such a situation is expected to be a general use environment.

  In this case, when the MIMO matrix is subjected to singular value decomposition, the absolute value of the first singular value becomes a very high value using the line-of-sight component, but the high order singular values after the second singular value are the first It tends to be very small relatively to the singular value. In other words, as spatial multiplexing transmission utilizing MIMO channels, there is much room for channel gain for the first path corresponding to the first singular value, but for higher order paths higher than the second singular value Is a relatively inefficient situation.

  In order to avoid this problem, for example, it is ideal not to concentrate the antenna of the base station apparatus in a small-sized housing, but to ensure spatial spread. For example, if the base station apparatus antennas are assembled in a linear array in which the antennas are equally arranged in a linear shape of about 10 m, even if the line-of-sight wave is dominant, the MIMO channel is used as a MIMO channel due to the spatial spread of the base station apparatus antenna. Can be expected to increase the absolute value of the high-order singular value greater than or equal to the second singular value and to contribute to the increase in capacity. However, such a large scale structure is not preferable in terms of the structure of the antenna installation. For example, when the number of antenna elements on the base station apparatus side is 256 elements, individually installing 256 antennas at intervals of about 4 cm increases the load of installation work. On the other hand, when a structure assembled in a linear array is installed on the wall of a building, it may be difficult to install it on the wall of the building in order to increase the size of the structure. Furthermore, in order to transmit in a coordinated manner with all the antenna elements, it is necessary to distribute radio frequency signals from one case of a base station apparatus to antenna elements having a spatial extent of about 10 m by a cable, in particular Cable losses in signal transmission in high frequency bands are not negligible. For example, assuming 20 GHz as a radio frequency, a cable loss of 10 dB or more at 10 m may result in loss of acquired line gain due to an increase in the number of antenna elements at the angle.

  For such reasons, in large-scale MIMO operation where line-of-sight waves dominate, it is required to miniaturize at least the antenna on the wireless station apparatus side. Making it difficult.

[For wireless entrance]
Although the above situation has been described focusing on the small cell environment, the same problem remains in the wireless entrance line under the same condition that the line of sight wave is dominant. For example, in the case where a small cell base station device for a fifth generation mobile phone is installed on the wall of a building, a large capacity entrance line is required there. Generally, the entrance circuit to the base station apparatus is generally provided using an optical fiber, but the installation of the optical fiber on the wall of the existing building is not easy. For example, if you pull the optical fiber to the roof or somewhere on the ground and install the cable along the wall of the building, you may lose the appearance of the already built building and get the consent of the building owner Hateful. With regard to the construction in which the wall of the building is pierced to provide an optical fiber from the inside of the building to the wall of the building, it may be difficult to obtain the consent of the building owner. In this case, for example, an optical fiber may be drawn to the roof of the building facing the building where the small cell base station device is installed, and the entrance line may be provided from the roof of the building to the small cell base station device on the building wall via a wireless circuit. Conceivable.

  The problem in this case is that the base station on the wall of the building is expected to be sufficiently miniaturized in the same manner as before, so that in the line of sight environment, higher order paths of the second or more singular value of the MIMO channel are space multiplexed transmission It is assumed that the contribution to the contribution is limited, and it is considered that it is effective for stable transmission to actively utilize the first singular value.

[Wireless entrance to moving objects such as trains]
Next, another major problem will be described in consideration of the usage form of the fifth generation mobile phone. As described above, when there are a large number of users in a very narrow area, it is important to be able to provide a large capacity radio channel to the large number of users. The small cell, as the name implies, constitutes a relatively small service area, so the target for providing a large-capacity wireless channel is a user moving at a speed as low as walking speed. Therefore, it is not assumed that the majority of users switch small cells (or between a small cell and a macro cell) or the like, and the load of exchange of control information of the switching does not matter so much. However, assuming, for example, users moving in a train or the like, the timing at which each user hands over is substantially the same, and an instantaneous large-scale processing is required of the network side. This load of signal processing and compression of transmission capacity due to generation of a large amount of control signals are inefficient, and usually each user in the train temporarily aggregates traffic in the train, and this aggregated traffic is It is preferable to bundle and secure the entrance line to the external network as seen from the train by a wireless line. In this way, it is possible to avoid the load due to the useless huge amount of handover and the compression of the transmission capacity due to the huge amount of control information. This concept is captured in the concept of a moving cell in the sense that the cell, which is the service area of the fifth generation mobile phone, moves with the train.

  The entrance line to the small cell base station device of the 5th generation mobile phone on the building wall, which I mentioned earlier, could not be provided by optical fiber if impossible. Assuming that “train moving cell” is used, it is impossible to use an optical fiber as the entrance line, and it is extremely fifth generation to establish an efficient construction method of wireless entrance to this train moving cell. It becomes important in mobile phones.

  Here, for example, a train in an urban area is taken as an example to estimate the necessary line capacity. For example, in the Yamanote Line, it is assumed that about 100 passengers per vehicle are on board in an 11-car train. If it is commuting time zone, it is assumed that most businessmen and university students who carry smartphones etc. are occupied, and as many as 550 users are trying to access the Internet by wireless communication with about half of all vehicles. In the age of the 5th generation, if one user is communicating at an average of 10 Mbit / s due to the increase in the capacity of content, a capacity of about 5.5 Gbit / s is required. If the line capacity is secured about twice in consideration of the temporal variation, a capacity of about 10 Gbit / s is required as the wireless entrance of the train moving cell.

  In order to provide such a large capacity line, it is naturally necessary to provide a line in a high frequency band such as a millimeter wave or a quasi-millimeter wave, and unlike the small cell described above, a high speed of 100 km / h or more Broadband and high-capacity transmission to mobiles assuming movement is required. As mentioned above, in order to avoid frequent handovers, it is preferable that at the wireless entrance of a train moving cell, sections of several hundred meters can be served by the same base station apparatus, and free space propagation loss in high frequency bands It is also necessary to secure channel gain by large-scale MIMO if

  Here, when realizing a transmission capacity of about 10 Gbit / s as described above, for example, assuming an efficiency of about 3 bits / Hz · s as the frequency utilization efficiency, a bandwidth of about 3 to 4 GHz without spatial multiplexing I need. For example, in the case of using an e-band band of 70 to 80 GHz, when one channel occupies a bandwidth of 3 to 4 GHz, only one channel can be secured. On the other hand, however, considering dividing by frequency channel for each of a plurality of parallel routes, the bandwidth of one channel needs to be reduced to about 1 GHz, and operation with a plurality of channels is required. In this case, it is necessary to secure the insufficient capacity by space multiplexing, and in the above example, space multiplexing of about 4 multiplexing is required.

  As described above, the application of large-scale MIMO is essential even in a train moving cell, but in that case, the channel information necessary for directivity formation for high-scale MIMO gain expansion can be performed with high accuracy. You need to know. However, in the high frequency band, the wavelength is short, and the time variation of the channel appears relatively large with respect to the movement of the wireless station apparatus. Furthermore, when the opposing trains pass each other, the external environment fluctuates more violently than the fluctuation of the channel accompanying the normal linear movement, and the reflected wave component is violently disturbed. When transmitting toward the train from the base station side, channel information is essential for directivity formation, but if channel information is required to have a high degree of estimation accuracy, channel information is very frequently used. It is necessary to mutually exchange However, despite aiming to increase the transmission capacity, the fact that the overhead for control information, that is, channel information feedback is large is a fall, and avoids the decrease in transmission capacity associated with the feedback of channel information. It is required that a large capacity space multiplexing can be realized stably even in an environment where the time fluctuation is large.

[Device configuration example of MIMO transmission]
(Whole circuit configuration)
FIG. 5 is a schematic block diagram showing an example of the configuration of base station apparatus 80 in a multi-user MIMO system. Although a multi-user MIMO system will be described here, basically, if it is understood that a plurality of antenna elements in the same wireless station perform a plurality of signal sequences instead of terminal stations where targets to be spatially multiplexed differ. Are identical to single-user MIMO systems.

  As shown in FIG. 5, the base station apparatus 80 includes a transmitter 81, a receiver 85, an interface circuit 87, a MAC (Medium Access Control) layer processing circuit 88, and a communication control circuit 820. The MAC layer processing circuit 88 includes a scheduling processing circuit 881.

  The base station apparatus 80 inputs and outputs data with an external device or a network via the interface circuit 87. The interface circuit 87 detects data to be transferred on the wireless channel among the input data, and outputs the detected data to the MAC layer processing circuit 88. The MAC layer processing circuit 88 performs processing related to the MAC layer in accordance with an instruction of the communication control circuit 820 that performs management control of the entire operation of the base station device 80. Here, the processing relating to the MAC layer includes data input / output by the interface circuit 87, conversion of data transmitted / received on the wireless channel, addition of header information of the MAC layer, and the like. In this process, the scheduling process circuit 881 performs various scheduling processes including a combination of terminal stations that perform space multiplexing simultaneously in multi-user MIMO transmission. The scheduling processing circuit 881 outputs the scheduling result to the communication control circuit 820. In multi-user MIMO, a plurality of signal sequences are output from the MAC layer processing circuit 88 to the transmission unit 81 in order to transmit signals to a plurality of terminal stations at one time.

(Circuit configuration of transmission unit 81)
FIG. 6 is a schematic block diagram showing an example of a configuration of transmission section 81 in base station apparatus 80 in a multi-user MIMO system. As shown in FIG. 6, the transmission unit 81 includes transmission signal processing circuits 811-1 to 811-N _SDM (N _SDM is an integer of 2 or more), and addition synthesis circuits 812-1 to 812-812 _BS-Ant (N _BS-Ant is an integer of 2 or more), IFFT (Inverse Fast Fourier Transform: Inverse Fast Fourier Transform) & GI (Guard Interval: Guard Interval) applying circuit 813-1 to 813-N _BS-Ant , D / A (digital / Analog) converter 814-1 to 814-N _BS-Ant , local oscillator 815, mixers 816-1 to 816-N _BS-Ant , filters 817-1 to 817-N _BS-Ant , high power includes amplifier (HPA) and _{818-1~818-N BS-Ant,} an antenna element _{819-1~819-N BS-Ant,} and a transmission weight processing unit 830 That. The transmission signal processing circuits 811-1 to 811-N _SDM and the transmission weight processing unit 830 are connected to the communication control circuit 820 shown in FIG.

The transmission weight processing unit 830 includes a channel information acquisition circuit 831, a channel information storage circuit 832, and a multi-user MIMO (MU-MIMO) transmission weight calculation circuit 833. Here, _{N SDM} subscript of the transmission signal processing circuit _{811-1~811-N SDM} in FIG. 6 represents a multiplex number for performing spatial multiplexing simultaneously. Further, _{N BS-Ant} subscript circuit from additive synthesis circuit _{812-1~812-N BS-Ant} to the antenna element _{819-1~819-N BS-Ant} is the number of antenna elements provided in the base station apparatus 80 Represents N _BS-Ant is, for example, 100.

In multi-user MIMO, in order to transmit signals to a plurality of terminal stations at one time, signal sequences of a plurality of systems are input from the MAC layer processing circuit 88 to the transmitter 81, and the input signal sequences of a plurality of systems are transmission signals Processing circuits 811-1 to 811-N are input to the _SDM . The transmission signal processing circuit 811-1 to 811-N _SDM receives the data (data input # 1 to #N _SDM ) to be transmitted to each of the destination terminal station apparatuses from the MAC layer processing circuit 88 by the wireless channel A radio packet to be transmitted is generated and modulation processing is performed. Here, in the case of using, for example, the OFDM modulation scheme, the signal of each signal sequence is modulated for each subcarrier. Furthermore, the base band signal subjected to the modulation processing is multiplied by the transmission weight for each subcarrier. The signal multiplied by the transmission weight corresponding to each antenna element 819-1 to 819-N _BS-Ant is subjected to the remaining signal processing as necessary, and added and synthesized as sampling data of the transmission signal in the baseband. -1 to 812-N _BS-Ant is input.

The signals input to the adding and combining circuits 812-1 to 812-N _BS-Ant are combined for each subcarrier. The synthesized signal is converted from the signal on the frequency axis to the signal on the time axis by the IFFT & GI application circuits 813-1 to 813-N _BS-Ant , and further insertion of guard intervals or between OFDM symbols (SC-FDE If it is Single-Carrier Frequency Domain Equalization), processing such as waveform shaping between blocks of block transmission is performed, and D / A converter 814-1 is performed for each antenna element 819-1 to 819-N _BS-Ant. At 814-N _BS-Ant , the digital sampling data is converted to a baseband analog signal. Further, each analog signal is multiplied by the local oscillation signal input from the local oscillator 815 by the mixers 816-1 to 816 -N _BS-Ant , and up-converted to a radio frequency signal. Here, since the upconverted signal includes the signal in the region outside the band of the channel to be transmitted, the filter 815-1 to 817-N _BS-Ant removes the out _- of-band component and transmits the electricity to be transmitted. Generate a dynamic signal. The generated signal is amplified by the high-power amplifier _{818-1~818-N BS-Ant,} and transmitted from the antenna elements _{819-1~819-N BS-Ant.}

In FIG. 6, after the addition synthesis of the signal of each subcarrier is performed by the addition synthesis circuit 812-1 to 812-N _BS-Ant , the processing such as IFFT processing, guard interval insertion, and waveform shaping is performed. However, even if the transmission signal processing circuit 811-1 to 811-N _SDM performs these processings, the IFFT & GI application circuits 813-1 to 813-N _BS-Ant may be omitted in this position in terms of function allocation. Good. In this case, the rest of the signal processing necessary after transmission weight multiplying in the transmission signal processing circuit _{811-1~811-N SDM} refers IFFT processing, insertion of the guard interval, the process of waveform shaping or the like.

Further, transmission weight to be multiplied by the transmission signal processing circuit _{811-1~811-N SDM} is the time of signal transmission processing is acquired from the multi-user MIMO transmission weight calculating circuit 833 provided in the transmission weight processing unit 830. In the transmission weight processing unit 830, in the channel information acquisition circuit 831, the channel information acquired by the reception unit 85 is separately acquired via the communication control circuit 820, and the channel information storage circuit 832 is sequentially updated. Remember. At the time of signal transmission, multi-user MIMO transmission weight calculation circuit 833 reads channel information corresponding to the destination terminal station apparatus from channel information storage circuit 832 according to an instruction from communication control circuit 820, and based on the read channel information. The transmission weight corresponding to the combination of the destination station apparatus is calculated. Multi-user MIMO transmission weight calculation circuit 833 outputs the calculated transmission weights to transmission signal processing circuits 811-1 to 811-N _SDM .

  Further, the communication control circuit 820 manages control related to the entire communication, such as management of a terminal station apparatus as a destination and overall timing control. The communication control circuit 820 outputs information indicating a terminal station apparatus or the like as a destination to the transmission weight processing unit 830 that performs signal processing related to the calculation of the transmission weight described above.

(Circuit configuration of the receiving unit 85)
FIG. 7 is a schematic block diagram showing an example of the configuration of the receiving unit 85 in the base station apparatus 80 in a multi-user MIMO system. As shown in FIG. 7, the receiving unit 85 includes antenna elements 851-1 to 851-N _BS-Ant , low noise amplifiers (LNA) 852-1 to 852-N _BS-Ant , a local oscillator 853 and a mixer 854. -1 to 854-N _BS-Ant , filters 855-1 to 855-N _BS-Ant , A / D (analog / digital) converters 856-1 to 856-N _BS-Ant , FFT (Fast Fourier) Transform (Fast Fourier Transform) circuits 857-1 to 857-N _BS-Ant , received signal processing circuits 858-1 to 858-N _SDM, and a reception weight processing unit 860. The reception signal processing circuits 858-1 to 858 -N _SDM and the reception weight processing unit 860 are connected to the communication control circuit 820 shown in FIG. 5. The reception weight processing unit 860 includes a channel information estimation circuit 861 and a multiuser MIMO (MU-MIMO) reception weight calculation circuit 862.

The signals received by the antenna elements 851-1 to 851-N _BS-Ant are amplified by the low noise amplifier 852-1 to 852-N _BS-Ant . The amplified signal and the local oscillation signal output from local oscillator 853 are multiplied by mixers 854-1 to 854-N _BS-Ant , and the amplified signal is down converted from a radio frequency signal to a baseband signal. Ru. Since the downconverted signal includes the signal also in the region outside the frequency band to be received, the out _- of-band component is removed by the filters 855-1 to 855-N _BS-Ant . The signals from which the out-of-band components have been removed are converted into digital baseband signals by A / D converters 856-1 to 856 -N _BS-Ant . The digital baseband signals are all input to the FFT circuits 857-1 to 857-N _BS-Ant, and the signals on the time axis are taken along the frequency axis at predetermined symbol timings determined by the timing detection circuit which is not described here. Convert to upper signal (split into signals of each subcarrier). The signals separated into the respective subcarriers are input to the reception signal processing circuits 858-1 to 858 -N _SDM and are also input to the channel information estimation circuit 861.

The channel information estimation circuit 861 includes an antenna element of each terminal station apparatus and a base station apparatus 80 based on a known signal for channel estimation (such as a preamble signal added to the beginning of a radio packet) separated into each subcarrier. The channel information between each antenna element 851-1 to 851-N _BS-Ant is estimated for each subcarrier, and the estimation result is output to multiuser MIMO reception weight calculation circuit 862. The multiuser MIMO reception weight calculation circuit 862 calculates, for each subcarrier, the reception weight to be multiplied based on the input channel information. At this time, the reception weights for combining the signals received by the respective antenna elements 851-1 to 851-N _BS-Ant differ for each signal sequence, and the reception signal processing circuit 858-1 corresponding to the signal sequence to be extracted 858-N _SDM is input to each.

In received signal processing circuits 858-1 to 858-N _SDM , the signal for each subcarrier input from FFT circuits 857-1 to 857-N _{BS-Ant is} input from multi-user MIMO reception weight calculation circuit 862 The reception weights are multiplied, and the signals received by the respective antenna elements 851-1 to 851-N _BS-Ant are added and synthesized for each subcarrier. The reception signal processing circuits 858-1 to 858 -N _SDM perform demodulation processing on the added and synthesized signal, and output the reproduced data to the MAC layer processing circuit 88. In this demodulation processing, for example, soft decision of the received signal may be performed once, deinterleave processing may be performed as necessary, and error correction processing may be performed thereafter to perform final signal detection.

Here, in different received signal processing circuits 858-1 to 858 -N _SDM , signal processing of different signal sequences is performed. In addition, the MAC layer processing circuit 88 performs processing related to the MAC layer (for example, conversion between data input to and output from the interface circuit 87 and data transmitted and received on a wireless channel, termination of header information of the MAC layer, etc. Do. In this process, the scheduling process circuit 881 performs various scheduling processes including a combination of terminal stations that perform space multiplexing simultaneously in multi-user MIMO transmission, and outputs the scheduling result to the communication control circuit 820. The received data processed by the MAC layer processing circuit 88 is output to an external device or network via the interface circuit 87.

  Further, the communication control circuit 820 manages control related to the entire communication, such as management of the terminal station apparatus of the transmission source and overall timing control. Further, information indicating the terminal station apparatus or the like of the transmission source is input from the communication control circuit 820 to the reception weight processing unit 860 that performs signal processing related to the above-described calculation of the reception weight.

As for signal reception, as in the case of transmission, in the wideband system using the OFDM modulation method or SC-FDE method, the multiplication of the above-mentioned reception weight is performed for each subcarrier. That is, for the signals output from A / D converters 856-1 to 856-N _BS-Ant , FFT circuits 857-1 to 857-N _BS-Ant perform FFT to separate into subcarriers, and the sub Signal processing in channel information estimation circuit 861 and received signal processing in received signal processing circuits 858-1 to 858-N _SDM are performed for each carrier.

The above is the description of the configurations of the base station apparatus 80, the transmitter 81, and the receiver 85 in the multiuser MIMO system. As described above, for example considers the transmission signal processing circuit 811-1～N _SDM and the reception signal processing circuit 858-1～N _SDM respectively a signal processing circuit for the signal sequence of _{N SDM} systems of single terminal station device, further If the multi-user MIMO transmission weight calculation circuit 833 and the multi-user MIMO reception weight calculation circuit 862 are regarded as transmission / reception weight calculation circuits for single-user MIMO, the base station apparatus 80 and transmission in the single-user MIMO system are basically described above. The configuration of the unit 81 and the receiving unit 85 is shown.

The important point here is that the local oscillator 815 in the transmission unit 81 is shared by the mixers 816-1 to 816 -N _BS-Ant in each antenna system of the transmission unit 81, the local oscillator 853 in the reception unit 85 is a reception unit The mixers 854-1 to 854-N _BS-Ant in each of the 85 antenna systems are common points. Although the phases of the transmit and receive signals are adjusted by each antenna, the phase relationship of the signals input from the respective local oscillator 815 or the local oscillator 853 is always constant, so that transmission and reception can be performed in any phase relationship. It can be determined whether the weight should be multiplied. When the local oscillator uses a plurality of asynchronous ones in the transmitter 81 or in the receiver 85, at least the transmitter 81 does not effectively function as a directivity control that multiplies transmission weights. Care should be taken in this regard in the design of the device.

[Features of Line-of-sight Dominant MIMO Channel]
First, the spatial multiplexing characteristics in the propagation path in which the sight wave is dominant are organized. The channel matrix when the transmitting station and the receiving station are in the line-of-sight environment is H _LOS , and the channel matrix in which each component of the matrix is independent and uncorrelated is H _{i. i. d} . For simplicity, H _LOS is substituted by a matrix in which each component is all “1”, and the following channel matrix is 16 transmit antennas, 16 receive antennas, 256 transmit antennas, and 16 receive antennas. For one case, the distribution of the absolute values of the 16 singular values of the channel matrix given by the following equation (9) is evaluated.

FIG. 8 shows the distribution characteristic of the absolute value of the singular value for each channel matrix. FIG. 8 (a) shows the result of _{Hi. i. In} the case of only _d. (iid channel), FIG. 8 (b) shows the case of the Rice channel (Rician channel) expressed by the equation (9) in the case of Rice coefficient K = 10 dB In the case, the right side is the case where there are 256 transmitters. When the number of transmitting and receiving antennas is equal to 16 as well, the distribution of the absolute values of the singular values tends to widen, and the gap between the absolute values of the first singular value and the sixteenth singular values tends to widen. However, when the number of transmitting or receiving antennas becomes redundant, for example, 256 transmitting antennas, the absolute value of the singular value is H _{i. i.} There is almost no difference between the distribution probability 0% and the 100% value without being influenced by the value of the random number for each evaluation of _d . This is common to FIGS. 8 (a) and 8 (b) _{. i. While} the gap from the first singular value to the sixteenth singular value becomes very small in the case of _d. only, the gap between the first singular value and the second singular value is in the case of the rice channel in FIG. The gap between the absolute values is 20 dB, which is 10 dB greater than the Rice coefficient, while the gap between the second singular value and the sixteenth singular value is small. That is, it can be understood from FIGS. 8 (a) and 8 (b) that the top of FIG. 2 (c) corresponds to the first singular value even if the number of antenna elements is increased when the line-of-sight wave is dominant. The line gain concentrates too much on the transmission line corresponding to the pipe of (λ ₁ ), and the second (λ ₂ ) and third (λ ₃ ) pipes from the top of FIG. It means that the corresponding transmission line can hardly be used.

  On the other hand, when using a millimeter wave or the like, the free space propagation loss increases depending on the frequency, so for example, a gain of about 24 dB must be obtained somewhere at 80 GHz for 5 GHz. For this purpose, it is effective to increase the size of the antenna, but it is necessary to carry out the increase in size of the antenna for spatial multiplexing and the increase in size of the antenna for gaining line gain from different viewpoints .

[About channel information estimation and feedback]
In large-scale MIMO, the improvement of channel gain by utilizing a large number of antenna elements is an effect obtained by performing transmit / receive directivity control (beamforming) to the last, and for the directivity control, a highly accurate channel Information is needed. However, since the signal between one antenna element pair and one antenna element is a signal at a stage where the line gain is not improved before directivity formation, the estimation accuracy of the acquired channel information is generally low. . It is possible to improve the SNR with repeated reception and averaging of the same signal, but this increases the overhead of channel feedback. For example, using odd or even subcarriers of all subcarriers and intermittently assigning either odd or even subcarriers to two antenna elements respectively and using them, forward and backward with respect to unassigned subcarriers. If channel estimation is performed on the basis of channel information obtained by using the subcarriers, channel estimation of two antennas can be performed simultaneously, and also the SNR of subcarriers performing channel feedback can be improved by about 3 dB. It is. Although it is possible in principle to improve the SNR of subcarriers on which channel feedback is performed further by changing the period of intermittent subcarriers from 2 subcarrier intervals to a larger subcarrier interval as described above, In the case of general channel information, there is a problem that the frequency dependency is very large, and the accuracy of channel interpolation between discontinuous subcarriers is significantly degraded. Conversely, if the number N _SC of subcarriers to be thinned out is limited in order to improve the interpolation accuracy of the channel between the subcarriers, the amount of improvement in SNR obtained there is limited. Therefore, there is a need for techniques to provide high accuracy channel estimation and high efficiency channel feedback in high SNR environments.

  For such a problem, for example, in Non-Patent Document 5, a base station apparatus combines a large number of antennas with predetermined transmission and reception weights, and a high gain superdirective beam having a very narrow beam width in horizontal and vertical directions. It has been proposed to cope with this by forming a large number of steps at a predetermined angle. For example, in the case where the terminal device exists in an area of ± 60 degrees in the horizontal direction (120 degrees in total), forming beams in increments of 5 degrees can cover the horizontal direction with 25 directional beams. The vertical direction can be covered with seven directional beams, for example, if the beam is formed in steps of 5 degrees if the user is only within an area of about 30 degrees. In order to cover the vertical and horizontal directions simultaneously, a total of 175 fixed directional beams may be used. If this step size is set to 10 degrees, 52 patterns of fixed directional beams can similarly cover the whole. If a plurality of such fixed beams are selected and considered as a MIMO channel between as many virtual antenna elements as that number and the antenna elements of the terminal station apparatus, it is possible to If channel gain can be sufficiently secured, channel information can be obtained. However, if the training signal is not transmitted by the number of virtual antenna elements, the terminal station apparatus side can not know which virtual antenna element can be used to secure a good communication environment. Overhead required to transmit the target antenna element is required. In this way, although it is possible to obtain MIMO channel information for realizing high-order spatial multiplexing while securing sufficient channel gain, it is still possible to efficiently obtain it with a small overhead. , Further devise is needed.

[About calibration and implicit feedback]
In an actual wireless communication apparatus, in signal processing on the transmission side, signal amplification is often performed by a high power amplifier immediately before transmission. In this case, there is an error in the amplification factor due to the individual difference of the high power amplifier, and in the high power amplifier, the complex phase may rotate at different values for each high power amplifier. Similarly, in signal processing on the reception side, signal amplification is often performed by a low noise amplifier immediately after reception. In this case, there is an error in the amplification factor due to the individual difference of the low noise amplifier, and in the low noise amplifier, the complex phase may rotate at a different value for each low noise amplifier. Strictly speaking, individual differences also occur in circuits of the transmission system and reception system including the RF system circuits such as other filters.

  Generally, the amplification factor and the amount of phase rotation of the high power amplifier and the low noise amplifier have frequency dependency. When individual differences between the amplification factor with frequency dependence and the amount of rotation of the complex phase are so large that they can not be ignored, it is necessary to perform calibration processing when estimating downlink channel information from uplink channel information. is there. If the error in the amplification factor and the amount of phase rotation is substantially stable in time, the error in the amplification factor and the amount of phase rotation is measured in advance, and the error is canceled using the coefficient to cancel the effect of the error. Convert link channel information to downlink channel information. Generally, a channel estimation method using such a property is called implicit feedback, and it is distinguished from explicit feedback in which channel information estimated in the downlink is accommodated as digital data in control information and notified in the uplink. ing.

  An example of the calibration process will be described below. FIG. 9 is a diagram showing asymmetry of channel information on uplink and downlink. In FIG. 9, reference numerals 25-0 to 25-2 denote wireless modules, reference numerals 21-0 to 21-2 denote high power amplifiers (HPAs), and reference numerals 22-0 to 22-2 denote low noise amplifiers (LNAs). , And symbols 23-0 to 23-2 indicate time division switches (TDD-SW), and symbols 24-0 to 24-2 indicate antenna elements.

Here, since only the function affecting the channel information is extracted in the base station apparatus, configurations other than those illustrated are omitted, but the wireless modules 25-0 to 25-2 also include other functions. In addition, when the signal passes through each of the high power amplifiers 21-0 to 21-2, the amplitude and the complex phase are Z _{HPA # 0} (f _k ), Z _{HPA # 1} (f _k ), Z _{HPA # 2} (f _k ) It shall change. In addition, when it is assumed that the amplitude is A and the complex phase is θ when passing through various amplifiers, the influence can be represented by a coefficient of Z = A · e ^jθ . When the signal passes through each of the low noise amplifiers 22-0 to 22-2, the coefficients of the influence by the amplitude and the complex phase are Z _{LNA # 0} (f _k ), Z _{LNA # 1} (f _k ), Z _{LNA # 2} ( It is assumed that f _k ). Here, it is assumed that there is frequency dependency as a general condition, and the frequency “(f _k )” for the kth subcarrier is described.

Here, for example, channel information in the case of transmitting a signal from the wireless module 25-1 and the wireless module 25-2 to the test wireless module 25-0 will be described. Here, channel information on the space between the antenna element 24-1 of the wireless module 25-1 and the antenna element 24-3 of the wireless module 25-0 is represented by h ₁ (f _k ), and the wireless module 25 is represented. The channel information on the space between the antenna element 24-2 of No. 2 and the antenna element 24-0 of the wireless module 25-0 is represented by h ₂ (f _k ).

At this time, channel information at the time of actually transmitting a signal from the wireless module 25-1 to the wireless module 25-0 indicates a change associated with the passage of the high power amplifier 21-1 in h ₁ (f _k ) in space. It is observed as a value multiplied by the coefficient Z _{HPA # 1} (f _k ) and the coefficient Z _{LNA # 0} (f _k ) indicating a change accompanying the passage of the low noise amplifier 22-0.

Similarly, channel information at the time of transmitting a signal from the wireless module 25-2 to the wireless module 25-0 has a coefficient Z indicating a change associated with the passage of the high power amplifier 21-2 to h ₂ (f _k ) in space. It is observed as a value multiplied by _{HPA # 2} (f _k ) and a coefficient Z _{LNA # 0} (f _k ) indicating a change accompanying passage of the low noise amplifier 22-0.

Therefore, the channel from the wireless module 25-1 to the wireless module 25-0 is represented by Z _{HPA # 1} (f _k ) · h ₁ (f _k ) · Z _{LNA # 0} (f _k ). Further, a channel from the wireless module 25-2 to the wireless module 25-0 is represented by Z _{HPA # 2} (f _k ) · h ₂ (f _k ) · Z _{LNA # 0} (f _k ). Therefore, in addition to the difference between the channel information h ₁ (f _k ) and h ₂ (f _k ) between the wireless module 25-1 and the wireless module 25-2, the relative value of Z _{HPA # 2} (f _k ) / Z _{HPA # 1} (f _k ) difference occurs.

This situation is the same in reverse communication, and when the signal transmitted from the wireless module 25-0 is received by the wireless module 25-1, the channel information is high power in h ₁ (f _k ) in space A value obtained by multiplying a coefficient Z _{HPA # 0} (f _k ) indicating a change accompanying the passage of the amplifier 21-0 and a coefficient Z _{LNA # 1} (f _k ) indicating a change accompanying the passage of the low noise amplifier 22-1 It is observed.

Similarly, when the signal transmitted from the wireless module 25-0 is received by the wireless module 25-2, the channel information changes h ₂ (f _k ) in space as the high power amplifier 21-0 passes. It is observed as a value obtained by multiplying the indicated coefficient Z _{HPA # 0} (f _k ) and the coefficient Z _{LNA # 2} (f _k ) indicating the change accompanying the passage of the low noise amplifier 22-2.

Therefore, the channel from the wireless module 25-0 to the wireless module 25-1 is represented by Z _{HPA # 0} (f _k ) · h ₁ (f _k ) · Z _{LNA # 1} (f _k ). A channel from the wireless module 25-0 to the wireless module 25-2 is represented by Z _{HPA # 0} (f _k ) · h ₂ (f _k ) · Z _{LNA # 2} (f _k ). Therefore, in addition to the difference between the channel information h ₁ (f _k ) and h ₂ (f _k ) between the wireless module 25-1 and the wireless module 25-2, the relative value of Z _{LNA # 2} (f _{k) is determined.} ) / Z _{LNA # 1} (f _k ) occurs.

  As described above, although the base station apparatus can acquire channel information including changes due to the low noise amplifiers 22-1 to 22-2 connected to each antenna element in the uplink, the base station apparatus is not capable of downlink It is not possible to directly obtain channel information in Therefore, downlink channel information is acquired by converting from uplink channel information. For this conversion, the influence of the individual differences of the low noise amplifiers 22-0 to 22-2 and the high power amplifiers 21-0 to 21-2 connected to the respective antenna elements 24-0 to 24-2 is cancelled. There is a need.

  Therefore, in the manufacturing stage of the base station apparatus, the test wireless module 25-0 is prepared, and the antenna terminal of the test wireless module 25-0 and the antenna terminal of the wireless modules 25-1 and 25-2 are prepared. And directly with a cable. In an environment where channel information on the propagation path has a common value, channel information including changes by the high power amplifiers 21-0 to 21-2 and low noise amplifiers 22-0 to 22-2 is measured, and the measured channel information is used. Make corrections.

  FIG. 10 is a diagram showing an outline of the calibration. In FIG. 10, reference numerals 26-0 to 26-2 denote antenna terminals, and reference numeral 27 denotes a coaxial cable. The same functional units as those shown in FIG. 9 are denoted by the same reference numerals.

  FIG. 10A shows a configuration in which the wireless module 25-0 and the wireless module 25-1 are connected by a coaxial cable. FIG. 10B shows a configuration in which the wireless module 25-0 and the wireless module 25-2 are connected by a coaxial cable. While FIG. 9 shows a state in which the signal is propagated in the actual space, FIG. 10 shows a state in which the signal is propagated on the coaxial cable without passing through the antenna element.

The channel information of the coaxial cable 27 as a propagation path connecting the wireless modules 25-1 and 25-2 and the wireless module 25-0 is h ₀ (f _k ).
At this time, the channel information from the wireless module 25-1 to the wireless module 25-0 is represented by Z _{HPA # 1} (f _k ) · h ₀ (f _k ) · Z _{LNA # 0} (f _k ). The channel information from the wireless module 25-2 to the wireless module 25-0 is represented by Z _{HPA # 2} (f _k ) · h ₀ (f _k ) · Z _{LNA # 0} (f _k ).

Further, channel information from the wireless module 25-0 to the wireless module 25-1 is represented by Z _{HPA # 0} (f _k ) · h ₀ (f _k ) · Z _{LNA # 1} (f _k ), and the wireless module 25 is represented. The channel information from −0 to the wireless module 25-2 is represented by Z _{HPA # 0} (f _k ) · h ₀ (f _k ) · Z _{LNA # 2} (f _k ).
Therefore, after these pieces of channel information are measured, calibration coefficients C ₁ (f _k ) and C ₂ (f _k ) expressed by the following equations (10) and (11) are calculated.

Earlier, the channel information from the wireless module 25-0 to the wireless module 25-1 is represented by Z _{HPA # 0} (f _k ) · h ₁ (f _k ) · Z _{LNA # 1} (f _k ), and the wireless module 25- It has been described that the channel information from 0 to the wireless module 25-2 is represented by Z _{HPA # 0} (f _k ) h ₂ (f _k ) Z _{LNA # 2} (f _k ). If these are multiplied by the calibration coefficients C ₁ (f _k ) and C ₂ (f _k ) of the equations (10) and (11), the following equations (12) and (13) are obtained.

  The right side of Equations (12) and (13) is the channel information from the wireless module 25-1 to the wireless module 25-0 described above and the channel information from the wireless module 25-2 to the wireless module 25-0. Matches.

  In this manner, calibration coefficients corresponding to the equations (10) and (11) are obtained in the manufacturing stage of the base station apparatus, and stored in the base station apparatus, whereby these calibrations are performed. The downlink channel information can be calculated from the uplink channel information using the coefficients.

  In the following description, these calibration coefficients are obtained in advance, and the values are used mainly in digital signal processing. Naturally, on the analog circuit, the inside of the base station apparatus and all the calibration coefficients have substantially constant values (the absolute value itself may have a difference if the complex phase is a constant value). If adjustment is performed in the terminal station apparatus, it is possible to read it as a process in which all calibration coefficients are considered to be 1. In particular, since the frequency dependency of the amplification factor of the amplitude is directly linked to the waveform distortion, it is generally industrially designed so that the characteristic of the amplifier is approximately a constant (flat waveform) on the frequency axis.

  Similarly, even when the complex phases of the uplink and the downlink are constant values, even if they are collectively adjusted in the latter stage (during reception) of the digital signal processing and the former stage (during transmission) of the IFFT in digital signal processing, As a result, since the complex phase of the calibration coefficient represented by the equation (10) and the equation (11) becomes substantially constant in all the antenna elements, the same effect can be obtained.

  The above description is about calibration for one-to-plural SIMO channels, but in the case of calibration of channel matrices of multiple antennas vs. multiple antennas, it is necessary to explain this in a form slightly corrected. It is. For example, as shown in FIG. 11, the case of two transmitting and receiving antenna elements is considered. In order to simplify the description, the left side is regarded as the base station apparatus 41, the right side is regarded as the terminal station apparatus 51, and the downlink which is communication from left to right and the uplink which is communication from right to left. Organize the relationship of the channel matrix.

  When considering the downlink first, the channel information observed by the antenna element of the terminal station apparatus 51 on the receiving side is added to each component of the channel matrix in the space between the transmitting antenna end and the receiving antenna end, and the base station apparatus The amount of rotation of the amplification factor and complex phase in the high power amplifier for transmission 41 and the amount of rotation of the amplification factor and complex phase in the low noise amplifier of the terminal station apparatus are multiplied. These rotation amounts differ for each high power amplifier corresponding to a plurality of antenna elements of the base station apparatus, and also differ for each low noise amplifier corresponding to a plurality of antenna elements of the terminal station apparatus. On the other hand, in the uplink, channel information observed by the antenna element of the base station apparatus on the receiving side is added to each component of the channel matrix in the space between the transmitting antenna end and the receiving antenna end, The amplification factor and the rotation amount of the complex phase in the high power amplifier for transmission are multiplied by the amplification factor and the rotation amount of the complex phase in the low noise amplifier of the base station apparatus. The amount of rotation differs for each high power amplifier corresponding to a plurality of antenna elements of the terminal station apparatus, and also differs for each low noise amplifier corresponding to a plurality of antenna elements of the base station apparatus. Here, regarding the channel matrix in the space between the transmit antenna end and the receive antenna end, there is no change in the value of each component itself because of the symmetry of the space, but the transmit antenna and the receive antenna are used in uplink and downlink. As the relationship between the base station apparatus and the terminal station apparatus is reversed, the spatial channel matrix can be converted between uplink and downlink by transposing the matrix. In addition to this, if the same processing as calibration in the above-described SIMO channel is performed, calibration for converting the channel state between uplink and downlink between any test transmitting / receiving apparatus It is possible to obtain the downlink channel matrix from the uplink channel matrix using the channel coefficients.

Here, θ BS-HPA, m (k) is the amount of phase rotation of the high power amplifier corresponding to the m-th antenna element of the base station apparatus, and A BS-HPA, m (k) is the m-th antenna of the base station apparatus Amplitude amplification amount of the high power amplifier corresponding to the element, θ BS -LNA, m (k) the phase rotation amount of the low noise amplifier corresponding to the mth antenna element of the base station apparatus, A BS -LNA, m ( Let k) be the amplification amount of the amplitude of the low noise amplifier corresponding to the m-th antenna element of the base station apparatus. Further, let θ MT-HPA, m (k) be the amount of phase rotation of the high power amplifier corresponding to the m-th antenna element of the terminal station device, and A MT-HPA, m (k) be the m-th antenna element of the terminal station device Amplitude amplification amount of the corresponding high power amplifier, θ MT -LNA, m (k) the phase rotation amount of the low noise amplifier corresponding to the mth antenna element of the terminal station apparatus, A MT -LNA, m (k) Is the amplification amount of the amplitude of the low noise amplifier corresponding to the m-th antenna element of the terminal station apparatus. Further, the phase rotation amount of the high-power amplifier station for testing θ Test-HPA (k), and the amplification of the amplitude of the high-power amplifier of A Test-HPA (k) station for testing, theta Test-LNA ( Let k) be the phase rotation amount of the test station, and A Test-LNA (k) be the amplification amount of the low noise amplifier of the test station. If BS (base station apparatus) and MT (terminal station apparatus) suffixed to each coefficient are generalized and shown as STA (base station apparatus or terminal station apparatus), the calibration coefficient for the mth antenna element of STA can be expressed as It defines as 14), and it becomes possible to acquire a downlink channel matrix from an uplink channel matrix like Formula (15) using this.

In general, the coefficients by which all the elements of the channel matrix are equally multiplied have no effect on the calculation of the transmission / reception weight matrix (vector) even when evaluating the channel characteristics indicated by the channel matrix. It is possible to treat it as an unknown factor. In this sense, in equation (15), the coefficients θ _Test-HPA ^(k) , A _Test-HPA ^(k) , θ _Test-LNA ^(k) , and A _Test-LNA ^{(k )} Although a contains, for all of which are coefficients equal multiplied to all the elements of the channel matrix, this value is not particularly affect especially when obtaining reception weight matrix (vector). Furthermore, the local oscillator signal input to the mixer when converting a baseband signal to a radio frequency at the transmitting station side, and the local oscillator input when mixing a radio frequency signal to a baseband signal at the receiving station side The state of the complex phase of each of the signals in, etc. actually affects the complex phase of each component of the observed channel matrix, but all coefficients are multiplied equally by all antennas, so Also, all components of the channel matrix will be multiplied equally as coefficients. Since this coefficient also has no influence on the transmission / reception weight matrix (vector), it is possible to carry out the evaluation ignoring these effects. In this way, if the calibration coefficients are known in advance, it is possible to convert the channel matrix between the uplink and the downlink based on the calibration coefficients. Similarly, if a reception weight vector can be obtained, it becomes possible to calculate a transmission weight vector by applying a calibration coefficient to the reception weight vector.

[Wireless entrance in the mobile front hall]
The same applies to the 4th generation mobile phones, but most of the base station devices are supposed to improve the maintainability of the base station devices that need to be installed in a large number, coordinated transmission from a plurality of base station devices, etc. (BBU: Base Band Unit) in one place, (Function 1) Generation of analog baseband signal by D / A conversion from digital sampling data, (Function 2) Analog baseband signal to radio frequency Up-conversion to band signals, (function 3) power amplification and transmission from antenna, (function 4) antenna reception and power amplification, (function 5) down conversion of radio frequency band signals to analog baseband signals , (Function 6) A remote radio head with only functions such as generation of digital sampling data by A / D conversion from analog baseband signal A cloud-type radio access network (C-RAN: Centralized Radio Access Network or Cloud Radio Access Network) in which only RRH (Remote Radio Head) is distributed is effective. This is a technology called "front hall" or "mobile front hall" in contrast to so-called "backhaul circuit", and CPRI (an interface standard for transmitting digital sampling data by optical fiber) Common Public Radio Interface) is standardized.

Here, in order to perform joint transmission (JT: Joint Transmit) or joint reception (JR: Joint Reception) with a plurality of RRHs, delay jitter is suppressed with an accuracy of 10 ^-9 order, and on an optical fiber. The bit error rate is required to have a high quality on the order of 10 ^{-12 and} a low delay of 100 μs or less. This mobile front hole can be provided not only by an optical fiber but also by a wireless circuit, in which case it is a wireless circuit, but with a delay jitter of 10 ^-9 , a bit error rate of 10 ^-12 , and a transmission delay of less than 100 μs. We must achieve high quality.

Among existing wireless entrance devices, large-scale parabolic antennas form ultra-high gain and high directivity of 40 dBi to 50 dBi, and this antenna is opposed to this to eliminate multi-path components almost without distortion. There is a transmission path which makes it possible to realize a bit error rate of 10 ^-12 equivalent to that of an optical fiber in one transmission without retransmission control. However, when securing directivity gain by utilizing large-scale MIMO as described above, the redundancy of the antenna is used, and multipath components are used in the same manner as in ordinary space multiplexing, and thus multiple streams of It was also possible to realize multiplexing. However, when such a multipath component is taken in, the effect of nonlinear distortion can not be ignored, and in general, the error rate characteristic draws an error floor and achieves a sufficiently low error rate even in a high SNR environment. It was difficult. In this case, although it is necessary to improve the error rate performance by performing retransmission control, in general, retransmission control involves a large transmission processing delay, and it is difficult to complete retransmission control with a low delay of 100 μs or less. Furthermore, when performing high-efficiency retransmission control in ultra-wide band large-capacity transmission, the process of selective retransmission control is complicated, and additionally, due to the restriction on the delay of 100 μs or less, regardless of transmission success / failure 2-3 times At the upper limit of the number of times of transmission, processing to stop retransmission and discard is also required. In the end of the retransmission, this may cause a mismatch between the sender and the receiver, but this mismatch between the sender and the receiver leads to a communication deadlock condition. However, if the negotiation process for avoiding such a state is performed, the process will be further complicated, which is not preferable. It is required to realize ultra-high-speed and highly-efficient retransmission control while avoiding the complication of the signal processing.

  In this mobile front hall, as described above, between BBU and RRH, digital sampling data of the signal to be transmitted from the antenna element of RRH is transmitted, and the above-mentioned (Function 1) to (Function 6) To carry out the function of) and transmit and receive the signal of the actual radio frequency. However, when using a large scale antenna in the 5th generation small cell base station apparatus, if the conventional method of transmitting digital sampling data on an optical fiber is followed, each antenna element Needed to transmit digital sampling data. Generally, the transmission capacity of digital sampling data is said to be about 16 times the amount of user data to be actually transmitted in the wireless section, and it is a case where data transmission of 1 Gbit / s is temporarily performed in the wireless section. Even in the mobile fronthaul section, 16 Gbit / s worth of digital sampling data must be transmitted. However, if large-scale MIMO technology using antenna elements of 256 elements is adopted in the 5th generation small cell base station apparatus, the transmission capacity required in the mobile front hall will be 4 Tbit / s, and the transmission medium Even if it is an optical fiber, it is an unrealistic area at present. Generally, a relatively inexpensive and usable system is a widely spread system having a low transmission rate, and at present is a communication system of about 10 Gbit / s. Even if future technological innovation is considered, the data capacity to be transmitted on the mobile front hall is not allowed to diverge as the number of antenna elements increases, and at the same time digital sampling for the number of signal systems to be spatially multiplexed・ It is required to fit within the capacity of data.

  From this point of view, a review of the function allocation between BBU and RRH is currently under consideration, and in addition to the functions (function 1) to (function 6) described above, the BBU side is originally implemented. In addition to the modulation function and the demodulation function of the physical layer in the radio signal processing, it is considered to implement complex MIMO signal processing, FFT / IFFT signal processing, etc. on the RRH side. This will significantly reduce the amount of information to be transmitted on the mobile front hall, which is far from the original mobile front hall concept. In other words, since (Function 1) to (Function 6) do not include functions specialized to the modulation / demodulation system specific to the wireless system to be adopted, only when changing the BBU side when upgrading the system version in the future If the RRH can be handled without modification, if the function allocation is reviewed as described above, the version is upgraded even if the basic design parameters such as the change in the number of subcarriers are changed. For this reason, remodeling of RRH itself will be forced.

  Therefore, there is a need for a device that can cope with large-scale MIMO but does not increase the transmission capacity of digital sampling data to be transmitted on the mobile front hall.

  The following is a description of the mobile front hall based on the figure. FIG. 12 is a diagram outlining the function sharing of the mobile fronthaul and the mobile backhaul in the prior art. In particular, FIG. 12 (a) represents the mobile fronthaul and FIG. 12 (b) represents the mobile backhaul. Here, only functions related to the transmission of signals in the direction from the network side to the user (in the case of a mobile front hall, from the BBU to the RRH) are extracted. In the figure, 401 is a MAC layer processing circuit, 402 is a transmission signal processing circuit, 403 is a time axis signal generation circuit, 404 is an optical interface circuit, 405 is an optical fiber, 406 is an optical interface circuit, 407 is a D / A converter, An RF processing circuit 408, an antenna element 409, a BBU 411-1, an RRH 412-1, an optical interface circuit 414, an optical fiber 415, and an optical interface 416 are illustrated. The MAC layer processing circuit 401, the transmission signal processing circuit 402, and the time axis signal generation circuit 403 collectively constitute areas 400 and 410 for performing baseband signal processing related to radio.

  In FIG. 12 (a), when a signal to be transmitted to BBU 441-1 is input from the network side, MAC layer processing circuit 401 performs signal processing of the MAC layer, and a frame format used for transmission and reception in a wireless section It converts and terminates the frame format of data flowing on the side, and inputs a signal of the format of a wireless packet to the transmission signal processing circuit 402. The transmission signal processing circuit 402 performs transmission signal processing of a wireless signal. Here, the transmission scheme in the wireless section is not particularly limited. For example, if OFDM is used, error correction coding, interleaving, modulation processing for each subcarrier, and the like are performed as needed. The signal generated in this manner is converted into a time axis signal by the time axis signal generation circuit 403. For example, in the case of OFDM as described above, IFFT is performed to convert a signal on the frequency axis into a signal on the time axis, insert a guard interval, and perform waveform shaping processing between symbols. As a result, each sampling value of the digital baseband signal is converted into a continuous signal in time series. The data of these sampling values are converted into a predetermined frame format by the optical interface circuit 404, converted from an electrical signal into an optical signal, and output to the optical fiber 405. The signal output to the optical fiber 405 is transmitted to the RRH 442-1 side, and in the RRH 442-1, the optical interface circuit 406 converts the optical signal into an electrical signal to terminate a signal of a predetermined format, and as a digital baseband signal Generate an information string of sampling values. The D / A converter 407 converts this to an analog baseband signal at a predetermined clock rate, the RF processing circuit 408 converts it to a radio frequency signal with an up converter, and the filter removes high frequency radiation from the out-of-band radiation signal. It is amplified by a power amplifier and transmitted from the antenna 409 into space. As described above, the functions equivalent to the base station apparatus of the radio signal as a whole are divided and accommodated in the BBU 441-1 and the RRH 442-1 intervened by the optical fiber. The feature here is that since wireless digital baseband signal processing is concentrated in the BBU provided in the network building on the network side, even if there is any change in the wireless communication method, all changes are made on the BBU side. There is a merit that it is finished.

  On the other hand, in the mobile backhaul, the configuration shown in FIG. 12B is adopted as a configuration corresponding to FIG. For example, with respect to a base station apparatus installed on a roof of a building or on a telegraph pole, user data is directly transmitted by an optical fiber, and all are closed in the base station apparatus to perform radio signal processing. Specifically, on the network side, the optical interface circuit 414 transmits user data to be transmitted by a wireless channel to the optical interface circuit 416 in the base station apparatus via the optical fiber 415. The optical interface circuit 416 converts an optical signal into an electrical signal, terminates a signal of a predetermined format, and outputs user data to the MAC layer processing circuit 401. The MAC layer processing circuit 401 performs signal processing of the MAC layer, converts and terminates the frame format used for transmission and reception in the wireless section, and the frame format of data flowing through the network side, and transmits the signal of the wireless packet format to the transmission signal processing Input to the circuit 402. The transmission signal processing circuit 402 performs transmission signal processing of a wireless signal. Here, the transmission scheme in the wireless section is not particularly limited. For example, if OFDM is used, error correction coding, interleaving, modulation processing for each subcarrier, and the like are performed as needed. The signal generated in this manner is converted into a time axis signal by the time axis signal generation circuit 403. For example, in the case of OFDM as described above, IFFT is performed to convert a signal on the frequency axis into a signal on the time axis, insert a guard interval, and perform waveform shaping processing between symbols. As a result, each sampling value of the digital baseband signal is converted into a continuous signal in time series. The data of these sampling values are input to the D / A converter 407, the D / A converter 407 converts it into an analog baseband signal at a predetermined clock rate, and the RF processing circuit 408 converts the radio frequency signal by the up converter. And then remove the out-of-band radiation signal with a filter, then amplify with a high power amplifier, and transmit it from the antenna 409 to space. The description of the signal processing and circuit configuration of the base station apparatus described so far is conscious of this mobile backhaul.

  In the description of FIG. 12 above, the antenna element included in the RRH 442-1 is one element, but in the mobile network in the future, application of Massive MIMO technology using multi-element antennas is assumed, The amount of information to be transmitted will increase. FIG. 13 is a diagram outlining the function sharing of the mobile fronthaul and the mobile backhaul in the prior art when using a multi-element antenna. In particular, FIG. 13 (a) represents the mobile fronthaul and FIG. 13 (b) represents the mobile backhaul. Here, as in FIG. 12, only the functions relating to the transmission of signals in the direction from the network side to the user (in the case of a mobile front hall, from the BBU to the RRH) are extracted. In FIG. 13, the same symbols are assigned to the same components as those in FIG. In FIG. 13, 412 is a transmission signal processing circuit, 424 is an optical interface circuit, 425 is an optical fiber, 426 is an optical interface circuit, 427 is a D / A converter, 428 is an RF processing circuit, 429 is an antenna element, and 430 is a transmission. The weight multiplication circuit 431 is a transmission weight processing circuit, 433 is a time axis signal generation circuit, 441-2 is a BBU, and 442 is an RRH. The MAC layer processing circuit 401, the transmission signal processing circuit 412, the transmission weight multiplication circuit 430, the transmission weight processing circuit 431, and the time axis signal generation circuit 433 collectively constitute an area 420 for performing baseband signal processing related to radio. Similarly, the MAC layer processing circuit 401, the transmission signal processing circuit 412, the transmission weight multiplication circuit 430, the transmission weight processing circuit 431, and the time axis signal generation circuit 433 collectively constitute an area 420 or 440 for performing baseband signal processing related to radio. Do. In FIG. 13A, when a signal to be transmitted is input to BBU 441-2 from the network side, MAC layer processing circuit 401 performs signal processing of the MAC layer, and a frame format used for transmission and reception in a wireless section, and the network It converts and terminates the frame format of data flowing on the side, and inputs a signal of the format of a wireless packet to the transmission signal processing circuit 412. The transmission signal processing circuit 412 performs transmission signal processing of a wireless signal. Here, the transmission scheme in the wireless section is not particularly limited. For example, if OFDM is used, error correction coding, interleaving, modulation processing for each subcarrier, and the like are performed as needed. In addition, for example, reception weight information and the like collected on the reception side of the BBU not shown here are provided to the transmission weight processing circuit 431 via the transmission signal processing circuit 412. The transmission weight processing circuit 431 performs calibration processing on the reception weight information and the like to calculate the transmission weight, and outputs this to the transmission weight multiplication circuit 430. The transmission weight multiplication circuit 430 multiplies the transmission signal input from the transmission signal processing circuit 412 by the transmission weight. The processing here is performed on the frequency axis in the case of using OFDM, for example. That is, using different transmission weights for each subcarrier calculated based on different reception weights for each subcarrier, transmission signals different for each subcarrier are multiplied for each subcarrier. The signal generated in this manner is converted by the time axis signal generation circuit 433 into a time axis signal for each antenna system. For example, taking the case of OFDM as an example, an antenna sequence that performs IFFT to convert a signal on the frequency axis into a signal on the time axis, inserts a guard interval, and performs waveform shaping between symbols, etc. Implement every time. As a result, each sampling value of the digital baseband signal is converted into a signal for each successive antenna element in time series. The data of these sampling values are converted into a predetermined frame format by the optical interface circuit 424, converted from an electrical signal into an optical signal, and output to the optical fiber 425. Here, unlike in the case of FIG. 12 mentioned above, it is necessary to simultaneously accommodate the signals for the number of antenna elements on the optical fiber, and the capacity can be increased by any method such as wavelength multiplexing, TDM, or using a plurality of optical fibers. The RRH 442-2 is transmitted on the transmission path of the optical fiber where it was made. In the RRH 442, an optical interface circuit 426 converts an optical signal into an electrical signal, terminates a signal of a predetermined format, and generates an information string of sampling values as a digital baseband signal. As described above, here, sampling signals of a system equivalent to the number of antenna elements are reproduced. The D / A converter 427 converts this to an analog baseband signal for each antenna system at a predetermined clock rate, and the RF processing circuit 428 uses an upconverter using a common local oscillator for each antenna system to generate a radio frequency signal. After the out-of-band radiation signal is removed individually by a filter, it is individually amplified by a high power amplifier, and this is transmitted to space from individual antenna elements 429. As described above, the functions equivalent to the base station apparatus of the radio signal as a whole are divided and accommodated in the BBU 441-2 and the RRH 442-2 mediated by the optical fiber.

  The above description is based on the original mobile fronthaul policy, but considering the transmission capacity of the signal actually flowing on the optical fiber, the number of antenna elements is limited to only a few, for example, a sector antenna of several sectors. It is thought that In the case of 100 or more antenna elements assumed in Massive MIMO etc., it is a general idea that the cost will be enormous in consideration of the number of optical fibers to be laid and the capacity of data to be transmitted.

  On the other hand, in the mobile backhaul, the configuration shown in FIG. 13B is adopted as a configuration corresponding to FIG. Referring to FIG. 12B, the area 410 for performing radio signal processing is replaced by an area 440 for performing radio signal processing of the same function as the area 420 for performing radio signal processing of FIG. The process after the A converter 427 is the same as in FIG. 13A, except that it is performed for each antenna system. The feature here is that although the number of antenna elements of the base station apparatus increases, the information capacity on the transmission path from the optical interface circuit 414 to the optical interface circuit 416 in the base station apparatus via the optical fiber 415 increases. It is not doing. Therefore, the problem as shown in FIG. 13 (a) does not occur. However, there is no solution to the problem in which the mobile front hole as shown in FIG. 12 (a) is useful, but in general, the mobile front hole can not be used while the transmission capacity on the optical fiber is reduced. There is a need for a configuration that is compatible with the advantages.

  In such a study, a review of the functional allocation between the BBU and the RRH is being made as a mobile midhaul between the fronthaul and the backhaul. FIG. 14 shows a configuration example of a mobile midhole in the prior art. Here, as in FIG. 12, only the functions relating to the transmission of signals (BBU to RRH direction) regarding the direction from the network side to the user are extracted. The same reference numerals are given to the same components as those in FIGS. 12 and 13. The MAC layer processing circuit 401, the transmission signal processing circuit 412, the transmission weight multiplication circuit 430, the transmission weight processing circuit 431, and the time axis signal generation circuit 433 collectively have the areas 445-1 and 445-2 for performing baseband signal processing related to radio. Although the configuration is different from FIGS. 12A and 13A, the MAC layer processing circuit 401 in the area 445-1 on the BBU side, the transmission signal processing circuit 412, the transmission weight multiplication circuit 430, the transmission weight processing The circuit 431 and the time axis signal generation circuit 433 are separately arranged in the area 445-2 on the RRH side, and the entire function distribution is reviewed.

  By reviewing the above function allocation, in the case of the mobile mid-hole shown in FIG. 14, the transmission capacity on the optical fiber is about the same as that shown in FIG. 12 (a), and the amount of information is large as shown in FIG. Without increasing it. In this sense, the suppression effect of the transmission capacity on the optical fiber is sufficiently obtained. However, from the viewpoint of eliminating the processing dependent on the radio transmission method from the RRH 442-3 side, which is originally the purpose of the mobile fronthole, and concentrating the processing dependent on all radio methods on the BBU 441-3 side, The arrangement of the processing in the area 445-2 on the RRH 442-3 side is not sufficiently suitable for the purpose of the front hole as compared with the configuration of the mobile front hole shown in FIG. 12 (a). In this sense, it is also called a mid-hole as an intermediate between the front hole and back hole, but ideally it depends on the wireless transmission method on the BBU side, while mounting many antenna elements on the RRH side It is preferable that the processing to be performed be integrated, and that the information capacity flowing on the optical fiber can be suppressed to the same extent as the backhaul.

[Other issues]
It is known that techniques such as OFDM and SC-FDE are generally effective for removing multipath components in a multipath environment. However, in the existing OFDM and SC-FDE systems, the base station apparatus and the radio station apparatus generally use their own asynchronous clocks and local oscillators, and perform transmission processing in burst mode. In the burst mode, even if there is a clock error, there is no problem because the process is reset for each burst. Furthermore, with regard to clock errors and local oscillator frequency errors, if the errors do not break the orthogonality of each subcarrier at the time of FFT, higher precision synchronization can not be required. Therefore, if a certain degree of frequency error is detected, the frequency error is corrected by AFC processing on the transmission side and the reception side, and the residual frequency error still remains, for example, in the case of OFDM, the phase rotation amount is estimated by pilot subcarriers. , It was possible to perform the correction process.

However, as described above, in the case of mobile ^fronthaul , the jitter of delay is required to be 10 ^-9 or less, and in block transmission techniques such as OFDM and SC-FDE, it is necessary to satisfy the jitter of delay required. could not.

  Furthermore, assuming that the e-band band of 70 to 80 GHz is used, the bandwidth in which phase noise is present becomes wider than the subcarrier spacing of OFDM, SC-FDE, etc., and within a time shorter than the OFDM symbol period. Due to phase noise, which is an indefinite fluctuation of complex phase, components that break orthogonality between subcarriers such as OFDM can not be ignored. This is a problem because FFT processing is performed at once, and single carrier transmission signal processing is performed on a single carrier in a time cycle shorter than the block length which is a unit of block transmission such as OFDM or SC-FDE. It was possible to correct the fluctuation component of the phase before the accumulation of the phase noise became large. However, if spatial multiplexing transmission is performed utilizing a MIMO channel as described above, the transmission and reception weights for separating the signals of each stream have different values for each subcarrier, and the frequency dependence of this weight is In order to perform signal processing reflecting the above, it is necessary to once convert the received signal into a signal on the frequency axis by FFT. As described above, although FFT processing is essential for signal separation of space multiplexing, if FFT is performed, phase noise can not be removed, which is a problem. However, before performing the FFT processing and performing space separation on the frequency axis, single stream signals of multiple streams interfere with each other, making it difficult to detect single carrier signals. As described above, if signal separation can be performed on the time axis without performing FFT processing, then phase noise compensation can be performed by signal processing of a single carrier signal, and such a technique is required.

Atsushi Ota, Atsushi Kurosaki, Ikki Maruta, Takuto Arai, Masataka Iizuka, "B-5-175 Proposal of Large-scale Antenna Wireless Entrance System", Proceedings of the IEICE General Conference 2013 (Communications_1), 2013 March 5, p. 585 Takuto Arai, Atsushi Ota, Atsushi Kurosaki, Itsuki Maruta, Masataka Iizuka, "B-5-176 Large-scale Antenna Wireless Entrance System Transmission / reception Weight Calculation Method", Proceedings of the IEICE General Conference 2013 (Communications_1) , March 5, 2013, p. 586 Kazuki Maruta, Atsushi Ota, Atsushi Kurosaki, Takuto Arai, Masataka Iizuka, "B-5-177 Interference Suppression among Users in Large-scale Antenna Wireless Entrance System," Proceedings of the 2013 IEICE General Conference ), March 5, 2013, p. 587 Ryo Kurosaki, Atsushi Ota, Ikki Maruta, Takuto Arai, Masataka Iizuka, "B-5-178 Low Correlation Scheduling Method for Large-scale Antenna Wireless Entrance System", Proceedings of the IEICE General Conference 2013 (Communications_1) , March 5, 2013, p. 588 Akinori Ohara, Atsushi Suyama, Shin Kiyun, Yukihiko Okumura, "Combined processing of fixed beamforming and eigenmode precoding in ultra-fast Massive MIMO transmission using RCS 2013-349 high frequency band", IEICE Technical Report, Wireless Communication System (RCS), Vol. 113, No. 456, February 24, 2014, p. 259-p. 264

  As an environment where it is required to increase the capacity of the transmission path in the mobile network in the future, (Environment 1) In the access system to mobile users, large capacity transmission from the base station apparatus to the mobile station of the mobile user, (Environment 2) When installing a cell radio base station apparatus, for example, on a wall of a building, large capacity transmission at a radio entrance that makes the entrance line to the base station apparatus wireless for construction reasons, (Environment 3) A large capacity transmission of a wireless entrance as an entrance line may be considered, in which data of a large number of mobile users are collected once in a train, and the collected data are collectively transferred to the ground network. In the case of (environment 1), for the purpose of efficient transmission to users present in a relatively narrow area, a service area is formed in a direction looking down from a high place such as a wall or roof of a building, In the station apparatus, the line-of-sight environment is generally secured from the physical characteristics that radio waves from above reach, and the antenna element mounted by the terminal station apparatus also has some directivity considering the arrival of radio waves from above If an antenna with gain is used, the line of sight results in a very dominant environment. Similarly, in the case of (environment 2), since the installation place is fixedly installed, it is expected that the line-of-sight environment is mutually secured and the antenna element also uses a high gain directional antenna (environment 1) The outlook wave has become the dominant situation above. In the case of (environment 3), for example, although a train or the like is a moving object, a directional antenna as high as (environment 2) can not be used, but in consideration of the feature of one-dimensional movement of trains, It is difficult to think of using a directional antenna commensurate with the one-dimensional path, and it is hard to think that an obstacle is intruding on the line, and it is expected that the line-of-sight wave is more dominant than (Environment 1).

  Here, as described above, in general, in spatial multiplexing transmission using a MIMO channel matrix, a multi-path environment in which a large number of reflected waves exist is effective, and so suitable when the direct line-of-sight component is dominant. It is not known that. In order to avoid this problem and perform high-order spatial multiplexing transmission in an environment in which line of sight waves dominate, it is necessary to make the antenna elements of the base station apparatus spatially large (expand the antenna aperture length). It is one choice. That is, it is possible to approach the multipath environment in a pseudo manner by forming a situation where the path lengths on the transmission path between the antenna elements are randomly different.

  On the other hand, as mentioned above, in the future mobile networks are required to increase the capacity of transmission paths, and considering the operation with a wide frequency bandwidth, operation in high frequency bands such as quasi-millimeter wave and millimeter wave Is forced. The point to be noted here is that when a high frequency band is used, each device or between the device and the antenna is connected by a coaxial cable etc., and when transmitting and receiving signals etc. are transmitted on this cable, transmission on the cable The problem is that the loss is very large. Specifically, although a 60 GHz band signal or the like depends on the cable, a loss of about 15 dB is expected per 1 m.

  Even if a large number of antennas are installed to secure the line gain while using the millimeter wave band which is disadvantageous as a gain in line design, even if a large number of cables are connected between spatial spreads of the antennas It does not make sense if a cable loss occurs. Furthermore, when the number of antenna elements is increased in order to secure the line gain, a huge number of cables will be distributed between the spatial spread of the antenna, and the transmission loss between If low-loss cables are used to reduce the cable routing between them, the installation freedom such as bending of the cables is also low, and the restriction on the installation of the base station equipment becomes very large and impractical.

That is, to realize high capacity transmission by high-order space multiplexing using high frequency bands such as millimeter waves even in the future in the environment where the sight wave is dominant in the wireless entrance or access system in the mobile network in the future Needs to solve these problems and build a wireless system.
Also, in addition to the above-discussed discussions specific to wireless links, the above wireless trends can not be ignored in providing mobile front-haul links using optical fiber to base station devices equipped with a large number of antenna elements. It is a situation. As described above, transmission of sampling data in the mobile fronthaul circuit originally requires an optical circuit having a capacity of about 16 times the amount of information of actual user data. In 5G mobile, the goal is to increase the capacity of 10 Gbps as user data by utilizing (quasi-) millimeter wave band etc. However, if it is 16 times that, 160 Gbps of capacity is required for mobile fronthaul circuits, and further As the number of antenna elements increases, it is theoretically possible to increase the capacity to 160 Gbps × the number of antenna elements. Even if the speed of increasing the capacity of the wireless circuit is equal to the speed of technology development of increasing the capacity of the optical circuit, the capacity increasing by the number of antenna elements far exceeds the allowable limit of the optical circuit. Yes, not very cost effective.
In this way, in the future use of an optical fiber-based mobile front hole circuit in a mobile network, these problems are solved for economical laying of a wireless base station device provided with a large number of antenna elements. There is a need to build a mobile network in the future that combines light and wireless.

  In view of the above circumstances, the present invention aims to provide a wireless communication system and a wireless communication method capable of increasing transmission capacity by MIMO in an environment where a line-of-sight environment is dominant.

  One aspect of the present invention relates to the first embodiment, and is a wireless communication system including a first wireless station apparatus and a second wireless station apparatus, wherein the first wireless station apparatus is a first antenna. A plurality of first signal processing units having an element group; and a second signal processing unit that executes signal processing of wireless communication associated with the plurality of first signal processing units, The second wireless station apparatus includes a second antenna element group and a transmitting / receiving unit that performs wireless communication with the first wireless station apparatus via the second antenna element group, The signal processing unit of the present invention is at least one of a transmission weight vector and a reception weight vector for a Multiple Input Multiple Output (MIMO) channel matrix used for wireless communication between the first antenna element group and the second antenna element group. , The MIMO channel matrix Calculated based on at least one of the first right singular vector and the first left singular vector corresponding to the first singular value, or the approximate solution of the first right singular vector and the approximate solution of the first left singular vector And wireless communication in which a plurality of the first signal processing units spatially multiplex and transmit independent signal sequences using at least one of the transmission weight vector and the reception weight vector corresponding to the first signal processing unit. It is a system.

  One embodiment of the present invention relates to the first embodiment, wherein the first signal processing unit is a mixer for performing up conversion and down conversion between a radio frequency signal and a baseband signal or an intermediate frequency signal. The same local signal is input to the mixer included in the same first signal processing unit, and the different local signal is input to the mixer included in the different first signal processing unit It is a wireless communication system.

  One embodiment of the present invention relates to the first embodiment, wherein one single first signal processing unit is provided with a plurality of signal sequences of one system when using a single polarization antenna at the same time and the same frequency. This is a wireless communication system that transmits or receives one system or two systems of signal sequences for each terminal station apparatus when using different types of polarization antennas simultaneously.

  One embodiment of the present invention relates to the first embodiment, and the first signal processing unit suppresses interference components by performing processing for improving directivity gain on a desired signal from a spatially multiplexed signal sequence. The second signal processing unit is a wireless communication system that performs signal processing for extracting the signal sequence that is a desired signal from a signal in which a residual interference component remains without being removed by the first signal processing unit. is there.

  One aspect of the present invention relates to the first embodiment, and is a wireless communication method in a wireless communication system including a first wireless station apparatus and a second wireless station apparatus, wherein the first wireless station apparatus is A plurality of first signal processing units having a first antenna element group, and a second signal processing unit that executes signal processing of wireless communication associated with the plurality of first signal processing units. When the second wireless station apparatus includes a second antenna element group and a transmitting / receiving unit that executes wireless communication with the first wireless station apparatus via the second antenna element group A transmit weight vector and a receive weight vector for a Multiple Input Multiple Output (MIMO) channel matrix used for wireless communication between the first antenna element group and the second antenna element group by the first signal processing unit. At least one of At least one of an approximate solution of at least one of a first right singular vector and a first left singular vector corresponding to a first singular value of the MIMO channel matrix or an approximate solution of the first right singular vector and an approximate solution of the first left singular vector A step of calculating based on one and a plurality of the first signal processing units are independent signal sequences using at least one of the transmission weight vector and the reception weight vector corresponding to the first signal processing unit. And D. space multiplexing transmission.

  The present invention enables a wireless communication system and a wireless communication method to increase transmission capacity by MIMO in an environment where a line-of-sight environment is dominant.

FIG. 1 shows an overview of an example of MIMO transmission. It is a conceptual diagram of the example of eigenmode transmission. FIG. 1 shows an overview of an example of a large scale antenna system. It is a figure showing the example of the impulse response in line-of-sight environment and non-line-of-sight environment. It is a schematic block diagram which shows an example of a structure of the base station apparatus in a multiuser MIMO system. It is a schematic block diagram which shows an example of a structure of the transmission part in the base station apparatus in a multiuser MIMO system. It is a schematic block diagram which shows an example of a structure of the receiving part in the base station apparatus in a multiuser MIMO system. It is a figure which shows the example of the distribution characteristic of the absolute value of the singular value for every channel matrix. It is a figure which shows the example of asymmetry of channel information of an uplink and a downlink. It is a figure which shows the outline | summary of the example of a calibration. It is a figure which shows the example of a calibration of several pairs and several channel matrices. FIG. 1 shows an overview of the functional sharing of mobile fronthaul and mobile backhaul. FIG. 1 is a diagram outlining the functional sharing of mobile fronthaul and mobile backhaul in the prior art when using a multi-element antenna. It is a figure which shows the structural example of a mobile midhole. FIG. 7 is a diagram showing an example of a linear array in which 100 antenna elements of a base station apparatus according to the first embodiment are arranged at equal intervals. FIG. 7 is a diagram showing an example of a linear array in which 100 antenna elements of a base station apparatus according to the first embodiment are divided into four groups of 25 each. It is a schematic block diagram which shows an example of a structure of the base station apparatus in 1st Embodiment. It is a schematic block diagram which shows an example of a structure of the 1st transmission signal processing part of a base station apparatus in 1st Embodiment. It is a schematic block diagram which shows an example of a structure of the 1st received signal processing part of a base station apparatus in 1st Embodiment. It is a schematic block diagram which shows an example of a structure of the terminal | end station apparatus in 1st Embodiment. It is a schematic block diagram which shows the 1st example of a structure of the transmission part of a terminal | end station apparatus in 1st Embodiment. It is a schematic block diagram which shows the 1st example of a structure of the receiving part of a terminal | end station apparatus in 1st Embodiment. It is a schematic block diagram which shows the 2nd example of a structure of the receiving part of a terminal | end station apparatus in 1st Embodiment. FIG. 5 is a schematic block diagram showing an example of the device configuration of a second received signal processing unit of a base station device or a second received signal processing circuit of a terminal station device in the first embodiment. It is a schematic block diagram which shows the 2nd example of a structure of the transmission part of a terminal | end station apparatus in 1st Embodiment. It is a flowchart which shows the outline | summary of the signal processing flow at the time of signal transmission in 1st Embodiment. It is a flowchart which shows the outline | summary of the 1st example of the signal processing flow at the time of signal reception in 1st Embodiment. It is a flowchart which shows the outline | summary of the 2nd example of the signal processing flow at the time of signal reception in 1st Embodiment. It is a figure which shows the 1st example of the case which mounted four parabola antennas in the base station apparatus in 16th Embodiment, and 16 antenna elements in the terminal station apparatus in 2nd Embodiment in the linear array form. It is a figure which shows the example of the path | route difference for every antenna element by the side of the terminal | end station apparatus from the parabola antenna by the side of a base station in 2nd Embodiment. It is a figure which shows the 2nd example of the case which mounted four parabola antennas in the base station apparatus in 2nd Embodiment, and 16 antenna elements in the terminal station apparatus in linear array form. It is a figure which shows the 1st example which implement | achieves transmission / reception of the signal from a supervising base station apparatus to a satellite base station apparatus in 3rd Embodiment by a radio channel. It is a figure which shows the 2nd example which implement | achieves transmission / reception of the signal from a supervising base station apparatus to a satellite base station apparatus in 3rd Embodiment by a radio channel. It is a figure which shows the outline | summary of the signal processing between a supervising base station apparatus and a satellite base station apparatus in 3rd Embodiment, and the signal processing between a satellite base station apparatus and a terminal station apparatus. It is a figure which shows the circuit structure of a satellite base station apparatus in 3rd Embodiment. The outline | summary of a structure of the radio | wireless communications system in 4th Embodiment is shown. It is a figure which shows the outline | summary of the signal processing in the train moving cell which used the virtual transmission path corresponding to the 1st singular value in 4th Embodiment in multiple systems. It is a figure which shows the outline | summary of the offline signal processing performed after acquiring the coordinate information and channel information in 4th Embodiment. It is a figure which shows the outline | summary of another signal processing in the train moving cell which used the virtual transmission line corresponding to the 1st singular value in 4th Embodiment in multiple systems. It is a figure which shows the outline | summary of another offline signal processing performed after acquiring coordinate information and channel information in 4th Embodiment. It is a figure which shows the example of the signal processing in the case of acquiring a transmission-and-reception weight vector from the coordinate information in service in the radio | wireless communications system of 4th Embodiment. It is a block diagram which shows the structural example of the base station apparatus in 4th Embodiment. It is a block diagram which shows the structural example of the terminal | end station apparatus in 4th Embodiment. It is a graph which shows the function F ((alpha)) in 5th Embodiment. It is a figure explaining the removal method of the interference signal in 5th Embodiment. It is a flowchart which shows the process of transmission / reception weight acquisition of time-axis beam forming in the radio | wireless communications system of 5th Embodiment. It is a figure which shows the structural example of the 1st transmission signal processing part with which the base station apparatus in 5th Embodiment is provided. It is a block diagram which shows the structural example of the 1st received signal processing part of the base station apparatus in 5th Embodiment. It is a block diagram which shows the example of a circuit structure of the transmission part with which the terminal | end station apparatus in 5th light embodiment is equipped. It is a block diagram which shows the structural example of the receiving part of the terminal | end station apparatus in 5th Embodiment. It is a figure which shows the outline | summary of the received signal received by several antenna elements. FIG. 7 is a diagram showing an outline of channel estimation with limited subcarriers. It is a flowchart which shows the processing operation of the approximate solution method of channel matrix acquisition in 6th Embodiment. It is a flowchart which shows the processing operation of the approximate solution method of the channel matrix which used multiple antennas. It is a flowchart which shows the processing operation which calculates | requires the approximate solution of a channel matrix from the bidirectional | two-way channel estimation result. It is a figure which shows the result of having evaluated the distribution characteristic by simulation. It is a figure which shows the gain characteristic in the transmission / reception weight vector approximate solution using the antenna away from the gravity center. It is a figure which shows the example of the antenna element used for transmission of the training signal at the time of using a linear array. It is a figure which shows the example of the antenna element used for transmission of a training signal in the case of using the closest packing 2D antenna arrangement. It is a figure which shows the example in the case of adding the antenna element for training signal transmission near the gravity center of a linear array. It is a figure which shows the specific example of linear interpolation of an approximate weight value. It is a flowchart which shows operation | movement of the determination processing of the complex phase offset in linear interpolation of an approximate weight value. It is a flowchart which shows the processing operation of the approximate solution method of the weight vector corresponding to several 1st singular value. It is a figure which shows the apparatus structure which performs simultaneous channel estimation between several small cells which exist in the same area. It is a figure showing the example of the frame composition in a 8th embodiment. It is a flowchart which shows the processing operation for the communication start between the terminal | end station apparatus and the base station apparatus which newly entered in the area. It is a flowchart which shows the transmission / reception signal processing operation of a base station apparatus. It is a flowchart which shows the transmission / reception signal processing operation of a terminal | end station apparatus. It is a figure which shows the frame structure used as a prerequisite. It is a figure which abbreviate | omits a training signal and shows the example which communicates by only a data payload. It is a flow chart which shows processing operation of other 2nd receiving weight matrix calculation. It is a flowchart which shows the other signal processing operation of time-axis beam forming. It is a figure which shows the structure of the base station apparatus at the time of multiuser MIMO application in 10th Embodiment, and a terminal station apparatus. It is a figure showing the circuit configuration of the 1st transmitting signal processing part of the base station device at the time of multi user MIMO application in a 10th embodiment. It is a figure which shows the circuit structure of the 1st received signal processing part of the base station apparatus at the time of multiuser MIMO application in 10th Embodiment. It is a figure which shows the outline | summary of function sharing of the mobile front hall using the virtual transmission line corresponding to the 1st singular value in 11th embodiment. It is a figure which shows the outline | summary (downlink) of the function sharing of the mobile front hall using the virtual transmission line corresponding to the several 1st singular value in 11th Embodiment. It is a figure which shows the outline | summary (uplink) of the function sharing of the mobile front hall using the virtual transmission line corresponding to the several 1st singular value in 11th Embodiment. It is a figure which shows the outline | summary of the correlation detection circuit in 11th Embodiment. FIG. 33 is a diagram showing an overview (uplink) of function sharing of another mobile front hall using virtual transmission paths corresponding to a plurality of first singular values in the eleventh embodiment. It is a figure which shows the outline | summary (downlink) of function assignment at the time of multiuser MIMO application of the mobile front hall using the virtual transmission line corresponding to the several 1st singular value in 11th Embodiment. FIG. 33 is a diagram illustrating an overview (uplink) of functional sharing in multi-user MIMO application of mobile front hall using virtual transmission paths corresponding to a plurality of first singular values in the eleventh embodiment. It is a figure which shows the example of the structure in the case of applying the 1st signal processing part and the 2nd signal processing part to an access system in an embodiment.

Embodiments of the present invention will be described in detail with reference to the drawings.
The symbols in the following embodiments will be described.
N _SDM : Space multiplexing number.
N, M, N ', M': natural numbers.
i, j, m, n: Mainly serial numbers of antenna elements etc. (general integers).
k: Subcarrier number (frequency component number).
N _BS-Ant : The total number of antenna elements of the base station apparatus.
_N'BS-Ant : The number of antenna elements provided in the first transmission signal processing unit or the first reception signal processing unit of the base station apparatus.
N _MT-Ant : Number of antenna elements of the terminal station apparatus.
N _Ant : Number of antenna elements of the base station apparatus or terminal station apparatus.
N _SC : Number of subcarriers.
N _FFT : Number of points of FFT.
L: Distance.
K: Rice coefficient.
λ _k : the wavelength of the k-th subcarrier.
r _ji : Distance between the ith antenna element on the transmitting side and the jth antenna element on the receiving side.
h _ji , h ' _ji : Channel information between the i-th antenna element on the transmitting side and the j-th antenna element on the receiving side (because it has frequency dependency, it is the k-th frequency component if necessary in the description In some cases, this is explicitly stated.
d: distance between antenna elements.
Δd _mn : distance between the n-th antenna element and the m-th antenna element.
ΔL _m : Path length difference of the m-th antenna element based on the first antenna element.
c: speed of light (3 × 10 ⁸ m / s).
f _c : Center frequency of wireless signal [Hz].
f _k : frequency [Hz] of the k th subcarrier of the baseband signal.
t: Time.
W: Bandwidth [Hz].
Δt: Sampling period (Δt = 1 / W).
ψ _j (t), Φ _j (t): Received signal (sampled value) at the j-th antenna element at time t.
φ _j ^(k) (t): A received signal of the kth subcarrier at the jth antenna element at time t (a value focused on a predetermined subcarrier in the sampling value).
η _k : Offset value of the k-th subcarrier in consideration of the complex phase of 2π period when using the least squares method.
u _m : m th left singular vector.
v _m : m th right singular vector.

First Embodiment
[Space multiplexing transmission using virtual transmission paths corresponding to a plurality of first singular values]
(Outline of Basic Principle according to First Embodiment)
As described in FIG. 8 as well, in the environment where the line-of-sight wave is dominant as shown in FIG. 8B, the gap between the absolute values of the first singular value and the second singular value becomes large, and the second singularity When a transmission path corresponding to a singular value greater than the value is used, Hi _{. i.} Communication will be performed with a slight line gain gained using the components of _d . However, for example, in the case of irradiating the area of the limited small cell below from the base station apparatus installed on the wall of the building, the base station apparatus side mounts an antenna with high directivity gain. Furthermore, in a situation where it is expected that a small antenna element capable of obtaining a directional gain by utilizing features such as a short wavelength millimeter wave is expected to be mounted on the terminal station side, the transmitting side and the receiving side The multipath component is expected to be very limited as compared to a microwave band system or the like in which both mount omnidirectional antennas. Therefore, the characteristics of the MIMO transmission in the case where only the line of sight wave is considered are organized.

FIG. 15 is a diagram showing an example of a linear array in which 100 antenna elements of a base station apparatus are arranged at equal intervals. In FIG. 15, reference numeral 40 denotes a wireless communication system, reference numeral 301 denotes a base station apparatus, and reference numeral 302 denotes a terminal station apparatus. In FIG. 15, the 100 antenna elements of the base station apparatus 301 are mounted in a linear array. 100 antenna elements of the base station apparatus 301 is arranged at regular intervals over the length D _1. Further, 16 of the antenna element of the terminal station device 302 is equally spaced linear array shape over the length D _2.

FIG. 16 is a diagram showing an example of a linear array in which 100 antenna elements of the base station apparatus are divided into four groups of 25 each. In FIG. 16, reference numeral 50 is a wireless communication system, reference numeral 303 is a base station apparatus, reference numeral 302 is a terminal station apparatus, reference numerals 304-1 to 304-4 are first signal processing units, and reference numeral 305 is a second signal. A processing unit (strictly, it includes other base station apparatus functions such as an interface circuit, a MAC layer processing circuit, and a communication control circuit). In FIG. 16, 100 antenna elements of the base station apparatus 303 are divided into four groups of 25 antenna elements. 25 antenna elements in the same group, in a very narrow interval as compared with the case of FIG. 15, over a short length D ₃ than the length D _1, are arranged in a linear array form.

In FIG. 16, the 100 antenna elements of the base station apparatus 303 are mounted in a linear array for each group (25 antenna elements). That is, the 100 antenna elements of the base station apparatus 303 are mounted in a linear array for each of the first signal processing units 304. The first signal processing units 304-1 to 304-4 form one directional beam for each group (25 antenna elements) by signal processing. Further, 16 of the antenna element of the terminal station device 302 is equally spaced linear array shape over the length D _2.

  Here, in each of the two cases shown in FIG. 15 and FIG. 16, the transmission characteristics in the case of transmitting four signal systems by space multiplexing (four multiplexing) are compared. The characteristics of the transmission can be grasped by the gain of each transmission line shown in FIG. 2C, which corresponds to the square of the absolute value of the singular value obtained by the singular value decomposition of the channel matrix. In the case shown in FIG. 15, assuming, for example, downlink, if the base station apparatus 301 is the transmitting station 11 and the terminal station apparatus 302 is the receiving station 12, then the size of the channel matrix is 16 × 100. Perform singular value decomposition on this matrix.

On the other hand, in the case shown in FIG. 16, the following procedure is assumed to grasp its characteristics. First, base station apparatus 303 is based on a 16 × 25 channel matrix (in the case of downlink) formed of 25 antenna elements of base station apparatus 303 and 16 antenna elements of terminal station apparatus 302. It is assumed that singular value decomposition is performed to transmit using the first right singular vector. Specifically, the base station device 303 is a portion between each of 25 antenna elements connected to the first signal processing units 304-1 to 304-4 and 16 antenna elements of the terminal station device 302. singular value decomposition of the channel matrix _H 1 to H _4. The partial channel matrices H _{1 to} H ₄ are shown in equation (16).

Each partial channel matrix H _{1 to} H ₄ here is a 16 × 25 matrix. Therefore, v _ij forming each right singular vector is a 25-dimensional vector, and represents the right singular vector corresponding to the j-th singular value in the i-th group of four antenna groups. Similarly, u _ij forming each left singular vector is a 16-dimensional vector, and represents the left singular vector corresponding to the j-th singular value in the i-th group of the four antenna groups, where An overall channel matrix between all the antenna elements of the base station apparatus 303 and the terminal station apparatus 302 is shown in equation (17).

The transmission weight matrix W _Tx here is shown in equation (18).

Here for convenience of notation, is used a Hermitian conjugate representation of the transmission weight matrix W _Tx, the size of the transmit weight matrix W _Tx itself is 100 × 4. As a result, the product of the entire channel matrix and the transmission weight matrix is shown in equation (19).

Here, H _i v _i1 is a 16 × 1 matrix (column vector), which matches λ _i u _{i1 according} to equation (16). As a result, the overall size of the product of the entire channel matrix and the transmission weight matrix is 16 × 4. In general, since the first left singular vectors of the partial channel matrices H _{1 to} H ₄ are not orthogonal to each other, reception weights for signal separation are formed and multiplied at the time of reception. However, when the first left singular vectors of the partial channel matrices H _{1 to} H ₄ are in an environment substantially orthogonal to each other, the matrix represented by the product of the entire channel matrix and the transmission weight matrix is subjected to singular value decomposition The square value of the absolute values of the four singular values approximately matches the channel gain of the transmission line of FIG. 2 (c). In the evaluation here, it is assumed that each element of the channel matrix is expressed by the following equation (20) by the free space propagation model in which only the line of sight wave is considered.

Here, r _ij represents the distance between the i-th antenna on the transmitting side and the j-th antenna on the receiving side, and λ represents the wavelength. In order to grasp the entire feature, the coefficients to be multiplied by the coefficients as a whole are omitted here for the sake of simplification.

Therefore, _L = 100 m in FIG. _{16, D 1 = 12m, D} 2 = 10cm, D 3 = 30cm, for the case of frequency 80 GHz, and compares the characteristics of Figure 15 installed in the antenna aperture length comparable. Here, assuming that the absolute value of the singular value is X as the channel gain, the channel gain is evaluated as 20 Log (X) [dB]. At this time, while the gains of the four lines in FIG. 15 are respectively -56.5 dB, -83.4 dB, -118.3 dB and -157.2 dB, the above-described processing is applied to FIG. Are respectively -62.5 dB. In the case of FIG. 15, only the line with the largest gain corresponding to the first singular value in FIG. 2 (c) has a large value, and the gain of the line corresponding to the remaining singular values is relatively small. Although dependent also on the value of a parameter such as antenna gain, it is in a situation where only the line corresponding to the first singular value can be used. On the other hand, it can be seen that in the case of FIG. 16, four transmission paths can be used almost equally. Here, although the gain difference with respect to the first singular value in FIG. 15 and the gain difference with respect to the singular value in FIG. 16 is 6 dB, the number of antenna elements used for directional beam forming is 1/100 to 25 in FIG. It corresponds to 4 and its corresponding 10 Log (1/4) =-6 dB. In other words, although the efficiency is reduced to 1⁄4 by dividing the antenna element group into four, it is better to multiplex four signal sequences to the channel capacity due to the Shannon limit than to improve the SNR by 6 dB. It is overwhelmingly efficient in terms of growth.

  If the received signal power of the reflected wave component is strong enough that the line gain of the line corresponding to the singular value after the second singular value can be effectively used by setting the values of parameters such as transmission power and antenna gain Generally, if the number of antenna elements is increased to compensate for the decrease in line gain due to the use of high frequency bands such as millimeter waves, the line gain of the line corresponding to the singular value after the second singular value is sufficient. In general, it is difficult to think about the situation where there is a problem, and the transmission capacity is increased in the case of FIG. 16 in which transmission for four lines is performed in parallel compared to the case of FIG. It can be viewed as being suitable for It is effective to group antennas in this manner and efficiently use virtual transmission paths corresponding to the first singular value in each group.

(Singular value decomposition and calibration)
Here, the case where the amplification factors of the low noise amplifier and the high power amplifier in the above-mentioned equation (14) do not differ for each antenna element (or can be approximated as a constant value) will be considered. In order to simplify the equation, let H _{UL be} the uplink channel matrix, H _{DL be} the downlink channel matrix, C _{BS be} the calibration matrix at the base station side and C _{MT be} the calibration matrix at the terminal station side. Further, singular value decomposition results of the uplink channel matrix H _UL are shown in equations (21) and (22).

Here, the absolute values of the diagonal terms in equation (21) are all equal in each matrix. In this case, both expressions of equation (21) become unitary matrices, and this term behaves as rotation of coordinate axes to the last.
Here, the downlink channel matrix H _DL is represented by using the calibration matrices C _BS and C _MT and the uplink channel matrix H _UL , and equation (22) is substituted to obtain equation (23).

Here, the equations on both sides of D _{UL on} the right side are shown in equations (24) and (25).

  If the absolute values of each component of the calibration matrix are approximately equal, calibration is performed on the uplink channel matrix and then singular value decomposition is performed to obtain transmission and reception weights, and the uplink channel matrix is set to the singular value The components of the first right singular vector and the first left singular vector obtained by decomposing are multiplied by the calibration coefficient and matched with the result of performing the calibration process, as in the equation (24) and the equation You can see from (25).

  That is, if singular value decomposition is performed on the uplink channel matrix, calibration processing is performed on each component of the obtained reception weight vector (or matrix) without performing singular value decomposition on the downlink. For example, it can be understood that it is possible to obtain a desired transmission weight vector (or matrix). For this reason, singular value decomposition needs to be performed only once.

(About the circuit configuration of the base station apparatus in the present invention)
Hereinafter, the circuit configuration of the base station apparatus 303 in the first embodiment of the present invention will be described according to the drawings.
FIG. 17 is a schematic block diagram showing an example of the configuration of the base station apparatus 70 in the MIMO system of the present invention. Although FIG. 16 illustrates the case of point-to-point type one-to-one communication with one base station apparatus 303 and one terminal station apparatus 302, it goes without saying that there are a plurality of terminal station apparatuses 302. It does not matter. Transmission and reception of the signals in FIG. 16 communicate with only one terminal station apparatus 302 at the same time in the focused subcarrier, and is in the form of single-user MIMO transmission, and the communication target is one terminal station apparatus 302 by scheduling. Is selected. If OFDMA is used in access control, different terminal station apparatuses 302 may be allocated to each subcarrier, but if attention is paid to each subcarrier, the allocation is limited to one terminal station apparatus 302. Further, when the terminal is fixed in the point-to-point type communication, the process of selecting the terminal station apparatus 302 of the communication counterpart in the scheduling becomes unnecessary. In the first embodiment of the present invention, point-to-multipoint type one-to-multipoint communication multi-user MIMO transmission can also be used in variations of the form, but in the following description, such variations are not relevant. , A general single-user MIMO transmission will be described. Further, in the following description, although a wideband system is assumed to simplify the explanation, signal processing for each subcarrier on the frequency axis such as OFDM or SC-FDE will be described, but other systems will be described. It can also be extended (for example, a narrow band single carrier system).

  As shown in FIG. 17, the base station device 70 corresponding to the base station device 303 includes the first transmission signal processing units 181-1 to 181-4, the second transmission signal processing unit 71, and the first reception. The signal processing units 185-1 to 185-4, the second received signal processing unit 75, the interface circuit 77, the MAC (Medium Access Control) layer processing circuit 78, and the communication control circuit 120 are provided. The MAC layer processing circuit 78 has a scheduling processing circuit 781.

  The base station device 70 inputs and outputs data with an external device or a network via the interface circuit 77. The interface circuit 77 detects data to be transferred on the wireless channel among the input data, and outputs the detected data to the MAC layer processing circuit 78. The MAC layer processing circuit 78 performs processing related to the MAC layer in accordance with an instruction of the communication control circuit 120 that performs management control of the entire operation of the base station device 70. Here, the processing related to the MAC layer includes conversion of data input / output by the interface circuit 77, data transmitted / received on the wireless channel, addition of header information of the MAC layer, and the like. In this process, the scheduling process circuit 781 performs various scheduling processes of the terminal station apparatus 302 that performs space multiplexing. The scheduling processing circuit 781 outputs the scheduling result to the communication control circuit 120. In MIMO transmission, since signals of a plurality of signal sequences are spatially multiplexed and transmitted at one time, signal sequences of a plurality of systems are output from the MAC layer processing circuit 78 to the second transmission signal processing unit 71.

  Although the operation of the second transmission signal processing unit 71 will be described later, basically, predetermined modulation processing is performed on a plurality of series of signals from the MAC layer processing circuit 78, and if necessary, some precoding processing (on the transmission side Equalization processing, signal separation processing, and the like), and the like, and output to the first transmission signal processing units 181-1 to 181-4. At this time, regardless of using OFDM or SC-FDE, when performing signal processing on the frequency axis by the first transmission signal processing units 181-1 to 181-4, the second transmission signal processing unit A signal on the frequency axis is generated in 71 and output to the first transmission signal processing units 181-1 to 181-4. When signal processing on the time axis is performed by the first transmission signal processing unit as in the fifth embodiment described later, a signal of the time axis may be output. Each of the first transmission signal processing units 181-1 to 181-4 is connected with a plurality of antennas as shown in FIG. 16, and outputs transmission signals to the respective antennas. At this time, a signal obtained by multiplying the transmission weight vector corresponding to the first singular value among the antenna element groups grouped for each of the first transmission signal processing units 181-1 to 181-4 (strictly, For example, in the case of OFDM, a signal obtained by combining signals of respective subcarriers is converted into a time axis component, and a signal obtained by up-converting the signal into a radio frequency is transmitted from each antenna.

  Next, at the time of reception, the signals received by the plurality of antennas connected to the first received signal processing units 185-1 to 185-4 (precisely, the received radio frequency signals are downconverted to baseband signals For example, in the case of OFDM, this time-axis signal is converted to a frequency-axis signal by FFT) and multiplied by a predetermined reception weight vector to convert one subcarrier into a complex scalar quantity for each subcarrier, and Output to the reception signal processing unit 75 of FIG. The second received signal processing unit 75 refers to four received signal sequences in this example, and first performs channel estimation for each subcarrier using a known training signal given at the beginning of the received signal, A × 4 MIMO channel matrix is obtained for each subcarrier. The reception weight matrix is calculated based on this channel matrix, and the detection processing of the signal transmitted based on the acquired reception weight matrix is performed. For example, an inverse matrix of ZF (Zero Forcing) type is used, or a reception weight matrix of Maximum Mean Square Error (MMSE) type is used. If there is enough time for signal processing, MLD (Maximum Likelihood Detection) or simplified MLD (QR-MLD) using QR decomposition may be used. The signal detected by this reception signal processing is output to the MAC layer processing circuit 78, is subjected to processing of a predetermined MAC layer, and is output to the network side through the interface circuit 77.

FIG. 18 is a schematic block diagram showing an example of a configuration of the first transmission signal processing unit 181 in the base station device 70 of the first embodiment of the present invention. As shown in FIG. 18, the first transmission signal processing unit 181 includes a first transmission signal processing circuit 111 and an IFFT (Inverse Fast Fourier Transform) & GI (Guard Interval: guard interval) application circuit 813. -1 to 813-N ' _BS-Ant , D / A (digital / analog) converter 814-1 to 814- (N' _BS-Ant ), local oscillator 815, and mixers 816-1 to 816- N'BS _-Ant ), filters 817-1 to 817- (N'BS _-Ant ), high power amplifiers (HPA) 818-1 to 818- (N'BS _-Ant ), antenna element 819-1 Through 819- (N ' _BS -Ant) and the first transmission weight processing unit 130. _N'BS-Ant is a value obtained by dividing the total number of antenna elements of the base station apparatus 70 by the number of space multiplexing (= _NBS-Ant / _NSDM ). In short, N ′ _{BS -Ant} represents the number of antenna elements included in one first transmission signal processing unit 181. The first transmission signal processing circuit 111 is connected to the second transmission signal processing unit 71 shown in FIG. Further, the first transmission signal processing circuit 111 and the first transmission weight processing unit 130 are connected to the communication control circuit 120 via the second transmission signal processing unit 71 shown in FIG. In the example of FIG. 17, the four first transmission signal processing units (181-1 to 181-4) are connected to the base station apparatus 70. The description focusing on one of them will be described below.

The first transmission weight processing unit 130 includes a first channel information acquisition circuit 131, a first channel information storage circuit 132, and a first transmission weight calculation circuit 133. Here, IFFT & GI imparting circuit 813-1~813- (N _'BS-Ant) from the antenna element 819-1~819- (N' _BS-Ant) of the circuit to the subscript (N _'BS-Ant) is Represents the number of antenna elements provided in the first transmission signal processing unit 181 of the base station device 70.

In the present invention according to the first embodiment, since signals of a plurality of systems N _SDM (= 4) are spatially multiplexed and transmitted to one terminal station apparatus 302, a plurality of signal sequences are transmitted from the MAC layer processing circuit 78. A transmission signal is input to each of the first transmission signal processing units 181-1 to 181-4 via the second transmission signal processing unit 71. When the data to be transmitted to the destination terminal station apparatus 302 is input from the MAC layer processing circuit 78, the second transmission signal processing unit 71 generates a radio packet to be transmitted by a radio channel and performs modulation processing. Here, in the case of using, for example, the OFDM modulation scheme, the signal of each signal sequence is modulated for each subcarrier. The signal subjected to the modulation processing is subjected to precoding processing as needed. The precoding processing here may be multiplication of a transmission weight matrix for suppressing leakage of signals between virtual transmission paths corresponding to a plurality of first singular values. Alternatively, such precoding processing may not be performed.
The signal of the N _SDM system generated in this manner is input to each of the first transmission signal processing units 181-1 to 181-4. In each of the first transmission signal processing units 181-1 to 181-4, the input digital baseband signal input #i (i is 1 to N _SDM ) is input to the first transmission signal processing circuit 111-i. Ru. The first transmission signal processing circuit 111 basically performs transmission weight multiplication and signal processing of the remaining physical layer. For example, in the case of using the OFDM modulation scheme, the input modulation-processed baseband signal is multiplied by the transmission weight for each subcarrier. The signal multiplied by the transmission weight corresponding to each antenna element 819-1 to 819- (N ' _BS-Ant ) is subjected to the remaining signal processing as necessary, and IFFT & GI adding circuits 813-1 to 813- ( The signal on the frequency axis is converted to the signal on the time axis at N ′ _{BS -Ant} The converted signal is further subjected to processing such as insertion of guard intervals and waveform shaping between OFDM symbols (between SC-FDE (Single-Carrier Frequency Domain Equalization) block transmission blocks), and antenna elements 819- The digital sampling data is converted to a baseband analog signal by the D / A converters 814-1 through 814- (N'BS _-Ant ) every 1 to 819- (N'BS _-Ant ). Further, each analog signal is multiplied by the local oscillation signal input from the local oscillator 815 by the mixers 816-1 to 816-(N ′ _BS -Ant), and up-converted to a radio frequency signal. Here, since the up-converted signal includes the signal outside the band of the channel to be transmitted, the out _- of-band signal is removed by the filters 817-1 to 817-(N'BS _-Ant ) and transmitted. To generate an electrical signal. The generated signal is 'is amplified by _(BS-Ant, antenna elements 819-1~819- (N high-power amplifier 818-1~818- N)' is transmitted from the _BS-Ant).

  The transmission weight vector to be multiplied by the first transmission signal processing circuit 111 is acquired from the first transmission weight calculation circuit 133 provided in the first transmission weight processing unit 130 at the time of signal transmission processing. In the first transmission weight processing unit 130, in the first channel information acquisition circuit 131, the channel information acquired by the first reception signal processing units 185-1 to 185-4 is transmitted via the communication control circuit 120 (strictly Is separately acquired by the second reception signal processing unit 75 and the second transmission signal processing unit 71), and is stored in the first channel information storage circuit 132 while being sequentially updated. At the time of signal transmission, the first transmission weight calculation circuit 133 reads channel information corresponding to the destination terminal station apparatus from the first channel information storage circuit 132 according to the instruction from the communication control circuit 120, and the read channel information The transmission weight vector is calculated based on

  There are several variations in the channel estimation method and transmission / reception weight calculation method in the case of utilizing a virtual channel corresponding to the first singular value, and the details of a method for efficiently acquiring this will be described later. As an example, the transmission weight vector may use, for example, a first right singular vector obtained by performing singular value decomposition on the acquired channel matrix.

Alternatively, focusing on one of the central portions of the antennas on the terminal station apparatus 302 side, the one antenna and one of the received signal processing units (one of 185-1 to 185-4) of the base station apparatus 70 Each component of the transmission weight vector may be determined by equation (7) on the basis of the channel vector between the antenna elements 819-1 to 819-4 and N ′ _BS-Ant provided (weight of maximum ratio combining) , Or all the absolute values may be made constant with respect to the value given by equation (7) (equal gain combining weight).

  Or, when the transmitting side multiplies a plurality of antenna elements by a predetermined transmission weight vector and transmits a signal, although one signal is actually transmitted from a plurality of transmitting antennas, it is effectively one. Since it is equivalent to the one transmitted from the virtual antenna element of the above, a training signal is transmitted from this one virtual antenna element by multiplying a predetermined transmission weight vector, and this one virtual antenna element and each A vector of channel information to and from the receiving antenna may be obtained, and each component of the receiving weight vector may be obtained by equation (7) based on this vector (weight of maximum ratio combining) or by equation (7) It is also possible to make all absolute values constant with respect to a given value (weight of equal gain combination).

  If the channel vector at the time of reception is known, it is possible to obtain uplink channel information by the implicit feedback method, and based on the uplink channel vector determined in this way, the transmission weight vector is obtained. You may calculate similarly. Similarly, based on the uplink reception weight vector, the transmission weight vector may be calculated by an implicit feedback method in which calibration processing is directly performed on the uplink reception weight vector.

  The first transmission weight calculation circuit 133 outputs the transmission weight vector calculated in this manner to the first transmission signal processing circuit 111. Further, the communication control circuit 120 manages control related to the entire communication, such as management of a terminal station apparatus as a destination and overall timing control. The communication control circuit 120 outputs information indicating a terminal station apparatus or the like as a destination to the first transmission weight processing unit 130 that performs signal processing related to the calculation of the transmission weight described above.

  In the above description, in the first channel information acquisition circuit 131, the channel information acquired by the first received signal processing units 185-1 to 185-4 is acquired via the communication control circuit 120, and this channel information is acquired. Although it has been described as sequentially updating, it is not necessary to update the channel information frequently if the line-of-sight environment in which the channel time variation can be ignored is high. The first channel information acquisition circuit 131 acquires channel information in advance, for example, before starting service operation, and further stores a transmission weight vector calculated from the value of the channel information (not shown in the figure). However, in this case, the "transmission weight storage circuit" may be implemented and recorded (implemented) and used repeatedly. Also, as an intermediate of these, while the transmission weight vector is basically read out from the first transmission weight storage circuit, the transmission weight vector is updated based on the sequentially acquired channel information, and the updated channel information is updated. It is also possible to have the transmission weight vector updated based on predetermined time intervals.

FIG. 19 is a schematic block diagram showing an example of a configuration of the first received signal processing unit 185 in the base station device 70 in the first embodiment of the present invention. As shown in FIG. 19, the first received signal processing section 185, antenna element 851-1~851- (N 'and _BS-Ant), a low noise amplifier (LNA) 852-1~852- (N' _{BS- Ant} ), local oscillator 853, mixers 854-1 to 854- (N'BS _-Ant ), filters 855-1 to 855- (N'BS _-Ant ), A / D (analog / digital) conversion Devices 856-1 to 856- (N'BS _-Ant ), an FFT (Fast Fourier Transform) circuit 857-1 to 857- (N'BS _-Ant ), and a first reception weight processing unit 160. And a first received signal processing circuit 158. The first received signal processing circuits 158-1 to 158-N _SDM (= 4) are connected to the second received signal processing unit 75 shown in FIG. In addition, the first received signal processing circuits 158-1 to 158-N _SDM (= 4) and the first received weight processing unit 160 are connected via the second received signal processing unit 75 shown in FIG. The communication control circuit 120 is connected. The first reception weight processing unit 160 includes a first channel information estimation circuit 161 and a first reception weight calculation circuit 162. As in the description of the first transmission signal processing unit 181, four first reception signal processing units (185-1 to 185-4) are connected to the base station device 70 in the example of FIG. However, an explanation focusing on one is given below.

First, the signals received by the antenna elements 851-1 to 851- (N'BS _-Ant ) are amplified by the low noise amplifiers 852-1 to 852- (N'BS _-Ant ). The amplified signal and the local oscillation signal output from the local oscillator 853 are multiplied by the mixers 854-1 to 854- (N ' _BS -Ant), and the amplified signal is converted from a radio frequency signal to a baseband signal. It is downconverted. Since the downconverted signal includes a signal outside the frequency band to be received, the out _- of-band component is removed by the filters 855-1 to 855- (N ' _BS -Ant). The signal from which the out-of-band component has been removed is converted to a digital baseband signal by A / D converters 856-1 to 856- (N'BS _-Ant ). For example, in the case of OFDM, digital baseband signals are all input to FFT circuits 857-1 to 857- (N'BS _-Ant ), and predetermined symbol timings determined by a circuit for timing detection whose description is omitted here. The signal on the time axis is converted to the signal on the frequency axis (separated to the signal of each subcarrier). The signal separated into each subcarrier is input to the first received signal processing circuit 158 and also input to the first channel information estimation circuit 161. Although the circuit for symbol timing detection of OFDM is omitted in FIG. 19, the symbol timing can be grasped by some existing method.

In the first channel information estimation circuit 161, an antenna element of each terminal station apparatus 302 based on a known signal for channel estimation (such as a preamble signal added to the beginning of a radio packet) separated into each subcarrier; Channel information between each antenna element 851 of base station apparatus 70 is estimated for each subcarrier, and the estimation result is output to first reception weight calculation circuit 162. The first reception weight calculation circuit 162 calculates, for each subcarrier, a reception weight vector to be multiplied based on the input channel information. Under the present circumstances, the receiving weight which synthesize | combines the signal received by each antenna element 851-1 to 851- (N'BS _-Ant ) is 1st received signal processing part 185-1-185-N _SDM (= 4) The first reception signal processing units 185-1 to 185-N _SDM are individually calculated.

In the first reception signal processing circuit 158, the signal for each subcarrier input from the FFT circuits 857-1 to 857- (N'BS _-Ant ) (precisely, signals from a plurality of antenna elements are elements) A reception signal vector) is multiplied by the reception weight (precisely, a reception weight vector having reception weights corresponding to a plurality of antenna elements) input from the first reception weight calculation circuit 162, and the multiplication result The signals received by the respective antenna elements 851-1 to 851- (N'BS _-Ant ) are added and synthesized for each subcarrier. The first reception signal processing circuit 158 outputs the signal obtained by the addition and synthesis to the second reception signal processing unit 75. Here, addition and combining means addition after multiplication of each component of the vector in the vector product for each subcarrier, and the multiplication of the received signal and the reception weight and the entire addition and combination thereof are mathematically vector Corresponds to product processing.

  The reception weight vector to be multiplied by the first reception signal processing circuit 158 is acquired from the first reception weight calculation circuit 162 provided in the first reception weight processing unit 160 at the time of signal reception processing. The first reception weight processing unit 160 calculates the reception weight vector in the first reception weight calculation circuit 162 using the channel information acquired in the first channel information estimation circuit 161. For example, if the terminal station device 302 is transmitting signals to the plurality of antenna elements 851 of the first received signal processing unit 185 in a virtual transmission path corresponding to the first singular value, the terminal station device 302 Is like transmitting toward each first received signal processing unit 185 using one virtual antenna element using the transmission weight for the virtual transmission line corresponding to the first singular value. The channel information on the reception side between the one antenna and the plurality of antenna elements 851 of the first reception signal processing unit 185 is determined, and each component of the reception weight vector for this channel vector is expressed by Expression (7). It may be determined (weight of maximum ratio combination) or all absolute values may be fixed with respect to the value given by equation (7) (weight of equal gain combination). Alternatively, the channel matrix between the antenna element of the terminal station apparatus 302 and the plurality of antenna elements 851 of the first received signal processing unit 185 of the base station apparatus 70 can be obtained by singular value decomposition One left singular vector may be used as a reception weight vector.

  The first reception weight calculation circuit 162 outputs the reception weight vector calculated in this manner to the first reception signal processing circuit 158. Further, the communication control circuit 120 manages control related to the entire communication, such as management of the transmission source station and overall timing control.

  In the above description, although it is described that the reception weight is sequentially calculated using the channel information acquired in the first channel information estimation circuit 161, if the line-of-sight environment where channel time fluctuation can be ignored is frequent, it is frequently Channel information need not be updated. For example, the first channel information estimation circuit 161 acquires channel information in advance before starting service operation, and stores the reception weight vector calculated from the value of the channel information (not shown in the figure). However, in this case, the “first reception weight storage circuit” may be implemented by a configuration that is mounted and recorded at the subsequent stage of the first reception weight calculation circuit 162), and it may be used repeatedly. In this case, the communication control circuit 120 outputs information indicating the terminal station apparatus or the like of the transmission source to the first reception weight storage circuit that outputs the reception weight. Also, as an intermediate of these, while the reception weight vector is basically read out from the first reception weight storage circuit, the reception weight vector is updated based on the sequentially acquired channel information, and the updated channel information is updated. It is also possible to have a configuration in which the reception weight vector is updated at a predetermined time interval based on that.

  In addition, in the second received signal processing unit 75 in FIG. 17, received signals obtained by multiplying the received weight vectors from the first received signal processing units 185-1 to 185-4 described above are aggregated into one system. Although they are input, they are substantially equivalent to the reception signal of the 4 × 4 MIMO channel, and it is possible to process the signal sequence spatially multiplexed by the reception signal detection processing of the prior art. Specifically, 4 × 4 in the case where four sets of signal sequences transmitted on the transmitting side are received by four sets of virtual antennas configured with a plurality of antennas on the receiving side (base station apparatus 70 side) For the MIMO channel of (4), a 4 × 4 channel matrix is acquired for each subcarrier with a known training signal for channel estimation attached to the head of the received signal. The reception weight matrix is calculated based on this channel matrix, and the detection processing of the signal transmitted based on the acquired reception weight matrix is performed. For example, an inverse matrix of ZF (Zero Forcing) type is used, or a reception weight matrix of Maximum Mean Square Error (MMSE) type is used. If there is enough time for signal processing, MLD (Maximum Likelihood Detection) or simplified MLD (QR-MLD) using QR decomposition may be used. Also, in the signal detection processing here, for example, soft decision of the received signal is performed once, deinterleave processing is performed as necessary, and then error correction processing is performed to perform final signal detection. good. The signal detected by this reception signal processing is output to the MAC layer processing circuit 78, is subjected to processing of a predetermined MAC layer, and is output to the network side through the interface circuit 77.

  In addition, the MAC layer processing circuit 78 performs processing related to the MAC layer (for example, conversion of data input to and output from the interface circuit 77 and data to be transmitted and received on a wireless circuit, that is, wireless packet, termination of header information of the MAC layer Etc.). Although the scheduling processing circuit 781 is substantially unnecessary in the case of point-to-point communication, in the case of performing point-to-multipoint communication with a plurality of terminal stations 302, It performs various types of scheduling processing to select the terminal station apparatus 302 that performs communication, and outputs the scheduling result to the communication control circuit 120. The received data processed by the MAC layer processing circuit 78 is output to an external device or network via the interface circuit 77.

  Further, the communication control circuit 120 manages control related to the entire communication, such as management of the terminal station apparatus 302 of the transmission source and overall timing control. Further, information indicating the terminal station apparatus or the like of the transmission source is input from the communication control circuit 120 to the first reception weight processing unit 160 that performs signal processing related to the calculation of the reception weight described above.

As for signal reception, as in the case of transmission, in the wideband system using the OFDM modulation method or SC-FDE method, the multiplication of the above-mentioned reception weight is performed for each subcarrier. That is, the FFT circuits 857-1 to 857- (N'BS _-Ant ) perform FFT on the signals output from the A / D converters 856-1 to 856- (N'BS _-Ant ) to each subcarrier. Signal processing in the first channel information estimation circuit 161 and reception signal processing in the first reception signal processing circuit 158 are performed for each of the separated and separated subcarriers.

The above is the description of the base station device 70 in the first embodiment of the present invention. Here, it is important that the local oscillator 815 in the first transmission signal processing unit 181 is common to the mixers 816-1 to 816-(N ′ _BS -Ant) in each antenna system in the same first transmission signal processing unit 181. On the other hand, the local oscillator 815 is not shared between the first transmission signal processing units 181 which are different. Similarly, the local oscillator 853 in the first received signal processing unit 185 is shared by the mixers 854-1 to 854- (N ' _BS -Ant) in each antenna system in the same first received signal processing unit 185. On the other hand, it is also important that the local oscillator 853 is not shared between the different first received signal processing units 185. As shown in FIG. 16, generally, the first transmission signal processors 181-1 to 181-4 (the first signal processors 304-1 to 304-4 in FIG. 16) are physically separated by several meters. In a high frequency band such as a millimeter wave, it is assumed that the loss is about 10 dB per meter or more when routed through a cable. Since there are a large number of antenna elements connected to the first transmission signal processing unit 181 and the first reception signal processing unit 185, it is necessary to branch and input signals to the mixers 816 and 854 in a plurality of systems, Considering the drop in the level due to this branching, the loss of the cable length of several meters can not be ignored, so it is closed in the individual first transmission signal processing unit 181 and the individual first reception signal processing unit 185 to be local. A configuration in which the oscillators 815 and 853 are shared and not shared between different first transmission signal processing units 181 and first reception signal processing units 185 is effective.

  Here, although each antenna adjusts the phase of the transmission / reception signal for directivity control, the same first transmission signal processing units 181-1 to 181-4 (the first signal processing unit in FIG. 16) In 304-1 to 304-4), it is easy to make the phase relationship of the signal inputted from each of the local oscillator 815 or the local oscillator 853 constant at all times. It is possible to determine what phase relationship should be multiplied by the transmission / reception weight so that the part does not depend. However, when the local oscillator 815 uses a plurality of asynchronous ones in the first transmission signal processing unit 181 (or the local oscillator 853 in the first reception signal processing unit 185), at least the first transmission signal When multiplying the transmission weight in the processing unit 181, it is necessary to adjust in consideration of the complex phase relationship between the plurality of local oscillators 815 (or 853), and if this adjustment is not made, directivity control functions effectively I will not do. Care should be taken in this regard in the design of the device. In the first embodiment of the present invention, the same first transmission signal processing unit 181 shares the local oscillator 815 for the reason described above, and the same first reception signal processing unit 185 uses the local oscillator 853 for the reason described above. There is no need to implement complete signal separation at the transmit side as in multi-user MIMO, as signal separation among four spatially multiplexed signal sequences can be carried out at the receive side, although commonality is common. .

  Although a plurality of signal sequences can basically be separated by signal processing on the reception side in this way, for example, the phases among the first transmission signal processing units 181-1 to 181-4 by feedback of a channel or the like. If the relationship is known, it is also possible to multiply (ie, transmit precoding) the transmit weight matrix of signal separation beforehand at the transmit side. In this case, the second transmission signal processing unit 71 of the base station device 70 multiplies this transmission weight matrix. The transmission weight matrix is calculated using calibration processing based on uplink channel information on the four signal sequences after multiplication by the reception weight vector acquired by the second reception signal processing unit 75. It does not matter. However, as described above, the terminal station apparatus 302 on the receiving side can acquire the channel matrix by the training signal given to the transmission signal even in the downlink, so if the signal processing on the receiving side is used, The multiplication of the transmission weight matrix in the transmission signal processing unit 71 is not necessary.

(About the circuit configuration of the terminal station device 60 corresponding to the terminal station device 302 in the present invention)
FIG. 20 is a schematic block diagram showing an example of a configuration of the terminal station device 60 corresponding to the terminal station device 302 in the first embodiment of the present invention. As shown in FIG. 20, the terminal station device 60 includes a transmitting unit 61, a receiving unit 65, an interface circuit 67, a MAC (Medium Access Control) layer processing circuit 68, and a communication control circuit 121.

  The terminal station device 60 inputs / outputs data to / from an external device or network via the interface circuit 67. The interface circuit 67 detects data to be transferred on the wireless channel among the input data, and outputs the detected data to the MAC layer processing circuit 68. The MAC layer processing circuit 68 performs processing related to the MAC layer in accordance with an instruction of the communication control circuit 121 which performs management control of the entire operation of the terminal station device 60. Here, the processing related to the MAC layer includes conversion of data input / output by the interface circuit 67, data transmitted / received on the wireless circuit, that is, wireless packet, addition of header information of the MAC layer, and the like. In MIMO transmission, since signals are space-multiplexed and transmitted to one terminal station apparatus 60, a plurality of signal sequences are output from the MAC layer processing circuit 68 to the transmission unit 61.

FIG. 21 is a schematic block diagram showing an example of the configuration of the transmission unit 61 in the terminal station device 60 in the first embodiment of the present invention. As illustrated in FIG. 21, the transmission unit 61 includes transmission signal processing circuits 811-1 to 811-N _SDM (N _SDM is an integer of 2 or more), and addition synthesis circuits 812-1 to 812-N _MT-Ant (N _MT-Ant is an integer of 2 or more), IFFT (Inverse Fast Fourier Transform: Inverse Fast Fourier Transform) & GI (Guard Interval: Guard Interval) applying circuit 813-1 to 813-N _MT-Ant , D / A (digital / Analog) converter 814-1 to 814-N _MT-Ant , local oscillator 815, mixers 816-1 to 816-N _MT-Ant , filters 817-1 to 817-N _MT-Ant , high power amplifier (HPA) and _{818-1~818-N MT-Ant,} an antenna element _{819-1~819-N MT-Ant,} a first transmission weight processing unit 140 It is provided. The transmission signal processing circuits 811-1 to 811-N _SDM and the first transmission weight processing unit 140 are connected to the communication control circuit 121 shown in FIG.

The first transmission weight processing unit 140 includes a channel information acquisition circuit 141, a channel information storage circuit 142, and a first transmission weight calculation circuit 143. Here, _{N SDM} subscripts of the transmission signal processing circuit _{811-1~811-N SDM} in FIG 21 represents a multiplex number for performing spatial multiplexing simultaneously. In addition, the N _MT-Ant of the suffix of the circuit from the addition synthesis circuit 812-1 to 812-N _MT-Ant to the antenna element 810-1 to 819-N _MT-Ant is the number of antenna elements included in the terminal station apparatus 60. Represents N _{MT -Ant} is, for example, 16.

In the first embodiment, since one terminal station apparatus 60 spatially multiplexes and transmits signals to the base station apparatus 70, signal sequences of a plurality of systems are input from the MAC layer processing circuit 68 to the transmitting unit 61 and are input. The signal sequences of the plurality of systems are input to the transmission signal processing circuits 811-1 to 811-N _SDM . The transmission signal processing circuits 811-1 to 811-N _SDM transmit data (wireless input) from the MAC layer processing circuit 68 to data (data inputs # 1 to #N _SDM ) to be transmitted to the destination base station apparatus 70 When a packet is input, modulation processing is performed on this. Here, in the case of using, for example, the OFDM modulation scheme, the signal of each signal sequence is modulated for each subcarrier. Furthermore, the base band signal subjected to the modulation processing is multiplied by the transmission weight for each subcarrier. The signal multiplied by the transmission weight corresponding to each antenna element 819-1 to 819-N _MT-Ant is subjected to the remaining signal processing as necessary, and each transmission signal processing as sampling data of the transmission signal in the baseband The circuits 811-1 to 811 -N _SDM input to the adding and combining circuits 812-1 to 812 -N _MT -Ant.

The signals input to the adding and combining circuits 812-1 to 812-N _MT-Ant are combined for each subcarrier. The synthesized signal is converted from the signal on the frequency axis to the signal on the time axis by the IFFT & GI application circuits 813-1 to 813-N _MT-Ant , and further insertion of guard intervals or between OFDM symbols (SC-FDE If it is Single-Carrier Frequency Domain Equalization), processing such as waveform shaping between blocks of block transmission is performed, and D / A converter 814-1 is carried out for each antenna element 819-1 to 819-N _MT-Ant. At 814-N _MT-Ant , digital sampling data is converted to a baseband analog signal. Furthermore, each analog signal is multiplied by the local oscillation signal input from the local oscillator 815 by the mixers 816-1 to 816 -N _MT-Ant , and is upconverted to a radio frequency signal. Here, since the upconverted signal includes the signal in the region outside the band of the channel to be transmitted, the filter 817-1 to 817-N _MT-Ant should remove the signal outside the band and transmit it. Generate a signal. The generated signal is amplified by the high-power amplifier _{818-1~818-N MT-Ant,} and transmitted from the antenna elements _{819-1~819-N MT-Ant.}

In FIG. 21, after the addition synthesis of the signals of the respective subcarriers is performed by the addition synthesis circuit 812-1 to 812-N _MT-Ant , processes such as IFFT processing, guard interval insertion, and waveform shaping are performed. , Perform these processes in the transmission signal processing circuit 811-1 to 811-N _{SDM, and} combine the IFFT-processed sampling signals on the time axis using the addition / synthesis circuit 812-1 to 812-N _MT-Ant. The IFFT & GI applying circuits 813-1 to 813-N _MT-Ant may be omitted (strictly speaking, they may be included in the transmission signal processing circuits 811-1 to 811 -N _SDM ). In this case, the rest of the signal processing necessary after transmission weight multiplying in the transmission signal processing circuit _{811-1~811-N SDM} refers IFFT processing, insertion of the guard interval, the process of waveform shaping or the like.

In addition, the transmission weights to be multiplied by the transmission signal processing circuits 811-1 to 811-N _SDM are acquired from the first transmission weight calculation circuit 143 provided in the first transmission weight processing unit 140 at the time of signal transmission processing. Do. In the first transmission weight processing unit 140, in the channel information acquisition circuit 141, the channel information acquired by the reception unit 65 is separately acquired via the communication control circuit 121, and while the channel information is successively updated, the channel information storage is performed. It is stored in the circuit 142. At the time of signal transmission, the first transmission weight calculation circuit 143 reads channel information corresponding to the destination station from the channel information storage circuit 142 in accordance with an instruction from the communication control circuit 121, and calculates transmission weight based on the read channel information. Do. The first transmission weight calculation circuit 143 outputs the calculated transmission weight to the transmission signal processing circuits 811-1 to 811-N _SDM . Note that, in normal communication, since the other party with which the terminal station apparatus communicates is limited to a specific base station apparatus, in the above description, management regarding the terminal station apparatus to be the destination is explicitly shown. Naturally, it is also possible to perform the processing as being one.

  The feature of the present invention according to the first embodiment is that the calculation of the transmission weight is performed between the terminal station device 60 and the first transmission signal processing units 181-1 to 181-4 of the base station device 70. It is to use a virtual transmission line corresponding to the first singular value. There are several variations in the channel estimation method and transmission / reception weight calculation method in the case of utilizing the virtual transmission path corresponding to the first singular value, and the method for efficiently acquiring this will be described in detail later. Do. For example, the first right singularity when singular value decomposition is performed on each channel matrix in the uplink from the terminal station device 60 to the first transmission signal processing units 181-1 to 181-4 of the base station device 70. A vector may be used as a transmission weight vector. In this case, the first transmission weight calculation circuit 143 has a function of calculating the first right singular vector.

  Alternatively, when the base station apparatus 70 multiplies the plurality of antenna elements of each of the first transmission signal processing units 181-1 to 181-4 by a predetermined transmission weight vector for signal transmission, in fact, Equivalently to one in which each of the first transmission signal processing units 181-1 to 181-4 transmits from one virtual antenna element in spite of being transmitted from a plurality of transmission antennas It is. Therefore, the vector of channel information between this one virtual antenna element and each receiving antenna of the terminal station apparatus 60 is acquired, and the uplink processing is performed by the implicit feedback method of performing calibration processing on this channel vector. It is also possible to obtain channel information of The first transmission weight calculation circuit 143 calculates a vector obtained by taking the complex conjugate of the channel vector as shown in equation (7) based on the uplink channel vector thus obtained, or each component of the vector. Any of vectors in which the absolute values of all are made constant may be used as the transmission weight vector.

  As an alternative to this, the first transmission weight calculation circuit 143 includes the terminal station apparatus 60 and one antenna element in the first transmission signal processing units 181-1 to 181-4 of the base station apparatus 70. The channel vector of the channel information on the reception side is determined between the vectors, and a vector obtained by taking the complex conjugate of this channel vector as shown in equation (7) May be used as a transmission weight vector. Further, the communication control circuit 121 manages control related to the entire communication such as the entire timing control.

FIG. 22 is a schematic block diagram showing an example of the configuration of the reception unit 65 in the terminal station device 60 in the first embodiment of the present invention. As shown in FIG. 22, the receiving unit 65 includes antenna elements 851-1 to 851-N _MT-Ant , low noise amplifiers (LNA) 852-1 to 852-N _MT-Ant , a local oscillator 853 and a mixer 854. -1 to 854-N _MT-Ant , filter 855-1 to 855-N _MT-Ant , A / D (analog / digital) converters 856-1 to 856-N _MT-Ant , FFT (Fast Fourier) Transform (Fast Fourier Transform) circuits 857-1 to 857-N _MT-Ant , received signal processing circuits 145-1 to 145-N _SDM, and a first reception weight processing unit 144. The reception signal processing circuits 145-1 to 145-N _SDM and the first reception weight processing unit 144 are connected to the communication control circuit 121 shown in FIG. The first reception weight processing unit 144 includes a first channel information estimation circuit 146 and a first reception weight calculation circuit 147.

First, signals received by antenna elements _{851-1~851-N MT-Ant} is amplified by the low noise amplifier _{852-1~852-N MT-Ant.} The amplified signal and the local oscillation signal output from local oscillator 853 are multiplied by mixers 854-1 to 854-N _MT-Ant , and the amplified signal is downconverted from the radio frequency signal to the baseband signal. Ru. Since the downconverted signal includes a signal outside the frequency band to be received, the out _- of-band component is removed by the filters 855-1 to 855-N _MT-Ant . The signal from which the out-of-band component has been removed is converted into a digital baseband signal by A / D converters 856-1 to 856-N _MT-Ant . For example, in the case of using OFDM, the digital baseband signal is input to FFT circuits 857-1 to 857-N _MT-Ant , and the time axis is determined at a predetermined symbol timing determined by the circuit for timing detection omitted here The upper signal is converted into a signal on the frequency axis (separated into the signal of each subcarrier). The signals separated into the respective subcarriers are input to the reception signal processing circuits 145-1 to 145 -N _SDM and also to the first channel information estimation circuit 146.

The first channel information estimation circuit 146 transmits a first transmission signal of the base station apparatus 70 based on a known signal for channel estimation (such as a preamble signal added to the beginning of a radio packet) separated into each subcarrier. A channel of channel information between a virtual antenna element formed by each transmission weight vector of processing units 181-1 to 181-4 and each antenna element 851-1 to 851-N _{MT-Ant of} terminal station apparatus 60 A vector is estimated for each subcarrier, and the estimation result is output to the first reception weight calculation circuit 147. The first reception weight calculation circuit 147 calculates the reception weight to be multiplied for each subcarrier based on the input channel information. For this reception weight, for example, as described above, a ZF type pseudo inverse matrix is used, or an MMSE type reception weight matrix is used. At this time, the reception weight vector for combining the signals received by each of the antenna elements 851-1 to 851-N _MT-Ant differs for each signal sequence, and the above-mentioned ZF pseudo inverse matrix or MMSE type reception It corresponds to a row vector such as a weight matrix and is respectively input to the received signal processing circuits 145-1 to 145-N _SDM corresponding to the signal sequence to be extracted.

In received signal processing circuits 145-1 to 145-N _SDM , the signal for each subcarrier input from FFT circuits 857-1 to 857-N _{MT-Ant is} input from first reception weight calculation circuit 147. The reception weights are multiplied, and the signals received by the respective antenna elements 851-1 to 851-N _MT-Ant are added and synthesized for each subcarrier. The received signal processing circuits 145-1 to 145 -N _SDM perform demodulation processing on the added and synthesized signal, and output the reproduced data to the MAC layer processing circuit 68.

Here, different received signal processing circuits 145-1 to 145 -N _SDM perform signal processing of different signal sequences. Further, MLD, a simplified MLD using QR decomposition, or the like may be used as received signal processing across a plurality of received signal processing circuits 145-1 to 145-N _SDM . In addition, the MAC layer processing circuit 68 performs processing related to the MAC layer (for example, conversion of data input to or output from the interface circuit 67 and data to be transmitted / received on a wireless circuit, that is, wireless packet, termination of header information of the MAC layer Etc.). The received data processed by the MAC layer processing circuit 68 is output to an external device or network via the interface circuit 67. Further, the communication control circuit 121 manages control related to the entire communication such as the entire timing control.

  Here, as in the case of the transmitting side, the terminal station apparatus 60 at the time of reception can also perform signal processing in which a virtual transmission path corresponding to the first singular value is consciously used. An example of another structure of the receiving part 65 in the terminal | end station apparatus 60 in 1st Embodiment is shown in FIG.

In FIG. 23, reference numeral 154 denotes a first reception weight processing unit, reference numeral 155 denotes a first reception signal processing circuit, reference numeral 156 denotes a first channel information estimation circuit, and reference numeral 157 denotes a first reception weight calculation circuit, reference numeral 159 Denotes a second reception signal processing circuit, and the other is the same as FIG. In the above description, although it has been described that the first reception weight calculation circuit 157 calculates the reception weight for direct signal separation of the signal sequence of the N _SDM system, it is temporarily assumed that it corresponds to the first singular value. While performing signal separation in the transmission path, it is also possible to remove residual interference between signal sequences that still remain in two steps.

  In this case, in the first reception weight calculation circuit 157, for example, the antenna elements of the first transmission signal processing units 181-1 to 181-4 of the base station apparatus 70 are directed to the antenna elements of the terminal station apparatus 60. When channel information can be acquired, a first left singular vector at the time of singular value decomposition is calculated as a reception weight vector for a channel matrix having this channel information as a component. Alternatively, in the case where signal transmission is performed using one virtual antenna element formed by multiplying the transmission weight vectors by the first transmission signal processing units 181-1 to 181-4, A channel vector between the corresponding virtual antenna element and each antenna element of the terminal station device 60 may be determined, and each component of the reception weight vector may be determined by equation (7) based on this vector (maximum ratio combining) Of all the absolute values with respect to the value given by equation (7) or (7) may be made constant (weight of equal gain combination).

Then, the reception weight vector obtained in this manner is input to each of the first reception signal processing circuits 155-1 to 155-N _SDM . However, it is not necessary to frequently update the reception weight vector in real time in order to perform signal processing for separating residual interference in the second reception signal processing circuit 159 in this latter stage, and a line of sight wave of about 100 ms period, for example. It is also possible to reuse the common receive weight vector in the time domain where it is expected that the channel information will not fluctuate rapidly.

The first channel information estimation circuit 156 and the first reception weight calculation circuit 157 do not update the reception weight sequentially from such a viewpoint, for example, the channel estimation result of a certain degree is The channel estimation accuracy is improved by averaging, and the first reception weight calculation circuit 157 calculates the first reception weight vector at a predetermined cycle based on the averaged channel information, and calculates the first reception weight vector. It is also possible to input to the received signal processing circuits 155-1 to 155-N _SDM . In this case, in averaging, channel information may be considered as relative channel information (or each channel information divided by channel information of the reference antenna) based on the complex phase of the reference antenna (for example, the first antenna) It is preferable to utilize good).

In the first received signal processing circuits 155-1 to 155 -N _SDM , signals from virtual transmission paths corresponding to these first singular values are input to the second received signal processing circuit 159. The division of functions between the first received signal processing circuits 155-1 to 155-N _SDM and the second received signal processing circuit 159 is performed by the first received signal processing unit 185 and the second received signal processing unit 185 shown in FIG. Is similar to the relationship of the reception signal processing unit 75 of FIG. That is, a signal obtained by reducing the signal of a huge reception signal corresponding to the number NMT _-Ant of antenna elements into the number N _SDM of signal sequences spatially multiplexed by the first reception signal processing circuits 155-1 to 155-N _SDM . To be input to the second reception signal processing circuit 159, and the second reception signal processing circuit 159 performs general MIMO signal processing in a space whose dimension is reduced.

Specifically, the second received signal processing circuit 159 refers to a known training signal added at the beginning of the received signal, and acquires a channel matrix of N _SDM × N _SDM for the signal sequence of the N _SDM system. And perform reception signal detection processing based on the channel matrix. As described above, the second reception signal processing circuit 159 multiplies an inverse matrix of ZF type or a linear reception weight matrix of MMSE type, or non-linear such as MLD or simple MLD (QR-MLD etc.) Signal processing can also be performed. The second reception signal processing circuit 159 performs demodulation processing on the signal of the N _SDM system thus signal-separated, and outputs the reproduced data to the MAC layer processing circuit 68. This is equivalent to the signal processing of the second received signal processing unit 75 of the base station apparatus.

Here, an example of an apparatus configuration in the second received signal processing unit 75 of the base station apparatus 70 or the second received signal processing circuit 159 of the terminal station apparatus 60 (basically, processing is common to the base station apparatus and the terminal station apparatus Is shown in FIG. The basic operation is as described above, the signal sequence of N _SDM sequence as a received signal of the virtual transmission path corresponding to the first singular value of N _SDM book of the second received signal processing circuit 190 (FIG. 23 When the signal (corresponding to each subcarrier signal is inputted to perform the same signal processing although not explicitly shown here) is input to the second reception signal processing circuit 159, the channel matrix acquisition circuit 191 Then, with reference to a known training signal attached to the head of the received signal, a channel matrix of N _SDM × N _SDM is acquired for a signal sequence of N _SDM system. The reception weight matrix calculation circuit 192 calculates a reception weight matrix as a ZF inverse matrix or an MMSE linear reception weight matrix based on the channel matrix, and inputs this to the reception weight matrix multiplication circuit 193. The reception weight matrix multiplication circuit 193 multiplies subsequent data by the reception weight matrix to suppress an interference signal which is a crosstalk component between different virtual transmission paths. The signal detection circuit 194 performs signal detection on each signal whose SINR characteristic has been enhanced. Signal detection here is intended for general demodulation processing. For example, soft decision of the received signal is performed, error correction is performed after deinterleaving, and final signal detection is performed. Data expanded and parallel-transmitted into a plurality of signal sequences are converted into data of one sequence by parallel / serial conversion, and these are output to the MAC layer processing circuit. Although an example of linear signal processing is shown here as a typical example, the signal detection circuit 194 can also perform non-linear signal processing such as MLD to QR-MLD.

In addition, as shown in FIG. 25, the second transmission signal processing unit 71 of FIG. 17 is explicitly shown in the description of the configuration of the base station apparatus 70, and the second station shown in FIG. Add the transmission signal processing circuit 148, the transmission signal processing circuit _{811-1~811-N SDM} replaces the first transmission signal processing circuit _{821-1~821-N SDM,} further first transmission weight calculating circuit 143 Can be replaced with the transmission weight calculation circuit 149. In this case, the second transmission signal processing circuit 148 shown in FIG. 21 has the function of the second transmission signal processing unit 71 in FIG. 17, and the second transmission signal processing circuit 148 generates a radio packet to be transmitted by a radio channel. And a transmission weight matrix for compensating for signal leakage between virtual transmission paths corresponding to each first singular value from the transmission weight calculation circuit 149, and the N _SDM system The transmission signal vector of (1) is multiplied by the transmission weight matrix (transmission precoding) and output to the first transmission signal processing circuit 821. On the other hand, in the first transmission signal processing circuits 821-1 to 821-N _SDM , it is assumed that each of the transmission signals of the N _SDM system input from the second transmission signal processing circuit 148 corresponds to the first singular value. The transmission weight for forming a transmission path is multiplied. Here, as a function of the transmission weight calculation circuit 149, the first transmission signal processing circuits 821-1 to 821-N _SDM calculate transmission weights for forming a virtual transmission line corresponding to the first singular value. And a function for calculating a transmission weight matrix for compensating for signal leakage between virtual transmission paths corresponding to the respective first singular values. The transmission weight matrix in this case is obtained based on channel information acquired by, for example, a method using implicit feedback with calibration processing or a method using direct explicit feedback.

  A feature of the present invention according to the first embodiment is that a virtual corresponding to the first singular value is performed between the terminal station device 60 and the first transmission signal processing units 181-1 to 181-4 of the base station device 70. It is to use a dynamic transmission line. Therefore, with respect to each channel matrix from terminal station apparatus 60 toward first transmission signal processing units 181-1 to 181-4 of base station apparatus 70, the transmission weights are set to the first right singular vector when singular value decomposition is performed. By using a vector, the first transmission weight calculation circuit 143 has a function of calculating the first right singular vector. As an approximate solution of this first right singular vector, although it will be described in detail later, between the one of the first transmission signal processing units 181-1 to 181-4 of the base station apparatus 70 and one antenna element. A channel vector may be determined, and either a vector obtained by taking a complex conjugate of this channel vector or a vector in which all absolute values of the components of the vector are fixed may be used as the transmission weight vector. Essentially, the antenna element groups of the first transmission signal processing units 181-1 to 181-4 of the base station device 70 form a virtual directional antenna as a whole, but as an alternative, for example, this method It is equivalent to considering the case where it represents with one antenna which exists near the center physically among antenna element groups as an approximate solution. In this case, although the weight of the approximate solution is different from the weight of the exact solution, the gain obtained as a result of evaluation by simulation does not necessarily have an extremely large deterioration as described later.

The above is the description of the configurations of the terminal station device 60, the transmitting unit 61, and the receiving unit 65 in the first embodiment. The important point here is that the local oscillator 815 in the transmitter 61 is shared by the mixers 816-1 to 816 -N _MT-Ant in each antenna system of the transmitter 61, and the local oscillator 853 in the receiver 65 is a receiver The mixers 854-1 to 854-N _MT-Ant in each of the 65 antenna systems are common points. In directivity control, the phases of transmission and reception signals are adjusted for each antenna element, but by making the phase relationship of the signals input from each local oscillator 815 or local oscillator 853 always be constant, It is possible to determine whether the transmission / reception weights should be multiplied with the same phase relationship. On the other hand, when the local oscillator 815 or the local oscillator 853 uses a plurality of asynchronous ones in the transmission unit 61 or in the reception unit 65, directivity control in which at least the transmission weight is multiplied in the transmission unit 61 functions effectively. It disappears. Care should be taken in this regard in the design of the device. It is also possible to share the local oscillator 815 and the local oscillator 853.

(About extension to Point-to-Multipoint)
In the above description, since the explanation has been made centering on the point-to-point type wireless entrance circuit, if the arrangement of the first signal processing unit 304 is operated in an optimized state by some method, the first signal is processed. Spatial multiplexing transmission as many as the number of processing units 304 is possible. The optimization method for this purpose may be an antenna arrangement method to be described later, or it may be set by searching for an arrangement in which virtual transmission paths are substantially orthogonal while setting it at several locations when installing the antenna. However, when one base station apparatus 70 and a plurality of terminal station apparatuses 60 simultaneously perform point-to-multipoint communication, an ideal antenna arrangement on the base station apparatus 70 side is common to all the terminal station apparatuses. Configuration is difficult, so a redundant number of first signal processing units 304 are installed, and the configuration is such that communication is performed by selecting a different combination of first signal processing units 304 for each terminal station apparatus 60 to communicate with. It does not matter. In this case, since the communication control circuit determines the optimum combination of the first signal processing units 304, the communication control circuit combines the optimum first signal processing units 304 for each terminal station apparatus 60. It is necessary to have functions such as a database for information.

(About the processing flow of signal processing in the present invention)
The processing flow of signal processing in the first embodiment of the present invention will be described below. Although the signal processing flow of the base station apparatus and the terminal station apparatus is basically the same, the names and code numbers of circuits to be referred to are different, so here, the signal processing flow regarding the base station apparatus will be described as an example. Also, as described above, the signal processing in the second transmission signal processing unit 71 (corresponding to the second transmission signal processing circuit 148 in FIG. 25 with explicit description in the case of the terminal station apparatus) It is also possible not to apply weight matrix multiplication (transmission precoding), and in this case, the part regarding the signal processing can be omitted.

  FIG. 26 shows an outline of a signal processing flow at the time of signal transmission in the first embodiment. When the transmission process is started, the communication control circuit 120 instructs the first signal processing unit 304 and the second signal processing unit 305 to read the first transmission weight vector and the second transmission weight matrix (steps S2601) (step S2602).

  The reading of the transmission weight vector and the transmission weight matrix here is, for example, in the case of the point-to-multipoint communication and in the case of the base station apparatus 70, the terminal station apparatus 60 which is the communication counterpart station is different each time communication Therefore, every time transmission is performed, read processing is performed. On the other hand, in the case of fixed point-to-point type communication, or when the communication partner station is always fixed to the base station device 70 like the terminal station device 60, the process is omitted even if it is not read every time. Is also possible. However, when the time variation of the channel can not be ignored and the transmission weight is changed each time, it may be read out each time transmission, even if the communication partner is fixed. Alternatively, the transmission weights updated in a predetermined cycle may be read. In addition, when the matrix multiplication in the second transmission weight multiplying circuit in the second signal processing unit 305 is omitted as described above, the reading of the transmission weight matrix in the second signal processing unit 305 becomes unnecessary. .

The second signal processing unit 305 generates transmission signals for N _SDM systems for spatial multiplexing (step S2603), and multiplies the transmission signals by the second transmission weight matrix (step S2604). The transmission data of the N _SDM system after multiplication is transferred to the corresponding first signal processing unit 304 (step S2605). The second signal processing unit 305 determines whether transmission of transmission data is completed (step S2606). If the transmission of transmission data is not completed (step S2606: NO), the second signal processing unit 305 returns the process to step S2603. If the transmission of transmission data has been completed (step S2606: YES), the second signal processing unit 305 ends the signal processing flow at the time of signal transmission.

On the other hand, the first signal processing unit 304 multiplies the transmission signal (one-dimensional signal) by the transmission weight vector using the transmission weight vector for each first signal processing unit 304 read out in advance (step S2607) The transmission antenna number is converted into a transmission signal vector of N _Ant dimensions, and transmission signal processing is performed for each antenna system (step S2608).

  For example, when assuming the OFDM modulation scheme, transmission signal generation is performed for each subcarrier and for each OFDM symbol, and multiplication of a transmission weight matrix, multiplication of a transmission weight vector, etc. are all performed separately for each subcarrier. To be done. As transmission signal processing here, IFFT processing is performed on the transmission signal for each subcarrier on the frequency axis, guard intervals are added, waveform shaping between symbols is added as necessary, and so on. After sampling data is D / A converted to generate an analog baseband signal, this is up-converted by a mixer, then out-of-band components are removed by a filter, and the signal amplified by a high power amplifier is transmitted from an antenna .

  The above is an example in the case of using the OFDM modulation scheme, but the same is true for other schemes such as SC-FDE, and various conventional communication schemes can basically be applied. Further, the signal processing on the frequency axis is not necessarily required. As described later, in the case where the multiplication of the transmission weight vector is performed on the time axis, sampling using a single transmission weight vector (and matrix) is performed. The transmission weight vector (and matrix) may be multiplied for each data. Further, processing such as error correction coding and interleaving is included in the generation processing of the transmission signal, and is omitted in the description here.

  Next, FIG. 27 shows an outline of a first example of signal processing flow at the time of signal reception in the first embodiment. When the signal is received, the communication control circuit 120 instructs the plurality of first signal processing units 304 to read out the respective first reception weight vectors (step S2701).

  As in the case of the transmitting side, reading of the receiving weight vector here is, for example, point-to-multipoint communication and in the case of the base station apparatus 70, each time the terminal station apparatus 60 which is the communication counterpart station performs communication. Because it is different, each time transmission is performed, read processing is performed. On the other hand, in the case of fixed point-to-point type communication, or when the communication partner station is always fixed to the base station device 70 like the terminal station device 60, the process is omitted even if it is not read every time It is also possible. However, when the time variation of the channel can not be ignored and the reception weight is changed each time, it may be configured to read out each time transmission, even if the communication partner is fixed. Alternatively, the reception weight updated at a predetermined cycle may be read out.

  Thereafter, when an actual signal is received, signal reception processing is performed for each antenna system (step S2702). The signal reception processing here is, for example, amplification of a received signal by a low noise amplifier, down conversion processing by a mixer, removal of out-of-band frequency components by a filter, and digital baseband by an A / D converter. Convert to signal sample values. In these received signal processing, for example, in the case of the OFDM modulation method, the signal is cut out for each OFDM symbol, the guard interval is removed, and the signal is converted to a signal on the frequency axis for each subcarrier component by FFT.

  For the signal subjected to the above reception signal processing, the reception weight vector is multiplied to the reception signal vector having the signal of each antenna element as the vector component (step S2703), and the result is subjected to the second signal processing. It transfers to the part 305 (step S2704). The first signal processing unit 304 determines whether reception of reception data is completed (step S2705). Here, if the reception data continues further (step S2705: YES), the process returns to the reception signal processing shown in step S2702 to continue the processing, and when the reception data is finished (step S2705: NO) The processing of the signal processing unit 304 of 1 ends.

On the other hand, in the second signal processing unit 305, it is determined whether or not the signal transferred from the first signal processing unit 304 is, for example, a channel estimation training signal added to the beginning of the wireless packet (step S2706). If it is a training signal (step S2706: YES), channel estimation is performed (step S2707), and a second reception weight matrix is generated based on the obtained channel matrix (step S2708). On the other hand, if it is a data portion which is not a training signal (step S2706: NO), the second reception weight matrix is multiplied by the space multiplex number N _SDM dimension reception signal vector (step S2709), and after signal separation. A signal detection process is performed (step S2710). The signal detection process here means, for example, a process of estimating a transmission signal through soft decision processing, deinterleaving processing, error correction processing and the like on the transmission signal, and the MAC layer after the signal processing of these physical layers is completed. It is output to the processing circuit.

  In addition to the OFDM modulation method, general conventional techniques such as the SC-FDE method can be similarly applied to received signal processing. Furthermore, although the description has been made as signal processing for each subcarrier, it is not necessary to be signal processing on the frequency axis, and multiplication of the reception weight vector and the reception weight matrix is performed on the time axis as described later. In such a case, the reception weight vector and the reception weight matrix may be multiplied for each sampling data using a single reception weight vector and a reception weight matrix. For MIMO signal processing equivalent to, for example, step S2709 of multiplying the second reception weight matrix (and step S2710 of performing signal detection processing after signal separation), the second reception weight matrix is necessarily multiplied as linear signal processing. It is not necessary to do, and it is also possible to apply general MIMO signal detection processing such as QR-MLD or MLD.

In the above description, it is assumed that the local oscillator 853 used in the first signal processing unit 304 is asynchronous for each first signal processing unit 304 as in the base station apparatus 70, and the second reception weight matrix is a signal. In the case of the terminal station apparatus 60, for example, since the local oscillator 853 used for upconversion processing and / or downconversion processing of all antenna elements is shared, in the case of the terminal station apparatus 60, reception is not always necessary. The second reception weight matrix does not change every time. As described above, when the local oscillator 853 is generally shared and channel fluctuation can be ignored, it is also possible to use the second reception weight matrix acquired in the past. FIG. 28 shows an outline of a second example of signal processing flow at the time of signal reception in the first embodiment of the present invention. Steps S2801 to S2805 illustrated in FIG. 28 are the same as steps S2701 to S2705 illustrated in FIG. The difference from FIG. 27 is that, in the signal processing of the second signal processing unit 305, the second signal processing unit 305 generates the second reception weight matrix from the communication control circuit 120 when reception processing of the signal is started. In response to the read instruction, the second reception weight matrix is read (step S2806). After that, when the reception signal is transferred from the first signal processing unit 304, the spatial multiplexing number N _SDM dimensional reception signal vector is multiplied by the second reception weight matrix (step S2807), and then signal detection is performed. Processing is performed (step S2808). The other signal processing is the same as in FIG.

  As described above, the wireless communication system 50 according to the first embodiment includes the base station device 70 (first wireless station device) and the terminal station device 60 (second wireless station device). The base station device 70 includes a plurality of first transmission signal processing units 181, a plurality of first reception signal processing units 185, a second transmission signal processing unit 71, and a second reception signal processing unit 75. . The plurality of first transmission signal processing units 181 have a plurality of antenna elements 819. The plurality of first reception signal processors 185 have a plurality of antenna elements 851. The second transmission signal processing unit 71 performs signal processing of wireless communication associated with the plurality of first transmission signal processing units 181. The second received signal processing unit 75 performs signal processing of wireless communication associated with the plurality of first received signal processing units 185. The terminal station device 60 includes a plurality of antenna elements 819, a plurality of antenna elements 851, a transmission unit 61, and a reception unit 65. The transmission unit 61 performs wireless communication with the plurality of first reception signal processing units 185 via the plurality of antenna elements 819. The transmitter 61 wirelessly transmits the plurality of first received signal processors 185 and the second received signal processor 75 associated with the first received signal processor 185 via the plurality of antenna elements 819. Communication may be performed. The receiving unit 65 performs wireless communication with the plurality of first transmission signal processing units 181 via the plurality of antenna elements 851. The receiving unit 65 wirelessly transmits the plurality of first transmission signal processing units 181 via the plurality of antenna elements 851 and the second transmission signal processing unit 71 associated with the first transmission signal processing unit 181. Communication may be performed. The first transmission signal processing unit 181 and the first reception signal processing unit 185 transmit weight vectors and reception weights for the MIMO channel matrix used for wireless communication between the first antenna element group and the second antenna element group. At least one of the vectors is at least one of a first right singular vector and a first left singular vector corresponding to a first singular value of the MIMO channel matrix or an approximate solution of the first right singular vector and an approximate solution of the first left singular vector Calculated based on at least one of The plurality of first transmission signal processing units 181 and the first reception signal processing unit 185 correspond to the first transmission signal processing unit 181 or the first reception signal processing unit 185 of the transmission weight vector and the reception weight vector. Each of the independent signal sequences is spatially multiplexed and transmitted using at least one of them.

  Thus, the base station apparatus 70, the terminal station apparatus 60, the wireless communication system 50, and the wireless communication method of the first embodiment can increase the transmission capacity by MIMO in an environment where the line-of-sight environment is dominant. For example, the base station device 70, the terminal station device 60, the wireless communication system 50, and the wireless communication method use the high-order spatial multiplexing and the high frequency band while the line-of-sight environment is dominant in the future wireless communication system in the mobile network. Thus, it is possible to realize a large capacity.

  The base station apparatus 70 as the base station apparatus 303 according to the first embodiment includes a first signal processing unit 304 in which antenna elements are bundled in a narrow area, and a second signal processing unit 305 that aggregates them. The first signal processing unit 304 performs closed beamforming in each signal processing unit and does not perform beamforming across different first signal processing units 304. The individual first signal processing unit 304 generates transmission / reception weights for the “virtual transmission line corresponding to the first singular value” that makes full use of the “line of sight wave”, and performs a plurality of first signal processing Spatial multiplexing transmission is performed using the individual transmission and reception weights in the unit 304.

  Here, in the above description, it has been described that the line-of-sight environment is the dominant environment as a typical embodiment, but even if it is not a perfect line-of-sight environment, for example, a very strong reflected wave comes from a specific direction. In the case where a diffracted wave that is comparable to the line-of-sight wave arrives, a virtual transmission path corresponding to the first singular value is formed in the incoming direction. In this sense, the virtual transmission line corresponding to the first singular value is not necessarily the transmission line formed by the line-of-sight wave, and in the first embodiment of the present invention, the virtual transmission line corresponding to the first singular value The first embodiment of the present invention is applicable even in a non-line-of-sight environment, as it is to carry out spatial multiplexing transmission by actively utilizing a plurality of transmission paths. The circuit configuration and the processing flow in that case can be applied as they are without changing the above description at all.

Second Embodiment
[Antenna placement condition for orthogonalization of line-of-sight MIMO transmission]
(Outline of Basic Principle according to Second Embodiment)
Here, the conditions for the plurality of virtual transmission paths to be substantially orthogonal are organized. In FIG. 16 described above, since one directional beam is effectively formed for every 25 antenna elements, this is approximately one antenna with a very high directional gain (for example, a parabolic antenna) Is equivalent to using Therefore, FIG. 29 shows an example of a case where four parabola antennas are mounted on the base station apparatus and sixteen antenna elements are mounted on the terminal station apparatus in a linear array. In FIG. 29, reference numeral 306 denotes a base station apparatus, reference numeral 302 denotes a terminal station apparatus, and reference numerals 307-1 to 307-4 denote parabola antennas. The terminal station device 302 corresponds to the terminal station device 60. The base station device 306 corresponds equivalently to the base station device 70. Basically, transmission equivalent to the parabola antennas 307-1 to 307-4 shown in FIG. 29 is realized by grouping a large number of antenna elements of the base station apparatus 70.

  Here, the condition in which the four singular values of the MIMO channel in FIG. 29 in which the line-of-sight wave is dominant becomes substantially equally large values as in the case described above is organized. For example, the conditions under which MIMO transmission in a line-of-sight environment is established are defined in the patent document (Japanese Patent No. 5488894), but the basic idea of these conventional techniques is the antenna elements provided in the base station apparatus and the terminal station apparatus Interval as much as possible, and as a result, the distance between each transmitting antenna and receiving antenna is dispersed, and the relationship of the difference of the distance becomes pseudo random relationship (or predetermined relationship) Try to adjust to

  On the other hand, for example, in the phased array antenna technology, if the distance between the antenna elements of a linear linear array is d, the value of d is set small, as seen from an antenna at a distance in the direction in which the directivity should be directed. Radio waves arriving at (or sent out) antenna elements of individual linear arrays can be approximated as plane waves, and if the direction of incoming waves is represented by angle θ, the path difference for each antenna element is high at d · sin θ It can be approximated to the accuracy. However, in order to set the distance between the antenna elements as wide as possible according to the basic idea of the above-mentioned prior art, such plane wave approximation can not be performed, and the approximation needs to be performed under different conditions. The As a result, the accuracy of the prior art approximation is reduced. In fact, when performing MIMO communication in a line-of-sight environment, both transmitting and receiving stations assumed large parabolic antennas. Usually, since the size of the parabolic antenna is much larger than the wavelength, it is not possible to approximate the plane wave with the accuracy with which the above-mentioned path length difference can be ignored with respect to the wavelength.

However, while the effective aperture length of the antenna on the base station apparatus 70 side is extended, the distance between the antenna elements on the terminal station apparatus 60 side is about several wavelengths (the distance between the base station apparatus 70 and the terminal station apparatus 60). For example, if it is set to the order of 3 wavelengths or less, the path difference for each antenna element on the terminal station apparatus 302 side from the parabola antenna on the base station apparatus 306 side in FIG. It becomes possible to approximate. Therefore, as shown in FIG. 30, a terminal station corresponding to a base station apparatus antenna #j (310-j) (1.ltoreq.j.ltoreq.M) of the base station apparatus 70 corresponding to the base station apparatus 306 and a terminal station apparatus 302. Assuming that the distance between the terminal station apparatus antenna # 1 (320-1) of the apparatus 60 is L _j and the angle difference between the terminal station apparatus antenna # 1 (320-1) and the front direction is θ _j , The channel information between the channel matrix (from the jth antenna on the base station apparatus 70 side to the ith ( _{1.ltoreq.i.ltoreq.N MT -Ant} ) antenna of the terminal station apparatus 60 using the antenna spacing d of the terminal station apparatus 60 is _hij If 表 す is expressed, it can be approximately expressed by the following equation (26). In Equation (26), N _{MT -Ant} is denoted as N in order to avoid complication. Further, although the number of parabolic antennas 307-1 to 307-4 included in the base station apparatus 306 is four in FIG. 29, the number of antenna elements is M as a general number here.

Here, the (j, j) component of the second item indicates channel information between the j-th antenna of the base station 70 and the first antenna of the terminal station 60 although it is displayed in the form of a product of matrices. The difference information of each component of the N _{MT -Ant} antennas of the terminal station device 60 corresponds to the column vector of the j-th column of the first item. Since the (j, j) component of the second item is a common term applied to all components of the column vector, if the column vectors of the first item are orthogonal, each singular value is stably large and Become. The condition that the i-th column vector of the first term and the j-th column vector are orthogonal is expressed by the following equation (27) that the inner product of the complex vectors is zero.

The above equation transformation uses the formula of the sum of geometric progressions. Since this last condition is sufficient if the numerator is zero under the condition that the denominator is not zero, the condition that the term in the parentheses of exp of the numerator is an integral multiple of 2πi is conditional expression (28) below Led. In the following formulas, N is described as the original number of antennas N _{MT -Ant} .

Constant _{K 1} of the formula (28) is an integer not the number of antennas _N a multiple of _MT-Ant terminal station 60. Here, as shown in FIG. 30, assuming that the j-th antenna on the base station apparatus 70 side has displacement d _{j in} the lateral direction from directly in front of the antenna on the terminal station apparatus 60 side, the difference between the i and j antennas is Δd _ij I assume. The distance between the base station apparatus 70 and the terminal station apparatus 60 is L, and L / d is multiplied by both sides of the first equation of Expression (28). Although the distance between each antenna element of base station apparatus 70 and the first antenna of terminal station apparatus 60 slightly differs, if distance L is sufficiently larger than displacement d _j , it can be approximated as LLL _j .

That is, with respect to the number of antennas _{N MT-Ant} terminal station _{60, N MT-Ant} to integer _{K 2} not be zero at integer multiples, the interval d of the antenna elements of the terminal station 60, terminal station 60 and the base With respect to the distance L of the station device 70 and the wavelength λ of the signal wave of wireless communication, the distance between any two antenna elements may be set to be the distance satisfying the equation (29). If this condition is geometrically interpreted, the condition can be easily realized, for example, under the following condition. For example, as the simplest conditions, on a line linear array and the distance of the terminal station 60 side L away and directly opposite, allows any offset in the linear direction, N in λL / N _{MT-Ant d} spacing _{MT- The} point of the _Ant point may be set linearly, the M point may be selected from the N _MT-Ant points, and the M antenna elements of the base station apparatus 70 may be arranged. This means that any two integers are selected from among consecutive N _{MT -Ant} integers, and their absolute value becomes (N _{MT -Ant} -1) or less whenever K is a difference, and K ₂ is N _MT It utilizes that it can avoid becoming an integral multiple of _Ant . In addition, it is possible to select the antenna installation point of the base station apparatus from other wider conditions by searching and setting an arrangement in which the difference between any two individual integers is not an integer multiple of N _MT-Ant. It is.

As shown in FIG. 29, when the antenna element on the terminal station apparatus 60 side is miniaturized, the antenna element number N _MT-Ant on the terminal station apparatus 60 side is increased accordingly, while the antenna of the base station apparatus 70 is generally In order to set the number of elements M to the upper limit of the assumed number of spatial multiplexing, it is assumed that the value of M is smaller than the number of antenna elements N _MT-Ant on the terminal station apparatus 60 side. That is, when N _{MT -Ant} > M, the M antennas on the base station apparatus 70 do not have to be equally spaced even under the simplest conditions described above, and intermittently from among the above N _{MT -Ant} points. The arrangement position of the antenna element may be set to Note that although the actual number of antenna elements is 100 in the case of FIG. 16, M can be 4 in FIG. 16 because it can be considered as four virtual antenna elements by grouping into 25 units. Should be considered. Moreover, although the approximation with very high accuracy is performed until derivation | leading-out of Formula (28), the approximation used in Formula (29) has a little low precision. However, although it depends on the setting of the parameters, if the distance between the antenna elements of base station apparatus 70 in equation (29) is temporarily about 1 m, even if there is an error of several% of about several cm, they are in almost orthogonal state There is no difference in things. The same applies to the first embodiment of the present invention, but basically, even if there are interference components between a plurality of virtual transmission paths on the receiving side, it is possible to suppress the interference by signal processing on the receiving side. If such signal processing is assumed, the loss can be minimized by making the signal substantially orthogonal to the last, so the approximation accuracy of equation (29) does not have a large influence on the system operation.

  If such properties are taken into consideration, if each grouped antenna element group shown in FIG. 16 is regarded as one virtual antenna, that central point of the physically spread multi-element antenna is Assuming that the physical position of the dynamic antenna and installing the plurality of antenna element groups in the above-described antenna arrangement, using a weight vector represented by the first singular vector, a plurality of signal sequences with desired characteristics are obtained. It becomes possible to transmit by space multiplexing.

In the above description, the antenna element on the base station apparatus and the antenna element on the terminal apparatus face each other as shown in FIG. As shown in FIG. 31, it is assumed that there is no facing relationship. This is due to the restriction on the installation places of the base station apparatus 306 and the terminal station apparatus 302. In such a case, the desired effect can be derived by converting the parameters slightly. In FIG. 31, when the terminal station apparatus 302 is viewed from the base station apparatus 306, the terminal station apparatus 302 exists in a direction shifted by an angle θ from the front, and when the base station apparatus 306 is viewed from the terminal station apparatus 302. The base station device 306 exists in a direction deviated from the front by an angle θ ′. In this case, the distance _{D 1} of the parabolic antenna 307-1 and 307-4 of the base station apparatus 306 is equivalently visible state narrowed to _{D 1 '(= D 1 cosθ} ). Similarly, the width D ₂ of the linear array 312 of the terminal station device 302 appears to be equivalently narrowed to D ₂ ′ (= D ₂ cos θ ′). In other words, it can be understood that the antenna element distance d is converted to dcosθ × cosθ ′ times. Assuming that the distance between the terminal station apparatus 302 and the vicinity of the approximate center of gravity of the parabolic antennas 307-1 to 307-4 of the base station apparatus 306 is L, after performing these conversions, optimum antenna arrangement using the above-mentioned conditional expression The condition of can be calculated. That is, the base station apparatus 70 virtually projects each linear array on an axis orthogonal to the straight line in the line-of-sight direction. Similarly, the terminal station device 60 virtually projects each linear array on an axis orthogonal to the straight line in the line-of-sight direction. The d of the equation (29) is converted into d cos θ × cos θ ′ based on the virtually projected spacing of the on-axis antenna elements. On the basis of the spacing of the virtually projected on-axis antenna elements, d in equation (29) is converted by a conversion device or a person.

  Incidentally, although the distance between the terminal station apparatus 302 and the vicinity of the approximate center of gravity of the parabolic antennas 307-1 to 307-4 of the base station apparatus 306 is L in FIGS. 29 and 31, the exact distances of the individual antenna elements are common. Rather than the distance L, each will have an error. However, as understood from the use of approximation in the middle stage of the calculation of equation (29), errors in the distance L here are also acceptable to some extent, and the antennas facing each other as shown in FIG. The distance between the elements (the distance between two straight lines of the antenna elements arranged in parallel straight lines) may be used, or as shown in FIG. 31, the distance may be approximately the center of gravity of the antenna elements. Thus, the distance L may be treated as an approximate value or an approximate value.

In the description of FIG. 29 and FIG. 30 in the present description, the case where the terminal station apparatus is configured by a linear array is shown as a typical example. In this feature, parabola antennas 307-1 to 307-4 or 310-1 to 310-M (corresponding to antenna element groups 304-1 to 304-4 in FIG. 16) on the base station apparatus side are disposed on a straight line Similarly, the antenna elements on the terminal station side are arranged in a linear array on a straight line. In the calculation of the orthogonalization condition described above, the two straight lines are described as being in a parallel relationship. Specifically, assuming a horizontal axis parallel to the wall of the building, for example, on the wall of the building or on the roof of the building, a plurality of first signal processing units and antenna elements (groups) 307 along the horizontal axis. It is preferable to arrange the linear array of the terminal station apparatus also on a horizontal axis parallel to the wall surface of the opposite building separated from the road when the No.-1 to 307-4 are installed. Here, a vertical axis orthogonal to the horizontal axis on the terminal station side is set, and a grid parallel to the horizontal axis and the vertical axis is assumed, and the linear array of the terminal station described above is N in the vertical direction. Consider the case of using a stacked square grid array. At this time, the total number of antenna elements of the terminal station apparatus becomes N × N ′ elements. However, looking at the antenna elements (groups) 307-1 to 307-4 of the plurality of first signal processing units of the base station apparatus from this terminal station apparatus, although there are angular differences in the horizontal direction, they are perpendicular. As there is no difference between the antenna element (groups) 307-1 to 307-4 with respect to the elevation angle of the direction, each of the N 'stages of the above-mentioned square lattice array is not regarded as an independent antenna element, and is substantially vertical. It is understood that N 'elements arranged in a direction behave equivalently as one virtual antenna element, and the virtual antenna elements are arranged in a linear array on the horizontal axis in N elements. In fact, this effect is also confirmed in simulation evaluation, and the antenna element of the terminal station apparatus is parallel to the axis connecting the antenna element (groups) 307-1 to 307-4 on the base station apparatus side. When arranging an N × N ′ element in a lattice formed of an axis and an axis orthogonal to the axis, the N × N ′ elements are parallel to the axis connecting antenna elements (groups) 307-1 to 307-4 on the base station apparatus side. If it is regarded as a linear array of N elements on the axis and the element intervals of the N elements, the element intervals of the antenna elements (groups) 307-1 to 307-4 on the base station apparatus side can be optimized by applying equation ( 29 ) good. Naturally, when the antenna elements (groups) 307-1 to 307-4 on the base station apparatus side are vertically aligned with the wall of the building, etc., the antenna elements on the terminal station apparatus side are also N elements in the vertical direction. Equation ( 29 ) may be applied by regarding N ′ elements arranged in a horizontal direction in a lattice, and regarding this as a linear array of equivalent N elements aligned in the vertical direction. Furthermore, although the antenna arrangement on the terminal station side has been described as a square lattice array here, the antenna arrangement need not necessarily be a square lattice, and the element spacing in the horizontal direction and the vertical direction may be different in a rectangular shape. In this case, the antenna element of the equation ( 29 ) can be obtained by setting the distance between the antenna elements arranged on the axis on the terminal station side parallel to the axis on which the antenna elements (groups) 307-1 to 307-4 on the base station side are aligned. It may be calculated as the interval d.

As described above, the wireless communication system 53 of the second embodiment includes the base station device 306 (first wireless station device) and the terminal station device 302 (second wireless station device). The base station apparatus 306 includes antenna elements (parabola antennas) 307-1 to 307-4, and the terminal station apparatus 302 includes a terminal station apparatus antenna element group 312. The plurality of parabola antennas (antenna elements) 307-1 to 307-4 of the base station apparatus 306 have a distance Δd _mn between the m-th element and the n-th element as shown in equation (29), and a terminal station apparatus 302 based on the distance L between the base station 306 and the base station 306, the wavelength λ of the signal wave of wireless communication, the number of antennas N _MT-Ant of the terminal station 302, and the spacing d of the antenna elements of the terminal station 302 Ru. The parabolic antennas 307-1 to 307-4 are, for example, each a single high gain antenna element (single antenna element). The high gain antenna element is, for example, a parabolic antenna 307. The parabolic antennas 307-1 to 307-4 may be antenna element groups instead of a single high gain antenna element. When using the antenna element group in this manner, the base station apparatus 306 may be the base station apparatus 303 shown in FIG. 16 and the base station apparatus 17 in FIG. The base station apparatus 70 in FIG. 17 includes a plurality of first transmission signal processing units 181, a plurality of first reception signal processing units 185, a second transmission signal processing unit 71, and a second reception signal processing unit 75. And. The plurality of first transmission signal processing units 181 have a plurality of antenna elements 819. The plurality of first reception signal processors 185 have a plurality of antenna elements 851. The second transmission signal processing unit 71 performs signal processing of wireless communication associated with the plurality of first transmission signal processing units 181. The second received signal processing unit 75 performs signal processing of wireless communication associated with the plurality of first received signal processing units 185. The terminal station device 60 includes a plurality of antenna elements 819, a plurality of antenna elements 851, a transmission unit 61, and a reception unit 65. The transmission unit 61 performs wireless communication with the plurality of first reception signal processing units 185 via the plurality of antenna elements 819. The receiving unit 65 performs wireless communication with the plurality of first transmission signal processing units 181 via the plurality of antenna elements 851.

  That is, the distance L between the antenna elements or the distance between the antenna element groups is the distance L between the base station device 306 (first wireless station device) and the terminal station device 302 (second wireless station device), and the signal wave of wireless communication And the number N of linear array antenna elements constituting the second antenna element group or the number N of antenna elements in either the vertical direction or the horizontal direction arranged in a lattice, and the second antenna element Each single antenna element constituting the plurality of first antenna groups or an integral multiple of the value calculated based on the spacing d of either the vertical direction or the horizontal direction of the antenna elements constituting the group Antenna element groups are arranged.

  Thus, the base station device 306 or 70, the terminal station device 302 or 60, the wireless communication system 53 or 50, and the antenna element arrangement method of the second embodiment transmit the transmission capacity by MIMO in an environment where the line of sight environment is dominant. It is possible to increase. The base station device 306 or 70, the terminal station device 302 or 60, the wireless communication system 53 or 50, and the antenna element arrangement method according to the second embodiment spatially multiplex and transmit a plurality of signal sequences with desired characteristics. Becomes possible. The base station apparatus 306 or 70, the terminal station apparatus 302 or 60, the wireless communication system 53 or 50, and the antenna element arrangement method of the second embodiment can improve the SIR characteristics. The base station apparatus 306 or 70, the terminal station apparatus 302 or 60, the wireless communication system 53 or 50, and the antenna element arrangement method of the second embodiment can realize higher spatial multiplexing.

  The base station apparatus 306 or 70, the terminal station apparatus 302 or 60, the wireless communication system 53 or 50, and the antenna element arrangement method according to the second embodiment are the side of the base station apparatus 306 or 70 in an environment dominated by line of sight. In a MIMO channel of a plurality of parabolic antennas 307 and a plurality of element antennas on the terminal station apparatus 302 or 60 side, antenna installation conditions for orthogonalizing each channel are defined. However, here the conditions for mounting a parabola antenna on the base station side are shown to simplify the discussion, but it goes without saying that a high directivity antenna other than the parabola antenna is used, and furthermore, the base station side has multiple antenna elements Even in the case where the first signal processing unit as shown in FIG. 16 configured as shown in FIG. 16 is used, orthogonalization between the respective transmission paths can be generally achieved under the same conditions. This makes it possible to improve the communication quality on the transmission path and to increase the transmission capacity.

Third Embodiment
[Applied technology to access systems of a plurality of virtual transmission paths corresponding to the first singular value]
(Basic principle according to the third embodiment)
In the above description, in the transmission path between the base station apparatus and the terminal station apparatus, an environment dominated by a line of sight wave has been taken. As such an example of application, it may be considered that the terminal station device 60 is regarded as a relay station, and the above-mentioned technique is used in an entrance line from the base station device 70 to the relay station. However, the virtual transmission line corresponding to the first singular value is not limited to the use in the wireless entrance line. Also in the above description, it is assumed that the terminal station device 60 corresponding to the relay station is small and that a large number of small antennas are arranged in a very narrow area. Therefore, for example, if a large number of antennas are mounted on a smartphone or the like, and in particular if reception at a small cell of the 5th generation mobile communication system (5G) is assumed, utilization of millimeter waves or quasi-millimeter waves is assumed, It can be regarded as an environment in which the antenna is condensed and disposed in a narrow place just as in the case of the above-mentioned wireless entrance except that the user moves. That is, as an example, when the small cell base station is disposed at a relatively high location such as a wall of a building, the base station apparatus 70 and the terminal station apparatus 60 are generally in a line of sight environment from almost directly upward. Considering the situation in which the antenna with the directivity gain is implemented, it is expected that communication will be performed in an environment where the line of sight is dominant. In such a case, it is effective to group the antennas of the base station apparatus into a plurality of groups as in FIG. 16 and to actively use virtual transmission paths equivalent to the first singular value in each group similarly. Become.

  Here, considering the difference between the case of FIG. 16 and the case of the access system, in FIG. In general, the three-dimensional positional relationship of each antenna is a combination of the antenna arrangement of the smartphone, the angle at which the user holds the smartphone, the direction in which the user is facing, etc. It is expected that they are not parallel to each other but twisted. In this case, although the optimum value as shown in equation (29) does not generally exist, for example, an antenna having a group of grouped antenna elements slightly redundantly and an optimum antenna of a combination with low correlation in selection diversity An element group may be selected and used. However, in this case, it is necessary to set the distance between the antenna element groups slightly larger so that the correlation of the channels can be reduced as much as possible. In that case, it may be effective to install antennas across a plurality of structures instead of arranging all the antenna element groups only on the wall surface of the same structure (building or the like). In such a case, it is difficult to connect the second signal processing unit 305 and the first signal processing units 304-1 to 304-4 with a wired cable (including an optical fiber) in FIG.

  Therefore, the antenna element group (first signal processing unit 304) of the base station apparatus 303 is installed as a satellite base station, and the general base station apparatus corresponding to the second signal processing unit 305 of the base station apparatus 303 It is effective to realize transmission and reception of signals to and from the satellite base station apparatus by a wireless channel. FIG. 32 shows a third embodiment of the present invention. In FIG. 32, reference numeral 31 denotes a general base station apparatus, reference numerals 32-1 to 32-3 denote satellite base station apparatuses, and reference numeral 33 denotes a terminal station apparatus. The terminal station device 33 corresponds to the terminal station device 60. Reference numeral 37 represents a structure. Reference numeral 38 represents a structure. The overall base station apparatus 31 is disposed, for example, on the top surface (the roof of a building or the like) of the structure 37. The satellite base station device 32 is disposed, for example, on the side surface (wall such as a building) of the structure 38. The terminal station device 33 is disposed, for example, in the cell 39 between the structure 37 and the structure 38.

  The overall control base station apparatus 31 and the satellite base station apparatuses 32-1 to 32-3 realize the functions of the above-described base station apparatus 70 as a whole. The central base station apparatus 31 includes a large number of antenna elements, and the overall arrangement of the antenna elements is small so that the distance between the antenna elements is about the same as the wavelength (for example, between several wavelengths and 1/2 wavelength). On the other hand, the satellite base station apparatus 32 has a small size in which a large number of antenna elements are arranged at the same interval (for example, between several wavelengths and a half wavelength) as the first signal processing unit 304 in FIG. Equipped with an antenna. Communication is performed between the integrated base station apparatus 31 and the satellite base station apparatuses 32-1 to 32-3 using the first singular vector corresponding to the first singular value of each antenna. The terminal station device 33 also has a plurality of antennas. For example, if a smartphone or the like is assumed, a plurality of antennas will be mounted in a relatively narrow area. Here, also between the terminal station apparatus 33 and the satellite base station apparatuses 32-1 to 32-3, communication is performed using the first singular vector corresponding to the first singular value of each antenna as the transmission / reception weight vector. Furthermore, communication is also performed between the satellite base station apparatuses 32-1 to 32-3 and the central base station apparatus 31 using the first singular vector corresponding to the first singular value of each antenna as the transmission / reception weight vector. As a result, stable MIMO transmission with small channel correlation can be achieved by utilizing a plurality of different satellite base station apparatuses 32-1 to 32-3 while utilizing transmission / reception weight vectors that maximize the line-of-sight gain. It will be possible.

  Next, FIG. 33 shows another example of the third embodiment of the present invention. In FIG. 33, reference numerals 34-1 to 34-2 denote a supervising base station apparatus, reference numerals 35-1 to 35-6 denote satellite base station apparatuses, and reference numerals 36-1 to 36-2 denote terminal station apparatuses. FIG. 33 is the same as FIG. 32 described above, but a number of satellite base station devices 35-1 to 35 arranged on the side surface (the wall of the building) of the structure 38 and the top surface of the structure 37 (the roof of the building) The difference from FIG. 32 is that the correspondence of the central base station apparatuses 34-1 to 34-2 arranged in the is not one-to-one correspondence. The satellite base station apparatuses 35-1 to 35-6 can be flexibly combined with a plurality of integrated base station apparatuses 34-1 to 34-2 to provide an access link. For example, when viewed from the overall control base station device 34-1 near the terminal station device 36-1, the nearest satellite base station devices are the satellite base station devices 35-1 to 35-3. However, assuming that the satellite base station devices with small correlation from the terminal station device 36-1 are the satellite base station devices 35-1, 35-2, 35-4, the general base station device 34-1 is the next general base station. The satellite base station apparatus 35-4 closer to the station apparatus 34-2 may be used to provide communication to the terminal station apparatus 36-1 using a virtual transmission path corresponding to the first singular value. become.

  Incidentally, one of the reasons for utilizing the virtual transmission line corresponding to the first singular value in such an access system is that channel information of the virtual transmission line corresponding to the first singular value is other higher order. As compared with the channel information of the virtual transmission line corresponding to the singular value, it is mentioned that the time variation is relatively small relatively. That is, the first right (left) singular vector in singular value decomposition represents a path corresponding to a line-of-sight wave, and there is no large change in the transmission / reception weight to be applied with some movement. However, since the high-order right (left) singular vector corresponds to a path containing a large amount of reflected wave components, a slight movement of the terminal significantly changes the channel state after synthesis, and as a result, the right (left) singular vector Is a big fluctuation. That is, if the channel information changes significantly, the transmit / receive weight vector for securing the gain also fluctuates, and if the fluctuation is ignored and transmission / reception is performed with the value as it is, the gain corresponding to the fluctuation decreases and the characteristic degrades. Unavoidable. In particular, if the terminal station device 36 is a small portable terminal such as a smartphone, the channel correlation between the antennas may be large, and it may be difficult to increase the capacity in single-user MIMO. In such a case, as shown in an embodiment described later, it is effective to increase the total total capacity by multi-user MIMO that simultaneously accommodates a large number of terminals with small transmission capacity, but channel time fluctuation is not Mutual interference, which degrades SINR characteristics, leading to a decrease in transmission capacity and destabilization of communication. However, by actively using a virtual transmission path corresponding to the first singular value with a relatively small channel time variation, the mutual interference between terminals is significantly increased even in the case of utilizing multi-user MIMO. It is also possible to obtain secondary effects that can be reduced.

 Generally, multi-user MIMO means that one base station device performs space multiplex transmission with a plurality of terminal stations, but here, as shown in FIG. 32, a plurality of integrated base station devices It can be understood including the case where services 34-1 to 34-2 simultaneously provide services to a plurality of terminal stations 36-1 to 36-2. In this case, satellite base station apparatuses 35-1 to 35- may use the same frequency channel for communication, and may utilize them, although each general base station apparatus and terminal station apparatus may actually be one-to-one communication. In the case where the satellite base station devices 35-1 to 35-6 are used while mutually using the resources so as to allow partial duplication as well as one another, a certain satellite base station device 32 can simultaneously receive a plurality of terminal station devices 36. However, since such a mode of operation can also be regarded as multi-user MIMO in a broad sense, since the signal is to be transmitted, it is possible to effectively reduce the above-mentioned effect of channel time fluctuation even in such a case. Become.

(Outline of non-regeneration re-beamforming relay in satellite base station equipment)
A basic operation when relay transmission is performed via a plurality of satellite base station apparatuses 32 using a virtual transmission path corresponding to the above-described first singular value will be described with reference to the drawings. In the configuration using the satellite base station apparatus 32 shown in FIG. 32, the second signal processing unit 305 and the first signal processing units 304-1 to 304- of the base station apparatus 303 shown in FIG. 16 described above are used. The configuration of No. 4 corresponds to the function of the base station as a whole constituted by the general base station 31 and the satellite base stations 32-1 to 32-3 in FIG. 32, and the second signal processor 305 in FIG. 32 and the first signal processing units 304-1 to 304-4 are connected by wire, in FIG. 32, the general base station apparatus 31 and the satellite base station apparatuses 32-1 to 32-3 are connected. It corresponds to having been configured to be connected wirelessly. In FIG. 16, the signal transmission between the second signal processing unit 305 and the first signal processing units 304-1 to 304-4 secures a situation in which there is no signal interference since the transmission medium is wired. Therefore, in FIG. 32, when the wired part is completely rewired to the radio, each satellite base station apparatus 32 is also used for the radio signal between the central base station apparatus 31 and the satellite base station apparatuses 32-1 to 32-3. -1 to 32-3 will ensure a situation where there is no signal interference. However, in the third embodiment, since the signal processing of signal separation is performed in the terminal station apparatus 33 (in the case of downlink) or the general base station apparatus 31 (in the case of uplink), the relay is not necessarily associated with the wired case. The point in this embodiment is that there is no need to completely separate the signal in part. However, when signal transmission / reception between the general control base station apparatus 31 and the satellite base station apparatuses 32-1 to 32-3 is performed with no directivity, the line gain corresponding to the first singular value can not be obtained. Transmission / reception signal processing is performed between the base station device 31 and the satellite base station devices 32-1 to 32-3 using the first singular vector for transmitting and receiving weights for utilizing the virtual transmission path corresponding to the first singular value. .

In FIG. 34, signal processing between the master base station apparatus 31 and the satellite base station apparatus 32 and signal processing between the satellite base station apparatus 32 and the terminal station apparatus 33 in the third embodiment of the present invention Provides an overview of In FIG. 34, reference numeral 31 denotes a general base station apparatus, reference numeral 32 denotes a satellite base station apparatus, reference numeral 33 denotes a terminal station apparatus, and reference numeral 52 denotes a wireless communication system, which correspond to the same numbers in FIG. A plurality of satellite base station apparatuses 32 exist, and a plurality of satellite base station apparatuses 32 are shown vertically in the center of FIG. In Figure 34, the general base station apparatus 31 and the satellite base station apparatus 32 performs communication using the frequency f _1, the satellite base station apparatus 32 and the terminal station is communicating at the frequency f _2. However, assuming communication in FDD (Frequency Division Duplex: Frequency Division Duplex), the central base station apparatus 31 and the satellite base station apparatus 32 communicate on the frequencies f _1-1 and f _1-2 and the satellite base The station apparatus 32 and the terminal station apparatus may communicate with each other at frequencies _f2-1 and _f2-2 . Here, for the sake of simplicity, assuming that TDD (Time Division Duplex), uplink and downlink in each section use the same frequency.

Also, as can be seen in Figure 34, except for the differences frequencies _{f 1} and _{f 2,} generally has a folded state symmetric, satellite base station apparatus 32-1 to 32-3 are first in FIG. 16 The base station apparatus 31 is similar to the terminal station apparatus 302 in FIG. 16 and is similar in circuit configuration and signal processing. Is configured to correspond to the terminal station device 302. Therefore, in the following description of the circuit configuration, the description will be made with reference to the reference numerals shown in FIG.

  Furthermore, for example, when using OFDM, it is necessary to perform individual signal processing in each subcarrier, but in order to simplify the explanation, the notation for identification of the subcarrier in each signal and transmission / reception weight is omitted here. Do.

FIG. 34 particularly shows signal processing at the time of transmission of a signal from the central base station apparatus 31 toward the terminal station apparatus 33. Although not particularly shown here, it should be noted that all of these signal processings are performed for each subcarrier as in the above processing. However, as described later, in the case where the same transmission / reception weight can be used within the entire frequency bandwidth, it is possible to extend the processing to perform signal processing on the time axis for each sampling. First, when transmitting the signal sequences S _{1 to} S _N of the N _SDM system (here, N = N _SDM here for convenience of the notation of subscripts) in the central base station apparatus 31, the general base station apparatus 31 In the second transmission signal processing circuit 148, in addition to the generation of the signal sequences S _{1 to} S _N as general transmission signal processing, between satellite base station apparatuses 32 as signal processing shown in (i) in FIG. _Assuming that the transmission weight matrix for signal separation of is _Wtx , signal conversion is performed by the following equation (30).

Here, although FIG. 25 is also described, in the signal processing corresponding to the second transmission signal processing circuit of FIG. 25, it is not necessary to introduce the transmission weight matrix W _tx , and W _tx is regarded as a unit matrix. The transmission signal series t _{1 to} t _N may remain as the signal series S _{1 to SN} . Next, directivity formation corresponding to the first singular value is performed on each satellite base station apparatus 32-1 to 32-N _SDM . Specifically, the channel matrix for each antenna of the kth _S satellite base station apparatus 32 to which attention is paid from each antenna of the supervising base station apparatus 31 is given by the first right singular vector at the time of singular value decomposition of the matrix. The transmission weight vectors w _tx, k S _{, 1 to} w t x _, k S _{, Nc} (here, for convenience of description _, N c is the number of antenna elements included in the central base station apparatus) are given by The transmission signal t _kS is multiplied as shown in ii).

As shown in equation (32) ((iii) in FIG. 34), the signals of the N _SDM system addressed to each satellite base station apparatus 32 are added and synthesized for each antenna element to generate a time axis signal.

Furthermore, the IFFT & GI addition circuit 813 of the central base station apparatus 31 performs IFFT conversion on the signal on the frequency axis shown in Equation (32) to convert it to a signal on the time axis (strictly, for example, it may be OFDM). For example, insertion of guard intervals and signal processing such as wave formation are also included, but the description is omitted here). The D / A converter 814 of the dominating base station apparatus 31 D / A converts the result of the IFFT conversion to generate an analog baseband signal. The mixer 816 of the general base station apparatus 31, an analog baseband signal, multiplies the signal from the local oscillator 815 of the frequency of f _1, upconverts the analog baseband signal to a frequency band of the signal f _1. Filter 817 of general base station apparatus 31 removes out-of-band signals from the signal in the frequency band of f _1. The HPA 818 of the central base station apparatus 31 amplifies the signal in the frequency band of f ₁ and transmits the amplified signal from the antenna element 819.

Antenna elements of the satellite base station apparatus 32 receives a signal in the frequency band of f _1. Low-noise amplifier of the satellite base station apparatus 32 (LNA) amplifies the signal in the frequency band of _{f 1.} The mixer of the satellite base station apparatus 32 downconverts the signal in the frequency band of f ₁ into an analog baseband signal. The filter of the satellite base station apparatus 32 removes out-of-band signals from the analog baseband signal. The A / D converter of the satellite base station apparatus 32 converts an analog baseband signal into a digital baseband signal. The FFT circuit of the satellite base station apparatus 32 converts the digital baseband signal into a subcarrier received signal based on the symbol timing of the sampled signal.

Here, in the satellite base station apparatus 32, a first left singular vector at the time of singular value decomposition of the channel matrix for each antenna of the satellite base station apparatus 32 to which attention is paid from each antenna of the integrated base station apparatus 31. The reception weight vector w ′ _{rx, kS, 1 to} w ′ _{rx, kS, M} given by is given by the equation (33) ((iv) in FIG. 34), and the reception signal vector r ′ _{kS, 1 to} r ′ Multiply _{kS, M.} Here, M is the number of antenna elements of the satellite base station apparatus 32, and represents the number of antenna elements performing communication with the central base station apparatus 31.

The scalar quantity signal _R'kS obtained here is not for estimating the transmission signal of the general base station apparatus 31, but an intermediate one-dimensional signal for extracting the virtual transmission line of the first singular value to the last And is different from regenerative relay. However, unlike ordinary non-regenerative relaying, it corresponds to generating the above-mentioned intermediate signal _R'kS and performing reception directional beamforming and transmission directional beamforming separately. With respect to the above-mentioned intermediate signal _R'kS obtained in this manner, in transmission from the satellite base station apparatus 32 to the terminal station apparatus 33 corresponding to the last access system, Signal transmission is performed utilizing a virtual transmission path corresponding to the first singular value. Specifically, with respect to the channel matrix for each antenna of the terminal station device 33 from each antenna of the satellite base station device 32 of interest, a transmission weight vector given by the first right singular vector at the time of singular value decomposition of the matrix. _{w 'tx, kS, 1 ~w} ' tx, kS, the _M, multiplying the equation (34) the received signal vector R _'kS as shown in ((v) in FIG. 34).

Here, in the satellite base station apparatus 32, the number of antenna elements for the terminal station apparatus 33 has been described as M, but this does not have to be the same as the number of antenna elements on the general base station apparatus 31 side. The number M of antenna elements may be understood as the number m of antenna elements. The transmission signal vector of each subcarrier obtained in this manner is converted to a time axis signal by IFFT (strictly, for example, in the case of OFDM, insertion of guard intervals, signal processing such as wave forming system, etc. Also, the description is omitted here), D / A conversion is performed to generate an analog baseband signal, which is multiplied with the signal from the local oscillator of the frequency of f ₂ by the mixer to obtain the frequency band of f ₂ The signal is up-converted to a signal of (1) and the out-of-band signal is removed by a filter, and then amplified by a high power amplifier and transmitted.

Here, in the equations (33) and (34), the reception signal vector is multiplied by the reception weight vector, and the R ′ _{k S} obtained thereafter is multiplied by the transmission weight vector. Is configured to generate a transmission weight matrix by previously multiplying these transmission weight vectors and reception weight vectors, and directly multiplying the reception signal vectors r ′ _{kS, 1 to} r ′ _{kS, M.} I do not care. However, the amount of operation in this case is 2 × M complex multiplications when equations (33) and (34) are performed separately, but when performed collectively, generation of transmission weight matrix and matrix × Both vector multiplications require a total of 2 × M ² complex multiplications, and it is more efficient to divide them into individual operations.

Next, signal processing for the signal received by the terminal station device 33 will be described. Antenna elements of the terminal station 33 851, receives a signal in the frequency band of _{f 2.} The LNA 852 amplifies a signal in the f ₂ frequency band. The mixer 854 downconverts the signal in the f ₂ frequency band to an analog baseband signal. The filter 855 removes out-of-band signals from the analog baseband signal. The A / D converter 856 converts an analog baseband signal into a digital baseband signal. The FFT circuit 857 converts the digital baseband signal into a subcarrier received signal based on the symbol timing of the sampled signal.

Here, in the terminal station apparatus 33, first, the first channel information estimation circuit 146 or the first channel information estimation circuit 156 is a virtual antenna element formed by each antenna element of each satellite base station apparatus 32 by a transmission weight vector Channel information with each antenna of the terminal station device 33 is acquired from the training signal added to the received signal. The first reception weight calculation circuit 147 or the first reception weight calculation circuit 157 generates a reception weight from the acquired channel information. Here, the reception weight may be a ZF type or MMSE type reception weight matrix for directly separating the signal sequence of the N _SDM system in one step (corresponding to FIG. 22), which corresponds to the first singular value. It is also possible to use a two-stage reception weight multiplication process (corresponding to FIG. 23) which temporarily separates the signals of the virtual transmission line and then cancels the mutual interference between the virtual transmission lines. In the latter case, for each channel vector between a virtual antenna element formed by each antenna element of each satellite base station apparatus 32 by a transmission weight vector and each antenna of the terminal station apparatus 33, this vector is used as a basis. Each component of the reception weight vector may be obtained by Equation (7) (weight of maximum ratio combination), or all absolute values may be made constant with respect to the value given by Equation (7). (Equal gain combining weights). The reception weight _{w rx, kS, 1 ~w rx} , kS, Nm ( notational convenience of subscript here, _{N m} is the number of antenna elements _{N MT-Ant} provided in the terminal station device), and the terminal station Assuming that the signal vector received by each antenna element of the device 33 is Rx, the signal is once separated into virtual transmission paths by the equation given by the equation (35) ((vi) in FIG. 34).

Here r _kS is the received signal of the virtual transmission channel via the first k _S satellite base station apparatus 32. At this time, since it is a signal of MIMO transmission of N _SDM × N _SDM , general MIMO signal processing is performed in the second reception signal processing circuit 159 of the terminal station device 33 thereafter. Specifically, the channel matrix of N _SDM × N _SDM is acquired for the signal sequence of N _SDM system with reference to the known training signal attached to the head of the received signal, and the received signal is detected based on the channel matrix Do the processing. As described above, it is also possible to multiply a ZF type inverse matrix or an MMSE type linear reception weight matrix, or to perform non-linear signal processing such as MLD. Demodulation processing is performed on the N channels of signals thus separated, and the reproduced data is output to the MAC layer processing circuit 68. As an example, in the case of using a linear reception weight matrix W _rx of ZF type or MMSE type, the operation shown in Equation (36) ((vii) in FIG. 34) is performed. Here, N _SDM is expressed as N in the equations (36) and (37) to avoid complexity.

Note that, as described above, direct signal separation may also be performed using the reception signal of each antenna element of the terminal station apparatus 33 in one step instead of the two-step signal processing of Equation (35) and Equation (36) It is possible. This is expressed by equation (37) using a linear reception weight matrix ~ W _rx (... is written on the matrix W _rx ) of ZF type or MMSE type which directly separates the signal to the reception signal vector Rx It corresponds to performing signal processing.

  The above-described features of the embodiment of the present invention are not limited to the satellite base station apparatus 32 having the regenerative relay function in the prior art, and on the other hand, is not limited to the normal non-regenerative relay function only. Implements intermediate processing such as reception weight vector multiplication and further multiplication of the signal by the transmission weight vector, and that transmission / reception weight vector basically makes effective use of virtual transmission paths corresponding to a plurality of first singular values It is the point that it becomes the directivity control to do.

  FIG. 35 shows a circuit configuration of the satellite base station apparatus according to the embodiment of the present invention. Reference numeral 700 in FIG. 35 denotes an antenna element. The code | symbol 701 shows TDD-SW. Reference numeral 702 indicates a low noise amplifier (LNA). The code | symbol 703 shows a mixer. Reference numeral 704 indicates a filter. Reference numeral 705 denotes an A / D converter. Reference numeral 706 denotes an FFT circuit. Reference numeral 707 indicates a reception weight multiplication circuit. Reference numeral 708 denotes a transmission weight multiplication circuit. Reference numeral 709 denotes an IFFT & GI application circuit. Reference numeral 710 denotes a D / A converter. Reference numeral 711 indicates a mixer. Reference numeral 712 indicates a filter. Reference numeral 713 indicates a high power amplifier. The code | symbol 714 shows TDD-SW. Reference numeral 721 denotes an antenna element. Reference numeral 722 indicates a low noise amplifier (LNA). Reference numeral 723 indicates a mixer. Reference numeral 724 indicates a filter. Reference numeral 725 indicates an A / D converter. Reference numeral 726 indicates an FFT circuit. Reference numeral 727 denotes a reception weight multiplication circuit. Reference numeral 728 indicates a transmission weight multiplication circuit. Reference numeral 729 indicates an IFFT & GI application circuit. Reference numeral 730 denotes a D / A converter. Reference numeral 731 indicates a mixer. Reference numeral 732 indicates a filter. Reference numeral 733 indicates a high power amplifier. Reference numeral 740 indicates a transmission weight processing unit. Reference numeral 741 denotes a local oscillator. Reference numeral 742 indicates a local oscillator. Reference numeral 743 denotes a communication control circuit. Hereinafter, for convenience of explanation, the left side will be described as a side that communicates with the central base station apparatus 31, and the right side will be a side that communicates with the terminal station apparatus 33.

  First, the downlink direction (overall base station apparatus 31 → satellite base station apparatus 32 → terminal station apparatus 33 direction) will be described. The signal received by each antenna element 700 is input to the low noise amplifier 702 via the TDD-SW 701. Here, the communication control circuit 743 performs switching of the TDD-SW 701 in accordance with transmission / reception timing. The weak signal input to the low noise amplifier 702 is amplified, and the amplified signal is multiplied by the local oscillation signal output from the local oscillator 741 by the mixer 703, and the amplified signal is converted from the radio frequency signal to the baseband Down-converted to a signal of Since the downconverted signal also includes frequency components outside the frequency band to be received, the filter 704 removes the out-of-band components. The signal from which the out-of-band component has been removed is converted to a digital baseband signal by an A / D converter 705. The digital baseband signals are all input to the FFT circuit 706, and the signals on the time axis are converted into signals on the frequency axis at predetermined symbol timings determined by the timing detection circuit (not shown). Separate) into signals. The signal separated into each frequency component is input to the reception weight multiplication circuit 707 and also input to the transmission / reception weight processing unit 740-1.

  In the reception weight multiplication circuit 707, the reception weight vector of the first singular value is multiplied corresponding to the virtual transmission line carried by the satellite base station apparatus 32, and the scalar signal of the plurality of antenna elements is converted into one scalar system. It is converted to a signal. This signal is input to the transmission weight multiplication circuit 708. The transmission weight multiplication circuit 708 corresponds to the virtual transmission path corresponding to the first singular value from the satellite base station apparatus 32 to which attention is paid to the terminal station apparatus 33. Are multiplied to generate signals corresponding to the respective antenna elements 721, these are input to the IFFT & GI application circuit 709 of each antenna element system, and the IFFT & GI application circuit 709 generates a signal on the time axis from the signal on the frequency axis. Further, processing such as insertion of guard intervals and waveform shaping between OFDM symbols (between blocks of block transmission if SC-FDE (Single-Carrier Frequency Domain Equalization) is performed) is performed, and for each antenna element 721, Digital sampling data to baseband analog at D / A converter 710 It is converted to issue. Furthermore, each analog signal is multiplied by the local oscillation signal input from the local oscillator 742 by the mixer 711 and up-converted to a radio frequency signal. Here, since the upconverted signal includes a signal in a frequency component outside the band of the channel to be transmitted, the filter 712 removes the frequency component out of the band to generate an electrical signal to be transmitted. . The generated signal is amplified by the high power amplifier 713 and transmitted from the antenna element 721 through the TDD-SW 714.

  Next, the uplink direction (terminal station apparatus 33 → satellite base station apparatus 32 → general base station apparatus 31 direction) will be described. The signal received by each antenna element 721 is input to the low noise amplifier 722 via the TDD-SW 714. Here, the communication control circuit 743 performs switching of the TDD-SW 714 in accordance with the timing of transmission and reception. The weak signal input to the low noise amplifier 722 is amplified, and the amplified signal is multiplied by the local oscillation signal output from the local oscillator 742 by the mixer 723 and the amplified signal is converted from the radio frequency signal to the baseband Down-converted to a signal of Since the down-converted signal includes frequency components outside the frequency band to be received, the filter 724 removes the out-of-band components. The signal from which the out-of-band component has been removed is converted to a digital baseband signal by an A / D converter 725. The digital baseband signals are all input to the FFT circuit 726, and the signals on the time axis are converted into signals on the frequency axis at predetermined symbol timings determined by the timing detection circuit (not shown). Separate) into signals. The signal separated into each frequency component is input to the reception weight multiplication circuit 727 and also to the transmission / reception weight processing unit 740-2.

  In the reception weight multiplication circuit 727, the reception weight vector of the first singular value is multiplied corresponding to the virtual transmission line carried by the satellite base station apparatus 32, and a vector of signals from multiple antenna elements is multiplied by a scalar of one system. It is converted to a signal. This signal is input to the transmission weight multiplication circuit 728. In the transmission weight multiplication circuit 728, the transmission weight is made to correspond to the virtual transmission path corresponding to the first singular value from the satellite base station 32 to which attention is paid The vectors are multiplied to generate signals corresponding to the respective antenna elements 700, which are input to the IFFT & GI application circuit 729 of each antenna element system, and the IFFT & GI application circuit 729 generates a signal on the frequency axis from the signal on the time axis. It is converted into a signal, and further processing such as insertion of a guard interval and waveform shaping between OFDM symbols (between blocks of block transmission if SC-FDE (Single-Carrier Frequency Domain Equalization) is performed) is performed. , D / A converter 730 to analyze the baseband signal from digital sampling data. It is converted into grayed signal. Further, each analog signal is multiplied by the local oscillation signal input from the local oscillator 741 by the mixer 731 and up-converted to a radio frequency signal. Here, since the upconverted signal includes a signal in a frequency component outside the band of the channel to be transmitted, the filter 732 removes the frequency component out of the band to generate an electrical signal to be transmitted. . The generated signal is amplified by the high power amplifier 733 and transmitted from the antenna element 700 through the TDD-SW 701.

  The transmission and reception weight vector multiplied by the reception weight multiplication circuit 707 and the transmission weight multiplication circuit 728 is calculated by the transmission and reception weight processing unit 740-1. The transmission and reception weight vector multiplied by the reception weight multiplication circuit 727 and the transmission weight multiplication circuit 708 is calculated by the transmission and reception weight processing unit 740-2. The reception weight vector is calculated based on the channel information acquired using the downlink and uplink FFT circuits, and the transmission weight vector in the opposite direction is calculated by performing calibration processing. In the transmission and reception between the general base station apparatus 31 and the channel, the channel is fixed, and once channel information is acquired from the viewpoint of a line-of-sight environment at a high place, that value may be used continuously thereafter. It is possible.

  Therefore, it is possible to repeatedly perform channel estimation before the start of service operation and use the channel estimation result of high accuracy (in this case, using relative channel information) by averaging and the transmission / reception weight vector calculated therefrom. It is possible. On the other hand, channel information between the satellite base station apparatus 32 and the terminal station apparatus 33 changes due to movement of the terminal station apparatus 33 and its surroundings or environmental fluctuations, so it is necessary to obtain channel information and update weight vectors in real time. It becomes. Any means may be used for this channel information acquisition (channel feedback) method. Then, with respect to the channel information acquired in this manner, the transmission / reception weight processing units 740-1 and 740-2 transmit / receive weight vectors of virtual transmission paths corresponding to the first singular value using, for example, a method such as singular value decomposition. It is calculated and input to reception weight multiplication circuits 707 and 727 and transmission weight multiplication circuits 708 and 728. The communication control circuit 743 recognizes transmission / reception timing and symbol timing, and gives various instructions to the TDD-SWs 701 and 714 and various circuits. The basis for the communication control circuit 743 to grasp such timing can also be grasped by synchronizing with the general base station apparatus 31 in communication with the general base station apparatus 31, and GPS or other wireless systems (for example, It may be performed using information obtained from the signal from the base station of the macro cell.

  As described above, the wireless communication system 52 according to the third embodiment includes the central base station apparatus 31 (first wireless station apparatus), a plurality of satellite base station apparatuses 32 (second wireless station apparatus), and a terminal And a station apparatus 33 (third radio station apparatus). The central base station apparatus 31 includes a first wireless station signal processing unit that performs wireless communication with a plurality of satellite base station apparatuses 32 via the first antenna element group. The satellite base station apparatus 32 executes wireless communication with the central base station apparatus 31 via the second antenna element group, and executes wireless communication with the terminal station apparatus 33 via the third antenna element group. 2 has a radio station signal processing unit.

  The second wireless station signal processing unit of the satellite base station apparatus 32 (second wireless station apparatus) of the third embodiment is used for wireless communication between the first antenna element group and the second antenna element group. At least one of a first right singular vector and a first left singular vector corresponding to a first weight of the MIMO channel matrix, or an approximate solution of the first right singular vector and a first left It is calculated based on at least one of the approximate solutions of the singular vector. The second radio station signal processing unit generates a transmission signal sequence by multiplying the signal received via the second antenna element group by the reception weight vector. The second wireless station signal processing unit transmits a signal based on the generated signal sequence to the terminal station device 33 (third wireless station device) via the third antenna element group.

  The second wireless station signal processing unit of the satellite base station apparatus 32 (second wireless station apparatus) of the third embodiment includes the fourth antenna element group and the satellite of the terminal station apparatus 33 (third wireless station apparatus). A transmission weight vector for a MIMO channel matrix used for wireless communication with the third antenna element group of the base station apparatus 32 (second wireless station apparatus) is a first right corresponding to a first singular value of the MIMO channel matrix Calculated based on at least one of the singular vector and the first left singular vector, or at least one of the approximate solution of the first right singular vector and the approximate solution of the first left singular vector, and the third antenna element group Signal series of the plurality of systems are generated by multiplying the received signals by the transmission weight vector, and signals based on the signal series of the plurality of systems are transmitted through the second antenna element group. And it transmits to end station device 33 (third radio station).

  The first wireless station signal processing unit of the wireless communication system 52 of the third embodiment transmits the transmission weight vector for the MIMO channel matrix used for wireless communication between the first antenna element group and the second antenna element group, At least one of a first right singular vector and a first left singular vector corresponding to a first singular value of the MIMO channel matrix, or an approximate solution of the first right singular vector and an approximate solution of the first left singular vector The signal sequence is generated for each second radio station apparatus, the generated signal sequence is multiplied by the transmission weight vector corresponding to each signal sequence, and the multiplication result is added and synthesized for all signal systems. , And transmit a signal based on the additively synthesized signal sequence to the satellite base station apparatus 32 (second wireless station apparatus) via the first antenna element group.

  The first wireless station signal processing unit of the wireless communication system 52 of the third embodiment is configured to receive weight vectors for a MIMO channel matrix used for wireless communication between the first antenna element group and the second antenna element group, At least one of a first right singular vector and a first left singular vector corresponding to a first singular value of the MIMO channel matrix, or an approximate solution of the first right singular vector and an approximate solution of the first left singular vector By calculating the satellite base station apparatus 32 (second radio station apparatus) based on the above and multiplying the signal received via the first antenna element group by the reception weight vector for each second radio station apparatus A plurality of signal sequences are generated, and residual interference components are removed from signals based on the plurality of signal sequences.

  Thus, the general control base station apparatus 31, satellite base station apparatus 32, terminal station apparatus 33, radio communication system 52, and radio communication method of the third embodiment increase the transmission capacity by MIMO in an environment where the line-of-sight environment is dominant. It is possible to In the general base station apparatus 31, satellite base station apparatus 32, terminal station apparatus 33, radio communication system 52, and radio communication method of the third embodiment, the transmission capacity can be increased by the non-regeneration re-beam forming relay. It becomes. In the integrated base station apparatus 31, satellite base station apparatus 32, terminal station apparatus 33, radio communication system 52, and radio communication method of the third embodiment, a plurality of signal sequences may be spatially multiplexed and transmitted with desired characteristics. It will be possible.

  In the first embodiment, the base station apparatus 70, the terminal station apparatus 60, the wireless communication system 50, and the wireless communication method use space multiple access using a plurality of virtual transmission paths corresponding to the first singular value. It is also possible to However, since the coordinates of the terminal station device 60 can not be designed in a fixed manner, it is preferable to arrange a plurality of first signal processing units 304 on the base station device 70 side in a distant place in space multiplexing. For this reason, when arranging a function equivalent to the first signal processing unit 304 as the satellite base station apparatus 32, it is preferable to relay wirelessly rather than wire routing between them. In order to realize this mode of use, signal processing in which only beamforming is performed at the relay station while being non-regenerative relay is effective, and an implementation method therefor is defined.

Fourth Embodiment
[Train moving cell using multiple virtual transmission lines corresponding to the first singular value]
(Outline of train moving cell and basic principle)
As described in the third embodiment above, in the future wireless access for 5G (fifth generation), in particular, instead of accommodating the traffic of a large number of users in the train in a macro cell, these traffics are processed in the train It is effective to aggregate and collectively offload user data to the wired network at the wireless entrance in order to effectively utilize the macrocell's frequency resources and to improve the total transmission capacity of the entire system. is there. Here, no distinction is made between trains and trains, and all vehicles such as conventional lines and Shinkansen are marked as trains.

  However, in the case of trains, the moving speed is generally high, and in the case of Shinkansen, the wireless entrance circuit is configured assuming transmission speeds exceeding 300 [km / h] and 100 [km / h] or more for conventional lines. Must. In particular, in order to provide a large capacity line, it is necessary to use a high frequency band such as millimeter wave, and in line design, the gain loss proportional to the square of the wavelength and the high power amplifier of the millimeter wave band In view of the fact that the output is very small compared to the microwave band and the high output is expensive, it is necessary to secure the line gain by making the large-scale antenna using many low-output antenna elements. In particular, in consideration of the moving speed of the train, in order to reduce the frequency of handover and reduce the control load, it is necessary to cover the area of about 500 [m] intervals with the base station apparatus on the entrance side. In order to secure the gain of a large-scale antenna, it is necessary to perform feedback of channel information with high accuracy and to perform directivity control with high accuracy. At this time, it is necessary to follow the fluctuation with high accuracy and to form the directivity at high speed.

  As mentioned above, in general, the time variation of the MIMO channel tends to be relatively small in the virtual channel corresponding to the larger singular value obtained when singular value decomposition of the channel matrix is performed. On the contrary, in a virtual transmission line corresponding to a small singular value, channel time variation tends to be very large. Here, since it is assumed that the transmitting and receiving stations are in the mutual line-of-sight environment, the virtual transmission line corresponding to the first singular value is a transmission line corresponding to the line-of-sight wave. In this case, in consideration of the fact that the track of the train is approximately linear, a high directivity antenna is installed in the direction parallel to the track in the vicinity of the upper part of the track (such as a pillar). If radio signals are transmitted and received facing each other in the front direction, the line-of-sight environment can be stably maintained for a long time. The channel information between the antenna elements changes very rapidly because the moving direction of the train and the incoming direction of radio waves are almost parallel. On the other hand, the relative relationship (complex phase difference) of the channel information of each path between each antenna element is almost the same as a plane wave, so there is almost no change with time. That is, although the channel itself fluctuates at high speed, the transmission / reception weight itself of the virtual transmission line corresponding to the first singular value changes extremely slowly, and it is almost ideal even if the constant transmission / reception weight value is kept for a certain period Directivity can be maintained.

  On the other hand, the virtual transmission line corresponding to the second singular value or later actively uses many reflected waves, and incident waves from various directions which are not parallel to the moving direction dominate It becomes a target. At this time, if waves from various angular directions are combined, the situation is clearly different from that of plane waves, and the relative relationship of the channel information corresponding to each path between each antenna element is also significantly changed with time fluctuation. Do. Therefore, the transmission / reception weights need to be updated at high speed in order to follow changes in the relative relationship of channel information, and appropriate directivity can not be formed unless the update can be performed at high speed. Here, even if implicit feedback is introduced to reduce the overhead of feedback of channel information, it is necessary to exchange training signals for channel estimation extremely frequently to follow this rapid channel information fluctuation. It is difficult to achieve the accuracy that is practically required.

  Here, various circuits (high power amplifier, low noise amplifier, filter, mixer, etc.) for transmission and reception included in the terminal station apparatus mounted on the train and the base station apparatus at the wireless entrance are complex that changes due to temperature characteristics etc. It is assumed that it is possible to handle the rotation amount of the phase as almost constant without time fluctuation by calibration processing with high accuracy. In this case, if attention is paid to a certain train, the movement route of the train (strictly speaking, each antenna element of the terminal station device mounted on the train) exhibits one-dimensional displacement with extremely high accuracy if the passing line is determined. If the coordinates on the one-dimensional movement route can be grasped, the channel information in the case where there is a train at the same place each time is almost the same channel every time.

  Of course, assuming, for example, 80 GHz, the wavelength is 3.75 mm, and a deviation of coordinates of only 1 mm results in an error of 25% or more of the wavelength, and absolute channel information changes greatly. It is relative channel information required when calculating transmission and reception weights. In the case of a line-of-sight wave coming in a plane wave, as long as the plane and the plane on which the antenna element is two-dimensionally arranged are almost parallel, the relative channel information is not greatly affected by the error of that degree. Generally, regarding the fluctuation of the transmission / reception weight, the transition in the vertical direction is compared with the transition in the direction perpendicular to the line-of-sight direction (the direction parallel to the straight line connecting the transmission / reception antennas) and the transition in the direction parallel to the line of sight. In this case, the amount of fluctuation of the transmission and reception weights is larger. Of course, the play between the track and the wheel of the train may be several mm or more, so it does not necessarily pass exactly the same coordinates, but with some error even in the direction perpendicular to the line of sight. Become. However, as described above, in the case of the virtual transmission line corresponding to the first singular value, it can be approximated by a plane wave, and if it is an error of that degree, the change in transmission / reception weight is within the allowable range. On the other hand, when transitioning in a direction parallel to the line-of-sight direction, the accuracy of the determination of the position of the train is low due to the high moving speed. However, in the case of plane waves, the transmission and reception weights are completely insensitive to the case where trains transition in the vertical direction, and can be sufficiently coped with.

  Although the transmission and reception weights themselves may have small time variations, the complex phase of the signal synthesized using the transmission and reception weights fluctuates with the coordinates, and is observed as a Doppler shift. However, since the Doppler shift itself is essentially the same as the frequency error, it does not reach the orthogonality of OFDM if it is within a small value compared to the subcarrier spacing even when using OFDM Therefore, it is possible to cope with the residual frequency error compensation by performing tracking processing or the like.

  Therefore, first, a mechanism for determining the position (coordinates) of a train in one-dimensional train movement with high accuracy is introduced. For example, a mark for specifying a certain coordinate is disposed at a predetermined interval on the side of the track, the mark is photographed by a camera or the like, and the image is measured on the image to measure the coordinate. For example, markers of geometric coordinate information in the form of barcodes are arranged at intervals of 10 [m], and an image captured by a camera is analyzed to specify coordinates of the markers from the barcode. Although there is a time lag between the time when the mark actually passes and the time when the coordinates of the mark can be specified by image analysis, it is possible to obtain in advance how long this time lag will be. On the other hand, since this time lag is estimated to be substantially the same for all the marks, if the marks are placed at an interval of 10 [m], the time difference to recognize the marks by image analysis It is possible to estimate the instantaneous moving speed with high accuracy from the relationship of moving distance of 10 [m]. Furthermore, from the traveling speed and the time lag mentioned above, it is possible to estimate with high accuracy which location the train is actually at the moment when the coordinates of the mark are determined, and also the coordinates when a certain minute time has passed from there. Can be estimated.

  In this way, it is possible to accurately obtain instantaneous coordinates in one-dimensional movement. Here, although the bar code is shown as an example, an optical signal by an LED or the like may be used, or the coordinates may be specified using an electromagnetic wave signal such as a wireless tag. Although the coordinate estimation accuracy of GPS at the present time is not very high, when GPS correction value information is acquired via a wireless line (different frequency bands or via different systems may be separately set as Assisted GPS) The accuracy of GPS positioning is known to be reduced to about 5 to 10 [m]. Furthermore, in the case of a train, the travel route is one-dimensional in order to travel on the track, and even if GPS or the like is used, the accuracy is further improved if limited to the location on the one-dimensional route (on the track). . In the future, it is said that coordinate identification using a quasi-zenith satellite is possible with an accuracy of about several centimeters, and it is naturally possible to use such future GPS functions. Furthermore, it is also possible to acquire the movement distance from the number of rotations of the wheel, as with a car speedometer or distance meter, and perform high-precision coordinate correction when stopping at the station's home, etc. The measurement accuracy is dramatically improved by using a combination such as using the travel distance of the vehicle and GPS information or other coordinate acquisition means in combination. This makes it possible to estimate coordinates with high accuracy without using a camera and an image recognition function as described above. The present embodiment is equally applicable to acquiring coordinate information by any of these means.

  Although the case where the coordinates are estimated on the train side has been described in the above description, it is also necessary to specify the coordinates of the train that is the other party of communication on the base station apparatus side as well. Basically, the base station apparatus acquires the coordinates of the train using the coordinate specifying means of the traveling train connected via the base station apparatus and the wired network. In this way, when specifying the coordinates of the train on the wired network side, for example, the light emitter and its light receiver may be arranged along the track, and the light interception may be used when passing the train. A light emitter may be installed on the train side, and the light from the light emitter may be detected by a light receiver on the side of the line. Although it is unrealistic in terms of cost to install a large number of cameras along the track, it is possible to obtain the coordinates with high accuracy by other methods. Alternatively, if the base station device side can be notified of coordinate information acquired by the train via a low delay separately set wireless line (it may be a different frequency band or via a different system), the wired network It is also possible to specify the coordinates of the train on the base station apparatus side without using the acquisition means of coordinate information connected in the above manner.

  Next, in a state in which coordinates of such a train can be identified, for example, while running an actual vehicle in a time zone out of the normal service such as midnight or before service start, the channel matrix at that time is It is associated with the coordinate information, acquired in advance, and stored in the base station apparatus or the terminal station apparatus. If the measurement is performed several times and the average value is calculated, the correspondence between the channel matrix and the coordinates of the train can be obtained with higher accuracy. When obtaining this average value, instead of absolute channel information as measured, relative channel information based on the complex phase of the reference antenna is used, and transmission / reception weights are calculated for this relative channel information. Do. The measurement of the channel matrix is also performed by the base station apparatus on the wired network side in addition to the train side. Alternatively, uplink channel information may be obtained from downlink channel information by a technique such as implicit feedback.

  As described above, before the actual service operation starts, a database of the relationship between train coordinates and uplink and downlink channel matrices is constructed in advance. Specifically, based on the channel matrix, the transmission and reception weights of the virtual transmission path corresponding to the first singular value are calculated in advance, and the transmission and reception weights are stored in the database. For example, on the train side, the transmission weight vector used in the uplink of each virtual transmission path and the reception weight vector used in the downlink are stored in the database in association with discrete coordinate information. Similarly, on the base station side, the reception weight vector used in the uplink of each virtual transmission path and the transmission weight vector in the downlink are stored in the database in association with discrete coordinate information.

  FIG. 36 shows the outline of the configuration of the wireless communication system in the fourth embodiment. In the figure, reference numeral 600 is a base station apparatus, reference numerals 601-1 to 601-8 are first signal processing units of the base station apparatus, reference numerals 602-1 to 602-8 are geometrical coordinate information, and reference numeral 603 is a train. Reference numeral 604 denotes a terminal station apparatus, reference numeral 605 denotes a camera, reference numeral 606 denotes a weight matrix / coordinate database on the train side, and reference numeral 607 denotes a weight vector / coordinate database on the base station apparatus side. When any one of the first signal processing units 601-1 to 601-8 is indicated, the first signal processing unit 601 is described.

  In the train 603, a terminal station apparatus 604 provided with a large number of antenna elements is installed above the leading end of the leading car in the traveling direction or at the upper end of the trailing end of the last car. The installation location here may be configured to transmit and receive signals from inside the train 603 via the window, or may be installed outside the train 603 and covered with a cover (redome) to further reduce the influence of rain wind. good. The base station apparatus 600 further includes a plurality of first signal processing units 601-1 to 601-8 such as the first signal processing units 304-1 to 304-4 shown in FIG. 16 in the first embodiment. The first signal processing units 601-1 to 601-8 are physically spaced apart and installed. For example, a large number of first signal processing units 601-1 to 601-8 are provided at intervals of 1 m or 2 m. The base station apparatus 600 also includes the second signal processing unit 305 illustrated in FIG. 16 in addition to the first signal processing units 601-1 to 601-8. The first signal processing units 601-1 to 601-8 included in the base station apparatus 600 are installed, for example, on a wire post or the like provided to cross the line in a direction perpendicular to the line of the train 603. . Similar to the first signal processing units 304-1 to 304-4 shown in FIG. 16, the first signal processing units 601-1 to 601-8 have a large number of antenna elements and a circuit for signal processing.

  Similarly, the terminal station device 604 comprises a large number of antenna elements. Each of the antenna elements of the terminal station apparatus 604 and the antenna elements of the first signal processing units 601-1 to 601-8 have a configuration in which the directivity is improved in the forward direction as seen from the antenna elements to secure the line gain. The movement of the train 603 may not be a perfect straight line but may involve a curve, so it is not a good idea to have an excessively high directional gain, and the base station 600 looks at the terminal station 604 installed in the train 603. It is ideal that the antenna element has high directivity gain within a range in which the directivity gain does not decrease with respect to the existing direction and the direction in which the terminal station device 604 of the train 603 exists from the base station device 600. is there.

  In the example shown in FIG. 36, the camera 605 is mounted on the train 603, the geometric coordinate information 602 on the side of the track is photographed, and the device for grasping the coordinates of the train 603 by the image analysis function not specified 603 is provided. Further, the base station apparatus 600 includes a weight vector / coordinate database 607, and the terminal station apparatus 604 of the train 603 includes a weight matrix / coordinate database 606. Geometrical coordinate information 602-1 to 602-8 are installed on the side of the track at substantially equal intervals, and are arranged at positions where photographing can be performed by the camera 605 mounted on the train 603. In the geometrical coordinate information 602-1 to 602-8, coordinate information indicating the coordinate at which the geometrical coordinate information 602-1 to 602-8 is installed or the geometrical coordinate information 602-1 to 602-8. The information which can identify can be recorded by a bar code or the like.

  For example, when the train 603 travels, the camera 605 captures geometric coordinate information 602-1 to 602-8 with the camera 605 and recognizes an image of the geometric coordinate information 602-7, the geometric coordinate information The coordinates of the geometrical coordinate information 602-7 are specified from the information of 602-7. In practice, if there is a time lag in image recognition, the time lag time ΔT is grasped at the time of factory shipment or product design. The geometrical coordinate information 602-1 to 602-8 are sequentially image-recognized by the camera 605, so the geometrical coordinate information 602-8 is recognized at a certain time t 1 [s] and another time t 2 [s] If recognition of the geometrical coordinate information 602-7 is performed by the following equation, v = x / with respect to the distance x [m] between the geometrical coordinate information 602-8 and the geometrical coordinate information 602-7. The second velocity [m / s] is grasped at (t2−t1), and at time t2 + Δt when minute time Δt has elapsed from time t2, only v × ΔT [m] ahead of the coordinate of the geometrical coordinate information 602-7 Judged as being located in

  On the other hand, prior to the start of service operation, while the train 603 travels on a predetermined track, the training signal transmitted from each of the first signal processing units 601-1 to 601-8 of the base station apparatus 600 is used as a terminal station. Each antenna element of apparatus 604 receives and obtains channel information of its MIMO channel. Based on the acquired channel information, reception weight vectors and transmission weight vectors used in communication using a virtual transmission path corresponding to the above-described first singular value are first signal processing units 601-1 to 601-8. Are acquired and recorded in a weight matrix / coordinate database 606. When the reception weight vector and the transmission weight vector are acquired a plurality of times, it is possible to improve the accuracy by obtaining an average value thereof or the like. In addition, in this averaging, for example, the transmission weight vector of the antenna element included in the terminal station apparatus 604 is determined based on the complex phase of the transmission / reception weight vector of a predetermined antenna element in the antenna element included in the terminal station apparatus 604. It is preferable to perform processing capable of averaging even information acquired at different times, such as using the respective components of the above as relative transmission and reception weights.

  A similar database is also created on the base station apparatus 600 side and stored in the weight vector / coordinate database 607. In the above description, the case where the training signal is transmitted from the base station apparatus 600 side has been described, but the coordinates of the train 603 and time information are recorded on the train 603 (or terminal station apparatus 604) side by some method described later A training signal is transmitted from the terminal station apparatus 604 side, received by each antenna element of the first signal processing units 601-1 to 601-8 of the base station apparatus 600, and the channel information of the MIMO channel together with the reception time information Record. Based on the recorded time information, train coordinate information, and the relationship between time information and channel information, reception for a predetermined train coordinate used in communication using a virtual transmission path corresponding to the above-described first singular value The weight vector and the transmission weight vector are obtained for each of the first signal processing units 601-1 to 601-8, and are stored in the weight vector / coordinate database 607. Alternatively, as long as downlink channel information can be acquired on the side of the train 603, uplink channel information can also be acquired using a calibration technique, so the training signal transmitted from the base station apparatus 600 side is used. After acquiring one kind of channel information on the train 603 side, uplink channel information is acquired off-line by calibration processing, and based on the channel information, the first signal processing units 601-1-601- The transmission weight vector and the reception weight vector may be calculated every eight, and this may be converted into a database and recorded in the weight vector / coordinate database 607.

  In the above process, information acquired on the base station apparatus 600 side and information acquired on the terminal station apparatus 604 side should be comprehensively processed and recorded in the weight matrix / coordinate database 606 and weight vector / coordinate database 607. In order to generate information, necessary information is temporarily collected in an environment such as another computer, and the information subjected to processing is separately stored in weight vector / coordinate database 607 of base station apparatus 600 and terminal station apparatus 604. The configuration may be such that the weight matrix / coordinate database 606 is recorded.

  Here, the interval of the coordinates corresponding to the transmission / reception weight vector in the weight matrix / coordinate database 606 and the weight vector / coordinate database 607 matches the interval of the position where the geometrical coordinate information 602-1 to 602-8 is installed. There is no necessity. As described above, coordinate information can be acquired from the moving speed of the train and the elapsed time even at any position between the geometrical coordinate information 602-1 to 602-8, and the channel information acquired at the arbitrary position Since it is also possible to calculate the transmission / reception weight vector at any position based on, the interval of the position (coordinate) corresponding to the transmission / reception weight vector in weight matrix / coordinate database 606 and weight vector / coordinate database 607 is arbitrary. It is possible to set to

  In the weight matrix / coordinate database 606 of the terminal station apparatus 604 and the weight vector / coordinate database 607 on the base station apparatus 600 side, transmission / reception weight vectors for discrete coordinates are recorded, and for continuous data transmission / reception. It is necessary to obtain the transmission / reception weight vector between the coordinates corresponding to this discrete transmission / reception weight vector. However, if the step size of the coordinates corresponding to the transmission / reception weight vector / matrix recorded in weight matrix / coordinate database 606 and weight vector / coordinate database 607 is sufficiently short, transmission / reception weight vector / matrix corresponding to each successive coordinate If the difference of is sufficiently small, it will not cause much problem even if the same transmission / reception weight vector / matrix is continuously used until the next transmission / reception weight vector / matrix is recorded. If there is no restriction on the storage capacity of weight vector / coordinates database 606 and weight vector / coordinates database 607, it is possible to continue to use the same transmission / reception weight vectors / matrices until such newly recorded coordinate points. Preferably, the transmission / reception weight vector / matrix of the coordinate information is recorded in the weight vector / coordinate database 606 and the weight vector / coordinate database 607. However, when the storage capacity is limited, for example, interpolation is performed in a section sandwiched between the coordinates of two points using transmission / reception weight vectors / matrices corresponding to the coordinates of two consecutive points, for example, Alternatively, the transmission and reception weight vectors / matrices in the section may be calculated.

  When the train 603 is moving at high speed, a Doppler shift occurs in the frequency of the received signal observed by the base station apparatus 600 and the terminal station apparatus 604. Therefore, the channel information measured in advance before the start of service operation in the above description naturally includes the influence of the Doppler shift that depends on the moving speed of the train 603. In this Doppler shift, wavelength shift occurs with frequency offset, so the value of the channel information itself varies depending on the speed of the train 603. In normal train 603 operation, it is expected to travel at the same speed at the same location, but in fact there is no denying the possibility of slowing down due to some operating trouble, and in such a situation the train It is necessary to avoid deterioration in the communication quality of the entrance line, which causes problems when the user in 603 accesses the Internet.

  However, the antenna elements of the first signal processing units 601-1 to 601-8 of the base station apparatus 600 and the antenna elements of the terminal station apparatus 604 respectively correspond to the first singular value. When using a virtual transmission line, it is assumed that the antenna element is concentrated in a very narrow area. In this case, the path length difference of the first signal processing units 601-1 to 601-8 of the base station apparatus 600 and the antenna elements of the terminal station apparatus 604 has a small value. In addition, the viewing directions of the respective antenna elements are the same with very high accuracy. In this case, the amount of Doppler shift generated in each antenna element is substantially the same value, and the complex phase difference of the relative channel state between the antenna elements is a value with high accuracy and not dependent on the moving speed of the train 603. . That is, although channel information (and channel vectors having channel information as components) takes different values due to the moving speed of the train 603, the transmission / reception weight vector corresponding to the channel vector does not depend on the moving speed of the train 603. It will be. Therefore, it is possible to calculate the moving speed of the train 603 by the method described above, and it is also possible to estimate the amount of Doppler shift from the relation between the moving speed and the coordinates and to correct the channel vector for any moving speed. In the wireless communication system according to the fourth embodiment, although it is possible from the viewpoint of the fourth embodiment, which actively uses the virtual transmission path corresponding to the first singular value without performing correction processing up to that point, It is possible to use the transmission / reception weight vector made into a database as it is without the correction of.

(Outline of Signal Processing in Wireless Communication System of Fourth Embodiment)
An outline of signal processing performed in the wireless communication system of the fourth embodiment will be described below with reference to the drawings. The training signal used here may be, for example, a training signal of an OFDM signal from which channel information of each frequency component can be acquired, or a training signal with small autocorrelation to calculate time correlation. Also, even when using an OFDM signal as a training signal, it is not necessary to always provide a guard interval as in a general OFDM signal, and it is possible to use a training signal in which each frequency component does not have a guard interval. is there. This is because absolute channel information (or channel matrix, channel vector) is not necessarily required in calculation of transmission and reception weight vectors, and it is sufficient if only relative relationships of complex phases between antenna elements can be obtained. This is because there is no necessity to detect the symbol timing of the OFDM signal with high accuracy if the phase relationship with respect to any one reference antenna element can be acquired. Also, even in the case of OFDM signals, there is no need to be a training signal using all subcarriers used as described later, and the number of subcarriers assigned as training signals is limited to increase transmission power per subcarrier. The training signal may be used in a state where the SNR on the receiving side is increased.

<When transmitting a training signal from the base station device side>
FIG. 37 is a diagram showing an outline of signal processing in a train moving cell using a plurality of virtual transmission paths corresponding to the first singular value in the fourth embodiment. Here, an example is shown in which a training signal is transmitted from the base station apparatus to the terminal station apparatus side and the coordinate information and channel information are recorded on the terminal station apparatus side. Also, although not explicitly described in the following description, in the case of a wideband system such as in the case of using OFDM, for example, it is necessary to obtain channel information for each subcarrier and to calculate transmission / reception weights. However, in order to simplify the description, the description and detailed explanation of subscripts and the like regarding subcarriers etc. will be omitted.

  FIG. 37 (a) is a flowchart showing processing performed by the base station device 600. When the base station device 600 starts processing, the base station device 600 transmits training signals continuously (step S3701). The base station device 600 terminates the process after continuously transmitting the training signal for a predetermined period. The base station device 600 detects, for example, that the train 603 exists in the area of the base station device 600, and continuously transmits training signals over the period in which the train 603 exists in the area. The terminal station device 604 included in the train 603 receives a training signal transmitted from the base station device 600. The detection of the train 603 by the base station device 600 is performed based on the operation information of the train 603 or the like, or performed by some other method.

  FIG. 37 (b) is a flowchart showing processing performed by the terminal station device 604 in the train 603. When the processing is started, the terminal station device 604 acquires coordinate information by any method such as analysis of a captured image captured by GPS or the camera 605 while traveling (step S3711), and arbitrary in accordance with acquisition of coordinate information. Time information is also acquired by a method (step S3712). The terminal station apparatus 604 associates and stores the coordinate information acquired in step S3711 and the time information acquired in step S3712 corresponding to each coordinate information (step S3713), and ends the processing. The time acquisition method in this process may use, for example, information from GPS or other macro radio communication system, or implement a clock whose time has been set in advance, and use the clock epoch. It may be recorded as it is. Here, when the coordinate information and the time information are associated and recorded in step S3713, not the time information after the time lag for analyzing the coordinate information but the raw information including the coordinate information before the time lag occurs (for example, the camera Time information at the time of acquisition of the image and GPS signal), and when the time information and coordinate information after analysis are recorded together, the time lag relates to the time lag when processing the information in the offline processing described later. There is no need to make corrections. Note that the correspondence between the time information and the coordinate information does not have to be recorded by the terminal station device 604 of the wireless communication system, and it is possible to record the time information and the coordinate information even in another completely independent system. For example, any device may be used.

  FIG. 37 (c) is also a flowchart showing processing performed by the terminal station device 604 in the train 603. The terminal station apparatus 604 performs the process shown in FIG. 37 (c) in addition to the process shown in FIG. 37 (b). When the process is started, the terminal station apparatus 604 receives a training signal (step S3721), and acquires time information in accordance with the reception of the training signal (step S3722). The terminal station apparatus 604 performs received signal processing on the received training signal and performs channel estimation (step S3723). As received signal processing here, for example, the received signal is amplified by a low noise amplifier, converted from a radio frequency to a baseband signal through a mixer, and a sampling signal on the time axis is generated by A / D conversion, OFDM In the case of the modulation method, FFT processing is performed at predetermined symbol timing, and channel estimation is performed by comparison with a known training signal. If the training signal is a continuous signal not including a guard interval, FFT processing may be performed at any timing. Also, the channel estimation result may be divided by the channel estimation result of the reference antenna element or its complex phase, and converted to relative channel information actually required for calculation of the transmission / reception weight vector.

  The terminal station apparatus 604 stores the channel information acquired in step S 3723 in association with the time information acquired in step S 3722 (step S 3724), and ends the processing. Here, the timing at which the channel information can be acquired in step S3723 is different from the time when the training signal was received in step S3721, but there is a time lag, but the time information acquired in step S3722 becomes the time when the training signal is received in step S3721 If managed in a similar manner, even if a time lag occurs by the time the channel information can be acquired in step S3723, it is possible to eliminate the time lag from the time information recorded in association with the channel information in step S3724. In the description, it is assumed that the acquisition time of channel information does not include a time lag. Note that since the above-described signal processing and channel estimation can be separately performed as offline processing by changing the time, raw sampling data may be recorded as it is. However, in the following description, it is assumed that the channel estimation result (relative channel information) is recorded for convenience.

  FIG. 38 is a diagram showing an outline of off-line signal processing performed after acquiring coordinate information and channel information in the fourth embodiment. Here, as in FIG. 37, an example of off-line processing is shown in which a training signal is transmitted from the base station apparatus 600 to the terminal station apparatus 604, and coordinate information and channel information are recorded on the terminal station apparatus 604 side. There is. This off-line signal processing is performed by a signal processor. The signal processing apparatus may be provided in the terminal station apparatus 604, or may be provided as an independent apparatus capable of acquiring coordinate information, time information, and channel information acquired by the terminal station apparatus 604.

  When the off-line signal processing is started, the signal processing device reads out the data of the combination of the recorded time information and the coordinate information (step S3801), and the difference between the coordinates with the coordinate information corresponding to the previous time series. The (moving distance) is calculated (step S3802). The signal processing apparatus also calculates an elapsed time as a difference between time information and time information corresponding to the previous time (step S3803). The signal processing device calculates the moving speed by dividing these differences (step S3804), and from the time lag required to acquire coordinate information actually measured separately, the distance traveled by the train 603 during that time lag is The coordinate information is calculated by adding only the amount of movement (step S3805). Here, the time lag means, for example, the time required for the image processing etc. when acquiring the coordinate information by image processing of the video taken by the camera, etc., and the actual image acquisition and the coordinate information are recorded. It corresponds to the time until Since the acquisition time of the channel information and the acquisition time of the coordinate information do not necessarily coincide with each other, the signal processing device converts the correspondence between the time and the coordinates from the relationship between the coordinate information, the time information, and the moving speed here. Formulas are extracted and managed (step S3806). However, when the time information and the coordinate information are correlated and recorded in the form of excluding the time lag in the above-described pre-processing, the correction process regarding the time lag in step S3805 can be omitted.

  On the other hand, when the signal processing apparatus reads out the combination of channel information and time information in parallel with the processing from step S3801 to step S3806 (step S3807), it estimates coordinate information from the time information according to the above conversion formula (step S3808). The signal processing apparatus calculates a channel matrix from the channel information (step S3809), and performs singular value decomposition on the channel matrix (step S3810). The signal processing device records the first left singular vector obtained by the singular value decomposition in step S3810 as the reception weight vector of the terminal station apparatus 604 in association with the coordinate information (step S3811), and the first obtained by the singular value decomposition The right singular vector is recorded as the transmission weight vector of the base station 600 in association with the coordinate information (step S3812).

  Further, the signal processing apparatus performs calibration processing on the first left singular vector (reception weight vector) and the first right singular vector (transmission weight vector) calculated in step S3810 (step S3813). The signal processing apparatus records the vector obtained by the calibration process for the reception weight vector as the transmission weight vector of the terminal station 604 in association with the coordinate information (step S3814), and obtains the calibration process for the transmission weight vector. The received vector is recorded as the received weight vector of the base station 600 in association with the coordinate information (step S3815). The above processing is performed for each virtual transmission path corresponding to each first singular value related to the first signal processing units 601-1 to 601-8 on the base station apparatus 600 side, and information recording is performed on each of them. Is done.

  In the processing so far, processing for each virtual transmission line corresponding to each first singular value related to all the first signal processing units 601-1 to 601-8 on the base station apparatus 600 side has been described. In a situation where the distance between the base station apparatus 600 and the terminal station apparatus 604 changes every moment by the movement of the first signal processing unit 601-according to the specific distance as shown in the second embodiment. The spacing of 1 to 601-8 can not be optimized, and as a result, not all virtual transmission paths are generally orthogonal. Therefore, the signal processing apparatus selects a part of efficient virtual transmission paths among the first signal processing units 601-1 to 601-8 on the base station apparatus 600 side with a slightly redundant number (step S3816). ). Data transmission between base station apparatus 600 and terminal station apparatus 604 is performed using the transmission / reception weight vector corresponding to the virtual transmission path selected in step S3816.

For example, when the target spatial multiplexing number is N _SDM , the correlation among the reception weight vectors on the terminal station apparatus 604 side corresponding to the virtual transmission path related to the first signal processing unit 601 on the base station apparatus 600 side is Choose a combination of N _SDM receive weight vectors with small _{か ら} from a plurality of recorded receive weight vectors. This selection is performed, for example, by searching for a combination of N _SDM reception weight vectors for which the correlation value is minimum from a plurality of recorded reception weight vectors. As an example of this search method, when, for example, N _SDM reception weight vectors are selected, among the correlation values calculated for each combination of two reception weight vectors included in the N _SDM reception weight vectors. The maximum correlation value may be used as the correlation value in the combination of the N _SDM reception weight vectors, and the combination of the reception weight vectors that minimizes the correlation value may be selected. The correlation value calculated for the combination of two reception weight vectors here corresponds to the inner product value of the normalized vector. Note Besides, for example, the correlation values in N _SDM number of receiving weight vectors, Shun power sum of the absolute value of the correlation value calculated for each combination of the two receiving weight vectors included in N _SDM number of receiving weight vector , You may select a combination that minimizes the power sum.

  In this way, a combination of reception weight vectors with small correlation is selected based on some criteria, and the combination information (that is, this is among all the first signal processing units 601-1 to 601-8 on the base station apparatus 600 side). The transmission / reception weight vector for the terminal station apparatus 604 on the train 603 side and the base station apparatus 600 side in correspondence with the above-mentioned transmission / reception weight vector, which corresponds to the combination of which first signal processing unit is selected and communicated. And transmit / receive weight vectors and combination information in the first signal processing units 601-1 to 601-8 to be used are recorded and managed. The recording management interval is a constant value between small coordinates where the time variation of the transmit / receive weight vector due to the time variation of the channel can be ignored, and the coordinate information and the transmit / receive weight vector information are each approximately significant difference You do not care about record management.

In the above description, the number of virtual transmission paths used, that is, the number of space multiplexes is fixed as N _SDM , but this number is variable, and the correlation value (standard The maximum number of virtual transmission paths may be utilized in the range in which the maximum value of the inner product value of the converted vector falls below a predetermined threshold value. Similarly, the maximum number of virtual transmission paths may be utilized within the range in which the power sum of the absolute values of the correlation values is less than a predetermined threshold. In the case of a wideband system, since there is no necessity that the multiplexing number for each subcarrier is constant, it is possible to manage as different multiplexing numbers for each subcarrier. In these cases, together with the information on the transmission / reception weight vector to be made into a database, the number of these multiplexes and which virtual transmission path related to the first signal processing unit 601 will be managed.

  In this case, the communication control circuit provided in the base station apparatus 600 refers to the weight vector / coordinates database 607 and instructs the first signal processing unit used for communication in the coordinates for each coordinate to instruct the signal transmission / reception It is good also as composition to carry out. Also, the coordinate information used in the database here does not have to represent the two-dimensional or three-dimensional geographical coordinates, but it is one-dimensional like the distance from an arbitrary starting point on the track. Any information may be used as long as the information can be matched with the channel information.

  Furthermore, although the previous coordinate information and the previous time information are required here, with regard to the top data of the database, the moving speed information can not be acquired because there is no previous coordinate and time information. However, if there is overlap between the area covered by one base station apparatus 600 and the area covered by the adjacent base station apparatus 600, and in fact, if data acquisition is performed before the service area is switched, There is no problem in ignoring the information at the beginning of the recorded data. For example, in the case of the base station apparatus 600 including the station home in the service area, there is no problem in discarding the head data if data acquisition is performed while the station home is stopped. Alternatively, it is possible to obtain the moving speed etc. directly by measuring the number of rotations of the train wheels etc. In this case, it is possible to directly obtain the moving speed without calculating the difference between the coordinates and the time. It is also possible to record the obtained value as the moving speed at the time of measurement of the channel information and use this value. In this way, by analyzing the acquired data offline, it becomes possible to process and prepare necessary information.

<When transmitting a training signal from the terminal station device side>
FIG. 39 is a diagram showing an outline of another signal processing in a train moving cell using a plurality of virtual transmission paths corresponding to the first singular value in the fourth embodiment. In the signal processing shown in FIG. 39, unlike the signal processing shown in FIG. It shows signal processing in the case of recording channel information.

  FIG. 39A is a flowchart of processing performed by the terminal station device 604. When the terminal station device 604 starts processing with the operation of the train 603, it transmits training signals continuously (step S3901). While the terminal station apparatus 604 is collecting channel information between the terminal station apparatus 604 and the base station apparatus 600 before the start of service operation, the terminal station apparatus 604 ends the process after continuously transmitting training signals.

  The terminal station apparatus 604 performs the process of FIG. 39 (b) in parallel with the process of FIG. 39 (a). When the terminal station apparatus 604 starts the process of collecting channel information before the start of service operation, the terminal station apparatus 604 acquires coordinate information and time information by some means (steps S3911, S3912). The terminal station apparatus 604 records data in which the acquired coordinate information is associated with time information (step S3913), and ends the process.

  On the other hand, when detecting that the train 603 is present in the area of the base station apparatus 600, for example, the base station apparatus 600 starts the process of FIG. (Step S3921). The base station device 600 acquires time information as well as reception of the training signal (step S3922). After the reception signal processing similar to the reception signal processing in the terminal station apparatus 604 described above, the base station apparatus 600 performs channel estimation based on the training signal (step S3923). The base station device 600 associates the obtained channel information with the time information and records them (step S3924), and ends the processing. The detection of the presence of the train 603 in the area of the base station device 600 is performed using some method such as a method using operation information.

  These processes are simultaneously performed in parallel by all the first signal processing units 601-1 to 601-8 on the base station apparatus 600 side. The time information may be acquired by each of the first signal processing units 601-1 to 601-8, or the entire base station apparatus 600 including the plurality of first signal processing units 601-1 to 601-8. , And may be notified of time information to all the first signal processing units connected by wire. Alternatively, the channel information acquired by the plurality of first signal processing units 601-1 to 601-8 is aggregated in the base station apparatus 600 via wire communication, and the time information acquired by the base station apparatus 600 and the plurality of first signal processing The channel information from the units 601-1 to 601-8 may be associated and recorded / managed.

  FIG. 40 is a diagram showing an outline of another offline signal processing performed after acquiring coordinate information and channel information in the fourth embodiment. Here, similarly to FIG. 39, a training signal is transmitted from the terminal station apparatus 604 to the base station apparatus 600 side, coordinate information is acquired on the terminal station apparatus 604 side, and channel information is recorded on the base station apparatus 600 side An example of offline signal processing is shown. The basic process shown in FIG. 40 is the same as the signal process shown in FIG. The processing from step S3801 to step S3810, the processing in step S3813, and the processing in step S3816 are the same. Here, duplicate explanations are omitted, and steps S4011, S4012, S4014, and S4015 will be described.

  The process shown in FIG. 40 is a process based on the training signal in the uplink, and the process shown in FIG. 38 is a process based on the training signal in the downlink. Due to this difference, the first left singular vector is used as the reception weight vector of the terminal station 604 on the side of the train 603 in step S3811 in FIG. 38, whereas the first left singular in step S4011 in the process shown in FIG. A vector is used as a reception weight vector of the base station apparatus 600. In addition, while the first right singular vector was used as the transmission weight vector of the base station apparatus 600 in step S3812, the first right singular vector is used as the transmission weight vector of the terminal station apparatus 604 on the train 603 side in step S4012. And

  Also, the handling of the first right singular vector obtained by singular value decomposition in step S4010 of FIG. 40 and the vector obtained by the calibration process for the first left singular vector are also different. In step S 4014 in FIG. 40, the first right singular vector, that is, the vector obtained by the calibration process for the reception weight vector of base station apparatus 600 is used as the transmission weight vector of base station apparatus 600. Further, in step S4015 in FIG. 40, the first left singular vector, that is, the vector obtained by the calibration process for the transmission weight vector of the terminal station apparatus 604 is used as the reception weight vector of the terminal station apparatus 604.

  In the above description of FIG. 38 and FIG. 40, the transmission weight vector and the reception weight vector on the base station apparatus 600 side and the transmission weight vector and the reception weight vector on the terminal station apparatus 604 side in each figure by calibration processing. Although the calculation was performed as a matter of course, it is possible to substitute for example only FIG. 38 or FIG. 40, and the processing after step S3813 in FIG. 38 and FIG. The process of obtaining the remaining transmission / reception weight vectors by calibration may be omitted in the processes after step S4013.

  In the above description, although the first signal processing units 601-1 to 601-8 are physically independent on the base station apparatus 600 side, they are described as transmission / reception weight vectors. As in the base station apparatus 600, independent first signal processing units 601-1 to 601-8 are not present on the side, so they are expressed as transmission / reception weight vectors in the description, but in actuality these are described It is correctly represented as a transmission / reception weight matrix composed of vectors, and in fact, it is represented as a weight matrix / coordinate database 606 of the terminal station apparatus 604. However, since this is not so essential in understanding the present embodiment, this point is expressed in accordance with the preceding and following explanations.

<Outline of signal processing during train operation>
FIG. 41 is a diagram showing an example of signal processing in the case of acquiring a transmission / reception weight vector from coordinate information during service operation in the wireless communication system of the fourth embodiment. Here, the transmission and reception weight vectors require both transmission weight vectors and reception weight vectors, and both the base station apparatus 600 side and the terminal station apparatus 604 side need to acquire transmission and reception weight vectors, and this processing flow is as follows. It represents the process common to both. Of course, the weight vector / coordinates database 607 of the base station 600 and the weight matrix / coordinates database 606 of the terminal station 604 used in this case are different, and the method of acquiring coordinate information of the train 603 is the train 603 side. And the base station apparatus 600 side. For example, when acquiring the coordinate information of the train 603 on the base station apparatus 600 side, a plurality of sensors may be provided beside the track as described above to detect the position, or the coordinate information acquired on the train 603 side It does not matter as a structure transmitted via the radio link (It may be a different frequency band like a macro cell, or may be via different systems) set separately. Even if the coordinate information of the train 603 is acquired by any of these methods, for example, if there is a difference in the time lag of information acquisition, any time that can adjust the time lag and estimate the coordinate information at the time of information acquisition with high accuracy Even if a method is used, it is possible to use essentially in the present embodiment as well.

  First, the current coordinate information of the train 603 is acquired by any method in any of the terminal station 604 or the base station 600 on the train side 603 (step S4101). Also, time information at that time is acquired (step S4102). The difference between the previous coordinate information and the previous time information is taken (steps S4103 and S4104), the movement speed is calculated by dividing it (step S4105), and the current coordinates are corrected by taking into consideration the processing time lag. Is calculated (step S4106), and the corresponding transmission / reception weight vector is read out from the weight matrix / coordinate database 606 or the weight vector / coordinate database 607 using the coordinate information as an argument (step S4107). The above-described processes of steps S4101 to S4107 are repeatedly performed each time acquisition of coordinate information of a train or at a predetermined cycle, and transmission / reception weight vectors / matrices of that moment are always acquired.

  When receiving a signal in parallel with the above processing, virtual transmission paths corresponding to the plurality of first singular values (the plurality of first signal processing units 601-1 to 601-8 of the base station apparatus 600) The reception weight vector of the uplink or downlink transmission path corresponding to one of the above is sequentially acquired by step S4107 (step S4111), and the reception signal processing is performed in symbol units at the time of signal reception (step S4112), For example, in the case of an OFDM signal, the signal on the frequency axis separated into each frequency component is multiplied by the reception weight vector for each frequency component (step S4113), and the reception signal on each virtual transmission line is extracted individually . Here, although signals from a plurality of virtual transmission paths are extracted, if it is necessary to separate these signals, the received signals on the respective virtual transmission paths are obtained using a training signal provided at the head of a series of received signals. Channel estimation is performed on the channel matrix, and signal detection is performed by arbitrary signal detection processing using the channel matrix of the MIMO channel obtained by this channel estimation (step S4114), and the above processing is repeated if there is subsequent data ( Step S4115: NO, if the data is completed (step S4115: YES), the reception process is ended.

  Similarly, when transmitting a signal in parallel with the above processing, virtual transmission paths corresponding to a plurality of first singular values (corresponding to a plurality of first signal processing units of the base station apparatus 600) Of the transmission weight vector read in step S4107 (step S4121), and generation of the transmission signal to be transmitted on each virtual transmission line (step S4122), with the transmission weight vector for each virtual transmission line The multiplication is performed (step S4123), the transmission signal processing of the signal is further performed (step S4124), the process is continued if the transmission data remains (step S4125: NO), and the process ends when the data ends ( Step S4125: YES).

  The above description collectively describes the processing performed by the base station apparatus 600 and the terminal station apparatus 604, but the details are slightly different. For example, in the base station apparatus 600, signal processing on one virtual transmission line is performed for each of the plurality of first signal processing units 601-1 to 601-8, but the terminal station apparatus 604 is used for all virtual transmission lines. The signal processing for the signal is performed in parallel in the terminal station device. For example, at the time of received signal processing, each received signal received by each antenna element is multiplied by different received weight vectors, and signal detection processing of MIMO transmission is performed on a plurality of signal sequences obtained as a result. Also, at the time of transmission signal processing, individual transmission weight vectors are multiplied for each signal sequence of each virtual transmission path. However, in the terminal station apparatus 604, in actual transmission signal processing, the signal components related to all the virtual transmission paths are added and synthesized for each antenna element for the signal multiplied by the transmission weight vector, and transmission signal processing is performed on the synthesis result. Will be given. Further, in base station apparatus 600, a first signal used for transmission / reception of an actual signal in first signal processing sections 601-1 to 601-8 installed redundantly with respect to the actual number of spatial multiplexing. The processing unit manages which one of the first signal processing units 601-1 to 601-8, and in transmission and reception, the communication control circuit or the like refers to the database and the corresponding first signal processing unit Signal processing instructions are issued, and various signal processing is performed in each of the first signal processing units. With the exception of such slight differences, both the base station apparatus 600 and the terminal station apparatus 604 can explain the signal processing flow in this processing flow.

(Selection of the first signal processing unit of a train moving cell using a plurality of virtual transmission paths corresponding to the first singular value)
Here, although the eight first signal processing units 601-1 to 601-8 are shown in the configuration of the wireless communication system according to the present embodiment shown in FIG. It does not necessarily mean to do. For example, when the transmission weight vectors corresponding to the respective first signal processing units 601-1 to 601-8 recorded in the weight vector / coordinate database 606 on the side of the train 603 and the reception weight vectors show high correlation. Means that the communication between another first signal processing unit 601 and the terminal station apparatus 604 interferes with the communication between a certain first signal processing unit 601 and the terminal station apparatus 604. The example shown in FIG. 29 in the second embodiment shows the optimum condition of the spacing of the antenna element (corresponding to the first signal processing unit) on the base station apparatus side by design. The distance L between the station apparatus and the terminal station apparatus is included. That is, in the wireless communication system of the present embodiment in which the distance between the base station apparatus and the terminal station apparatus changes with the movement of the train 603, the first signal processing unit always satisfies the relationship of equation (29). It is impossible to install 601.

Therefore, a situation may occur where the correlation between the transmission and reception weight vectors corresponding to the individual first signal processing units 601 is high. In such a case, since efficient spatial multiplexing can not be performed, it is used when simultaneously performing spatial multiplexing transmission by using a combination of transmission weight vectors or reception weight vectors with low correlation among a large number of first signal processing units 601. A predetermined number of first signal processing units 601 are selected. This selection can also be performed in advance before starting service operation, for example, if the target number of spatial multiplexing is 4, among 70 patterns of ₈ C ₄ combinations, each transmission / reception weight It is a combination pattern in which the maximum value of the absolute value of the inner product of vector normalized vectors (the absolute value of six inner products is calculated by ₄ C ₂ ) is the smallest, or the power sum of the absolute value of the inner product is the minimum The spatial multiplexing transmission is performed using the transmission / reception weight vector which is a combination pattern of the above and combinations having relatively low correlation. The combination pattern of the first signal processing unit 601 used in the combination is also recorded on the weight vector / coordinate database 607 side of the base station apparatus 600 side, and the combination pattern of the recorded first signal processing unit 601 is used. You may communicate.

  In addition, since a wide band communication is normally performed, the optimum combination pattern of the first signal processing unit 601 is different for each frequency component to be used, so the combination pattern of the first signal processing unit 601 is different for each subcarrier. May be used. Alternatively, the same combination pattern of the first signal processing unit 601 may be used for all subcarriers, and in this case, for example, the absolute value of the inner product of each subcarrier may be the inner product of all subcarriers. Alternatively, a combination pattern of transmission / reception weight vectors that minimizes the maximum absolute value may be selected, or a combination pattern of transmission / reception weight vectors that minimizes the sum of all frequency components of the power of absolute value of each inner product may be selected. It is good.

(Base station apparatus configuration example of a train moving cell using a plurality of virtual transmission paths corresponding to the first singular value)
FIG. 42 is a block diagram showing a configuration example of a base station apparatus 600 in the fourth embodiment. The base station device 600 includes an interface circuit 77, a MAC layer processing circuit 78, a second transmission signal processing unit 71, first transmission signal processing units 181-1 to 181-4, and first reception signal processing. The unit 185-1 to 185-4, the second received signal processing unit 75, the communication control circuit 614, the weight vector / coordinate database 607, the coordinate information acquisition circuit 612, and the time information acquisition circuit 613.

  Also in the radio communication system of the fourth embodiment, in order to use spatial multiplexing using virtual transmission paths corresponding to a plurality of first singular values, the base station apparatus 600 in FIG. The configuration is the same as that of the base station apparatus 70 shown, but the following configuration is different. The base station apparatus 600 in the fourth embodiment includes a weight vector / coordinate database 607, a coordinate information acquisition circuit 612, and a time information acquisition circuit 613, and a communication control circuit 614 instead of the communication control circuit 120. Are different from the base station device 70.

  The weight vector / coordinate database 607 stores the transmission / reception weight vector associated with the coordinate information acquired as described above. When some of the plurality of first signal processing units 601 included in the base station apparatus 600 are selected and used, information indicating the first signal processing unit 601 to be used in addition to the transmission / reception weight vector Are associated with each other and stored in the weight vector / coordinate database 607. Each of the plurality of first signal processing units 601 includes one each of a first transmission signal processing unit 181 and a first reception signal processing unit 185.

  The coordinate information acquisition circuit 612 acquires coordinate information indicating the position of the running train 603 by the means described above. The time information acquisition circuit 613 acquires time information indicating the current time. The communication control circuit 614 reads out from the weight vector / coordinate database 607 the transmission / reception weight vector corresponding to the coordinate information acquired by the coordinate information acquisition circuit 612. The communication control circuit 614 outputs the transmission weight vector among the read transmission / reception weight vectors to the first transmission signal processing units 181-1 to 181-4 via the second transmission signal processing unit 71. The communication control circuit 614 instructs the first transmission signal processing units 181-1 to 181-4 to transmit using the transmission weight vector. The communication control circuit 614 may output the transmission weight vector directly to the first transmission signal processing units 181-1 to 181-4 without passing through the second transmission signal processing unit 71. Further, the communication control circuit 614 outputs the reception weight vector among the read transmission and reception weight vectors to the first reception signal processing units 185-1 to 185-4 via the second reception signal processing unit 75. The communication control circuit 614 instructs the first reception signal processing units 185-1 to 185-4 to perform reception using the reception weight vector. The communication control circuit 614 may output the reception weight vector directly to the first reception signal processing units 185-1 to 185-4 without passing through the second reception signal processing unit 75.

  Therefore, the first transmission weight processing unit 130 included in the first transmission signal processing units 181-1 to 181-4 in the fourth embodiment is the first transmission shown in FIG. 18 in the first embodiment. Like the first transmission weight processing unit 130 included in the signal processing unit 181, it is necessary to include the first channel information acquisition circuit 131, the first channel information storage circuit 132, and the first transmission weight calculation circuit 133. Instead, the transmission weight vector instructed from the communication control circuit 614 is output to the first transmission signal processing circuit 111. Similarly, the first reception weight processing unit 160 provided in the first reception signal processing units 185-1 to 185-4 in the fourth embodiment is the same as the first reception weight processing unit illustrated in FIG. 19 in the first embodiment. There is no need to include the first channel information estimation circuit 161 and the first reception weight calculation circuit 162 as in the first reception weight processing unit 160 included in the reception signal processing unit 185, and an instruction from the communication control circuit 614 The received weight vector is output to the first received signal processing circuit 158.

  However, if the base station 600 acquires channel information before starting service operation (corresponding to FIG. 39), it is necessary to perform channel estimation on the received signal, so the first reception weight processing is performed. The unit 160 may be configured to include the first channel information estimation circuit 161. Similarly, in order to calculate the reception weight vector, the first reception weight processing unit 160 may be configured to include the first reception weight calculation circuit 162. However, since the channel estimation and the calculation of the reception weight vector can also be performed by off-line processing as pre-processing before the start of service operation, it is not necessary to have a configuration that implements them. As described above, if the first reception weight processing unit 160 implements only the first channel information estimation circuit 161, the first reception weight processing unit 160 uses the first channel information estimation circuit 161. The estimated channel information is output to the communication control circuit 614. The communication control circuit 614 records the channel information acquired from the first reception weight processing unit 160 in association with the time information acquired in the process of step S3922 shown in FIG. 39 (c). In FIG. 42, it is assumed that this recording function is also implemented in the communication control circuit 614, and it is not shown explicitly.

  When the first reception weight processing unit 160 does not include the first channel information estimation circuit 161, the first reception weight processing unit 160 uses the acquired sampling value of the training signal for channel estimation. It outputs to the communication control circuit 614. The communication control circuit 614 may record the sampling information acquired from the first reception weight processing unit 160 in association with the time information acquired in the process of step S3922 shown in FIG. 39C. . In this case, for example, instead of the processing of step S4007 of FIG. 40, sampling data may be read out and processing of channel estimation may be performed.

  The time information acquisition circuit 613 is required when acquiring channel information in advance before starting service operation, so this function can be omitted in actual service operation. In addition, since the communication of the train moving cell is basically point-to-point communication, the scheduling processing circuit 781 can be omitted. However, since terminal station apparatus 604 is mounted on a plurality of trains with respect to fixedly installed base station apparatus 600, the information recorded in weight vector / coordinate database 607 may be slightly different for each of the trains. Therefore, in this case, the weight vector / coordinate database 607 records a database concerning all the trains that may pass the place, and the scheduling processing circuit 781 refers to the operation information etc. It is also possible to have a function of managing whether it is a passing timing.

  In addition, although it was decided to record various information acquired in the communication control circuit 614 in the acquisition of channel information etc. before the start of service operation, it goes without saying that such information is recorded in other circuits. The information may be read out via the communication control circuit 614.

(Example of terminal station apparatus configuration of a train moving cell using a plurality of virtual transmission paths corresponding to the first singular value)
FIG. 43 is a block diagram showing a configuration example of the terminal station apparatus 604 in the fourth embodiment. The terminal station device 604 includes an interface circuit 67, a MAC layer processing circuit 68, a transmission unit 61, a reception unit 65, a communication control circuit 624, a weight matrix / coordinate database 606, a coordinate information acquisition circuit 622, time And an information acquisition circuit 623.

  Also in the wireless communication system of the fourth embodiment, the terminal station apparatus 604 is shown in FIG. 20 in the first embodiment in order to use spatial multiplexing using virtual transmission paths corresponding to a plurality of first singular values. Although the configuration is the same as that of the terminal station device 60, the following configuration is different. The terminal station apparatus 604 in the fourth embodiment includes a weight matrix / coordinate database 606, a coordinate information acquisition circuit 622, and a time information acquisition circuit 623, and a communication control circuit 624 instead of the communication control circuit 121. Are different from the terminal station device 60.

  The weight matrix / coordinate database 606 stores the transmission / reception weight matrix (vector) associated with the coordinate information acquired as described above. As for base station apparatus 600, in the case where some of the plurality of first signal processing units 601 provided in base station apparatus 600 are selected and used, the first to be used in addition to the transmission / reception weight vector Although information indicating the signal processing unit 601 is associated with coordinate information and stored in the weight matrix / coordinate database 606, the terminal station apparatus 604 selects the physically different first signal processing unit 601. Since it does not exist, the information of the vector constituting the transmission / reception weight matrix includes which first signal processing unit 601 in the base station apparatus 600 is to be used, and explicitly which first signal processing is performed It is not necessary to record whether to use the part 601. However, the combination of the first signal processing unit 601 actually used in the combination of the first signal processing unit 601 assumed to be used by the base station apparatus 600 and the transmission / reception weight matrix assumed to be used by the terminal station apparatus 604 Must be consistent with each other.

  The coordinate information acquisition circuit 622 acquires coordinate information indicating the position of the running train 603 by the means described above. The time information acquisition circuit 623 acquires time information indicating the current time. The communication control circuit 624 reads from the weight matrix / coordinate database 606 the transmission / reception weight matrix associated with the coordinate information acquired by the coordinate information acquisition circuit 622. The communication control circuit 624 outputs the read transmission weight matrix to the transmission unit 61, and causes the transmission unit 61 to perform transmission using the transmission weight matrix. At the time of reception, the communication control circuit 624 outputs the read reception weight matrix to the reception unit 65, and causes the reception unit 65 to perform reception using the reception weight matrix.

Therefore, the first transmission weight processing unit 140 included in the transmission unit 61 in the fourth embodiment is the same as that of the first transmission weight processing unit 140 included in the transmission unit 61 shown in FIG. 21 in the first embodiment. Similarly, the channel information acquisition circuit 141, the channel information storage circuit 142, and the first transmission weight calculation circuit 143 do not necessarily need to be provided (if implemented, they will be used only in pre-processing before the start of service The transmission weight vector (the column vector forming the transmission weight matrix) instructed from the communication control circuit 624 is output to the transmission signal processing circuit 811-1 to 811-N _SDM , which is not required in the normal operation. Similarly, the first reception weight processing unit 44 provided in the reception unit 65 in the fourth embodiment is the first reception weight processing unit 144 provided in the reception unit 65 shown in FIG. 22 in the first embodiment. As described above, the reception weight vector instructed from the communication control circuit 624 is output to the reception signal processing circuit 145 during service operation.

  However, when the terminal station apparatus 604 on the train 603 side acquires channel information before the start of service operation (when the process shown in FIG. 37 is performed), the terminal station apparatus 604 selects a channel based on the received training signal. Since it is necessary to perform estimation, the first reception weight processing unit 144 may be configured to include the first channel information estimation circuit 146. However, since the channel estimation and the calculation of reception weight vectors (matrices) can be performed even by off-line processing, it is not necessary to necessarily have a configuration for implementing these. As described above, when the first reception weight processing unit 144 includes only the first channel information estimation circuit 146, the first reception weight processing unit 144 is estimated by the first channel information estimation circuit 146. The channel information is output to the communication control circuit 624. The communication control circuit 624 records the channel information acquired from the first reception weight processing unit 144 in association with the time information acquired in the process of step S3722 shown in FIG. 37 (c). In FIG. 43, it is assumed that this recording function is also implemented in the communication control circuit 624, and is not shown explicitly. In the first embodiment, with regard to another configuration example of the receiving unit 65 of the terminal station device 60 shown in FIG. 23 and another configuration example of the transmitting unit 61 of the terminal station device 60 shown in FIG. Except for the difference, it is possible to cope with the same configuration.

  As described above, the terminal station apparatus 604 included in the train 603 moving at high speed when traffic of information terminals and the like used by a user who rides the train 603 is aggregated and offloading from the macro cell is attempted at the wireless entrance. Channel time variation between the base station apparatus 600 and the base station apparatus 600 becomes a problem. Therefore, in the wireless communication system according to the fourth embodiment, in an environment in which the incoming direction of radio waves and the moving direction of the train 603 are substantially the same, the time variation of the transmission line corresponding to the first singular value tends to be smaller Use for Specifically, a plurality of first signal processing units 601 are installed in the overhead wire column, while the relationship between coordinate information and channel information is acquired in advance and made into a database, and weight matrix / coordinate database 606 and weight vector / The coordinate database 607 is recorded and used. The terminal station apparatus 604 provided in the train 603 sequentially acquires its own coordinates, reads out the transmission / reception weight matrix from the weight matrix / coordinate database 606 using this as an argument, and enables stable communication to enable communication. It is possible to increase the capacity. Similarly, the base station device 600 is stabilized by sequentially acquiring coordinate information of the train 603 and reading out the transmission / reception weight vector used by the first signal processing unit 601 from the weight vector / coordinate database 607 as an argument. Communication can be enabled to enable a large capacity of communication lines.

  In the wireless communication system of the present embodiment, each of the first transmission signal processing units 181 multiplies the transmission weight vector by the baseband signal obtained by performing the modulation processing and the like in the second transmission signal processing unit 71. Generate components of the signal vector. The signal vector formed of the components of the signal generated in each of the first transmission signal processing units 181 is transmitted from the antenna element provided in the first transmission signal processing unit 181. The component of the signal vector transmitted from each antenna element is the component of the signal vector obtained by multiplication with the transmission weight vector corresponding to the antenna element.

  In the wireless communication system of the present embodiment, each of the first reception signal processing units 185 acquires baseband signal vectors from the reception signals received by the plurality of provided antenna elements, and acquires the acquired baseband signal vectors and reception weights. A signal sequence of one system is acquired by multiplying by a vector. The second received signal processing unit 75 reproduces the data transmitted from the terminal station apparatus from each of the signal sequences of one system acquired by the first received signal processing units 185.

  In the radio communication system of the fourth embodiment, the base station apparatus 600 spatially multiplexes data sequences # 1 to #L of a plurality of systems with the terminal station apparatus 604. May be transmitted to and received from the terminal station device 604.

  When the antenna element included in the first transmission signal processing unit 181 is a single polarization antenna element, the first transmission signal processing unit 181 may transmit a signal sequence of one system. . When the antenna element included in the first transmission signal processing unit 181 is a plurality of types of polarization antenna elements, and a plurality of types of polarization antenna elements are used simultaneously, the first transmission signal processing unit 181 includes one system. Only one or up to two signal sequences may be transmitted. In addition, when the antenna element included in the first received signal processing unit 185 is a single polarization antenna element, the first received signal processing unit 185 acquires a baseband signal of only one system from the received signal. You may do it like this. When the antenna element included in the first reception signal processing unit 185 is a plurality of types of polarization antenna elements and a plurality of types of polarization antenna elements are used simultaneously, the first reception signal processing unit 185 includes one system. Only one or up to two base band signals may be acquired.

Fifth Embodiment
[About time axis beamforming]
(Summary of basic principles)
First, consider two antenna elements (a first antenna element and a second antenna element) having a distance of d [m]. Assuming that the transmission signal from the communication partner station has very strong directivity and is in the line-of-sight environment, the signals arriving at the two first and second antenna elements can be approximated by plane waves, and the first, The received signal at time t of the second antenna element is represented by φ ₁ (t) and φ ₂ (t). Assuming that the signal component at time t of the k-th frequency component is S _k (t), the distance between the transmitting station and the first antenna element is L, the center frequency is f _c , and the frequency bandwidth is W and f _c Assuming that the frequency of the k-th frequency component between W / 2 and + W / 2 is f _k and the wavelength of the k-th frequency component in the radio frequency is λ _k , the reception signal φ ₁ (t) is expressed by equation be able to.

Here, the direction of arrival and an angle theta, (the c here the speed of light) the received signal phi _{2 (t)} is the received signal phi _{1 (t)} is (dsin) / c in the case where it can approximate plane wave only delayed Since it can be considered that the signal arrives, the received signal φ ₂ (t) can be approximated as shown in equation (39).

  Here, Δt represents a sampling interval Δt [s] (= 1 / W). The coefficient α is expressed by equation (40) using the bandwidth W [Hz] of the signal, the angle θ in the incoming direction, the antenna element interval d [m], and the speed of light c [m / s].

Comparing this equation (38) with the equation (39), weights (coefficients) w _k ^{(freq-domain)} on the frequency axis shown in equation (41 ⁾ are multiplied in each frequency component k of equation (39) There is.

Since this coefficient is a coefficient on the frequency axis, if it is converted to a coefficient on the time axis by IFFT, the weight (coefficient) of the n-th (0 ≦ n ≦ N _FFT −1, N _FFT is the number of FFT points) on the time axis w _n ^{(time-domain)} is expressed by equation (42).

Here, a value representing the ratio of the sum of the square of the absolute value of the time axis weight at the beginning n = 0 and the square value of the absolute value of the weight from n = 1 to N _FFT −1 in [dB] Is expressed by the following equation (43) as a function F (α) of the coefficient α.

  FIG. 44 is a graph showing the function F (α) in the fifth embodiment. In the figure, the vertical axis indicates the value of the function F (α), the relative power ratio, and the horizontal axis indicates the value of α. The function F (α) corresponds to the ratio of the received power of the leading wave to the received power of all delay waves other than the leading wave. From the figure, it can be seen that when α is sufficiently small, almost one component of the preceding wave occupies most of the entire power, and conversely, the delay wave component increases as α increases.

Further, as can be seen from the equation (40), although depending on the bandwidth W, in general, if the antenna element distance d is sufficiently narrow, α has a small value, and thus the received signal can be sufficiently narrowed An approximate value of φ ₂ (t) can be obtained by multiplying the value of the reception signal φ ₁ (t) by the coefficient of the preceding wave component. That is, since weight w _k ^{(freq-domain)} on the frequency axis is different for each frequency component, conventionally, weight is multiplied after the signal on the time axis is converted to the signal on the frequency axis by FFT. The However, if the antenna element spacing is narrowed to make the coefficient α smaller so that the function F (α) becomes equal to or greater than a predetermined value (eg, 30 [dB]) in FIG. it is possible to calculate as only the receiving signal phi _{1 (t)} from the received signal phi _{2 (t)} the equation multiplied (44).

Although the relationship between the received signal φ ₁ (t) and the received signal φ ₂ (t) is shown here, assuming that N antenna elements are disposed at an interval d in a linear array, the m-th antenna element The received signal φ _m (t) can be approximated by equation (45) using the received signal φ _m-1 (t) of the adjacent (m-1) antenna element.

That is, the reception signal φ _m (t) of the _m- th antenna element can be approximated by equation (46) using the reception signal φ ₁ (t) of the first antenna element.

However, _cm in equation (46) is given by equation (47).

  Here, the mathematical meanings of the above description are organized using Equation (41) and Equation (43). First, the situation where energy concentrates on one time axis component when performing IFFT or inverse Fourier transform means that a delta function behavior is seen on the time axis. It is known that Fourier transform of the delta function results in a constant on the frequency axis. Conversely, if IFFT-appearing components appear on the frequency axis, mathematical descriptions can be made with a single time-axis component with high precision on the time-axis.

The first half of the right side of the equation (41) is a constant which does not depend on the frequency component, and focusing on the exponent of the natural logarithm of the second half, it is −2πjf _k αΔt. The frequency f _k of the kth frequency component takes on a value between −W / 2 and + W / 2, and the sampling interval Δt is 1 / W, so f _k Δt has a value of −1⁄2 to 1/2. take. The complex phase of Exp (-2πjf _k αΔt) is -απ to + απ, and if α is sufficiently small, the complex phase is approximately 0, and Exp (-2πjf _k αΔt) can be approximated by a constant of 1. That is, if α is sufficiently small, w _k ^{(freq-domain)} has negligible frequency dependence. If α is large to some extent, the frequency dependency can not be ignored, and the frequency component obtained after inverse Fourier transform deviates from the delta function type behavior, and the delay wave component can not be ignored.

Here, in order to discuss the value of the coefficient α quantitatively, a frequency bandwidth of 1 GHz is assumed in the 80 GHz band. The wavelength is 3.75 mm, and antenna elements are arranged at intervals of 5 mm, and d = 0.005. Assuming that the angle range of the incoming wave is ± 10 degrees, the value of α is 0.005 × sin (10 °) × 10 ⁹ / (3 × 10 ⁸ ) = 0.002894 from the equation (40). This is 1.04 degrees at an angle, and it can be seen that this is a constant having almost no frequency dependency. That is, it can be understood that the value of the function F (α) in the equation (43) becomes a very large value in the equation (43) if the frequency dependency is small, and the function is more delta function-like.

Thus, in a situation where the coefficient α is sufficiently small and Exp (−2πjf _k αΔt) can be approximated by 1, equation (42) becomes a non-zero value only when n = 0, and equation (48) And so on.

  Substituting this into equation (47), the coefficient of the m-th antenna element is expressed by equation (49).

  In this way, if the relationship of the complex phase in the reception signal of each of the N antenna elements can be acquired, the reception weight at the time of combining the reception signals of the respective antenna elements can be determined as in equation (50).

  Note that although the reception weight vector of this combination is the weight of maximum ratio combination, it is expected to be approximately the same reception level in the line-of-sight environment, so it is the weight of equation (51) of equal gain combination. It is good.

That is, the signal from the target transmitting antenna element is extracted by multiplying the receiving signal vector of each antenna element by the receiving weight vector of (̃w ₁ , ̃w ₀ ,..., ̃W _N ). Is possible.

  Here, α in the equation (40) is a coefficient depending on the path length difference, and in the case of a linear array, it can be expressed in the form of the equation (49), and also in the case of other antenna arrangements. If geometric features are used, such reception weights can be described in the same procedure.

Note that this coefficient _cm can also be determined directly instead of using equation (47). In particular, in the case other than the linear array, the periodic training signal transmitted from the transmitting station is acquired over one period, and the training signal is It is also possible to calculate it as Formula (52) based on a cross correlation. However, j = m (j, m is a value indicating the number of the antenna element, and is expressed as j instead of m).

Here, S _j (n) represents the n-th sampling value of the j-th antenna element. Here, since signal processing is performed for the signal of one antenna element pair, it is not necessary to use the value in one symbol as it is for the line gain, but using the average value of many times It is desirable to increase the SNR value at For example, if averaging of 100 symbols is performed, an SNR improvement of 20 dB can be expected, so that averaging may be performed in accordance with the required channel estimation accuracy. In the averaging here, it is also preferable to use relative channel information with reference to the complex phase of the antenna element serving as a reference. By the way, in high frequency bands such as millimeter waves, the effect of phase noise becomes a problem, but since the randomness of phase can be expected to some extent, the effect of phase noise is also suppressed by averaging processing here. It is possible to obtain the correlation value in the form.

  Here, the above-mentioned training signal may be a signal with a guard interval of OFDM, and may be a continuous signal without a guard interval. In order to obtain absolute channel information, it is necessary to accurately obtain symbol timing and measure the channel, but for calculation of transmission and reception weights, complex phase is required with respect to the reference antenna element. This is because it is only necessary to relatively know what kind of relationship it is. Also, other training signals may be used as long as they have low temporal autocorrelation.

  The value of the reception weight may be determined by equation (50) or equation (51) based on the coefficient determined in this manner. For the reception weight determined in this way, for example, as reception signal processing, the signal sequence ^ S (n) on the time axis of one system may be calculated by equation (53).

Here, in the equation (53), ̃w ₁ is 1, and in the equation (53), a process of adding ̃w _j S _j (n) from j = 2 to N ′ _BS-Ant to S ₁ (n) It is carried out. The signal sequence ^ S (n) of one system obtained in this manner is regarded as a signal received by one virtual receiving antenna for receiving the signal of the virtual transmission channel corresponding to the first singular value. , Receive signal processing.

  In general, directivity can be directed in the direction of angle θ by combining signals with a complex phase difference of 2π (d · sin θ / c) / λ with respect to the direction of angle θ for narrow band signals. However, for a wide band signal having a bandwidth of 1 GHz, it is necessary to change the complex phase difference for each frequency component. In addition, in order to achieve signal separation with sharp directivity with a narrow beam width, it is preferable to widen the antenna element distance, but in this case, the complex phase difference for each frequency component is made larger (that is, frequency dependency is enhanced). It appeared as an effect. Therefore, in general, communication is performed by widening the antenna element distance to widen the antenna aperture length and narrowing the directional beam while forming directivity stably over a wide band with different weights for each frequency component I was aiming for that. However, if directivity can be narrowed by increasing the number of antenna elements, the antenna aperture length is rather narrowed, and delays other than the leading wave are caused by the evaluation function F (α) shown in equation (43). If the antenna element interval and the range of the direction θ of the incoming wave are limited (that is, the value of α is limited) so that the contribution of the wave component can be determined as a negligible level, By multiplying the received signal by a predetermined coefficient, a signal arriving from a predetermined direction can be efficiently extracted, and signal separation at the time of spatial multiplexing can be realized on the time axis using the coefficient.

  For example, when space multiplexing is performed using a plurality of virtual transmission paths corresponding to a plurality of first singular values as shown in FIG. 16 in the first embodiment, the equations (49) to (52) The coefficients corresponding to are calculated corresponding to individual virtual transmission paths, and the weight vector is calculated using equations (50) to (51) for that, to spatially multiplex a plurality of signal sequences. It is possible to calculate a time-based reception weight vector for the first stage signal separation.

Also, calibration processing similar to calibration processing on the frequency axis is also possible on the time axis, and if each reception weight vector is multiplied by the calibration coefficient on the time axis, the time axis for transmission is similarly obtained. It is also possible to obtain transmission weight vectors.

(Outline of residual interference removal method)
As described above, it is possible to extract the signal of the virtual transmission line corresponding to the target first singular value, but between virtual transmission lines that are not completely orthogonal to each other, Although weak, interference signals will leak. Below, the removal method of this interference signal is demonstrated. The following interference signal removal method corresponds to signal processing in the second received signal processing unit 75 in the base station apparatus. Also in the description of the first embodiment described above, interference signals that can not be completely removed by the first reception signal processing unit 185 are eliminated in two steps using the second reception signal processing unit 75. Is equivalent to its signal processing.

  In the following description, for example, as shown in FIG. 16 in the first embodiment, a plurality of first signal processing units provided with a plurality of antenna element groups are mounted on the base station apparatus side, and the terminal station apparatus side In this case, it is assumed that there is only one antenna element group including a plurality of antenna elements. FIG. 45 is a diagram for explaining a method of removing an interference signal in the fifth embodiment. As shown in the figure, the wireless communication system in the fifth embodiment includes a base station apparatus 204 and a terminal station apparatus 206. The base station apparatus 204 includes a plurality of first signal processing units, and the first signal processing unit includes a plurality of antenna elements. The plurality of antenna elements provided in each first signal processing unit form virtual antenna elements 201, 202, and 203, respectively. The terminal station device 206 includes an antenna element group 205 formed of a plurality of antenna elements. In the wireless communication system according to the fifth embodiment, the communication path between the base station apparatus 204 and the terminal station apparatus 206 is a communication path between the virtual antenna elements 201, 202, 203 and the antenna element group 205. It can be understood as a system to be

FIG. 45 (a) is a diagram showing interference components generated in the downlink in the wireless communication system of the fifth embodiment. The antenna element group 205 of the terminal station device 206 directs the virtual beam to the directional beam (transmission beam) directed to the terminal station device 206 formed by the virtual antenna element 201 of the base station device 204 Consider the case of receiving with a directional beam (received beam). In this case, the receive beam towards a virtual antenna elements 201 of the antenna element group 205, the signal transmitted from the virtual antenna elements 202 in addition to the signal [psi ₁ transmitted from virtual antenna elements 201 [psi _When a part of ₂ leaks in, consider removing a part of the signal ₂ as an interference component. This is equivalent to the removal of the interference component in the uplink shown in FIG. 45 (b), and can be processed in the same way in terms of signal processing. In the case of transmitting the signal light ₁ with the transmission beam directed to the virtual antenna element 201 which is a transmission beam formed in the antenna element group 205 of the terminal station apparatus 206 in the uplink, the virtual antenna element 201 is A part of the signal line ₂ transmitted to the virtual antenna element 202 by the antenna element group 205 leaks into the reception beam directed to the antenna element group 205 of the terminal station device 206, and the leaked signal line ₂ It is equivalent to removing a part of H as an interference component.

First, taking the downlink as an example, for the incoming wave from the direction of the angle θ toward the virtual antenna element 201 in the antenna element group 205, the reception weight vector shown by equations (48) to (52) etc. I will stand by. Here, the signal ψ ₁ from the virtual antenna element 201 is awaited, but the signal ψ ₂ from the virtual antenna element 202 in the direction of the angle θ ′ leaks into here. At this time, assuming that θ 'in equation (40) is substituted for θ' as a coefficient β, the coefficient α in equation (49) is replaced with a coefficient β for each antenna element of the antenna element group 205 of the terminal station device 206. The signal ψ ₂ is received by the factor. However, each element of the reception weight vector to be multiplied when the antenna element group 205 of the terminal station device 206 stands by with the directional beam directed to the virtual antenna element 201 is an equation in which the coefficient α is not replaced by the coefficient β. It is a receiving weight represented by Formula (50) which took the complex conjugate of (49). Therefore, the complex phase is not originally canceled to 0 by the multiplication of equation (49) and equation (50) in the multiplication of the reception weight. When the sum is calculated by multiplying coefficients m {{2πj f _c (α-β) x (m-1) Δt}} in the m-th antenna element constituting antenna element group 205, respective complex phases are synthesized with different complex phases. It will be done. It is to be noted that m = 1, 2 _,.

However, although a combination of different complex phases has been described here, since the combination is actually regular, it satisfies the “antenna arrangement condition for orthogonalization of line-of-sight MIMO transmission” described in the second embodiment. In the situation, this summed value converges to approximately zero. By the way, when the receiving weight shown in the equation (50) or the equation (51) is applied to the signal ψ ₁ from the virtual antenna element 201, all the antenna elements constituting the antenna element group 205 receive The complex phase has a constant value. This makes it possible to perform additive synthesis of signals in the same phase, the amplitude of the signal becomes N _{MT -Ant} (the number of antenna elements) times, and a large gain is obtained.

According to such a procedure, when directivity is directed to the j-th virtual antenna element of the base station device 204 on the receiving side, the signal from the i-th virtual antenna element of the base station device 204 is received You just need to figure out In the above-mentioned example, the signal representing how the signal ψ ₂ transmitted from the virtual antenna element 202 leaks in the state of waiting with the directional beam directed to the virtual antenna element 201 is the equation (49) Is obtained by replacing .alpha. With (.alpha .-. Beta.) And summing over all m. Since this sum is given as the sum of geometric series as in the left side of equation (54), it is expressed by the following formula by the formula of the sum of geometric series.

Here, when f _c (α-β) N _{MT -Ant} Δt is an integer, the sum of geometric series is zero and residual interference disappears, but in general residual interference will remain Therefore, it is necessary to suppress the interference component.

On the other hand, ψ ₁ (t) is transmitted by the directional beam directed from the virtual antenna element 201 to the antenna element group 205, and awaits by the directional beam directed from the antenna element group 205 to the virtual antenna element 201. In the signal ψ ₁ (t) received in the state, the relationship between the signal ψ ₁ (t) transmitted from the antenna element group 205 to the first virtual antenna element and the first virtual antenna element is The desired signal component is included as ^ h ₁₁ × ψ ₁ (t) using the coefficient ^ h ₁₁ shown. Similarly, the signal [psi _{j (t)} transmitted from the j virtual antenna elements, a signal ^ [psi _{i (t)} from the i-th virtual antenna element is received by the receiving beam toward the antenna element group 205 It can be expressed that the component of the signal ψ _i (t) is included by only h h _ij × ψ ₁ (t), using the coefficient h h _ij indicating the relationship between

However, the point to be noted here is that when the weight on the frequency axis of equation (41) does not have frequency dependency, only the time axis weight of equation (42) is nonzero n = 0, and all the others are not Although the above description can be made because it becomes zero, when the weight of equation (41) remains frequency-dependent, the term weight n of the time axis weight of equation (42) and subsequent terms are also included. It has a value, and as a result it can not be expressed in a simple expression so far. Therefore, if this signal is converted to a signal on the frequency axis to be understood, in general, both the coefficient h h _ij and the signal ψ ₁ (t) and the signal ψ _i (t) are decomposed into frequency components, For the kth subcarrier, using the coefficient ^ h _ij ^(k) , the signal ψ ₁ ^(k) (t) and the signal ^ ψ _i ^(k) (t), the signal ^ ψ _i ^(k) (t) Should be treated as if the components of the signal ψ ₁ ^(k) (t) are contained ^ h _ij ^(k) × ψ ₁ ^(k) (t).

However, when the weight on the frequency axis in equation (41) hardly has frequency dependency, the term weight n of the time axis weight in equation (42) approximates 0 with high precision. In this case, signal processing on the time axis becomes possible also for the subsequent processing. In this way, whether the signal processing on the frequency axis is necessary or the simple signal processing on the time axis can cope with it depends on the frequency dependency of the weight on the frequency axis of equation (41). It depends on various parameters of the target system. Therefore, when the frequency dependency of all the coefficients h _ij can be neglected in system design, and separation of each signal is possible by signal processing between each sampling value on the time axis, FIG. The state of communication shown can be expressed by equation (55). Here, M represents the space multiplexing number (virtual transmission path number) for convenience of notation.

This is the same format as the conventional signal processing of MIMO on the frequency axis, so for example, a signal vector (^ ψ ₁₎ observed as a slight interference signal leaks out by the inverse processing of this channel matrix or linear processing by MMSE. A signal vector (ψ ₁ (t), ψ ₂ (t),..., Ψ _M ) in which the interference component is canceled from (t), ^ ψ ₂ (t), ..., ^ ψ _M (t) (T)) can be determined. In this way, signals between multiple virtual transmission paths can be separated.

  The important point here is the intention of this equation, and time t in equation (55) is applicable to any time, and if this relationship is understood at any time as an instantaneous value, this is all This means that this relationship holds between sampling values at the same time with respect to sampling values. In other words, this means that signal separation can be performed with the sampling value as it is without performing FFT or the like after sampling.

  Therefore, in the above description, the explanation was made on the assumption that the OFDM modulation scheme was mainly applied, but a training signal is added to the beginning of the transmission signal, and equation (55) is given by some method using this training signal. If a corresponding channel matrix can be obtained in advance, signal separation can be performed on the time axis without performing FFT, and in the case of performing single carrier transmission, directly on the signal separated by equation (55), It is possible to perform signal processing of single carrier transmission.

  In general, phase noise can not be ignored in the case of high frequency bands such as millimeter waves, and when using a wide band OFDM signal, etc., the phase noise component breaks orthogonality between OFDM subcarriers by FFT processing, It is expected that the frequency components will be dispersed as noise and crosstalk between subcarriers, making it difficult to correct phase noise. In the case of single carrier transmission, it is possible to estimate and compensate the phase noise component sequentially from the relationship between the signal detected for each symbol and the received signal, and it is possible to cope with relatively large phase noise. However, in the prior art, when multiple series of single carrier transmission signals are combined by the spatial MIMO channel, signal separation can not be performed unless they are once converted into signals on frequency, and this is why Implementation of the FFT was essential. However, since the implementation of the FFT transforms the phase noise into an uncompensable state, a signal separation technique that can be realized without the implementation of the FFT has been necessary for the phase noise compensation.

  (55) for the MIMO channel on the time axis, in addition to the theoretical analysis operation as shown in equation (54), for example, using a training signal for one period equivalent to one OFDM symbol, correlation operation is performed. It is also possible to directly determine each component of the channel matrix. In signal processing for multiplying received signal vectors received by a plurality of antenna elements by received weight vectors, the SNR of the original signal was low, so some measure such as averaging was required. However, since the signal obtained by multiplying the reception weight vector for extracting the signal of the virtual transmission line corresponding to the first singular value is already improved in SNR by the directional gain, the reception weight vector is obtained therefrom. When performing the task of calculating 相関 by correlation, it is not necessary to average multiple times, and there is no problem even if it is acquired in one process using a training signal of one cycle equivalent to one OFDM symbol .

Moreover, although N _FFT number of sample points are used in this equation, since there is no necessity to use FFT in particular, N _{FFT in} equation (56) does not have to be the number of FFT points in OFDM, and it is arbitrary value It is also possible.

  The condition for approximation here is that the virtual transmission line corresponding to the first singular value is generally assumed to be such that interference components are suppressed by directivity control of both transmission and reception, in other words, a pair of MIMO channel When the absolute value of the angular component is relatively large compared with the absolute value of the non-diagonal component, approximation can be performed with high accuracy.

By using the coefficients ^ h _ij obtained in this way, the channel matrix of equation (55) is subjected to operations such as multiplying both sides by linear weights according to the ZF method, MMSE method, etc. If, for example, single-carrier signal processing is performed on the resulting time-axis signal, even if phase noise becomes a problem, high-accuracy signals can be obtained using existing single-carrier phase noise countermeasures. It becomes possible to carry out signal detection processing.

  In the above description, when the frequency dependence of the weight on the frequency axis of the equation (41) can be almost ignored, that is, n = 1 of the time axis weight of the equation (42) and the subsequent terms are 0 with high accuracy. Although the description has been made centering on the case where it can be approximated, when this condition is not satisfied, the FFT processing is performed straightly and the processing for signal separation is individually performed on the frequency axis.

(Process flow of time axis beam forming)
FIG. 46 is a flowchart of a process of acquiring transmit / receive weights of time axis beamforming in the wireless communication system according to the fifth embodiment. The flow chart shown in FIG. 46 (a) is a flow chart of the process of acquiring the transmission and reception weights of the first stage of time axis beamforming. Here, for example, not the method of obtaining the coefficient on the time axis from various parameters as shown in the equation (49) but the case of using the correlation coefficient as shown in the equation (52) will be described. However, these values are different only in the method of calculating the weight, and analytical methods can be applied similarly if the calculation method is replaced. Further, acquisition of transmission / reception weight vectors is similarly required on both the base station apparatus side and the terminal station apparatus side, and the processing described herein is to be carried out in both directions. Below, the process performed by a downlink is demonstrated.

  First, the base station apparatus transmits a training signal for channel estimation (step S4601). The training signal here is a signal for performing channel estimation and does not depend on the transmission scheme used in data communication after channel estimation. For example, it may be a continuous OFDM signal without a guard interval, or it may have less autocorrelation. It may be a training signal having a predetermined periodicity. As a matter of course, it is also possible to use the training signal used in the transmission method used in data communication as it is.

When the terminal station apparatus receives the training signal transmitted from the base station apparatus by each antenna element (step S4611), the received signal is individually down converted from the radio frequency to the baseband to perform A / D conversion. Record the sampling value after. The training signal is acquired in a predetermined cycle (for example, 100 cycles), and the signals of each cycle are added and combined in accordance with the periodicity to perform averaging (step S4612). Assuming that the averaged sampling value of the n-th sample of the j-th antenna element is S _j (n), the correlation coefficient c _j of the j-th antenna element with respect to the first antenna element is obtained by equation (52) (step S4613) ).

The reception weight of the j-th antenna element is calculated by taking the complex conjugate of the correlation coefficient c _j obtained in step S4613 (or further dividing the value by its absolute value), and combining these to obtain a reception weight vector (Step S4614). In particular, when spatial multiplexing is not performed, the reception weight vector is simply calculated in this manner on the directivity formation time axis. However, in the case of spatially multiplexing signals of a plurality of systems as shown in FIG. 16 shown in the first embodiment, the above-described reception weight vector for each of the first signal processing units 304-1 to 304-4 in FIG. Calculate

  If the calibration coefficient at the time of implicit feedback does not have frequency dependency, the calibration process is performed on the reception weight vector on the time axis using a common calibration coefficient in the frequency direction, The transmission weight vector is acquired (step S4615). The obtained transmission and reception weight vector is recorded and managed (step S4616), and the transmission and reception weight vector is used for subsequent signal transmission and reception. If the training signal can be received in a high SNR state by some method, the averaging process can be omitted. Also, even if the calibration coefficient has frequency dependency, if the deviation for each frequency is relatively small, the average value of the calibration coefficients for all subcarriers is regarded as the calibration coefficient common to all subcarriers. You may process it. In this manner, it is possible to obtain the first stage reception weight vector and transmission weight vector for extracting the virtual transmission path corresponding to each first singular value.

  FIG. 46 (b) is for acquiring the second stage (transmission) reception weight matrix for residual interference cancellation with respect to the signal of each virtual transmission path received by increasing the directivity gain as described above It is a flow chart which shows processing of. First, the base station apparatus transmits a training signal using the transmission weight vector obtained by the processing shown in FIG. 46 (a) (step S4621). The training signal here is a signal for performing channel estimation, and may not be dependent on a transmission scheme used in data communication after channel estimation.

  When the terminal station apparatus receives the training signal transmitted from the base station apparatus by each antenna element (step S4631), the received signal is individually down converted from radio frequency to baseband, and A / D conversion is performed. Get the sampling value by. The reception signal vector at each of these antenna elements is multiplied by the reception weight vector (step S4632). The cross-correlation of the training signal for each of the reception weight vectors thus acquired (that is, for each virtual channel corresponding to the first singular value) is acquired by equation (56) (step S4633), and a channel matrix is acquired. (Step S4634). The terminal station apparatus calculates a reception weight matrix based on the acquired channel matrix (step S4635), and records and manages the calculated reception weight matrix as a second stage time axis weight matrix (step S4636).

  In the case of a wireless entrance where both the base station apparatus and the terminal station apparatus are fixedly installed, the weights of the first and second stages hardly change in time, so before the start of service operation. If you acquire it in advance, you can continue to use it. On the other hand, for example, in a train moving cell etc., the channel fluctuates at high speed and time, but the reception weight vector of the first stage is acquired in advance and made into a database, so that each stage of communication There is no need to calculate the transmission / reception weight vector of the eye. However, with regard to the second stage reception weight matrix (vector), it is for removing interference signals remaining in different states depending on the imperfection of the first stage reception weight vector. These need to be updated sequentially. In this case, the processing from step S4631 to step S4636 may be performed each time the signal of the wireless packet is received.

  In addition, when using a plurality of virtual transmission paths corresponding to the first singular value in the access system, not only the second stage reception weight matrix (vector) but also the first stage transmission / reception weight vector is successively updated There is a need to. In the frame configuration shown in FIG. 65 described in the eighth embodiment described later, in the example shown in the eighth embodiment described later as a slot for accommodating a training signal for acquiring a transmission / reception weight vector therefor, Use the slot at the beginning of the frame and the four slots from the end of the frame. However, although it is possible to place training signals for calculating the second stage reception weight matrix in the above-mentioned slots, it is possible to use the subdivided sub-slots in the intermediate slots other than these slots. It is also possible to place it at the beginning (that is, at the beginning of the wireless packet) and calculate the second reception weight matrix based on this.

(Circuit configuration of time axis beam forming)
FIG. 47 is a diagram illustrating a configuration example of the first transmission signal processing unit 381-j included in the base station apparatus in the fifth embodiment. In addition, j shows the serial number in the some 1st transmission signal processing part 381 with which a base station apparatus is provided. As shown in the figure, the first transmission signal processing unit 381-j includes an IFFT & GI applying circuit 813-j, a first transmission signal processing circuit 311-j, and D / A converters 814-1 to 814-. N'BS _-Ant , local oscillator 815, mixers 816-1 to 816-N ' _BS-Ant , filters 817-1 to 817-N' _BS-Ant , high power amplifier (HPA) 818-1 818-N ' _BS-Ant , antenna elements 819-1 to 819-N' _BS-Ant, and a first transmission weight processing unit 330. The first transmission weight processing unit 330 includes a first channel information acquisition circuit 332, a first channel information storage circuit 333, and a first transmission weight calculation circuit 334. The configuration of the base station apparatus in the fifth embodiment has the same configuration as that of the base station apparatus 70 (FIG. 17) in the first embodiment, but is shown in FIG. 47 instead of the first transmission signal processing unit 181. The second embodiment differs in that the first transmission signal processing unit 381 is provided.

  When the first transmission signal processing unit 381-j in the fifth embodiment applies time-axis beamforming to the first transmission signal processing unit 181-j shown in FIG. 18 in the first embodiment The circuit configuration of The difference between the first transmission signal processing unit 381-j and the first transmission signal processing unit 181-j shown in FIG. 18 is that a large number of IFFT & GI addition circuits 813 are integrated into one, A first transmission signal processing circuit 311 and a first transmission weight processing unit 330 in place of the transmission signal processing circuit 111-j and the first transmission weight processing unit 130, and the IFFT & GI application circuit 813 is a first This is a point disposed in the front stage of the transmission signal processing circuit 311.

  Also, in the first transmission signal processing unit 181-j shown in FIG. 18, multiplication of the transmission weight vector on the frequency axis is essential, and so regardless of OFDM or SC-FDE, on the frequency axis As transmission signal processing, IFFT processing was essential in any arrangement. On the other hand, since the first transmission signal processing unit 381-j in the fifth embodiment can perform signal processing of directional beam formation in sampling data units on the time axis, the IFFT & GI addition circuit 813 can be used. It is also possible to omit the signal processing of pure single carrier. In this sense, the IFFT & GI applying circuit 813 shown in FIG. 47 is represented by a dotted square.

The operation of the first transmission signal processing unit 381-j included in the base station apparatus of the fifth embodiment will be described. When a digital baseband signal is input from the second transmission signal processing unit 71 to the first transmission signal processing unit 381-j, if it is a signal on the frequency axis, the IFFT & GI provision circuit 813-j performs the frequency conversion. The on-axis signal is converted to a on-time signal. In the case of a system in which sampling data on the time axis is input like a pure single carrier signal, the digital IFFT & GI application circuit 813-j is not used (the IFFT & GI application circuit 813-j is omitted), digital The baseband signal is input to the first transmission signal processing circuit 311-j. The first transmission signal processing circuit 311-j multiplies the signal sequence of one system by the transmission weight vector on the time axis for each sampling value, and each component of the transmission signal vector as a result is D / D for each antenna element system. A converter 814-1 to 814-N 'It outputs to _BS-Ant . The subsequent processing is equivalent to the processing performed in the first transmission signal processing unit 181-j shown in FIG.

  The transmission weight vector by which the signal sequence is multiplied in the first transmission signal processing circuit 311-j is the same as in the first transmission signal processing unit 181-j shown in FIG. It is acquired from the first transmission weight calculation circuit 334 provided in the transmission weight processing unit 330. In the first transmission weight processing unit 330, the first channel information acquisition circuit 332 acquires channel information acquired by the reception unit via the communication control circuit 120, and the first channel information storage circuit 333 Store and update sequentially. At the time of transmission, in accordance with an instruction from communication control circuit 120, first transmission weight calculation circuit 334 reads channel information corresponding to the destination terminal station apparatus from first channel information storage circuit 333 and uses the read channel information as a base. To calculate the transmission weight vector. The transmission weight vector can be calculated using any method as described above.

  Strictly speaking, the communication control circuit 120 shown in FIG. 17 and the communication control circuit 120 shown in FIG. 47 have slightly different details such as channel information transferred via the communication control circuit 120. For example, in FIG. 17 it is necessary to transfer different channel information for each subcarrier since a general communication scheme is targeted, but in the case of FIG. 47 single channel information on the time axis (ie, frequency dependent It is sufficient to transfer only common channel information) to all subcarriers without any gender. However, except for such specific information differences, the other functions are equivalent, and therefore, they will be described using the same reference numerals.

  Furthermore, while the first transmission signal processing unit 181-j shown in FIG. 18 obtains individual transmission weight vectors of each frequency component, the first transmission signal processing unit 381-j in the fifth embodiment is simple. The point from which only the transmission weight vector in one time axis component is determined is the difference from the first transmission signal processing unit 181-j in FIG. 18. In addition, when the radio station apparatus of the communication partner is fixedly installed, the transmission weight vector itself is calculated and recorded in advance, and this is simply read to transmit to the first transmission signal processing circuit 311 It may be configured to notify weight vectors, and further, in the case of point-to-point type one-to-one communication, the transmission weight vector may be stored directly in the first transmission signal processing circuit 311. good.

FIG. 48 is a block diagram showing a configuration example of the first received signal processing unit 385-j of the base station apparatus in the fifth embodiment. As shown in the figure, the first reception signal processing unit 385-j includes antenna elements 851-1 to 851-N ' _BS-Ant and low noise amplifiers (LNA) 852-1 to 852-N' _BS-Ant. , Local oscillator 853, mixers 854-1 to 854-N ' _BS-Ant , filters 855-1 to 855-N' _BS-Ant , A / D converters 856-1 to 856-N ' _{BS- An Ant} , a first reception signal processing circuit 358-j, an FFT circuit 857-j, and a first reception weight processing unit 360. The first reception weight processing unit 360 includes a first channel information estimation circuit 362 and a first reception weight calculation circuit 363.

  When the first received signal processing unit 385-j in the fifth embodiment applies time-axis beamforming to the first received signal processing unit 185-j shown in FIG. 19 in the first embodiment The circuit configuration of The difference from the first received signal processing unit 185-j shown in FIG. 19 is that a large number of FFT circuits 857 are integrated into one, the received signal processing circuit 858 and the first received weight processing unit 160. Instead, the first reception signal processing circuit 358 and the first reception weight processing unit 360 are provided, and further, the FFT circuit 857 is disposed downstream of the first reception signal processing circuit 358.

  Also, in the first received signal processing unit 185-j shown in FIG. 19, multiplication of the reception weight vector on the frequency axis is essential, and so regardless of whether it is OFDM or SC-FDE, the multiplication on the frequency axis may be performed. As received signal processing, FFT processing is essential in any arrangement. On the other hand, since the first reception signal processing unit 385-j in the fifth embodiment can perform signal processing of directional beam forming in sampling data units on the time axis, the first reception signal can be processed. It is also possible to omit the FFT circuit 857-j from the processing unit 385-j and perform pure single carrier signal processing. Furthermore, as shown in FIG. 46 (b), the FFT circuit 857-j is omitted here also in the case of using the reception weight matrix of the time axis in the weight following the two-step weight multiplication, Furthermore, it is also possible to continue signal processing on the time axis. In this sense, the FFT circuit 857-j shown in FIG. 48 is represented by a dotted square.

The operation of the first received signal processing unit 385-j included in the base station apparatus of the fifth embodiment will be described. Antenna elements 851-1~851-N 'of the processing for the received signal in the _BS-Ant, low noise amplifier 852-1~852-N' from _BS-Ant A / D converter 856-1~856-N ' _The processing performed up to _BS-Ant is the same processing as the processing in the first received signal processing unit 185-j of the first embodiment. By this processing, the received signal is converted into a digital baseband signal (sampling data). The A / D converters 856-1 to 856 -N ′ _BS-Ant input a reception signal vector composed of digital baseband signals in each sampling to the first reception signal processing circuit 358-j. According to an instruction from communication control circuit 120, first reception signal processing circuit 358-j multiplies the reception signal vector on the time axis received from first reception weight processing unit 360 by the reception signal vector for each sampling value. And convert it into a single signal sequence.

  As in the case of OFDM and SC-FDE, the first received signal processing is required if signal processing on the frequency axis is required after that, or if processing using a reception weight matrix on the frequency axis is performed in the latter stage. The circuit 358-j outputs the signal series of one system obtained by the conversion to the FFT circuit 857-j, and the FFT circuit 857-j converts the signal on the time axis into a signal on the frequency axis. In the case of a system that performs all signal processing on the time axis like a pure single-carrier signal, or in the case where the reception weight matrix of the time axis is subsequently multiplied later, through the FFT circuit 857-j This signal is output (FFT circuit 857-j is omitted). The FFT circuit 857-j performs FFT on the time axis at a predetermined symbol timing determined by the circuit for timing detection, the description of which is omitted here, from the signal received from the first received signal processing circuit 358-j. The signal is converted into a signal on the frequency axis (digital baseband signal for each subcarrier). The FFT circuit 857-j outputs the digital baseband signal for each subcarrier to the second received signal processing unit 75. When the first received signal processing unit 385-j does not include the FFT circuit 857-j as described above, the first received signal processing circuit 358-j determines the received weight vector on the time axis for each sampling value. And the signal of one system obtained by multiplying the received signal vector by the signal vector is output to the second received signal processing unit 75.

If each sampled received signal vector acquired by each A / D converter 856-1 to 856 -N ′ _BS-Ant is based on a channel estimation training signal, this received signal vector Is input to the first channel information estimation circuit 362. The first channel information estimation circuit 362 uses the antenna elements 819-1 to 819-N _MT-Ant of each terminal station apparatus and the antenna elements 851-1 to 851-N of the base station apparatus 70 by any method as described above. 'Channel information between the antenna element and the virtual antenna element configured by _BS-Ant is estimated on the time axis (for example, corresponding to the correlation operation shown in equation (52)) It is output to the calculation circuit 363. The first reception weight calculation circuit 363 calculates a complex conjugate value or a complex conjugate value obtained by dividing each component for each component on the time axis to be multiplied, based on the input channel information on the time axis. The reception weight vector is calculated. While the first reception signal processing unit 185 shown in FIG. 19 in the first embodiment determines individual reception weight vectors of each frequency component, the first reception signal of the base station apparatus in the fifth embodiment The point with which the processing unit 385 finds only the reception weight vector of a single time axis component is the difference from the first reception signal processing unit 185. The first reception weight calculation circuit 363 outputs the reception weight vector calculated in this manner to the first reception signal processing circuit 358-j. The calculation of the reception weight vector by the first reception weight calculation circuit 363 corresponds to the processing described in FIG.

  Further, the communication control circuit 120 manages control related to the entire communication, such as management of the transmission source station and overall timing control. Here, when the terminal station apparatus at the other end of communication is fixedly installed, the reception weight vector itself is calculated and recorded in advance, and this is simply read out to the first reception signal processing circuit 358-j. Alternatively, the reception weight vector may be notified, or in the case of point-to-point one-to-one communication, the reception weight vector may be stored directly in the first reception signal processing circuit 358. May be

  Although the above description is the signal processing for only the virtual transmission line corresponding to the first singular value, while effectively utilizing the virtual transmission line corresponding to the first singular value mainly considering the line of sight wave Also, for example, in the case of using a relatively clean reflected wave component of a large structure or the like as an auxiliary, several samples other than addition synthesis by multiplying coefficients between sampling values at the same time between multiple antenna elements are also used. It may be configured to include a delay component. For example, in the case where the first antenna is used as the reference antenna, and the coefficient by which the delay wave for one sampling clock leaks to the jth antenna is obtained, the following equation (57) obtained by extending the equation (52) may be used.

Here, S _j (n) represents the n-th sampling value of the j-th antenna element, as in equation (52). Here, since signal processing is performed on the signal of one antenna element pair, one antenna element, it is not necessary to use the value in one symbol as it is as the line gain, and using the average value of multiple times It is desirable to increase the value. Based on this coefficient, the value of the reception weight may be determined in the same manner as in equation (50) or equation (51). For the reception weight obtained in this manner, for example, as reception signal processing, a signal sequence on one time axis may be calculated by the following equation (58).

Here, ~ w ' _j is the reception weight calculated using equation (50) or equation (51) for the coefficient c' _j . In this way, it is also possible to perform processing taking into consideration a delay wave for one sampling clock. Similarly, in the case of considering a delay wave for two sampling clocks, the following equation (59) may be applied.

Here, ̃w ′ ′ _j is the reception weight calculated using the equation (50) or the equation (51) for the coefficient c ′ ′ _j . Similar extension makes it possible to consider additional delayed wave components, but both of the equations (58) and (60) can be used for the respective antenna elements of the same time as shown in the equation (53). It is a common feature that it includes a component that adds and synthesizes a signal obtained by multiplying the signal of sampling value by the reception weight ̃w _j , and the essence of this embodiment is here.

  In this embodiment, in the synthesis of transmission and reception signals of a large number of antenna elements, transmission and reception weights having frequency dependency are used, and different transmission and reception weights are multiplied for each subcarrier on the frequency axis. Instead, directivity control on the time axis is made possible for each sampling value by using a common transmission / reception weight for all subcarriers, and this may be similarly used in both the base station apparatus and the terminal station apparatus. It is also possible to implement only the base station apparatus or the terminal station apparatus. Although the above description is about the base station apparatus, the following description is about the terminal station apparatus. Here, as in the case of the base station apparatus, it is possible to apply time axis beamforming on both the transmitting side and the receiving side.

FIG. 49 is a block diagram illustrating an example of a circuit configuration of the transmission unit 361 included in the terminal station apparatus in the fifth embodiment. As shown in the figure, the transmission unit 361 includes the second transmission signal processing circuit 348, the IFFT & GI provision circuits 813-1 to 813-N _SDM , and the first transmission signal processing circuits 331-1 to 331-N _SDM. , Addition synthesis circuits 812-1 to 812-N _MT-Ant , D / A converters 814-1 to 814-N _MT-Ant , local oscillator 815, and mixers 816-1 to 816-N _MT-Ant , Filters 817-1 to 817-N _MT-Ant , high power amplifiers (HPA) 818-1 to 818-N _MT-Ant , antenna elements 819-1 to 819-N _MT-Ant , and the first And a transmission weight processing unit 340. The first transmission weight processing unit 340 includes a first channel information acquisition circuit 341, a first channel information storage circuit 342, and a first transmission weight calculation circuit 343. The configuration of the terminal station apparatus in the fifth embodiment is the same as the configuration of the terminal station apparatus 60 (FIG. 20) in the first embodiment, except that a transmitting unit 361 is provided instead of the transmitting unit 61.

  The transmission unit 361 in the fifth embodiment has a circuit configuration in the case where time axis beamforming is applied to the transmission unit 61 shown in FIG. 25 in the first embodiment. The difference between the transmitting unit 361 and the transmitting unit 61 shown in FIG. 25 is that a large number of IFFT & GI applying circuits 813 are integrated into one, and instead of the first transmission weight processing unit 140, a first transmission weight is generated. The point that the processing unit 340 is provided, and the point that the IFFT & GI application circuit 813 is disposed at the front stage of the first transmission signal processing circuit 331.

  In addition, since transmission unit 61 shown in FIG. 25 requires transmission weight vector multiplication on the frequency axis, regardless of whether it is OFDM or SC-FDE, IFFT processing is performed as transmission signal processing on the frequency axis. It was essential in any arrangement. On the other hand, the transmitting unit 361 in the fifth embodiment can perform signal processing of directional beam forming in units of sampling data on the time axis, so the IFFT & GI applying circuit 813 is also omitted and a pure single It is also possible to use carrier signal processing. In this sense, the IFFT & GI application circuit 813 in the figure is represented by a dotted square.

The operation of the transmission unit 361 included in the terminal station apparatus of the fifth embodiment will be described. When data (data inputs # 1 to #N _SDM ) for the number of space multiplexes N _SDM systems is input from the MAC layer processing circuit 68 to the second transmission signal processing circuit 348, the second transmission signal processing circuit 348 The transmission signal processing is performed on each input data system to generate a digital baseband signal for each data system. This signal processing is general and may be any type of radio signal, either OFDM, single carrier, SC-FDE or others. When arbitrary precoding is performed on the transmission signal, the second transmission signal processing circuit 348 performs precoding signal processing. If the signal output from the second transmission signal processing circuit 348 is a signal on the frequency axis, the IFFT & GI application circuit 813 converts the signal on the frequency axis into a signal on the time axis for each signal system. In the case of a system in which sampling data on the time axis is input like a pure single carrier signal, the second transmission is performed without intervention of the IFFT & GI application circuit 813 (ie, the configuration in which the IFFT & GI application circuit 813 is omitted) The signal processing circuit 348 inputs the signal of each data system obtained by the signal processing to the first transmission signal processing circuits 331-1 to 331 -N _SDM .

The first transmission signal processing unit 381-j included in the base station apparatus shown in FIG. 47 performs signal processing on a single signal sequence. On the other hand, in order to process a plurality of signal systems in the terminal station apparatus, the terminal station apparatus is configured to include a plurality of first transmission signal processing circuits 331-1 to 331-N _SDM . The first transmission signal processing circuits 331-1 to 331 -N _SDM multiply the transmission weight vector on the time axis for each sampling value by the signal series of the corresponding data system, and each transmission signal vector of this result is obtained. The components are output to the adding and combining circuit 812-1 to 812-N _MT-Ant . Adder / synthesizer circuits 812-1 through 812-N _MT-Ant adds and synthesizes the signals of all signal sequences input from respective first transmission signal processing circuits 331-1 through 331-N _SDM , and the same system Output to the D / A converters 814-1 through 814-N _SDM . The subsequent processing is the same as the processing performed in transmission unit 61 shown in FIG. The transmission weight vector on the time axis multiplied by the signal series of each data system in the first transmission signal processing circuits 331-1 to 331 -N _SDM is the same as the transmission unit 61 at the time of signal transmission processing, It is acquired from the first transmission weight calculation circuit 343 provided in the first transmission weight processing unit 340. Alternatively, as a configuration in which the transmission weight vector on the time axis acquired in advance as described above is stored in the first transmission weight processing unit 340, the first transmission signal processing circuit 331-1 will be described. .About.331-N _SDM may be instructed.

In the first transmission weight processing unit 340, the first channel information acquisition circuit 341 separately acquires channel information acquired by the reception unit 365 via the communication control circuit 121, and while sequentially updating this, It is stored in the first channel information storage circuit 342. At the time of signal transmission, the first transmission weight calculation circuit 343 reads the channel information corresponding to the destination terminal station apparatus from the first channel information storage circuit 342 according to the instruction from the communication control circuit 120, and the read channel information The transmission weight vector on the time axis is calculated based on. The transmission weight vector can be calculated using any method similar to that of the above-described base station apparatus. Furthermore, while the transmission unit 61 shown in FIG. 25 obtains individual transmission weight vectors of each frequency component, the first transmission weight calculation circuit 343 in the fifth embodiment is a transmission weight with a single time axis component. The point at which only the vector is determined is the difference from the transmission unit 61. In addition, when the terminal station apparatus is fixedly installed, the transmission weight vector itself is calculated and stored in advance, and the first transmission signal processing circuit 331-1 to 331 -N _SDM is simply read out. In the first transmission signal processing circuit 331-1 to 331 -N _SDM , the transmission weight vector may be notified to the transmission weight signal, or in the case of point-to-point type one-to-one communication. The transmission weight vector may be stored directly in.

FIG. 50 is a block diagram showing a configuration example of the receiving unit 365 of the terminal station apparatus in the fifth embodiment. As shown in the figure, the reception unit 365 includes antenna elements 851-1 to 851-N _MT-Ant , low noise amplifiers 852-1 to 852-N _MT-Ant , a local oscillator 853, and mixers 854-1 854-N, filters 855-1 to 855-N, A / D converters 856-1 to 856-N, first received signal processing circuits 355-1 to 355-N _SDM, and an FFT circuit 857-. 1 to 857-N _SDM , a second reception signal processing circuit 359, and a first reception weight processing unit 354. The first reception weight processing unit 354 includes a first channel information estimation circuit 356 and a first reception weight calculation circuit 357.

The receiving unit 365 in the fifth embodiment has a circuit configuration in the case of applying time-axis beamforming to the receiving unit 65 shown in FIG. 23 in the first embodiment. The difference from the reception unit 65 shown in FIG. 23 is that a large number of FFT circuits 857 are integrated into one, the first reception signal processing circuit 155, the second reception signal processing circuit 159, and the first reception weight processing. A first reception signal processing circuit 355, a second reception signal processing circuit 359, and a first reception weight processing unit 354 in place of the unit 154, and FFT circuits 857-1 to 857-N _SDM ; Is a point disposed at the rear stage of the first reception signal processing circuit 355.

Also, in the receiver 65 shown in FIG. 23, since multiplication of the reception weight vector on the frequency axis is essential, regardless of whether it is OFDM or SC-FDE, FFT processing is performed as reception signal processing on the frequency axis. It was essential in any arrangement. On the other hand, the reception unit 365 in the fifth embodiment can perform signal processing of directional beam formation in units of sampling data on the time axis, so that the FFT circuits 857-1 to 857-N _SDM can also be used. It is also possible to omit and use pure single carrier signal processing. In this sense, the FFT circuits 857-1 to 857-N _SDM shown in FIG. 50 are represented by dotted-line squares.

The operation of the reception unit 365 provided in the terminal station apparatus of the fifth embodiment will be described. Among processing for signals received by the antenna elements 851-1 to 851-N _MT-Ant , low noise amplifiers 852-1 to 852-N _MT-Ant to A / D converters 856-1 to 856-N _MT-Ant The processing performed up to is the same processing as the processing in the receiving unit 65 of the first embodiment. The A / D converters 856-1 to 856-N _MT-Ant input received signal vectors formed by digital baseband signals in each sampling to the first received signal processing circuits 355-1 to 355-N _SDM . . In the base station apparatus, as shown in FIG. 48, the first received signal processing unit 385-j has processed the signal series of one system. On the other hand, in order to process a plurality of signal systems in the terminal station apparatus, the receiving unit 365 is configured to include a plurality of first reception signal processing circuits 355-1 to 355-N _SDM . According to the instruction from communication control circuit 121, first received signal processing circuits 355-1 to 355-N _SDM multiplies the received signal vector on the time axis corresponding to each signal sequence with the received signal vector for each sampling value. , Convert to the signal of each signal sequence. The reception weight vector corresponding to each signal sequence is input from the first reception weight processing unit 354. The first received signal processing circuits 355-1 to 355-N _SDM output the obtained signals of the respective signal sequences to the FFT circuits 857-1 to 857-N _SDM corresponding to the signal sequences.

  Strictly speaking, the communication control circuit 121 shown in FIG. 20 and the communication control circuit 121 shown in FIG. 50 have slightly different details of channel information transferred via the communication control circuit 121. For example, in FIG. 20, it is necessary to transfer different channel information for each subcarrier because a general communication method is targeted, but in the case of FIG. 50, single channel information on the time axis (ie, frequency dependence). It is sufficient to transfer only common channel information) to all subcarriers without any gender. However, except for such specific information differences, the other functions are equivalent, and therefore, they will be described using the same reference numerals.

If signal processing on the frequency axis is required after that, as in OFDM and SC-FDE, the FFT circuits 857-1 to 857-N _{SDM use} the signal of each signal series on the time axis as the frequency axis. Convert to the above signal. In the case of a system that performs all signal processing on the time axis like a pure single carrier signal, or when performing signal processing using the reception weight matrix on the time axis also in the latter stage, the FFT circuit 857-1 The signals of the respective signal series are output to the second reception signal processing circuit 359 without passing through 857 -N _SDM (that is, the FFT circuits 857-1 to 857 -N _SDM are omitted).

For example, in the case of performing signal processing on the time axis in the second received signal processing circuit 359, the first received signal processing circuit 355-1 to 355-N _{SDM is} directed on the time axis of the spatial multiplexing number N _SDM. The signal of each signal sequence is input, multiplied by the reception weight matrix on the time axis, signal separation of the spatially multiplexed signal sequence is performed on the time axis, and the separated signals are input as necessary. A signal detection process is performed, and the reproduced data sequence is output to the MAC layer processing circuit 68.

Here, the reception weight matrix with which the second reception signal processing circuit 359 multiplies the signal of each signal sequence is a reception weight matrix obtained by the processing shown in FIG. Here, the reception weight matrix acquired in advance may be used, or the head of a signal (wireless packet) input from the first received signal processing circuit 355-1 to 355-N _SDM each time a wireless packet is received. The channel matrix relating to the channel information is obtained by the operation shown in equation (56) using the training signal for channel estimation given in, and the general MIMO signal processing is performed by the ZF method, MMSE method, etc. A linear weight may be calculated and used as a reception weight matrix. The signal of the N _SDM system separated in this way may be single carrier signal processing if it is a single carrier signal, and if it is OFDM or SC-FDE, timing detection is omitted here. The signal on the time axis is converted to the signal on the frequency axis (digital baseband signal for each subcarrier) by FFT at a predetermined symbol timing determined by the circuit for the normal demodulation processing, and the reproduced data The sequence is output to the MAC layer processing circuit 68.

On the other hand, when the second reception signal processing circuit 359 performs signal processing on the frequency axis, the FFT circuits 857-1 to 857-N _SDM output the signal on the frequency axis to the second reception signal processing circuit 359. The received weight matrix may be input and obtained in advance, or may be the head of a signal (wireless packet) input from the first received signal processing circuit 355-1 to 355-N _SDM each time a wireless packet is received. The channel matrix is obtained by performing channel estimation based on the relationship between the information on the training signal portion for channel estimation given in the above and the known training signal, and ZF and MMSE are obtained as general MIMO signal processing. , Etc., may be used as a reception weight matrix, or signal detection processing may be performed by non-linear processing such as MLD or simplified QR-MLD. The data sequence reproduced by performing such signal detection processing is output to the MAC layer processing circuit 68. Note that the signal detection processing or demodulation processing in the above description may be configured to include deinterleaving, error correction processing, and the like as necessary.

If the received signal vector having each sampling value obtained by each A / D converter 856-1 to 856 -N _MT-Ant as a component corresponds to a training signal for channel estimation, The received signal vector is input to the first channel information estimation circuit 356. The first channel information estimation circuit 356 uses the antenna elements 851-1 to 851-N _MT-Ant of each terminal station apparatus and the virtual antennas of the base station apparatus 70 by any method similar to the above-mentioned base station apparatus 70. Channel information between itself and the element is estimated on the time axis, and the estimation result is output to the first reception weight calculation circuit 357. The first reception weight calculation circuit 357 calculates a reception weight vector to be multiplied based on the input channel information on the time axis. While the receiver 65 shown in FIG. 23 determines the individual transmission weight vector of each frequency component in the first embodiment, the receiver 365 of the terminal station apparatus in the fifth embodiment has a single time axis. The point at which only the transmission weight vector in the component is determined is the difference from the reception unit 65. The first reception weight calculation circuit 357 outputs the reception weight vector thus calculated to the first reception signal processing circuits 355-1 to 355-N _SDM . Alternatively, as a configuration in which the reception weight vector on the time axis acquired in advance as described above is stored in the first reception weight processing unit 354, the first reception signal processing circuit 355-1 is obtained therefrom. -355-N _SDM may be instructed.

Further, the communication control circuit 121 manages control related to the entire communication, such as management of the transmission source station and overall timing control. Here, when the terminal station apparatus is fixedly installed, the reception weight vector itself is calculated in advance and recorded, and the reception weight vector for the first reception signal processing circuit 355 is simply read out. In the case of point-to-point type one-to-one communication, the first reception signal processing circuits 355-1 to 355-355-N can directly receive weight vectors in the _SDM . You may remember. The above is the description of the configuration example of the reception unit 365 of the terminal station device in the fifth embodiment.

  As a point of the fifth embodiment, in the case where a plurality of antenna elements are provided, the difference in the path length of the line-of-sight wave becomes sufficiently small by narrowing the spacing between the antenna elements. The frequency dependency tends to be small. At this time, when the reception weight is converted to the time domain by IFFT or the like, almost the first wave component (component of the line-of-sight wave) occupies most of the total received power. Under such circumstances, MIMO signal separation performed in the frequency domain can be performed in the time domain. Further, MIMO signal separation that can be performed in the time domain using only the first wave component can be performed for each sampling value.

  Therefore, in the wireless communication system according to the fifth embodiment, for example, at the time of signal reception, the first received signal processing unit 385 provided in the base station apparatus and the first received signal processing in the receiving unit 365 provided in the terminal station apparatus. The circuit 355 performs first stage signal separation for separating the signal of the virtual transmission line corresponding to the first singular value obtained by singular value decomposition of the channel matrix from the reception signal for each sampling value on the time axis Do. Thereafter, the second received signal processing unit 75 provided in the subsequent stage of the first received signal processing unit 385 in the base station apparatus and the second received signal processing circuit 359 in the receiving unit 365 which can be provided in the terminal station apparatus are virtual. The second stage signal separation is performed to separate the signals between the transmission paths. In the apparatus on the receiving side of the wireless communication system of the fifth embodiment, when the second stage signal separation is performed on the time axis, the reception weight matrix used in signal separation is a signal received for each antenna element It is possible to calculate based on the channel matrix obtained by finding the correlation value of

  By performing MIMO signal separation in the time domain, it is possible to eliminate the FFT required for signal separation, and it is possible to reduce computational load and circuitry. In addition, since it is not necessary to convert signals on the frequency axis when performing spatially multiplexed single carrier transmission, the effect of phase noise, which is a problem in high frequency bands such as millimeter waves, can be reduced to the existing pure single It is possible to divert phase noise compensation techniques in carrier transmission. Also, while reducing the calculation load for MIMO signal separation in the base station apparatus and the terminal station apparatus, as in the first to fourth embodiments, band expansion is realized in a situation where direct waves in a line-of-sight environment are dominant be able to.

  In the wireless communication system of the fifth embodiment, the configuration for obtaining the reception weight / transmission weight from the correlation coefficient has been described. However, channel information may be fed back to some of the subcarriers, and the reception weight / transmission weight of all frequency bands may be calculated using the average value.

Sixth Embodiment
[Approximate solution for channel matrix acquisition when line of sight wave dominates]
(Outline of Basic Principle According to Sixth Embodiment)
In the above description, spatial multiplexing transmission using virtual transmission paths corresponding to a plurality of first singular values has been described. In this space multiplex transmission, it is necessary to carry out singular value decomposition of the channel matrix and calculate its first right singular vector or first left singular vector. Also, in order to obtain the original channel matrix used for this singular value decomposition, channel estimation and feedback of channel information are necessary. Here, in order to improve the gain by using a large number of antenna elements, it is necessary to obtain a component of a channel matrix corresponding to the product of the numbers of transmitting antennas and receiving antennas. Generally, the overhead accompanying the training signal transmission for channel estimation increases in proportion to the number of components of the channel matrix, and the amount can not be ignored.

  In implicit feedback, for example, when the uplink channel matrix can be obtained, the calibration process can estimate the downlink channel matrix and reduce the overhead associated with the increase in the number of antenna elements of the base station apparatus. it can. On the other hand, if there are multiple antenna elements on the terminal side, the uplink channel matrix can not be acquired unless the training signal is individually transmitted from each antenna element, and the overhead accompanying the increase in the number of antenna elements on the terminal side is It can not be reduced.

  In the case where the number of antenna elements on the terminal side is relatively small, for example, if two antennas are used, each antenna transmits a training signal on even and odd subcarriers, and becomes independent in two subcarrier periods on the frequency axis. Channel estimation is possible. In this case, channel information of a subcarrier between which channel estimation has not been performed is estimated by interpolation. It is possible to efficiently perform channel estimation by using a plurality of orthogonalized preamble signals. However, in the case of including 16 or 32 multi-element antenna elements, for example, there is a problem that the subcarrier interval is too wide, and the accuracy of channel estimation by interpolation is extremely reduced. There are limitations in channel estimation and its feedback.

  Furthermore, even when a large-scale antenna is applied to improve the channel gain, a single element antenna can not expect a high SNR (signal-to-noise ratio) that allows effective channel estimation on the channel design. For this reason, for example, long-time measurement and averaging processing on the assumption that there is no channel time fluctuation are required. However, long-time averaging can not be employed in the case of time-varying cases, for example, as in the case of a train moving cell, and it is necessary to instantly acquire channel information of all necessary paths. In a general MIMO system, for example, in the case of providing N antenna elements on the transmission side, channel estimation is performed using N training symbols of one OFDM symbol. However, when both transmitting and receiving stations are equipped with a large number of antenna elements as described above, there is a possibility that the channel may be time-varying during N channel estimation. For this reason, a channel estimation method is required that finishes acquiring necessary channel information in as short a time as possible.

  Here, when the virtual transmission path for the first singular value described above is actively used, the antenna group of the terminal device and the antenna group of each first signal processing unit of the base station device are very narrow. It is arranged intensively. Therefore, the channel information changes minutely due to the slight positional deviation. However, if attention is focused only on the line-of-sight component, the change is expected to change in a manner that maintains regularity among the components of the channel vector. For example, when a signal transmitted from an antenna element on the transmitting side is received by the antenna group on the receiving side, channel information changes depending on the difference in path length of the signal. Therefore, it is necessary to obtain what this regularity is like on a propagation model that takes into account only "line-of-sight waves" and to carry out channel estimation efficiently by actively using the regularity. Can. Below, the method to implement this channel estimation efficiently is demonstrated.

FIG. 51 is a diagram schematically showing received signals received by a plurality of antenna elements. In this figure, reference numerals 221-1 to 221-5 denote antenna elements, and another antenna element is located at a distance L from the antenna element 221-1 in the angle θ direction with respect to the antenna elements 221-1 to 5-5 (in the figure, a diagram The situation when receiving the signal from (not shown) is shown. Here, the received signal at time t at the m-th antenna element 221-m is represented as _{m m} (t). When plane wave approximation is performed by setting the distance from the transmitting antenna to the first antenna element 221-1 of the incoming wave from the direction of angle θ as L, the wavefront is a line indicated by a dotted line.

Therefore, the difference between the arrival distances of the first antenna element 221-1 and the m-th antenna element 221-m is ΔL _m as shown in the figure. If the antenna element spacing is d, the path difference between the mth and (m + 1) th antenna elements is d · sin θ. However, in this case, in order to generalize also including the arrangement other than the linear array, the difference in path length is defined based on the first antenna element 221-1 using ΔL _m . Here, if the reception signal of the m-th antenna element 221-m is divided into the k-th subcarrier φ _m ^(k) (t), the reception signal Φ _m (n) at time t at the m-th antenna element 221-m t) can be expressed by the following equation (61) as a sum of subcarrier components.

Here, if the transmission signal component at time t is _represented by S _k (t), the reception signal of the radio frequency f _c + f _k with respect to the center frequency f _c can be represented by the following equation (62).

Here, c _m ^(k) is a coefficient (relative channel information) of the received signal at the m-th antenna element 221-m when the received signal at the first antenna element 221-1 is based on the k-th subcarrier Yes, it is given by the following equation (63).

Here, α _m is a constant relating to the m-th antenna element 221-m, and is defined by the following relational expression (Expression (64)).

The right side of equation (63) is expressed by the product of two natural logarithms, but the former is a constant which does not have frequency dependence for each subcarrier, while the antenna element takes different values, while the latter is a frequency. Show the dependency. Considering that W is a frequency bandwidth, if x = f _k / W, then when α _m is smaller than 1, it is complex when x changes by a value between -1/2 and +1/2. The phase changes linearly at 2πα _m x. If this complex phase linearly changing feature is used, linear interpolation of complex phase components is performed on all subcarriers if some subcarriers are determined without calculating reception weights on all subcarriers. It is possible to calculate the reception weight of

Although the right side of the equation (63) is a complex number having an absolute value of 1 in the above description, the actual measurement result does not strictly become a complex number having an absolute value of 1 due to the influence of noise and reflected waves. In such cases, observed _c ^m a ^{_{(k) c m (k)}} / | c m (k) | with substituted on normalized value and performing the following processes. In addition, when the reception weight is calculated using the complex conjugate value of the channel information standardized in this manner, the weight can be obtained as a weight for equal gain combination instead of the maximum ratio combination. Since the line-of-sight waves predominately assume an environment, it is expected that there will be no significant difference in characteristics between equal gain combining and maximum ratio combining.

FIG. 52 is a diagram showing an outline of channel estimation with limited subcarriers. In this figure, channel estimation is performed on the m-th antenna element for four subcarriers f _{A to} f _D within the frequency bandwidth W, and normalized relative channel information c _m obtained by measurement. A value obtained by dividing the value obtained by taking the natural logarithm (the logarithm using the base e of the natural logarithm) with respect to ^(fk) and dividing by 2πj is Y _m (f _k ) (equation (65)).

Referring to equation (65), when f _k / W is plotted on a graph with x and Y _m (f _k ) as y, the graph on the right in FIG. 52 is obtained for x between −1/2 and +1/2. Y _m (f _k ) observed in the vicinity of the straight line y = ax + b is linearly plotted as in FIG. Therefore, if this straight line is fitted by the least squares method, the frequency dependency of channel information can be obtained. Conventionally, there has been a technique for performing channel estimation by making the subcarriers transmitting the training signal defective. However, since they have no information on the frequency dependence of channel information in the entire band, linear interpolation between the two observed subcarriers and frequency dependence of the two before and after In consideration of this, it has been limited to processing such as non-linear interpolation. In the present embodiment, in the case of focusing only on the line-of-sight component, the linear phase interpolation utilizing the frequency dependence is shown over the entire band, taking advantage of showing linear frequency dependence in complex phase in all frequency bands. To do. However, it should be noted here that when taking natural logarithm of complex number, Y _m (f _k ) takes the same value even if an integer multiple of 2πj is added to the exponentiation part of e. There is a point that correction must be made.

For example, in the example shown in FIG. 52, the least squares method is a regression line and four points (f _A / W, Y _m (f _A )), (f _B / W, Y _m (f _B )), (f _C / The operation is performed on W, Y _m (f _C ), and (f _D / W, Y _m (f _D )). In addition to this, for example, for η _A , _{B B} , η _C , and _{D D} , each of which is given by an offset value of ± 1 and 0, respectively, a regression line and four points (f _A / W, Y _{m (f A) + η A)} , (f B / W, Y m (f B) + η B), (f C / W, Y m (f C) + η C), (f D / W, Y m (f Perform the least squares method between _D ) + _{D D} ) to calculate a and b for each combination of offset values, and among them, the combination of a and b that minimizes the following equation (66) Think of it as a true regression line.

Here, since the offset value can take values of −1, 0, and +1 in each of the four subcarriers, the 3 ⁴ = 81 least squares method is implemented as a combination of all of these. The channel information of the following equation (67) may be calculated for an arbitrary frequency f _k using a and b obtained in this manner, and the reception weight may be calculated as this complex conjugate.

If added slightly, the coefficient a, which is the slope of this regression line, is a physical quantity corresponding to α _m in equation (64). If this is, for example, a value sufficiently smaller than 1, the line of y = ax + b will have a change of y sufficiently smaller than 1 when x changes between -1/2 and 1/2, noise and reflected waves The offset amount is considered to be sufficient in the range of ± 1 even in consideration of the influence of However, if the value of ΔL _m is increased to some extent, the change of y will exceed 1 and in such a case, it can be coped with by expanding the selection range of 0, ± 1, ± 2, etc. It becomes.

If the value of [Delta] L _m is small to some extent, on the other hand, for example, _f B / and _Y m _{(f B)} observed in W, is observed at the front and rear of _f A / W and _f C / W _Y m ( f _A ) and Y _m (f _C ) do not make a big difference. In this case, both values of | Y _m (f _A ) -Y _m (f _B ) + η _B | and | Y _m (f _C ) -Y _m (f _B ) + η _B | are predetermined values (for example, 0) An offset amount η _{B which} is larger than 1) (eg, a value smaller than 1 such as .5 or 0.3) can be excluded from consideration. Here, for example, since Y _m (f _A ), Y _m (f _D ), etc. are observation points at both ends, instead of comparison with observation points at both ends, comparison with only one may be performed. In this way, it is possible to limit the range of the search while considering the offset amount, and it is possible to suppress an increase in the amount of calculation involved in performing the individual least squares method.

  In the above channel estimation, since it is sufficient to transmit the training signal with limited subcarriers from a certain transmitting antenna, the transmission power is focused on a small number of subcarriers as compared to transmitting training signals to all the subcarriers. You will be able to For example, assuming that the total number of subcarriers is 1000, when focusing on four subcarriers, transmission can be performed with 250 times the transmission power as compared to transmission on all subcarriers, so 10 Log 250 ≒ 24 [dB] You can earn circuit gain. Therefore, the channel estimation accuracy will be improved accordingly.

Furthermore, by using four subcarriers f _A 'to f _D ' of subcarriers different from the subcarriers f _A to f _D described above, it is possible to simultaneously perform similar channel estimation for different antenna elements. It is. For example, even if an antenna of 250 elements is provided on the transmitting side, if each transmits on only four subcarriers, a total of 1000 subcarriers will be sufficient. That is, even if a MIMO channel is configured with 250 antenna elements and a large number (for example, 100 elements) of antenna elements, channel information of a 250 × 100 MIMO channel matrix of all subcarriers in one channel estimation Can be obtained.

  Here, if the number of effective subcarriers (subcarriers used in data communication) is not a multiple of 4, it is necessary to at least cope with adjustment of the fraction. For example, a part of a training signal to which only a few lesser sub-carriers are allocated may be used for a certain antenna element, or a part of effective sub-carriers may be adjusted without assigning a training signal. good. This area depends on the system parameters.

  Here, for example, as shown in FIG. 16, a case where a plurality of first signal processing units 304 are provided with a plurality of antenna elements is considered. Assuming that each first signal processing unit 304 is provided with 50 antenna elements, and further provided with five first signal processing units 304-1 to 5, the total number of antenna elements is 250. . When channel information in the downlink is acquired using these antennas as described above, they can all be acquired collectively. In addition, when spatial multiplexing is performed using a plurality of virtual transmission paths corresponding to the first singular value, transmission directivity control across different first signal processing units 304 is not performed, so that the first temporary In the case where ten signal processing units 304 are provided, the same processing is performed for the first to fifth first signal processing units 304, and the sixth to tenth first signal processing units 304, and The channel estimation may be divided into two.

  It is important to simultaneously perform channel estimation on antenna elements related to the same first signal processing unit 304, and for antenna elements belonging to different first signal processing units 304, perform channel estimation collectively. do not have to. Incidentally, if there are 2000 subcarriers, then 4 subcarriers × 50 antennas × (5 sets + 5 sets) results in 2000, so channel estimation is possible if divided into two. As described above, the total number of subcarriers, the number of subcarriers used for channel estimation at one time, and the like are parameters on the system design, and the present embodiment is applicable to any other values as well.

Here, points to be noted when the transmitting side performs channel estimation using a plurality of antenna elements will be described. First, in the above description, as shown in equation (63), the reception signals of the plurality of antenna elements on the reception side are divided by the reception signal of the antenna element (here, the first antenna element is assumed) serving as a reference. That is, relative channel information is used as a relative relationship of channel information based on the first antenna element, not absolute channel information. If this is explicitly shown, the relative channel matrix _Hrelative obtained by the above-described procedure sets the component of the channel information between the jth transmit antenna and the ith receive antenna of the kth subcarrier as h _ij ^(k) Then, it is given by the following equation (68).

The feature of the relative channel matrix _Hrelative is that each component of the row vector in the first row is 1 and all information is based on the first antenna element on the receiving side. The relative channel information for each subcarrier determined by the above-mentioned least squares method is all about the receiving antenna after the second antenna, and the channel information regarding the first antenna element on the receiving side is out of the discussion above. Of course, since sampling information on Φ ₁ (t) is acquired, absolute channel information can be obtained through FFT processing or the like based on this information over one OFDM symbol. However, as described above, for example, the signal transmitted from the first antenna on the transmitting side is only the signal on subcarrier f _A to f _D , for example, the signal on another subcarrier f _A ′ to f _D ′ is on the transmitting side It becomes a signal to be transmitted from the second antenna.

That is, only the channel information of the first transmission antenna can be acquired for the same subcarrier f _A , and only the channel information of the second transmission antenna can be acquired for the subcarrier f _A ′. For example, when changing antenna elements between even and odd subcarriers, channel information of subcarriers for which training signals are not transmitted can also be predicted by linear interpolation, but a small number of By limiting the training signal to the carrier, linear interpolation between them becomes impossible.

  However, as described later, when singular value decomposition is performed on this matrix, the left singular matrix is affected regardless of the diagonal component of the second term of the product on the right side of equation (68). Will produce the same left singular matrix.

The reason is as follows. First, when all components of a matrix are multiplied by a predetermined coefficient, when singular value decomposition is performed on the matrix, all singular values are simply multiplied by the predetermined coefficient, and the right singular matrix and the left singular matrix have the same value. It becomes. Next, and the antenna element in 1 / h 1j ^_(k) is expected waves of the respective diagonal components of the second term of the right side of formula (68) to the vicinity respectively, corresponds to the amplitude of the received signal | h _1j ^(K) becomes approximately equal values.

That is, when the whole matrix is multiplied by the coefficient | h ₁₁ ^(k) |, each diagonal component of the second term on the right side of equation (68) becomes | h ₁₁ ^(k) | / h _1j ^(k) , The diagonal terms all have an absolute value of 1 and differ only in complex phase. In this matrix, arbitrary column vectors are orthogonal to each other, and the diagonal term of the complex conjugate of this matrix is the inverse of the original diagonal term, and the product of the complex conjugate matrix and the original matrix is It is an identity matrix. That is, since this is a unitary matrix, when singular value decomposition is performed on the matrix of the first term of the right side of equation (68), the left singular matrix and the matrix having singular values in the diagonal term are common. Is the only matrix transformed to the right singular matrix rotated by the unitary matrix described above.

Even if the entire channel matrix, which is the matrix of the first term on the right side of Equation (68), is not known, it is sufficient to know the relative channel matrix _Hrelative to calculate the reception weight vector for reception. Incidentally, if the reception weight vector is known, the transmission weight vector can also be acquired by calibration processing. As a result, if channel estimation is performed by the above-described method, both the reception weight vector and the transmission weight vector are received on the side receiving the training signal. , Which means that both can be obtained.

  As described above, according to the above-mentioned method, it is possible to obtain the reception weight vector and the transmission weight vector (or the weight matrix formed by combining the vectors) of the terminal station apparatus by performing channel estimation in the downlink, for example. However, in order to obtain the reception weight vector and the transmission weight vector (or the weight matrix formed by combining the vectors) on the base station apparatus side, a training signal is similarly transmitted from the terminal station apparatus side and the uplink The same process should be performed.

(Optional feature to improve channel estimation accuracy)
In the system to which the above-described method is applied, it is assumed that an antenna element with high directivity gain is used since the case where the arrival direction of radio waves is in a relatively narrow range is targeted. In this case, since the line-of-sight wave is dominant in the line-of-sight environment, it is expected that the channel information exhibits linear behavior on the frequency axis as described above. Is slightly expected, and due to that effect, the frequency selective distortion may show a behavior deviated from the straight line. The degree of this deviation depends on the ratio of the power of the reflected wave, but if, for example, regression calculation is performed using only four subcarriers, the accuracy may be low. Therefore, it is considered to comprehensively carry out regression calculation for the entire channel information of multiple antenna elements.

For example, in consideration of a linear array as shown in FIG. 51, it is assumed that the path length difference ΔL _m between the first antenna element and the m-th antenna element will be the following relational expression (69).

That is, in equation (64), α _m is given by the following equation (70).

In this case, it is sufficient to acquire the entire channel information with the constraint of Expression (70). In addition, several variations can be considered as how to apply this constraint condition. For example, a and b are individually determined by the above-mentioned least squares method for all antennas, and since the slope a of this straight line coincides with α _m , a / (m−1) is calculated for 2 or more m an average value, an average value ^ alpha ₂ of alpha ₂ of the formula (70) with respect to two or more m. Next, for the m-th antenna, (m-1) × ^ α ₂ may be substituted for a, and the Y intercept at the inclination may be determined for each antenna element by the least squares method.

Other methods include, for example, the following methods. If the arrival direction of the angle θ and the antenna element spacing d of the radio wave is obtained, alpha ₂ is because it can estimate the approximate value d · sinθ × W / c, α 2 value for testing a plurality of values in the vicinity And a = α _m (that is, the slope of the regression line) by (m−1) × α ₂ , and using the a by the least squares method that introduces the offset value η _k for each antenna as described above The value b _m of the Y intercept of the m-th antenna and the offset value group {η _k } used at that time are determined together, and this α ₂ and the corresponding {b _m } {η _k } are used to obtain Calculate F (α ₂ , {b _m }, {η _k }) given by 71).

Here, Σ of the sum with respect to _k means the sum with respect to only the subcarrier used in each antenna element, and η _k means the offset value of the k th subcarrier used in the m-th antenna. _{F (α 2, {b m} }, {η k}) for such alpha ₂ in a, _{F (α 2, {b m} }, {η k}) and alpha ₂ when is minimum, It is possible to obtain channel information of each subcarrier of each antenna element using the set {b _m } of Y intercepts at that time.

As a supplement here, since the value of this Y-intercept is originally given by f _c ΔL _m / c, it should have the periodicity of the equation (69) for the antenna number m. However, for example, assuming that the center frequency is 80 GHz, f _c / c is about 267 and the complex phase is one revolution when the value of ΔL _m is 3.35 _mm , even when dividing by 3 × 10 ^8. It will be. As f _c ΔL _m / c + η, it is observed in the state where the aliasing offset is given by exceeding 2π, and since the offset value が can be a relatively large value, equation (70) It becomes impossible to grasp such proportional relationships.

  Therefore, the constraint condition as in the equation (70) which is set to the value of b of the regression line for each antenna element can not be expressed in a simple form, and each has a relation to be evaluated as an independent Y intercept value. This is the reason why it is difficult to calculate the exact solution of this problem. However, it is possible to apply the present embodiment with high accuracy if the reception weight vector is calculated using channel information acquired by applying measures to improve the accuracy of the above-mentioned least squares method in consideration of such effects. It is possible.

  In the example of the above-mentioned description, although an example was shown about a case where four subcarriers are used, it may be naturally the number of subcarriers other than four. If the number of subcarriers used here is small, the accuracy of the least squares method will fall, but conversely if the number of subcarriers is increased, the transmission power per subcarrier will decrease, so the SNR of channel estimation will decrease. The estimation accuracy is degraded. Although it is possible to improve the accuracy by correcting the slope a of the regression line of the least squares method by using a plurality of receiving antennas in combination, the above method can not improve the accuracy with respect to b of the Y intercept . For this reason, in an actual system design, the number of subcarriers used for channel estimation will be optimized by balancing these.

In the above description, channel estimation by transmission of one training signal has been described. However, as is clear from equation (63), different sub-signals may be used to obtain relative channel information with the reference antenna element. It is not necessary to perform channel estimation in carriers at the same time. For example, after channel estimation is performed for only subcarriers f _A to f _D, if channel estimation is performed also for other subcarriers f _A ′ to f _D ′ by changing time, for eight subcarriers. Information can be obtained. By repeating these processes, it is possible to improve the accuracy of regression line fitting by using more subcarriers.

(Supplement on how to get the complete channel matrix)
In the above description, it is assumed that, for example, in the case of downlink, the terminal station apparatus alone calculates the reception weight and the transmission weight on the terminal station apparatus side, each component of the matrix of the second term of the right term of equation (68) Was indeterminate. However, if it is applied to a fixedly installed wireless entrance or a train moving cell, it is possible to combine the information acquired in the downlink and the information acquired in the uplink and grasp the both, and then execute the process. It is.

For example, assuming that equation (68) is downlink information, the components of the first column vector of the relative channel matrix _Hrelative obtained in the uplink are subjected to calibration processing. Assuming that the component of channel information between the jth transmit antenna and the ith receive antenna of the kth subcarrier of the downlink converted by this calibration process is h ′ _ij ^(k) , the above process for the uplink is performed. ^{_{h '12 (k) / h}} ' 11 (k), h '13 (k) / h' 11 (k), become ^{_{···, h '1M (k)}} / h' 11 (k) that is obtained . 'When multiplying ₁₁ ^(k), the (j, j) of the matrix of the second term of the right side diagonal elements h' coefficient h to the components of the relative channel matrix _{H relative} downlink ₁₁ ^(k) _{/ h 1j} It becomes ^(k) . If this is possible approximated by ^{_{h '11 (k) / h}} ' 1j (k), which h obtained for uplink ^{_{'12 (k) / h'}} 11 (k), h '13 (k) / h ^{_{'11 (k), ···,}} h' corresponds to the reciprocal of the ^{_{1M (k) / h '11}} (k). Therefore, it is possible to obtain all channel information by using this information.

  It should be noted that the above-mentioned approximate solution method for channel estimation is a channel estimation method using a condition in which only the sight wave is dominant, and it goes without saying that the absolute value of the first singular value in the single MIMO channel matrix is natural. It is expected that the absolute value of the singular value less than or equal to the second singular value with respect to the value will be a very small value. Therefore, the MIMO channel itself is not suitable for spatially multiplexing a plurality of signal sequences. However, if multiple virtual transmission paths corresponding to the first singular value are used in parallel by using a plurality of different antenna groups, efficient space multiplex transmission is possible. If used for such an application, such an approximate solution of channel estimation will function effectively.

Furthermore, with regard to “an optional function for improving channel estimation accuracy”, the accuracy considering the regularity of the path length difference for each antenna element is utilized by using the case where the antenna elements on the receiving side are arranged in a linear array. Signal processing for improvement has been described. As for the prior art, as the path length difference is described as ΔL _{m in} general, no restriction is placed on the arrangement of the antenna element. This discussion is based on the premise that the line-of-sight component is dominant to the last, so if it is possible to extend the range of the offset value _{k k} somewhat, even if the distance between the antenna elements is somewhat wide, this technique is applied It is possible. In particular, when the antenna element spacing increases, the absolute values of the inner product values of the column vectors or row vectors of the relative channel matrix H _relative to one another remain relatively high, but their respective components change almost irregularly. However, since restrictions on the antenna configuration and arrangement are not particularly strongly limited as application conditions, the present embodiment is applicable under a wide range of conditions if the sight wave is dominant.

(Processing flow of approximate solution for channel matrix acquisition when line of sight wave dominates)
The processing operation of the approximate solution for channel matrix acquisition will be described below with reference to the drawings. Here, no distinction is made between uplink and downlink, and either the base station apparatus or the terminal station apparatus transmits a training signal for channel estimation, and the other receives the training signal to obtain a channel matrix. It is possible to obtain a bidirectional channel matrix by performing the same process in the reverse direction after acquiring the channel matrix in one direction. When performing transmission and reception on the virtual transmission line corresponding to the first singular value, a complete channel matrix is not necessarily required, and it is only necessary to obtain a relative channel matrix _Hrelative as in equation (68). The method of acquiring the channel matrix will be described. Further, for convenience of explanation, the case where a training signal is transmitted in downlink will be described as an example.

  Next, with reference to FIG. 53, the processing operation of the approximate solution of channel matrix acquisition in the present embodiment will be described. FIG. 53 is a flowchart showing the processing operation of the approximate solution for channel matrix acquisition in the present embodiment. First, when the base station apparatus transmits a training signal (step S5301), the terminal station apparatus receives a training signal at each antenna element (step S5302). In the reception here, a signal is amplified by a low noise amplifier, down-converted from a radio frequency to a baseband signal by a mixer, signal components out of band are removed by a filter, and sampling is performed by A / D conversion. This signal is subjected to FFT processing to convert it into a signal on the frequency axis.

The training signal here is a training signal in which a transmission signal is present only on a specific subcarrier on the frequency axis in order to increase transmission power per subcarrier, and the terminal station apparatus is supposed to use the discontinuous subcarrier. Among the sub-carriers, the sub-carriers including the signal component are extracted as channel estimation results of the sub-carriers (step S5303). Then, the terminal station apparatus divides this result by the same channel estimation result of the antenna serving as a reference to acquire c _m ( _k ) in equation (63) (step S5304).

Next, the terminal station device obtains Y _m (f _k ) by equation (65) (step S5305), and calculates a and b to minimize equation (66) by the least squares method taking into account the offset value _{k k} (Step S5306). Subsequently, the terminal station apparatus substitutes equation (67) based on a and b obtained here and performs channel estimation of each subcarrier (step S5307), and further performs calibration to perform bi-directional communication. The relative channel matrix is calculated (step S5308), and this is recorded and managed as a relative channel matrix (step S5309).

  On the other hand, recording management of transmission / reception weight vectors is more important than recording management of the channel matrix in practice. Value decomposition is performed, and a reception weight vector is acquired as a first left singular vector (step S5310). Next, the terminal station apparatus performs calibration processing on this reception weight vector (step S5311), and acquires a transmission weight vector. Then, the terminal station apparatus records and manages the transmission / reception weight vector acquired in this manner (step S5312).

The above is the case of performing channel estimation by closing to a single antenna element, but it is also possible to improve the estimation accuracy in consideration of a plurality of antenna elements as described above. FIG. 54 is a flow chart showing processing operation of approximate solution of channel matrix using a plurality of antennas. Similar to the processing operation shown in FIG. 53, when each antenna element receives a training signal in the terminal station apparatus, predetermined reception processing is performed (steps S5401-1 to S5401-3), and signal components among discontinuous subcarriers Are extracted as sub-carriers including the sub-carriers as channel estimation results of the sub-carriers (steps S5402-1 to S5402-3). Subsequently, the terminal station apparatus divides this result by the same channel estimation result of the reference antenna to obtain c _m ( _k ) in equation (63) (steps S5403-1 to S5403-2), Y _m (f _k ) is determined for each antenna element according to equation (65) (steps S5404-1 to 5404-2).

Here, the terminal station apparatus sets, for example, a value of α ₂ with a predetermined step width, and inclines the regression line as a = (m−1) α _{2 with} respect to Y _m (f _k ) of the m-th antenna element. are fixed and determine the b by the least square method in consideration of the offset _{value, F (α 2, {b} m}, {η k}) of the formula (66) determine the alpha ₂ that minimizes (step S5405) , the least squares method considering the offset value at each antenna element by using the alpha ₂ was performed using equation (71) (step S5406), where the resulting a, formula based on b (67) The channel estimation value of each subcarrier is substituted and implemented (step S5407), calibration is further performed (step S5408), a bidirectional relative channel matrix is determined, and this is recorded and managed (step S5409).

  On the other hand, recording management of transmission / reception weight vectors is more important than recording management of the channel matrix in practice, in which case the terminal station apparatus obtains channel estimation results for each subcarrier in equation (67), Value decomposition is performed, and a reception weight vector is acquired as a first left singular vector (step S5410). Next, the terminal station apparatus performs calibration processing on this reception weight vector (step S5411), and acquires a transmission weight vector. Then, the terminal station apparatus records and manages the transmission / reception weight vector acquired in this manner (step S5412).

  Next, with reference to FIG. 55, the processing operation for obtaining an approximate solution of the channel matrix from the bidirectional channel estimation result will be described. FIG. 55 is a flowchart showing a processing operation for obtaining an approximate solution of a channel matrix from bidirectional channel estimation results. First, the downlink will be described. When the base station apparatus transmits a training signal (step S5501), the terminal station apparatus receives a training signal at each antenna element (step S5502).

The training signal here is a training signal in which a transmission signal is present only on a specific subcarrier on the frequency axis in order to increase transmission power per subcarrier, and the terminal station apparatus is supposed to use the discontinuous subcarrier. Among the sub-carriers, the sub-carriers including the signal component are extracted as channel estimation results of the sub-carriers (step S5503). Then, the terminal station apparatus divides this result by the same channel estimation result of the antenna serving as a reference to acquire c _m ( _k ) in equation (63) (step S5504).

Next, the terminal station device obtains Y _m (f _k ) by equation (65) (step S5505), and calculates a and b to minimize equation (66) by the least squares method taking into account the offset value _{k k} (Step S5506). Subsequently, the terminal station apparatus substitutes equation (67) based on a and b obtained here to carry out channel estimation of each subcarrier (step S5507).

  Next, the uplink will be described. When the terminal station device transmits a training signal (step S5508), the base station device receives the training signal at each antenna element (step S5509).

The training signal here is a training signal of missing teeth on the frequency axis in order to increase transmission power per subcarrier, and the base station apparatus includes a signal component among the discontinuous subcarriers. Subcarriers are extracted and used as channel estimation results for the subcarriers (step S5510). Then, the base station apparatus divides this result by the same channel estimation result of the antenna serving as a reference to acquire c _m ( _k ) in equation (63) (step S5511).

Next, the base station device obtains Y _m (f _k ) according to equation (65) (step S5512), and calculates a and b to minimize equation (66) by the least squares method taking into account the offset value _{k k} Step S5513. Subsequently, the base station apparatus substitutes equation (67) based on a and b obtained here to carry out channel estimation of each subcarrier (step S5514).

  Next, an absolute channel matrix is calculated from the relative channel matrix of uplink and downlink according to equation (68) (step S5515). Then, calibration is performed (step S5516), and this is recorded and managed as a bidirectional channel matrix (step S5517).

  The calibration process in step S5516 is basically the process of steps S5502 to S5507 and the individual processes of steps S5509 to S5516 to obtain row vector components of the first row of equation (68). It can be acquired except ambiguity (relative relationship of channel information). Therefore, if the respective information are butted in step S5515 to compensate for the ambiguity of the first row row component in both the uplink and the downlink, calibration is not performed (ie, step S5516 is omitted It is also possible to obtain a bi-directional channel matrix.

As described above, although the above description is a procedure for obtaining a relative channel matrix, if information on both uplink and downlink is known, not only the relative channel matrix but also the channel matrix as an absolute value It can be asked. Similar to the above description, after receiving the training signal by each antenna element, channel estimation of each subcarrier is performed by equation (67) by predetermined processing. This is acquired separately on the downlink terminal station side and the uplink base station side. Collected both obtained _{^{results, (h 11 (k) /}} h 11 (k), h 12 (k) / h 11 (k), h 13 (k) / h 11 (k), ···, h _By multiplying a matrix having _1M ^(k) / h ₁₁ ^(k) ) as a diagonal component from the right of the relative channel matrix, 1 / h ₁₁ ^(k) is added to the matrix of the first item on the right side of equation (68) The multiplied channel channel can be obtained. If calibration processing is performed on this, a bidirectional channel matrix can be obtained and recorded and managed.

  As described above, in the line-of-sight environment and when the antenna element spacing is narrow, it is expected that the complex channel phase of relative channel information exhibits linear frequency dependence even when the channel information has frequency dependence. it can. Moreover, it is possible to improve the channel estimation precision before directivity formation by limiting the subcarrier used for a training signal and raising the transmission power per subcarrier and transmitting. By using these features and performing channel estimation with a small number of subcarriers, it is possible to provide a method of estimating accuracy while estimating channel information of unacquired subcarriers in a least square method. . Furthermore, by using different sub-carriers for each antenna element, it is possible to transmit training signals simultaneously from multiple antenna elements, and it becomes possible to cope with multiple transmit antennas that can not be solved by implicit feedback, and at the same time Channel information can be efficiently fed back simultaneously to a plurality of antenna elements or antenna element groups.

  Note that since the actual channel contains a reflected wave, the channel information does not form a straight line on the frequency axis but contains many errors. It is expected that it is possible to extract features.

Seventh Embodiment
[Approximate solution of weight vectors corresponding to a plurality of first singular values]
(Outline of Basic Principle According to the Seventh Embodiment)
In the case of ordinary MIMO transmission, it has been considered ideal to separate the antenna element intervals as much as possible for the purpose of reducing the correlation between the antenna elements. However, in order to actively use the virtual transmission path corresponding to the first singular value as described above, it is better to narrow the spacing between the antenna elements instead and to enhance the correlation. Of course, if the antenna element spacing is set to 1/2 wavelength or less, mutual coupling between the antenna elements can not be ignored, so it is necessary to set an interval of about 1⁄2 wavelength, but the spacing of that degree If this is the case, it is ideal to narrow the element spacing of the antenna to enhance the correlation. Although the above-mentioned approximate solution of channel estimation described an applicable method without too much restriction on the antenna arrangement, it is different if this narrows the element spacing of the antenna and limits it to a state where the correlation is enhanced It is possible to calculate the transmission and reception weights by a simpler method.

  Here, although the channel correlation of the channel matrix itself is large, for example, when the channel information between the first transmitting antenna and the first receiving antenna and the transmission at the same first antenna are compared with the channel information of the second antenna, In most cases, the values show completely different values due to the difference in path length. In order to perform singular value decomposition, the same operation is required regardless of whether the channel correlation is strong or weak, so even if only the right singular vector or the left singular vector related to the first singular value is determined, The load required for the operation is not too small. In particular, since the number of antenna elements is huge, the matrix size is huge, and it is necessary to calculate the right singular vector and the left singular vector corresponding to the first singular value in a simple calculation.

  Here, considering that the correlation of the antenna is high, approximate solutions of the right singular vector and the left singular vector corresponding to the first singular value are determined by the following method. First, in the case where the first left singular vector is used as the reception weight vector, one virtual antenna element is formed by multiplying the reception signal of each reception antenna element by the weight vector. Since this forms a beam tuned to a line of sight wave, it is expected that the beam is directed from the receiving antenna group to the transmitting antenna group side. Since the antenna elements on the transmission side are spaced apart by at least a half wavelength or more, strictly speaking, the first right singular vector of singular value decomposition is formed in consideration of the spread thereof However, since it is close to a state in which one virtual antenna element is disposed substantially in a very narrow area, reception directed to a specific antenna installed near the center of gravity of these transmitting antenna groups It can be expected that the weight vector can be roughly approximated.

Therefore, the receiving weight vector directed to one specific transmitting antenna element near the center of gravity of the transmitting antenna group is determined, and the transmission side uses the first right singular vector obtained by singular value decomposition for transmission, and Consider the case where the right and left singular vectors corresponding to the first singular value are used for both the transmit and receive weight vectors. First, as an example, it is assumed that a channel matrix in which the (i, j) component of the matrix is given by the line-of-sight model of equation (20) by 31 elements antennae for transmission and reception. Let v _{1 be} the first right singular vector when singular value decomposition is performed as in equation (4), and u ₁ be the first left singular vector. The gain when forming the transmission / reception weight vector based on these is given by the following equation (72).

On the other hand, the SIMO channel vector (h _1,16 , h _2,16 , h _3,16 ,..., H _31,16 ) ^T between the 16th element at the center of the 31 elements of the transmit antenna group and the receive antenna group On the other hand, a vector obtained by normalizing this Hermite conjugate vector is approximately received weight vector w _rx-app. Consider the case of (formula (73)).

  The gain in this case is as shown in the following equation (74).

  Further, the training signal is limitedly transmitted from the antenna element in the vicinity of the center of gravity of the antenna group not only in one direction but also in both directions to obtain both reception weight vectors, and from there, the transmission weight vector is obtained by calibration processing In this case, it is possible to simplify the transmission weight vector and the reception weight vector as much as possible from the feedback of the same channel information to the calculation processing of the transmission and reception weight vectors. The gain at this time is given by the following equation (75).

Here, _Grx-app. G _ideal and G _app. The results of evaluating the distribution characteristics of -G _ideal are shown in FIG. The evaluation conditions are: ± 5 [5] in the x-axis direction centering on a point where the antenna 31 elements are arranged in parallel to the y-axis as a linear array on a xy two-dimensional plane with 1 wavelength interval m], the installation point was determined with uniform random numbers in the range of ± 5 [m] in the y-axis direction, and the CDF distribution of the gain difference was evaluated. As parameters, the frequency is 20 GHz, and the distance L in the x direction is 20 m, 30 m, and 100 m.

From these results, there is almost no difference between using approximate weight vectors and using ideal weight vectors by singular value decomposition, and the closer the distance between the transmitting and receiving stations is, the larger the gain gap becomes. It can be seen that the gain difference is within 0.25 dB even at 20 m. That is, in the present embodiment, when performing signal transmission using a virtual transmission path corresponding to the first singular value, the reception weight vector w _rx-app. And calibration processing is performed on the transmission weight vector w _tx-app. By performing signal transmission and reception in accordance with the above, it is possible to perform communication as good as in the case of using the optimal transmission and reception weight vector obtained by almost singular value decomposition. Here, although the 20 GHz band has been described as an example of the frequency, if the frequency becomes higher such as 80 GHz band, the antenna element spacing can be shortened by shortening the wavelength, and it is possible to use approximately one point physically. Since the antenna is placed near, the approximation accuracy is further enhanced and the gain gap can be ignored.

(Option for approximate solution of weight vector corresponding to multiple first singular values)
The above is the basic principle of the simple weight vector calculation method, but some problems remain if this is done. For example, when gain on line design is obtained by using a large-scale antenna as described above, it can not be said that the accuracy of channel estimation for one antenna to one antenna is very high. Therefore, similar to the technique used in “approximate solution for channel matrix acquisition when line of sight wave is dominant” (sixth embodiment), the number of subcarriers used for channel estimation is significantly limited, and there per subcarrier It is applied to increase the transmission power of to improve the channel estimation accuracy for that subcarrier. Furthermore, by using the fact that the amount between antenna elements is narrow, it is possible to maintain channel estimation with a fairly good accuracy even if the transmitting antenna transmitting the training signal is slightly away from the center of gravity of the antenna group in channel estimation. Become.

  For example, when performing the same evaluation as in FIG. 56, it is considered to obtain transmission / reception weight vectors using an antenna element slightly away from the center of gravity without using an antenna near the center of gravity. As an example, in the case of disposing 61 elements in a linear array at 1⁄2 wavelength intervals in the same manner as in the evaluation of FIG. 56 and using transmit / receive weight vectors using 31st antenna elements corresponding to the center of gravity When elements are used, the gain in the case of using the 61st antenna element at the end of the linear array is evaluated as the amount of deterioration from the case of using ideal transmission and reception weights using singular value decomposition for both transmission and reception.

  FIG. 57 is a diagram showing gain characteristics in a transmission / reception weight vector approximate solution using an antenna far from the center of gravity. As can be seen from the figure, even if the frequency is 20 GHz and the distance is 20 m, the median value of the gain gap (CDF 50% value) is 0.5 dB for the 46th antenna element, even for the 61st antenna element. It is 0.8 dB. If this degree of gain reduction is allowed, training signals of different sub-carriers are transmitted using all antennas 61 elements, and training signals of all frequency components for which training signals are transmitted by any antenna element The transmission / reception weight vector at the receiving station can be easily obtained.

  For example, if training signals of four subcarriers are transmitted from each antenna element and the number of antenna elements is 64, for example, channel estimation can be performed simultaneously for 256 subcarriers. In the above example, the linear array has been described as an example for the sake of simplicity, but if a 16 × 16 square array antenna (256 elements in total) in the form of a square lattice with half wavelength intervals is used, FIG. Since similar characteristics are expected, it is possible to estimate 1024 subcarriers at a time by simultaneously using 256 elements.

  Assuming that uplink is assumed here, it may not be possible to arrange so many antenna elements on the terminal station side, but even in that case, it has been explained in the "approximate solution for channel matrix acquisition when line of sight wave is dominant" It is known that the weight value between subcarriers of each component (corresponding to an individual antenna element) of a vector of transmission / reception weight vectors similarly exhibits a gradual fluctuation in complex phase, and the horizontal axis represents subcarriers and the vertical axis In the graph of the complex phase, if the offset value is corrected taking into consideration the 2π cycle aliasing, the weight value is considered to be contained in a linear gradual change within the entire band and calculated. It is also possible to obtain weight values of subcarriers that can not be obtained by performing linear interpolation while extending the complex phase of weight values out of the range of −π to + π. In this case, it is preferable that the channel information be relative channel information based on the complex phase of the reference antenna, and the reception weight be determined from the relative channel information.

  Here, an example of use of the antenna element near the center of gravity in the entire antenna will be described with reference to FIG. FIG. 58 is a diagram showing an example of antenna elements used to transmit a training signal when using a linear array. In this figure, a 31-element linear array is taken as an example, and its center of gravity antenna is the middle sixteenth antenna (shown by a double circle in the figure). Basically, only the 16th element, which is the antenna element at the center of gravity, is used to transmit the training signal as shown in FIG. 58 (a), but in this case the transmission of the training signal is performed to improve the transmission power per subcarrier. If the number of subcarriers used for the channel information is limited, the number of subcarriers from which channel information can be obtained decreases, and the accuracy of interpolation of channel information in the meantime decreases. Therefore, for example, even if the training signal is transmitted within the range of ± 5 elements of the center of gravity as shown in FIG. 58 (b) or the training signal is transmitted within the range of ± 10 elements as shown in FIG. Good. As shown in FIG. 57, if there is no problem in channel estimation accuracy, it is also possible to use all elements for transmitting training signals as shown in FIG. 58 (d).

  Next, FIG. 59 is a view showing an example of an antenna element used for transmission of a training signal in the case of using a close-packed two-dimensional antenna arrangement. In this figure, the case of a two-dimensional array of 37 elements is taken as an example, and symbols a to z and A to K are given for convenience of explanation. The center-of-gravity antenna is the a-th antenna in the middle (indicated by a double circle in FIG. 59). Since the antenna element at the center of gravity is basically the antenna element as shown in FIG. 59 (a), only the a-th element is used to transmit the training signal, but in this case, in order to improve the transmission power per subcarrier, If the subcarriers used for transmission are limited, the number of subcarriers from which channel information can be obtained decreases, and the accuracy of interpolation of channel information in the meantime decreases. Therefore, for example, as shown in FIG. 59 (b), in addition to the a-th element, the training signal is transmitted in the range of the b-th element (indicated by a black circle in FIG. 59). The K element antenna is used for communication but not for channel feedback. Alternatively, all these antenna elements may be used for both transmission of training signals and data communication.

  The same concept can be applied to the case where the number of antenna elements available for transmitting the training signal is small. For example, when using a one-dimensional linear array for transmission and reception for some reason, only antenna elements near the center of gravity can be used for channel estimation accuracy as shown in Fig. 58 (b) for transmitting training signals. It is also possible to add and operate an antenna element used only for training signal transmission separately. FIG. 60 is a diagram showing an example in the case of adding an antenna element for transmitting a training signal in the vicinity of the center of gravity of the linear array. As shown in this figure, antenna elements E, D, C, B, A, z, r, g, a, d, l, t, u, v, w, x, y constitute a linear array. The center of gravity is the a-th element, and the a-s elements are arranged in a close-packed manner at the center of this element. Antenna elements b, c, e, f, h, i, j, k, m, n, o, p, q, s that are out of the linear array are not used for data communication, but for channel estimation, A training signal is transmitted using an as element.

  This is an example, and even in the case of using a close-packed two-dimensional antenna arrangement as shown in FIG. 59, for example, the training signal is 1 around the a-th element while the close-packing is performed at intervals of 5 wavelengths. The configuration may be such that the antenna elements are concentratedly arranged in close packing at a / 2 wavelength interval and used. In this manner, the density of the antenna elements may be increased near the center of gravity, and transmission of the training signal may be performed using only the antenna elements near the center of gravity. Also, in the description of FIG. 60, the antenna elements b, c, e, f, h, i, j, k, m, n, o, p, q, s which are out of the linear array are not used for data communication, There is no problem at all in using these antenna elements for data communication.

FIG. 61 is a diagram showing a specific example of linear interpolation of approximate weight values. The horizontal axis of the graph represents subcarriers, and the vertical axis represents the complex phase of weight values. For example, it is assumed that weight values of f ₁ , f ₉ , f ₁₆ , and f ₂₅ can be obtained as indicated by plot points 291 to 294 at intervals of eight subcarriers. Since this complex phase is given so as to be in the range of π to + π, for example, when linear interpolation is performed based on plot points 291 to 295, the complex phase of the weight value of f ₂₅ exceeds + π, and the actual observation The plotted point 294 has a value near −π. However, since the complex phase can not change by nearly −2π when the subcarrier changes from f ₁₆ to f ₂₅ , the plot point 294 in which an offset of + 2π is inevitably added to the plot point 294 is an actual complex It can be considered as a phase.

By using a complex phase value corrected by adding an offset in this manner, it is also possible to obtain a complex phase between them by linear interpolation. For example, regarding subcarriers f _{2 to} f _8, it is also possible to interpolate by linear interpolation between plot points 291 and plot points 292. Similarly, by applying the least squares method using plot points 291 to 293 and plot point 295, a linear regression line can be obtained, and the weight value of each subcarrier may be given by the regression line. At this time, although it is possible to use the weight values of already acquired subcarriers f ₁ , f ₉ , f ₁₆ and f ₂₅ as they are, on the other hand, instead of using them as they are, they are replaced with linearly interpolated prediction values In some cases, estimation error components such as noise can be suppressed by giving a carrier weight value. Furthermore, in consideration of offsets of 2π from weight values of nearby subcarriers, the amount of change is extremely large in addition to other plot points (for example, the amount of change is ± π / 2 or more). It is also possible to exclude from linear interpolation as the one with low accuracy to improve the accuracy.

  Next, with reference to FIG. 62, an operation of determination processing of complex phase offset in linear interpolation of approximate weight values performed by the base station apparatus or the terminal station apparatus will be described. FIG. 62 is a flow chart showing operation of determination processing of complex phase offset in linear interpolation of approximate weight values. Here, it is assumed that the base station apparatus performs the processing operation shown in FIG. In the case where the number of subcarriers used for channel estimation can be secured to a certain extent (including the case where training signals are transmitted from a plurality of antenna elements), the interval between subcarriers for which channel estimation is performed can be relatively narrow, The difference between the complex phases of the channel estimation results of adjacent subcarriers is considered to be sufficiently small.

  Therefore, after performing channel estimation on discrete subcarriers, the base station apparatus reads out each component of the transmission / reception weight vector (or channel vector or channel matrix) for complex phase information interpolation (step S6201). Then, the base station apparatus reads the weight values Wk and Wk 'of consecutive subcarriers (step S6202). At this time, for each vector or matrix, the channel information acquired is read out in order from, for example, the smaller subcarrier number. Then, the base station apparatus obtains the complex phase θ (Wk) of Wk and Wk ′ (not necessarily because k and k ′ are adjacent because they are discrete) of consecutive acquired subcarriers Wk and Wk ′. Theta (Wk ') is acquired (step S6203). Subsequently, the base station apparatus adds + 2π if the difference between the complex phase θ (Wk ′) and θ (Wk) is less than or equal to −π (steps S6204 and S6205). By subtracting 2π (steps S6206 and S6207), the absolute value of the difference between the corrected complex phase θ (Wk) and θ (Wk ′) is set to π or less.

  Basically, this alone is sufficient, but when it is judged that the information on the complex phase difference of ± π / 2 or more is low in credibility, the complex phases θ (Wk) and θ (Wk ′) after correction The differences are compared again (steps S6208 and S6209), and if the difference is less than -.pi./2 or more than .pi. / 2, the channel information is judged to be unreliable and the weight value Wk 'is discarded (step S6210). If all the acquired subcarriers have been checked, the processing is ended, and if not checked, the next subcarrier is read out, and the same processing is performed between the values of the previously processed subcarriers. It carries out (step S6211).

  Although the correction process is performed as the complex phase, the actual weight value is given by Exp (jθ (k), so that the correction of the complex phase 2π does not cause any difference as the weight. It is possible to apply an arbitrary interpolation process on the complex phase corrected in this way, and perform linear interpolation from the complex phases of two adjacent points. It is also possible to perform linear interpolation in a least-squares manner from complex phases of multiple points, or other non-linear interpolation, and in addition to interpolation, outside of the two points Extrapolation may be used.

  In the various descriptions above, the “offset value” was introduced by linear interpolation (regression calculation using the least squares method) such as channel vector, channel matrix, transmission / reception weight vector, etc. The correction of the complex phase performed without abrupt complex phase change between subcarriers is basically equivalent to the introduction of the “offset value”, and the process shown in FIG. The introduction of “offset value” (processing such as regression calculation using a least squares method for a plurality of offset value candidates) is not necessarily required, and can be omitted. The above-mentioned correction processing can be similarly used in the processing in the sixth embodiment.

(Processing of approximate solution of weight vector corresponding to a plurality of first singular values)
Next, with reference to FIG. 63, the processing operation of the approximate solution method of the weight vector corresponding to the plurality of first singular values will be described. FIG. 63 is a flowchart showing processing operation of approximate solution of weight vectors corresponding to a plurality of first singular values. Here, regardless of the uplink and the downlink, either the base station apparatus or the terminal station apparatus transmits a training signal for channel estimation, and the other receives the training signal to obtain a channel vector, and as a result, Use to calculate the reception weight vector. Furthermore, a transmission weight vector is also acquired by calibration processing. It is possible to obtain bidirectional transmission and reception weight vectors by acquiring the transmission and reception weight vectors in one direction and then performing the same process in the opposite direction. Also, for convenience of explanation, the following description will be made by taking a case of transmitting a training signal in downlink as an example.

  The base station apparatus transmits a training signal using a part of subcarriers in the whole in order to increase transmission power per subcarrier (step S6301). At this time, the training signal is transmitted by the antenna element around the center of gravity of the antenna group. The terminal station apparatus receives this training signal and performs predetermined reception processing (step S6302). In the reception processing here, a signal is amplified by a low noise amplifier, down-converted from a radio frequency to a baseband signal by a mixer, signal components out of band are removed by a filter, and sampling is performed by A / D conversion This signal is subjected to FFT processing to convert it into a signal on the frequency axis. The terminal station apparatus takes a complex conjugate with the channel vector obtained in this way (and, if necessary, includes a normalization process) to calculate a reception weight vector (step S6303).

  Subsequently, the terminal station device implements correction processing of the complex phase (processing shown in FIG. 62) (step S6304). The terminal station apparatus acquires channel vectors of all subcarriers by linear interpolation or the like (step S6305). Subsequently, the terminal station apparatus predicts the reception weight vector of the subcarrier which can not be acquired by various interpolation processing such as linear interpolation and non-linear interpolation, calculates the transmission weight vector by calibration processing after acquiring all reception weight vectors. (Step S6306). Finally, the terminal station device records and manages the transmission weight vector (step S6307).

  As described above, when acquiring transmission / reception weights for using a virtual transmission line corresponding to the first singular value in the line-of-sight environment, use is made of the fact that the antenna element group is limited to a very narrow place. The reception weight can be calculated extremely easily by obtaining the reception signal vector using a single antenna element near the center of gravity of the antenna group and taking its complex conjugate. The transmission weight can also be acquired if calibration processing is performed, and if this processing is performed in both directions, the base station apparatus and the terminal station apparatus can easily approximate the transmission and reception weights of the virtual transmission path corresponding to the first singular value. Can be asked. Further, while only the antenna near the center of gravity is used for training signal transmission, it is also possible to improve the line gain and to improve the spatial multiplexing characteristic by using the entire antenna element whose aperture length is expanded for transmitting and receiving data.

  It should be noted that if the transmission weight vector is known on the transmission side, it is not necessary to transmit the training signal using only the antenna near the center of gravity and limiting the subcarriers to be used. It is also possible to transmit a training signal by increasing the gain by directivity formation using transmission weight vectors in all antenna elements using a carrier. Therefore, the present embodiment is used as a method for acquiring the initial value of the transmission weight vector, and after acquiring the transmission weight vector, directivity is formed using the transmission weight vector to transmit a training signal, and so on. Channel estimation in a state where the channel gain is enhanced, and the reception weight vector is determined with high accuracy by the complex conjugate of the channel vector, and calibration processing is performed on the high accuracy reception weight vector. A procedure such as obtaining a transmission weight vector may be taken. By repeating such processing sequentially, it is possible to perform communication in a more stable situation in a steady communication state.

Eighth Embodiment
[About simultaneous channel estimation in multiple radio stations]
(Outline of Basic Principle according to Eighth Embodiment)
In the above-mentioned “approximate solution for channel matrix acquisition when line of sight wave is dominant” and “approximate solution for weight vector corresponding to a plurality of first singular values”, 1 is achieved by limiting subcarriers transmitting training signals 1 It is shown that the channel estimation accuracy of this antenna to one antenna can be improved. Also, with these techniques, different training signals of different subcarriers can be simultaneously transmitted from multiple antennas, thereby obtaining channel information on different antenna elements.

  In the description of these techniques, for example, a training signal is transmitted using different subcarriers from a plurality of antenna elements of the same terminal station apparatus, or a plurality of first signals in the base station apparatus 303 as shown in FIG. The case where the processing unit 304 is included has been described. In this case, the transmission control of the training signal from the antenna element of the first signal processing unit 304 is adjusted by the communication control circuit etc. in the base station apparatus 303, and the timings are aligned and transmitted simultaneously. It was controlled to do. On the other hand, it may be effective to simultaneously perform channel estimation of a plurality of antenna elements under other conditions. As an example thereof, in addition to the application of the present embodiment to the wireless entrance channel as described above, it may be used for the access system of the small cell base station apparatus.

  FIG. 64 is a diagram showing an apparatus configuration for performing simultaneous channel estimation among a plurality of small cells existing in the same area. In FIG. 64, in the left figure, for example, a plurality of small cell base station apparatuses 231 to 234 are installed on a building wall of a building, and an environment where the service areas are spread around the base station apparatuses 231 to 234 is shown. ing. In this service area, terminal stations 235 to 239 that perform wireless communication with the base stations 231 to 234 exist. The terminal station apparatus 235 is under the control of the small cell base station apparatus 231, the terminal station apparatus 236 is under the control of the small cell base station apparatus 232, the terminal station apparatus 237 is under the control of the small cell base station apparatus 233, and the terminal station The device 238 and the terminal station device 239 are subordinate to the small cell base station device 234.

  As described in the background art, small cells are installed to offload traffic from areas where traffic is concentrated locally from the macro cells. A small cell in a narrow area is set in an area where traffic is concentrated, and the small cell base station apparatus performs communication limited to terminal stations in the small cell. In the case of a macro cell, it is general to strictly perform station design and manage the level of mutual interference between macro cells to reuse frequency resources. However, small cells are installed as spots according to requirements such as traffic concentration, and station placement design is not necessarily implemented in a planned manner.

  In such an environment, a small cell configuration technique using Massive MIMO technology has attracted attention for the purpose of reducing mutual interference in accordance with the requirement of securing channel gain. In this small cell, in order to reduce mutual interference, although the transmission power per element is not high, a huge line gain is secured by the directional gain formed by a large number of antenna elements. However, to ensure the channel gain, it is necessary to know the MIMO channel matrix between each transmit antenna element and receive antenna element, and a channel feedback technique for this is important.

  Furthermore, although the accuracy of channel estimation is also important in order to obtain this channel gain efficiently, when a training signal is transmitted for channel estimation, interference waves from neighboring small cells on the same subcarrier are used. If it is received, the interference signal significantly degrades the channel estimation accuracy and not only can not ensure a sufficient line gain, but also the radiation in other than the desired direction becomes large, so there is a risk that mutual interference increases. There is also. This problem is called Pilot Contamination, and the problem of training signal contamination due to mutual interference is an important problem.

  To simply avoid this problem, shift the transmission timing of the training signal for each small cell (If the training signal is transmitted in some small cells in the neighborhood, refrain from communicating in other small cells) Was considered valid. However, in this method, overhead due to a training signal for channel estimation becomes large, and transmission efficiency in the MAC layer is reduced. In particular, in a small cell in 5G, since it is assumed that one base station device accommodates a very large number of terminal stations at one time, those terminal stations frequently transmit training signals, and for each small cell. If the transmission timing is divided, the MAC efficiency will be greatly reduced.

  Therefore, as shown in the right diagram of FIG. 64, among a large number of subcarriers in the entire frequency bandwidth W, each base station apparatus 231 to 234 divides the subcarriers and divides them at the same time on the downlink. Send training signals. Similarly, the terminal station devices 235 to 238 of each small cell also divide the subcarriers among a large number of subcarriers in the entire frequency bandwidth W, respectively, and align training signals at the same time on the uplink. Send. For example, base station apparatus 231 and terminal station apparatus 235 use a plurality of subcarrier groups 241 in the right figure, and base station apparatus 232 and terminal station apparatus 236 use a plurality of subcarrier groups 242 and base station apparatus 233 and The terminal station device 237 uses a plurality of subcarrier groups 243 in the right diagram, and the base station device 234 and the terminal station device 238 use a plurality of subcarrier groups 244 in the right diagram. Since these do not overlap on the frequency axis, there is no interference of the same frequency (same subcarrier) at the same time, and the problem of Pilot Contamination does not occur.

  For the uplink, for example, the terminal station apparatus 238 and the terminal station apparatus 239 subordinate to the base station apparatus 234, for example, the terminal station apparatus 238 uses the subcarrier group 244, while the terminal station apparatus 239 (base station apparatus Transmitting a training signal using subcarrier group 243 and assuming channel information of terminal station apparatus 238 and terminal station apparatus 239 at the same time, assuming that 233 and terminal station apparatus 237 do not communicate in subcarrier group 243 It is also possible to configure.

  In general, while the base station apparatus is allowed to increase the size of the apparatus, the terminal station apparatus can not increase the apparatus size much. In addition, as shown in FIG. 16, when the plurality of first signal processing units 304 are provided, and each of the plurality of antenna elements is provided, the terminal station apparatus The total number of antennas is expected to be small. In that case, it makes sense to simultaneously perform channel estimation with multiple terminal stations in the same cell (or different cells may be included) as described above.

  Specifically, for example, it is assumed that the base station apparatus includes five first signal processing units 304, and each includes 50 antenna elements. In the case of transmitting training signals on four subcarriers per one antenna element, channel estimation of as many as 1000 subcarriers can be simultaneously performed with 4 × 50 × 5 = 1000. Furthermore, if the total number of subcarriers is 2000, for example, base station apparatus 231 and base station apparatus 233 perform channel estimation using even-numbered subcarriers, and base station apparatus 232 and base station apparatus 234 using odd-numbered subcarriers. Is possible. Depending on the directivity characteristics of the antenna, etc., if each antenna element on the base station side has directivity of a certain degree of strength, it may give interference to the adjacent small cell, but the next adjacent In the small cell of, there may be a case where interference can be ignored, and in such a case, the above setting will be established.

  On the other hand, assuming that the number of antenna elements included in the terminal station apparatus is 25, the number of subcarriers in one terminal station apparatus will be 100 in total, assuming that four subcarriers are used per element. However, since one base station apparatus uses 1000 subcarriers for channel estimation, even if terminal stations of 10 stations transmit uplink channel estimation training signals simultaneously at the same time (same time) There is no interference on the same subcarrier), and channel estimation can be performed simultaneously on these. With regard to the downlink, when the base station apparatus transmits a training signal once, all terminal stations that have received it can simultaneously estimate the downlink channel for the own station. However, for the uplink, each terminal station apparatus must transmit a training signal individually, and how to prevent the reduction in MAC efficiency due to the overhead accompanying the transmission of the training signal, while how to improve the channel estimation accuracy Is an issue.

  Here, in order to simultaneously transmit and receive the above-described channel estimation by changing subcarriers between different small cells or between different terminal stations in the same small cell, (1) Training that each terminal station should transmit It is important to be able to grasp the subcarrier of the signal and (2) to be able to grasp the timing at which each terminal station apparatus should transmit the training signal.

FIG. 65 is a diagram showing an example of a frame configuration in the present embodiment. In FIG. 65, reference numerals 251-1 to 251-3 denote slots for transmitting a training signal on the downlink by the base station apparatus, and reference numerals 252-1 to 252-3 denote data communication between the base station apparatus and the terminal station apparatus. It is a slot to do. Also, reference numerals 253-1 to 253-3, 254-1 to 254-3, 255-1 to 255-3, and 256-1 to 256-3 respectively indicate that the terminal station apparatus transmits a training signal on the uplink. Represents a slot. Also, the time length from the slot 251-1 to the slot 256-1, the time length from the slot 251-2 to the slot 256-2, and the time length from the slot 251-3 to the slot 256-3 are the basic frame periods, respectively. It is T _b-frame . Strictly speaking, the guard time is changed because switching from uplink to downlink is performed between the end of slot 256-1 and slot 251-2, between the end of slot 256-2 and slot 251-3, and so on. The predetermined time may be set as

  Also, in the example shown in FIG. 65, a superframe structure having a long period of repeating the basic frame N times is used, and slots 253-1 to 253-3 and 254-1 to 254-3 in different basic frames in this superframe, Different terminal stations from 255-1 to 255-3 and 256-1 to 256-3 are assigned to transmit uplink training signals. Focusing on any one of slots 253-1 to 253-3, 254-1 to 254-3, 255-1 to 255-3, and 256-1 to 256-3, each is divided into subcarrier groups It is done. In the above-described example, it is assumed that one terminal station apparatus transmits training signals on four subcarriers for each of fourteen subcarriers for a total of 1000 subcarriers. Therefore, for example, the (10 × m + j) subcarrier is used for the subcarrier m of the j-th (j is an integer of 1 to 10) group for the integer m of 0 to 99. (As shown in the figure, four slots 253 to 256 are allocated, so n ′ is an integer of 1 to 4) The terminal station apparatus is allocated by using the j-th group of subcarriers in the slot.

  In FIG. 65, the slots 251-1 to 251-3 are allocated to the top of each basic frame every frame, but it is sufficient if at least one long frame is included in the super frame structure, and the super frame If it is arranged at the beginning of a predetermined basic frame in the inside, correspondence is possible even if it is not arranged at the beginning of all the basic frames. Similarly, the slots 253-1 to 253-3, 254-1 to 254-3, 255-1 to 255-3, 256-1 to 256-3, etc. are described as the same number of slots in each basic frame. However, it is not necessary to necessarily have the same number, and in some cases there may be basic frames that do not contain any.

  Also, if it is assumed that time-division TDD is assumed on the assumption that the data communication slots 252-1 to 252-3 perform implicit feedback, for example, the bandwidth request from the terminal station apparatus or the base station from the network side Allocation of bands in the data communication slots 252-1 to 252-3 is adaptively performed according to the data arriving at the station apparatus. At this time, the uplink and downlink time lengths may be set fixedly, or the allocation may be variable according to the traffic conditions. In any case, the present embodiment is applicable. The data communication slots 252-1 to 252-3 are further subdivided to perform transmission and reception in units of small slots.

  In each base station apparatus and terminal station apparatus, it is assumed that the start timings of the super frame and the basic frame are known by, for example, GPS (Global Positioning System) or some other means. A signal other than the GPS may be used by the macro cell signal, or timing information may be notified by another wireless system. In addition, although the terminal station apparatus newly entering the area needs to access the base station for belonging processing first, it performs downlink channel estimation for the access, and from the estimation result, If the obtained reception weight vector and uplink transmission weight vector are not obtained, it is not possible to secure channel gain by using a large number of antennas.

  In a general wireless system, it is common to add a signal available for timing detection to the beginning of a wireless packet, but in the present embodiment, the signals in slots 256-1 to 256-3 are not good in terms of subcarriers. In general, the signal is not very suitable for timing detection because it is an incomplete signal that is not transmitted and is not a directivity signal. Also, even if the base station apparatus transmits another signal for timing detection, if the directional beam is not formed on the terminal station apparatus side using the reception weight vector, the line gain is insufficient, and the base station apparatus receives a signal. I can not receive a signal for timing detection. Therefore, it is preferable that some mechanism for making the base frame and the top of the super frame known be implemented in this embodiment, and in the following description, these timing information be known. This timing information may be grasped using other wireless systems such as GPS, or may be grasped using macrocell information (that is, synchronized with periodicity of macrocell communication).

(Process flow of simultaneous channel estimation by multiple radio stations)
Here, the processing operation for the start of communication between the terminal station apparatus and the base station apparatus newly entering the area will be described. FIG. 66 is a flow chart showing processing operation for starting communication between the terminal station apparatus and base station apparatus newly entering the area. When the terminal station apparatus starts communication (for example, when the user turns on the terminal station apparatus), the terminal station apparatus acquires the start timings of the basic frame and the super frame by predetermined means (step S6601) and the frame start Training signal is received (step S6602). Then, the terminal station apparatus acquires the reception weight vector on the downlink and the transmission weight vector on the uplink from the received slots 251-1 to 251-3 (step S6603).

  Next, the terminal station apparatus is configured to receive one of the slots 253-1 to 253-3, 254-1 to 254-3, 255-1 to 255-3, and 256-1 to 256-3 and any one of them. A combination of subcarriers of the condition allocated to the system as a slot for random access among the group of subcarriers and also the subcarriers of the condition allocated to the system as a subcarrier for random access similarly to the slot for random access It is regarded as (step S6604), and a training signal for channel estimation is transmitted there by random access (step S6605). For example, predetermined subcarriers of the slots 256-1 to 256-3 at the end of the basic frame (in the above example, divided into 10 groups, all groups or some groups may be random) are random A slot for access may be used, or a slot for random access may be arranged only on a predetermined group of subcarriers in the last slot 256-1 in, for example, the top basic frame in a superframe.

  Each time the base station apparatus receives the random access slot (step S6621), the base station apparatus performs FFT processing, and acquires the reception level of a subcarrier that may contain a training signal as the random access slot (step S6622). ). Then, if the subcarrier of the received signal has a predetermined level (step S 6623), the base station device regards it as an initial access from a new terminal station device, and performs uplink channel estimation from the training signal. (Step S6624). Subsequently, the base station apparatus acquires the downlink transmission weight vector by calibration together with the calculation of the uplink reception weight vector (step S6625). The base station apparatus, using the transmission weight vector acquired here, transmits uplink signals to the terminal station apparatus in any of the subdivided slots in the data communication slots 252-1 to 252-3. Signal transmission of a slot assignment instruction for transmission is performed (step S6626). Here, since the base station apparatus does not know the identity of the terminal station apparatus, the identification number of the terminal station apparatus can not be used for the assignment instruction, so as an instruction for initial access to the randomly accessed terminal station apparatus, For example, the terminal station apparatus specifies information such as a slot, a subcarrier, etc. which transmitted a training signal by random access, and informs the terminal station apparatus that it is an assignment to the own station.

  In response to this, the terminal station apparatus forms a directional beam with the acquired reception weight vector after the above-described random access training signal transmission, and determines a time-out (step S6606). It waits (step S6607). When the slot allocation instruction signal transmitted by the base station apparatus can be received using the above-mentioned reception weight vector (step S6608), the control information containing the information of the own station is transmitted using the instructed slot (Step S6609), and makes a request for belonging to that base station.

  The base station apparatus awaits signal reception from the terminal station apparatus while determining a timeout using the reception weight vector acquired in the previous random access slot in the slot allocated by the own station (step S6627), the terminal station apparatus And the base station apparatus performs signal reception with high quality by utilizing the channel gain secured by forming the directional beam (step S6628). After these information exchanges, the base station apparatus manages the attribution of the apparatus on its own database (step S6629).

  Also, the base station apparatus may be any one of the slots 253-1 to 253-3, 254-1 to 254-3, 255-1 to 255-3, and 256-1 to 256-3 and any one of the sub-slots. A group of carriers is allocated as a slot for channel feedback with the terminal station apparatus (step S6630), and the contents are similarly notified by the subdivided slots in data communication slots 252-1 to 252-3. Do. In response to this, the terminal station apparatus grasps a channel estimation slot for periodically transmitting a training signal to the base station apparatus on the uplink (step S6610). Thereafter, bandwidth allocation and bandwidth request are performed in the subdivided slots in the data communication slots 252-1 to 252-3, and the base station apparatus and the terminal station apparatus appropriately communicate while updating the transmission / reception weight vector Continue.

  Although detailed description is omitted in the above processing, in order to avoid stagnation of processing without normal information exchange due to a code error, the timer which is a general technique is stretched and the presence or absence of response is timed out. It is also possible to apply a combination of management techniques. Also, as described in the first embodiment, when the base station apparatus implements a plurality of first signal processing units, the amount of information of control information such as a request for attribution is limited, and thus the plurality of first There is no necessity to communicate while performing spatial multiplexing using a signal processing unit, and it is possible to select one specific first signal processing unit and use only the transmission weight vector directed to that first signal processing unit. It is also possible to transmit control information. Similarly, also in the transmission of control information from the base station apparatus, a specific first signal processing unit may be selected to transmit the control information without performing spatial multiplexing.

  Next, with reference to FIG. 67, transmission / reception signal processing operation of the base station apparatus in the present embodiment will be described. FIG. 67 is a flowchart of the transmission / reception signal processing operation of the base station apparatus. In FIG. 67, (a) shows reception processing, and (b) shows transmission processing. First, the reception process will be described. When the base station apparatus acquires frame timing (step S6701), it recognizes the reception timing of the radio packet from the terminal station grasped due to base station centralized control in the data communication slot following the slot at the head of the basic frame. (Step S6702). Then, the base station apparatus reads the reception weight vectors in the plurality of first signal processing units 304 of the own station with respect to the terminal station apparatus from the memory (step S6703), and receives the reception signal vectors of each first signal processing unit The weight vector is multiplied (step S6704).

  Subsequently, the base station apparatus aggregates one system of signals for each virtual transmission path, and based on the channel estimation result for the training signal received in the leading area of each wireless packet (step S6705), it is linear The second reception weight matrix is calculated (step S6706), and signal detection processing is performed using this (step S6707). Then, the base station apparatus repeats signal detection processing while data is spread over a plurality of symbols (step S6708), and determines whether or not the frame ends after the data end (step S6709), and the frame continues. The above process (signal reception from another terminal station apparatus) is repeated. At the end of the frame, the base station receives a slot for transmitting a training signal from the terminal station side at the end of the frame (step S6710), and based on the received signal, calculates a transmission / reception weight vector for the terminal station. This is recorded and managed, and the processing is ended (step S6711).

  Next, transmission processing will be described. When the base station apparatus acquires frame timing (step S6721), the terminal station apparatus transmits a training signal for calculating the reception weight vector used by the first signal processing unit in the slot at the head of the basic frame (step S6722). . Then, the base station apparatus recognizes the transmission timing of the radio packet to the terminal station apparatus grasped because of base station centralized control (step S6723), and, from the memory, the plurality of first local stations with respect to the terminal station apparatus. The transmission weight vector is read out by the signal processing unit 304 (step S6724).

  Subsequently, after generating the transmission signal (step S6725), the base station apparatus multiplies the first transmission weight vector for each first signal processing unit (step S6726), and performs signal transmission (step S6727). When the data continues, the base station device repeats the above processing (step S 6728), and when the data ends, determines whether the frame is ended (step S 6729), and the frame is ended. If not, the next transmission processing or the reception processing of FIG. 67 (a) is repeated. If the frame ends, the process ends.

  Next, with reference to FIG. 68, the transmission / reception signal processing operation of the terminal station apparatus in the present embodiment will be described. FIG. 68 is a flowchart showing transmission / reception signal processing operation of the terminal station apparatus. In FIG. 68, (a) shows reception processing and (b) shows transmission processing. First, the reception process will be described. The terminal station apparatus acquires frame timing (step S6801) and receives the training signal transmitted by the base station apparatus in the slot at the head of the basic frame (step S6802), and based on this, the first reception weight vector is virtualized. The calculation is performed for each target transmission path (for each of the plurality of first signal processing units) (step S6803). The terminal station apparatus receives a wireless packet from the base station apparatus (step S6804), reads the reception weight vector in the plurality of first signal processing units 304 of the terminal station apparatus from the memory, and generates virtual reception signal vectors. The reception weight vector corresponding to the transmission path is multiplied (step S6805).

  Next, the terminal station apparatus aggregates one system of signals for each virtual transmission path, and generates a linear signal based on the channel estimation result (step S6806) for the training signal received in the leading area of each wireless packet. The second reception weight matrix is calculated (step S6807), and the signal detection processing is performed using this (step S6808). The terminal station apparatus repeats the signal detection process in the same manner while the data passes over a plurality of symbols (step S6809). After the end of data, the terminal station apparatus determines whether the end of the frame is reached (step S6810). If the frame continues, the above-described reception processing is repeated if there is an assignment to the own station. The process ends when the frame ends.

  Next, transmission processing will be described. The terminal station apparatus acquires a frame timing (step S6821), performs calibration processing on the reception weight vector obtained by the above-described reception processing, and calculates a first transmission weight vector (step S6822). Then, the terminal station apparatus recognizes the transmission timing of the own station (step S6823), generates a transmission signal (step S6824), and then multiplies the signal for each virtual transmission path by the first transmission weight vector. (Step S6825) The result of multiplication is subjected to addition synthesis for each antenna element to perform signal transmission (step S6826).

  Next, the terminal station apparatus repeats the above processing when the data continues (step S6827), and when the data ends, determines whether or not the frame ends (step S6828), and the frame If it has not been completed, the next transmission process is repeated if there is an assignment for the own station. If the frame is over, the terminal station apparatus transmits a training signal in the slot and subcarrier allocated to the local station for the training signal at the end of the frame (step S6829), and ends the processing.

  Although details are omitted here, the above processing is basically performed for each subcarrier on the frequency axis. The above-described process is the case of the configuration shown in FIG. 69 as a precondition. FIG. 69 is a diagram showing a frame configuration as a precondition. In this figure, reference numerals 257-1 to 257-3 represent training signals, and 258-1 to 258-3 represent data payloads. In the description of FIGS. 67 and 68 described above, first, channel estimation for signal separation of MIMO transmission is performed on signal sequences for each of a plurality of virtual transmission paths, which are aggregated by multiplying the first reception weight vector. The training signals 257-1 to 257-3 are used for this. Using uplink and downlink adaptively in a time division manner, as it is not clearly indicating whether the training signals 257-1 to 257-3 and the data payloads 258-1 to 258-3 are uplink or downlink. It is possible. For example, the training signals 257-1 to 257-2 and the data payloads 258-1 to 258-2 are downlink signals (the base station transmits), and the training signal 257-3 and data payload 258-3 are uplink signals. (A terminal station may transmit), all may be downlink, and all may be uplink. In any case, since the training signals 257-1 to 257-3 are signals with secured line gain multiplied by the first transmission weight vector regardless of uplink or downlink, they are high at a sufficient reception level. Channel estimation is possible with accuracy.

  On the other hand, since the training signals 251-1 to 251-3 are all signals for which directivity formation has not been performed by the first transmission weight vector, the channel is channeled although transmission power is concentrated on a small number of subcarriers. The estimation accuracy is relatively low. However, it is possible to calculate the second reception weight matrix using the training signals 251-1 to 251-3 as long as channel estimation accuracy without any problem in operation can be secured. In this case, the training signals 257-1 to 257-3 are omitted as shown in FIG. 70 instead of FIG. 69, and communication is performed using only the data payloads 258-1 to 258-3. FIG. 70 is a diagram showing an example in which the training signal is omitted and communication is performed only with the data payload.

  Training signals 257-1 to 257-3 shown in FIG. 69 are for channel estimation for signal separation in MIMO transmission, and for example, if the number of spatial multiplexing is large, training signals orthogonal to that number are required. Yes, and the overhead of the corresponding number of symbols is necessary. However, in FIG. 70, when channel estimation for signal separation in MIMO transmission is performed using the training signals 251-1 to 251-3 at the beginning of a frame as much as possible, spatially multiplexed signal sequences are clearly separated. It is sufficient to perform only channel estimation necessary for signal detection and demodulation processing of the independent signal sequence, and all training signals are simultaneously transmitted (for example, in one symbol in the case of OFDM), and on the receiving side, as if after signal separation. Reception signal processing can also be performed as if it is a signal of SISO (Single Input Single Output). In this sense, also in FIG. 70, a training signal for signal detection and demodulation of the SISO signal may be included.

  Next, with reference to FIG. 71, the processing operation of another second reception weight matrix calculation in the present embodiment will be described. FIG. 71 is a flowchart showing the processing operation of another second reception weight matrix calculation. This processing operation is applicable to both the processing of the base station apparatus in the uplink and the processing of the terminal station apparatus in the downlink. Also, in the case of the uplink, it is performed on the training signal transmitted from the terminal station device at the end of the frame, and in the case of the downlink, it is performed on the training signal transmitted from the base station device at the head of the frame. Or you may implement with respect to the training signals 257-1 to 251-3 shown in FIG. Here, it demonstrates as operation | movement of a base station apparatus.

  First, when receiving a training signal (step S7101), the base station apparatus calculates a first reception weight vector for each virtual transmission path (step S7102). The first reception weight vector may be any one of the slots 253-1 to 253-3, 254-1 to 254-3, 255-1 to 255-3, and 256-1 to 256-3 and the slots thereof. The reception weight vector calculated using any subcarrier group may be substituted. The base station apparatus then multiplies the training signal used to calculate the first reception weight vector by the first reception weight vector for each virtual transmission path (step S7103). A signal sequence of the number of virtual transmission paths is acquired by this multiplication. Since reception of the training signal for each virtual transmission line is divided into subcarriers for each virtual transmission line, crosstalk components between the respective virtual transmission lines can also be extracted individually, and therefore, it is possible to individually The extracted channel estimation result corresponds to each component of the channel matrix of equation (55). The second reception weight matrix is calculated based on the channel matrix obtained in this manner (step S7104).

For example, the case where the number of virtual transmission paths is three will be described as an example. As an example, the first virtual transmission path is a multiple of three subcarriers, the second virtual transmission path is a multiple of three + 1 subcarriers, and the third virtual transmission path is a multiple of three + 2 sub-carriers. It is assumed that a training signal is transmitted using a carrier. At this time, attention is paid to a signal obtained by multiplying each subcarrier component of the reception signal by the reception weight of the first virtual transmission path. Focusing on 3 multiple subcarriers, since the first receiving weight vector may perform channel estimation of the desired signal after the multiplication, h ₁₁ components of the channel matrix of 3 × 3 can be obtained.

On the other hand, focusing on subcarriers of multiples of 3 + 1, it is possible to extract signal components that the signal of the second virtual transmission line leaks into the first virtual transmission line, so h of the 3 × 3 channel matrix ₁₂ ingredients can be obtained. Similarly, focusing on subcarriers of multiples of 2 and 3, it is possible to extract signal components that the signal of the third virtual transmission line leaks into the first virtual transmission line, so h ₁₃ of the 3 × 3 channel matrix You can get the ingredients.

Similarly, attention is paid to the signal multiplied by the reception weight of the second virtual transmission line. Focusing on subcarriers of multiples of 3 + 1, since channel estimation of the desired signal after multiplication of the second reception weight vector can be performed, h ₂₂ components of the 3 × 3 channel matrix can be obtained. On the other hand, focusing on subcarriers of multiples of three, signal components of the signal of the first virtual transmission line leak into the second virtual transmission line can be extracted, so h ₂₁ of the 3 × 3 channel matrix You can get the ingredients. Similarly, focusing on subcarriers of multiples of 2 and 3, it is possible to extract signal components that the signal of the third virtual transmission line leaks into the second virtual transmission line, so h ₂₃ of the 3 × 3 channel matrix You can get the ingredients.

Furthermore, similarly, attention is paid to the signal multiplied by the reception weight of the third virtual transmission line. Focusing on subcarriers of multiple of 3 and 2 subcarriers, channel estimation of the desired signal after multiplication of the third reception weight vector can be performed, so _h33 components of the 3 × 3 channel matrix can be obtained. On the other hand, focusing on subcarriers of multiples of three, it is possible to extract signal components that the signal of the first virtual transmission line leaks into the third virtual transmission line, so h ₃₁ of the 3 × 3 channel matrix You can get the ingredients. Similarly, focusing on subcarriers of multiples of 3 + 1, it is possible to extract signal components that the signal of the second virtual transmission line leaks into the third virtual transmission line, so h ₃₂ of the 3 × 3 channel matrix You can get the ingredients.

  In this way, all 3 × 3 matrix elements can be obtained. Here, the case where the number of virtual transmission paths is three has been described as an example, but it is possible to acquire a channel matrix for any number of virtual transmission paths as well. If these channel matrices can be obtained, the signal processing from that point onward is similar to space-multiplexed signal processing using an ordinary MIMO channel. It is possible to carry out 1000 signal detection by arbitrary signal processing including non-linear signal processing such as MLD. In the above description of FIGS. 67 and 68, although an example in the case of using a linear reception weight matrix has been shown, it is of course possible to apply signal detection processing of any MIMO channel as described above.

  Here, an example is shown in which orthogonalization is achieved by dividing subcarriers used for training signals for each virtual transmission path, but if it is possible to apply other orthogonalized training signals to each signal sequence to be spatially multiplexed, Similar signal processing can also be performed using training signals 257-1 to 257-3 shown in FIG. 69 using the orthogonality. For example, if OFDM is used with 3 OFDM symbols and the training signals for each signal sequence are shifted in time, it is naturally possible to perform channel estimation using all subcarriers.

(Utilization of time axis beam forming)
As described above in the term of time axis beamforming, in general, when the antenna element is present in a very narrow area, the frequency dependency of the reception weight for each antenna element (that is, each component of the reception weight vector) Is expected to be relatively small. In order to increase the degree of signal separation between the virtual transmission paths, it is possible to align the complex phases of the reception signals of the respective antenna elements with the respective subcarriers and add and combine them by using the optimum reception weight as much as possible. preferable. However, if crosstalk components between virtual transmission paths can be separated with high accuracy using the second stage reception weight matrix in the latter stage, the major problem is that the accuracy of the first reception weight vector is low to some extent. It does not.

  For example, consider the addition of Sin (θ) and Sin (θ + δ). The following relational expression (76) is obtained from the formula of the trigonometric function.

  This equation means that if the efficiency at the time of addition is incomplete in-phase synthesis and a phase error δ exists, the in-phase synthesis causes the amplitude to be 2 × Cos (δ / 2) instead of 2 times. Do. Taking δ as an example at 30 degrees, the amplitude becomes Cos (30/2) = 0.965..., resulting in a gain loss of about 3.4%. That is, if the error is within 30 degrees for all antennas, the loss of gain due to this error is only about 0.3 dB. Even if δ is 60 degrees, it is only about 1.2 dB with a gain loss of about 13.4%.

  Therefore, even in the case where F (α) in FIG. 44 is not in a range exceeding 30 dB, the first signal processing unit 304 performs the signal separation processing on the frequency axis at the subsequent stage. Even if the beamforming on the time axis is performed with low accuracy, it means that there is no problem if it can tolerate a limited gain reduction. That is, after receiving weight vectors on the frequency axis with the training signals 251-1 to 251-3 in FIG. 65, IFFT processing is performed on the reception weight for each frequency for each element of the receiving weight vector, on the time axis. Convert to weights. Then, using only the value of this leading component (corresponding to the line-of-sight component), a reception weight vector on the time axis is constructed. In each sampling value, when the reception signal vector of the sampling value constituted by the sampling value of each antenna element is multiplied by the reception weight vector of the time axis, the sampling signal in the state where the line gain is enhanced by directivity control is virtually transmitted. It can be acquired for each road. In this way, it is possible to obtain a received signal in a state in which timing detection is possible.

  It should be noted that if the speed of the channel time variation is relatively slow with respect to the basic frame period, relative channel information extracted from a plurality of signals (ie, If channel information using relative complex phase based on the complex phase of the reference antenna is averaged, it is also possible to suppress noise components and improve channel estimation accuracy. If averaging is not performed, it is also possible to use absolute channel information instead of relative channel information as it is.

  Next, with reference to FIG. 72, another signal processing operation in the case of applying time-axis beamforming to the present embodiment will be described. FIG. 72 is a flowchart showing another signal processing operation of time axis beamforming. In the description of the fifth embodiment, in consideration of the fact that the SNR between one antenna element on the transmitting side and one antenna element on the receiving side is significantly short in line design, the equation (52) is used. ) Has been described, but according to the techniques shown in the sixth, seventh, and eighth embodiments, one antenna element on the transmitting side and one antenna element on the receiving side Since it becomes possible to obtain channel information between antenna elements with high accuracy on the frequency axis, channel information on the time axis used for time axis beamforming by IFFT processing using channel information on the frequency axis It will be possible to get Here, although the process of the terminal station apparatus in the downlink is shown as an example, the same process is also possible in the uplink. First, the base station apparatus transmits a training signal for channel estimation in the frame start area (step S7201). The terminal station apparatus receives this training signal and performs received signal processing (step S7211). As described above, in this signal processing, a training signal having a length slightly longer than the OFDM symbol period not including a guard interval is assumed, and a sampling signal cut out with an appropriate FFT window even with rough frame timing as a reference Perform FFT processing using.

  Next, the terminal station apparatus performs channel estimation on the signal subjected to the FFT processing with reference to a known training signal, and channel information for each virtual transmission path (or for each virtual transmission path for the reference antenna). Get relative channel information). The channel information obtained here is in a state in which information is present only in specific subcarriers for each virtual transmission line, so any method such as linear interpolation may be used to obtain unacquired subcarrier components. Similarly, the channel estimation result is obtained (step S7212). Further, based on the acquired channel information, a reception weight vector on the frequency axis is calculated (step S7213).

  Next, the terminal station apparatus applies IFFT processing to the subcarriers of the weight information of each antenna element (step S 7214). Then, the terminal station apparatus extracts the value of the leading component (corresponding to the line of sight wave component) in the weight information on the time axis obtained as a result, and calculates the reception weight on the time axis of each antenna element, Reception weight vectors having these components as vectors are acquired for each virtual transmission path (step S7215). Calibration processing is performed on the reception weight vector of this time axis (step S7216). Subsequently, the terminal station apparatus calculates transmission weight vectors, and records and manages transmission / reception weight vectors on the time axis of these first stages (step S7217).

  The above description is the time axis weight calculation operation of the first stage, but the frequency axis weight calculation of the second stage is performed as follows. At the stage of calculating the receiving weight vector on the time axis (step S7215), the terminal station apparatus uses the sampling value for each antenna element as an element for each sampling value of the above received training signal used for the receiving weight calculation. The reception signal vector (sampling value) to be received is multiplied by the reception weight vector for each virtual transmission path, and the signal sequence is separated for each virtual transmission path (step S 7221). Subsequently, the terminal station apparatus individually performs FFT processing on signal sequences of sampling values corresponding to the obtained number of virtual transmission paths, and performs channel estimation with reference to a known training signal (step S7222). ). The channel estimation result here is that only the subcarrier to which the training signal is assigned for each virtual channel is the channel estimation result of the desired virtual channel, and the subcarrier in which the training signal is not present is from different virtual channels. Represents the crosstalk component of

  Here, the case of extracting and using the value of the leading component (corresponding to the line-of-sight component) in the weight information on the time axis has been introduced, but the time axis of the delay component of one sample shown in equation (57) The reception weight of can be calculated based on the second component subjected to IFFT, and the reception weight on the time axis of the delay component of two samples shown in equation (59) can be calculated based on the third component subjected to IFFT. Similarly, it is possible to use more delayed wave components.

Next, _let h j i ^(k) be channel information in which the terminal station apparatus leaks the signal transmitted to the i th virtual transmission path in the k th subcarrier to the j th virtual transmission path. Since information is present only in subcarriers, interpolation processing is performed on subcarriers that have been missed by linear interpolation etc., and a complete channel matrix including crosstalk components from different virtual transmission paths is acquired (Step S7223). Then, the terminal station apparatus calculates a reception weight matrix for signal separation based on the reception weight matrix obtained in this way (step S7224), which is used as the reception weight matrix on the second stage frequency axis. Are recorded and managed as (step S7225).

  The time axis transmission / reception weight vector of the first stage and the frequency axis reception weight matrix of the second stage acquired as described above can be continuously used in the subsequent frames.

  As described above, in obtaining transmission / reception weights for using a virtual transmission line corresponding to the first singular value in the line-of-sight environment, channel estimation is performed with a small number of subcarriers, the antenna element spacing is narrow, and the line-of-sight wave is When dominant, it is possible to estimate the transmission / reception weights of the remaining subcarriers by linear interpolation or the like, using the characteristic that the frequency dependence of channel information is reduced. For this reason, it becomes possible to perform channel estimation simultaneously and in parallel in a form in which subcarriers do not overlap in a plurality of wireless stations. In order to do this efficiently, a training signal for channel estimation by the base station apparatus and the individual terminal station apparatus in the frame is used as a fixed slot in the frame configuration with periodicity and on the same subcarrier. In order to avoid interference with each other, the timing and subcarriers are divided, slot allocation is performed, and channel feedback is periodically performed in that slot.

  In the training signal for performing feedback of channel information in the prior art, it is general to assign the identification number of the transmitting station or the like to the payload part following the training signal. Moreover, since it is general that the frequency dependency of channel information is large, it has been designed so that estimation of channel information for each antenna can be performed on all subcarriers. These overheads are very large and cause significant loss of MAC efficiency. However, in the eighth embodiment, since this training signal is assigned to a known fixed location (slot and subcarrier) for each base station apparatus or terminal station apparatus, the identification number of the transmitting station, etc. is given to the subsequent payload. There is also a feature that it is possible to simultaneously perform multiple channel estimation and channel feedback simultaneously while dividing subcarriers by one slot.

The ninth embodiment
The above description of the eighth embodiment has focused on the case where channel information or transmission / reception weights have frequency dependency. However, as shown in the fifth embodiment, it is possible to extend even when channel information or transmission / reception weights are not frequency-dependent.

In the fifth embodiment described above, channel information is acquired from the evaluation of the correlation value on the time axis of the received training signal by using, for example, the equation (52). (52) means that if there is no frequency dependency in the channel information, S _j (n) S ₁ (n) ^* at each sampling time in Σ in equation (52) excludes the noise component For example, since all the values become constant values, noise components are averaged with N _FFT sampling values, and as a result, relative channel information (correlation value for each antenna) can be obtained with high channel estimation accuracy.

  However, in the eighth embodiment, since signals from different antenna elements are mixed on the frequency axis, the correlation between received signals in which signals from different terminal stations and different antenna elements are mixed is determined as a specific transmitting antenna. The correlation value in the state in which signals from different terminal stations and different antenna elements are mixed is acquired instead of the correlation for each receiving antenna of the signal from the signal, and a specific terminal station apparatus or a specific It is not possible to extract correlation values for antenna elements. The state where signals from different terminal stations and different antenna elements are mixed can be easily separated by performing FFT. In this case, the channel estimation result of each acquired subcarrier is It is required that the channel estimation accuracy is sufficiently high (ie, sufficient SNR characteristics).

  In the sixth, seventh, and eighth embodiments described above, the channel estimation accuracy is improved by limiting the number of subcarriers used as an example to 4 as an example. If there is no frequency dependency, it is possible to further average these channel estimation results and obtain common channel information for all subcarriers by the average value.

Here, c _j (f _k ) represents channel information on the frequency axis of subcarrier f _k . Alternatively, relative channel information based on the complex phase of the reference antenna may be used.

Also, here, taking the case of using four subcarriers as an example, a coefficient of 1⁄4 is explicitly stated for the purpose of averaging in the four used subcarriers f _k, but the number of other subcarriers is Average according to the number. The channel information obtained as a result is equivalent to that obtained by equation (52), and the reception weight is calculated using equation (50) or equation (51).

  The same applies to the seventh embodiment in addition to the eighth embodiment, but a case where a training signal including four subcarriers is used as a training signal for acquiring channel information is illustrated and described. However, the number of subcarriers is not four and may be any other number of subcarriers. As the ultimate condition, it is possible to set the number of used subcarriers to one.

  In the fifth embodiment described above, the spacing between the antenna elements of each first signal processing unit on the base station apparatus side is narrowed to create a situation where the frequency dependency of channel information or transmission / reception weights is very small, and as a result The common channel information or transmission / reception weight is used for subcarriers. Although it is possible to expect suppression of measurement error by using multiple subcarriers, there are some merits to using a single subcarrier as an ultimate condition.

  As a training signal using only a single subcarrier, if the purpose is to acquire relative channel information for each antenna element (ie, without using a purpose such as timing detection), temporarily using OFDM However, it is not necessary to give a guard interval, but to use continuous transmission of a mere sine wave (that is, a non-modulated signal) of the frequency of the allocated subcarrier. The feature of this signal is that the ratio PAPR (Peak to Average Power Ratio) of the temporal variation of the average transmission power and the instantaneous transmission power is 1 because the signal is completely constant in amplitude. .

  Generally, when OFDM is used, the amplitude of the signal fluctuates with time in order to combine independent signals for each subcarrier even when using a modulation scheme with a constant amplitude such as QPSK. Therefore, PAPR has a relatively large value, and the peak transmission power has a very large value compared to the average transmission power. Since the linearity of the high power amplifier is limited, when the peak power exceeds the linear region of the high power amplifier, non-linear distortion occurs to deteriorate the communication characteristics. Therefore, in OFDM, a transmission power margin called a so-called backoff is estimated, and the transmission power is reduced by the margin to perform signal transmission.

  In terms of channel design, for example, in a system using 1024 point FFT (using about 1000 as an effective subcarrier), it is necessary to expect a backoff of at least about 10 dB. However, if the sine wave is completely constant in amplitude, it is possible to increase the effective transmission power by the backoff value and transmit, since it is not necessary to anticipate such backoff. Note that raising the transmission power by this back-off value can also be used except when using a single subcarrier, and for example, even when using 4 subcarriers, the initial phase of each subcarrier is adjusted to an arbitrary value. By configuring the training signal so as to minimize PAPR in one symbol period, it is possible to operate with a lower backoff value according to the PAPR value at that time.

  When transmitting the training signal, when the minimum back-off value used in other data communication is x dB, it is desirable to set the back-off value at the time of transmitting the training signal to (x / 2) dB or less. Specifically, it is desirable to set the backoff value at the time of transmission of the training signal to 0 dB.

As a result, for example, if transmission power of 1000 subcarriers is focused on one subcarrier, it becomes possible to gain 30 dB (= ₁₀ Log ₁₀ 1000) of channel gain there, and a circuit of approximately 10 dB of backoff Channel estimation in good condition with gain added is possible. In other words, an improvement of about 40 dB can be expected. Originally, in the high frequency band such as the millimeter wave band, the line design was severe because of the high frequency. For example, if the frequency is 10 times, the increase in loss due to the frequency dependent term of the free space propagation loss is 20 dB. In addition, the output of the high power amplifier also decreases relatively.

  Furthermore, since Massive MIMO requires 100 or more RF devices to be mounted, the price of each device is expected to be reduced, and the use of inexpensive component materials also reduces the linearity of the amplifier. As a result, there is a problem that the channel estimation accuracy is lowered unless the channel gain of 30 dB or more is additionally secured, but a channel in which transmission power is increased by focusing on one subcarrier or four subcarriers and suppressing the PAPR value. The estimation can solve these problems.

  When channel estimation is performed using only one subcarrier per one antenna element, actually, under the influence of various reflected wave components, the channel estimation result in that subcarrier is channel information based on only the line-of-sight component. May indicate different values. However, as shown in the seventh embodiment, the wireless station side transmitting the training signal is provided with a plurality of antenna elements, and a plurality of antenna elements transmit unmodulated continuous signals of different subcarriers. Then, the wireless station receiving the training signal will receive training signals of a plurality of subcarriers. If relative channel information is obtained from the obtained channel information, the influence of different transmitting antenna elements is relatively small, and the relative channel information obtained by the plurality of antenna elements is substantially constant on the frequency axis. It should be considered. If averaging processing is performed using this feature, relative channel information on the time axis in a form in which the reflected wave components that are affected differently for each subcarrier are relatively suppressed and in effect the line of sight wave component is extracted It becomes possible to calculate the transmission-and-reception weight of a time-axis based on this.

  Although the averaging process here basically intends to average the complex phase, it is expected that the amplitude is generally constant and that the fluctuation range of the complex phase is also expected to be small. The channel information itself given by complex numbers may be simply averaged on the complex plane.

Tenth Embodiment
[Multi-user MIMO transmission using virtual transmission line corresponding to first singular value]
(Outline of Basic Principle according to Tenth Embodiment)
In the above description, for example, as shown in FIG. 16, each first signal processing unit 304 transmits and receives a single signal sequence between a single terminal station apparatus 302 and a base station apparatus 303, and Spatial multiplex transmission is performed as a whole using the signal processing unit 304 of FIG. This corresponds to communication between the base station apparatus 303 and the terminal station apparatus 302 using single-user MIMO transmission. However, as a matter of course, the present invention is also applicable to the case of performing space multiplex transmission (that is, multi-user MIMO) between a plurality of terminal stations 302 and the base station 303.

  In normal multi-user MIMO, when performing signal transmission simultaneously on the same frequency to a plurality of terminal stations, null formation is performed on the downlink so that mutual signals do not interfere with each other's terminal stations. I was doing signal transmission while doing. However, the null formation can be broken if there is time variation of the channel and residual interference can occur. This is because the transmission weight used for transmission directivity control is calculated based on the result of channel feedback performed in a predetermined cycle, but in order to suppress the overhead accompanying the feedback of this channel, feedback of the channel is carried out so frequently. It can not be performed, and as a result, the transmission weight can not but be calculated based on the channel information acquired in the past. Therefore, no matter how precisely the transmission weight is calculated, it is meaningless if there is a fluctuation of the channel with the passage of time, and conversely, a transmission method that is less susceptible to the fluctuation of the channel is required.

  When the virtual transmission line corresponding to the first singular value shown in the first embodiment is actively used, the value of the line gain obtained on the transmission line is on another transmission line using a reflected wave. When using a large number of antenna elements on both the base station side and the terminal side, which is relatively higher than the gain, it is considered that an additional line gain can be obtained according to the product of the number of antenna elements. Then, it is possible to obtain a desired SIR characteristic without intentionally forming perfect null control.

  FIG. 73 is a diagram showing configurations of a base station apparatus and a terminal station apparatus at the time of applying multi-user MIMO in the tenth embodiment. The difference from FIG. 16 is that the terminal station apparatus is only the terminal station apparatus 302 in FIG. 16, but in FIG. 73 the terminal station apparatus is added to the terminal station apparatus 302 and the terminal station apparatuses 308 and 309 are added. In each of the first signal processing units 304-1 to 304-4 of the base station apparatus 303, virtual transmission paths corresponding to the first singular value are formed toward the terminal station apparatuses 302, 308, and 309, respectively. Multi-user MIMO transmission.

  In the prior art, the plurality of first signal processing units 304-1 to 304-4 have not been implemented, so when multiple signal sequences are simultaneously spatially multiplexed to a single terminal station apparatus, One transmission / reception signal processing circuit performs multi-user MIMO signal processing using different transmission / reception weight vectors. Here, the transmission side needs to perform complete null control so that mutual interference does not occur between the terminal stations.

On the other hand, in the embodiment of the present invention, unlike the conventional multi-user MIMO, each of the first signal processing units 304-1 to 304-4 uses one transmission / reception weight vector for one terminal station apparatus. It only performs signal processing for a minute. For example, one first signal processing unit 304-1 transmits signals for three channels in total to one terminal station apparatus 302, 308, and 309. At this time, transmission weight vectors directed to virtual transmission paths corresponding to the first singular values for the respective terminal station apparatuses 302, 308, and 309 are respectively designated v ₁₁ , v ₁₂ , and v ₁₃ . These are not orthogonal to each other because they are transmission weight vectors for individual single-user MIMO as described above. In contrast, transmission weight vectors v actually used for multi-user _{MIMO '11, v' 12,} v '13 , the vector v' ₁₁ are as orthogonal to _{v 12} and _{v 13,} the vector v _{'12 v 11} The vector v ′ ₁₃ is set to be orthogonal to v ₁₁ and v _{12 so as} to be orthogonal to v ₁₃ and v ₁₃ . Specifically, the transmission weight vector v ′ ₁₁ for the terminal station device 302 is expressed by the following equation (78) using Gram-Schmidt orthogonalization.

The same process may be performed on the terminal station devices 308 and 309 to obtain v ′ ₁₂ and v ′ ₁₃ . The above is the description for the first signal processing unit 304-1, but the same processing may be performed for the first signal processing units 304-2 to 304-3. In this manner, the first signal processing unit transmission weight vector v _'11 ~v' ₁₃ for the 304-1, the first transmission to the signal processing section 304-2 weight vectors v _'21 ~v' _23, the first The transmission weight vectors v ′ _{31 to} v ′ ₃₃ for the signal processing unit 304-3 and the transmission weight vectors v ′ _{41 to} v ′ ₄₃ for the first signal processing unit 304-4 may be obtained. By the above processing, the first signal processing units 304-1 to 304-4 can transmit signal sequences of three systems, respectively, and can perform space multiplex transmission of twelve systems in total. The supplement Here, unlike the conventional multi-user MIMO, for example, j ≠ transmission weight vector v _'11 and v' _2j for one becomes j is not orthogonal, become a complete null formation Not. However, since these are different combinations of the first signal processing unit 304 of the transmission source and the terminal stations to receive different combinations, mutual interference is relatively small because they are virtual transmission paths with small correlation. As described above, this reason is attributed to the high directivity gain of the pinpoint that comes from the product of the base station device and the many antenna elements of the terminal station device, and the point that the absolute value of the first singular value exhibits a high gain. Do. Therefore, the orthogonality of the collateral transmission weight vector v _'11 and v' _2j are become unnecessary.

The same processing may be performed similarly to the processing on the base station apparatus side in the uplink. That is, although the above-mentioned formula (78) shows the orthogonalization process regarding a transmission weight vector, the completely same orthogonal relation is applied to a reception weight, and the 1st singularity with respect to each terminal station apparatus 302, 308, 309 The reception weight vectors for virtual transmission paths corresponding to the values are replaced with v ₁₁ , v ₁₂ , and v ₁₃ respectively, and the reception weight vectors v ′ ₁₁ , v ′ ₁₂ , and v ′ ₁₃ used for multi-user MIMO are represented by vector v _'11 so as to orthogonal to the _{v 12} and _{v 13,} the vector v' as the ₁₂ orthogonal to the _{v 11} and _{v 13,} it Rure be set so that orthogonal to the vector v _{'13 v 11} and _{v 12} . Equation (78) can be applied as it is to the reception weight calculation at this time. Also, the wireless station apparatus side may be completely the same in processing and circuit configuration as the wireless station apparatus in the first embodiment described above without being aware of the fact that multi-user MIMO is employed.

  The circuit configuration of the base station apparatus 303 in this case needs to be changed as compared with FIG. FIG. 74 is a first transmission corresponding to the first transmission signal processing units 181-1 to 181-4 of the base station device 70 corresponding to the base station device 303 at the time of multi-user MIMO application in the embodiment of the present invention. FIG. 7 is a diagram illustrating a circuit configuration of a signal processing unit 182. In FIG. 74, as shown in FIG. 73, the number of terminal stations to be simultaneously spatially multiplexed is three as multiuser MIMO transmission. The configuration of FIG. 74 is similar to that of FIG. 6, but the first transmission signal processing circuits 113-1 to 113-3 correspond to three terminal station apparatuses that perform space multiplexing simultaneously, and further, The transmission weight processing unit 150 generates and manages transmission weight vectors for multi-user MIMO used for transmission on a virtual transmission line corresponding to the first singular value with respect to the terminal stations of three stations that simultaneously perform spatial multiplexing. Specifically, in the transmission weight processing unit 150, in the channel information acquisition circuit 151, the channel information acquired by the reception unit is separately acquired via the communication control circuit 120, and the channel information storage circuit is sequentially updated. Store in 152. At the time of signal transmission, according to an instruction from communication control circuit 120, multi-user MIMO (MU-MIMO) transmission weight calculation circuit 153 reads channel information corresponding to the destination station from channel information storage circuit 152, and based on the read channel information. The transmission weight vector is calculated as in the above equation (78). Here, it should be noted that in the case of single-user MIMO, the first right singular vector is calculated and used as a transmission weight vector as it is, but in the case of multi-user MIMO, each is calculated within the same first signal processing unit. In order to orthogonalize the transmission weight vector between the terminal stations, for example, from the first right singular vector for the terminal station 302, the component of the first right singular vector for the terminal station 308 and the first for the terminal station 309 The components of the right singular vector are subtracted, and signal transmission is performed in a form orthogonal to each other. The multiuser MIMO transmission weight calculation circuit 153 outputs the transmission weights calculated in this manner to the first transmission signal processing circuits 113-1 to 113-3.

Next, FIG. 75 corresponds to the first received signal processing units 185-1 to 185-4 of the base station apparatus 70 corresponding to the base station apparatus 303 in multi-user MIMO application in the tenth embodiment. The circuit configuration of the reception signal processing unit 186 of FIG. Also in FIG. 75, as shown in FIG. 73, the number of terminal stations to be simultaneously spatially multiplexed is three as multi-user MIMO transmission. The configuration of FIG. 73 is similar to that of FIG. 7, but the first received signal processing circuits 114-1 to 114-3 correspond to three terminal station apparatuses that perform spatial multiplexing simultaneously, and further, The reception weight processing unit 170 generates and manages reception weight vectors for multi-user MIMO used for reception on a virtual transmission line corresponding to the first singular value with respect to terminal stations of three stations that simultaneously perform spatial multiplexing. Specifically, in accordance with an instruction from communication control circuit 120 in reception weight processing section 170, known signal for channel estimation separated into each subcarrier based on the information inputted from FFT circuit 857 in channel information estimation circuit 172. The antenna elements of each of the terminal station apparatuses 60 corresponding to the terminal station apparatuses 302, 308, and 309 and the antennas of the base station apparatus 70 corresponding to the base station apparatus 303 from (a preamble signal or the like added to the beginning of the wireless packet) Channel information to and from element 851 is estimated for each subcarrier, and the estimation result is output to multi-user MIMO reception weight calculation circuit 173. The multiuser MIMO reception weight calculation circuit 173 calculates, for each subcarrier, a reception weight vector to be multiplied based on the input channel information. Here, equation (78) describes the process using transmission weight vector v in the case of single-user MIMO, but if this is similarly replaced with reception weight vector u in the case of single-user MIMO, equation (78) is used as it is. Is applicable. At this time, the reception weight vector for combining the signals received by the antenna elements 851-1 to 851-N _MT-Ant differs depending on the terminal station apparatus 60, and the first reception corresponding to the terminal station apparatus 60 to be extracted The signal processing circuits 114-1 to 114-3 are individually calculated. Although the second received signal processing unit 75 in the latter stage of this signal processing removes the crosstalk component between the signal sequences from the same wireless station apparatus, the terminal station apparatus is processed in the process of the former stage. There is no need to be aware of multi-user MIMO when calculating the reception weight matrix here, since it is expected that mutual interference components between them can be sufficiently reduced. MIMO signal detection processing is individually performed for each wireless station apparatus. You should do it. However, since different wireless station apparatuses require separate MIMO signal detection processing, it is necessary to provide the second received signal processing unit 75 in parallel for the wireless station apparatuses that perform spatial multiplexing simultaneously. .

  Thus, the information between the first transmission signal processing unit 181 and the second transmission signal processing unit 71, and the information between the first reception signal processing unit 185 and the second reception signal processing unit 75. Although there is a difference in that the system is extended to a plurality of systems, the point that space multiplex transmission is performed with a single terminal station apparatus using the plurality of first transmission signal processing units 181 and the first reception signal processing unit 185 Therefore, the configuration is different from that of the prior art.

  As described above, the wireless communication system 90 according to the tenth embodiment includes the base station device 303 (first wireless station device) and a plurality of terminal station devices such as the terminal station device 308 (second wireless station device). And The base station apparatus 303 according to the tenth embodiment performs signal processing of wireless communication associated with a plurality of first signal processing units 304 having a first antenna element group and a plurality of first signal processing units. A second signal processing unit 305 to be executed (strictly, it includes other base station apparatus functions such as an interface circuit, a MAC layer processing circuit, a communication control circuit, etc.). The terminal station device 308 according to the tenth embodiment includes a second antenna element group and a transmitting / receiving unit that performs wireless communication with the first signal processing unit via the second antenna element group. The same applies to the terminal station device 302 and the terminal station device 309.

  The first signal processing unit 304 of the tenth embodiment transmits the transmission weight vector of the MIMO channel matrix used for wireless communication between the first antenna element group and the second antenna element group as the first of the MIMO channel matrix. Calculated based on at least one of the first right singular vector and the first left singular vector corresponding to one singular value, or at least one of the approximate solution of the first right singular vector and the approximate solution of the first left singular vector . The first signal processing unit 304 generates a transmission signal for each terminal station apparatus, and generates a signal vector by multiplying the generated transmission signal by a transmission weight vector. The first signal processing unit 304 adds and synthesizes signal vectors for terminal stations that are space-multiplexed at the same time. The first signal processing unit 304 transmits a signal based on the additively synthesized signal vector from the first antenna element group to the plurality of terminal stations at the same time using the same frequency channel.

  The first signal processing unit 304 according to the tenth embodiment uses the reception weight vector of the MIMO channel matrix used for wireless communication between the first antenna element group and the second antenna element group as the first of the MIMO channel matrices. Second based on at least one of a first right singular vector and a first left singular vector corresponding to one singular value, or at least one of an approximate solution of the first right singular vector and an approximate solution of the first left singular vector Calculated for each wireless station device of The first signal processing unit 304 multiplies the signal vector based on the signal received via the first antenna element group by the reception weight vector for each terminal station apparatus, thereby forming one system of signal for each terminal station apparatus. Generate a series. The second signal processing unit 305 removes the residual interference component from the signal based on the generated signal sequence.

  That is, the first signal processing unit transmits the second transmission weight vector for the MIMO channel matrix used for wireless communication between the first antenna element group and the second antenna element group for each second wireless station apparatus. At least one of the first right singular vector and the first left singular vector corresponding to the first singular value of the MIMO channel matrix for each wireless station apparatus, or an approximate solution of the first right singular vector and the first left singular vector At least one of a first right singular vector and a first left singular vector corresponding to a first singular value of another second wireless station apparatus spatially multiplexing at the same time based on at least one of the approximate solutions, or It is calculated to be orthogonal to at least one of the approximate solution of the first right singular vector and the approximate solution of the first left singular vector. The first signal processing unit generates a transmission signal for each second wireless station apparatus, and generates a transmission signal vector by multiplying the generated transmission signal by a transmission weight vector. The first signal processing unit adds and synthesizes the transmission signal vectors for the second wireless station apparatus spatially multiplexing at the same time, and generates a plurality of signals based on each component of the added and synthesized signal vector from the corresponding first antenna element To the second radio station apparatus of

  In addition, the first signal processing unit may be configured to perform second reception weight vectors for the MIMO channel matrix used for wireless communication between the first antenna element group and the second antenna element group for each second wireless station apparatus. At least one of the first right singular vector and the first left singular vector corresponding to the first singular value of the MIMO channel matrix for each wireless station apparatus, or an approximate solution of the first right singular vector and the first left singular vector At least one of a first right singular vector and a first left singular vector corresponding to a first singular value of another second wireless station apparatus spatially multiplexing at the same time based on at least one of the approximate solutions, or It is calculated to be orthogonal to at least one of the approximate solution of the first right singular vector and the approximate solution of the first left singular vector. The first signal processing unit multiplies each of the second wireless station apparatuses by a reception weight vector by a signal vector based on a signal received via the first antenna element group, for each second wireless station apparatus. Generates a single signal sequence. The second signal processing unit removes the residual interference component from the signal based on the generated plurality of signal sequences for each first signal processing unit.

  Thus, the base station apparatus 70, the terminal station apparatus 60, the wireless communication system 90, and the wireless communication method of the tenth embodiment can increase the transmission capacity by MIMO in an environment where the line-of-sight environment is dominant. The base station device 70, the terminal station device 60, the wireless communication system 90, and the wireless communication method according to the tenth embodiment can increase transmission capacity in an environment where line-of-sight environment is dominant even when performing multi-user MIMO. It becomes possible.

Eleventh Embodiment
[Mobile front hall configuration method using virtual transmission line corresponding to the first singular value]
(Outline of Basic Principle according to Eleventh Embodiment)
In the fifth embodiment, using time axis beamforming, sampling data of one digital baseband is multiplied by the transmission weight for each antenna system and each sampling value, and thereby it is possible to virtually transmit the first singular value. The directivity was formed for transmission on the transmission line. Since multiplication of the transmission weight here can be performed directly on the sampling data, as in the configuration of the mobile front hall shown in FIG. 12A, generation of sampling data of digital baseband is performed on the BBU side. It is possible to implement all of the signal processing described above, transmit sampling data of one system of digital baseband on an optical fiber, and carry out the remaining processing on the RRH side. On the RRH side, transmission weight multiplication processing for time axis beamforming is required, but this transmission weight information is transferred on an optical fiber, and on the RRH side, the transferred transmission weight is converted to sampling data of digital baseband. Since only multiplication and transmission are performed, all processes dependent on the wireless communication system are basically implemented on the BBU side. The only factor in RRH that depends on the wireless communication system is that transmission directivity control does not have frequency dependence, and it is possible to form a very narrow directivity beam by means of a multi-element antenna to communicate substantially. Means to do. The directional beam is preferably a beam directed to the communication partner station in the line-of-sight environment, but if a very strong reflected wave arrives even if the line-of-sight does not occur, It is possible to cope by forming a directed directional beam. In a situation where very many reflected waves are gathered together, resulting in a reasonable reception level, the line gain of the virtual transmission line corresponding to the first singular value is degraded compared to the line-of-sight environment. However, originally there is a tendency to use an antenna element with high directivity gain because it is originally in a high frequency band where the distance attenuation is large and the line design is strict, and as a practical matter there is rarely It is expected to be. Therefore, especially as a mobile front hole in a wireless system using a high frequency band, it is possible to use a large scale antenna while performing time axis beamforming by using a virtual transmission path corresponding to the first singular value. It is possible to reduce the information transmission capacity on the fiber.

  The information capacity at the time of transferring this transmission weight information on the optical fiber is estimated. For example, in the case of using OFDM, the number of FFT points at that time is set to 1024, for example. In this case, assuming a guard interval of 256 samples, for example, the number of samples of one OFDM is 1280 samples. On the other hand, in the case of using an antenna element of 256 elements, for example, 256 pieces of transmission weight information are required (since the transmission weight vector has 256 dimensions). Assuming that the number of bits of one piece of transmission weight information and the number of bits of one sample are the same, the transmission weight information corresponds to information of 20%. That is, even in the case of transmitting transmission weight information every time for every OFDM symbol, it will be 20% more than the configuration of the mobile front hole shown in FIG. 12 (a). This is overwhelmingly less than the information amount of 256 times in the configuration of the mobile front hall shown in FIG. 13 (a), and is almost the same as the configuration shown in FIG. 12 (a). Furthermore, it is possible to further compress the information amount of the transmission weight information by transmitting the transmission weight information every several OFDM symbols instead of transmitting the transmission weight information every every OFDM symbol. . Alternatively, the transmission weight information may be transferred only when the transmission weight information is changed. In this way, the time-base beam for consolidating the signal processing function depending on the wireless transmission method on the BBU side and configuring the virtual transmission path corresponding to the first minimum singular value necessary on the RRH side It becomes possible to make it the composition which mounts a forming function.

  FIG. 76 is a diagram showing an outline of function sharing of the mobile front hall using a virtual transmission path corresponding to the first singular value in the eleventh embodiment. Here, only the functions related to the transmission of signals (BBU to RRH direction) regarding the direction from the network side to the user are extracted. In the figure, reference numeral 401 is a MAC layer processing circuit, reference numeral 402 is a transmission signal processing circuit, reference numeral 403 is a time axis signal generation circuit, reference numeral 454 is an optical interface circuit, reference numeral 455 is an optical fiber, reference numeral 456 is an optical interface circuit, reference numeral 457 Is a time axis transmission weight multiplication circuit, 427 is a D / A converter, 428 is an RF processing circuit, 429 is an antenna element, 432 is a transmission weight processing circuit, 441-4 is a BBU, 442 is an Represents RRH. The MAC layer processing circuit 401, the transmission signal processing circuit 412, the time axis signal generation circuit 403, and the time axis transmission weight multiplication circuit 457 collectively constitute an area 446-1 and an area 446-2 for performing baseband signal processing related to radio.

17 and 47 and FIG. 76, the MAC layer processing circuit 78 of FIG. 17 corresponds to the MAC layer processing circuit 401 of FIG. The second transmission signal processing unit 71 in FIG. 17 corresponds to the transmission signal processing circuit 412 in FIG. The IFFT & GI application circuit 813 in FIG. 47 corresponds to the time axis signal generation circuit 403. The first transmission signal processing circuit 311 in FIG. 47 corresponds to the time axis transmission weight multiplying circuit 457 in FIG. The D / A converters 814-1 to 814-N ' _BS-Ant in FIG. 47 correspond to the D / A converter 427 in FIG. The mixers 816-1 to 816-N ' _BS-Ant , filters 817-1 to 817-N' _BS-Ant , and high power amplifiers 818-1 to 818-N ' _BS-Ant in FIG. 47 correspond to the RF processing circuit 428. Do. The antenna elements 819-1 to 819 -N ′ _BS-Ant correspond to the antenna element 429. The first transmission weight processing unit 330 in FIG. 47 corresponds to the transmission weight processing circuit 432 in FIG.

  In FIG. 76, when a signal to be transmitted to BBU 441-1 is input from the network side, MAC layer processing circuit 401 performs signal processing of the MAC layer, and the frame format used for transmission and reception in the wireless section flows through the network side. It converts and terminates the frame format of data, and inputs the signal of the format of the wireless packet to the transmission signal processing circuit 412. The transmission signal processing circuit 412 performs transmission signal processing of a wireless signal. Here, the transmission scheme in the wireless section is not particularly limited. For example, if OFDM is used, error correction coding, interleaving, modulation processing for each subcarrier, and the like are performed as needed. The signal generated in this manner is converted into a time axis signal by the time axis signal generation circuit 403. For example, in the case of OFDM as described above, IFFT is performed to convert a signal on the frequency axis into a signal on the time axis, insert a guard interval, and perform waveform shaping processing between symbols. As a result, each sampling value of the digital baseband signal is converted into a continuous signal in time series. On the other hand, for example, the reception weight information and the like collected in the reception weight processing circuit on the reception side of the BBU not shown here are transmission weight processing via (or directly) the communication control circuit etc. A circuit 432 is provided. The transmission weight processing circuit 432 performs calibration processing on the reception weight information and the like to calculate the transmission weight. Each sampling value of the digital baseband signal generated by the time axis signal generation circuit 403 and the transmission weight information calculated by the transmission weight processing circuit 432 are converted by the optical interface circuit 454 into a predetermined frame format, and the electric signal is converted to light. The signal is converted and output to the optical fiber 455.

  The signal output to the optical fiber 455 is transmitted to the RRH 442-4, and the RRH 442-4 converts the optical signal into an electrical signal in the optical interface circuit 456, terminates a signal of a predetermined format, and a digital baseband signal The transmission weight information is further reproduced as the sampling value information sequence. The sampling value and the transmission weight information are both input to the time axis transmission weight multiplication circuit 457 as the digital baseband signal. The time base transmission weight multiplication circuit 457 multiplies the sampling value as the digital baseband signal by the transmission weight for each antenna system in sampling value units. Each sampling value multiplied by the transmission weight is input to the D / A converter 427. The D / A converter 427 converts the signal for each antenna system into an analog baseband signal at a predetermined clock rate, and the RF processing circuit 458 In each antenna system, the up converter converts the signal into a radio frequency signal, filters out the out-of-band radiation signal in each antenna system, and amplifies it in a high power amplifier for each antenna system. Send.

  As described above, the functions equivalent to the base station apparatus for wireless communication as a whole are divided and accommodated in the BBU 441-4 and the RRH 442 that are mediated by an optical fiber. The feature here is that since wireless digital baseband signal processing is integrated in BBU 441-4 provided in the network-side station building, even if there is any change in the wireless communication system, all can be BBU 441-4. There is a merit that only the side change is sufficient.

  The above description corresponds to the case where there is only one first transmission signal processing unit 181 in FIG. 17, but it goes without saying that the same configuration can be made also in the case where a plurality of first transmission signal processing units 181 are provided. It is possible. FIG. 77 is a diagram showing an outline (downlink) of function sharing of the mobile front hall using virtual transmission paths corresponding to a plurality of first singular values in the eleventh embodiment. Here, only functions relating to the transmission of signals (direction from BBU 441-5 to RRHs 442-4a and 442-4b) in the direction (downlink) from the network side to the user are extracted. In the figure, reference numeral 461 is a MAC layer processing circuit, reference numeral 462 is a transmission signal processing circuit, reference numerals 403a to 403b are time axis signal generation circuits, reference numerals 454a to 454b are optical interface circuits, reference numerals 455a to 455b are optical fibers, reference numeral 456a. 456a is an optical interface circuit, 457a to 457b are time axis transmission weight multiplication circuits, 427a to 427b are D / A converters, 428a to 428b are RF processing circuits, 429a to 429b are antenna elements, and 470 is an antenna element Reference numeral 441-5 denotes a BBU, and reference numerals 442-4a to 442-4b denote RRHs.

  In FIG. 77, the same components as those in FIG. 76 are denoted by the same reference numerals. As RRHs 442-4a to 442-4 b in FIG. 77 have the same plural systems as RRH 442-4 in FIG. 76, suffixes a and b are added for identification. Similarly, the time axis signal generation circuits 403a to 403b, the optical interface circuits 454a to 454b, and the optical fibers 455a to 455b are used as many as the number of RRHs 442 to 442b as in FIG. The signal processing is the same as that of FIG. 76, but the MAC layer processing circuit 461 and the transmission signal processing circuit 462 process signal sequences for the number of RRHs 442-4a to 442-4b. For example, if spatial multiplexing is performed for one terminal station apparatus using MIMO, the MAC layer processing circuit 461 processes the MAC layer as a signal of one terminal station apparatus, while the transmission signal processing circuit 462 performs space processing. It will be serial-to-parallel converted into a plurality of signals in consideration of multiplexing, and transmit signal processing will be performed on each. The transmission weight processing circuit 470 may, for example, receive weight information or the like collected in the reception weight processing circuit on the reception side of the BBU not shown here via the communication control circuit etc. (not shown here). ) To get information. Processing such as calibration is performed based on the reception weight information for the plurality of systems, transmission weights are calculated, and transmission weight information is transferred to RRHs 442-4a to 442-4b corresponding to the transmission weights. The transmission weight processing circuit 470 may acquire the transmission weight in the same manner as the first transmission weight processing units 330 and 340 in the fifth embodiment.

  Next, FIG. 78 is a diagram showing an outline (uplink) of function sharing of the mobile front hall using virtual transmission paths corresponding to a plurality of first singular values in the eleventh embodiment. Here, only functions related to transmission of signals (direction from RRH 444-1a and 444-1b to BBU 443-1) in the direction (uplink) from the user side to the network are extracted. In the figure, reference numeral 471 is a MAC layer processing circuit, reference numeral 472 is a reception signal processing circuit, reference numerals 474a to 474b are optical interface circuits, reference numerals 475a to 475b are optical fibers, reference numerals 476a to 476b are optical interface circuits, reference numerals 473a to 473b. Is a time axis reception weight multiplication circuit, 477a to 477b are A / D converters, 478a to 478b are RF processing circuits, 479a to 479b are antenna elements, 480 is a reception weight processing circuit, and 481a to 481b are correlations. A calculation circuit, reference numeral 443-1 indicates BBU, and 444-1a to 444-1b indicate RRH. Although FIG. 78 describes a plurality of RRHs 444-1 a to 444-1 b as corresponding to FIG. 77, it may be only one system as in FIG.

  In FIG. 78, when antenna elements 479a to 479b receive a signal, the RF processing circuits 478a to 478b amplify the signal with a low noise amplifier for each antenna system, and a mixer multiplies the signal by a common local oscillator signal to obtain a signal Are down-converted and out-of-band signals are filtered out to generate an analog baseband signal for each antenna system. This signal is sampled by A / D converters 477a to 477b to generate a data string of sample values of digital baseband signals for each antenna system. On the other hand, the time axis reception weight multiplication circuits 473a to 473b multiply the reception weights to perform addition combination (the reception signal vector is multiplied by the reception weight vector), and convert it into one digital baseband signal. On the other hand, in the A / D converters 477a to 477b, data strings of sample values of the digital baseband signal are also output to the correlation calculation circuits 481a to 481b in accordance with each other.

The correlation calculation circuit 481A～481b, the sampled values of the received signals of the second to the _{N Ant} antenna elements in the antenna elements 479A～479b, a complex conjugate value of the sampling values of the received signal of the first antenna element Multiplication is performed, and the operation result of a predetermined period is added. Similarly, multiplying the complex conjugate of the sample values in the sampled values of the received signals of the first to N _Ant antenna elements of the antenna elements 479A～479b, adds the calculation result of the predetermined time period. Based on the above results, the correlation calculation is performed using equation (52), and the reception weight of each antenna element is calculated by taking its complex conjugate. The correlation calculation circuits 481a to 481b are supplied with timing indication signals from the reception weight processing circuit 480 on the BBU 443-1 side via the optical interface circuits 474a to 474b, the optical fibers 475a to 475b, and the optical interface circuits 476a to 476b. Be done.

  Each time the correlation calculation circuits 481a to 481b receive the timing signal, the correlation calculation circuits 481a to 481b perform the correlation operation using Expression (52), and reset the memory value of each addition. The correlation calculation circuits 481a to 481b collectively transfer the correlation information acquired here to the reception weight processing circuit 480 via the optical interface circuits 476a to 476b, the optical fibers 475a to 475b, and the optical interface circuits 474a to 474b. Do. The reception weight processing circuit 480 calculates reception weight information based on the correlation information received here, records and manages the calculated reception weight information, and transmits correlation information or reception weight information to the transmission weight processing circuit 470 as necessary. It notifies, and performs calibration processing etc. here and uses for transmission weight calculation.

  Further, the reception weight processing circuit 480 grasps the scheduling information of the reception processing of the BBU 443-1 and the RRHs 442-4a to 442-4b (for example, by the communication control circuit not shown here or the MAC layer processing circuit 471). Determine the timing at which the reception weight should be switched by grasping with reference to scheduling information to be managed etc.), and at which timing to switch the reception weight, the optical interface circuits 474a to 474b, the optical fibers 475a to 475b, the optical interface The correlation calculation circuits 481a to 481b are output via the circuits 476a to 476b. This timing instruction takes into consideration the time lag caused by passing through the optical interface circuits 474a to 474b, the optical fibers 475a to 475b, and the optical interface circuits 476a to 476b. When the correlation calculation circuits 481a to 481b receive this switching timing instruction, they output reception weight information calculated based on the correlation information calculated immediately before that to the time axis reception weight multiplication circuits 473a to 473b. The time axis reception weight multiplying circuits 473a to 473b continue to multiply the reception signal instructed at the end by the reception weight information until the switching instruction is given.

FIG. 79 is a diagram showing an outline of the correlation detection circuit 481 (481a, 481b) in the eleventh embodiment. In the figure, reference numerals 491-2 to 491-N _Ant and 495-1 to 495-N _Ant are multipliers, reference numerals 492-2 to 492-N _Ant and reference numerals 496- 1 to 496-N _Ant are adders, Reference numerals 493-2 to 493-N _Ant and 497-1 to 497-N _Ant denote memories, reference numerals 498-1 to 498-N _Ant denote square root acquisition circuits, and reference numeral 494 denotes a correlation operation control unit. The correlation calculation circuit 481 is connected to the A / D converter 477, the time axis reception weight multiplication circuit 473, and the optical interface circuit 476. Furthermore, the reception weight processing circuit 480 on the BBU 443-1 side is also connected via the optical interface circuit 476, the optical fiber 475, and the optical interface circuit 474.

When the sampling value of each antenna system is input from the A / D converter 477, the correlation calculation circuit 481 multiplies the sampling value of the reception signal of the first to N- _ant antenna elements in the antenna element 479. vessel in _{491-2~491-N Ant,} multiplied by the sampled value of the second to _{N Ant} antenna elements in the first complex conjugate value of the sampling values of the antenna and the antenna element 479 of the antenna element 479, a multiplication result Are output to the adders 492-1 to 492-N _Ant . Adder _{492-2~492-N Ant} adds the value of the input value and the memory _{493-2~493-N Ant,} stored by entering the addition result to the memory _{493-2~493-N Ant} Let By sequentially rotating the calculated values between the adders 492-2-492 -N _Ant and the memories 493-2-493 -N _Ant , the correlation results are sequentially added, and the accumulated value is determined. When an instruction to reset the correlation value is input from the reception weight processing circuit 480 on the BBU 443-1 side to the memories 493-2 to 493-N _Ant via the optical interface circuit 474, the optical fiber 475, and the optical interface circuit 476, the memory 493-2 to 493-N _Ant outputs the accumulated value to the correlation calculation control unit 494, and resets its own value to zero.

When sampling values of each antenna system are input from the A / D converter 477 in parallel with the above-described processing, sampling values of reception signals of the first to N- _ant antenna elements in the antenna element 479 are multipliers. _495-1~495-N to _Ant is input, the sampling value of the _{1 to N Ant} antenna of the antenna element 479, the complex conjugates and are multiplied by the multiplier _{495-1~495-N Ant,} sampling The square of the absolute value of the value is output to the adders 496-1 to 496 -N _Ant . Adder _{496-1~496-N Ant} adds the value of the input value and the memory _{497-1~497-N Ant,} stored by entering the addition result in the memory _{497-1~497-N Ant} Let By rotating the calculated values sequentially between the adders 496-1 to 496-N _Ant and the memories 497-1 to 497-N _Ant , the square value of the absolute value of the sampling value is sequentially added, and the accumulated value is calculated. It will be sought.

When an instruction to reset the correlation value is input from the reception weight processing circuit 480 on the BBU 443-1 side to the memories 497-1 to 497-N _Ant via the optical interface circuit 474, the optical fiber 475, and the optical interface circuit 476, the memory _{497-1~497-N Ant} outputs the accumulated value to the square root acquisition circuit _{498-1~498-N Ant,} reset their values to zero. The square root obtaining circuits 498-1 to 498 -N _Ant obtain the square root of the accumulated value, and output the value to the correlation operation control unit 494. The correlation calculation control unit 494 calculates the reception weight by performing correlation calculation according to equation (52) and further taking its complex conjugate. The reception weight information aggregates information on all antenna systems, and transfers the information to the reception weight processing circuit 480 via the optical interface circuit 476, the optical fiber 475, and the optical interface circuit 474. Similarly, this reception weight is also input to the time axis reception weight multiplication circuit 473.

  FIG. 80 is a diagram showing an outline (uplink) of function sharing of another mobile front hall using virtual transmission paths corresponding to a plurality of first singular values in the eleventh embodiment. Here, only functions relating to transmission of signals (direction from RRH 444-2a to 444-2b to BBU 443-2) regarding the direction (uplink) from the user side to the network are extracted. In the figure, reference numerals 484a to 484b indicate optical interface circuits, reference numerals 485a to 485b indicate optical fibers, reference numerals 486a to 486b indicate optical interface circuits, reference numeral 482 indicates reception weight processing circuits, reference numerals 481a to 481b indicate correlation calculation circuits, reference numerals 443- 2 represents BBU, and 444-2a to 444-2b represent RRH.

  Further, in the figure, the same reference numerals are given to the same components as those in FIG. The difference between FIG. 80 and FIG. 78 is the reception weight input to the time base reception weight multiplication circuit 473, from the reception weight processing circuit 482 on the BBU 443-2 side to the optical interface circuit 484, the optical fiber 485, the optical interface circuit Is a point to notify through. Therefore, it is possible to instruct to use the reception weight acquired in the past managed by the reception weight processing circuit 482 without necessarily using the reception weight calculated with reference to the training signal received immediately before. There is.

  As another variation, in FIG. 78, the correlation calculation circuits 481a and 481b have reception weight storage / management functions implemented, and instead of notifying reception weight information itself with the optical fiber 475 as shown in FIG. 80, correlation calculation is performed. The identification number etc. of the reception weight managed by the circuit 481 is notified from the BBU 443-1 to the RRHs 444-1a and 444-1b, and the reception weight information stored in the correlation calculation circuits 481a and 481b based on the identification number May be read out and output to the time axis reception weight multiplying circuit 473.

  Next, an embodiment in the case of applying the mobile fronthaul configuration in this embodiment to multi-user MIMO will be described. FIG. 81 is a diagram showing an outline (downlink) of function sharing in multi-user MIMO application of mobile front hall using virtual transmission paths corresponding to a plurality of first singular values in the eleventh embodiment. . Here, only the functions related to the transmission of signals (direction from BBU 441-6 to RRH 442-6) in the direction (downlink) from the network side to the user are extracted. In the figure, reference numeral 463 denotes a transmission signal processing circuit, reference numeral 464 denotes a transmission weight processing circuit, reference numeral 465 denotes an addition synthesis circuit, reference numeral 441-6 denotes a BBU, and reference numeral 442-6 denotes an RRH.

  In the figure, the same reference numerals as in FIG. 77 denote the same parts. In the description so far, basically, a plurality of RRHs may perform space multiplex communication as single-user MIMO with one terminal station apparatus, or as mutual interference is limited, communication with different terminal station apparatuses is performed. It has been explained that it may be done. On the other hand, FIG. 81 is expanded to multi-user MIMO in which signal transmission is performed from one RRH 442-6 to a different terminal station apparatus, and FIG. While explaining. However, although FIG. 74 does not necessarily presuppose time-axis beamforming, it will be described here based on the example of FIG. 74 in the case of performing time-axis beamforming.

In the configuration of first transmission signal processing section 181 of base station apparatus 70 corresponding to base station apparatus 303 at the time of multi-user MIMO application shown in FIG. The transmission signal processing circuit 463 in FIG. 81 carries out the processing corresponding to the transmission signal processing circuits 113-1 to 113-3. Further, the adding and combining circuit 465 in FIG. 81 carries out the processing corresponding to the adding and combining circuits 812-1 to 812-N ' _BS-Ant in FIG. Further, the processing corresponding to the transmission weight processing unit 150 in FIG. 74 is implemented by the transmission weight processing circuit 464 in FIG. Further, the outputs of the time axis transmission weight multiplying circuits 457a to 457b of the plurality of RRHs 442-4a to 442-b in FIG. 77 are input to the adding and combining circuit 465. The addition synthesis circuit 465 adds and synthesizes signals transmitted by the same antenna element, and outputs the addition result to the D / A converter 427a. The addition / synthesis circuit 465 combines two transmission signals into one signal.

  Further, in the description of FIG. 77, the two RRHs 442-5a to 442-b may transmit signals to one terminal station apparatus or may transmit signals to different terminal station apparatuses. Because of MIMO, each different signal sequence is to be transmitted to a different terminal station apparatus. Except for the above differences, basically the same signal processing as that of FIG. 77 is performed. Here, with regard to the optical interface circuits 454a to 454b, the optical fibers 455a to 455b, and the optical interface circuits 456a to 456b, the BBU 441-6 and the RRH 442-6 are connected via physically different optical fibers and optical interface circuits. Although an example of the case has been shown, if optical fibers capable of realizing large-capacity transmission are used, it is possible to physically exchange information by aggregating them in the same optical interface circuit and the same optical fiber. In this sense, what is shown in FIG. 81 may be understood as a “logical transmission line”.

  FIG. 82 is a diagram showing an outline (uplink) of function sharing in multi-user MIMO application of mobile front hall using virtual transmission paths corresponding to a plurality of first singular values in the eleventh embodiment. . Here, only functions related to transmission of a signal (direction from RRH 444-3 to BBU 443-3) in a direction (uplink) from the user side to the network side are extracted. In the figure, reference numeral 466 denotes a reception signal processing circuit, reference numeral 467 denotes a reception weight processing circuit, reference numeral 443-3 denotes a BBU, and reference numeral 444-3 denotes an RRH. In the figure, the same reference numerals as in FIG. 80 denote the same parts. Also in the description here, the extension to multi-user MIMO in which signals from different terminal stations are received by one RRH will be described using FIG. 75 for explaining multi-user MIMO. However, although FIG. 75 does not necessarily presuppose time-axis beamforming, it will be described here based on an example of FIG. 75 in the case of performing time-axis beamforming.

  In the configuration of first received signal processing section 186 of base station apparatus 70 corresponding to base station apparatus 303 at the time of multi-user MIMO application shown in FIG. 75 in order to support multi-user MIMO, the first of FIG. The time axis reception weight multiplying circuit 483 in FIG. 82 carries out the processing corresponding to the reception signal processing circuits 114-1 to 114-3. In the fifth embodiment, the first received signal processing unit 385 of the base station apparatus performs signal processing for extracting a virtual transmission path corresponding to the first singular value of the MIMO channel matrix. Further, as in the tenth embodiment, when multi-user MIMO is applied, the first reception signal processing unit 186 performs signal processing to extract a virtual transmission path corresponding to the first singular value of the MIMO channel matrix. In the first embodiment, interference components that could not be completely removed by this signal processing are removed by the second received signal processing unit 75 in the subsequent stage. On the other hand, in the eleventh embodiment, second-stage signal processing is performed to remove interference components that the reception signal processing circuit 466 could not completely remove. By performing two-step signal processing in this manner, it is possible to perform signal separation efficiently.

  The processing corresponding to the reception weight processing unit 170 in FIG. 75 is performed by the reception weight processing circuit 467 in FIG. Furthermore, while in FIG. 75 the signal is output to only one time axis reception weight multiplying circuit 473 in one RRH 444-1, a plurality of outputs from one A / D converter 477a are shown in FIG. Are output to the time axis reception weight multiplying circuits 483a to 483b.

  The plurality of time axis reception weight multiplying circuits 483a to 483b multiply the reception weight vectors on the time axis corresponding to different terminal stations by the same reception signal vector. The multiplication results are input to the reception signal processing circuit 466 via the optical interface circuits 486a to 486b, the optical fibers 485a to 485b, and the optical interface circuits 484a to 484b. Except for the above difference, the same signal processing is performed basically in the same configuration as that of FIG. Here, with regard to the optical interface circuits 486a to 486b, the optical fibers 485a to 485b, and the optical interface circuits 484a to 484b, the RRH 444-3 and the BBU 443-3 are connected via physically different optical fibers and optical interface circuits. Although an example of the case has been shown, if optical fibers capable of realizing large-capacity transmission are used, it is possible to physically exchange information by aggregating them in the same optical interface circuit and the same optical fiber. In this sense, what is shown in FIG. 82 may be understood as a “logical transmission line”.

  In the wireless communication system of the eleventh embodiment, the BBU includes the optical interface circuit 454 as the first transmission means and the third transmission means. The optical interface circuit 454 transmits the sampling data sequence generated by the time axis signal generation circuit 403 and transmission weight information (transmission weight vector) to the RRH. In addition, the BBU includes reception weight processing circuits 480, 482, and 467 as reception weight acquisition means. The reception weight processing circuit acquires reception weight information (reception weight vector) that the RRH uses for reception from the terminal station apparatus. The reception weight information is calculated based on the correlation value calculated in the RRH. Further, the reception weight processing circuit instructs the RRH to calculate the correlation value and the reception weight information.

  Further, in the radio communication system of the eleventh embodiment, the RRH includes a time axis reception weight multiplying circuit 473 as a second transmission means. The time-base reception weight multiplication circuit 473 sequentially multiplies the reception sampling signal vector train, which is a data train of sample values of the digital baseband signal output from the A / D converter 477, for each sampling data, The multiplication results are added and synthesized to generate one system sampling data sequence, and the one system sampling data sequence is transmitted to the BBU through the optical interface circuit 456. The RRH includes an optical interface circuit 456 as a fourth transmission unit. The optical interface circuit 456 collectively transmits the correlation information calculated by the correlation calculation circuit 481 for all the antenna sequences, to the BBU via the optical interface circuit 476.

  In the radio communication system of the eleventh embodiment, the RRH includes optical interface circuits 456 and 476 as the first receiving means and the second receiving means. The optical interface circuit 456 receives transmission weight information (transmission weight vector) and a sampling data string from the BBU. The optical interface circuit 476 receives reception weight information (reception weight vector) from the BBU. The RRH includes a correlation calculation circuit 481 as reception weight calculation means. The correlation calculation circuit 481 uses any one of the plurality of antenna elements 479 as a reference antenna element based on the signal when the known signal transmitted from the terminal station apparatus is received by each of the antenna elements 479, and the reference antenna element A correlation value is calculated between the known signal received in and the known signal received by the other antenna element 479 over a predetermined number of samples, and a reception weight vector is calculated based on the calculated correlation value.

  According to the above correspondence, according to the radio communication system in the eleventh embodiment, also in the downlink from the BBU to the RRH, the uplink from the RRH to the BBU, and also the extension to the multiuser MIMO. Therefore, the basic aim of the mobile front hall can be achieved with an impact that the amount of information slightly increases compared with the conventional one without increasing the information capacity flowing on the optical fiber by providing the multi-element antenna in the RRH. It becomes possible to optimize the function allocation as it is maintained, and it is possible to realize band expansion in a situation where the RRH and the terminal station device are in a line-of-sight environment and the direct wave is dominant.

  In the radio communication system according to the eleventh embodiment, the optical fiber is used as the communication medium between the BBU and the RRH. A line may be used.

[Supplemental Matters Concerning Relationships of Various Embodiments]
First, in the embodiment of the present invention, it is common to spatially multiplex a plurality of signal sequences by utilizing virtual transmission paths corresponding to the plurality of first singular values shown in the first embodiment as the basis thereof. It is characterized. The relationship with each specific embodiment is organized as follows.

(Combination with the Second Embodiment of the Present Invention)
The second embodiment of the present invention is a specific embodiment regarding optimization of the arrangement interval of the first signal processing unit on the base station apparatus side in order to efficiently realize the first embodiment of the present invention. It is shown. In the case of the fourth embodiment, since the distance between the base station apparatus and the train changes with time, the optimization conditions shown in the second embodiment can not be used directly, but other embodiments are possible. In the second embodiment, it is possible to utilize the second embodiment. Specifically, in the third embodiment, since the satellite base station apparatus and the control base station apparatus are fixedly installed, the satellite base station apparatus is regarded as the first signal processing unit, and the control base station apparatus is considered. If the satellite base station apparatus is assumed to be a terminal station apparatus, and spatial multiplexing transmission is performed between the satellite base station apparatus and the general base station apparatus using virtual transmission paths corresponding to a plurality of first singular values, The third embodiment and the second embodiment are arranged linearly if they are arranged so that the arrangement condition satisfies a predetermined condition with respect to the average distance between the general base station apparatus and the satellite base station apparatus. Operation combining the form and the first embodiment is possible. Further, as opposed to the satellite base station apparatus of the third embodiment as shown in FIG. 83, the first signal processing shown in the second embodiment for the average distance between the first signal processing unit and the terminal station apparatus on the ground. By optimizing the arrangement interval of parts, although a good orthogonal relationship can be maintained although it deviates from the perfect orthogonal condition, it is possible to stabilize the communication between each terminal station apparatus and the base station apparatus. become. Therefore, even when the present technology is applied to an access system, an operation combining the second embodiment and the first embodiment is possible. The eighth embodiment and the tenth embodiment are also applicable in terms of application to the access system, and an operation combining the second embodiment and the first embodiment with these embodiments is also possible. It is.

  In the description of the second embodiment, for example, as illustrated in FIG. 29 in which the antenna element on the base station apparatus side is replaced with a parabolic antenna, it is assumed that the terminal station apparatus has a plurality of antenna elements. However, the base station apparatus side does not necessarily have to include the multi-element antenna group as shown in the first signal processing unit of the first embodiment, and it is possible to use a single super high gain antenna element substantially. The terminal station apparatus side can form a virtual transmission path corresponding to the first singular value by combining with a plurality of antenna elements.

(Combination with the Third Embodiment of the Present Invention)
As described above, in the case of the fourth embodiment, since the present invention is specialized to direct communication between the terminal station apparatus on the train side and the base station apparatus, the third embodiment of the present invention performs two-hop relay transmission. Although it is hard to think of a combination with the form, it is possible to combine the third embodiment and the second embodiment of the present invention, and in the other operation conditions of the fourth embodiment, it is possible to combine with the other embodiments. It is applicable also about a combination. For example, the tenth embodiment can be applied to the third embodiment so that the first embodiment can be extended to multi-user MIMO transmission as well as single-user MIMO transmission as shown in the tenth embodiment. And the first embodiment can be combined to realize multi-user MIMO while utilizing the satellite base station apparatus of the third embodiment.

(Combination with the Fourth Embodiment of the Present Invention)
As described above, the fourth embodiment of the present invention is a kind of unique application form, but here, the signal processing of directivity formation on the time axis shown in the fifth embodiment may be performed in combination, It is also possible to use in combination with the sixth and seventh embodiments of the present invention in the process of channel estimation. Here, to the end, one-to-one communication between one terminal station apparatus on the train side and the base station apparatus is described as an example, and in this sense, single-user MIMO transmission is fundamental, but, for example, In the case where a plurality of trains exist simultaneously in the same section, in addition to changing the frequency channel for each train and dividing it, if combining and using multi-user MIMO shown in the tenth embodiment, Communication with a plurality of trains can be realized in parallel on the same channel, which makes it possible to effectively utilize frequency resources.

(Combination with the fifth embodiment of the present invention)
The intended technology of the fifth embodiment of the present invention can be applied to a system of single carrier transmission as a wireless communication system, a system of multi carrier such as OFDM, or a system of any condition. In the case of a system based on signal processing on the frequency axis such as OFDM, the direct merit of the application of the fifth embodiment of the present invention is small (however, the number of implemented FFT circuits and IFFT circuits can be reduced). Possible), for example, in the case of mounting a large-scale antenna on the RRH side in the mobile front hole as in the eleventh embodiment, even in a multi-carrier system such as OFDM, it is transmitted on an optical fiber In order to compress the information capacity and extend it to a large-scale antenna, the signal on the time axis is transmitted on the optical fiber, and on the RRH side, the time axis beamforming shown in the fifth embodiment is used. In this sense, the eleventh embodiment is premised on using the fifth embodiment (and the first embodiment) in combination.

(Combination with the sixth and seventh embodiments of the present invention)
The sixth and seventh embodiments of the present invention are embodiments showing means for feedback of channel information used for directivity formation, and can be used in combination in all the embodiments of the present invention. This can be used also in the typical fixed installation-type terminal station apparatus used in the description of the first embodiment, and is also applicable to an environment in which the channel varies with time in the access system. In the eighth embodiment of the present invention, a specific configuration method for simultaneously performing channel estimation among a plurality of adjacent cells or a plurality of terminal stations in the same cell, especially assuming use in an access system This is an embodiment that assumes a combination of the sixth embodiment and the seventh embodiment of the present invention. Furthermore, channel estimation between the terminal station apparatus on the train side and the base station apparatus shown in the fourth embodiment in which channel gain in channel estimation is usually insufficient can also be used to improve channel gain, It is also possible to use the fourth embodiment and the seventh embodiment (and the first embodiment) of the present invention in combination with the fourth embodiment.

  In particular, as a specific embodiment in which the sixth embodiment and the seventh embodiment are directly combined, training on discrete subcarriers without overlapping of subcarriers between antenna elements of a terminal station apparatus or a base station apparatus is performed. When allocating to a signal, relative channel information acquired for a plurality of subcarriers from a plurality of different antenna elements, assuming that each antenna element of the terminal station apparatus or the base station apparatus is transmitted from the same virtual antenna element On the other hand, relative channel information including subcarriers to which no training signal is assigned is acquired using the least squares method or the like, and the reception weight (and cancellation of the complex phase difference of the relative channel information based on this relative channel information) It is possible to calculate the transmission weight by calibration processing.

(Combination with the eleventh embodiment of the present invention)
The eleventh embodiment of the present invention is similar to the mode in which the first signal processing unit and the second signal processing unit in the first embodiment of the present invention are separated, and an interval between them is operated using an optical fiber. However, in the first embodiment, various types of signal processing of transmission and reception directivity formation are concentrated in the first signal processing unit, whereas in the eleventh embodiment, the second signal processing unit corresponds to the second signal processing unit. It has been consolidated on the BBU side. However, in the case where space multiplex transmission is performed by actively utilizing virtual transmission paths corresponding to a plurality of first singular values, the basic principle is shared, and in this point the eleventh embodiment Operation corresponding to the combination of the fifth embodiment and the first embodiment (in the sense that the virtual transmission path corresponding to the first singular value is used although the arrangement of the first signal processing unit is strictly different) It is considered to correspond to the first embodiment). Further, as shown in the eleventh embodiment, the extension to the multiuser MIMO transmission shown in the tenth embodiment is also possible, and in this case, the eleventh embodiment, the tenth embodiment and the tenth embodiment The operation corresponds to the combination of the fifth embodiment and the first embodiment. As described above, also with regard to the train moving cell shown in the fourth embodiment, a plurality of first signal processing units are regarded as RRH using the eleventh embodiment, and the first on the BBU side through an optical fiber If the configuration is such that information is exchanged between the two signal processing units, it is possible to use an operation corresponding to the combination of the eleventh embodiment, the fourth embodiment, and the first embodiment. Naturally, if this is combined with multi-user MIMO including terminal units of multiple trains, the tenth embodiment will be combined at the same time.

  As indicated above, the various embodiments of the present invention can be used in combination with one another.

[Other supplementary items of the embodiment according to the present invention]
In the following, some supplementary points regarding the embodiments according to the present invention will be explained.

In the description of the embodiments of the present invention, the receiving side of the antenna element 851-1~851- (N _'BS-Ant) and the transmission side of the antenna element 819-1~819- (N' _BS-Ant) is Although different numbers have been assigned to each of the above, the antenna elements (851-1 to 851- (N'BS _-Ant ) common to the transmitting system and the receiving system via TDD-SW or the like are naturally described. It is also possible to treat it as 819-1 to 819- (N'BS _-Ant )). In particular, in the case of performing implicit feedback, it is essential to share antennas of the transmitting system and the receiving system, and TDD control in which uplink and downlink are switched in time is fundamental. The switching of the TDD-SW here is managed and controlled by the communication control circuit 120 and the like.

Also, the reception antenna element 851-1 to 851- (N'BS _-Ant ) for the base station apparatus and the transmission antenna element 812-1 to 819- (N'BS _-Ant ) and in the side of the antenna element 851-1~851- _{(N MT-Ant)} and the transmission side of the antenna element 819-1~819- _{(N MT-Ant),} a common number in order to simplify the description, respectively To explain. However, the requirements such as the antenna element size and directivity gain for the base station apparatus and the requirements such as the antenna element size and directivity gain of the terminal station apparatus generally do not match, and the functions are the same. Although the same number is given as the number, in actual operation, antenna elements having different conditions in terms of size, directivity gain, etc. may be used. The same holds true for the other circuits, and the description is made using common numbers for the sake of simplicity. For example, high power amplifiers 818-1 to 818-(N ' _BS-Ant and N _{MT- Ant),} a low noise amplifier 852-1~852- (N _'BS-Ant and _{N MT-Ant)} even in such similarly amplification factor of the amplifier, power consumption, requirement requirements such as the size and permissible heating value is generally Although they do not match and are identical as a function, the same number is given as a number, but in actual operation, circuits of different conditions may be used.

  Further, the transmission weight vector in the embodiment of the present invention means each column vector (or a part of column vectors) of the transmission weight matrix, and is a vector-described transmission weight focused on one of the signal sequences to be spatially multiplexed simultaneously. is there. Specifically, each column vector of the transmission weight matrix at the time of spatially multiplexing a plurality of signal sequences is a vector representation of weights (components of the transmission weight vector) focusing on a certain signal sequence among the plurality of signal sequences. In the process of generation of the entire transmission weight matrix based on the channel vector or channel matrix of the terminal station apparatus to be spatially multiplexed. Therefore, the generation of the transmission weight vector (and also when words such as "calculation", "determination", "multiplication", and "component" follow) is equivalent to the generation of the transmission weight matrix as a whole, in particular When considering a procedure for sequentially generating row vectors or column vectors of a matrix, it may be marked as "generation of transmission weight vector" in a sense equivalent to "generation of transmission weight matrix".

  Similarly, the reception weight vector in the embodiment of the present invention means each row vector (or a part of the row vectors) of the reception weight matrix, and vector notation focused on one of the spatially multiplexed signal sequences simultaneously It is a reception weight. Specifically, each row vector of the reception weight matrix when a plurality of signal sequences are spatially multiplexed is a vector representation of weights (components of the reception weight vector) focusing on a certain signal sequence among the plurality of signal sequences. In the process of generation of the entire reception weight matrix based on the channel vector or channel matrix of the spatially multiplexed terminal station apparatus. Therefore, the generation of the reception weight vector (and also when words such as "calculation", "determination", "multiplication", and "component" follow) is equivalent to the generation of the reception weight matrix as a whole, in particular When considering a procedure for sequentially generating row vectors or column vectors of a matrix, it may be marked as "generation of reception weight vector" in a meaning equivalent to "generation of reception weight matrix".

  Furthermore, “vectors” in various forms such as “transmission weight vector”, “reception weight vector”, “channel (information) vector”, “transmission signal vector”, and “reception signal vector” in the embodiment of the present invention These are all vectors that use "transmission weight", "reception weight", "channel (information)", "transmission signal" and "reception signal" as components corresponding to each antenna element. Even if there is no explicit mention of “vector” in each embodiment, it is obvious that those “components” that have them as components in that embodiment are those “vectors”, if necessary. Should be understood by supplementing these contents. Furthermore, "transmission signal" and "reception signal" in "reception signal vector" and "reception signal vector" are transmission or reception signals corresponding to each antenna element or based on each antenna element, and are actually transmitted and received. It may be a digitized baseband (or intermediate frequency) signal instead of a radio frequency analog. This signal includes both a signal on the frequency axis and a sampling signal on the time axis. Therefore, although there are parts where explicit description is omitted in the description of the above embodiment, these signals are not only the signals themselves transmitted and received at the radio frequency, but also those signals and the signals transmitted and received at the radio frequency. Between the signals, some signal processing may be performed.

  Further, in the description of channel estimation in the embodiment of the present invention, in addition to the case of acquiring channel information in one channel estimation, the case of averaging a plurality of channel estimation results is also described. For example, considering the line gain by utilizing a large-scale antenna element, there is no problem if the line gain is insufficient in the line design between one antenna and one antenna, but the directivity by the large-scale antenna In order to gain the sex gain, appropriate transmit and receive weights are required, and the channel information required to calculate the transmit and receive weights is basically based on the channel estimation result between one antenna and one antenna. If the line gain is insufficient, the directivity gain can not be obtained thereafter. As a countermeasure for this, in Non-Patent Document 2 and the like, a training signal is received by a plurality of symbols, and the shortage of channel gain is compensated by averaging the received signals. As averaging of the plurality of estimation results, in the case where a periodic training signal continues over several symbols, a method of performing short-term averaging over several symbols utilizing the periodicity and discrete There is a method of performing long-time averaging in which channel estimation results performed at time are gathered for a plurality of times and averaging is performed. Here, in the case of short-time averaging, since a plurality of symbols to be averaged are continuous, it is considered that all symbol timings are preserved because of their periodicity. It was not necessary to treat it as a relative channel based on the complex phase. On the other hand, with regard to channel estimation results performed at discrete times, it is generally not necessary that the symbol timing be the same. Therefore, relative channel information is used to eliminate the influence associated with the error of the symbol timing. It was necessary to take a configuration to acquire and average. In this case, the complex phases of the channel information of the reference antenna element are all considered to be zero (ie, take real values). In calculation of these relative channel information, channel information of all antenna elements is multiplied by Exp (-jθ) by complex phase θ of the reference antenna, and channel information of all antenna elements is used as a reference It may be obtained by dividing by the channel information of the antenna element. Such utilization of relative channel information is generally essential for averaging channel estimation results with different symbol timings, but in the case where symbol timings are common, relative channel is generally used in calculating transmission / reception weights. There is no need to use information. It suffices to simply calculate the transmission / reception weight using the equation (1) or the equation (7) for the acquired channel information. However, even if the symbol timing is common, there is no problem even if the relative channel information is used, so sometimes the above description will be described as relative channel information or simple channel information. There is a case where it is used as it is. However, the difference is as described above, and it is not necessary to use relative channel information.

  Also, relative channel information can be treated as channel information obtained by correcting the complex phase with reference to the complex phase of the reference antenna, or the channel information of the antenna element as channel information of the antenna including the amplitude. Information may be divided. The amplitude of each piece of relative channel information has a meaning, for example, in the case of the reception weight of maximum ratio combination represented by equation (7), but in the case of the transmission / reception weight of equal gain combination represented by equation (1) do not have. Furthermore, in practice, in the line-of-sight environment, the deviation of the amplitude of each antenna is expected to be very limited, in which case there is no significant difference even if all are approximated with the same amplitude. In the first place, the absolute value of the transmit and receive weights has no significant meaning, and needs to be separately optimized to be an efficient value among the limited number of quantization bits, but it is related to the number of such quantization bits The discussion is completely different from the embodiment of the present invention, and optimization may be made in the existing technology. In that sense, the size (absolute value) of the transmit / receive weight vector as a vector does not have a special meaning here, and the transmit / receive weight vector multiplied by an arbitrary coefficient also has its statistical properties preserved here. It is explaining.

  By using the above relative channel information, various signal processing can be simplified in the embodiment of the present invention. On the other hand, in the conventional MIMO transmission technology, for example, the process of multiplying the reception weight matrix directly has been described as including the process of converting into a state suitable for signal detection. That is, even for SISO signals, for signal detection processing, the received signal is divided by the channel estimation result, and the signal point on the constellation where the I and Q axes are correctly set from the received signal subjected to channel distortion. Needed to be converted. In the embodiment of the present invention, the multiplication processing of transmission / reception weights performed in the first signal processing unit mainly aims at securing directivity gain and signal processing for rough signal separation. The received signal is divided by the channel estimation result, as it is possible to treat many antenna elements as one virtual antenna element by multiplication of weights, and the I / Q axis is obtained from the received signal subjected to channel distortion. Does not include the process of converting into signal points on the constellation set correctly. However, on the reception side, processing such as signal detection can also be performed in the subsequent stage, and the same signal processing as the signal processing performed in conventional MIMO transmission or SISO transmission can be applied to these processing. is there. Particularly, in the reception system of the base station apparatus, these processes are performed in the second signal processing circuit.

  Also, when a training signal composed of limited subcarrier components is transmitted from different antenna elements on the transmitting station side without overlapping of subcarriers allocated by each antenna element, there is no overlap on the frequency axis The nuclear training signal is combined in space and received at the receiving station as a training signal containing more subcarrier components. In this sense, the receiving station side is different from the training signal transmitted by each antenna element on the transmitting side, but the training signal synthesized in space (referred to as “combined training signal”) is regarded as a normal training signal. It becomes possible to perform signal processing. Also in the case of using such a synthetic training signal, it is effective to convert channel information into relative channel information which is a relative value with respect to the antenna element serving as a reference.

  In the above description, for simplicity, k (for example, the k-th frequency component etc.) representing a subcarrier is omitted, and further, there is a part where an explanation regarding an individual subcarrier is also omitted. The system assumed is a wide-band system, and all signal processing in channel information, transmission / reception weights, and further transmission and reception signals, etc. is basically the same except for signal processing on the time axis in the fifth embodiment, etc. In practice, each subcarrier should be defined and processed individually on the frequency axis. Therefore, in order to simplify the description, in many descriptions, subscripts that explicitly represent subcarriers have been omitted. However, these descriptions are actually performed individually for each subcarrier, and in that case, the descriptions can be interpreted exactly if they are understood by adding subscripts representing the subcarriers. Inside each signal processing circuit, for example, signal processing up to the previous stage of IFFT processing on the transmission side (in the case of OFDM modulation as an example, interleaving of bit strings, mapping of signal points, modulation of signals, transmission weight vector The multiplication and the like are all performed for each subcarrier, and the signal processing after FFT processing on the reception side (also assuming the OFDM modulation method, multiplication of reception weight, signal detection processing, signal de- Mapping, deinterleaving, etc.) are all performed for each subcarrier.

  Also, in terms of circuit configuration, separate circuits may be provided for each subcarrier, or since the same processing is performed, processing is performed serially for each subcarrier, and the circuit is shared for subcarriers. It is also possible to Furthermore, a plurality of circuits may be prepared intermediately, the subcarrier may be divided appropriately, and the parallel processing may be performed serially by the plurality of circuits. These are stories common to all the embodiments.

  Further, in the OFDM modulation scheme, all subcarriers are used for communication with the same terminal station apparatus, so the transmission / reception weights (averaged transmission / reception weight vector and real-time transmission / reception weight matrix) at that time are common to all subcarriers. The transmission and reception weights for the terminal station apparatus of the combination will be used. However, in OFDMA, assignments are made to different combinations of terminal stations in a patchwork manner on the time axis and the frequency axis, and therefore, they are assigned for each time (OFDM symbol) and frequency (subcarrier). It is necessary to use transmission / reception weights for the terminal station apparatus. However, except for the difference, OFDM and OFDMA can be processed in exactly the same way, and in the present specification, the explanation has been made focusing on OFDM, but the embodiment of the present invention is applied to OFDMA in exactly the same way. can do.

  In addition, although there are various operational variations regarding SC-FDE, received signal processing after signals transmitted from each antenna element are combined in space after multiplication by a transmission weight vector on the transmission side, and In each of the above configuration examples, the processing performed in the conventional SC-FDE can be applied as it is to any of the reception signal processing after the reception weight is multiplied on the reception side and the signals of the respective antenna elements are added and synthesized. Because of the configuration, it is applicable to all variations of SC-FDE. In this case, signal processing with a single carrier is performed instead of signal processing with the OFDM modulation scheme, and then, in the case of downlink, FFT processing is performed on the signal on the time axis of the single carrier, thereby each subcarrier. The following signal components may be generated, and these signal components may be regarded as signals of respective subcarriers generated by the OFDM modulation method and multiplied by the transmission weight vector generated according to the embodiment of the present invention. Similarly, in the case of the uplink, the signal obtained by subjecting the received signal to FFT processing is treated in the same manner as in the case of the OFDM modulation scheme, and signal separation is performed by multiplying the reception weight vector generated according to the present invention. The subcarrier signal may be converted to a single carrier signal on the time axis by performing IFFT processing. As described above, although there is a difference between part of signal processing between the OFDM modulation method and SC-FDE, generation of transmission / reception weights and multiplication processing are common, and the embodiment of the present invention It is applicable.

  Also, in spatial multiplexing transmission, signals of a plurality of sequences are transmitted in parallel, but in the above embodiment, processing such as error correction performed on these signal sequences is performed independently for each signal sequence as an example. As described above, it goes without saying that the signal after error correction coding is serial / parallel converted at the transmitting side to separate into a signal sequence to be spatially multiplexed, and the signal before the error correction processing is performed on the receiving side. Parallel / serial conversion may be performed, and then error correction processing such as Viterbi decoding of one system may be performed. Furthermore, any error correction process may be performed regardless of the essence of the embodiment of the present invention, including other variations. In this case, the exchange of signals between the MAC layer processing circuit and the second signal processing unit etc. is not performed in the space multiplex system, but, for example, information exchange is performed as a single system signal, and serial / A function corresponding to parallel conversion, parallel / serial conversion, and error correction is included in the second signal processing unit or the like.

  In the above description, although the right singular vector of singular value decomposition is used, the left singular vector obtained by singular value decomposition of the matrix obtained by transposing the matrix subjected to singular value decomposition is the singular value decomposition of the matrix not transposed It is equivalent to the right singular vector. Similarly, even if the original channel vector is multiplied by the Hermite conjugate vector to perform eigenvalue decomposition, a vector equivalent to the right singular vector or the left singular vector can be obtained. In this sense, the use of a mathematically equivalent vector to the “right singular vector” also falls within the intended “right singular vector” of the embodiment of the present invention. Similarly, utilizing a mathematically equivalent vector to the left singular vector also falls within the intended “left singular vector” range of the embodiment of the present invention.

  Furthermore, in the above description, only the virtual transmission path corresponding to the first singular value is used between each first signal processing unit on the base station side and the terminal station apparatus, and the singular value after the second singular value is used. It has been described that the virtual transmission line corresponding to is not used, but, for example, in the case of using a shared antenna relating to a plurality of polarizations such as V polarization and H polarization, each first signal processing unit In order to utilize the virtual transmission line corresponding to the first singular value in the polarization antenna, formally one virtual signal transmission line corresponding to two singular values is utilized in one first signal processing unit. It is equivalent to In particular, when there is a slight leak between polarizations, the first signal processing unit collectively accommodates and performs signal processing on all two types of polarization antennas to suppress crosstalk components between polarizations. In other words, using virtual transmission paths corresponding to the first singular value and the second singular value is superior in transmission efficiency. In this case, the mathematical expression corresponds to utilizing the first singular value and the second singular value, but in the practical sense, virtual transmission paths corresponding to the first singular value are provided for each polarization antenna group. It corresponds to using. The intention of the present invention is that, in the case of utilizing a plurality of antenna groups which are expected to be very low in correlation with each other as in the case of utilizing such polarized antennas, signal processing or circuit Even in the case where the same first signal processing unit is used, the case where the virtual transmission paths corresponding to the first singular value effectively corresponding to the respective antenna groups are used in parallel, Spatial multiplexing transmission utilizing a virtual transmission path corresponding to a first singular value in an effective sense is to be transmitted using a plurality of first signal processing units. In addition, in the plurality of first signal processing units included in the base station apparatus, upconversion and downconversion processing between the baseband and the radio frequency are basically performed using individual local oscillator signals. However, it is preferable that the signals of the respective local oscillators behave as quasi-synchronously as possible, and for this purpose, a reference signal such as a common reference clock may be supplied from the third signal processing unit. Alternatively, a radio signal for performing up conversion and down conversion between the baseband and the radio frequency by supplying a reference signal of an intermediate frequency from the second signal processing unit to the first signal processing unit and performing, for example, multiplication processing thereof. It is also possible to generate a signal of frequency. Depending on the value of the intermediate frequency used here and the stability of the circuit that performs multiplication processing, the uncertainty of the complex phase of the wireless signal among the plurality of first signal processing units is limited also in the transmission signal processing. In this case, it also becomes possible to perform precoding processing among the plurality of first signal processing units in the second signal processing unit. However, basically, the reference signals supplied from at least the second signal processing unit to the plurality of first signal processing units are on the cable between the plurality of first signal processing units in the second signal processing unit. In order to reduce the loss of the signal, it is fundamental to set the frequency to 1/2 or less of the frequency of the local signal used for upconversion and downconversion processing between baseband and radio frequency, and in this sense the local oscillator It becomes independent between the signal processing units. However, when the base station apparatus and the terminal station apparatus are arranged in a short distance, L in equation (29) becomes smaller, and as a result, the distance between the first signal processing units can be shortened. Depending on the value of the cable loss, the second signal processing unit may directly supply the signal of the local oscillator to the plurality of first signal processing units.

  Further, in realizing the third embodiment, the general control base station not only manages the communication between the satellite base station apparatus and the general control base station, but also manages the communication between the satellite base station apparatus and the terminal station apparatus as well. Will do. As described above, when the satellite base station apparatus capable of performing low correlation communication with the focused terminal station apparatus is selected from the plurality of satellite base station apparatuses to perform communication, etc. The satellite base station apparatus to be used is selected as scheduling processing, and the selection result is instructed to the satellite base station apparatus to perform communication. Furthermore, as shown in FIG. 33, especially when communication is performed across a plurality of small cells, etc., the distance to the satellite base station to be used varies, and there is a gain difference in line design, etc. Management of transmission power control may be required. These management functions are implemented on the side of the general base station apparatus to stably realize various wireless communication controls.

  Further, in the description of the tenth embodiment, the transmission weight is notified from the transmission weight processing circuits 432, 464, and 470 in the BBU to the RRH side via the optical fibers 455, 455a, and 455b. The reception weight is notified from the reception weight processing circuit 467, 480, 482 to the RRH side via the optical fibers 475a, 475b, 485a, 485b, and the correlation calculation circuit 481a, 481b, 483a, 483b from the optical fiber 475a, 475b In this configuration, the reception weight information or channel information on the reception side is notified to the BBU side via 485a and 485b. Here, the information exchanged between the BBU and the RRH does not necessarily indicate the transmission / reception weight information itself digitally. It does not have to be For example, if the beam patterns used by the RRH at the time of transmission and reception are menued, and the coefficients for forming a number of transmission and reception weights corresponding to each beam pattern are listed in the RRH, the beam patterns It is also possible to manage the number as an identifier of H as a codebook, and exchange the codebook value between BBU and RRH. In this case, the correlation value calculation circuits 481a, 481b, 483a, and 483b on the RRH side compare the calculated correlation value for each antenna element with the coefficient on the codebook, and the codebook value with the smallest error from the coefficient It is also possible to configure the function of selecting and notifying of the selected codebook value.

  Furthermore, particularly when the present invention is used in an access system, etc., the antenna element of the terminal station apparatus and the antenna element of the base station apparatus are in line-of-sight environment when the antenna installation position of the base station apparatus is installed low. It may be required to be able to perform stable communication even under conditions that are not In this case, the superiority of the first singular value to the second singular value and thereafter diminishes, and as a result, the entire antenna element is divided into a plurality of groups and signal processing results in sufficient gain due to the division loss. It is conceivable that it can not be secured. In such a case, for example, it is better in characteristics to combine all antenna elements in one place conventionally and perform MIMO transmission normally than when divided into a plurality of first signal processing units, for example. Is expected. When designing a system including such a case, as an example, a part (for example, one) of a plurality of first signal processing units is mounted with more antenna elements than the other first signal processing units. When the special first signal processing unit performs MIMO transmission utilizing the second singular value and subsequent ones according to the situation, and the first signal processing unit respectively corresponds to the first singular value It is also possible to operate with mode switching in the case of performing MIMO transmission using a target transmission path. For example, when it is determined that the number of antenna elements of the special first signal processing unit is 256 elements, the number of antenna elements of the other first signal processing units is 64, and it is determined that the line of sight with the terminal device can be generally secured. Performs four-multiplexing using four virtual transmission paths corresponding to the first singular value, and performs ordinary MIMO communication using the 256-element antenna element of the special first signal processing unit when the line of sight can not be secured You may At this time, the special first signal processing unit is not necessarily provided in the vicinity of the plurality of other first signal processing units. For example, the special first signal processing unit is low at the rest, A plurality of first signal processing units may be provided at high places, for example, to provide physically clear differences. In this case, the special first signal processing unit installed at a low place utilizes the virtual transmission path corresponding to the first singular value if the possibility of securing the line of sight with the terminal station apparatus is low. The first signal processing unit may be provided only at a high place. If mode switching is appropriately performed on these, communication can always be performed under optimum conditions in accordance with the environment. The device corresponding to the special first signal processing unit does not necessarily have to be accommodated in the same second signal processing unit. For example, apart from the base station apparatus performing the MIMO transmission utilizing the virtual transmission path corresponding to the first singular value shown in the embodiment of the present invention, the conventional base station apparatus also utilizing the second singular value and subsequent ones is also nearby. It is also possible for the terminal station apparatus to select an optimal base station apparatus to perform communication according to the situation while the two are coordinated.

  Further, in the second embodiment, the provision on the case where the first signal processing unit on the base station apparatus side or the directional antenna element is disposed on a straight line is defined, but in the other embodiments, the first signal is provided. The arrangement of the processing units is not necessarily arranged on a straight line, and may be an arrangement having a two-dimensional spread. In particular, when the present invention is used in an access system, etc., the correlation between each first signal processing unit is reduced by arranging the arrangement of the first signal processing unit on the wall of a building two-dimensionally. Is also possible.

  In each of the embodiments described above, the configuration assuming implicit feedback of Massive MIMO has been described. However, strictly speaking ignoring MAC efficiency, there is no problem with using explicit feedback with control information.

  Also, in the above description, the multiplication processing of the transmission / reception signal and the transmission / reception weight has been described as being performed on the baseband, but it shows a typical signal processing, and a signal processing equivalent thereto is a baseband signal It is of course also possible to implement on an intermediate frequency (IF) (Intermediate Frequency) signal between RF and RF signals. Although it is easier to perform processing in a lower frequency band in terms of signal processing, in the 5th generation mobile communication, since communication is performed in a bandwidth of several hundred to 1 GHz in a high frequency band such as a millimeter wave band, Even if the frequency is processed in the IF band of several GHz, the processing difficulty does not greatly change. The baseband signal or the baseband in the present specification is used in a broad sense of a frequency band capable of performing digital signal processing, and in this sense, an IF different from the baseband in a narrow sense. Even in the case of band-based signal processing, it is possible to apply the present invention essentially, and the claims of the present invention include such broad baseband signals and basebands.

  Further, according to the present invention, since the number of antenna elements included in the radio station apparatus is enormous, even if a few of the antenna elements among them are subjected to exceptional signal processing, the remaining majority of It is possible to obtain generally expected characteristics by the effect. However, even if such a part of antenna elements is processed exceptionally, the result of signal processing using the remaining antenna elements will determine most of the characteristics in most cases. Even if the application of the exception processing is performed, if the effect is limited without obtaining the expansion effect by the exception application, it should be the scope of the present invention.

  The base station apparatus and the terminal station apparatus in the embodiments described above may be realized by a computer. In that case, a program for realizing this function may be recorded in a computer readable recording medium, and the program recorded in the recording medium may be read and executed by a computer system. Here, the “computer system” includes an OS and hardware such as peripheral devices. The term "computer-readable recording medium" refers to a storage medium such as a flexible disk, a magneto-optical disk, a ROM, a portable medium such as a ROM or a CD-ROM, or a hard disk built in a computer system. Furthermore, the "computer-readable recording medium" dynamically holds a program for a short time, like a communication line when transmitting a program via a network such as the Internet or a communication line such as a telephone line. It may also include one that holds a program for a certain period of time, such as volatile memory in a computer system that becomes a server or a client in that case. Further, the program may be for realizing a part of the functions described above, and may further be capable of realizing the functions described above in combination with a program already recorded in a computer system. It may be realized using hardware such as PLD (Programmable Logic Device) or FPGA (Field Programmable Gate Array).

  Although the embodiments of the present invention have been described above with reference to the drawings, it is apparent that the above embodiments are merely examples of the present invention, and the present invention is not limited to the above embodiments. is there. Therefore, additions, omissions, substitutions, and other modifications of the components may be made without departing from the technical spirit and scope of the present invention.

1 ... Base station apparatus,
2 ... Radio station apparatus,
3 ... prospect wave,
4 ... Stable reflection wave by structure,
5 ... Multiple reflections near the ground,
6 ... Multiple reflections near the ground,
7 ... structure,
11 ... transmitting station,
12 ... receiving station,
21 ... High Power Amplifier (HPA),
22 ... Low noise amplifier (LNA),
23 ... time division switch (TDD-SW),
24 ... antenna element 25 ... wireless module,
26 ... antenna terminal,
27 Coaxial cable,
31 · · · General control base station equipment,
32: Satellite base station apparatus,
33: Terminal station apparatus,
34 · · · Base station equipment,
35: Satellite base station apparatus,
36 ... terminal station apparatus,
37 ... structure,
38 ... structure,
39 ... cell,
40 ... wireless communication system,
41: Base station apparatus,
50: Wireless communication system,
51: Terminal station apparatus,
52: Wireless communication system,
53 ... wireless communication system,
60 ... terminal station apparatus,
61: Transmission unit,
65: Reception unit,
67 ... interface circuit,
68 MAC layer processing circuit
70: Base station apparatus,
71 ... second transmission signal processing unit,
75: second received signal processing unit,
77: Interface circuit,
78 MAC layer processing circuit
80: Base station apparatus,
81: Transmission unit,
85: Reception unit,
87: interface circuit,
88 MAC layer processing circuit
90 ... wireless communication system,
111 ... first transmission signal processing circuit,
113 ... Transmission signal processing circuit,
114: Reception signal processing circuit,
120 ... communication control circuit,
121 ... communication control circuit,
130 ... first transmission weight processing unit,
131: First channel information acquisition circuit,
132: first channel information storage circuit,
133: first transmission weight calculation circuit,
140: first transmission weight processing unit,
141: Channel information acquisition circuit,
142: Channel information storage circuit,
143: first transmission weight calculation circuit,
144: first reception weight processing unit,
145: Reception signal processing circuit,
146: first channel information estimation circuit,
147: first reception weight calculation circuit,
148: second transmission signal processing circuit,
149: Transmission weight calculation circuit,
150: Transmission weight processing unit,
151: Channel information acquisition circuit,
152: Channel information storage circuit,
153: Multi-user MIMO transmission weight calculation circuit,
154: first reception weight processing unit,
155: first received signal processing circuit,
156: first channel information estimation circuit,
157: first reception weight calculation circuit,
158: first reception signal processing circuit,
159 ... second received signal processing circuit,
160: first reception weight processing unit,
161: first channel information estimation circuit,
162: first reception weight calculation circuit,
170: Reception weight processing unit
172: Channel information estimation circuit,
173: Multi-user MIMO reception weight calculation circuit,
181 ... first transmission signal processing unit,
182: first transmission signal processor,
185: first received signal processing unit,
186: first received signal processing unit,
190 ... second reception signal processing circuit,
191: Channel matrix acquisition circuit,
192 ... reception weight matrix calculation circuit,
193 ... reception weight matrix multiplication circuit,
194: Signal detection circuit,
201 ... virtual antenna element,
202 ... virtual antenna element,
203: Virtual antenna element,
204: Base station apparatus,
205: Antenna element group,
206: Terminal station apparatus,
221-1 to 5 ... antenna elements,
231, 232, 233, 234 ... base station apparatus,
235, 236, 237, 238, 239 ... terminal station apparatus,
241, 242, 243, 244 ... subcarrier,
251-1 to 3, 253-1 to 3, 254-1 to 3, 255-1 to 3, 256-1 to 3, ... slots,
252-1 to 3 ... slots for data communication,
257-1 to 3 ... training signal,
258-1 to 3 ... data payload,
291 to 295: plot points,
301: Base station apparatus,
302: Terminal station apparatus,
303 ... base station apparatus,
304: first signal processing unit,
305: second signal processing unit,
306 ... base station apparatus,
307 ... dish antenna,
308: Terminal station apparatus,
309 ... terminal station apparatus,
310 ... base station apparatus antenna,
311 ... first transmission signal processing circuit,
312 ... terminal station apparatus antenna element group,
320 ... terminal station apparatus antenna element group,
330 ... first transmission weight processing unit,
331: first transmission signal processing circuit,
332: First channel information acquisition circuit,
333 ... first channel information storage circuit,
334: first transmission weight calculation circuit,
340 ... first transmission weight processing unit,
341: first channel information acquisition circuit,
342: first channel information storage circuit,
343 ... first transmission weight calculation circuit,
348 ... second transmission signal processing circuit,
354 ... first reception weight processing unit,
355 ... first received signal processing circuit,
356: first channel information estimation circuit,
357 ... first reception weight calculation circuit,
358 ... first reception signal processing circuit,
359 ... second received signal processing circuit,
360 ... first reception weight processing unit,
361 ... transmission unit,
362 ... first channel information estimation circuit,
363 ... first reception weight calculation circuit,
365: Reception unit,
381 ... first transmission signal processing unit,
385 ... first received signal processing unit,
401: MAC layer processing circuit
403, 403a, 403b ... time axis signal generation circuit,
412: Transmission signal processing circuit,
427, 427a, 427b ... D / A converter,
428, 428a, 428b ... RF processing circuit,
429, 429a, 429b ... antenna elements,
432: Transmission weight processing circuit
441-4, 441-5, 441-6 ... BBU,
442-4, 442-4a, 442-4b, 442-6 ... RRH,
443-1, 443-2, 443-3 ... BBU,
444-1a, 444-1b, 444-2a, 444-2b, 444-3 ... RRH,
446-1, 446-2 ... area,
454, 454a, 454b ... optical interface circuit,
455, 455a, 455b ... optical fiber,
456, 456a, 456b ... optical interface circuit 457, 457a, 457b ... time axis transmission weight multiplication circuit,
461 ... MAC layer processing circuit,
462 ... Transmission signal processing circuit,
463 ... Transmission signal processing circuit,
464 ... Transmission weight processing circuit,
465 ... addition synthesis circuit,
466 ... received signal processing circuit,
467 ... Reception wait processing circuit,
470: Transmission weight processing circuit
471 ... MAC layer processing circuit,
472: Reception signal processing circuit,
473a, 473b ... time axis reception weight multiplication circuit,
474a, 474b ... optical interface circuit,
475a, 475b ... optical fiber,
476a, 476b ... optical interface circuit,
477a, 477b ... A / D converter,
478a, 478b ... RF processing circuit,
479a, 479b ... antenna elements,
480: Reception wait processing circuit,
481, 481a, 481b ... correlation calculation circuit,
482 ... Reception weight processing circuit,
483a, 483b ... time axis reception weight multiplication circuit,
484a, 484b ... optical interface circuit,
485a, 485b ... optical fiber,
486a, 486b ... optical interface circuit,
491-2, 491-3, 491-N _Ant ... multiplier,
492-2, 492-3, 492-N _Ant ... Adder,
493-2, 493-3, 493-N _Ant ... Memory,
494 ... correlation operation control unit,
495-1, 495-2, 495-3, 495-N _Ant ... multiplier,
496-1, 496-2, 496-3, 496-N _Ant ... Adder,
497-1, 497-2, 497-3, 497-N _Ant ... Memory,
498-1, 498-2, 498-3, 498-N _Ant ... square root acquisition circuit,
600 ... base station apparatus,
601, 601-1, 601-2, 601-3, 601-4, 601-5, 601-6, 601-7, 601-8 ... the first signal processing unit,
602-1, 602-2, 602-3, 602-4, 602-5, 602-6, 602-7, 602-8 ... geometric coordinate information,
603 ... train,
604 ... terminal station apparatus,
605 ... camera,
606 ... weight matrix / coordinates database,
607 ... weight vector / coordinate database,
612 ... coordinate information acquisition circuit,
613 ... time information acquisition circuit,
614 ... communication control circuit,
622 ... coordinate information acquisition circuit,
623 ... Time information acquisition circuit,
624 ... communication control circuit,
700 ... antenna element,
701 ... TDD-SW,
702 ... Low noise amplifier (LNA),
703 ... mixer,
704 ... filter,
705 ... A / D converter,
706 ... FFT circuit,
707: Reception weight multiplication circuit,
708 ... Transmission weight multiplication circuit,
709 ... IFFT & GI application circuit,
710 ... D / A converter,
711 ... mixer,
712 ... filter,
713 ... high power amplifier,
714 ... TDD-SW,
721 ... antenna element,
722 ... low noise amplifier (LNA),
723 ... mixer,
724 ... filter,
725 ... A / D converter,
726 ... FFT circuit,
727: Reception weight multiplication circuit
728 ... Transmission weight multiplication circuit,
729 ... IFFT & GI application circuit,
730 ... D / A converter,
731 ... mixer,
732 ... filter,
733 ... High Power Amplifier,
740 ... Transmission weight processing unit,
741 ... local oscillator 742 ... local oscillator 743 ... communication control circuit,
781 ... scheduling processing circuit,
811 ... Transmission signal processing circuit,
812 ... addition synthesis circuit,
813 ... IFFT & GI addition circuit,
814 ... D / A converter,
815 ... Local oscillator,
816 ... mixer,
817 ... filter,
818 ... High Power Amplifier,
819 ... antenna element,
820 ... communication control circuit,
821 ... first transmission signal processing circuit,
830 ... Transmission weight processing unit,
831 ... channel information acquisition circuit,
832: Channel information storage circuit,
833 ... multi-user MIMO transmission weight calculation circuit,
851 ... antenna element,
852 ... Low noise amplifier (LNA),
853 ... Local Oscillator,
854 ... mixer,
855 ... filter,
856 ... A / D converter,
857 ... FFT circuit,
858 ... received signal processing circuit,
860: Reception weight processing unit
861 ... channel information estimation circuit,
862 ... multi-user MIMO reception weight calculation circuit,
881 ... scheduling processing circuit

Claims (5)

A wireless communication system comprising a first wireless station apparatus and a second wireless station apparatus, the wireless communication system comprising:
The first radio station apparatus is
A plurality of first signal processing units having a first antenna element group;
A second signal processing unit that executes signal processing of wireless communication associated with the plurality of first signal processing units;
Have
The second radio station apparatus is
A second antenna element group,
A transmitting / receiving unit that performs wireless communication with the first wireless station apparatus via the second antenna element group;
Have
The first signal processing unit
Each of the first signal processing units is provided with a different local oscillator,
The transmission weight vector for MIMO (Multiple Input Multiple Output) channel matrix used in the wireless communication between the second antenna element group and the plurality of first antenna element group, the MIMO channel matrix is the local oscillator common Calculated based on the first right singular vector corresponding to the first singular value of the MIMO partial channel matrix obtained by dividing each individual first antenna element group to be converted or the approximate solution of the first right singular vector And
A wireless communication system, wherein the plurality of first signal processing units spatially multiplex and transmit independent signal sequences using the transmission weight vectors corresponding to the first signal processing units.   The first signal processing unit is configured to input a local signal to a mixer for performing up conversion and down conversion between a radio frequency signal and a baseband signal or an intermediate frequency signal as the same first signal. The wireless communication system according to claim 1, wherein the common local signal is input to a mixer included in the processing unit, and the different local signal is input to a mixer included in the different first signal processing unit.   When one single signal processing unit uses a single polarization antenna at the same time and the same frequency, one system of signal sequences is used, or one system or a plurality of types of polarization antennas are used simultaneously. The wireless communication system according to claim 1, wherein two signal sequences are transmitted or received for each terminal station apparatus. The first signal processing unit suppresses interference components by performing processing for improving directivity gain on a desired signal from a spatially multiplexed signal sequence;
The second signal processing unit performs signal processing for extracting the signal sequence, which is a desired signal, from a signal in which a residual interference component is not removed by the first signal processing unit. Wireless communication system. A wireless communication method in a wireless communication system comprising a first wireless station apparatus and a second wireless station apparatus,
The first radio station apparatus is
A plurality of first signal processing units having a first antenna element group;
A second signal processing unit that executes signal processing of wireless communication associated with the plurality of first signal processing units;
Have
The second radio station apparatus is
A second antenna element group,
A transmitting / receiving unit that performs wireless communication with the first wireless station apparatus via the second antenna element group;
If you have
The first signal processing unit
Each of the first signal processing units is provided with a different local oscillator,
The transmission weight vector for MIMO (Multiple Input Multiple Output) channel matrix used in the wireless communication between the second antenna element group and the plurality of first antenna element group, the MIMO channel matrix is the local oscillator common Calculated based on the first right singular vector corresponding to the first singular value of the MIMO partial channel matrix obtained by dividing each individual first antenna element group to be converted or the approximate solution of the first right singular vector Step to
A plurality of first signal processing units performing spatial multiplexing transmission of independent signal sequences using the transmission weight vectors corresponding to the first signal processing units;
Wireless communication method including: