JP2760254B2 - Multi-channel FM receiving method and apparatus - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a multi-channel used as a front end of a BS or CS tuner expected to have a wide band in the future, or a receiving front end used for CATV services in a local area. The present invention relates to an FM reception method and apparatus.

[0002]

2. Description of the Related Art Conventionally, this type of FM receiving method and apparatus use a down-conversion method. FIG.
1 shows a configuration block diagram of a conventional multi-channel FM receiver. In FIG. 9, reference numeral 1 denotes a high-pass filter for selecting a multi-channel FM signal from a received signal, and reference numeral 2 denotes a variable gain control unit for adjusting the FM signal obtained from the high-pass filter 1 to a predetermined level. Reference numeral 3 denotes a variable band-pass filter that selects an FM signal of a desired channel in accordance with a channel selection command from a controller (not shown). 4 is a broadband amplifier for amplifying the selected FM signal. Reference numeral 5 denotes a down-convert mixer that down-converts the FM signal obtained from the amplifier 4 in response to the local oscillation signal and generates an intermediate frequency signal. Reference numeral 6 denotes a SAW (Sur) for removing unnecessary components of the intermediate frequency signal obtained from the down-convert mixer 5.
face Acoustic Wave) filter.

[0003] Reference numeral 7 denotes a VCO in which the oscillation frequency of the output signal changes according to the DC control signal. Reference numeral 8 denotes a distributor which receives an output signal from the VCO 7 and distributes the signal to the mixer 5 to a local oscillation signal and a feedback signal for generating a DC control signal. Reference numeral 9 denotes a prescaler that divides the feedback signal by a factor of 128. Numeral 10 further divides the feedback signal divided by the prescaler 9 into 1 / N (N is a positive integer and changes according to the channel selection signal) by a channel selection command of a controller (not shown). It is a programmable prescaler that goes around.

A reference signal generator 11 generates a reference signal. Reference numeral 12 denotes a phase error detection circuit that detects a phase error between the feedback signal obtained from the programmable prescaler 10 and the reference signal and generates a phase error signal. Reference numeral 13 denotes an amplifier that amplifies the phase error signal, and reference numeral 14 denotes a low-pass filter that extracts a DC component of the phase error signal and uses the DC component as a DC control signal to the VCO 7. According to the configuration of FIG.
Reception by the single super reception method is performed.

Next, the operation of the conventional multi-channel FM receiver shown in FIG. 9 will be described. Now, about 9
BS (Broa) located from 50 MHz to 2000 MHz
It is assumed that a multi-channel FM signal conforming to the dcasting satellite (broadcasting satellite) format is received. The input multi-channel FM signal is filtered to remove unnecessary low-frequency components by a high-pass filter 1, adjusted to a predetermined level by a variable gain control unit 2, and selected only a desired channel by a variable frequency band-pass filter 3. , Suppresses image disturbance. After that, the FM signal of the selected channel is amplified by the amplifier 4 and the down-convert mixer 5
After being converted into a 2.78 MHz intermediate frequency signal and passed through the SAW filter 6, it is detected by a detector (not shown).

Further, a local signal for down-conversion is oscillated at a frequency of about 1350 MHz to 2400 MHz, which is controlled by a PLL synthesizer at a comparative frequency of about 10 kHz. That is, a loop circuit from the VCO 7 to the low-pass filter 14 constitutes a PLL circuit 15 controlled by a controller (not shown).

[0007]

However, in the conventional single super receiving method, when the number of channels is increased, the variable frequency range of the variable frequency band-pass filter 3 is widened, which is difficult to realize, and the oscillation of the VCO 8 is difficult. The band also requires a range exceeding one octave, which also has the problem that the complexity of the circuit cannot be avoided. Furthermore, since the local oscillation signal exists in the band of the input signal, when a plurality of tuners are driven in parallel, the leakage of the local oscillation signal is
There was also a problem of interfering with adjacent tuners.

The present invention solves the above-mentioned conventional problem. By tuning in a double super system, it is possible to reduce the influence of leakage between a plurality of tuners with a simple circuit configuration, and to reduce image interference. It is an object of the present invention to provide an excellent multi-channel FM receiving method and apparatus capable of suppressing the noise.

[0009]

According to the present invention, there is provided a multi-channel FM receiving method comprising the steps of: receiving a multi-channel FM input signal having a center frequency at a predetermined frequency interval; Generating a first local oscillation signal having an integer multiple of and having a variable frequency, receiving the first local oscillation signal, up-converting the input signal, and doubling the frequency of an arbitrary frequency of the spectrum of the FM input signal. And generates a first intermediate frequency signal that does not overlap any of the frequency band of the FM input signal and the first local oscillation signal frequency band, and generates the FM signal frequency band, the first local oscillation signal band, and A second intermediate frequency signal is generated by receiving a second local oscillation signal having a constant frequency that does not overlap with any of the first intermediate frequency bands and down-converting the first intermediate frequency signal.

In order to achieve the above object, a multi-channel FM receiver according to the present invention provides a multi-channel
PLL that makes C / N suppression performance by L as good as possible
A control unit, a first local oscillation signal generation unit controlled by the control unit, and a multi-channel FM input signal whose center frequency is formed at a predetermined frequency interval, avoiding a distortion frequency spectrum group generated by mutual interference between the channels. First intermediate frequency generating means for generating a second local oscillation signal having a constant frequency, receiving means for down-converting the first intermediate frequency signal in response to the second local oscillation signal. , A second intermediate frequency signal generating means for generating a second intermediate frequency signal.

[0011]

According to the multi-channel FM receiving method and apparatus of the present invention having the above-described configuration, the frequency band of the multi-channel FM input signal, the first local signal frequency band, and the first intermediate frequency band are arranged such that the center frequencies are arranged at predetermined frequency intervals. And the second local oscillation signal band are all set so that they do not overlap each other, and the reference frequency of the PLL of the first local oscillation signal generating means is raised to the theoretical limit. Can improve the C / N suppression performance as much as possible, and as a result, the evaluation noise of the video baseband signal after detection can be reduced.

[0012]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS A first embodiment of the present invention will be described below with reference to the drawings.

FIG. 1 is a block diagram showing the configuration of a multi-channel FM receiver according to a first embodiment of the present invention. In FIG. 1, reference numeral 21 denotes a high-pass filter that removes unnecessary components in a low frequency band from a received signal and selects a multi-channel FM signal. Reference numeral 22 denotes a variable gain control unit that adjusts the FM signal obtained from the high-pass filter 21 to a predetermined level. Reference numeral 23 denotes a broadband amplifier that amplifies the FM signal obtained from the variable gain control unit 22. Reference numeral 24 denotes a low-pass filter for removing unnecessary high-frequency components of the amplified FM signal. 25 receives the first local oscillation signal and generates a low-pass filter 2
4 is an up-converting mixer (hereinafter, referred to as a “first mixer”) as a first intermediate frequency generating unit that generates an intermediate frequency signal by up-converting the FM signal obtained from the signal No. 4.

Reference numeral 26 denotes a band-pass filter for removing unnecessary components of the intermediate frequency signal obtained from the first mixer 25, which comprises a half-wavelength type filter, a dielectric filter and the like. Reference numeral 27 denotes a narrow band amplifier for amplifying the first intermediate frequency signal obtained from the band pass filter 26. Reference numeral 28 denotes a band-pass filter for removing unnecessary components of the first intermediate frequency signal amplified by the amplifier 27, which is constituted by a half-wavelength type filter, a dielectric filter, and the like. 29 receives the second local oscillation signal and receives the first signal obtained from the band-pass filter 28.
A down-converter (hereinafter, referred to as a "second mixer") as a second intermediate frequency generating means for down-converting the intermediate frequency signal to generate a second intermediate frequency signal. Reference numeral 30 denotes a band pass filter for removing unnecessary components of the second intermediate frequency signal. Reference numeral 31 denotes a signal generator that generates a second local oscillation signal and supplies it to the second mixer 29.

The reference numeral 32 designates, according to the supplied DC control signal,
This is a VCO in which the oscillation frequency of the output signal changes. Reference numeral 33 denotes a distributor that distributes an output signal from the VCO 32 to a first local oscillation signal supplied to the first mixer 25 and a feedback signal for generating a DC control signal. Reference numeral 34 denotes a prescaler that divides the feedback signal obtained from the distributor 33 by ８. Numeral 35 denotes a feedback signal divided by the prescaler 34, which is further divided by a channel selection command from a controller (not shown in particular) to 1 / N (N is an integer from 107 to 192, and changes according to the channel selection signal). ) Is a programmable prescaler (hereinafter, referred to as a second frequency divider).

Reference numeral 36 denotes a reference signal generator for generating a reference signal. Reference numeral 37 denotes a phase error detection circuit (hereinafter, referred to as a phase comparator) that detects a phase error between a feedback signal obtained from the programmable prescaler 35 and a reference signal obtained from the reference signal generator 36 and generates a phase error signal. ). Reference numeral 38 denotes an amplifier for amplifying the phase error signal, and reference numeral 39 denotes a low-pass filter that extracts a DC component of the phase error signal and uses the DC component as a DC control signal to the VCO 32. That is, VCO
The loop circuit from 32 to the low-pass filter 39 is P
The LL circuit 40 is configured.

The PLL circuit 40 is realized by a PLL circuit having a double loop structure or a PLL circuit having a triple loop structure, which will be described later, in order to obtain a first local oscillation signal having good phase noise, which is a feature of the present system. You.

Next, the operation of the configuration shown in FIG. 1 will be described. FIG. 2 shows a frequency spectrum diagram of a multi-channel FM signal in one embodiment of the present invention. In FIG. 2, (a)
(B) is a diagram showing a multi-channel FM signal which is an input signal. In this input signal, as shown in FIG. 2A, the center frequency of each channel has a predetermined frequency interval of 38.36 MHz. As shown in FIG. 2B, the total number of input signal channels is 86,
The center frequency is f ₁ = 128.84 MHz to f ₈₆ =
Up to 3389.44 MHz, it is configured at a frequency interval of 38.36 MHz (this frequency configuration is called a channel plan).

After removing unnecessary signals of 100 MHz or less from the multi-channel FM signal by the high-pass filter 21 and absorbing the amplitude difference by the variable gain controller 22,
Amplified by 3. Further, unnecessary components such as high-band noise are removed by the low-pass filter 24, and the first mixer 25
Is supplied to the RF port. 38.36 from the VCO 32 via the distributor 33 to the LO port of the first mixer 25.
In MHz intervals, f _{L0 1 (1)} = first local oscillation signal corresponding to the 86 channels up f _{L01 (86)} = 7365.12MHz from 4104.52MHz is supplied.

In the first mixer 25, the input signal from the RF port is up-converted by the first local oscillation signal from the LO port, and the center frequency f _{IF (1)} = 3975.68 MHz.
Is generated. These frequency spectrum distributions are shown in FIG. This 3975.68M
The meaning of the intermediate frequency of Hz has the following two features.

One advantage is that the first mixer 25
This is to prevent the second harmonic component of the input signal from the RF port, which is generated in the above, from being generated in the band of the first intermediate frequency. That is, the half frequency 1 that has the greatest effect on the carrier having the center frequency of 3975.68 MHz.
987.84 MHz is 1987.84 MHz = 128.84 MHz + 38.36 MHz × 48.46, and the energy of this frequency component is equal to the frequency almost intermediate between f ₄₉ and f _{50 in} the channel plan shown in FIG. Therefore, the setting is made so that the interference is minimized.

Another advantage is that the frequency f _LO of the first local signal for generating this intermediate frequency signal is:
38.36 MHz. The reasons for the advantages are described below.

The frequency of the first local oscillation signal supplied to the first mixer 25 is supplied from the VCO 32. The oscillation frequency range of the VCO 32 is from 4.1 GHz to 7.4 GHz.
Although it does not exceed the octave, from the viewpoint of the phase noise of the oscillation spectrum, the influence of the phase noise during the free-run of the VCO 32 cannot be ignored.

Therefore, it is necessary to make the most of the C / N suppression effect of the PLL circuit 40.
It is preferable to increase the comparison frequency as much as possible. on the other hand,
The channel space is 38.36 MHz and the VCO32
The first-stage prescaler 34 that divides the frequency of the output has to be a fixed-type prescaler because the frequency-divided feedback signal is of a very high frequency. Therefore, the highest possible comparison frequency at this stage is 1/8 of 38.36 MHz, 4.79.
5 MHz is the limit. As a result, the frequency f _LO of the first local oscillation signal lockable at this frequency becomes an integral multiple of 38.36 MHz, and the first intermediate frequency f _IF becomes 3975.68 MHz.
It is convenient to limit to

For example, if the input signal has a frequency f ₁₁ = 51
Assuming a signal of the eleventh channel of 2.44 MHz, the first local oscillation frequency is given by the following equation, which is a multiple of 38.36 MHz.

The first intermediate frequency signal output from the first mixer 25 is 512.44 MHz + 3975.68 MHz = 4488.12 MHz = 117 × 38.36 MHz.
After passing through the band-pass filter 26, the narrow-band amplifier 27, and the band-pass filter 28, the signal is down-converted by the second mixer 29 to 402.78 MHz, which is the second intermediate frequency, and the band-pass filter 30 prevents interference of adjacent channels. It is detected by a detection circuit (especially not shown). Since the oscillation signal has a fixed frequency, the signal generator 31 can be constituted by a DRO oscillator having relatively good phase noise performance.

FIG. 3 is a block diagram showing the configuration of a PLL circuit applied to a multi-channel FM receiver according to a second embodiment of the present invention. In FIG. 3, reference numeral 41 denotes a reference signal generator for generating a reference signal, and reference numeral 42 denotes a distributor for distributing the reference signal. The distributor 42 has a function of outputting a digital reference signal (square wave) from the output D and an analog reference signal (sine wave) from the output A. Reference numeral 43 denotes a VCO whose oscillation frequency changes in accordance with a supplied DC voltage, ie, a tuning voltage. Reference numeral 44 denotes a prescaler that divides the frequency of an oscillation signal output from the VCO 43 as a feedback signal. Reference numeral 45 denotes a distributor for distributing the feedback signal divided by the prescaler 44 and sending out a first digital feedback signal from the output D1 and a second digital feedback signal from the output D2.

Reference numeral 46 denotes a digital phase comparator which detects a phase difference between the digital reference signal from the distributor 42 and the first digital feedback signal from the distributor 45 and outputs a digital phase error signal. Reference numeral 47 denotes a charge pump that removes the carrier component of the phase error signal and outputs a DC voltage. The digital phase comparator 46 and the charge pump 47 constitute digital phase error detecting means. 48 is a DC-isolated mixer,
An analog reference signal from the distributor 42 is supplied to the RF port, and a second digital feedback signal from the distributor 45 is supplied to the LO port. Further, a digital phase error signal is supplied as a DC signal from the charge pump 47 to a terminal 48a which is normally set to GND. Reference numeral 49 denotes a low-pass filter for removing a high-frequency component and a noise component of the output signal from the mixer 48.
43 and used as a tuning voltage.

Next, the operation of the PLL circuit of FIG. 3 will be described. If the analog reference signal V _s to be supplied to the RF port of the mixer 48, the phase of the second digital feedback signal V _f satisfies supplied to the LO ports are matched, the positive maximum of the analog reference signal V _s timing values are consistent with the rise timing of the second digital feedback signal V _f, the timing of the maximum negative value of the analog reference signal V _s is the timing of the fall of the second digital feedback signal V _f Matches. Thus, the analog reference signal V _s of pulses of the second digital feedback signal V _f satisfies the high level period is, the positive portion and the negative portion are equal, the integral value of this period is zero.

On the other hand, if the phase is shifted, the analog reference signal V _s of duration pulse is the high level of the second digital feedback signal V _f is the positive portion and the negative portion is different sizes Therefore, the integral value during this period does not become zero, and this integral value (a value including the sign) changes according to the amount of the phase shift. Therefore, by integrating the output signal from the IF port of the mixer 48 with the low-pass filter 49, a tuning voltage, which is an analog phase error signal corresponding to the phase error, is supplied to the VCO 43.

As shown in FIG. 3, the terminals 48 of the mixer 48
The DC phase error signal Vc output from the charge pump 47 is supplied to a. Therefore, a combined phase error signal in which the digital phase error signal is superimposed on the analog phase error signal is supplied to the VCO 43.

In general, when the phase comparator digitally performs phase detection, the higher the detection sensitivity, the more likely it is to be affected by disturbance noise such as ripple in the power supply voltage.
The C / N of the original phase information becomes worse. In this regard, analog phase detection can obtain better C / N. On the other hand, in the case of analog phase detection, C / N is improved because it can be constituted only by passive components, but it has a drawback that the output level of phase information decreases.

Therefore, as in this embodiment, by forming a double feedback loop of an analog system and a digital system in the PLL circuit, the analog phase error signal and the digital phase error signal are synthesized, and this is combined with the VCO signal. By using the analog phase error signal as a tuning voltage, a good C / N ratio can be obtained by the analog phase error signal, and a wide-band lock operation can be performed by the digital phase error signal.

FIG. 4 is a block diagram showing a configuration of a PLL circuit applied to a multi-channel FM receiver according to a third embodiment of the present invention. In FIG. 4, reference numeral 51 denotes a reference signal generator for generating a reference signal, and reference numeral 52 denotes a distributor for distributing the reference signal. The distributor 52 has a function of outputting a digital reference signal (square wave) from the output D and an analog reference signal (sine wave) from the output A. 53 is a VCO whose oscillation frequency changes according to the supplied DC control voltage tuning voltage, and 54 divides the frequency of the output oscillation signal from the VCO 53 as a feedback signal and has a frequency higher than the frequency of the reference signal. , An upper programmable counter for generating a first digital feedback signal. 5
A lower programmable counter 5 further divides the frequency of the first digital feedback signal divided by the programmable counter 54 to generate a second digital feedback signal having a lower frequency than the first digital feedback signal. 56
Is a distributor for distributing and outputting the second digital feedback signal divided by the lower programmable counter 55.

Reference numeral 57 denotes a digital phase comparator which detects a phase difference between the digital reference signal from the distributor 52 and the second digital feedback signal from the distributor 55 and outputs a digital phase error signal. A charge pump 58 amplifies and smoothes the phase error signal, removes the carrier component, and outputs a DC voltage. Digital phase comparator 57 and charge pump 58 constitute first phase error detecting means.

Numeral 59 denotes a lower-order mixer which is isolated in terms of direct current. Its RF port is supplied with an analog reference signal from the distributor 52, and its LO port is supplied with the second digital feedback signal from the distributor 55. Supplied. Further, a DC signal from the charge pump 58, that is, a digital phase error signal is supplied to a terminal 59a which is normally set to GND. Accordingly, the output port I of the lower mixer 59 is
From the F port, the digital phase error signal is superimposed on the phase error signal between the analog reference signal and the second digital feedback signal, and output as a first combined phase error signal. Reference numeral 60 denotes a low-pass filter that removes a high-frequency component and a noise component of the first phase error signal from the lower mixer 59. The lower mixer 59 and the low-pass filter 60 constitute a second phase error detecting means.

Numeral 61 denotes a frequency multiplier for converting the analog reference signal obtained from the distributor 52 into a higher harmonic analog reference signal containing the same frequency component as the output frequency of the higher-order programmable counter 54. Reference numeral 62 denotes a band-pass filter that passes the same frequency component as the output frequency of the higher-order programmable counter 54. This bandpass filter 62 limits the spectrum to a continuous spectrum corresponding to the desired lock frequency. An amplifier 63 amplifies the harmonic analog reference signal from the band pass filter 62.

Reference numeral 64 denotes a DC-isolated upper mixer similar to the lower mixer 59. The RF port of the upper mixer is supplied with a harmonic analog reference signal amplified to a predetermined level by an amplifier 63. The port is supplied with a first digital feedback signal from the programmable counter 54, ie, a harmonic digital feedback signal. Also,
A low-pass filter 6 is connected to a terminal 64a which is normally set to GND.
A first combined phase error signal from 0 is provided.

Therefore, from the IF port, which is the output port of the mixer 64, the first combined phase error signal is superimposed on the phase error signal between the harmonic analog reference signal and the second digital feedback signal, and Is output as a combined phase error signal. Reference numeral 65 denotes a low-pass filter that removes a high-frequency component and a noise component of the second combined phase error signal. The second combined phase error signal output from the low-pass filter 65 is used as a tuning voltage by the VCO 53
Will be fed back. The upper mixer 64 and the low-pass filter 65 constitute a third phase error detecting means.

As described above, according to the present embodiment, the digital phase comparison control loop, which is the first loop of the first phase error detecting means, and the second phase error detecting means of the second phase error detecting means.
A lower phase comparison mixer control loop, which is a loop of
A triple loop of the higher-order phase comparison mixer control loop, which is the third loop of the phase error detecting means, is formed.

Next, the reason for forming the triple loop of the third embodiment will be described. FIG. 5 is a configuration block diagram of a general single-loop PLL circuit.
Is divided by a 1 / N frequency divider 72 and input to a phase comparator 73. On the other hand, the reference signal from the reference signal generator 74 is also input to the phase comparator 73 to output a phase error signal. After being amplified by the amplifier 75, the comparison frequency is removed by the low-pass filter 76 to control the DC component. The signal is fed back to the VCO 71. Therefore, ideally, there is no problem with the configuration of the single loop shown in FIG.

However, there are actually various problems.
For example, only the phase noise signal at the time of free running existing in the output of the VCO 71 is directly frequency-divided into 1 / N on the frequency axis, and the frequency-divided phase noise signal is input to the phase comparator 73. , N are large, the frequency divider 7
2 will be added. This results in the loss of true phase noise information. Therefore, the smaller the value of N, the better.

FIGS. 6 (a) and 6 (b) are waveform diagrams showing the case where the comparison frequency is high and the case where the comparison frequency is low, respectively. In this figure, 81 is a waveform of the comparison frequency, and 82 is a waveform of the control voltage signal fed back to the VCO 71. In this figure, t _c indicates control timing, and t _d
Indicates the detection timing. As is clear from this figure, when the comparison frequency is increased in the phase comparator 73, finer control over time is possible. If the value of N is increased, the comparison frequency decreases.
The smaller the value, the better.

However, on the one hand, multi-channel FM
In the case of reception, as shown in FIG. 2, the upper limit of the comparison frequency always exists from the viewpoint of the channel space (38.36 MHz in this case). Raising it raises new problems.

Further, the cut-off frequency of the low-pass filter 76 is usually about 1/10 to 1/100 of the comparison frequency because the comparison frequency needs to be sufficiently reduced. As a result, the response of the control voltage signal fed back to the VCO 71 is reduced. Deteriorate. For example, the band characteristic of the phase noise signal during free-run of the VCO 71 is 1M as shown in FIG.
In the case of Hz, the improvement state of the phase noise changes depending on the characteristics of the low-pass filter 76. FIG. 7B shows a phase noise signal when the cutoff frequency of the low-pass filter 76 is 200 kHz, and FIG.
6 is a phase noise signal when the cutoff frequency is 800 kHz.

Therefore, when the phase noise signal of the VCO 71 during free-run is wideband and its frequency spectrum distribution is wide, there is a problem that sufficient control cannot be performed only by a single loop configuration.

Therefore, as in the third embodiment,
By using a triple loop structure in which two phase detection loops having different frequency spectra are provided in addition to the digital phase detection loop, it is possible to solve the above series of problems.

That is, in the first stage of the PLL control, the output level of the phase error signal obtained from the lower mixer 59 and the upper mixer 64 is extremely small.
It takes on the retraction operation. When the lock voltage is determined by the pull-in operation, the output frequency of the VCO 53 is first guided to a band close to the desired frequency. After that, in order to perform fine control, the first combined phase error signal
L control is performed. At the same time, PLL control is performed using second phase error signals having different frequency spectra.

In this case, the multiplier 61 is 4.79.
A harmonic signal of an integral multiple of 5 MHz is generated. FIG. 8 shows harmonic signals f _{S1 to} f _S extracted by the band-pass filter.
It is a frequency map of _S86 . Further, the harmonic signal f _S1 (frequency 513.
065 MHz) from the harmonic signal f _S86 (frequency 92
Frequency components up to 0.64 MHz) are extracted, amplified by the amplifier 63, and supplied to the upper-level mixer 64. As a result, in the upper mixer 64, the channel 1 is locked to the harmonic signal f _S1 and the channel 86 is locked to the harmonic signal f _S86 . Therefore, fine control in time is possible, and the cutoff frequency of the low-pass filter 65 can be increased. However, after locking, the tuning voltage, which is the control voltage signal, becomes 4.795.
MHz harmonics, the low-pass filter 65
Needs to have a cutoff characteristic for removing a side band of an integral multiple of 4.795 MHz.

In the present multi-channel FM receiving method, in the above-described embodiments (2) and (3), the frequency relationship is set so as to improve the C / N suppression characteristics. This is an extremely effective method.

[0051]

As is clear from the above description of the operation, the present invention is based on the channel plan according to the current BS format, and the multi-channel FM input signal is 3 to 128.84 MHz.
Of 86 channels up to 389.44 MHz, and
A method and an apparatus capable of selectively receiving an FM continuous wave having an interval of 38.36 MHz between adjacent channels without problems of image interference and local leak. The method and the apparatus include a second intermediate frequency and one wave of an input signal (397.36 MHz). ), Except for the interference occurring between
There is an effect that a wideband system that can exist independently from 8.84 MHz to 7365.12 MHz can be configured.

[Brief description of the drawings]

FIG. 1 shows a multi-channel FM according to a first embodiment of the present invention.
Configuration block diagram of receiving device

FIG. 2 is a frequency spectrum diagram of a multi-channel FM signal in one embodiment of the present invention.

FIG. 3 shows a multi-channel FM according to a second embodiment of the present invention.
Configuration block diagram of PLL circuit applied to receiver

FIG. 4 shows a multi-channel FM according to a third embodiment of the present invention.
Configuration block diagram of PLL circuit applied to receiver

FIG. 5 is a configuration block diagram of a general single-loop PLL circuit.

6A is a waveform diagram of a comparison frequency signal and a control voltage signal when a comparison frequency is high. FIG. 6B is a waveform diagram of a comparison frequency signal and a control voltage signal when a comparison frequency is low.

FIG. 7A is a band characteristic diagram of a phase noise signal at the time of free-run of the VCO. FIG.
(c) Cut-off frequency of low-pass filter 800 kHz
Phase noise signal characteristic diagram for z

FIG. 8 is a characteristic diagram of a harmonic signal extracted by a band-pass filter.

FIG. 9 is a configuration block diagram of an apparatus to which a conventional multi-channel FM reception method is applied.

[Explanation of symbols]

 Reference Signs List 21 high-pass filter 22 variable gain control unit 23 wide-band amplifier 24 low-pass filter 25 first mixer 26 band-pass filter 27 narrow-band amplifier 28 band-pass filter 29 second mixer 30 band-pass filter 31 signal generator 32 VCO 33 distributor 34 first Frequency divider 35 Second frequency divider 36 Reference signal generator 37 Phase comparator 38 Amplifier 39 Low-pass filter 40 PLL circuit

──────────────────────────────────────────────────続 き Continuing on the front page (72) Inventor Yasuhito Okawa 4-1-1 Tsunashima Higashi, Kohoku-ku, Yokohama-shi, Kanagawa Prefecture Inside Matsushita Communication Industrial Co., Ltd. Matsushita Communication Industrial Co., Ltd. (3-1) Matsushita Communication Industrial Co., Ltd. (72) Inventor Hiroshi Takahashi 4-3-1 Tsunashimahigashi, Kohoku-ku, Yokohama-shi, Kanagawa Prefecture Matsushita Communication Industrial Co., Ltd. JP, A) (58) Fields investigated (Int. Cl. ⁶ , DB name) H04B 1/16-1/28 H03D 7/16-7/18 H04N 7/10 H03L 1/00-7/26 H04N 7 / 20

Claims (4)

(57) [Claims] A multi-channel FM receiving a multi-channel FM input signal having a center frequency at a predetermined frequency interval.
A receiving method for generating a first local oscillation signal having an integer multiple of the predetermined frequency and a variable frequency, receiving the first local oscillation signal, upconverting the FM input signal, and
A gap of twice the frequency of an arbitrary frequency of the spectrum of the M input signal, and the frequency band of the FM input signal;
And generating a first intermediate frequency signal that does not overlap with any of the first local oscillation signal frequency band, and a constant frequency that does not overlap with any of the FM input signal frequency band, the first local oscillation signal band, and the first intermediate frequency band. Receiving the second local oscillation signal and down-converting the first intermediate frequency signal to generate a second intermediate frequency signal. 2. A multi-channel FM receiving a multi-channel FM input signal whose center frequency is formed at a predetermined frequency interval.
A receiving device, comprising: first local oscillation signal generating means for generating a first local oscillation signal having an integer multiple of the predetermined frequency and a variable frequency; and receiving the first local oscillation signal to upconvert the input signal. A first intermediate frequency signal that has a gap of twice the frequency of an arbitrary frequency of the spectrum of the FM input signal and does not overlap with any of the frequency band of the FM input signal and the first local signal frequency band. A first intermediate frequency generating means for generating, and a second station transmitting a second local signal having a constant frequency which does not overlap with any of the FM input signal frequency band, the first local signal band, and the first intermediate frequency band. Signal generating means, and second intermediate frequency generating means for receiving the second local oscillation signal and down-converting the first intermediate frequency signal to generate a second intermediate frequency signal. Multi-channel FM receiving apparatus that. 3. The multi-channel FM receiving apparatus according to claim 2, wherein said first local oscillation signal generating means is a PLL circuit having a double loop structure. 4. The multi-channel FM receiver according to claim 2, wherein said first local oscillator signal generating means is a PLL circuit having a triple loop structure.