JP2009171460A - Communication device, oscillator and frequency synthesizer - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a LINC transmitter that not only adopts a modulation system with such great envelop variations as caused in CDMA and OFDM but also strikes a balance between linearity and high efficiency of the transmitter. <P>SOLUTION: A communication device achieves wideband modulation where modulated data input in an oscillator is neither appeared for PLL to be disturbance nor limited by a PLL loop bandwidth by adopting a two-point modulation system in PLL 120a and 120b for carrying out frequency conversion of the modulated data. Further, replicas in the vicinity of sampling frequencies are significantly reduced by integration effect of the oscillator without adding an analog smoothing filter, so that the circuit reconfigurability is enhanced. <P>COPYRIGHT: (C)2009,JPO&INPIT

Description

  The present invention relates to a communication device, an oscillator, and a frequency synthesizer that are applied to advanced wireless communication and broadcasting such as digital wireless, and more particularly, a communication device that uses a modulation method with large envelope fluctuations such as CDMA and OFDM. , An oscillator, and a frequency synthesizer.

  More particularly, the present invention relates to a communication apparatus, an oscillator, and a frequency synthesizer using a LINC (Linear Amplifying Non-Linear Components) amplifier that achieves both linearity and high efficiency in a transmitter, and in particular, a broadband modulation signal. The present invention relates to a communication device, an oscillator, and a frequency synthesizer that handle and suppress spurious emission outside the band.

  Wireless technology plays a wide range of roles from high-capacity trunk lines such as terrestrial broadcasting, terrestrial microwave communication, satellite communication, and satellite broadcasting to access lines such as mobile communication. The wireless transmitter up-converts the transmission signal to a radio frequency (RF) band, further power-amplifies it, and releases it from the antenna into the air. Recently, modulation schemes with high frequency utilization efficiency such as CDMA and OFDM have been widely applied with the advancement of digitalization of wireless communication and broadcasting.

  Here, the code division multiple access (CDMA) system modulates a digital signal with a high-speed pseudo-random spreading code and spreads the frequency to transmit, but by giving a different spreading code to each user. One frequency channel can be shared by a plurality of users.

  Further, in OFDM (Orthogonal Frequency Division Multiplexing) modulation scheme, a plurality of data output by serial / parallel conversion for each symbol period is assigned to sub-carriers orthogonal to each other to perform inverse FFT. Thus, each subcarrier on the frequency axis is converted into a signal on the time axis and data is transmitted. On the receiving side, the reverse operation, that is, FFT (Fast Fourier Transform) is performed to convert the signal on the time axis into the signal on the frequency axis, and the subcarriers are demodulated in accordance with the respective modulation schemes. / Reproduce the information sent by the original serial signal after serial conversion. Here, subcarriers being orthogonal to each other means that the peak point of the spectrum of an arbitrary subcarrier always coincides with the zero point of the spectrum of another subcarrier, and there is no crosstalk. Therefore, since transmission data is distributed and transmitted to a plurality of carriers whose frequencies are orthogonal to each other, each carrier has a narrow band, has very high frequency utilization efficiency, and is resistant to frequency selective fading interference. The OFDM transmission system is applied to various broadband digital communication systems such as a wireless LAN (Local Area Network), terrestrial digital broadcasting, fourth generation mobile communication, and power line carrier communication.

  Here, CDMA and OFDM are modulation schemes that involve large envelope fluctuations. An amplifier that handles such a modulated signal is required to have very high linearity. This is because when a signal with a large envelope variation (ie, with a large peak factor) is amplified, signal distortion due to amplifier nonlinearity occurs at the peak where the envelope strength increases. Because it will end up. Usually, in order to obtain high linearity, the operating point of the amplifier is sufficiently backed off from the saturated state. For this reason, extra power other than the RF power that should be output is consumed as heat, and the power efficiency of the amplifier is significantly reduced. In a mobile radio communication terminal represented by a mobile phone or the like, a decrease in power efficiency directly leads to a decrease in battery duration, which impairs user convenience. In addition, reduction of power consumption is an important issue that should be considered from an environmental point of view, regardless of whether it is a moving object or a fixed object.

  LINC (Linear Amplifying Non-Linear Components) is one method that has been proposed to amplify linearly modulated waves with high efficiency by power combining to achieve both linearity and high efficiency (for example, Patent Documents). 1). LINC is based on the idea of expressing an arbitrary symbol point on the complex plane by combining two constant amplitude vectors, and operates the amplifier with saturated power to reduce the power consumption of the amplifier.

  FIG. 9 shows a configuration of a general LINC transmitter. The signal decomposing means 902 decomposes the linear amplitude signal into at least two constant amplitude angle modulation signal components. In the illustrated example, the signal decomposing means 902 decomposes the input linear modulation signal 901 into two constant amplitude signal components 905a and 905b. The amplifiers 903a and 903b amplify the constant amplitude signals 905a and 905b with a gain of G, respectively. The combining unit 904 combines the output signals 906a and 906b of the amplifiers 903a and 903b, and outputs a linear modulation signal 907. However, the amplifiers 903a and 903b are not required to have linearity, and non-linear and high-efficiency power amplifiers such as class C and class F can be used.

The operation of the illustrated LINC transmitter will be described with reference to FIG. A general modulation signal s (t) input to the signal decomposing means 902 can be expressed as the following equation (1). However, a (t) is an amplitude component with time fluctuation, ω _c is a carrier frequency, and φ (t) is a phase component due to modulation.

The signal decomposing means 902 generates constant amplitude signals s ₁ (t) and s ₂ (t) represented by the following equations (2) to (4). However, A _max is an assumed maximum amplitude, and is set to an appropriate value in advance according to the type of signal to be handled.

The constant amplitude signals s ₁ (t) and s ₂ (t) are amplified by the amplifiers 903a and 903b having the gain G, and then synthesized by the synthesizing means 904, whereby G · s (t) is output. can get.

Since each modulation signal component 906a and 906b has a constant amplitude, the output voltage of each of the width devices 903a and 903b has a constant value. The output power of the LINC amplifier has an amplitude component {cos θ (t)} ² and has a peak value when θ (t) = 0 degrees. Therefore, linear amplification can be performed by combining the two constant amplitude signals after amplification by the highly efficient and nonlinear amplifiers 903a and 903b.

  Here, the input signal 901 has been described as a signal whose RF carrier frequency is modulated. However, when the input signal 901 is a baseband signal, the constant amplitude signals 905a and 905b output from the signal decomposing means 902 are used. The signal is up-converted to an RF frequency band by a modulator (not shown) and then input to the amplifiers 903a and 903b. As the modulator, an orthogonal modulator, an angle modulator using a PLL (Phase Locked Loop), or the like can be used. Further, when the signal decomposing means 902 is mounted as a digital signal processing circuit such as a DSP (Digital Signal Processor), a D / A converter may be further required.

  For example, when the first amplitude data is larger than a predetermined value, the second amplitude data having the first data value equal to or larger than the predetermined value is generated, and the second amplitude data having the first data value is generated. The input signal is decomposed into the first and second output signals included to generate new phase data, and a signal having a large peak factor is amplified with high power efficiency without causing distortion. A proposal has been made on a data conversion apparatus that can perform the above (for example, see Patent Document 2).

  The present inventors consider that the conventional LINC transmitter as described above has several problems.

  First, when using a D / A converter (for example, when the signal decomposing means 902 is implemented by a digital signal processing circuit) and a quadrature modulator that upconverts a baseband signal to an RF frequency band, A replica (alias) near the sampling frequency output from the converter becomes a problem. In order to avoid such a problem, it is common to insert an analog smoothing filter (reconstruction filter) into the output of the D / A converter, but this leads to an increase in circuit scale. In addition, this is also a hindrance in achieving common circuit (or improving reconfigurability) between a plurality of different wireless systems, which has been rapidly demanded in recent years. If the sampling frequency is significantly increased, the smoothing filter may be unnecessary, but it is not preferable to increase the number of circuit blocks that operate at high speed.

  Second, when a PLL is used for the angle modulator for up-converting the baseband signal to the RF frequency band, the bandwidth of the modulation signal to be handled needs to be narrower than the loop bandwidth of the PLL. In other words, a broadband PLL is required. In the case of a LINC transmitter, the bandwidth of the constant amplitude signal is significantly wider than the bandwidth of the original linear modulation signal, which entails great difficulty.

  Third, when a PLL is used as an angle modulator for upconverting a baseband signal to an RF frequency band, sample points existing near the origin on the complex plane may not be accurately reproduced. As a result, there is a problem that distortion occurs in the synthesized spectrum. This problem can be serious when applied to systems with strict out-of-band emission standards.

JP 2000-349575 A JP 2005-39725 A

  An object of the present invention is to provide an excellent communication apparatus, an oscillator, and an LINC amplifier that can use a modulation scheme with large envelope fluctuations such as CDMA and OFDM, and uses a LINC amplifier that achieves both linearity and high efficiency. It is to provide a frequency synthesizer.

  It is a further object of the present invention to provide a communication device, an oscillator, and a frequency synthesizer that can handle a wideband modulated signal, have less spurious radiation out of the band, and have excellent reconfigurability.

The present invention has been made in consideration of the above problems, and the first aspect thereof is
A signal decomposing means for inputting a linear modulation signal of the first frequency band and decomposing it into at least two constant amplitude angle signal components;
At least two frequency conversion means for frequency-converting each constant amplitude angle signal component of the first frequency band to a second frequency band;
At least two amplifiers for power-amplifying each of the frequency-converted constant amplitude angle signal components,
It is a communication apparatus provided with a transmitter with combining means for combining the amplified constant amplitude angle signal components.

A second aspect of the present invention is an oscillator that is a component of a frequency synthesizer,
An input terminal that accepts a fixed-point digital frequency control signal;
A digital filter for filtering the input digital frequency control signal;
First resonance frequency variable means for discretely changing the resonance frequency of the oscillator controlled based on the integer part data of the digital frequency control signal after passing through the digital filter;
A dither processing unit that receives decimal part data of the digital frequency control signal and generates an integer data string having the same average value as the decimal part data;
Second resonance frequency variable means controlled by the dither processing unit;
An active circuit for sustaining oscillation;
It is an oscillator characterized by comprising.

  In addition, a third aspect of the present invention includes a PLL that connects a phase comparator, a low-pass filter, and the oscillator according to the second aspect of the present invention in a loop to generate a predetermined carrier frequency. A frequency synthesizer characterized in that

  In advanced wireless communication and broadcasting, modulation schemes with large envelope variations such as CDMA and OFDM are employed, and amplifiers that handle such modulation signals are required to have very high linearity. As a method of configuring an amplifier that realizes linearity without increasing power consumption, LINC that amplifies a linear modulation wave with high efficiency by power combining is known.

  The LINC transmitter to which the LINC amplifier is applied decomposes the linear modulation signal into two constant amplitude signal components, amplifies each, and then combines them to output a linear modulation signal. When the input linear modulation signal is a baseband signal, it is necessary to convert each low-amplitude signal after signal decomposition into an RF frequency band. Further, when the signal decomposition of the input signal is performed by digital signal processing, a D / A converter is further required.

  However, when frequency conversion of a modulation signal is performed using an angle modulator using a PLL, the bandwidth of the modulation signal to be handled needs to be narrower than the loop bandwidth of the PLL. In the case of a LINC transmitter, the bandwidth of the constant amplitude signal is significantly wider than the bandwidth of the original linear modulation signal, which entails great difficulty. In addition, the sample points existing near the origin on the complex plane may not be accurately reproduced, and as a result, there is a problem that distortion occurs in the spectrum after power combining.

  The communication device according to the present invention includes a LINC transmitter, and each frequency conversion means for converting a baseband constant amplitude angle signal component as a first frequency band into an RF frequency as a second frequency band includes a phase A comparator, a low-pass filter, and an oscillator are connected in a loop to form a PLL that generates a predetermined carrier frequency.

  An oscillator in the digital PLL circuit includes an input terminal for receiving a digital frequency control signal having a fixed-point number, a digital filter for filtering the input digital frequency control signal, and a digital frequency control after passing through the digital filter. First resonance frequency variable means for discretely changing the resonance frequency of the oscillator controlled based on the integer part data of the signal, and receiving the fractional part data of the digital frequency control signal to obtain the same average value as the fractional part data A digitally controlled oscillator comprising a dither processing unit for generating an integer data string, a second resonance frequency variable means controlled by the dither processing unit, and an active circuit for sustaining oscillation.

  In this way, replicas near the sampling frequency can be reduced by linearly interpolating data for controlling the oscillator with a digital filter such as an FIR filter. A system with strict standards for out-of-band radiation that avoids problems peculiar to LINC transmitters that use a PLL as a modulator (spectrum distortion due to sample points near the origin on the complex plane not being accurately reproduced) Application to is possible.

  Here, the active circuit for sustaining oscillation can be configured by an active circuit having a negative resistance value. Alternatively, the active circuit for sustaining oscillation may be a feedback circuit in which a plurality of delay elements are subordinate.

  Further, the first and second resonance frequency control means can change the resonance frequency by switching a plurality of capacitors.

  Alternatively, when the active circuit is a feedback circuit in which a plurality of delay elements are subordinated, the first and second resonance frequency control means can change the resonance frequency by switching the number of stages of the plurality of delay elements. it can. Further, the first and second resonance frequency control means can change the resonance frequency by switching the delay amounts of the plurality of delay elements.

  Further, in the LINC transmitter according to the first aspect of the present invention, each frequency converting means constituted by a digital PLL includes two carrier frequency data input terminals to the PLL and two frequency control signal input terminals to the oscillator. At the point, modulation is performed using each constant amplitude angle signal component of the first frequency band.

  In a normal PLL, the modulation signal is limited by the loop bandwidth because it acts to suppress disturbances input to the loop. On the other hand, in the present invention, by adopting the two-point modulation method as described above, the modulation data input to the oscillator is not seen as a disturbance to the PLL, and wideband modulation that is not limited by the loop bandwidth of the PLL is realized. It becomes possible to do.

  According to the present invention, it is possible to use a modulation method with large envelope fluctuations such as CDMA and OFDM, and an excellent communication apparatus, oscillator, and A frequency synthesizer can be provided.

  In addition, according to the present invention, it is possible to provide a communication device, an oscillator, and a frequency synthesizer that can handle a wideband modulated signal, have less spurious radiation outside the band, and have excellent reconfigurability.

  The communication apparatus according to the present invention employs a LINC transmitter to amplify a linear modulation wave with high efficiency to achieve both linearity and high efficiency while reducing the power consumption of the amplifier. In two systems, an angle modulator for converting a baseband input signal into an RF frequency band is configured by a PLL to which a two-point modulation method is applied. Therefore, the modulation data input to the oscillator does not appear to be a disturbance for the PLL, and it is possible to realize wideband modulation that is not limited by the loop bandwidth of the PLL.

  In addition, by adopting PLL for frequency conversion of the modulation data, the replica in the vicinity of the sampling frequency, which is a problem in the conventional LINC transmitter, is obtained by analog smoothing due to the integration effect of the oscillator that time-integrates the frequency dimension data. It is possible to greatly suppress without adding a filter, and it is possible to improve the reconfigurability of the circuit.

  In addition, it is possible to reduce replicas near the sampling frequency by linearly interpolating data for controlling the oscillator with an FIR filter.

  In addition, by linearly interpolating the data for controlling the oscillator with an FIR filter, there is a problem peculiar to a LINC transmitter that uses a PLL as a modulator (a spectrum caused by a sample point near the origin on the complex plane not being accurately reproduced). Distortion) and can be applied to systems with strict standards for out-of-band emissions.

  Other objects, features, and advantages of the present invention will become apparent from more detailed description based on embodiments of the present invention described later and the accompanying drawings.

  Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings.

  FIG. 1 shows the configuration of a LINC transmitter according to an embodiment of the present invention. The illustrated LINC transmitter includes a signal decomposing unit 110, PLLs 120 a and 120 b as phase modulators, amplifiers 130 a and 130 b, and a synthesizing unit 140. As described above, LINC is a system that amplifies a linear modulation wave with high efficiency by expressing an arbitrary symbol point on the complex plane by combining two constant amplitude vectors, thereby achieving both linearity and high efficiency. The amplifier is operated with saturated power to reduce the power consumption of the amplifier.

Baseband modulation signal data s (t) indicated by reference numeral 101 is input to the signal decomposing means 110 and decomposed into at least two constant amplitude angle signal components. The baseband modulation signal data s (t) is updated at a reference clock (not shown) cycle. In the illustrated example, the signal decomposing means 110 decomposes the baseband modulation signal data s (t) into two constant amplitude signals, and the modulation data s ₁ (t) and s ₂ indicated by reference numerals 102a and 102b, respectively. (T) and input to the subsequent PLLs 120a and 120b. These modulation data s ₁ (t) and s ₂ (t) are expressed in a manner suitable for controlling the frequencies of the PLLs 120a and 120b.

The PLLs 120a and 120b convert the baseband constant amplitude signal components 102a and 102b into an RF frequency band. These outputs 104a and 104b are RF phase modulation signals s _RF1 (t) and s _RF2 (t) modulated by the modulation data 102a and 102b with the frequency given by the carrier frequency data 103 as the center. It is preferable to perform a process (not shown) that compensates for variations in control sensitivity of the oscillators 124a and 124b.

  The outputs 104a and 104b of the PLLs 120a and 120b are input to the amplifiers 130a and 130b, respectively, and are amplified with a gain of G. Then, the synthesizing unit 140 synthesizes the output signals of the amplifiers 130a and 130b and outputs the linear modulation signal 105. The amplifiers 130a and 130b are not required to have linearity, and non-linear and high-efficiency power amplifiers such as class C and class F can be used.

  The PLL is a circuit that locks the carrier frequency to an accurate frequency. In this embodiment, the digital phase comparator, a low-pass filter, and a digital PLL in which an oscillator is connected in a loop are configured, and the digital phase comparator compares the phase of the reference signal and the output signal of the oscillator, It operates so as to reduce the phase difference. Further, as the oscillator, a digitally controlled oscillator (DCO) that switches frequencies discretely is used instead of the voltage controlled oscillator (VCO) that linearly changes the oscillation frequency according to the input voltage. A PLL circuit having a full digital configuration can be obtained. In the digital PLL, a time difference corresponding to a minor component of the frequency division ratio is converted into a digital value by a time-to-digital converter (TDC) circuit, and an integer component is converted into a digital value by an accumulator circuit. The digital value corresponding to the frequency ratio is fed back by various methods to control the oscillation frequency digitally.

  Here, in the normal PLL, the modulation signal is limited by the loop bandwidth because it acts to suppress disturbance input to the loop. On the other hand, in the present embodiment, a two-point modulation method in which frequency modulation data is injected in at least two places on a PLL loop used as a modulator is adopted, and the modulation signal is made invisible as a disturbance by the oscillator. As a result, wideband modulation that is not limited by the loop bandwidth is realized.

  In the example shown in FIG. 1, the PLLs 120a and 120b include reference phase accumulators 121a and 121b, phase comparators 122a and 122b, loop filters 123a and 123b, oscillators 124a and 124b, and variable phase accumulators 125a, respectively. , 125b. The digital phase comparators 122a and 122b perform phase comparison between the reference signal and the output signals of the oscillators 124a and 124b, respectively, and operate so that the phase difference becomes small. The reference phase accumulators 121a and 121b and the variable phase accumulators 125a and 125b generate phase data through an operation of adding or subtracting the phase data held therein and the input phase difference data.

Here, the modulation data s ₁ (t) and s ₂ (t) input to the oscillators 124a and 124b correspond to disturbances to the respective PLLs 120a and 120b. In each of the PLLs 120a and 120b, the modulation data 102a and 102b are added to the carrier frequency data 103 and input to the reference phase accumulators 121a and 121b. Since the reference signal to be compared is modulated, the modulation data input to the oscillators 124a and 124b cannot be seen as disturbance to the PLL. Therefore, it is possible to realize wideband modulation that is not limited by the loop bandwidth of the PLL.

  Let us consider the internal operation of the PLL 120a, 120b mathematically. The transfer function for the modulation data 102a and 102b from the input of the reference phase accumulators 121a and 121b to the PLL outputs 104a and 104b is a low-pass type, and is expressed by the following equation (5).

Where G (s) is an open loop gain. If the phase comparator gain is K _φ , the loop filter transfer function is Z (s), the oscillator gain is K _DCO , and the ratio of the output frequency to the reference frequency is N, then the gain G (s) is It is expressed as equation (6).

  On the other hand, for the modulation data 102a and 102b input to the oscillators 124a and 124b, the PLLs 120a and 120b exhibit a high-pass transfer function expressed by the following equation (7).

  It is obvious that the sum of the above equations (5) and (7) is the total response to the modulation signals of the PLLs 120a and 120b, and this is 1. That is, it becomes a transfer function of all band passing.

  In this way, the modulation data 102a and 102b are added to the carrier frequency data 103 input to the PLLs 120a and 120b and the input to the oscillators 124a and 124b (phase difference data after passing through the loop filters 123a and 123b), respectively. By adopting the two-point modulation method of performing wideband modulation, it is possible to apply wideband modulation that is not limited by the PLL bandwidth.

  Further, since the oscillators 124a and 124b time-integrate the frequency-dimensional data and output it as a phase, 1 / s is applied, and the output spectrum is suppressed with a slope of 6 dB / oct. As a result, replicas generated near the reference frequency are suppressed, and a replica suppression filter is unnecessary or simple.

  FIG. 2 shows the internal configuration of the oscillators 124a and 124b. The illustrated oscillators 124a and 124b include an FIR (Finite Impulse Response) filter 202 as a digital filter, an oscillator core 206 that is controlled by discrete control data and can take discrete oscillation frequencies, and a dither. A processing unit 207 and a frequency divider 210 are included.

  The oscillator control data 201 input to the oscillators 124a and 124 is (n + k) -bit data expressed by a fixed-point number consisting of an n-bit integer part and a k-bit decimal part, and the cycle of the reference clock. Updated every time.

  The FIR filter 202 operates by a clock 211 (higher speed than the reference clock) obtained by dividing the output of the oscillator core 206 by the frequency divider 210, upsamples the oscillator control data 201, and performs linear interpolation. The gist of the present invention is not limited to linear interpolation as an interpolation method, and other methods such as raised cosine may be used as long as the hardware scale is allowed.

  The output 203 of the FIR filter 202 is divided into n-bit integer part data 204 and k-bit fraction part data 205. The integer part data 204 controls resonance frequency control means (described later) provided in the oscillator core 206 to determine the oscillation frequency.

  The least significant bit (LSB) of the integer part data 204 corresponds to the minimum control unit of the resonance frequency control means and is input to the oscillator core 206 as it is. For example, when the resonance frequency control means is composed of a plurality of capacitors connected in parallel, each bit of the integer part data 204 corresponds to one corresponding capacitor. The fractional part data 205 represents one capacitor or less, and can be set to a desired value by inputting to the dither processing unit 207 and averaging.

  Similar to the FIR filter 202, the dither processing unit 207 operates by a clock 211 obtained by dividing the output of the oscillator core 206 by the frequency divider 210. The dither processing unit 207 outputs m-bit data 208 corresponding to the decimal part data 205 as an input to the oscillator core 206. The output 208 of the dither processing unit 207 controls the resonance frequency control means of the oscillator core 206 and realizes on average a frequency setting finer than the minimum control unit (a capacitor corresponding to LSB) of the resonance frequency control means. The dither processing unit 207 can be realized by using, for example, a ΔΣ modulator.

  If necessary, the oscillator core 206 can have a mechanism for dynamically removing non-linearity caused by mismatch of elements constituting the resonance frequency control means.

  FIG. 3 shows an implementation example of the oscillator core 206. The illustrated oscillator core 206 includes a negative resistance active circuit formed by a cross couple of n-type MOS (Metal Oxide Semiconductor) transistors M1 and M2, and a p-type MOS transistor M3 for supplying a bias current to the active circuit. And a tank circuit composed of the inductor L1 and the varactor bank 301, and a resonance type oscillator composed of a control unit 302 that converts the integer part data 204 and the output 208 of the dither processing unit 207 into a varactor control signal 303. is there.

  As shown in the figure, the varactor bank 301 is formed by connecting a plurality of MOS varactors using the change in gate capacitance according to the voltage applied to the gate of the MOS transistor in parallel. The tank circuit is a resonance circuit whose resonance frequency is determined by the inductance of the inductor L1 and the capacitance of the tank circuit. The active circuit has an action of sustaining this oscillation, and the resonance frequency becomes the output frequency of the oscillator core 206. Then, the control unit 302 controls the capacitance switching of each MOS varactor according to the integer part data 204 to discretely change the resonance frequency, and also has an integer data string having the same average value as the fractional part data 205 (that is, The capacitance switching of the MOS varactor is controlled according to the output 208) of the dither processing unit 207.

  Although FIG. 3 shows a configuration with a cross couple of n-type MOS transistors, a configuration in which a cross couple of p-type MOS transistors is further added, and a configuration in which a bias current is supplied by a tail current source of an n-type MOS transistor. It should be appreciated that several variations, such as a configuration using varactor diodes instead of MOS varactors, can be readily implemented by those skilled in the art.

  FIG. 4 shows another implementation example of the oscillator core 206. The oscillator core 206 shown in the figure is a ring oscillator composed of a delay chain 401 and a multiplexer 404 in which a plurality of delay elements are cascade-connected, and a control for converting the integer part data 204 and the dither processing part output 208 into a multiplexer control signal 303. The unit 302 is configured.

  As the delay element of the delay chain 401, for example, an inverter or a buffer can be used. Since the delay time for making a round of the ring oscillator differs depending on the path selected by the multiplexer 404, the oscillation frequency can be changed. That is, the feedback circuit in which a plurality of delay elements are subordinated has an action of sustaining oscillation as an active circuit, and the resonance frequency becomes the output frequency of the oscillator core 206.

  In the example shown in the figure, the method of controlling the number of stages of delay elements constituting the ring oscillator is shown. However, other implementations such as a method of changing the gate size of one stage of the delay element and a method of using both of them are used. You may take a method. Furthermore, as another realization means, an analog delay cell may be used, and a method of changing transconductance or a method of changing load capacitance may be adopted.

  In the LINC transmitter shown in FIG. 1, reference phase accumulators 121a and 121b, phase comparators 122a and 122b, loop filters 123a and 123b, which are elements constituting PLLs 120a and 120b, other than oscillators 124a and 124b, and loop filters 123a and 123b, and The variable phase accumulators 125a and 125b are preferably realized by digital hardware. This is because, when realizing the above-described two-point modulation with analog hardware, mismatch between elements becomes a problem, and it is often difficult to obtain sufficient performance. In addition, from the viewpoint of improving reconfigurability, the present inventors consider that realization with digital hardware is preferable. However, it should be appreciated that the subject matter of the present invention is not limited to the implementation of these circuit components in digital hardware.

  Next, the performance of the LINC transmitter shown in FIG. 1 will be described in detail with reference to simulation results.

  Here, an OFDM signal conforming to IEEE 802.11a was used as the test signal. The conditions in this simulation are 52 subcarriers, the primary modulation method is 64QAM (Quadrature Amplitude Modulation), the data rate is 54 Mbps, the modulation bandwidth is 20 MHz, and the reference clock is 8 times the modulation bandwidth (160 MHz). It is. For simplification of the simulation, a frequency equal to four times the reference clock is selected as the clock for driving the ΔΣ modulator as the dither processing unit 207. Similarly, the operation clock of the FIR filter 202 as a digital filter has a frequency equal to four times the reference clock. The parameters of the FIR filter 202 are such that the number of taps = 4 and all the tap coefficients = 1/4.

  FIG. 5 shows a spectrum simulation result when the LINC transmitter shown in FIG. 1 is modulated. For comparison, an output spectrum of a conventional LINC transmitter constituted by a D / A converter and a quadrature modulator is also shown.

  In the conventional LINC transmitter, as shown in the right of FIG. 5, replicas (aliases) near the sampling frequency output from the D / A converter exist at a large level, and there is no D / A converter output smoothing filter. The spectrum mask standard defined by IEEE802.11a cannot be satisfied.

  On the other hand, in the LINC transmitter of the two-point modulation method shown in FIG. 1, the replica is reduced as shown in the left of FIG. 5 by the integration effect that each oscillator 124a, 124b integrates the frequency dimension data. You will understand that analog circuits such as smoothing filters are not required. This means that it is very advantageous in sharing circuits (increasing reconfigurability) in devices corresponding to a plurality of different wireless systems. In addition, since fewer analog circuits are required, it is advantageous in terms of variation, and the effects of increasing the yield of the circuits and reducing the manufacturing cost can be obtained.

  FIG. 6 shows a simulation result showing the effect of frequency control data interpolation by the FIR filter 202 in the oscillator core 206.

  The upper part of FIG. 6 shows waveforms before and after the input of the FIR filter 202 of the oscillator control data 201 input to the PLLs 124a and 124b. The FIR filter has an effect of converging an output signal when an impulse is input to 0 in a finite time, and is often realized by a non-recursive difference equation.

  The lower part of FIG. 6 shows the phase of the output waveform of the oscillator core 206 based on the oscillator control data 203 after passing through the FIR filter 202. For comparison, a simulation result when frequency control data interpolation by the FIR filter 202 is not performed is also shown. In the case of no interpolation, the frequency data which has been zero-order held in the reference clock period is upsampled and linearly interpolated by four times by interpolating with the FIR filter 202, and as a result, at the output 209 of the oscillator 206. It can be seen that the phase waveform is smooth. That is, the component near the reference clock frequency is reduced.

  Further, as reference data, FIG. 7 shows a spectrum simulation result when data interpolation by the FIR filter 202 is not performed in the oscillator core 206. It can be seen that the component near the reference clock frequency is larger than the result of FIG. LINC transmitters that use PLL for frequency conversion of modulated data satisfy the IEEE802.11a specification sufficiently because reference replicas are inherently small due to the integration effect of the oscillator that integrates the frequency dimension data over time (described above). can do. However, considering that it corresponds to a system in which a stricter specification is defined, data interpolation by the FIR filter 202 has an effective effect in the oscillator core 206.

  FIG. 8 shows the configuration of a LINC transmitter according to another embodiment of the present invention. The illustrated LINC transmitter includes a signal decomposing unit 810, PLLs 820a and 820b as phase modulators, amplifiers 830a and 830b, and a synthesizing unit 140. The LINC transmitter shown in FIG. Symbol points are expressed and linear modulation waves are amplified with high efficiency to achieve both linearity and high efficiency (same as above).

The signal decomposing means 810 decomposes the baseband modulated signal data s (t) indicated by reference numeral 801 into two constant amplitude signals, and modulates data s ₁ (t) and s ₂ indicated by reference numerals 802a and 802b, respectively. (T) and input to the subsequent PLLs 820a and 820b. The PLLs 820a and 820b convert the baseband constant amplitude signal components 802a and 802b into an RF frequency band. These outputs 805a and 805b are RF phase modulation signals s _RF1 (t) and s _RF2 (t) modulated by the modulation data 802a and 802b around the frequency given by the carrier frequency data 803, and each amplifier 830a. , 830b and amplified with a gain of G, respectively. Then, the synthesizing unit 840 synthesizes the output signals of the amplifiers 830a and 830b and outputs a linear modulation signal 806.

  The LINC transmitter shown in FIG. 8 adopts a two-point modulation method in which frequency modulation data is injected at least two points on the loop of PLLs 820a and 820b used as modulators, and the modulation signal is a disturbance by the oscillator. By making it invisible, wideband modulation that is not limited by the loop bandwidth is realized, as in FIG. However, since 2-point modulation is different from FIG. 1, this point will be described in detail below.

  PLLs 820a and 820b that perform frequency conversion of the modulated data 802a 802b are phase comparators 821a and 821b, loop filters 822a and 822b, oscillators 823a and 823b, frequency dividers 824a and 824b, and ΔΣ modulators 825a and 825b, respectively. It is a digital PLL composed of In a loop composed of digital phase comparators 821a and 821b, low-pass filters 822a and 822b, and oscillators 823a and 823b, the phase comparators 821a and 821b perform phase comparison between the reference signal and the output signals of the oscillators 823a and 823b, Each operates so that the phase difference becomes small.

  The modulation data 802a and 802b output from the signal decomposing means 810 are added to the carrier frequency data 804 and then input to the ΔΣ modulators 825a and 825b. The ΔΣ modulators 825a and 825b modulate the frequency dividers 824a and 824b based on the input data. As a result, at the outputs 805a and 805b of the PLLs 820a and 820b, RF phase modulation signals modulated by the modulation data 802a and 802b with the frequency given by the carrier frequency data 804 as the center are obtained.

  Here, the transfer function for the modulation data 802a and 802b from the input of the frequency dividers 824a and 824b to the PLL outputs 805a and 805b is a low-pass type. The modulation data 802a and 802b are also input to the oscillators 823a and 823b. At this time, it is preferable to perform processing (not shown) for compensating for variations in control sensitivity of the oscillators 823a and 823b. For the modulation data 802a and 802b input to the oscillators 823a and 823b, the PLL exhibits a high-pass transfer function.

  In this manner, the modulation data 802a and 802b are added to the carrier frequency data 804 input to the PLLs 820a and 820b and the input to the oscillators 823a and 823b (phase difference data after passing through the loop filters 822a and 822b), respectively. By adopting the two-point modulation method of performing wideband modulation, it is possible to apply wideband modulation that is not limited by the loop bandwidth of the PLL.

  As the configurations of the oscillators 823a and 823b, any one of the digitally controlled oscillators shown in FIG. 3 or FIG. 4 may be selected according to the corresponding communication system.

  Further, the simulation result of the spectrum when the LINC transmitter is modulated and the simulation result indicating the effect of frequency control data interpolation by the FIR filter 202 in the oscillator core 206 are the same as those shown in FIGS. Therefore, explanation is omitted here.

  The present invention has been described in detail above with reference to specific embodiments. However, it is obvious that those skilled in the art can make modifications and substitutions of the embodiment without departing from the gist of the present invention.

  The present invention can be applied to various broadband wireless technologies such as wireless LAN, terrestrial digital broadcasting, fourth generation mobile communication, and power line carrier communication.

  In short, the present invention has been disclosed in the form of exemplification, and the description of the present specification should not be interpreted in a limited manner. In order to determine the gist of the present invention, the claims should be taken into consideration.

FIG. 1 is a diagram illustrating a configuration of a LINC transmitter according to an embodiment of the present invention. FIG. 2 is a diagram showing an internal configuration of the oscillators 124a and 124b. FIG. 3 is a diagram illustrating one implementation example of the oscillator core 206. FIG. 4 is a diagram showing another implementation example of the oscillator core 206. FIG. 5 is a diagram showing a spectrum simulation result when the LINC transmitter shown in FIG. 1 is modulated. FIG. 6 is a diagram showing a simulation result representing the effect of frequency control data interpolation by the FIR filter 202 in the oscillator core 206. FIG. 7 is a diagram showing a spectrum simulation result when data interpolation by the FIR filter 202 is not performed in the oscillator core 206. FIG. 8 is a diagram illustrating a configuration of a LINC transmitter according to another embodiment of the present invention. FIG. 9 is a block diagram showing a configuration of a general LINC transmitter. FIG. 10 is a diagram for explaining the operation of the LINC transmitter shown in FIG. 9, and specifically shows a state in which symbol points are expressed by combining two constant amplitude vectors on a complex plane. FIG.

Explanation of symbols

110 ... Signal decomposition means 120a, 120b ... PLL
121a, 121b ... reference phase accumulators 122a, 122b ... phase comparators 123a, 123b ... loop filters 124a, 124b ... oscillators 125a, 125b ... variable phase accumulators 130a, 130b ... amplifiers 140 ... synthesis means 202 ... FIR filters 206 ... Oscillator core 207 Dither processing unit 210 Frequency divider 301 Varactor bank 302 Control unit 401 Delay chain 402 Control unit 404 Multiplexer 810 Signal decomposition means 820a, 820b PLL
821a, 821b ... phase comparators 822a, 822b ... loop filters 823a, 823b ... oscillators 824a, 824b ... frequency dividers 825a, 825b ... [Delta] [Sigma] modulators 830a, 830b ... amplifiers 840 ... synthesis means

Claims (15)

A signal decomposing means for inputting a linear modulation signal of the first frequency band and decomposing it into at least two constant amplitude angle signal components;
At least two frequency conversion means for frequency-converting each constant amplitude angle signal component of the first frequency band to a second frequency band;
At least two amplifiers for power-amplifying each of the frequency-converted constant amplitude angle signal components,
Combining means for combining the amplified constant amplitude angle signal components with each other;
In the transmitter,
Each frequency converting means is composed of a PLL that connects a phase comparator, a low-pass filter, and an oscillator in a loop, and generates a predetermined carrier frequency,
The oscillator includes an input terminal for receiving a digital frequency control signal having a fixed-point number, a digital filter for filtering the input digital frequency control signal, and integer data of the digital frequency control signal after passing through the digital filter And a first resonance frequency variable means for discretely changing the resonance frequency of the oscillator controlled based on the following: an integer data string having the same average value as the decimal part data upon receiving the decimal part data of the digital frequency control signal; A dither processing unit that generates, a second resonance frequency variable means controlled by the dither processing unit, and an active circuit for sustaining oscillation,
A communication device. Each frequency conversion means uses each constant amplitude angle signal component of the first frequency band at two points of a carrier frequency data input terminal to the PLL and a frequency control signal input terminal to the oscillator. Apply modulation,
The communication apparatus according to claim 1. The active circuit for sustaining the oscillation is an active circuit having a negative resistance value.
The communication apparatus according to claim 1. The active circuit for sustaining the oscillation is a feedback circuit in which a plurality of delay elements are subordinate,
The communication apparatus according to claim 1. The first and second resonance frequency control means change the resonance frequency by switching a plurality of capacitors.
The communication device according to claim 3, wherein the communication device is a device. The first and second resonance frequency control means change the resonance frequency by switching the number of stages of the plurality of delay elements;
The communication apparatus according to claim 4. The first and second resonance frequency control means change the resonance frequency by switching delay amounts of the plurality of delay elements;
The communication apparatus according to claim 4. An oscillator that is a component of a frequency synthesizer,
An input terminal that accepts a fixed-point digital frequency control signal;
A digital filter for filtering the input digital frequency control signal;
First resonance frequency variable means for discretely changing the resonance frequency of the oscillator controlled based on the integer part data of the digital frequency control signal after passing through the digital filter;
A dither processing unit that receives decimal part data of the digital frequency control signal and generates an integer data string having the same average value as the decimal part data;
Second resonance frequency variable means controlled by the dither processing unit;
An active circuit for sustaining oscillation;
An oscillator comprising: The active circuit for sustaining the oscillation is an active circuit having a negative resistance value.
The oscillator according to claim 8. The active circuit for sustaining the oscillation is a feedback circuit in which a plurality of delay elements are subordinate,
The oscillator according to claim 8. The first and second resonance frequency control means change the resonance frequency by switching a plurality of capacitors.
The oscillator according to claim 9 or 10, wherein The first and second resonance frequency control means change the resonance frequency by switching the number of stages of the plurality of delay elements;
The oscillator according to claim 10. The first and second resonance frequency control means change the resonance frequency by switching delay amounts of the plurality of delay elements;
The oscillator according to claim 10. A phase comparator, a low-pass filter, and the oscillator according to any one of claims 8 to 13 are connected in a loop to generate a predetermined carrier frequency.
This is a frequency synthesizer. Adopting a two-point modulation method in which frequency modulation data is injected into at least two locations in the PLL loop;
The frequency synthesizer according to claim 14.