DK161172B - Installation for detection of an object within a surveillance area - Google Patents

Description

DK 161172 B

The invention relates to a system for detecting the presence of an object in a monitoring area comprising transmitting means for transmitting two radio frequency signals at two different frequencies in the monitoring area, the radio frequencies 5 differing from an average center frequency in each direction and with equal values, transponder means, detachably attached to protected objects and movable by an object in the monitoring area, said transponder means having antenna means tuned to receive the radio frequency signals transmitted at both frequencies, and a non-linear impedance element coupled to the antenna means wherein said transponder means re-radiating a return signal at the frequency equal to the sum of the frequencies of the two transmitted radio frequency signals, which facility further comprises antenna means for receiving the return signal, narrow-band receiving means for processing the received return signal. ignal and alarm means capable of responding to an output signal produced by the signal processing of the received return signal by the narrow band receiving means.

20

Known monitoring systems> of this type - cf. U.S. Patent No. 4,063,229 - transmits a single radio frequency sensed by an antenna on a transponder label, wherein a non-linear impedance such as a semiconductor diode generates a selected harmonic of the transmitted signal re-radiated for detection by a receiver circuit for eliminating the transmitted frequency. However, such systems are not satisfactory in practice, as they are not sufficiently sensitive to provide a reliable detection of the presence of a transponder in the monitoring area and may give rise to false alarms in dependence on other objects present. The nonlinear characteristics of the transmitter circuit and the elements often resulted in harmonics being transmitted together with the fundamental frequency, whereby the receiver could respond without a non-linear impingement element in the transponder. If the receiver's sensitivity is reduced to not responding to such directly transmitted 2

DK 161172 B

harmonics, weak harmonics that are re-radiated in certain circumstances can be masked. Although this problem can be minimized by appropriate shielding and RF filtering in both the transmitter and the receiver, the filters must still be provided with 5 extremely sharp cut-off characteristics such that even a small frequency drift in the transmitted signal - which is multiplied in the harmonic - can result in that the re-radiated frequency falls outside the receiver band of the receiver. Frequency operation can also result from Doppler power by rapid movement of the transponder within the monitoring area so that transmitter operation is changed.

On the other hand, such high frequency signals can easily propagate outside the intended monitoring area, thereby giving rise to false triggering of the alarm by a remote transponder. Protected objects as a result cannot be located or handled in the vicinity of the monitoring area. Even then, high-frequency energy can propagate through unpredictable reflections or through metal rods or energy conductors acting as waveguides to and from remote positions within the protected structure, thereby giving rise to false alarms of the alarm system.

Such systems are also subject to false triggering of metal objects such as umbrellas, prams and shopping carts, where a welding or contact point between dissimilar metals forms a non-linear diode which generates or radiates a harmonic of the transmitted signal. The receiver may also respond to false radio frequency noise from other sources such as mc-30 ignition systems and electronic equipment.

Conversely, the system cannot respond to the instantaneous presence of a transponder element in the monitoring area if the energy captured by yog is radiated as a harmonic is insufficient. Eg. this could happen if the transponder antenna was misaligned with respect to the polarization of the transmitted field, or if the antennas' electromagnetics 3

DK 161172 B

shielded from the transmitter of the human body or of a metal surface. The location of the transponder in the vicinity of the human body can also adjust the resonant frequency so that the harmonic energy for re-radiation to the receiver 5 is dissipated. Although a signal track circuit can adjust the frequency response of the receiver to compensate for the transmitter frequency operation, the efficiency of the transponder is nevertheless reduced if the tuned tank circuit is forced to oscillate at frequencies other than the normal resonant frequency.

10

Efforts to solve the problems of such systems have resulted in several solutions. In one of these (U.S. Patent No. 3,631,484), the individual radio frequency is transmitted to the transponder for re-radiation as a harmonic compared to the signal captured by the receiver for detecting Doppler frequency offsets caused by a movement of the transponder. Although this system eliminates problems associated with transmit frequency operation and false alarms from nearby stationary transponders, an object moving slowly through the monitoring area would still not produce a Doppler frequency shift that could trigger the alarm.

Systems have been investigated in which the non-linear impedance element of the transponder acted as a signal mixer for generating 25 sum and difference frequencies in dependence on two transmitted signals of different frequencies - cf. U.S. Patent No. 3,395,368. Such dual frequency mixing systems have several drawbacks, for example, limitations of the high frequency transmissions to the intended monitoring in range. To solve this problem, Gorden et al. Patent discloses the use of a dual-field system employing a high-frequency electromagnetic field in conjunction with a powerful low-frequency electrostatic field between discontinuous conductors disposed on either side of the monitoring area. The non-linear impedance element exposed to these two fields acts as a mixing step forming the sum and difference frequency re-radiated to the receiver for detection. Cen energy that 4

DK 161172 B

required to establish the required electrostatic field in the monitoring field is substantial and such low frequency electrostatic fields can be effectively shielded from the transponder by the human body or by a surrounding conductor or diverted from the transponder via the meta1 structure of a cart or the like. electrostatic fields can also be easily diverted via nearby pipes and other metal structures to remote locations, thereby giving rise to false trigging by means of markings without; 10 for the monitoring area. The problem of false alarms due to disparate metal connections in metal wagons and the like was exacerbated by concentration of the electrostatic field through such metal structures.

The object of the invention is to provide a plant of the above-mentioned type in which the stated disadvantages are eliminated. An installation of the kind mentioned initially is according to the invention in the n-dozen characteristic in that it has multiple antenna transmission means for both radio frequency signals, and which are arranged 20 in opposite positions, defining the monitoring area such that the ratio of the field strength of the two signals are substantially uniform across the monitoring area, and the narrow band receiving means signal processing is the received return signal to exclude the transmitted row in frequency signals and their harmonics to provide the output of the alarm means.

The transponder preferably takes the form of a folded dipole antenna with the impedance element e.g. a semiconductor diode coupled between the opposite sides of a closed loop section at one end to form a tuned tank circuit with a resonant frequency twice that of a selected center frequency. The longer antenna portion extending from the diode is approximately equal to a quarter of a wavelength at the selected center frequency, e.g. can be 915 MHz. The resonant frequency of the tank circuit, which is determined by the capacitance of the diode and the inductance of the present closed loop section of the antenna, is the double 5

DK 161172 B

belt of the selected center frequency (eg 1830 MHz). Two different radio frequency signals are both emitted from bellows radiating dipole antennas located on either side of a monitoring area. One of these signals is produced as a continuous wave from a very stable crystal oscillator source at a fixed frequency (e.g., 905 MHz) offset from the selected center frequency by about 1%. The second transmitted signal is tone modulated preferably with an audio signal in the range 1-20 kHz to produce a high frequency fluctuation of t 5 kHz 10 in the carrier frequency, which is also derived from a very stable crystal oscillator source at a frequency (e.g. 925 MHz). which is offset to the same degree from the selected center frequency on the opposite side such that the mean of the two signals is equal to the selected center frequency. Both transmitter signals are radiated across the monitoring region from di pole antenna segments which are oriented perpendicular to each other on the same side of the region, the respective di pol segments for radiating the same frequency from opposite sides are also oriented perpendicular to each other. This leads to cross-clarification in the monitoring region of the radio frequencies emitted from opposite sides, thereby ensuring that the radiation of both frequencies in the monitoring region between the transmitters is sufficient in all directions for any orientation of the tag, while propagating the two signals. from the antennas on only the one side to the same remote location and outside the monitoring area is substantially decreased due to different polarizations.

Dual frequency operation substantially decreases the efficiency of transmit frequency operation and increases the bandwidth of the system relative to the efficiency of the transponder by re-emitting incident radio frequency signals. In particular, the frequency to which the transponder is tuned may lie anywhere between the two transmitted frequencies without thereby reducing the efficiency of the transponder 35. This eliminates any need for precise antenna sizing. Furthermore, the problems are minimized by 1-body mood, with the normal 6

DK 161172 B

mood of the transponder is shifted downward in frequency due to the di-electric load effect of a human body in contact with or near the label. For example, the transponder antenna is tuned downward from the selected center frequency, this only increases the efficiency of the transponder relative to the lower transmitted frequency, and the overall mixing effect is not very much affected by the fact that the proper mixing occurs with a power ratio of ten to one or more. Similarly, the effects of transmitter frequency drift are minimized in that a shift 1 of one of the transmitters is not multiplied as re-radiated harmonics in single frequency systems, and any operation in one of the transmitters can be offset by an opposite operation in the other transmitter.

j 15 The strength and frequency stability of the re-emitted transponder signal and the likelihood of triggering a false reproduction due to transponders outside the monitoring area allow for a maximum receiver sensing down and a minimum receiver bandwidth. Signals received from circularly polarized receiver antennas on each side are passed through a very narrow bandpass that rejects transmit frequencies and then amplifies such that in the on-tone can be derived by conventional demodu 1 in lighting techniques. The audio tone (e.g., 2 kHz) is preferably used to frequency modulate the RF carrier so that the filtered and amplified signal from the receiving antenna can be applied to a passive dual balanced mixing stage receiving a sub-side injection signal (e.g., 1808,600 MHz) generated by of a stable local oscillator to provide a suitable flour at 1 em frequency, for example 21.4 MHz at the mixing output. This MF output signal from the mixing stage is amplified and applied to another precision filter with a narrow pass band (e.g., 30 kHz) defining the pre-detection bandwidth. Detection of the modulat ion tune is performed by a narrow-band (e.g., 30 kHz) crystal discrimination whose output is fixed to ground until its input is of a strength sufficient to generate an AGC detector voltage which exceeds - 7 increases a preselected reference level which is adjusted to adjust the system sensitivity. With the fixation open, the tone is applied to a phase-locked s phetone decoder circuit whose voltage controlled oscillator has a free-running frequency equal to the frequency of the tone and capable of achieving a stable tone within a narrow frequency range (e.g., ± 10%). When the loop obtains the tone signal, a quadrature detector senses the phase-locked state and produces a DC output voltage to operate an operational amplifier with a capacitive feedback that maintains an output signal for triggering an alarm for a minimal time period, e.g., 3 sec. how short the duration of the detected tone is. By means of this means, the alarm is activated no matter how short the transponder stays in the monitoring area when the detected signal is once of sufficient strength and has the proper modulated frequency content. This eliminates false alarms due to weak return signals from transponders outside the monitoring area and due to signals from unauthorized sources which can produce signals corresponding to the 20 re-radiated frequency but lacking the necessary tone modulation.

The invention will be explained in more detail below with reference to the drawing, wherein 25 f in g. 1 shows an object monitoring device according to the invention, comprising a transmitter and a receiver; FIG. 2 shows a more detailed illustration of the transmitter antenna segments, further showing a non-linear impedance element of a transponder; FIG. 3 is a diagram of a narrow-band tone modulated RF transmitter in the monitoring system; FIG. 4 a diagram of a C.W. RF transmitter in the monitoring system, 35 8

DK 161172 B

FIG. 5 is a diagram of linear amplifiers in the monitoring system in g. 1 and FIG. 6 is a diagram of the narrow band tone modulated receiver of FIG. 1, wherein the transmitted signal is frequency modulated.

FIG. 1 illustrates an apparatus according to the invention for monitoring objects. Rows of transmitter and receiver antennas are mounted on freestanding pedestals 10 and 12 or on or in existing door frames on either side of a monitoring area, typically at the entrance or exit of a retail establishment so that anyone entering or exiting , must pass the space between the antennas. Although slightly illustrated for the purposes of the illustration, the respective antenna rows on each side usually face directly against each other with the respective antenna elements arranged in parallel vertical planes. Both rows of transmitting antennas consist of orthogonally arranged pairs of metal strips segments 18, 19, 20 and 21 mounted on a vertical flat backing on either side of the 20 protected approach or protected area. Each strip extends from a central hub region with individual pairs aligned to form a conventional center-fed dipole antenna that is approximately a quarter of a wavelength long for the frequency being transmitted and may advantageously be oriented as shown to extend say horizontally and vertically. The individual strips 18-21 may be cut from conventional cloth tape like copper coating of the type commonly used in conjunction with printed circuit boards and applied to a non-conductive dielectric substrate with suitably small losses on the pedestal or door frame. , or the four strip rows can simply be etched out by removing the surrounding conductive plate on a printed circuit board. A conductive metal panel or small mesh grid (not shown) may be located behind and parallel to the plane 35 of the antenna strips 18-21 for reflection and concentration of the transmitted signal energy and radiation pattern over the protected space for greater efficiency and sub-9.

DK 161172 B

printing of radiation from other side to areas behind pedestals 10 and 12. In the preferred embodiment, the copper coating strips are applied to the surface of a G-10 fiberglass panel which is fixed by adhesive in a 5 1 one-weight anodized aluminum frame covering the entire back side of pedestal 10 or 12 and structurally supports the antenna mount with associated circuit elements.

On each side are also mounted antenna receivers 22 and 24, 10 which are circularly polarized, for example the cross-folded dipole configuration, usually referred to as a "turn-style" antenna or a coiled in antenna. The length of each receiver dipole segment must be a quarter of a wavelength of the frequency of the re-radiated signal, which, as the d will be described below, is equal to the sum of the two transmitted frequencies.

Two different radio frequency signals 1s f 1 and f 2 are generated for radiation from the respective dipole strip segments 20 '19, 20 ° 9 21, which form the transmitter antenna rows 14 and 16. The f 1 signal is a narrow band modulated radio frequency generated by of a very stable? oscillator 26 coupled to the vertical dipole strip segments 18 of the transmitter antenna array 14 on one side and also via a linear amplifier 28 25 to the opposite horizontal strip segments 21 of the end terminal array 16 on the other side of the monitoring region -t. The second transmitter signal f2 is similarly generated at a fixed radio frequency by a very stable oscillator 3C, i.e. to the horizontal strip segments 19 of the transmitter antenna array 14 on one side and on the other side via a linear amplifier 32 to the vertical strip segments 20 disposed in the transmitter's antenna array 16. Both oscillators 26 and 30 preferably use temperature compensated crystal oscillators with cascade coupler couplers and narrow-band bandpass fi lms for generating the continuous wave f2 and the radio frequency carrier of the tone modulated signal f1 as will be described in more detail in connection with Figs. 3 and 4.

10

DK 161172 B

The distance between the antenna segments 18-21 of the metal strip and the adjacent reflecting surface of the conductive panel or lattice, which depends on the thickness of the low dielectric constant, is selected to provide a standwave ratio (VSWR) adapted to the antenna input impedance with the output impedance of the respective transmitter signal source at the transmitted frequency so as to provide an effective radiation pattern with an approximately 60 ° beam extending from the transmitter antenna rows 14 and 16 on each side.

Thus, both radio frequencies f1 and f2 are radiated from the transmit rows 14 and 16 on either side with opposite polarizations for intersection and penetration from both sides by a transponder 15 34 located in the monitoring area between the two pedestals 10 and 12. The transponder 34 is shown in FIG. 1 as a circularly polarized helical in-antenna loop with a diode 36 coupled over a short closed section of the loop. As shown in more detail in FIG. 2, in its preferred embodiment, transponder 34 consists of an elongated flat metal antenna loop 38 with a central gap on one side to form a folded dipole configuration. The total antenna length is ideally a quarter of a wavelength of the center-center frequency between the two transmitted radio frequencies f ^ and f2. The non-linear impedance element 36, in the form 25 of a semiconductor diode, is interposed between opposite sides of the loop near one end approximately midway from the side gap so that the capacitance of the diode 36 with the inductance of the nearby closed end of the conducting loop forms a tank circuit with a resonant frequency equal to or approximately equal to the sum of the two transmission frequencies fj and f2 / or in other words a resonant frequency twice the frequency of the selected center center frequency of the transmitted signals. The precise location of the diode 36 on the antenna loop 38 to produce the desired resonant frequency for the tank circuit is not essential and is, for the most part, empirically determined by the capacity of the selected diode and the wiring characteristics of the antenna loop. The short straight line

DK 161172 B

The π number segment of the di odes in that of the gap serves as a quarter-wave d in pole antenna at the resonant frequency of the tank circuit.

Maximum transponder efficiency and selectivity are achieved 5 where the frequency difference between the two transmitter frequencies fi and f 2 is about 1% of their center 1 center frequency. In the power version of the system, the frequency of the continuous wave signal f2 is generated by the source 30 at a frequency of 905 MHz, while the frequency of the tone modulated carrier for the second transmitted signal f from the source 26 is at 925 MHz. The mean center frequency is thus 915 MHz, while the center frequency of the tank circle is 1.830 MHz. These frequencies are chosen to be within the range available for such purposes in the United States. In order to satisfy international broadcasting standards, the system can e.g. is dimensioned with a resonant frequency for the tank circuit of about 4,900 MHz at transmitter frequencies of about 2,420 and 2,480 MHz.

20 When both transmitted signals fi and f 2 are received by transponder antenna loop 38, due to the non-linear impedance of semiconductor diode 36 to initiate oscillations at the resonant frequency of the tank circuit, the resonant frequency is equal to the sum of fi and f 2. Mixing and efficiency are improved by a planar ode with fast interrupt, low RF threshold and low bias. Cheap germanium diodes are preferred due to their relatively low threshold of about 0.3 V compared to more expensive silicon diodes with threshold values of about 0.6 V.

30

The frequency separation of approximately 2% between the transmitted signals provides significant advantages in maximizing the efficiency of the transponder and in terms of the ability of the system to avoid false alarms as a result of the return signal of the transponder 35 being separated from the signal which could possibly be produced by various metal objects such as sunglasses, shopping carts and the like, which items tended to 12

DK 161172 B

causing false alarms by the previous systems. In particular, the bandwidth of transponder 34 relative to the incoming radio frequencies is expanded without reducing the efficiency as the receiver antenna 33 can be tuned to any frequency between the two transmitting frequencies, which also minimizes the effects of tuning the downward frequency offset as a result. of such dielectric load effects are easily accommodated within this range. This is because tuning or tuning of the antenna 38 10 more towards one transmitter frequency than against the other transmitter frequency only serves to increase the signal strength at this frequency without reducing the efficiency of the mixer due to the fact that proper radio frequency mixtures can be made with power ratios of 10 to one or more between the signals.

15

Due to the cross-polarization of the two frequencies transmitted from each of the antennas 14 and 16, their propagation from one transmitter position to distant positions outside the monitoring area is rarely the same for both signals. A strange reflection pattern that can result in one transmitting signal being concentrated by a transponder in a remote position will almost never result in the other opposite polarized transmission reflecting with the same pattern and reaching the same region with a significant power . If only one signal is received, the non-linear impedance of the diode 36 can only provide a frequency doubling instead of the required mixing power such that the resulting return signal is at a frequency much offset from the frequency of the desired transponder feedback. . With the current system parameters, a transponder will produce frequency doublings at 1,810 or 1,850 MHz, both offset 20 MHz from the normal return frequency at 1,830 MHz. These offset frequencies can be significantly attenuated in tuned tank forces and are facilitated by conventional filtering techniques. distinguishable from a true mixed frequency response at 1.830 MHz.

In this regard, signals are collected by means of the receiver antenna 22 and 24 on each side supplied via a convention in o-13.

DK 161172 B

The mixing of the two transmitted signals in the return signal of the transponder allows a restriction of the reproduction of the receiver 42 to a very narrow band operation which serves to eliminate false alarms due to unauthorized noise or transmission signals. from other sources. The required receive bandwidth is, for the most part, dependent only on the frequency stability of the transmitters 26 and 30. A very narrow detection window corresponding to the possible operation of the transmit frequency 10 is thus possible. With very stable transmitter oscillators, which will be described below, the bandwidth of the received signals available for detecting the modulated tone can be very small and the bandwidth of the receiver (after detection) can be narrowed by accurate detection of the modu-15 1 that in the onstone. The reliability and sensitivity of the system is further enhanced by the receiver 42 having an output signal for activating only an alarm 44 when the strength of the detected tone modulation signal exceeds a selected minimum amplitude level for a predetermined fixed interval to ensure the 20 instantaneous presence of a transponder in the detection zone.

Referring now to FIG. 3. The preferred embodiment during operation generates the transmitter signal f 1 as a very stable narrow band frequency modulated signal to maximize system sensitivity and selectivity. A stable tone generator 46 of conventional design, e.g. can be of a simple Retype, generating a fixed tone frequency in the audio range of 1-20 kHz. This tone, which is in the current system at 2 kHz, is applied as a modulation signal to a voltage controlled crystal oscillator 48 for frequency modulation of its output. In the preferred embodiment, the crystal oscillator 48 is of conventional design with accurate temperature compensation capable of maintaining a frequency stability of 0.7 35 ppm from 5 ° C to 45 ° C at a frequency of approximately 51.4 MHz. The amplitude of the modulation signal from the tone generator 46 applied to the voltage controlled signal is adjusted to

DK 16117 2 B

u produce a maximum frequency deviation of only plus or minus about 0.25 to 0.30 kHz. This results in a very narrow band modulation of the carrier oscillator. The modulated output signal 1 of the oscillator 48 is then applied to 5 a conventional frequency multiplier 50 which triples the oscillator frequency and then supplies the triple frequency to a narrow band bipolar bandpass 52. This filtered multiplied signal is then applied to a second frequency multiplier 54. again triples the frequency, which triples frequency is applied to another narrow bandpass filter 55. The filtered output of bandpass filter 56 is then applied to a second frequency multiplier 58 which doubles the input frequency to produce the desired modulated output signal fi at 925 MHz The output of plus or minus 5 kHz which is then fed to an RF amplifier 60 with variable amplifier and a power amplifier 62. The amplified transmitter signal fi is passed through a narrow band three-pole bandpass filter 64 to a power divider 66 which supplies the transmitter signal to d e vertical antenna 20 strips 18 on transmitter row 14 of pedestal 10 and via a lightweight cable connection to linear amplifier 28 on second pedestal 12.

Referring now to Fig. 4, the second transmitting frequency f2 is generated in a corresponding fashion by means of a conventional temperature compensated crystal oscillator 68 capable of maintaining the frequency of 0.5 ppm from 5 ° C. 45eC with an output frequency of about 50.3 MHz. This output frequency is tripled by frequency multiplier 70 and filtered 30 by a two-band bandpass 72. The narrow band output of filter 72 is then applied to another frequency multiplier 74 which in turn triples the frequency, which triples frequency through another bipolar bandpass filter 76. The filtered output frequency is then doubled in a last frequency multiplier 78 to produce the desired f2 signal at 905 MHz. The f2 signal is applied to the input of a variable gain RF amplifier 80 and 15

DK 161172 B

to the additional amplifier stage 82 to obtain a desired power level. The amplified output is then filtered via a narrow band three-pole bandpass filter 84 for removing distortions or harmonics and supplying the signal to a power divider 86 for supply to the antenna strips 19 and transmitter sack 14 on the pedestal 10 and via an appropriate RF connection to the respectively linear amplifier 32 on the opposite pedestal 12. Due to the high efficiency and high sensitivity, the transmitting power of these signals is of a larger order than that required in previous systems. Thus, the possible effects of microwave transmissions on tissue and health can be disregarded.

In FIG. 5, the respective fj and ^ 2 signal outputs of the power divider 66 and 86 can be connected to the respective linear amplifiers 28 and 32 of the facing antenna pedestal 12 by simple wire conductors or lightweight cables. This eliminates the need for expensive and difficult installations of the heavy and heavy RF cable connections required by 20 previous systems to avoid losses. Linear amplifiers 28 and 32 consist of a variable RF amplifier step 88, the output of which is passed through a narrow-band three-pole bandpass filter 90 to remove signal distortions or noise collected on the connection line or generated during the gain. The splash of amplifier stage 88 is adjusted to restore the transmit signal strength to the same level applied to the transmit antenna segments on the opposite side.

Ten are now referenced! FIG. 6. In the preferred embodiment, using the narrow band frequency modulation of the transmit signal, the signals are collected by the receiver antennas 22 and 24 and via the mixing step 40 a narrow band four-pole bandpass filter 92 whose passband is centered at the average frequency of the mixed transponder returns igna1 - e.g. at 35 1830 MHz. In the particular system described, a valid return signal from transponder 34 is frequency modulated with a single fixed audio tone, preferably at 2 kHz, provided

DK 161172 B

bringing a maximum deviation of only 5 kHz on each side of the 1830 MHz carrier frequency. The bandpass is adapted to reject the 1-frequency transmit signals with a minimum of 60 dB to prevent internal mixing as a result of circuit 1 obesity. A filtered output of the bandpass filter 92 is applied to a double balanced mixing step 94 for mixing with a lower side injection frequency f3 at 808,600 MHz, e.g. from a stable local oscillator to produce an average frequency output (IF) of about 21.4 MHz at its output 10 in the presence of a valid transponder return signal. The lower side injection frequency is also generated by stable temperature-compensated crystal oscillator 96, operated at about 50.24 MHz. This oscillator frequency is initially quadrupled in a frequency multiplier 98 and is thereafter passed through two triplicate frequency multiplicers 100 and 102 to a four-pole narrow bandpass filter 104 for supplying the lower s of the injection signal to the mixing stage 94.

The intermediate frequency output of the balanced mixing stage 94 20 is applied to the low noise amplifier 106 such that the total receiver noise number becomes 12 dB. The signal is then passed to a monolithic crystal band pass 108, preferably model 1619-1622 manufactured by Piezo Technology, Inc. under its registered trademark "COMLINE" where the reproduction of the amplitude 25 as a function of frequency is - 3 dB at 30 kHz. The crystal band pass filter 108 determines the pre-detection bandwidth and, together with the 12 dB noise number and modulation index of five, provides a total receiver sensitivity of -113 dBm for a 20 dB S + N / N ratio at the output of a crystal disc iminator 110, which will be described. in the following. The output of the crystal band pass filter 108 is passed through successive RF amplifier steps 112 and 114, each provided on a chip with AGC to provide the desired input level to the crystal descriptor 110. The output of each step 35 112 and 114 cause the respective AGC circuits to generate a direct current proportional to the amplitude of the output. The respective AGC levels from the individual steps 112 17

DK 161172 B

and 114 are summed to act as a total automatic gain detector 116, the output of which is a direct current proportional to the combined output amplitude of each step, which is an indication of the initial strength of the transponder signal from the bandpass filter 108. This combined AG two detector The output ignal is fed to a low-pass filter 118. The output charge of 1pass in the first 118 is output to a comparator circuit 120 for comparison with a predetermined threshold level established by sensitivity settings of a 10 potentiometer 122.

In the preferred embodiment of the system, the crystalline generator 110 consists of a monolithic crystal fan of the type 2378F from Piezo Technology, Inc. combined with an integrated RCA-15 circuit model CA 3089E to produce an extremely narrow band stable discriminator with a bandwidth of the order of 30 kHz. With a valid transponder return signal, the output of the discriminator 110 produces the modulating audio tone, which is at 2 kHz. The output of the discriminator 110 is held at ground potential by means of a fixation circuit 124 until a trigger output signal from the comparator circuit 120 indicates that the charge build-up signal in the adapter 118 exceeds the selected sensitivity setting 11 of the potentiometer 122. The system can thereby be set to sensitivity level causing short or weak return signals from remote transponders or other sources not to be detected.

Once the fixation circuit 124 is open, the 2 kHz audio tone 30 is passed through a low-pass filter 126 for decoding using phase-locked s 1 eye techniques using a quadrature detector 128 and a phase detector 130 to obtain a sustained tone within 10¾. of the modulation tone frequency. established as the free-running frequency of its voltage controlled oscillator 132.

The output of the phase detector 130 is applied to a loop filter 134 to generate a signal for adjusting the frequency and phase of the voltage controlled oscillator 132 with

Claims (11)

Although the system has been described in connection with a lined back embodiment using the specially described circuit elements and techniques with operating parameters relevant to an existing preferred audio tone frequency modulation embodiment, it is understood that 20 various modifications and variations of the circuit elements and techniques, without departing from the spirit of the invention. Eg. For example, the system may be modified to use amplitude modulation of one of the transmitted radio frequencies instead of frequency modulation, or use modu1 to operate in onstones outside the audio range without sacrificing the basic benefits of the unique unique system. Claims. 30 -------------------- An apparatus for detecting the presence of an object in a monitoring area comprising transmitting means (25, 30) for transmitting two radio frequency signals1 is at two different frequencies (fi, f2) in the monitoring area, the rad in the frequencies differing from one agent center frequency in each direction and with equal values, transponder means (34) which are detachably attached to protected objects and which can be moved by an object in the monitoring area, which transponder means (34) have antenna means (38) tuned to receive the radio frequency signals transmitted at both 5 frequencies, and a non-linear impedance element (36) coupled to the antenna means, the transponder means (34) radiating a return signal at the frequency equal to the sum of the frequencies of the two transmitted radio frequencies said system further comprising antenna means (22, 24) for receiving the return signal, narrow band reception means (42) for processing the received return signal and alarm means (24) capable of responding to an output signal produced by the signal processing of the received return signal by the narrow band receiving means, characterized in that it has multiple antenna transmission means (14, 16). for both radio frequency signals and arranged in opposite positions defining the monitoring region such that the ratio of the field strength of the two signals is substantially uniform over the monitoring region and the narrow band receiving signal processing is the received return signal. to exclude the transmitted rad in the frequency ignores and their harmonics to provide the output signal of the α1 arm means. Installation according to claim 1, characterized in that the two different frequencies (f ^, f2) differ from each other by at least 2% of the center-center frequency. System according to claim 1, characterized in that the modulated row of frequency in gna1 is frequency modulated by a fixed tone frequency. System according to claim 1, characterized in that each of the two radio frequency signals is produced by a temperature-compensated crystal controlled oscillator (48), frequency multiplication means (50, 54, 58) and narrow band filter means (52, 56). . DK 161172 B System according to claim 1, characterized in that the latter means contain signal source means (26, 30), antenna means (14, 16) spaced from the signal source means, linear amplifier means (28, 32) in the vicinity of the antenna means and past parts. means for delivering a signal from the signal source means to the linear amplifier means (28, 32). System according to claim 1, characterized in that the antenna of the transponder means is tuned to a frequency between the two different frequencies, the non-linear impedance element (36) being connected to the antenna means, thereby providing a tank circuit with a resonant frequency which is equal to the sum of the two different frequencies, for re-emitting a return signal at the resonant frequency. The system of claim 1, characterized in that the receiving means comprises a phase-locked loop (128, 130, 132, 134) for decoding the modulated radio frequency signal (fi). System according to claim 1, characterized in that the narrow-band receiver means includes receiver antenna means (22, 24) for collecting the return signal, filterogans (92, 108) for rejecting the signals captured by the antenna apart from the signals within a narrow one. pass bands at the frequency of the return signal, signal amplitude detection means (116) for generating a comparison level indicating the amplitude of the filtered return signal, and demodulation means (110, 124, 126) capable of responding to the comparison level by detecting the modulation only when the comparison level exceeds a preselected level setting 11 (122). System according to claim 8, characterized in that the signal amplifier in the nozzle-detecting means (116) contains a local oscillator, mixing means for emitting the intermediate frequency signal and a bandpass for the medium frequency signal. DK 161172 B System according to claim 1, characterized in that one of the radio frequency signals (f 1) is modulated by a fixed radio frequency tone to produce a narrow band frequency modulation, while the second radio frequency signal 5 is transmitted as an unmodulated signal at a fixed radio frequency (f 2). and that the receiver means includes a receiver antenna (22, 24), filter means (92) for rejecting signals received by the antenna, and which is outside a narrow passband at the resonant frequency, mixing means (94) 10 for generating an intermediate frequency for demodulation of passport band signals, amplifier means (112, 114) for amplifying the intermediate frequency signal and generating a comparison level indicating the amplitude of the intermediate frequency signal, narrow band discriminator means (110, 124, 126) capable of responding to the comparison level by demodulating only the intermediate frequency to indicate the low-frequency modulation when the amplitude of the sam a level 1 exceeds a preselected threshold, a phase-locked loop detector (128, 130, 132, 134) tuned to the frequency of the fixed audio tone to generate an alarm output signal upon detection of the fixed audio tone, and operation amplifier means , which is coupled to receive the alarm output signal for activating an alarm for a fixed period of time after initiating such an alarm output signal. 25 System according to claim 1, characterized in that the receiver means (42) are arranged to decode the return signal without the use of the reference signal1 derived from the transmitting means. 30 35