WO2024040742A1 - 一种多极少槽单元化永磁轮毂电机及协同控制系统和方法 - Google Patents

一种多极少槽单元化永磁轮毂电机及协同控制系统和方法 Download PDF

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WO2024040742A1
WO2024040742A1 PCT/CN2022/128230 CN2022128230W WO2024040742A1 WO 2024040742 A1 WO2024040742 A1 WO 2024040742A1 CN 2022128230 W CN2022128230 W CN 2022128230W WO 2024040742 A1 WO2024040742 A1 WO 2024040742A1
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motor
permanent magnet
torque
zone
area
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PCT/CN2022/128230
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English (en)
French (fr)
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朱孝勇
蒋敏
项子旋
王宝国
郑诗玥
全力
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江苏大学
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Publication of WO2024040742A1 publication Critical patent/WO2024040742A1/zh

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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02K—DYNAMO-ELECTRIC MACHINES
    • H02K1/00—Details of the magnetic circuit
    • H02K1/06—Details of the magnetic circuit characterised by the shape, form or construction
    • H02K1/12—Stationary parts of the magnetic circuit
    • H02K1/14—Stator cores with salient poles
    • H02K1/146—Stator cores with salient poles consisting of a generally annular yoke with salient poles
    • H02K1/148—Sectional cores
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02K—DYNAMO-ELECTRIC MACHINES
    • H02K21/00—Synchronous motors having permanent magnets; Synchronous generators having permanent magnets
    • H02K21/02—Details
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02K—DYNAMO-ELECTRIC MACHINES
    • H02K21/00—Synchronous motors having permanent magnets; Synchronous generators having permanent magnets
    • H02K21/12—Synchronous motors having permanent magnets; Synchronous generators having permanent magnets with stationary armatures and rotating magnets
    • H02K21/22—Synchronous motors having permanent magnets; Synchronous generators having permanent magnets with stationary armatures and rotating magnets with magnets rotating around the armatures, e.g. flywheel magnetos
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02K—DYNAMO-ELECTRIC MACHINES
    • H02K1/00—Details of the magnetic circuit
    • H02K1/06—Details of the magnetic circuit characterised by the shape, form or construction
    • H02K1/22—Rotating parts of the magnetic circuit
    • H02K1/27—Rotor cores with permanent magnets
    • H02K1/2786—Outer rotors
    • H02K1/2787—Outer rotors the magnetisation axis of the magnets being perpendicular to the rotor axis
    • H02K1/2789—Outer rotors the magnetisation axis of the magnets being perpendicular to the rotor axis the rotor consisting of two or more circumferentially positioned magnets
    • H02K1/279—Magnets embedded in the magnetic core
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02K—DYNAMO-ELECTRIC MACHINES
    • H02K15/00—Methods or apparatus specially adapted for manufacturing, assembling, maintaining or repairing of dynamo-electric machines
    • H02K15/0006—Disassembling, repairing or modifying dynamo-electric machines
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02K—DYNAMO-ELECTRIC MACHINES
    • H02K16/00—Machines with more than one rotor or stator
    • H02K16/04—Machines with one rotor and two stators
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02K—DYNAMO-ELECTRIC MACHINES
    • H02K21/00—Synchronous motors having permanent magnets; Synchronous generators having permanent magnets
    • H02K21/02—Details
    • H02K21/021—Means for mechanical adjustment of the excitation flux
    • H02K21/028—Means for mechanical adjustment of the excitation flux by modifying the magnetic circuit within the field or the armature, e.g. by using shunts, by adjusting the magnets position, by vectorial combination of field or armature sections
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02K—DYNAMO-ELECTRIC MACHINES
    • H02K21/00—Synchronous motors having permanent magnets; Synchronous generators having permanent magnets
    • H02K21/12—Synchronous motors having permanent magnets; Synchronous generators having permanent magnets with stationary armatures and rotating magnets
    • H02K21/22—Synchronous motors having permanent magnets; Synchronous generators having permanent magnets with stationary armatures and rotating magnets with magnets rotating around the armatures, e.g. flywheel magnetos
    • H02K21/222—Flywheel magnetos
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02K—DYNAMO-ELECTRIC MACHINES
    • H02K41/00—Propulsion systems in which a rigid body is moved along a path due to dynamo-electric interaction between the body and a magnetic field travelling along the path
    • H02K41/02—Linear motors; Sectional motors
    • H02K41/03—Synchronous motors; Motors moving step by step; Reluctance motors
    • H02K41/031—Synchronous motors; Motors moving step by step; Reluctance motors of the permanent magnet type
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P25/00—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details
    • H02P25/16—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details characterised by the circuit arrangement or by the kind of wiring
    • H02P25/18—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details characterised by the circuit arrangement or by the kind of wiring with arrangements for switching the windings, e.g. with mechanical switches or relays
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
    • H02P27/04—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
    • H02P27/06—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02K—DYNAMO-ELECTRIC MACHINES
    • H02K2213/00—Specific aspects, not otherwise provided for and not covered by codes H02K2201/00 - H02K2211/00
    • H02K2213/03—Machines characterised by numerical values, ranges, mathematical expressions or similar information
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02T—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO TRANSPORTATION
    • Y02T10/00—Road transport of goods or passengers
    • Y02T10/60—Other road transportation technologies with climate change mitigation effect
    • Y02T10/64—Electric machine technologies in electromobility

Definitions

  • the invention belongs to the field of permanent magnet motors, and specifically refers to a permanent magnet hub motor suitable for electric vehicles, ship propulsion, electric tractors, etc. that require high efficiency and multi-working condition operating characteristics.
  • Permanent magnet hub motors have shown potential application prospects in direct drive fields such as electric vehicles, ship propulsion, and electric tractors due to their advantages of high torque density and high power density.
  • the permanent magnet magnetic field is constant and difficult to adjust, and deep field weakening is difficult, resulting in this type of motor having a small speed regulation range and low high-speed efficiency, making it difficult to be suitable for multi-working conditions such as electric vehicles.
  • Chinese Patent No. 202210042232.8 proposes a thrust-type hybrid excitation motor.
  • the motor uses a hybrid excitation method of permanent magnet magnets and excitation current.
  • the adjustable excitation current reduces the difficulty of field weakening of the motor's excitation field.
  • the multi-working condition operation of the motor is realized; however, the introduction of excitation current increases the copper consumption of the motor and reduces the operating efficiency of the whole machine.
  • the Chinese patent number 201410768272.6 proposes a stator magnetization hybrid permanent magnet memory motor. This motor uses an excitation method in which low coercivity soft magnetic materials and rare earth permanent magnets work together.
  • the soft magnetic materials The magnetic potential of the motor changes with the change of the pulsating winding current, and the intensity of the excitation magnetic field also changes accordingly, thereby reducing the difficulty of field weakening of the motor and enabling the motor to operate in multiple working conditions; however, the addition of soft magnetic materials and pulsating windings increases the size of the motor. and weight, reducing the motor power density. At the same time, the introduction of pulse windings also increases motor losses and reduces motor operating efficiency.
  • the purpose of the present invention is to propose a multi-slot, few-pole unitized permanent magnet hub motor and a multi-unit collaborative control system and method for the motor in order to meet the needs of the permanent magnet hub motor.
  • the in-wheel motor improves operating efficiency and expands the high-efficiency operating area under the premise of multi-working operating requirements.
  • a unitized permanent magnet hub motor with multiple poles and few slots is that it is composed of N identical motor units evenly distributed along the circumferential direction of the radial cross section, and each motor unit includes an outer rotor of 1/N , 1/N inner stator and 1/N concentrated winding.
  • the inner stator is coaxially sleeved inside the outer rotor.
  • the inner stator is wound with concentrated windings.
  • the concentrated windings in each motor unit are three-phase symmetrical and The distribution is the same; the outer rotor has a rotor core, and 2a permanent magnet steel groups are evenly distributed on the rotor core along the circumferential direction.
  • Each permanent magnet steel group is composed of the first and second rectangular permanent magnet steel and arc
  • the first and second rectangular permanent magnets have the same structure and are rectangular in radial cross-section.
  • the inner and outer diagonals are in the length direction of the rectangle. They are arranged in a V-shape with the opening facing the air gap side on the arc-shaped permanent magnet.
  • the outside of the magnet is symmetrical with respect to the center line of the arc-shaped permanent magnet along the diameter direction; the magnetizing direction of the first and second rectangular permanent magnets is perpendicular to its own length direction, and the magnetizing direction and center of the arc-shaped permanent magnet are The line directions are consistent.
  • the first and second rectangular permanent magnets and the arc-shaped permanent magnets belonging to the same permanent magnet group are magnetized in directions pointing toward or away from the air gap at the same time.
  • Each of the 1/N outer rotors is divided into the same M rotor segments along the axial direction.
  • the M rotor segments are arranged in a mechanical offset angle sequentially rotated along the same rotation direction, 20mm ⁇ l ef /M ⁇ 120mm , l ef is the axial length of the motor.
  • each arc-shaped permanent magnet is surrounded by an outer long side of the arc-shaped permanent magnet, an inner long side of the arc-shaped permanent magnet and two short sides of the arc-shaped permanent magnet in the radial cross-section.
  • the arc center of the outer long side of the arc-shaped permanent magnet and the inner long side of the arc-shaped permanent magnet are the same as the center of the outer rotor.
  • the short side of the arc-shaped permanent magnet is consistent with the diameter direction of the outer rotor.
  • each of the first and second rectangular permanent magnets is provided with an inner magnetic barrier at the end close to the air gap side, and is provided with an outer magnetic barrier at the end far from the air gap side.
  • Each arc-shaped permanent magnet is provided with an inner magnetic barrier.
  • Each of the tangential ends is provided with a virtual groove that becomes part of the air gap.
  • the technical solution adopted by the collaborative control system of the unitized permanent magnet hub motor with multiple poles and few slots according to the present invention is: it includes a battery, two control modules and N winding electronic switches, and one winding electronic switch controls one motor unit.
  • each control module is composed of a power electronic switch, a DSP controller and an inverter connected in series.
  • the input terminals of the two power electronic switches are connected to the output terminals of the battery respectively.
  • Each inverter The output end of the converter is connected to N/2 winding electronic switches respectively, and the output end of the centralized winding is connected to the battery through the rectifier.
  • the constant torque area is divided into the first area and the second area.
  • T p When the torque at the operating point T p ⁇ T b , it is the first area.
  • T p > T b it is the second area.
  • T b is the critical area.
  • Torque T b is N-2 times the peak torque T x of a motor unit (1); the maximum value of the operating point torque in the second zone does not exceed the torque corresponding to the external characteristic curve g;
  • both power electronic switches are closed, and at least N/2 motor units connected to each control module are closed.
  • Two winding electronic switches, T p1 is the torque at the operating point in the first zone; the two inverters output currents with the same amplitude and different phases; when the operating point is in the second zone, two power electronic switches and N winding electronic switches All closed.
  • the motor operating point speed n p > n b it is a constant power zone; the constant power zone is divided into the third to eighth zones.
  • ⁇ b is the boundary efficiency of the motor when a single control module is running; the area surrounded by the abscissa axis, the external characteristic curve g and a straight line j perpendicular to the abscissa passing through the highest speed point E in the fourth zone is the eighth zone; the torque is less than The area where the torque at the lowest speed point D in the fourth zone and the speed is smaller than the speed at point D is the area S 31 , which consists of the abscissa, a straight line k passing through point D perpendicular to the abscissa, the fourth zone point D, and the lower half of E
  • the area enclosed by the peripheral boundary line and straight line j is area S 32 , and the union of S 31 and S 32
  • the area S 51 is the area S 51 , and the area surrounded by the straight line k, the straight line j, the upper half-peripheral boundaries of the fourth area points D and E, and the lower half-peripheral boundaries of the sixth area points F and G is the area S 52 , and the union of S 51 and S 52
  • the set is the fifth area; the remaining area in the constant power area is the seventh area;
  • the power-increasing operation in the third zone is: the transition point P 3 ' (n p3 ,T p3 ' ) with the same speed as the third zone operating point P 3 (n p3 ,T p3 ).
  • a set S P3 is formed when The three-phase current output by the inverter at the power operating point H 3 (n H3 , T H3 ) feeds excess energy back to the battery; the power-increasing operation in the fifth area is the same as the power-increasing operation in the third area; n p3 , T p3 and ⁇ p3 are respectively the speed, torque and efficiency of the operating point P 3 (n p3 , T p3 ), T p3' and ⁇ p3' are respectively the torque and efficiency of the transition point, and ⁇ pg is the power generation efficiency.
  • the seventh zone there are two transition points that are equal to the rotation speed of the seventh zone operating point.
  • the sum of the torques of the two transition points is equal to the torque of the seventh zone operating point.
  • the system of the two transition points is calculated.
  • the sum of the total power consumed by the system with two operating points is equal to the minimum value of the sum of the total power consumed by the system at the two transition points, then the two operating points are determined by the corresponding Two control modules control the operation.
  • the present invention adopts the concept of multi-unit design for unitized motor structural design.
  • the spatially non-overlapping unit design makes all motor units independent of each other.
  • Each motor unit can operate and control independently, which greatly improves the operation of the motor.
  • With the degree of freedom of control the different combinations and working modes of its multiple units make the permanent magnet hub motor capable of operating in multiple working conditions.
  • the present invention adopts a structure with more poles and less slots, which relieves the traditional unitized motor's high requirements for pole-slot ratio, provides a new pole-slot ratio for the unitized motor, increases the number of rotor pole pairs, and ensures permanent
  • the magnetic hub motor has the performance characteristics of low speed and high torque, which better meets the performance requirements of the hub motor.
  • the rotor in the present invention adopts a segmented oblique pole structure in the axial direction, which is conducive to adjusting the harmonic phase distribution of the rotor torque and realizing the harmonic phase of the first rotor torque harmonic phase and the second rotor torque harmonic phase. Compensation eliminates the highest amplitude harmonic of the output torque, thereby greatly reducing the motor torque pulsation and improving the torque quality.
  • the long side of the arc-shaped permanent magnet steel in the present invention close to the air gap adopts a sinusoidal design.
  • it is beneficial to adjust the salient pole rate of the motor and improve the field weakening capability of the motor; on the other hand, it improves the sine degree of the permanent magnet magnetic potential.
  • reducing the harmonic complexity of the permanent magnet magnetic field is conducive to reducing the motor core loss, improving the motor operating efficiency while reducing the motor torque ripple, and improving the overall quality of the motor output performance.
  • the rectangular permanent magnet in the present invention is provided with a magnetic leakage circuit near the air gap end, so that as the q-axis current changes, the permanent magnet magnetic field of the motor can achieve "multiple flux leakage at light load and no flux leakage at heavy load" , that is, when the motor is heavily loaded, the permanent magnet flux is effective flux, which is beneficial to improving the motor's torque output capability and enabling the motor to operate under heavy load conditions; when the motor is at high speed, the q-axis current is reduced and the permanent magnet magnetic field is weakened, which is beneficial to Broaden the motor speed range and realize high-speed motor operation.
  • the mutually independent winding electronic switches realize independent control of the motor unit, which not only improves the freedom of motor control, but also improves the motor fault tolerance; the mutually independent power electronic switches decouple the two motors.
  • the control module improves the freedom of motor control, provides a hardware foundation for high-performance motor operation, and improves the operational reliability of the motor in the light load area.
  • the present invention proposes that when the motor works in the first low-speed load zone, the two inverters output currents with the same frequency, same amplitude and different phases, and the motor units of the two modules work at the same operating point. At this time, the two control The main harmonic phase difference of the corresponding output torque of the motor is 180°, and the two compensate each other, thereby reducing the motor torque ripple.
  • the present invention proposes that when the motor operates in the third high-speed and low-load zone, only one of the first power electronic switch and the second power electronic switch is closed at the same time, and all winding electronic switches are closed, corresponding to the controlled motor unit. Power operation, at this time, the operating efficiency of the motor is improved, and the excess energy is fed back to the battery through the rectifier, which is beneficial to improving the efficiency of the motor system.
  • the present invention proposes that when the motor operates in the fifth high-speed load zone, all power electronic switches are closed and all winding electronic switches are closed, and the motor unit operates with increased power. At this time, the operating efficiency of the motor is improved, and the excess energy passes through the rectifier. Feedback to the battery helps improve motor and control system efficiency.
  • the present invention proposes that when the motor operates in the seventh high-speed overload zone, all power electronic switches are closed and all winding electronic switches are closed.
  • the unit motors of the two modules operate at different operating points, and the output torques are algebraically superimposed. , the losses are also algebraically superimposed.
  • the total motor losses are reduced, thus improving the motor operating efficiency.
  • Figure 1 is a schematic diagram of the radial structure and unitized decomposition of the unitized permanent magnet hub motor with multi-pole and few slots according to the present invention
  • FIG 2 is an enlarged structural view of a motor unit 1 in Figure 1;
  • Figure 3 is a schematic diagram of the two-section rotor axial installation structure of the multi-pole and few-slot unitized permanent magnet hub motor of the present invention
  • Figure 4 is a radial view of Figure 3;
  • Figure 5 is a flow chart for calculating the mechanical misalignment angle ⁇ in Figure 4.
  • Figure 6 is the output torque diagram of the two-stage rotor in Figure 3.
  • Figure 7 is a composite diagram of the rotor torque of the two sections in Figure 6;
  • Figure 8 is an enlarged view of the rotor structure and a labeled view of the permanent magnet magnetization method in Figure 2;
  • Figure 9 is an enlarged view of the structure and geometric dimensions of a permanent magnet steel group in Figure 8.
  • Figure 10 is an enlarged view of the structure of the arc-shaped permanent magnet steel in Figure 9 and its geometric dimensions;
  • Figure 11 is an enlarged view of the structure of the first rectangular permanent magnet, the inner magnetic barrier and the outer magnetic barrier in Figure 8 and its geometric dimensions;
  • Figure 12 is an enlarged view of the structure and geometric dimensions of the three permanent magnet steel groups in Figure 8;
  • Figure 13 is an enlarged schematic diagram of the formation structure of the rotor virtual slot in Figure 12;
  • Figure 14 is an enlarged view of the stator structure in Figure 2;
  • Figure 15 is a schematic diagram of the working magnetic circuit of the unitized permanent magnet hub motor with multiple poles and few slots under light load according to the present invention.
  • Figure 16 is a schematic diagram of the working magnetic circuit of the unitized permanent magnet hub motor with multiple poles and few slots under heavy load according to the present invention.
  • Figure 17 is a structural block diagram of the multi-unit collaborative control system of the multi-pole and few-slot unitized permanent magnet hub motor of the present invention.
  • Figure 18 is a graph showing the constant torque area and constant power area divided according to the critical speed when the multi-unit collaborative control system is working;
  • Figure 19 is a distribution diagram of two sub-regions divided into the constant torque area in Figure 18;
  • Figure 20 is a distribution diagram of eight sub-areas divided into the constant power area in Figure 18.
  • FIG. 1 In Figure 1: 1. Motor unit; 2. Outer rotor; 3. Inner stator; 4. Centralized winding; 5. Rotating shaft; 2.1. First rotor section; 2.2. Second rotor section; 2.3. Rotor core; 2.4 .Permanent magnet steel group; 2.5. Inner magnetic barrier; 2.6. External magnetic barrier; 2.7. Virtual slot; 2.4.1. First rectangular permanent magnet steel; 2.4.2. Second rectangular permanent magnet steel; 2.4. 3. Arc-shaped permanent magnet steel; 2.4.3.1. The outer long side of the arc-shaped permanent magnet steel; 2.4.3.2. The inner long side of the arc-shaped permanent magnet steel; 2.4.3.3. The short side of the arc-shaped permanent magnet steel; 2.5 .1. The first side of the inner magnetic barrier; 2.5.2.
  • the multi-pole and few-slot unitized permanent magnet hub motor of the present invention is composed of N identical motor units 1 uniformly distributed along the circumferential direction of the radial cross section.
  • the two control modules are controlled independently.
  • the corresponding central angle ⁇ N of each motor unit 1 on the radial cross section is 2 ⁇ /N.
  • Each motor unit 1 is composed of an outer rotor 2 of 1/N in the circumferential direction of the radial cross section, an inner stator 3 of 1/N, a concentrated winding 4 of 1/N and a rotating shaft 5 of 1/N. Therefore, N
  • the permanent magnet hub motor composed of motor unit 1 consists of an outer rotor 2, an inner stator 3, a set of concentrated windings 4 and a rotating shaft 5.
  • the inner stator 3 is coaxial with the outer rotor 2 and is sheathed inside the outer rotor 2.
  • the center of the inner stator 3 is used to place the rotating shaft 5.
  • the inner stator 3 is wound with a concentrated winding 4.
  • the thickness of the air gap is related to the power level of the motor, the selected permanent magnet material, and the processing and assembly process of the outer rotor 2 and the inner stator 3.
  • the thickness of the outer rotor 2 and the inner stator 3 are related to each other.
  • Sub-3 is made of laminated silicon steel sheets with a thickness of 0.35mm, and the lamination coefficient is 0.95.
  • the 1/N outer rotor 2 of each motor unit 1 is divided into identical M sections along the axial direction, forming M rotor sections, which are the first rotor section and the second rotor section in sequence. ,..., Mth rotor section.
  • M rotor sections which are the first rotor section and the second rotor section in sequence. ,..., Mth rotor section.
  • the number of M should satisfy: 20mm ⁇ l ef /M ⁇ 120mm, where l ef is the axial length of the motor.
  • M rotor segments are arranged and installed sequentially rotated by a mechanical misalignment angle ⁇ along the same rotation direction, and two adjacent rotor segments differ by a mechanical misalignment angle ⁇ .
  • the mechanical misalignment angle ⁇ is determined as follows, as shown in Figure 5:
  • Step 1 Assign the initial mechanical dislocation angle ⁇ 0 to 0.
  • Step 2 Use finite element software to simulate and obtain the output torque T(t) waveform of a motor when the mechanical misalignment angle is ⁇ 0 , and calculate the motor torque ripple when the mechanical misalignment angle is ⁇ 0.
  • the motor torque ripple is the initial Torque ripple T rip0 .
  • the calculation method of the initial torque ripple T rip0 is: first calculate the average value of the output torque T(t), and make a difference between the maximum value and the minimum value of the output torque T(t). The difference accounts for the output torque T The percentage of the average value of (t) is the initial torque ripple T rip0 .
  • Step 3 Perform fast Fourier decomposition on the motor output torque T(t) waveform to obtain the harmonic order k of the highest amplitude harmonic component.
  • the motor output torque T(t) waveform is decomposed by Fast Fourier into the DC component T 0 , the highest amplitude harmonic component T k cos (kwt+ ⁇ k ) and the remaining harmonic components Sum.
  • T k is the amplitude of the highest amplitude harmonic component
  • ⁇ k is the phase
  • s is the order of the remaining harmonic component
  • its amplitude is T s
  • the phase is ⁇ s . Therefore, the fast Fourier decomposition expression of the motor output torque T(t) is:
  • t is time
  • w is the rotation speed of motor output torque T(t).
  • Step 4 Based on the harmonic order k of the highest amplitude harmonic component obtained in step 3, calculate the transition mechanical dislocation angle according to the following formula Among them, P r is the number of motor rotor pole pairs, and M is the number of rotor segments.
  • Step 5 Use finite element software to simulate the output torque waveform of the motor when the mechanical misalignment angle is the transition mechanical misalignment angle ⁇ 1 , and calculate the motor torque ripple at this time to obtain the transition torque ripple T rip1 .
  • the calculation method of the transition torque ripple T rip1 is the same as the calculation method of the initial torque ripple T rip0 in step 2.
  • Step 6 Compare the transition torque ripple T rip1 with the initial torque ripple T rip0 in step 2 to determine whether the transition mechanical misalignment angle ⁇ 1 can effectively reduce the torque ripple. If the transition torque ripple T rip1 is smaller than the initial torque ripple T rip0 , it is judged that the transition mechanical misalignment angle ⁇ 1 can effectively reduce the torque ripple, and then step 9 is performed; otherwise, if the transition torque ripple T rip1 is greater than or equal to the initial torque ripple If the torque ripple is T rip0 , it is judged that the transition mechanical misalignment angle ⁇ 1 cannot effectively reduce the torque ripple, and then step 7 is performed.
  • Step 7 Assign the transition mechanical misalignment angle ⁇ 1 to the initial mechanical misalignment angle ⁇ 0 .
  • Step 8 Based on the assigned initial mechanical misalignment angle ⁇ 0 in step 7, use finite element software to simulate and obtain the motor output torque T(t) waveform when the mechanical misalignment angle is the assigned initial mechanical misalignment angle ⁇ 0 ; then Loop through steps 3 to 6 until the transition torque ripple Trip1 is smaller than the initial torque ripple Trip0 in step 6, and the judgment result is that the torque ripple can be effectively reduced and step 9 is executed.
  • Step 9 Assign the transition mechanical misalignment ⁇ 1 to the mechanical misalignment angle ⁇ and output it.
  • the output torque T 2.1 (t) generated by the first rotor section 2.1 is a DC component. Highest amplitude harmonic component and the remaining harmonic components The sum of , the expression is:
  • the phases ⁇ k and ⁇ s of the second rotor section 2.2 change relative to the first rotor section 2.1.
  • the output torque T 2.2 (t) generated by the second rotor section 2.2 is a DC component. Highest amplitude harmonic component and the remaining harmonic components The sum of , the expression is:
  • the j-th rotor segment 2.j of the M rotor segments, j is a positive integer and j ⁇ M
  • the output torque T generated by the j-th rotor segment 2.j 2.j (t) is the DC component Highest amplitude harmonic component and the remaining harmonic components
  • the phases of the k-order harmonics in their output torque waveforms differ by That is to say, when the output torque waveforms of all rotor segments are synthesized, the k-order harmonics will form compensation, and the amplitude of the synthesized harmonics will be 0. That is, the output torque T'(t) of the motor's magnetic potential is the DC component. and the remaining harmonic components The sum of , the expression is:
  • the cogging torques of the first rotor section 2.1 and the second rotor section 2.2 oscillate evenly between [-6.4, 6.1], and the cogging torque of the first rotor section 2.1
  • the cogging torque of the second rotor section 2.2 takes a minimum value, and vice versa. It can be seen that the output torque of the first rotor section 2.1 and the output torque of the second rotor section 2.2 achieve peak and valley compensation, which is beneficial to eliminating the highest amplitude harmonic component, reducing motor torque pulsation, and improving motor torque. quality.
  • the abscissa is the rotor position in electrical angle; the ordinate is the cogging torque in Nm.
  • the solid curve in Figure 7 is the actual rotor torque curve after the output torque of the first rotor section 2.1 and the output torque of the second rotor section 2.2 are combined.
  • the rotor position changes in the electrical angle 0-360°, the rotor torque oscillates uniformly between [-1.1,1].
  • the peak value of the torque curve in Figure 7 is greatly reduced. Therefore, the mechanical misalignment angle ⁇ used in the present invention can significantly reduce the torque ripple of the motor and improve the torque quality.
  • the outer rotor 2 of the permanent magnet hub motor is composed of a rotor core 2.3, 2a permanent magnet steel groups 2.4, 4a inner magnetic barriers 2.5, 4a outer magnetic barriers 2.6 and 4a virtual slots 2.7, where a is a positive integer.
  • 2a permanent magnet steel groups 2.4 are evenly distributed along the circumferential direction on the entire rotor core 2.3.
  • each permanent magnet group 2.4 is composed of a first rectangular permanent magnet 2.4.1, a second rectangular permanent magnet 2.4.2 and an arc-shaped permanent magnet 2.4.3.
  • 2a arc-shaped permanent magnets 2.4.3 are embedded inside the inner surface of the outer rotor 2.
  • the 2a arc-shaped permanent magnets 2.4.3 are evenly arranged along the circumferential direction.
  • Each arc-shaped permanent magnet 2.4.3 The center line on the radial section is consistent with the diameter direction, and the center line of the permanent magnet steel group 2.4 coincides with the center line of the arc-shaped permanent magnet steel 2.4.3.
  • the first rectangular permanent magnet steel 2.4.1 and the second rectangular permanent magnet steel 2.4.2 have the same structure. Their radial cross-sections are both rectangular.
  • the first rectangular permanent magnet steel 2.4.1 and the second rectangular permanent magnet steel 2.4.2 in the magnetic steel group 2.4 are placed symmetrically with respect to the center line of the permanent magnet group 2.4.
  • the first rectangular permanent magnet steel 2.4. 1 and the second rectangular permanent magnet 2.4.2 are also placed symmetrically with respect to the center of the arc-shaped permanent magnet 2.4.3.
  • the first rectangular permanent magnet 2.4.1 and the second rectangular permanent magnet 2.4.2 belonging to the same permanent magnet group 2.4 are arranged in a "V" shape with the opening facing the air gap side, and the inner and outer oblique directions are The length of the rectangle.
  • the arc-shaped permanent magnet 2.4.3 is placed in the middle of the V-shaped opening formed by the first rectangular permanent magnet 2.4.1 and the second rectangular permanent magnet 2.4.2.
  • the three permanent magnets do not contact each other. .
  • the magnetizing direction of the first rectangular permanent magnet 2.4.1 and the second rectangular permanent magnet 2.4.2 belonging to the same permanent magnet group 2.4 is perpendicular to the long side of the magnet itself, and The direction of its own width is consistent, and the magnetizing direction of the arc-shaped permanent magnet 2.4.3 is consistent with the direction of the center line, pointing toward or away from the center of the circle.
  • the magnetizing directions of the first rectangular permanent magnet 2.4.1, the second rectangular permanent magnet 2.4.2 and the arc-shaped permanent magnet 2.4.3 belonging to the same permanent magnet group 2.4 are directed toward or away from the air gap at the same time.
  • the magnetizing directions of the two adjacent permanent magnet steel groups 2.4 are opposite.
  • the first rectangular permanent magnet steel 2.4.1 and the second rectangular permanent magnet steel 2.4.2 are respectively provided with a magnetic barrier at their ends close to the air gap side and far away from the air gap side, that is, the inner end of the rectangular permanent magnet steel. and the outer end are each provided with a magnetic barrier. They are: each of the first rectangular permanent magnet steel 2.4.1 and the second rectangular permanent magnet steel 2.4.2 has an inner magnetic barrier 2.5 at its inner end. Therefore, there are a total of 4a inner magnetic barriers 2.5; each The outer ends of the first rectangular permanent magnet 2.4.1 and the second rectangular permanent magnet 2.4.2 are each provided with an external magnetic barrier 2.6. Therefore, there are 4a external magnetic barriers 2.6 in total.
  • a virtual slot 2.7 is provided at both ends of each arc-shaped permanent magnet 2.4 along the tangential direction. Therefore, there are 4a virtual slots 2.7 in total, and the virtual slots 2.7 are connected to the air gap. Become one with the air gap and become part of the air gap.
  • the V-shape formed between the first rectangular permanent magnet 2.4.1 and the second rectangular permanent magnet 2.4.2 The angle ⁇ pm should satisfy: 40° ⁇ pm ⁇ 65°.
  • the first rectangular permanent magnet steel 2.4.1 and the second rectangular permanent magnet steel 2.4.2 in order to strengthen the strength of the first rectangular permanent magnet steel 2.4.1 and the second rectangular permanent magnet steel 2.4.2 as well as the processing difficulty and stress distribution, the first rectangular permanent magnet steel 2.4.1 and the second rectangular permanent magnet steel 2.4.
  • the long side length w pm and the wide side length h pm of magnetic steel 2.4.2 should satisfy: 2 ⁇ w pm /h pm ⁇ 4.
  • each arc-shaped permanent magnet 2.4.3 consists of an arc-shaped permanent magnet outer long side 2.4.3.1, an arc-shaped permanent magnet inner long side 2.4.3.2 and two arc-shaped permanent magnets.
  • Magnetic steel 2.4.3 is surrounded by the short side 2.4.3.3 of arc-shaped permanent magnetic steel that is symmetrical to the center.
  • the arc center of the outer long side 2.4.3.1 of the arc-shaped permanent magnet is the same as the center of the outer rotor 2.
  • the central angle corresponding to the inner long side 2.4.3.2 of the arc-shaped permanent magnet is the same as the central angle corresponding to the outer long side 2.4.3.1 of the arc-shaped permanent magnet.
  • the short side 2.4.3.3 of the arc-shaped permanent magnet is consistent with the diameter direction of the outer rotor 2, and is on a radius of the outer rotor 2.
  • the inner long side 2.4.3.2 of the arc-shaped permanent magnet steel is a half-cycle sinusoidal curve.
  • the independent variable ⁇ 1 of the sinusoidal curve ranges from [ ⁇ , 2 ⁇ ], which is the sine of the inner long side 2.4.3.2 of the arc-shaped permanent magnet steel.
  • the curve function is:
  • f 1max is the amplitude of the sinusoidal curve, which is determined by the specific performance requirements of the motor.
  • ⁇ 1 is 3 ⁇ /2
  • the corresponding point f 1 (3 ⁇ /2) on the sinusoidal curve is exactly located on the inner surface of the outer rotor 2. Therefore, the arc-shaped permanent magnet 2.4.3 is embedded in the outer rotor 2 as a whole. Inside this side of the inner surface, point f 1 (3 ⁇ /2) when ⁇ 1 is 3 ⁇ /2 coincides with the inner surface of the outer rotor 2 .
  • the design of the sinusoidal shape of the inner long side 2.4.3.2 of the arc-shaped permanent magnet changes the magnetic potential waveform of the arc-shaped permanent magnet 2.4.3.
  • the magnetic potential waveform of the arc-shaped permanent magnet 2.4.3 changes from the original square wave to the superposed waveform of rectangular wave and sine, which changes the magnetic potential harmonic distribution of the arc-shaped permanent magnet 2.4.3 and improves the arc shape.
  • the sinusoidal nature of the magnetic potential waveform of the permanent magnet 2.4.3 increases the amplitude of the fundamental wave of the magnetic potential, which is beneficial to the improvement of the motor's torque output capability.
  • this sinusoidal shape design also reduces the fundamental wave amplitude of the permanent magnet magnetic potential, which is beneficial to reducing the loss of the motor core.
  • the minimum width h pmin of the arc-shaped permanent magnet steel in the radial direction should satisfy: 6 ⁇ h pm /h pmin ⁇ 8.
  • the minimum width h pmin of the arc-shaped permanent magnet steel 2.4.3 is consistent with the arc-shaped permanent magnet.
  • the maximum width h pmax of magnetic steel 2.4.3 should satisfy: 1.5 ⁇ h pmax /h pmin ⁇ 2.
  • the inner magnetic barriers 2.5 respectively provided at the ends of the first rectangular permanent magnet 2.4.1 and the second rectangular permanent magnet 2.4.2 close to the air gap have the same structure and are arranged along the permanent magnet group.
  • the center line of 2.4 is symmetrically distributed.
  • the radial cross section of the inner magnetic barrier 2.5 is a "pentagon". It is surrounded by the second side of the inner magnetic barrier 2.5.2, the third side of the inner magnetic barrier 2.5.3, the fourth side of the inner magnetic barrier 2.5.4 and the fifth side of the inner magnetic barrier 2.5.5.
  • the first side 2.5.1 of the inner magnetic barrier is the extended side of the long side of the first rectangular permanent magnet 2.4.1 close to the air gap side
  • the second side 2.5.2 of the inner magnetic barrier is a circle coaxial with the outer rotor 2 of the motor. Curved edge.
  • the second side of the inner magnetic barrier 2.5 In order to construct a permanent magnet magnetic field leakage magnetic circuit, widen the motor speed regulation range and improve the motor's operating efficiency in the high-speed area, while controlling the motor's magnetic leakage degree, and improve the motor's torque output capability in the low-speed and light-load area, the second side of the inner magnetic barrier 2.5.
  • the distance h b1 between 2 and the inner surface of the outer rotor 2 should satisfy: 0.75 ⁇ h b1 /h pm ⁇ 0.9.
  • the third side 2.5.3 of the inner magnetic barrier is located on the radius of the outer rotor 2.
  • the fourth side 2.5.4 of the inner magnetic barrier is parallel to the first side 2.5.1 of the inner magnetic barrier, and is located outside the first side 2.5.1 of the inner magnetic barrier, that is, the side away from the air gap.
  • the first side 2.5.1 of the inner magnetic barrier is connected to the inner magnetic barrier.
  • the distance h b2 between the fourth side 2.5.4 is less than the width of the first rectangular permanent magnet 2.4.1 and the second rectangular permanent magnet 2.4.2, and should satisfy: 0.8h pm ⁇ h b2 ⁇ 0.9h pm .
  • the fifth side 2.5.5 of the inner magnetic barrier coincides with the short sides of the first rectangular permanent magnet 2.4.1 and the second rectangular permanent magnet 2.4.2 close to the air gap side.
  • the ends of the first rectangular permanent magnet 2.4.1 and the second rectangular permanent magnet 2.4.2 away from the air gap are respectively provided with external magnetic barriers 2.6.
  • the two external magnetic barriers 2.6 have the same structure and are located along the permanent magnet group 2.4. symmetrically distributed along the center line.
  • the radial cross section of the external magnetic barrier 2.6 is also a "pentagon", consisting of the first side 2.6.1 of the external magnetic barrier, the external magnetic barrier 2.6. It is surrounded by the second side of the outer magnetic barrier 2.6.2, the third side of the outer magnetic barrier 2.6.3, the fourth side of the outer magnetic barrier 2.6.4 and the fifth side of the outer magnetic barrier 2.5.5.
  • the first side 2.6.1 of the external magnetic barrier is located on the extension line of the long side of the first rectangular permanent magnet 2.4.1 close to the air gap side, and the second side 2.6.2 of the external magnetic barrier is located on the radius of the outer rotor 2 .
  • the third side 2.6.3 of the outer magnetic barrier is an arc side coaxial with the outer rotor 2. In order to reduce the uncontrollable flux leakage of the first rectangular permanent magnet steel 2.4.1 and the second rectangular permanent magnet steel 2.4.2 and at the same time reduce the q-axis magnetic resistance and maximize the torque, the third side of the external magnetic barrier 2.6.
  • the distance h b3 between 3 and the outer surface of the outer rotor 2 should satisfy: 0.2(R ro -R ri ) ⁇ h b3 ⁇ 0.35 (R ro -R ri ), where R ro is the outer diameter of the outer rotor 2, R ri is the inner diameter of the outer rotor 2.
  • the fourth side 2.6.4 of the external magnetic barrier is parallel to the first side 2.6.1 of the external magnetic barrier, and is located outside the first side 2.6.1 of the external magnetic barrier, that is, the side away from the air gap.
  • the fifth side 2.5.5 of the external magnetic barrier coincides with the short sides of the first rectangular permanent magnet 2.4.1 and the second rectangular permanent magnet 2.4.2 away from the air gap.
  • the two ends of the arc-shaped permanent magnet 2.4.3 are symmetrically arranged along the center line of the arc-shaped permanent magnet 2.4.3.
  • Each virtual slot 2.7 is a sinusoidal section.
  • point A the intersection point between the virtual slot 2.7 and the inner long side 2.4.3.2 of the arc-shaped permanent magnet steel 2.4.3 is point A. It is also the intersection point of the inner long side 2.4.3.2 of the arc-shaped permanent magnet steel and the short side 2.4.3.3 of the arc-shaped permanent magnet steel.
  • the intersection point of the second side 2.5.2 of the inner magnetic barrier and the third side 2.5.3 of the inner magnetic barrier is point B, and the intersection point of a radius passing through point B and the inner surface of the outer rotor 2 is point C.
  • point A to point C is a sinusoidal curve, forming a virtual slot 2.7.
  • the sinusoidal curve of virtual slot 2.7 has an independent variable ⁇ 2 in the range of [ ⁇ /2, ⁇ ].
  • the sinusoidal function of virtual slot 2.7 is:
  • f 2max is the function amplitude, which is determined by the specific performance requirements of the motor.
  • ⁇ 2 is ⁇ /2
  • point f 2 ( ⁇ /2) on the sinusoid coincides with point A.
  • point f 2 ( ⁇ ) on the sinusoid coincides with point C.
  • the virtual slot 2.7 can change the magnetic permeability waveform of the outer rotor 2 and the air gap magnetic potential waveform of the permanent magnet magnetic field.
  • the magnetic permeability changes from the original square wave to the superposition waveform of square wave and sinusoidal, and at the same time, the permanent magnetic steel group 2.4 is realized.
  • the polarization processing of the magnetic potential waveform improves the sinusoidality of the magnetic potential waveform of the arc-shaped permanent magnet 2.4.3, which increases the amplitude of the fundamental magnetic potential wave, which is beneficial to the improvement of the motor's torque output capability.
  • the virtual slot The design of 2.7 reduces the amplitude of the fundamental wave of the permanent magnet magnetic potential, which is beneficial to reducing the loss of the motor core.
  • the outer end of each stator salient pole 3.2 extends a stator pole piece 3.3 to both tangential sides.
  • the stator salient poles 3.2 are rectangular in radial cross-section and evenly distributed along the circumferential direction of the outer surface of the magnetic ring 3.1.
  • identical stator pole shoes 3.3 are respectively provided on both tangential sides of the outer end of the stator salient pole 3.2.
  • the radial width of the magnetic ring 3.1, the radial length and tangential width of the stator salient pole 3.2, and the tangential width of the stator pole piece 3.3 are determined by the motor power.
  • the number of rotor pole pairs P r , the number of stator slots N s , the number of motor phases m, and the number N of motor units 1 of the unitized permanent magnet hub motor with multiple poles and few slots according to the present invention should meet the following conditions: (1) More poles and fewer slots: The number of rotor pole pairs P r is greater than the number of stator slots N s ; the slot pitch angle ⁇ of the motor is the product of the quotient of the number of rotor pole pairs P r divided by the number of stator slots N s and 2 ⁇ .
  • Rotor unitization The unitization motor requires that the rotor can be divided into N parts, that is, the number of rotor pole pairs P r should be an integer multiple of the number N of motor units 1; (3) Stator unitization: It requires that the stator can be divided into N parts parts, then the number of stator slots N s should be an integer multiple of the number of phases m, and at the same time an integer multiple of the number N of motor units 1; (4) Control module unitization: In order to improve the degree of freedom of unitized motor control, improve the overall motor In terms of operating efficiency and operating performance, the multi-pole and few-slot unitized permanent magnet hub motor of the present invention is equipped with two control modules.
  • the number N of motor units 1 is required to be an integer multiple of 2;
  • Concentrated winding The winding is concentrated winding 4, that is, the winding span should be 1;
  • Winding unitization On the basis of stator unitization , it needs to be satisfied that the windings in each motor unit 1 are three-phase symmetrical and have the same distribution, that is, there are positive integers i, a, b, c, d, e so that the following equations are simultaneously established:
  • the condition for its light load operation is defined as the motor output torque T satisfying: T ⁇ 0.6T rated , where T rated is the rated output torque of the motor; heavy load
  • the operating conditions are that the motor output torque T satisfies: T>0.6T rated .
  • the condition for low-speed operation is defined as the motor speed n satisfying the condition: n ⁇ n rated , where n rated is the rated speed of the motor; the condition for high-speed operation is that the motor speed n satisfies the condition: n>n rated .
  • the two magnetic circuits in the motor at this time are the main magnetic circuit I and the leakage magnetic circuit II, and the main magnetic circuit I is connected in parallel with the magnetic leakage circuit II.
  • the magnetic flux path of the main magnetic circuit I is as follows: starting from the first rectangular permanent magnet 2.4.1 in the first permanent magnet group 2.4, passing through the rotor core 2.3 and the first arc-shaped permanent magnet in sequence 2.4.3, air gap, inner stator 3, air gap, second arc-shaped permanent magnet 2.4.3 adjacent to the first arc-shaped permanent magnet 2.4.3, rotor core 2.3, and the second arc-shaped permanent magnet 2.4.3.
  • One permanent magnet group 2.4 is adjacent to the second rectangular permanent magnet 2.4.2 and the rotor core 2.3 in the second permanent magnet group 2.4, and finally returns to the first rectangular permanent magnet 2.4. 1. Form a closed magnetic circuit. Due to the existence of the magnetic leakage circuit, there is a magnetic leakage circuit II in the motor that is different from the traditional magnetic circuit.
  • the magnetic flux path of the magnetic leakage circuit II is: from the first rectangular permanent magnet in the first permanent magnet group 2.4 Starting from steel 2.4.1, it passes through the rotor iron core 2.3, the second arc-shaped permanent magnet steel 2.4.3 adjacent to the first arc-shaped permanent magnet steel 2.4.3, the rotor iron core 2.3, and the first arc-shaped permanent magnet steel 2.4.3.
  • the second rectangular permanent magnet 2.4.2 and the rotor core 2.3 in the second permanent magnet group 2.4 adjacent to the first permanent magnet group 2.4 finally return to the first rectangular permanent magnet 2.4.1 , forming a closed magnetic circuit in the outer rotor 2. It can be seen that the main magnetic circuit I and the leakage magnetic circuit II are connected in parallel.
  • the main magnetic circuit I and the q-axis magnetic circuit III operate in parallel.
  • the main magnetic circuit I is the same as the main magnetic circuit I during light load operation in Figure 15.
  • the magnetic flux path of the q-axis magnetic circuit III is: starting from the inner stator 3, passing through the air gap and arc-shaped permanent magnet 2.4 in sequence. 3.
  • a collaborative control system is used to control the unitized permanent magnet hub motor with multiple poles and few slots according to the present invention.
  • the collaborative control system includes a battery, two control modules and N winding electronic switches.
  • One winding electronic switch is connected to a centralized winding 4 in a motor unit 1 to control the flow of the centralized winding 4 in a motor unit 1. break.
  • Each control module is composed of a power electronic switch, a DSP controller and an inverter connected in series. The input terminals of the two power electronic switches are connected to the output terminals of the battery respectively, and the output terminals of each inverter are connected respectively.
  • N/2 winding electronic switches that is, the first control module is composed of the first power electronic switch, the first DSP controller and the first inverter in series, and the second control module is controlled by the second power electronic switch and the second DSP
  • the converter and the second inverter are connected in series.
  • the power electronic switches in the two control modules are independent of each other.
  • the two control modules are independent of each other and have the same structure.
  • Each controls N/2 motor units 1, which reduces the coupling between the two modules and improves the unitization of multiple poles and few slots. The degree of freedom and control quality of permanent magnet hub motor control.
  • N winding electronic switches are divided into two independent groups, and each group of winding electronic switches controls N/2 motor units 1.
  • N winding electronic switches are divided into two groups and connected to two control modules respectively.
  • the first winding electronic switch is connected to the first motor unit 1
  • the second winding electronic switch is connected to the second motor unit 1
  • the N/2nd winding electronic switch is connected to the N/2nd motor unit 1.
  • the N/2+1th winding electronic switch is connected to the N/2+1th motor unit 1
  • the Nth winding electronic switch is connected to the Nth motor unit 1.
  • the input terminals of a total of N/2 winding electronic switches from the first winding electronic switch to the N/2th winding electronic switch are respectively connected to the first inverter in the first control module.
  • each motor unit 1 can be independently controlled on and off through the corresponding winding electronic switch, and the total armature magnetic field intensity of the motor can achieve multi-level transformation, which is beneficial to adjusting the motor magnetic field and realizing multi-working condition operation of the motor.
  • the control of mutually independent motor units 1 can further improve the operating freedom of multi-pole and few-slot unitized permanent magnet hub motors, and provides a hardware foundation for improving the working efficiency of the motor and its collaborative control system.
  • the output ends of the centralized windings 4 of all motor units 1 are connected to the battery through the feedback module.
  • the main component of the feedback module is the rectifier.
  • the feedback module is designed for the motor in the power-boosting operation mode. The energy generated by the motor operation is higher than that required for the wheel hub drive. of energy, the remaining energy is recycled into the battery through the feedback module.
  • the collaborative control system adopts the following control strategy to control the multi-pole and few-slot unitized hub permanent magnet motor proposed by the present invention:
  • Step 1 Use finite element software simulation to obtain the external characteristic curve of the motor when the control module is working.
  • the two power electronic switches and N winding electronic switches are all closed, and finite element software simulation is used to obtain the external characteristic curve g of the motor, as shown in Figure 18.
  • the two power electronic switches and N winding electronic switches are all closed, and finite element software simulation is used to obtain the external characteristic curve g of the motor, as shown in Figure 18.
  • the simulation is obtained At this time, the external characteristic curve f of the motor; the abscissa is the motor speed n, the unit is rpm; the ordinate is the motor output torque T, the unit is Nm.
  • Step 2 Based on the obtained external characteristic curve, divide the constant torque area and constant power area of the motor.
  • Step 2.1 First calculate the critical speed n b .
  • the maximum torque T max is obtained from the external characteristic curve f, and the critical speed n b is the maximum speed corresponding to the maximum torque T max , that is:
  • n i is the speed set of the operating point under the motor output peak torque.
  • Step 2.2 According to the critical speed n b , divide the constant torque area and the constant power area.
  • point P(n p ,T p ) be any operating point of the motor
  • n p be the rotation speed at the point
  • T p be the torque at the point.
  • the speed n p at this point satisfies n p ⁇ n b
  • this point belongs to the constant torque zone, that is, the speed n p ⁇ n b at the motor operating point in the constant torque zone; if the speed n p at this point satisfies n p > n
  • the vertical dotted line h is the dividing line between the constant torque area and the constant power area
  • the rotation speed n p n b at the dividing line.
  • Step 3 Then for the constant torque zone, divide the first zone low-speed load zone and the second zone low-speed overload zone according to the critical torque.
  • Step 3.1 Based on finite element software simulation, calculate the peak torque T x of a motor unit 1. On the premise that both the first power switch and the second power electronic switch are closed, that is, the first control module and the second control module are both working, and only one of the winding electronic switches is closed. Use finite element software to simulate this situation. Motor peak torque, which is the peak torque T x of a motor unit 1;
  • the critical torque T b is calculated based on the peak torque T x of one electric machine unit 1 .
  • the critical torque T b is N-2 times the peak torque T x of a motor unit 1, that is:
  • T b (N-2)T x .
  • Step 3.3 Refer to Figure 19.
  • the abscissa is the motor speed in rpm; the ordinate is the motor output torque in Nm.
  • T b Based on the critical torque T b , two zones are determined, namely the first low-speed load zone and the second low-speed overload zone.
  • the operating point P (n p ,T p ) meets the conditions: T p ⁇ T b , n p ⁇ n b , then the operating point P (n p , T p ) belongs to the first zone low-speed load zone, that is, in In the constant torque zone, if the torque T p of the operating point P(n p ,T p ) is equal to or less than the critical torque T b , then the operating point P(n p ,T p ) belongs to the first low-speed load zone S 1 ; on the contrary, P(n p ,T p ) satisfies the conditions: T p > T b , n p ⁇ n b , then point P at this time belongs to the second zone low-speed overload zone S 2 .
  • the maximum torque T p at the operating point P (n p , T p ) in the second zone low-speed overload zone S 2 does not exceed the torque of the external characteristic curve
  • the horizontal straight line j in Figure 19 is the dividing line between the first zone low-speed load zone and the second zone low-speed overload zone. Its function expression is:
  • T(n) T b ,n ⁇ [0,n b ].
  • the grid area is the first low-speed load area
  • the left slash area is the second low-speed and low-load area.
  • Step 4 Based on the zoning results of the constant torque zone in step 3, determine the control method of the first low-speed load zone.
  • the main control principle of the first low-speed load zone is the minimum torque ripple principle: both the first power electronic switch and the second power electronic switch are closed, and the first inverter and the second inverter output currents with the same amplitude and different phases. , adjust the output torque phase of the two control modules so that the valleys and peaks overlap and eliminate high-order amplitude harmonics.
  • P 1 n p1 ,T p1
  • T p1 is the torque at the point.
  • Step 4.1 According to the output torque T p1 required by the motor, calculate the number K of closed winding electronic switches connected to each control module, 1 ⁇ K ⁇ N/2.
  • the required output torque of the motor is T p1
  • the required torque output of a single control module is T p1 /2.
  • the peak torque of a single motor unit 1 obtained in step 3.1 is T x . Therefore, among the N/2 motor units 1 connected to each control module, the minimum number K of winding electronic switches that need to be closed is (T p1 /2T x ).
  • the number K of winding electronic switch closures in each control module is rounded to the nearest integer using the "one-step method", that is:
  • Step 4.2 Based on the number K of closed winding electronic switches in 4.1, use finite element software simulation to calculate the transition of the output of the two inverters when the output torque T 2 (t) when the number of closed winding electronic switches is K current.
  • the first power electronic switch and the second power electronic switch are both closed, any K switches from the first winding electronic switch to the N/2th winding electronic switch are closed, and any K switches from the (N/2+1)th winding electronic switch to the (N/2+1)th winding electronic switch are closed. Any K switches among the N winding electronic switches are closed.
  • the finite element software simulation is used to calculate the motor output torque waveform T 2 (t) and the transition current output by the two inverters at this time.
  • the transition current amplitude is I max0
  • the sinusoidal function of the transition current phase ⁇ 0 the expression is:
  • n p is the rotation speed of the operating point P (n p ,T p ), and P r is the number of rotor pole pairs.
  • Step 4.3 Based on the results of step 4.2, the waveform of the output torque T 2 (t) obtained by fast Fourier decomposition is used to obtain the harmonic order r of the main harmonic component.
  • the waveform of the output torque T 2 (t) is decomposed by Fast Fourier into the DC component T 20 , the main harmonic component T r cos (rw 2 t+ ⁇ r ) and the remaining harmonic components
  • the sum of where the amplitude of the main harmonic component T r cos(rw 2 t+ ⁇ r ) is T r and the phase is ⁇ r ; the remaining harmonic components
  • the degree is v, its amplitude is T v , and its phase is ⁇ v . Therefore, the fast Fourier decomposition expression of the output torque T 2 (t) is:
  • w 2 is the rotation speed of the output torque T 2 (t).
  • Step 4.4 Calculate the current dislocation angle ⁇ based on the harmonic order r of the main harmonic component T r cos (rw 2 t+ ⁇ r ).
  • the current misalignment angle ⁇ is the angle difference between the current phases output by the two inverters. Its calculation method is the quotient of ⁇ /2 and the harmonic order r of the main harmonic component, that is:
  • Step 4.5 Transition current based on the two inverter outputs in step 4.2 And the current misalignment angle ⁇ calculated in step 4.4 is used to determine the ABC three-phase current output by the two inverters.
  • the phase of the ABC three-phase current output by the first inverter is ahead of the above-mentioned excessive current.
  • Phase ⁇ /2 the ABC three-phase current output by the second inverter lags behind the excess current Phase ⁇ /2, that is, the expression is:
  • I 1A , I 1B and I 1C are the ABC three-phase currents output by the first inverter;
  • I 2A , I 2B and I 2C are the ABC three-phase currents output by the second inverter,
  • n p is the operating point P (n p ,T p ),
  • P r is the number of rotor pole pairs.
  • the three-phase currents I 1A , I 1B and I 1C , I 2A , I 2B and I 2C are stored in the first and second DSP controllers respectively, and the first inverter is controlled by the first and second DSP controllers respectively. and the second inverter outputs three-phase current.
  • Step 5 Based on the zoning results of the constant torque zone in step 3, determine the control method of the second zone low-speed overload zone. Since the second zone low-speed overload zone requires higher motor torque, all motor units 1 work together, that is, for any operating point P 2 (n p2 ,T p2 ) in the second zone low-speed overload zone, all The power electronic switch and all winding electronic switches are closed, and then finite element software simulation is used to calculate the ABC three-phase current output by the two inverters when the motor runs at point P 2 (n p2 , T p2 ).
  • Step 6 At the same time as step 3, based on the results in step 2, perform detailed partitioning processing of the constant power area.
  • the constant torque zone is divided into 6 sub-areas, divided into the third to eighth zones.
  • the third zone is a high-speed low-load zone, denoted as S 3 ;
  • the fourth zone is a high-speed and high-efficiency zone, denoted as S 4 ;
  • the fifth zone is a high-speed load zone, denoted as S 5 ; and the fifth zone is a double-efficiency zone.
  • the seventh area is the high-speed overload area, marked as S 7 ; the eighth area is the high-speed field weakening area, marked as S 8 .
  • the maximum value of the torque at the operating point P(n p ,T p ) in the constant power zone does not exceed the torque corresponding to the external characteristic curve g.
  • Step 6.1 Determine the scope of the fourth high-speed and high-efficiency zone.
  • Step 6.1.1 Simulate and calculate the efficiency map of the operation of a single control module. At this time, only one of the first power electronic switch and the second power electronic switch is closed, and all the winding electronic switches are closed. Under this premise, finite element software simulation is used to obtain the efficiency map of motor operation, and the maximum efficiency ⁇ max of the motor operating in a single control module is determined based on the efficiency map.
  • Step 6.1.2 Calculate the boundary efficiency ⁇ b based on the maximum efficiency ⁇ max of the motor running with a single control module:
  • Step 6.1.3 Determine the range of the fourth high-speed and high-efficiency area based on the boundary efficiency eta b .
  • the highest speed point E and the lowest speed point D of the fourth high-speed high-efficiency zone are obtained, as shown in Figure 20.
  • Step 6.2 Based on the results of step 6.1, determine the range of the eighth high-speed magnetic field weakening zone.
  • Step 6.2.1 The highest speed point E in the fourth high-speed and high-efficiency zone is not unique. Then take the lowest torque point among all the highest speed points as the required highest speed point E, and obtain a vertical line passing through the highest required speed point E. The straight line j on the abscissa.
  • Step 6.2.2 Determine the range of the eighth high-speed magnetic field weakening zone.
  • the area surrounded by the abscissa axis, curve g and straight line j is the eighth high-speed field weakening zone.
  • the rotation speeds of the eighth high-speed field weakening zone are higher than the rotation speed n E of the required highest speed point E.
  • Step 6.3 Based on the results in steps 6.1 and 6.2, determine the range of the third zone high-speed low-load zone.
  • Step 6.3.1 Based on the results in step 6.1, if the lowest speed point D(n D , T D ) in the fourth high-speed and efficient zone is not unique, where n D and T D are the lowest speed points D(n D) respectively. , T D ), then take the lowest torque point among all the lowest speed points D (n D , T D ) as the required lowest speed point D, and obtain a line perpendicular to the horizontal direction passing through the lowest required speed point D. coordinates of the straight line k.
  • Step 6.3.2 Based on the results in step 6.3.1, determine the range of the third high-speed low-load zone in the area where the rotation speed is less than point D, denoted as S 31 .
  • the area where the torque is less than the point D torque T D and the speed is less than the point D speed n D belongs to the third high-speed and low-load zone.
  • point P (n P ,T P ) meets the conditions: n p ⁇ n D and T P ⁇ T D
  • Step 6.3.3 Determine the range of the third high-speed low-load zone in the area where the rotational speed is greater than point D (n D , T D ) and less than point E (n E , TE ) , recorded as S 32 .
  • S 32 is the area surrounded by the abscissa axis, the straight line k, the lower half boundary line of the fourth zone high-speed and high-efficiency zone (the lower half connecting line between point E and point D) and the straight line j.
  • Step 6.3.4 Based on steps 6.3.2 and 6.3.3, determine the range of the third zone high-speed low-load zone.
  • the size of the above-mentioned third zone high-speed low-load zone S 31 and S 32 will change as the characteristics of the motor change.
  • some of the sets may be empty sets.
  • Step 6.4 At the same time as step 6.3, based on the high-speed and high-efficiency zone of the fourth zone determined in step 6.1, determine the double college zone of the sixth zone.
  • the Sixth District's Double High School Zone is a double expansion area of the Fourth District's high-speed and high-efficiency zone. Its expansion method is that the speed remains unchanged and the torque is doubled. That is, the speed of the Sixth District's Double High School Zone is the same as the original speed.
  • the speeds of the fourth high-speed and high-efficiency zone are the same.
  • the highest speed point G in the sixth zone double high school zone is on the straight line j, and the lowest speed point F is on the straight line k; for each operating point P in the sixth zone double high school zone 6 (n P6 ,T P6 ), whose torques are twice the torque of the same speed n P6 operating point in the fourth high-speed and high-efficiency zone.
  • n P6 ,T P6 the double high school zone in the sixth zone
  • it is a double expansion point of a certain point in the high-speed and high-efficiency zone in the fourth zone. That is, when point P(n P ,T P ) belongs to the sixth double-efficiency area, point P needs to meet the conditions: and P' is a point in the fourth high-speed high-efficiency zone, that is:
  • Step 6.5 Based on the results in steps 6.1 and 6.4, determine the range of the fifth zone high-speed load zone.
  • Step 6.5.1 The area where the torque is less than the torque T F of point F and higher than the torque of point D and the rotation speed is less than the rotation speed n F of point F belongs to the fifth zone high-speed load zone, recorded as S 51 . That is, when point P (n P ,T P ) meets the conditions: n p ⁇ n F and T D ⁇ T P ⁇ T F , point P (n P , T P ) belongs to the fifth zone high-speed load zone, that is :
  • Step 6.5.2 Determine the range of the fifth high-speed load zone in the area where the rotational speed is greater than point F (n F , T F ) and less than point G (n G , T G ), recorded as S 52 .
  • S 52 is the area composed of straight line k, straight line j, the upper half-peripheral boundary of the fourth area high-speed high-efficiency area and the lower half-peripheral boundary of the sixth area double-efficiency area.
  • Step 6.5.3 Based on steps 6.5.1 and 6.5.2, determine the range of the fifth zone high-speed load zone.
  • Step 6.6 Based on the results in steps 6.1 to 6.5, determine the range of the seventh high-speed overload zone.
  • the seventh zone range is the range in the constant power zone that does not belong to the third, fourth, fifth, sixth and eighth zones, that is, the remaining area of the constant power zone is the seventh zone.
  • point P(n P ,T P ) satisfies the conditions: Then point P(n P ,T P ) belongs to the seventh area high-speed overload area, that is:
  • the six sub-areas divided into the constant power area are shown with different labels in the figure.
  • the right diagonal area is the third high-speed low-load area
  • the dark gray area is the fourth high-speed high-efficiency area
  • the oblique grid area is the fifth high-speed load area
  • the white area is the sixth double-high efficiency area
  • the dot area is
  • the seventh zone is the high-speed overload zone
  • the light gray is the eighth zone, the high-speed weak magnetic zone.
  • Step 7 Based on the zoning results of the constant power zone in step 6, determine the control method of the third zone high-speed low load zone. Since the torque of the third high-speed low-load zone is lower than that of the fifth, sixth and seventh zones, and the speed is also lower than that of the eighth zone, for any point P in the third high-speed low-load zone 3 (n p3 ,T p3 ), n p3 and T p3 are the speed and torque at any point in the third zone, high speed and low load zone respectively.
  • the first power electronic switch and the second power electronic switch have and only one closed At the same time all winding electronic switches are closed.
  • the main control principle of the third high-speed and low-load zone is the principle of maximizing system efficiency.
  • the method is to use the method of increasing the power of the motor, that is, increasing the output torque without changing the motor speed, and the motor runs at the increased power operating point H. 3 (n H3 ,T H3 ), excess energy is fed back to the battery, thereby improving the efficiency of the motor system.
  • the control method is determined by the following steps:
  • Step 7.1 Calculate the total power W 3 consumed by the system at any point P 3 (n p3 , T p3 ) in the third zone high-speed low-load zone.
  • the motor runs at point P 3 (n p3 ,T p3 )
  • its efficiency is eta p3
  • the total power consumed by the system at point P 3 (n p3 ,T p3 ) W 3 is the product of speed and torque divided by 60 efficiency 2 ⁇ times the value, that is:
  • Step 7.2 At the same time as step 7.1, calculate the total power W 3 ' consumed by the system at the transition point P 3 '(n p3 ,T p3' ), and the transition point P 3 '(n p3 ,T p3' ) is the point P 3 (n p3 ,T p3 ) at any point with the same rotational speed.
  • Step 7.3 Based on steps 7.1 and 7.2, establish a mathematical model for power-increasing operation.
  • the mathematical model of power-increasing operation is the total power consumed by the system at transition point P 3 'W 3 ' minus the total power consumed by the system at point P 3 W 3 , which is:
  • Step 7.4 Based on step 7.3, determine the set SP3 . Let the mathematical model of power-increasing operation be greater than 0, then the above formula The following formula can be obtained: then all conditions that satisfy The set of transition points P 3 '(n p3 ,T p3' ) is S P3 .
  • S P3 may be an empty set, which mainly depends on the motor power and specific performance of the motor.
  • S P3 is an empty set, the motor power-increasing operation method is not used, but it is operated directly.
  • Step 7.5 Based on step 7.4, when S P3 may be a non-empty set and the motor is operated with increased power, further determine the increased power operating point H 3 (n H3 , T H3 ) in the set S P3 .
  • the power-increasing operating point H 3 (n H3 , T H3 ) needs to belong to S P3 and also needs to belong to the third zone high-speed low-load zone or the fourth zone high-speed and high-efficiency area, it is also necessary to satisfy that the torque TH3 of the power-increasing operating point H3 is greater than or equal to TP3 . That is to say, the power-increasing operating point H 3 (n H3 ,T H3 ) needs to satisfy:
  • Step 7.6 Use finite element software simulation to obtain the inverter output ABC three-phase current when the motor runs at point H 3 (n H3 , T H3 ).
  • Step 8 At the same time as step 7, based on the zoning result of step 6, determine the control method of the fourth zone high-speed and high-efficiency zone. Since the torque of the fourth high-speed and high-efficiency zone is lower than that of the fifth, sixth and seventh zones, and the speed is also lower than that of the eighth zone, for any point P 4 ( n p4 ,T p4 ), when one and only one of the first power electronic switch and the second power electronic switch is closed, all winding electronic switches are closed at the same time. In addition, the main control principle of the fourth high-speed and high-efficiency zone is the principle of highest system efficiency.
  • the operating point P 4 (n p4 ,T p4 ) works normally. Finite element software simulation is used to obtain the ABC three-phase current output by the inverter when the motor runs at P 4 (n p4 ,T p4 ).
  • Step 9 At the same time as step 8, based on the zoning result of step 6, determine the control method of the fifth zone high-speed load zone. Since the torque in the fifth zone high-speed load zone is higher than that in the third and fourth zones, for any point P 5 (n p5 ,T p5 ) in the fifth zone high-speed load zone, all power electronic switches are closed and all winding electronic switches are closed. The switch is closed. In addition, the main control principle of the fifth high-speed load zone is the principle of maximizing system efficiency.
  • the main method is to use the method of increasing the power of the motor, that is, increasing the output torque without changing the motor speed, and the motor runs at the increased power operating point H 5 (n H5 ,T H5 ) excess energy is fed back to the battery, thereby improving the efficiency of the motor system.
  • the control method is as follows:
  • Step 9.1 Calculate the total power W 5 consumed by the system at point P 5 (n p5 ,T p5 ).
  • the efficiency is ⁇ p5
  • the total power W 5 consumed by the system at point P 5 (n p5 , T p5 ) is divided by the product of rotation speed n p5 and torque T p5 Taking 2 ⁇ times the value of 60 efficiency, that is:
  • Step 9.2 At the same time as step 9.1, calculate the total power W 5 ' consumed by the system at the transition point P 5 '(n p5 ,T p5' ).
  • the transition point P 5 '(n p5 ,T p5' ) is the point P 5 (n p5 ,T p5 ) any point with the same rotational speed.
  • the motor runs at point P 5 '(n p5 ,T p5' ) its efficiency is eta p5
  • the total power consumed by the system at transition point P 5 '(n p5 ,T p5' ) is Consumption minus the efficiency fed back to the battery is:
  • Step 9.3 Based on steps 9.1 and 9.2, establish a mathematical model for power-increasing operation.
  • the mathematical model of power-increasing operation is the total power consumed by the system at transition point P 5 ' W 5 ' minus the total power consumed by the system at point P 5 W 5 , which is:
  • Step 9.4 Based on step 9.3, determine the set SP5 .
  • the mathematical model f(T p5' ) of power-increasing operation be greater than 0, and the following formula is obtained:
  • the transition point P 5 '(n p5 ,T p5' ) should satisfy the conditions of the above formula, and the transition points P 5 '(n p5 ,T p5' ) that satisfy the conditions form a set S P5 . It should be noted that for some points P 5 (n p5 ,T p5 ), S P5 may be an empty set, which mainly depends on the motor power and specific performance of the motor. When S P5 is an empty set, the fifth high-speed load zone does not use the motor power-up operation method, but runs directly.
  • Step 9.5 When the set S P5 is a non-empty set and the motor is operated with increased power, the increased power operating point H 5 (n H5 , T H5 ) is determined in the set S P5 .
  • the power-increasing operating point H 5 (n H5 , T H5 ) needs to belong to S P5 and also needs to belong to the fifth zone high-speed load zone or the sixth zone double efficiency
  • the area in addition, needs to satisfy T H5 to be greater than or equal to T P5 . That is to say, the power-increasing operating point H 5 (n H5 ,T H5 ) needs to satisfy:
  • Step 9.6 Based on the results in step 9.5, use finite element software simulation to obtain the ABC three-phase current output by the inverter when the motor runs at point H 5 (n H5 , T H5 ).
  • Step 10 Simultaneously with step 9, based on the zoning result of step 6, determine the control method of the sixth double-efficiency zone. Since the torque of the sixth double-efficiency zone is higher than that of the third, fourth and fifth zones, for any point P 6 (n p6 ,T p6 ) in the sixth double-efficiency zone, all power electronic switches When both are closed, all winding electronic switches are closed. In addition, the main control principle of the sixth double-efficiency zone is the principle of highest system efficiency. Since the operating point efficiency of the sixth zone double-efficiency zone is higher than that of the third, fifth, seventh and eighth zones, the operation Point P 6 (n p6 ,T p6 ) works normally. Finite element software simulation is used to obtain the ABC current output by the inverter when the motor runs at P 6 (n p6 ,T p6 ), and the DSP controller controls the output of the inverter.
  • Step 11 Simultaneously with step 10, based on the zoning result of step 6, determine the control method of the seventh zone high-speed overload zone. Since the torque of the seventh zone high-speed overload zone is higher than that of the third zone, fourth zone and fifth zone, for any operating point P 7 (n p7 ,T p7 ) in the seventh zone high-speed overload zone, all power electronic switches When both are closed, all winding electronic switches are closed. In addition, the main control principle of the seventh zone high-speed overload zone is the principle of maximizing system efficiency. Therefore, the seventh zone control strategy is to coordinate the work of the two control modules, that is, the output currents of the first inverter and the second inverter are different.
  • the first control module makes the motor work at point P 71 (n p71 ,T p71 ); the second control module makes the motor work at point P 72 (n p72 , T p72 ).
  • the control method is as follows:
  • Step 11.1 Establish the cooperating power function.
  • the seventh zone high-speed overload zone has two different points, namely the first transition point P 71 '(n p71' ,T p71' ) and the second transition point P 72 '(n p72' ,T p72' ).
  • the rotational speeds of each transition point are equal to the seventh zone operating point P 7 (n p7 ,T p7 ) and the sum of the torques is T P7 , that is The set of transition points that satisfy the conditions is denoted as SP7 .
  • the total power W 71 ' consumed by the system at the first transition point P 71 '(n p71' ,T p71' ) is 2 ⁇ times the product of the rotational speed n p71' and the torque T p71' divided by 60 efficiency, that is:
  • the total power W 72 ' consumed by the system at the second transition point P 72 '(n p72' ,T p72' ) is 2 ⁇ times the product of the speed and torque divided by 60 efficiency, that is:
  • the cooperative operation power function is the first transition point P 71 '(n p71' ,T p71' ) total power consumed by the system W 71' and the second transition point P 72 '(n p72' ,T p72' ) total power consumed by the system W
  • the sum of 72' is:
  • Step 11.2 Based on the cooperative operation power function W 7 (T p71 ', T p72 ') determined in step 11.1, determine the two operating points P 71 (n p71 , T p71 ) and P 72 (n p72 , T p72 ).
  • the total power consumed by the system at the first operating point P 71 (n p71 , T p71 ) is W 71
  • the total power consumed by the system at the second operating point P 72 (n p72 , T p72 ) is W 72
  • the two operating points conform to their cooperative operation power function W 7 (T p71 ,T p72 ) is equal to the cooperative operation power function W 7 ( of the two transition points P 71 '(n p71' ,T p71' ) and P 72 '(n p72' ,T p72' )
  • T p71 ', T p72 ') is the minimum value
  • the motor works at the two operating points P 71 (n p71 ,
  • Step 11.3 Based on the point P 71 (n p71 ,T p71 ) determined in step 11.2, determine the first inverter output current. At this time, when all the winding electronic switches are closed, only the first power electronic switch is closed, and the output current of the first inverter is obtained using finite element software simulation.
  • the current output by the second inverter is determined.
  • the second power electronic switch is closed, and the output current of the second inverter is obtained using finite element software simulation.
  • Step 12 Simultaneously with step 11, based on the partition result of step 6, determine the control strategy for the eighth high-speed field weakening zone. Since the rotation speed of the eighth high-speed field weakening zone is higher than that of all other sub-regions, for any point P 8 (n p8 ,T p8 ) in the eighth high-speed field weakening zone, all power electronic switches are closed and all winding electronic switches are closed, Finite element software simulation is used to obtain the inverter output current when the motor runs at P 8 (n p8 ,T p8 ).
  • the motor is divided into 8 working modes according to the characteristics of the sub-region:
  • both the first power electronic switch and the second power electronic switch are closed, and any K switches from the first winding electronic switch to the N/2th power electronic switch are closed, and Close any K switches from the (N/2+1)th winding electronic switch to the Nth power electronic switch.
  • the two inverters output currents with the same frequency, same amplitude and different phases.
  • the motor units 1 of the two parts work at the same operating point.
  • the torque waveform trough compensation greatly reduces the torque pulsation of the motor in this sub-region and improves the rotation speed. moment mass.

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Abstract

本发明公开一种多极少槽单元化永磁轮毂电机及协同控制系统和方法，电机由沿径向截面圆周方向均匀分布的N个相同的电机单元组成，每个1/N的外转子沿轴向上分为相同的M个转子段，M个转子段沿同一旋转方向依次地旋转一个机械错位角布置；将恒转矩区分成第一区和第二区，将恒功率区分成第三区至第八区，控制系统包括两个控制模块，每个控制模块均由一个电源电子开关，一个DSP控制器和一个逆变器依次串联组成，两个电源电子开关的输入端分别连接电池的输出端，每个逆变器的输出端分别连接N/2个绕组电子开关，一个绕组电子开关控制一个电机单元中集中式绕组的通断，相互独立的绕组电子开关实现电机单元的独立控制，提高电机控制自由度和运行效率。

Description

一种多极少槽单元化永磁轮毂电机及协同控制系统和方法 技术领域
本发明属于永磁电机领域，特指一种适用于电动汽车、轮船推进、电动拖拉机等需要高效率以及多工况运行特点的永磁轮毂电机。
背景技术
永磁轮毂电机因其有高转矩密度与高功率密度等优势在电动汽车、轮船推进、电动拖拉机等直驱领域显示出潜在的应用前景。但是，永磁磁场恒定且难以调节，深度弱磁较难导致该类电机存在调速范围小，高速效率低，难以适用于电动汽车等多工况运行场所。
中国专利号为202210042232.8的文献中提出了一种凸击式混合励磁电机，该电机采用永磁磁钢与励磁电流的混合励磁方式，励磁电流可调这一特点使得电机励磁磁场弱磁难度降低，实现了电机多工况运行；但是励磁电流的引入增加了电机铜耗，降低了整机运行效率。中国专利号为201410768272.6的文献中提出了一种定子聚磁式混合永磁记忆电机，该电机中采用了低矫顽力的软磁材料与稀土永磁磁钢共同作用的励磁方式，软磁材料的磁势随着脉动绕组电流的改变而改变，励磁磁场强度也随之改变，进而降低了电机弱磁难度，实现电机多工况运行；但是软磁材料与脉动绕组的加入增加了电机的体积与重量，降低了电机功率密度，同时脉冲绕组的引入也提高了电机损耗，降低电机运行效率。
因此，如何实现多工况运行的高效性是永磁轮毂电机亟需解决的问题。
发明内容
本发明的目的是针对现有永磁轮毂电机多工况运行存在的问题，提出一种多槽少极单元化永磁轮毂电机以及针对该电机的多单元协同控制系统和方法，以满足永磁轮毂电机在多工况运行需求的前提下提高运行效率，扩宽运行高效区。
本发明一种多极少槽单元化永磁轮毂电机采用的技术方案是：其由沿径向截面圆周方向均匀分布的N个相同的电机单元组成，每个电机单元包括1/N的外转子、1/N的内定子以及1/N的集中式绕组，内定子同轴心套在外转子内部，内定子上绕有集中式绕组，每个电机单元中的集中式绕组均为三相对称且分布相同；外转子具有一个转子铁芯，2a个永磁磁钢组沿圆周方向均匀分布在转子铁芯上，每个永磁磁钢组均由第一、第二矩形永磁磁钢和弧形永磁磁钢组成，第一、第二矩形永磁磁钢结构相同且径向截面均呈矩形，内外斜向为矩形长度方向，呈开口朝向气隙侧的V型布置于弧形永磁磁钢外侧，相对于弧形永磁磁钢沿直径方向的中心线对称；第一、第二矩形永磁磁钢充磁方向垂直于自身长度方向，弧形永磁磁钢充磁方向和中心线方向一致，属于同一个永磁磁钢组的第一、第二矩形永磁磁钢及弧形永磁体充磁方向同时指向或背离气隙，相邻两个永磁磁钢组充磁方向相反；转子极对数P r、定子槽数N s、电机相数m、电机槽距角τ以及N需同时满足：P r＞N s、P r＝Na、N s＝mNb、
Figure PCTCN2022128230-appb-000001
cτ＝d*2π、
Figure PCTCN2022128230-appb-000002
N＝2i，i,a,b,c,d,e均是正整数。
每个所述的1/N的外转子，沿轴向上分为相同的M个转子段，M个转子段沿同一旋转方向依次地旋转一个机械错位角布置，20mm≤l ef/M≤120mm，l ef为电机轴向长度。
进一步地，每个弧形永磁磁钢在径向截面上均由一条弧形永磁磁钢外长边，一条弧形永磁磁钢内长边以及两条弧形永磁磁钢短边围成，弧形永磁磁钢外长边和弧形永磁磁钢内长边的弧形中心与外转子中心相同，弧形永磁磁钢短边和外转子的直径方向一致，弧形永磁磁钢内长边为正弦曲线f 1(θ 1)＝f 1maxsin(θ 1)，θ 1∈[π,2π]，f 1max为幅值，当θ 1为3π/2时，点f 1(3π/2)位于外转子内表面上。
进一步地，每个第一、第二矩形永磁磁钢在靠近气隙侧的端部设有内磁障，远离气隙侧的端部设有外磁障，每个弧形永磁磁钢的切向两端各设有一个成为气隙一部分的虚拟槽。
本发明所述的多极少槽单元化永磁轮毂电机的协同控制系统采用的技术方案是：其包括一个电池、两个控制模块和N个绕组电子开关，一个绕组电子开关控制一个电机单元中集中式绕组的通断，每个控制模块均由一个电源电子开关，一个DSP控制器和一个逆变器依次串联组成，两个电源电子开关的输入端分别连接电池的输出端，每个逆变器的输出端分别连接N/2个绕组电子开关，集中式绕组输出端经整流器连接电池。
本发明所述的协同控制系统的控制方法采用的技术方案是：
闭合两个电源电子开关与N个绕组电子开关，以横坐标为电机转速，纵坐标为电机输出转矩，仿真获得电机的外特性曲线g；再断开其中一个电源电子开关，仿真获得电机的外特性曲线f；以外特性曲线f上最高转矩对应的最高转速作为临界转速n b，当电机运行点的转速n p≤n b，为恒转矩区；
将所述的恒转矩区分成第一区和第二区，当运行点的转矩T p≤T b，为第一区，当T p＞T b，为第二区，T b为临界转矩T b，是一个电机单元(1)峰值转矩T x的N-2倍；第二区的运行点转矩的最大值不超过外特性曲线g对应的转矩；
当运行点在第一区，两个电源电子开关均闭合，与每个控制模块连接的N/2电机单元中至少闭合
Figure PCTCN2022128230-appb-000003
个绕组电子开关，T p1为第一区运行点的转矩；两个逆变器输出同幅值不同相位的电流；当运行点在第二区，两个电源电子开关和N个绕组电子开关均闭合。
当电机运行点转速n p＞n b，为恒功率区；将所述的恒功率区分成第三至第八区，当运行点的效率η p≥η b，则为第四区，η b为单一控制模块运行时电机的边界效率；由横坐标轴、外特性曲线g与经过第四区速度最高点E的一条垂直于横坐标的直线j所围的区域为第八区；转矩小于第四区速度最低点D的转矩且转速小于点D的转速的区域为区域S 31，由横坐标、经过点D的一条垂直于横坐标的直线k、第四区点D、E下半周边界线、直线j所围成的区域为区域S 32，S 31与S 32的并集为第三区；转速与第四区转速相同且转矩是第四区转矩的两倍的区域是第六区，第六区的速度最高点G在直线j上，速度最低点F在直线k上；转矩小于点F的转矩、高于点D的转矩且转速小于点F转速的区域为区域S 51，由直线k、直线j、第四区点D、E上半周边界、第六区点F、G下半周边界所围成的区域为区域S 52，S 51与S 52的并集为第五区；恒功率区中的剩余区域为第七区；
当运行点在第三区，两个电源电子开关有且仅有一个闭合，N个绕组电子开关均闭合，且采用不改变电机转速的同时提高转矩的升功率运行；当运行点在第四区，两个电源电子开关有且仅有一个闭合，N个绕组电子开关均闭合；当运行点在第五区，两个电源电子开关均闭合，N个绕组电子开关均闭合且升功率运行；当运行点在第六区，两个电源电子开关均闭合，N个绕组电子开关均闭合；当运行点在第七区，两个电源电子开关均闭合，N个绕组电子开关均闭合，两个逆变器输出的电流不同；当运行点在第八区，两个电源电子开关均闭合，N个绕组电子开关均闭合。
进一步地，仿真计算K个绕组电子开关时电机输出转矩波形和两个逆变器输出的过渡电流，傅里叶分解所述的输出转矩波形，求取主要谐波分量的谐波次数r，计算出电流错位角
Figure PCTCN2022128230-appb-000004
则两个逆变器输出的三相电流幅值I max＝1.05I max0，I max0为过渡电流幅值，第一个逆变器输出的三相电流相位超前过渡电流相位β/2，第二个逆变器输出的三相电流相位滞后过渡电流相位β/2。
进一步地，第三区的升功率运行是：与第三区运行点P 3(n p3,T p3)转速相同的过渡点P 3’(n p3,T p3’)，当满足条件
Figure PCTCN2022128230-appb-000005
时形成集合S P3，集合S P3中属于第三区或第四区且转矩大于或等于T P3的运行点为升功率运行点H 3(n H3,T H3)，仿真获得电机运行于升 功率运行点H 3(n H3,T H3)时的逆变器输出的三相电流，将多余能量回馈至电池；第五区的升功率运行与第三区的升功率运行雷同；n p3、T p3和η p3分别为运行点P 3(n p3,T p3)的转速、转矩和效率，T p3’和η p3’分别为过渡点的转矩和效率，η pg为发电效率。
进一步地，在第七区，与第七区运行点转速相等有两个过渡点，两个过渡点转矩之和等于第七区运行点转矩，计算出所述的两个过渡点的系统消耗总功率之和，当有两个运行点的系统消耗总功率之和等于所述的两个过渡点系统消耗总功率之和的最小值时，则所述的两个运行点分别由对应的两个控制模块控制运行。
本发明采用上述技术方案后具备以下有益效果：
1.本发明采用了多单元设计的概念进行单元化电机结构设计，空间上不重合的单元化设计使所有电机单元相互独立，每个电机单元可独立运行、独立控制，大大提高了电机的运行与控制的自由度，其多单元的不同组合形式与工作模式使得永磁轮毂电机具备多工况运行的能力。
2.本发明采用了多极少槽的结构，解放了传统单元化电机对极槽配比的高要求，为单元化电机提供了新的极槽配比，提高了转子极对数，使得永磁轮毂电机获得了低速大转矩的性能特点，更加满足轮毂电机的性能需求。
3.本发明中的转子在轴向上采用分段斜极结构，有利于调整转子转矩谐波相位分布，实现第一段转子转矩谐波相位与第二段转子转矩谐波相位相互补偿，消除输出转矩最高幅值谐波，从而大幅度降低电机转矩脉动，提高转矩品质。
4.本发明中的弧形永磁磁钢靠近气隙侧长边采用正弦曲线设计，一方面有利于调整电机凸极率，提高电机弱磁能力；另一方面，提高永磁磁势正弦度，降低永磁磁场谐波复杂度，有利于降低电机铁芯损耗，在提高电机运行效率的同时降低电机转矩脉动，提高电机输出性能综合质量。
5.本发明中的矩形永磁磁钢靠近气隙端设置有漏磁磁路，使得随着q轴电流的变化，电机永磁磁场能够实现“轻载多漏磁，重载无漏磁”，即电机重载时，永磁磁通均为有效磁通，有利于提高电机转矩输出能力，实现电机重载工况运行；电机高速时，q轴电流减少，永磁磁场减弱，有利于拓宽电机转速范围，实现电机高速运行。
6.本发明提出的多单元协同控制系统，相互独立的绕组电子开关实现电机单元的独立控制，提高电机控制自由度的同时，提高电机容错率；相互独立的电源电子开关解耦了电机两个控制模块，提高电机控制自由度，为实现电机高性能运行提供硬件基础的同时，提高了电机轻载区域的运行可靠性。
7.本发明提出当电机工作于第一区低速负载区时，两个逆变器输出同频同幅值不同相位的电流，两个模块的电机单元工作在同一运行点，此时两个控制器对应的输出转矩主要谐波相位差为180°，两者相互补偿，从而降低电机转矩脉动。
8.本发明提出当电机工作于第三区高速低载区时，第一电源电子开关与第二电源电子开关有且仅有一个闭合的同时所有绕组电子开关闭合，对应所控制的电机单元升功率运行，此时，电机运行效率提高，且多余能量通过整流器回馈至电池，有利于提高电机系统效率。
9.本发明提出当电机工作于第五区高速负载区时，所有电源电子开关均闭合的同时所有绕组电子开关闭合，电机单元升功率运行，此时，电机运行效率提高，且多余能量通过整流器反馈至电池，有利于提高电机和控制系统效率。
10.本发明提出当电机工作于第七区高速过载区时，所有电源电子开关均闭合的同时所有绕组电子开关闭合，两个模块的单元电机工作在不同运行点，输出转矩代数叠加的同时，损耗也代数叠加，协同工作模式下，由于所有电机单元均工作在高效区，电机总损耗降低，因此提高了电机运行效率。
附图说明
图1为本发明多极少槽单元化永磁轮毂电机的径向结构及单元化分解示意图；
图2为图1中一个电机单元1的结构放大图；
图3为本发明多极少槽单元化永磁轮毂电机的两段转子轴向安装结构示意图；
图4为图3中的径向视图；
图5为图4中机械错位角α的计算流程图；
图6为图3中两段转子输出转矩图；
图7为图6中两段转子转矩合成图；
图8为图2中的转子结构放大图及永磁体充磁方式标注图；
图9为图8中一个永磁磁钢组的结构放大图及几何尺寸标注图；
图10为图9中弧形永磁磁钢的结构放大图及其几何尺寸标注图；
图11为图8中第一矩形永磁磁钢、内磁障与外磁障的结构放大图及其几何尺寸标注图；
图12为图8中三个永磁磁钢组的结构放大图及几何尺寸标注图；
图13为图12中转子虚拟槽的形成结构放大示意图；
图14为图2中的定子结构放大图；
图15为本发明多极少槽单元化永磁轮毂电机在轻载时的工作磁路示意图；
图16为本发明多极少槽单元化永磁轮毂电机在重载时的工作磁路示意图；
图17为本发明多极少槽单元化永磁轮毂电机的多单元协同控制系统结构框图；
图18为多单元协同控制系统工作时根据临界转速划分的恒转矩区和恒功率区的曲线图；
图19为对图18中的恒转矩区划分出的两个子区域分布图；
图20为对图18中的恒功率区划分出的八个子区域分布图。
图1中：1.电机单元；2.外转子；3.内定子；4.集中式绕组；5.转轴；2.1.第一转子段；2.2.第二转子段；2.3.转子铁芯；2.4.永磁磁钢组；2.5.内磁障；2.6.外磁障；2.7.虚拟槽；2.4.1.第一矩形永磁磁钢；2.4.2.第二矩形永磁磁钢；2.4.3.弧形永磁磁钢；2.4.3.1.弧形永磁磁钢外长边；2.4.3.2.弧形永磁磁钢内长边；2.4.3.3.弧形永磁磁钢短边；2.5.1.内磁障第一边；2.5.2.内磁障第二边；2.5.3.内磁障第三边；2.5.4.内磁障第四边；2.5.5.内磁障第五边；2.6.1.外磁障第一边；2.6.2.外磁障第二边；2.6.3.外磁障第三边；2.6.4.外磁障第四边；2.6.5.外磁障第五边；3.1.导磁环；3.2.定子凸极；3.3.定子极靴。
具体实施方式
参见图1和图2所示，本发明多极少槽单元化永磁轮毂电机由沿径向截面圆周方向均匀分布的N个完全相同的电机单元1组成，为保证N个电机单元1可以被两个控制模块独立操控，电机单元1的数量N应当满足N＝2i，i为正整数。每个电机单元1在径向截面上所对应的圆心角β N＝2π/N。
每个电机单元1由沿径向截面圆周方向上1/N的外转子2、1/N的内定子3、1/N的集中式绕组4以及1/N的转轴5组成，因此，N个电机单元1组成的永磁轮毂电机由一个外转子2、一个内定子3、一套集中式绕组4以及一个转轴5组成。内定子3与外转子2同轴心，套在外转子2内部，内定子3的中心用于安放转轴5，内定子3上绕有集中式绕组4。外转子2内壁与内定子3外壁之间具有气隙，气隙的厚度与电机的功率等级、所选取的永磁材料以及外转子2、内定子3加工和装配工艺有关，外转子2与内定子3都是由0.35mm厚度的硅钢片叠压而成，叠压系数为0.95。
参见图3与图4，每个电机单元1的1/N的外转子2，沿轴向上划分为完全相同的M段，形成M个转子段，依次为第一转子段、第二转子段，……，第M转子段。考虑到降低转矩脉动的同时降低外转子2的加工难度，M数量应当满足：20mm≤l ef/M≤120mm，其中l ef为电机轴向长度。M个转子段沿同一旋转方向依次地旋转一个机械错位角度α布置安装，相邻两个转子段相差一个机械错位角度α。图3中，以M＝2为例，仅示出了两个转子段，即第一转子段2.1和第二转子段2.2。如图4所示，第一转子段2.1和第二转子段2.2交错α角度，机械错位角度α为图3中的第一转子转2.1相对于第二转子段2.2逆时针旋转的α角 度。
为降低单元化永磁轮毂电机的转矩脉动，机械错位角α按以下方法确定，如图5所示：
步骤1：赋值初始机械错位角α 0为0。
步骤2：运用有限元软件，仿真获得机械错位角为α 0时的a电机输出转矩T(t)波形，计算机械错位角为α 0时的电机转矩脉动，该电机转矩脉动为初始转矩脉动T rip0。其中，初始转矩脉动T rip0计算方法为：先计算出输出转矩T(t)平均值，将输出转矩T(t)的最大值与最小值作差，该差值占输出转矩T(t)平均值的百分比即是初始转矩脉动T rip0。
步骤3：对所述的电机输出转矩T(t)波形进行快速傅里叶分解，求取最高幅值谐波分量的谐波次数k。电机输出转矩T(t)波形经快速傅里叶分解为直流分量T 0、最高幅值谐波分量T kcos(kwt+θ k)以及剩余谐波分量
Figure PCTCN2022128230-appb-000006
之和。其中，T k为最高幅值谐波分量幅值，θ k为相位；s为剩余谐波分量次数，其幅值为T s，相位为θ s。因此，电机输出转矩T(t)的快速傅里叶分解表达式为：
Figure PCTCN2022128230-appb-000007
其中，t为时间，w为电机输出转矩T(t)的旋转速度。
步骤4：基于步骤3中所获得的最高幅值谐波分量的谐波次数k，根据下式计算出过渡机械错位角
Figure PCTCN2022128230-appb-000008
其中，P r为电机转子极对数，M是转子段数量。
步骤5：运用有限元软件仿真机械错位角为过渡机械错位角α 1时电机的输出转矩波形，计算此时电机转矩脉动，即获得过渡转矩脉动T rip1。过渡转矩脉动T rip1的计算方法和步骤2中的初始转矩脉动T rip0的计算方法雷同。
步骤6：将过渡转矩脉动T rip1和步骤2中的初始转矩脉动T rip0作比较，判断出过渡机械错位角α 1是否能有效减小转矩脉动。若过渡转矩脉动T rip1小于初始转矩脉动T rip0，则判断过渡机械错位角α 1能有效减小转矩脉动，则执行步骤9；反之，若过渡转矩脉动T rip1大于或等于初始转矩脉动T rip0，则判断过渡机械错位角α 1不能有效减小转矩脉动，则执行步骤7。
步骤7：将过渡机械错位角α 1赋值到初始机械错位角α 0。
步骤8：基于步骤7的赋值后的初始机械错位角α 0，运用有限元软件，仿真获得机械错位角为赋值后的初始机械错位角α 0时的电机输出转矩T(t)波形；然后循环步骤3-步骤6，直至在步骤6中过渡转矩脉动T rip1小于初始转矩脉动T rip0，判断结果为能有效减小转矩脉动执行步骤9为止。
步骤9：将过渡机械错位α 1赋值给机械错位角α并输出。
此时，第一转子段2.1产生的输出转矩T 2.1(t)为直流分量
Figure PCTCN2022128230-appb-000009
最高幅值谐波分量
Figure PCTCN2022128230-appb-000010
以及剩余谐波分量
Figure PCTCN2022128230-appb-000011
之和，表达式为：
Figure PCTCN2022128230-appb-000012
第二转子段2.2相对于第一转子段2.1，相位θ k和θ s发生改变，第二转子段2.2产生的输出转矩T 2.2(t)为直流分量
Figure PCTCN2022128230-appb-000013
最高幅值谐波分量
Figure PCTCN2022128230-appb-000014
以及剩余谐波分量
Figure PCTCN2022128230-appb-000015
之和，表达式为：
Figure PCTCN2022128230-appb-000016
依次类推，相位θ k和θ s依次发生改变，M个转子段的第j段转子段2.j，j为正整数且j≤M，第j段转子段2.j产生的输出转矩T 2.j(t)为直流分量
Figure PCTCN2022128230-appb-000017
最高幅值谐波分量
Figure PCTCN2022128230-appb-000018
以及剩余谐波分量
Figure PCTCN2022128230-appb-000019
之和，表达式为：
Figure PCTCN2022128230-appb-000020
由此可知，对于M个转子段，其输出转矩波形中的k次谐波的相位，依次相差
Figure PCTCN2022128230-appb-000021
也就是说，当所有转子段输出转矩波形合成后，k次谐波将会形成补偿，其合成谐波的幅值为0。即电机磁势的输出转矩T’(t)为直流分量
Figure PCTCN2022128230-appb-000022
与剩余谐波分量
Figure PCTCN2022128230-appb-000023
之和，表达式为：
Figure PCTCN2022128230-appb-000024
由于k为最高幅值谐波分量的谐波次数，因此随着k次谐波的消失，转矩脉动将会降低。
参见图6，以M＝2为例，展示转子引入机械错位角α的效果。横坐标为转子位置，单位为电角度；纵坐标为齿槽转矩，单位为Nm。图6中点线为第一转子段2.1的齿槽转矩波形，虚线为第二转子段2.2的齿槽转矩波形。当转子位置在电角度0-360°变化时，第一转子段2.1和第二转子段2.2的齿槽转矩均在[-6.4,6.1]之间均匀震荡，且在第一转子段2.1的齿槽转矩取最大值时，第二转子段2.2的齿槽转矩取最小值，反之亦然。由此可知，第一转子段2.1的输出转矩与第二转子段2.2的输出转矩实现了峰谷补偿，这有利于消除最高幅值谐波分量，降低电机转矩脉动，提升电机转矩品质。
参见图7，其横坐标为转子位置，单位为电角度；纵坐标为齿槽转矩，单位为Nm。图7中实线曲线为第一转子段2.1输出转矩与第二转子段2.2输出转矩复合后的实际转子转矩曲线。当转子位置在电角度0-360°变化时，转子转矩在[-1.1,1]之间均匀震荡。相较于图6中的转矩曲线，图7中的转矩曲线的峰值大幅度降低，因此，本发明采用的机械错位角α能大幅度降低电机的转矩脉动，提高转矩品质。
参见图8，永磁轮毂电机的外转子2由一个转子铁芯2.3、2a个永磁磁钢组2.4、4a个 内磁障2.5、4a个外磁障2.6以及4a个虚拟槽2.7构成，其中a为正整数。2a个永磁磁钢组2.4沿圆周方向均匀分布在整个转子铁芯2.3上。
参见图9，每个永磁磁钢组2.4均由一个第一矩形永磁磁钢2.4.1、一个第二矩形永磁磁钢2.4.2以及一个弧形永磁磁钢2.4.3构成。2a个弧形永磁磁钢2.4.3嵌于外转子2的内表面的内部，2a个弧形永磁磁钢2.4.3沿圆周方向均匀布置，每个弧形永磁磁钢2.4.3在径向截面上的中心线与直径方向一致，永磁磁钢组2.4中心线与弧形永磁磁钢2.4.3的中心线重合一致。第一矩形永磁磁钢2.4.1与第二矩形永磁磁钢2.4.2的结构相同，径向截面均呈矩形，均位于弧形永磁磁钢2.4.3的外侧，属于同一个永磁磁钢组2.4中的第一矩形永磁磁钢2.4.1与第二矩形永磁磁钢2.4.2相对于永磁磁钢组2.4中心线对称放置，第一矩形永磁磁钢2.4.1与第二矩形永磁磁钢2.4.2也相对于弧形永磁磁钢2.4.3的中心对称放置。属于同一个永磁磁钢组2.4中的第一矩形永磁磁钢2.4.1与第二矩形永磁磁钢2.4.2呈开口朝向气隙侧的“V”型布置方式，内外斜向为矩形的长度方向。弧形永磁磁钢2.4.3放置在第一矩形永磁磁钢2.4.1与第二矩形永磁磁钢2.4.2形成的V形开口中间，三个永磁磁钢相互之间不接触。
如图8，属于同一个永磁磁钢组2.4中的第一矩形永磁磁钢2.4.1与第二矩形永磁磁钢2.4.2的充磁方向垂直于磁钢自身的长边，与自身宽度方向一致，弧形永磁磁钢2.4.3充磁方向和中心线方向一致，指向或背离圆心。属于同一个永磁磁钢组2.4的第一矩形永磁磁钢2.4.1、第二矩形永磁磁钢2.4.2以及弧形永磁体2.4.3充磁方向同时指向或背离气隙，相邻两个永磁磁钢组2.4的充磁方向相反。
第一矩形永磁磁钢2.4.1与第二矩形永磁磁钢2.4.2靠近气隙侧和远离气隙侧的端部各设有一个磁障，即矩形永磁磁钢的内端部和外端部各设有一个磁障。分别是：每个第一矩形永磁磁钢2.4.1与第二矩形永磁磁钢2.4.2的内端部各具有一个内磁障2.5，因此，共有4a个内磁障2.5；每个第一矩形永磁磁钢2.4.1与第二矩形永磁磁钢2.4.2的外端部各设有一个外磁障2.6，因此，共有4a个外磁障2.6。沿外转子2的内侧面上，在每个弧形永磁磁钢2.4的沿切线方向的两端各设有一个虚拟槽2.7，因此，共有4a个虚拟槽2.7，虚拟槽2.7和气隙相通，与气隙成为一体，成为气隙的一部分。
参见图9，为了兼顾永磁轮毂电机弱磁扩速能力与电机峰值转矩输出能力，第一矩形永磁磁钢2.4.1与第二矩形永磁磁钢2.4.2之间形成的V型夹角β pm应当满足：40°≤β pm≤65°。另外，为坚固第一矩形永磁磁钢2.4.1与第二矩形永磁磁钢2.4.2强度以及加工工艺难度与应力分布情况，第一矩形永磁磁钢2.4.1与第二矩形永磁磁钢2.4.2的长边长度w pm与宽边长度h pm应当满足：2≤w pm/h pm≤4。
参见图10，每个弧形永磁磁钢2.4.3均由一条弧形永磁磁钢外长边2.4.3.1，一条弧形永磁磁钢内长边2.4.3.2以及两条沿弧形永磁磁钢2.4.3中心对称的弧形永磁磁钢短边2.4.3.3围成。弧形永磁磁钢外长边2.4.3.1的弧形中心与外转子2中心相同。弧形永磁磁钢内长边2.4.3.2所对应的圆心角与弧形永磁磁钢外长边2.4.3.1所对应的圆心角相同。弧形永磁磁钢短边2.4.3.3和外转子2的直径方向一致，在外转子2的一条半径上。弧形永磁磁钢内长边2.4.3.2为半个周期的正弦曲线，正弦曲线的自变量θ 1范围为[π,2π]，即弧形永磁磁钢内长边2.4.3.2的正弦曲线函数为：
f 1(θ 1)＝f 1maxsin(θ 1)，θ 1∈[π,2π]，
其中f 1max为正弦曲线幅值，由电机具体性能要求决定。当θ 1为3π/2时，对应的正弦曲线上的点f 1(3π/2)正好位于外转子2的内表面上，因此，弧形永磁磁钢2.4.3整体嵌于外转子2的内表面这一侧的内部，在θ 1为3π/2时的点f 1(3π/2)与外转子2的内表面相重合。
弧形永磁磁钢内长边2.4.3.2的正弦曲线形状的设计，改变了弧形永磁磁钢2.4.3的磁势波形。弧形永磁磁钢2.4.3的磁势波形由原本的方波向矩形波与正弦的叠加波形转变，这改变了弧形永磁磁钢2.4.3的磁势谐波分布，提高弧形永磁磁钢2.4.3的磁势波形的正弦性， 使得磁势基波幅值提高，有利于电机转矩输出能力的提升。同时，这一正弦曲线形状的设计还降低了永磁磁势基波幅值，有利于电机铁芯损耗的降低。
参见图10，为增强电机永磁磁场的同时，减少主磁路磁阻以提高电机峰值转矩，弧形永磁磁钢沿径向上的最小宽度h pmin应当满足：6≤h pm/h pmin≤8。此外，为减小永磁磁场谐波降低电机铁耗的同时，减小电机平均气隙宽度，降低主磁路磁阻，弧形永磁磁钢2.4.3的最小宽度h pmin与弧形永磁磁钢2.4.3的最大宽度h pmax应当满足：1.5≤h pmax/h pmin≤2。
参见图11与图12，第一矩形永磁磁钢2.4.1与第二矩形永磁磁钢2.4.2靠近气隙的端部分别设置的内磁障2.5结构相同且沿永磁磁钢组2.4的中心线对称分布。以第一矩形永磁磁钢2.4.1端部的内磁障2.5为例说明：内磁障2.5的径向截面为“五边形”，由内磁障第一边2.5.1、内磁障第二边2.5.2、内磁障第三边2.5.3、内磁障第四边2.5.4以及内磁障第五边2.5.5围成。内磁障第一边2.5.1为第一矩形永磁磁钢2.4.1靠近气隙侧长边的延长边，内磁障第二边2.5.2为与电机外转子2同轴心的圆弧边。为在构建永磁磁场漏磁磁路，拓宽电机调速范围和提高电机高速区域运行效率的同时控制电机漏磁程度，提高电机低速轻载区域转矩输出能力，内磁障第二边2.5.2与外转子2的内表面之间的距离h b1应当满足：0.75≤h b1/h pm≤0.9。内磁障第三边2.5.3位于外转子2的半径上。内磁障第四边2.5.4平行于内磁障第一边2.5.1，且位于内磁障第一边2.5.1外侧，即远离气隙侧。为确保降低第一矩形永磁磁钢2.4.1和第二矩形永磁磁钢2.4.2不可控漏磁的同时保证其安装的可靠性，内磁障第一边2.5.1与内磁障第四边2.5.4之间的距离h b2小于第一矩形永磁磁钢2.4.1和第二矩形永磁磁钢2.4.2的宽度，应当满足：0.8h pm≤h b2≤0.9h pm。内磁障第五边2.5.5与第一矩形永磁磁钢2.4.1与第二矩形永磁磁钢2.4.2靠近气隙侧的短边重合。
第一矩形永磁磁钢2.4.1与第二矩形永磁磁钢2.4.2远离气隙的端部分别设置外磁障2.6，两个外磁障2.6结构相同且沿永磁磁钢组2.4的中心线对称分布。以第一矩形永磁磁钢2.4.1端部的外磁障2.6为例说明：外磁障2.6的径向截面也是“五边形”，由外磁障第一边2.6.1、外磁障第二边2.6.2、外磁障第三边2.6.3、外磁障第四边2.6.4以及外磁障第五边2.5.5围成。外磁障第一边2.6.1位于第一矩形永磁磁钢2.4.1靠近气隙侧长边的延长线上，外磁障第二边2.6.2位于外转子2的半径上。外磁障第三边2.6.3为一条与外转子2同轴心的圆弧边。为了在降低第一矩形永磁磁钢2.4.1和第二矩形永磁磁钢2.4.2不可控漏磁的同时降低q轴磁阻，实现转矩最大化，外磁障第三边2.6.3与外转子2的外表面之间的距离h b3应当满足：0.2(R ro-R ri)≤h b3≤0.35(R ro-R ri)，其中R ro为外转子2的外径，R ri为外转子2的内径。外磁障第四边2.6.4平行于外磁障第一边2.6.1，且位于外磁障第一边2.6.1外侧，即远离气隙侧。为降低第一矩形永磁磁钢2.4.1和第二矩形永磁磁钢2.4.2漏磁的同时保证其安装可靠性，外磁障第一边2.6.1与外磁障第四边2.6.4之间的距离h b4小于第一矩形永磁磁钢2.4.1和第二矩形永磁磁钢2.4.2的宽度，应当满足：0.8h pm≤h b4＝h b1≤0.9h pm。外磁障第五边2.5.5与第一矩形永磁磁钢2.4.1与第二矩形永磁磁钢2.4.2远离气隙侧的短边重合
参见图13，为提高气隙磁密正弦度，降低电机铁耗与转矩脉动，在弧形永磁磁钢2.4.3的两端沿弧形永磁磁钢2.4.3中心线对称设有两个虚拟槽2.7。每个虚拟槽2.7为一段正弦曲线，在径向截面上，虚拟槽2.7与弧形永磁磁钢2.4.3的弧形永磁磁钢内长边2.4.3.2的端部交点为点A，也是弧形永磁磁钢内长边2.4.3.2与弧形永磁磁钢短边2.4.3.3的交点。内磁障第二边2.5.2与内磁障第三边2.5.3交点为点B，过点B的一条半径与外转子2内表面的交点为点C。则由点A至点C是一段正弦曲线，形成一个虚拟槽2.7。虚拟槽2.7的这段正弦曲线，其自变量θ 2范围为[π/2,π]，虚拟槽2.7正弦曲线函数为：
Figure PCTCN2022128230-appb-000025
其中，f 2max为该函数幅值，由电机具体性能要求决定。当θ 2为π/2时，在正弦曲线上的点f 2(π/2)与点A重合，当θ 2为π时，在正弦曲线上的点f 2(π)与点C重合。
虚拟槽2.7能改变外转子2的磁导波形与永磁磁场的气隙磁势波形，磁导由原本的方波向方波与正弦的叠加波形转变，同时实现永磁磁钢组2.4的永磁磁势波形的削极处理，提高弧形永磁磁钢2.4.3的磁势波形的正弦性，使得磁势基波幅值提高，有利于电机转矩输出能力的提升，同时，虚拟槽2.7的设计降低了永磁磁势基波幅值，有利于电机铁芯损耗的降低。
参见图14，永磁轮毂电机的内定子3由一个导磁环3.1、B个定子凸极3.2以及2B个定子极靴3.3组成，其中B＝mj，m为电机相数，此处的j为正整数。每个定子凸极3.2的外端向切向两边延伸一个定子极靴3.3。定子凸极3.2在径向截面为矩形，沿导磁环3.1外表面圆周方向均匀分布。同时，在定子凸极3.2外端部的切向两边分别设有完全相同的一个定子极靴3.3。导磁环3.1的径向宽度、定子凸极3.2的径向长度与切向宽度，以及定子极靴3.3的切向宽度的大小由电机功率大小决定。
本发明所述的多极少槽单元化永磁轮毂电机整机的转子极对数P r，定子槽数N s，电机相数m，电机单元1的数量N应当满足以下条件：(1)多极少槽：转子极对数P r大于定子槽数N s；电机槽距角τ为转子极对数P r除以定子槽数N s之商与2π的乘积。(2)转子单元化：单元化电机要求转子可以分为N份，即转子极对数P r应当为电机单元1的数量N的整数倍；(3)定子单元化：要求定子可以分为N份，则定子槽数N s应该是相数m的整数倍，同时为电机单元1的数量N的整数倍；(4)控制模块单元化：为提高单元化电机控制的自由度，提高电机整体运行效率与运行性能，本发明所述的多极少槽单元化永磁轮毂电机配备了两个控制模块。则要求电机单元1的数量N为2的整数倍；(5)集中式绕组：绕组为集中式绕组4，即绕组跨距应当为1；(6)绕组单元化：在定子单元化的基础上，需要满足每个电机单元1中的绕组均为三相对称且分布相同，即存在正整数i,a,b,c,d,e使如下方程组同时成立：
Figure PCTCN2022128230-appb-000026
针对本发明所述的多极少槽单元化永磁轮毂电机，定义其轻载运行的条件为电机输出转矩T满足：T≤0.6T rated，其中T rated为电机额定输出转矩；重载运行条件为电机输出转矩T满足：T＞0.6T rated。并且定义低速运行的条件为电机转速n满足条件：n≤n rated，其中n rated为电机额定转速；高速运行的条件为电机转速n满足条件：n＞n rated。
参见图15，当本发明所述的多极少槽单元化永磁轮毂电机在轻载运行时，此时电机中的两条磁路是主磁路I与漏磁路II，并且主磁路I与漏磁路II并联。主磁路I的磁通路径如下：从第一个永磁磁钢组2.4中的第一矩形永磁磁钢2.4.1出发，依次经过转子铁芯2.3、第一个弧形永磁磁钢2.4.3、气隙、内定子3、气隙、与第一个弧形永磁磁钢2.4.3相邻的第二个弧形永磁磁钢2.4.3、转子铁芯2.3、与第一个永磁磁钢组2.4相邻的第二个永磁磁钢组2.4中的第二矩形永磁磁钢2.4.2、转子铁芯2.3，最终回到第一矩形永磁磁钢2.4.1，形成闭合磁路。由于漏磁磁路的存在，电机中存在区别与传统磁路的漏磁路II，漏磁路II的磁通路径为：从第一个永磁磁钢组2.4中的第一矩形永磁磁钢2.4.1出发，依次经过转子铁芯2.3、与第一个弧形永磁磁钢2.4.3相邻的第二个弧形永磁磁钢2.4.3、转子铁芯2.3，与第一个永 磁磁钢组2.4相邻的第二个永磁磁钢组2.4中的第二矩形永磁磁钢2.4.2、转子铁芯2.3，最终回到第一矩形永磁磁钢2.4.1，在外转子2中形成闭合磁路。由此可知主磁路I与漏磁路II并联。
参见图16，当本发明所述的多极少槽单元化永磁轮毂电机在重载运行时，电机中存在主磁路I与q轴磁路III这两条磁路，主磁路I与q轴磁路III并联运行。主磁路I与图15中的轻载运行时的主磁路I相同，q轴磁路III的磁通路径为：从内定子3出发，依次经过气隙、弧形永磁磁钢2.4.3、转子铁芯2.3、相邻的弧形永磁磁钢2.4.3、气隙、最后回到内定子3，形成闭合磁路。由此可知主磁路I与q轴磁路III并联。
结合图15与图16可以看出，漏磁路II与q轴磁路III存在部分路段重合且重合路段磁阻较大，极易饱和。当本发明所述的多极少槽单元化永磁轮毂电机在低速轻载运行时，电机转速与转矩均较低，因此集中式绕组4输出的电流幅值I较小，则q轴电流分量i q较小。也就是q轴磁路III磁通较弱，漏磁路II与q轴磁路III重合路段被漏磁路II的磁通饱和。当电机在高速轻载运行时，由于此时电机转矩较高，集中式绕组4输出电流幅值I较大的同时其电流角较大，q轴电流分量i q较小，其磁路分布与低速轻载运行时相同。同时，由于q轴磁路III磁通较弱，漏磁路II磁通较强，电机励磁磁场减弱，这一现象使得相对于传统电机来说，本发明电机在高速区域铜耗更低。此外，当本发明电机在重载运行时，随着q轴电流分量i q的增大，电机q轴磁路III磁通增强，漏磁路II磁通会逐渐减弱直至消失。此时，永磁磁钢组2.4所有磁通均经由主磁路I形成有效磁通，进而提高电机转矩输出能力。
参见图17，针对本发明所述的多极少槽单元化永磁轮毂电机，采用协同控制系统对其进行控制。该协同控制系统包括一个电池、两个控制模块和N个绕组电子开关，其中，一个绕组电子开关连接一个电机单元1中的集中式绕组4，以控制一个电机单元1中集中式绕组4的通断。每个控制模块均由一个电源电子开关，一个DSP控制器以及一个逆变器依次串联组成，两个电源电子开关的输入端均分别连接电池的输出端，每个逆变器的输出端分别连接N/2个绕组电子开关，即第一控制模块由第一电源电子开关，第一DSP控制器以及第一逆变器依次串联组成，第二控制模块由第二电源电子开关，第二DSP控制器以及第二逆变器依次串联组成。两个控制模块中的电源电子开关相互独立，两个控制模块相互独立，结构相同，各控制N/2个电机单元1，降低了两个模块之间的耦合，提高了多极少槽单元化永磁轮毂电机控制的自由度与控制品质。
N个绕组电子开关分为相互独立的两组，每组绕组电子开关控制N/2个电机单元1。N个绕组电子开关分为两组分别连接到两个控制模块上。如第一绕组电子开关与第一个电机单元1相连接，第二绕组电子开关与第二个电机单元1相连接，第N/2绕组电子开关与第N/2个电机单元1相连接，第N/2+1绕组电子开关与第N/2+1个电机单元1相连接，第N绕组电子开关与第N个电机单元1相连接。将第一绕组电子开关直至第N/2绕组电子开关共N/2个绕组电子开关的输入端分别与第一个控制模块中的第一逆变器相连接。将第N/2+1绕组电子开关直至第N绕组电子开关共N/2个绕组电子开关的输入端分别与第二个控制模块中的第二逆变器相连接。这样，每个电机单元1都可通过相应的绕组电子开关独立控制通断，电机总电枢磁场强度可实现多级变换，有利于调节电机磁场，实现电机多工况运行。对相互独立的电机单元1的控制能进一步提高多极少槽单元化永磁轮毂电机的运行自由度，为提高电机及其协同控制系统的工作效率提供了硬件基础。
另外，所有电机单元1的集中式绕组4输出端都经回馈模块连接电池，回馈模块主要部件为整流器，回馈模块针对电机在升功率运行模式下，电机运行所产生的能量高于轮毂驱动所需要的能量时，剩余能量通过回馈模块回收到电池中。
协同控制系统采用以下控制策略对本发明所提出的多极少槽单元化轮毂永磁电机实施控制：
步骤1：运用有限元软件仿真获得控制模块工作时电机的外特性曲线。
将两个电源电子开关与N个绕组电子开关均闭合，采用有限元软件仿真获得电机的外 特性曲线g，如图18所示。保持N个绕组电子开关均闭合，使第一电源电子开关与第二电源电子开关有且仅有一个闭合，即断开其中一个电源电子开关，只有一个控制模块工作，采用有限元软件，仿真获得此时电机的外特性曲线f；横坐标为电机转速n，单位为rpm；纵坐标为电机输出转矩T，单位为Nm。
步骤2：基于获得的外特性曲线，划分电机的恒转矩区与恒功率区。
步骤2.1：首先计算临界转速n b。从外特性曲线f得到最高转矩T max，临界转速n b为最高转矩T max对应的最高转速，也就是：
Figure PCTCN2022128230-appb-000027
其中，n i为电机输出峰值转矩下工作点的转速集合。
步骤2.2：根据所述的临界转速n b，划分恒转矩区与恒功率区。设点P(n p,T p)为电机任一运行点，n p为该点转速，T p为该点转矩。当该点转速n p满足n p≤n b时，则该点属于恒转矩区，即恒转矩区电机运行点的转速n p≤n b；若该点转速n p满足n p＞n b时，该点属于恒功率区。如图18中的垂直的虚线h即为恒转矩区与恒功率区的分界线，在分界线处的转速n p＝n b。
步骤3：再针对恒转矩区分区，根据临界转矩分出第一区低速负载区和第二区低速过载区。
步骤3.1：基于有限元软件仿真，计算出一个电机单元1的峰值转矩T x。在第一电源开关与第二电源电子开关均闭合的前提下，也就是第一控制模块和第二控制模块均工作，有且仅有任意一个绕组电子开关闭合，用有限元软件仿真此时的电机峰值转矩，该转矩为一个电机单元1的峰值转矩T x；
根据一个电机单元1的峰值转矩T x计算出临界转矩T b。临界转矩T b为一个电机单元1的峰值转矩T x的N-2倍，也就是：
T b＝(N-2)T x。
步骤3.3：参见图19，横坐标为电机转速，单位为rpm；纵坐标为电机输出转矩，单位为Nm。基于所述的临界转矩T b，确定出两个区，分别是第一区低速负载区和第二区低速过载区。若运行点P(n p,T p)满足条件：T p≤T b，n p≤n b时，则此时运行点P(n p,T p)属于第一区低速负载区，即在恒转矩区内，如果运行点P(n p,T p)的转矩T p等于或者小于临界转矩T b，则运行点P(n p,T p)属于第一区低速负载区S 1；反之，P(n p,T p)满足条件：T p＞T b，n p≤n b时，则此时点P属于第二区低速过载区S 2。第二区低速过载区S 2的运行点P(n p,T p)的转矩T p最大值不超过外特性曲线g的转矩。恒转矩区的分化结果为：
Figure PCTCN2022128230-appb-000028
图19中的水平直线j为第一区低速负载区与第二区低速过载区的分界线，其函数表达式为：
T(n)＝T b,n∈[0,n b]。
恒转矩区所分的两个子域用不同标记展现在图19中：网格区域为第一区低速负载区，左斜线区域为第二区低速低载区。
步骤4：基于步骤3中的恒转矩区的分区结果，确定出第一区低速负载区的控制方法。第一区低速负载区的主要控制原则为转矩脉动最小原则：第一电源电子开关与第二电源电 子开关均闭合，第一逆变器与第二逆变器输出同幅值不同相位的电流，调整两个控制模块的输出转矩相位，使其实现谷峰相叠，消除高次幅值的谐波。对于第一区低速负载区中任意一运行点P 1(n p1,T p1)，n p1为该点转速，T p1为该点转矩，控制方法按以下步骤实现：
步骤4.1：根据电机需求的输出转矩T p1，计算出与每个控制模块相连接的绕组电子开关的闭合个数K，1≤K≤N/2。
由于在运行点P 1(n p1,T p1)，电机需求的输出转矩为T p1，所以单一控制模块的需求转矩输出为T p1/2。又由步骤3.1获得的单一电机单元1的峰值转矩为T x，因此与每个控制模块连接的N/2电机单元1中，至少需要闭合的绕组电子开关个数K为(T p1/2T x)。为保证电机的转矩输出能力，这里每个控制模块中绕组电子开关闭合个数K采用“进一法”取整，也就是：
Figure PCTCN2022128230-appb-000029
步骤4.2：基于4.1的绕组电子开关的闭合个数K，采用有限元软件仿真计算出绕组电子开关的闭合个数为K时的输出转矩T 2(t)时两个逆变器输出的过渡电流。
第一电源电子开关与第二电源电子开关均闭合，在第一绕组电子开关到第N/2绕组电子开关中任意闭合K个开关，且在第(N/2+1)绕组电子开关到第N个绕组电子开关中任意闭合K个开关。此时，两个逆变器输出电流完全相同，利用有限元软件仿真计算出此时电机输出转矩波形T 2(t)和两个逆变器输出的过渡电流，为过渡电流幅值为I max0，过渡电流相位为θ 0的正弦函数，表达式为：
Figure PCTCN2022128230-appb-000030
Figure PCTCN2022128230-appb-000031
其中
Figure PCTCN2022128230-appb-000032
分别为第一逆变器输出的过渡电流；
Figure PCTCN2022128230-appb-000033
分别为第二逆变器输出的过渡电流，n p为运行点P(n p,T p)的转速，P r为转子极对数。
步骤4.3：基于步骤4.2的结果，快速傅里叶分解所获得的输出转矩T 2(t)的波形，求取主要谐波分量的谐波次数r。
输出转矩T 2(t)的波形经快速傅里叶分解为直流分量T 20，主要谐波分量T rcos(rw 2t+θ r)以及剩余谐波分量
Figure PCTCN2022128230-appb-000034
之和，其中，主要谐波分量T rcos(rw 2t+θ r)的幅值为T r，相位为θ r；剩余谐波分量
Figure PCTCN2022128230-appb-000035
次数为v，其幅值为T v，相位为θ v。因此，输出转矩T 2(t)的快速傅里叶分解表达式为：
Figure PCTCN2022128230-appb-000036
其中，w 2为输出转矩T 2(t)的旋转速度。
步骤4.4：基于所述的主要谐波分量T r cos(rw 2t+θ r)的谐波次数r，计算出电流错位角β。电流错位角β为两个逆变器输出的电流相位的角度差，其计算方法为π/2与主要谐波分量的谐波次数r的商，即：
Figure PCTCN2022128230-appb-000037
步骤4.5：基于步骤4.2中的两个逆变器输出的过渡电流
Figure PCTCN2022128230-appb-000038
Figure PCTCN2022128230-appb-000039
以及与步骤4.4计算出的电流错位角β，确定出两个逆变器输出的ABC三相电流。
为了保证电机转矩输出能力，电流幅值I max在过渡电流幅值I max0的基础上增加5％，即即I max＝1.05I max0。第一逆变器输出的ABC三相电流相位超前所述的过度电流
Figure PCTCN2022128230-appb-000040
Figure PCTCN2022128230-appb-000041
相位β/2，第二逆变器输出的ABC三相电流相位滞后所述的过度电流
Figure PCTCN2022128230-appb-000042
Figure PCTCN2022128230-appb-000043
相位β/2，即表达式为：
Figure PCTCN2022128230-appb-000044
Figure PCTCN2022128230-appb-000045
其中，I 1A，I 1B以及I 1C为第一逆变器输出的ABC三相电流；I 2A，I 2B以及I 2C为第二逆变器输出的ABC三相电流，n p为运行点P(n p,T p)的转速，P r为转子极对数。
将三相电流I 1A，I 1B以及I 1C，I 2A，I 2B以及I 2C分别保存在第一、第二DSP控制器中，经第一、第二DSP控制器分别控制第一逆变器与第二逆变器输出三相电流。
步骤5：基于步骤3中的恒转矩区的分区结果，确定第二区低速过载区控制方法。由于第二区低速过载区要求对电机转矩要求较高，因此使所有的电机单元1共同作用，即对于第二区低速过载区中任意一运行点P 2(n p2,T p2)，所有的电源电子开关和所有的绕组电子开关均闭合，然后利用有限元软件仿真计算出电机运行于点P 2(n p2,T p2)时两个逆变器输出的ABC三相电流。
步骤6：在步骤3同时的，基于步骤2中的结果，对恒功率区进行详细分区处理。针 对恒功率区运行工况较为复杂这一现状，将恒转矩区分为6个子区域，分成第三直至第八区。其中，第三区为高速低载区，记作S 3；第四区为高速高效区，记作S 4；第五区为高速负载区，记作S 5；第五区为双倍高效区，记作S 6；第七区为高速过载区，记作S 7；第八区为高速弱磁区，记作S 8。恒功率区的运行点P(n p,T p)转矩的最大值不超过外特性曲线g对应的转矩。
步骤6.1：确定第四区高速高效区范围。
步骤6.1.1：仿真计算单一控制模块运行的效率map图。此时，第一电源电子开关与第二电源电子开关有且仅有一个开关闭合，所有的绕组电子开关均闭合。在此前提下，利用有限元软件仿真获得电机运行的效率map图，根据效率map图确定出单一控制模块运行的电机最大效率η max。
步骤6.1.2：基于单一控制模块运行的电机最大效率η max，计算出边界效率η b：边界效率η b为单一控制模块运行的电机最大效率的95％，即：η b＝0.95η max。
步骤6.1.3：基于边界效率η b确定第四区高速高效区的范围。第四区高速高效区即为边界效率η b所包围的恒功率区范围。即，当恒功率区中的任意运行点P(n p，T p)的效率η p满足条件：η p≥η b，则点P(n p，T p)属于第四区高速高效区，也就是：S 4＝{P(n p,T p)|η p≥η b且n p＞n b}。
第四区高速高效区确定后，便获得第四区高速高效区的速度最高点E和速度最低点D，如图20所示。
步骤6.2：基于步骤6.1的结果，确定第八区高速弱磁区的范围。
步骤6.2.1：第四区高速高效区的速度最高点E不唯一，则取所有速度最高点中的转矩最低点为所需速度最高点E，得到经过所需速度最高点E的一条垂直于横坐标的直线j。
步骤6.2.2：确定第八区高速弱磁区的范围。由横坐标轴、曲线g与直线j所围区域为第八区高速弱磁区，第八区高速弱磁区的转速均高于所需速度最高点E的转速n E。
步骤6.3：基于步骤6.1与步骤6.2中的结果，确定第三区高速低载区范围。
步骤6.3.1：基于步骤6.1中的结果，若第四区高速高效区中速度最低点D(n D,T D)不唯一，其中，n D和T D分别为速度最低点D(n D,T D)的转速和转矩，则取所有速度最低点D(n D,T D)中转矩最低点为所需速度最低点D，得到经过所需速度最低点D的一条垂直于横坐标的直线k。
步骤6.3.2：基于步骤6.3.1中的结果，确定转速小于点D的区域内第三区高速低载区的范围，记作S 31。在恒功率区内，转矩小于点D转矩T D且转速小于点D转速n D的区域属于第三区高速低载区。即当点P(n P,T P)满足条件：n p＜n D且T P＜T D时，点P(n P,T P)归属于第三区高速低载区，也就是：S 31＝{P(n p,T p)|n b＜n p＜n D,0≤T p＜T D}。
步骤6.3.3：确定转速大于点D(n D,T D)且小于点E(n E,T E)的区域内第三区高速低载区的范围，记作S 32。S 32为横坐标轴、直线k、第四区高速高效区下半周边界线(点E和点D之间下半周连接线)与直线j所围成的区域。
步骤6.3.4：基于步骤6.3.2与步骤6.3.3，确定第三区高速低载区的范围。第三区高速低载区为S 31与S 32的并集，也就是：S 3＝S 31∪S 32。
值得注意的是，上述第三区高速低载区S 31与S 32的大小会随着电机特性的改变而改变，对于部分电机可能会出现其中部分集合为空集的情况。
步骤6.4：与步骤6.3同时的，基于步骤6.1中确定的第四区高速高效区，确定第六区双倍高校区。第六区双倍高校区为第四区高速高效区的双倍扩展区域，其扩展方法为转速不变，转矩扩展为原本的两倍；也就是，第六区双倍高校区的转速与第四区高速高效区的 转速相同，第六区双倍高校区的速度最高点G在直线j上，速度最低点F在直线k上；对于第六区双倍高校区的每一个运行点P 6(n P6,T P6)，其转矩均是第四区高速高效区中的相同速度n P6运行点的转矩的两倍。对于第六区双倍高校区中的任意一点P 6(n P6,T P6)均为第四区高速高效区中某一点的双倍拓展点。即当点P(n P,T P)归属于第六区双倍高效区时，点P需满足条件：
Figure PCTCN2022128230-appb-000046
且
Figure PCTCN2022128230-appb-000047
P'为第四区高速高效区某一点，也就是：
Figure PCTCN2022128230-appb-000048
步骤6.5：基于步骤6.1与步骤6.4中的结果，确定第五区高速负载区范围。
步骤6.5.1：转矩小于点F的转矩T F且高于D点的转矩且转速小于点F转速n F的区域属于第五区高速负载区，记作S 51。即当点P(n P,T P)满足条件：n p＜n F且T D≤T P＜T F时，点P(n P,T P)归属于第五区高速负载区，也就是：
S 51＝{P(n p,T p)|n b＜n p＜n F,T D≤T p≤T F}。
步骤6.5.2：确定转速大于点F(n F,T F)且小于点G(n G,T G)的区域内第五区高速负载区的范围，记作S 52。S 52为直线k、直线j、第四区高速高效区上半周边界与第六区双倍高效区下半周边界所构成的区域。
步骤6.5.3：基于步骤6.5.1与步骤6.5.2，确定第五区高速负载区的范围。第五区高速负载区为S 51与S 52的并集，也就是：S 5＝S 51∪S 52。
值得注意的是，上述第五区高速负载区S 51与S 52的大小会随着电机特性的改变而改变，对于部分电机可能会出现S 51与S 52中存在一个或两个为空集的情况。
步骤6.6：基于步骤6.1～步骤6.5中的结果，确定第七区高速过载区范围。第七区范围为恒功率区中不属于第三、四、五、六、八区的范围，即恒功率区的剩余区域为第七区。当点P(n P,T P)满足条件：
Figure PCTCN2022128230-appb-000049
则点P(n P,T P)归属于第七区高速过载区，也就是：
Figure PCTCN2022128230-appb-000050
参见图20，恒功率区所分的6个子区域用不同标记展现在图中。右斜线区域为第三区高速低载区，深灰色区域为第四区高速高效区，斜网格区域为第五区高速负载区，白色区域为第六区双倍高效区，点区域为第七区高速过载区，浅灰色为第八区高速弱磁区。
步骤7：基于步骤6中的恒功率区的分区结果，确定第三区高速低载区控制方法。由于第三区高速低载区的转矩相较于第五区、第六区以及第七区都较低，同时转速也低于第八区，对于第三区高速低载区中任意一点P 3(n p3,T p3)，n p3和T p3分别为第三区高速低载区中任意一点的转速和转矩，第一电源电子开关与第二电源电子开关有且仅有一个闭合的同时所有绕组电子开关闭合。另外，第三区高速低载区的主要控制原则为系统效率最高原则，方法是采用电机升功率运行的方法，即不改变电机转速的同时，提高输出转矩，电机运行于升功率运行点H 3(n H3,T H3)，多余能量回馈至电池，从而提高电机系统效率。对于第三区高速低载区中任意一点P 3(n p3,T p3)，控制方法确定分为以下步骤：
步骤7.1：计算第三区高速低载区中任意一点P 3(n p3,T p3)系统消耗总功率W 3。当电机运行于点P 3(n p3,T p3)时，其效率为η p3，则点P 3(n p3,T p3)系统消耗总功率W 3为转速转矩之积除 以60效率的值的2π倍，即：
Figure PCTCN2022128230-appb-000051
步骤7.2：与步骤7.1同时，计算过渡点P 3’(n p3,T p3’)系统消耗总功率W 3’，过渡点P 3’(n p3,T p3’)为点P 3(n p3,T p3)转速相同的任意一点。当电机运行于过渡点P 3’(n p3’,T p3’)时，其效率为η p3’，发电效率为η pg，则过渡点P 3’(n p3’,T p3’)系统消耗的总功率为此时电机运行消耗的消耗减去回馈至电池的效率，即：
Figure PCTCN2022128230-appb-000052
步骤7.3：基于步骤7.1与7.2，建立升功率运行的数学模型。升功率运行的数学模型为过渡点P 3’系统消耗总功率W 3’减去P 3点系统消耗总功率W 3，即为：
Figure PCTCN2022128230-appb-000053
步骤7.4：基于步骤7.3，确定集合S P3。令升功率运行的数学模型大于0，则由上式
Figure PCTCN2022128230-appb-000054
可以得到下式：
Figure PCTCN2022128230-appb-000055
则所有满足条件
Figure PCTCN2022128230-appb-000056
的过渡点P 3’(n p3,T p3’)的集合为S P3。
需要注意的是，对于部分点P 3(n p3,T p3)，S P3可能为空集，这主要取决于电机功率与电机具体性能。当S P3为空集时，就不采用电机升功率运行方法，而是直接运行。
步骤7.5：基于步骤7.4，在S P3可能为非空集采用电机升功率运行时，再进一步在集合S P3中确定出升功率运行点H 3(n H3,T H3)。
为在电机单元1正常运行范围内最大化提高电机系统效率，升功率运行点H 3(n H3,T H3)需要属于S P3的同时需要属于第三区高速低载区或第四区高速高效区，还需要满足升功率运行点H 3的转矩T H3大于或等于T P3。也就是说升功率运行点H 3(n H3,T H3)需要满足：
Figure PCTCN2022128230-appb-000057
需要注意的是，若S P3中存在符合上述条件的点H 3(n H3,T H3)不唯一，则取满足条件的点中升功率数学模型函数最大值f(T p3')点。
步骤7.6：采用有限元软件仿真获得电机运行于点H 3(n H3,T H3)时的逆变器输出ABC三相电流。
步骤8：与步骤7同时，基于步骤6的分区结果，确定第四区高速高效区的控制方法。由于第四区高速高效区的转矩相较于第五区、第六区以及第七区都较低，同时转速也低于第八区，对于第四区高速高效区中任意一点P 4(n p4,T p4)，第一电源电子开关与第二电源电子开关有且仅有一个闭合的同时所有绕组电子开关闭合。另外，第四区高速高效区的主要控制原则为系统效率最高原则，由于第四区高速高效区运行点效率高于第三区、第五区、第七区以及第八区，因此运行点P 4(n p4,T p4)正常工作。采用有限元软件仿真获得电机运行于P 4(n p4,T p4)时的逆变器输出的ABC三相电流。
步骤9：与步骤8同时，基于步骤6的分区结果，确定第五区高速负载区的控制方法。 由于第五区高速负载区转矩高于第三区与第四区，对于第五区高速负载区中任意一点P 5(n p5,T p5)，所有电源电子开关均闭合的同时所有绕组电子开关闭合。另外，第五区高速负载区的主要控制原则为系统效率最高原则，主要方法是采用电机升功率运行的方法，即不改变电机转速的同时，提高输出转矩，电机运行于升功率运行点H 5(n H5,T H5)多余能量回馈至电池，从而提高电机系统效率。对于第五区高速负载区中任意一点P 5(n p5,T p5)，控制方法为以下步骤：
步骤9.1：计算点P 5(n p5,T p5)系统消耗总功率W 5。当电机运行于点P 5(n p5,T p5)时，效率为η p5，则点P 5(n p5,T p5)系统消耗总功率W 5为转速n p5和转矩T p5之积除以60效率的值的2π倍，即：
Figure PCTCN2022128230-appb-000058
步骤9.2：与步骤9.1同时的，计算过渡点P 5’(n p5,T p5’)系统消耗总功率W 5’，过渡点P 5’(n p5,T p5’)为点P 5(n p5,T p5)转速相同的任意一点。当电机运行于点P 5’(n p5,T p5’)时，其效率为η p5，则过渡点P 5’(n p5,T p5’)系统消耗的总功率为此时电机运行消耗的消耗减去回馈至电池的效率，即：
Figure PCTCN2022128230-appb-000059
步骤9.3：基于步骤9.1与9.2，建立升功率运行的数学模型。升功率运行的数学模型为过渡点P 5’点系统消耗总功率W 5’减去P 5点系统消耗总功率W 5，即为：
Figure PCTCN2022128230-appb-000060
步骤9.4：基于步骤9.3，确定集合S P5。令升功率运行的数学模型f(T p5')大于0，获得下式：
Figure PCTCN2022128230-appb-000061
过渡点P 5’(n p5,T p5’)应当满足上式条件，将满足条件的过渡点P 5’(n p5,T p5’)的形成集合S P5。需要注意的是，对于部分点P 5(n p5,T p5)，S P5可能为空集，这主要取决于电机功率与电机具体性能。当S P5为空集时，第五区高速负载区就不采用电机升功率运行方法，而是直接运行。
步骤9.5：当集合S P5为非空集采用电机升功率运行时，在集合S P5中确定出升功率运行点H 5(n H5,T H5)。为在电机单元正常运行范围内，最大化提高电机系统效率，升功率运行点H 5(n H5,T H5)需要属于S P5的同时需要属于第五区高速负载区或第六区双倍高效区，此外，需要满足T H5大于等于T P5。也就是说升功率运行点H 5(n H5,T H5)需要满足：
Figure PCTCN2022128230-appb-000062
需要注意的是，若集合S P5存在符合条件的点H 5(n H5,T H5)不唯一，则取满足条件的点中升功率数学模型函数f(T p5')最大值点。
步骤9.6：基于步骤9.5中的结果，采用有限元软件仿真获得电机运行于点H 5(n H5,T H5)时的逆变器输出的ABC三相电流。
步骤10：与步骤9同时的，基于步骤6的分区结果，确定第六区双倍高效区的控制方 法。由于第六区双倍高效区的转矩高于第三区、第四区以及第五区，对于第六区双倍高效区中任意一点P 6(n p6,T p6)，所有电源电子开关均闭合的同时所有绕组电子开关闭合。另外，第六区双倍高效区的主要控制原则为系统效率最高原则，由于第六区双倍高效区运行点效率高于第三区、第五区、第七区以及第八区，因此运行点P 6(n p6,T p6)正常工作。采用有限元软件仿真获得电机运行于P 6(n p6,T p6)时的逆变器输出的ABC电流，由DSP控制器控制逆变器的输出。
步骤11：与步骤10同时的，基于步骤6的分区结果，确定第七区高速过载区的控制方法。由于第七区高速过载区的转矩高于第三区、第四区以及第五区，对于第七区高速过载区中任意一点运行点P 7(n p7,T p7)，所有电源电子开关均闭合的同时所有绕组电子开关闭合。另外，第七区高速过载区的主要控制原则为系统效率最高原则，因此第七区控制策略为两个控制模块协调工作，即第一逆变器与第二逆变器输出的电流不同，两个N/2的电机单元1工作在两个运行点，则第一控制模块使电机工作在点P 71(n p71,T p71)；第二控制模块使电机工作在点P 72(n p72,T p72)。对于第七区高速过载区中任意一点P 7(n p7,T p7)，控制方法为以下步骤：
步骤11.1：建立协同运行功率函数。第七区高速过载区有两个不同点，分别为第一过渡点P 71’(n p71’,T p71’)与第二过渡点P 72’(n p72’,T p72’)，该两个过过渡点转速均与第七区运行点P 7(n p7,T p7)相等且转矩之和为T P7，即
Figure PCTCN2022128230-appb-000063
满足条件的过渡点集合记作S P7。
第一过渡点P 71’(n p71’,T p71’)系统消耗总功率W 71’为转速n p71’和转矩T p71’之积除以60效率的值的2π倍，即：
Figure PCTCN2022128230-appb-000064
相似地，第二过渡点P 72’(n p72’,T p72’)系统消耗总功率W 72’为转速转矩之积除以60效率的值的2π倍，即：
Figure PCTCN2022128230-appb-000065
协同运行功率函数为第一过渡点P 71’(n p71’,T p71’)系统消耗总功率W 71’与第二过渡点P 72’(n p72’,T p72’)系统消耗总功率W 72’之和，即：
Figure PCTCN2022128230-appb-000066
步骤11.2：基于步骤11.1中确定的协同运行功率函数W 7(T p71',T p72')，确定两个运行点P 71(n p71,T p71)与P 72(n p72,T p72)。
第一运行点P 71(n p71,T p71)的系统消耗总功率为W 71，第二运行点P 72(n p72,T p72)的系统消耗总功率为W 72，则两个运行点P 71(n p71,T p71)和P 72(n p72,T p72)的协同运行功率函数W 7(T p71,T p72)＝W 71+W 72，当两个运行点符合其协同运行功率函数W 7(T p71,T p72)等于所述的两个过渡点P 71’(n p71’,T p71’)和P 72’(n p72’,T p72’)的协同运行功率函数W 7(T p71',T p72')的最小值时，则电机工作在该两个运行点P 71(n p71,T p71)和P 72(n p72,T p72)，运行点P 71(n p71,T p71)和P 72(n p72,T p72)分别由第一控制模块和第二控制模块控制运行。
步骤11.3：基于步骤11.2中确定的点P 71(n p71,T p71)，确定第一逆变器输出电流。此时所 有绕组电子开关闭合的同时有且仅有第一电源电子开关闭合，利用有限元软件仿真获得第一逆变器的输出电流。
同时，确定第二逆变器输出的电流。此时所有绕组电子开关闭合的同时有且仅有第二电源电子开关闭合，利用有限元软件仿真获得第二逆变器的输出电流。
步骤12：与步骤11同时的，基于步骤6的分区结果，确定第八区高速弱磁区的控制策略。由第八区高速弱磁区的转速高于其余所有子区域，对于第八区高速弱磁区中任意一点P 8(n p8,T p8)，所有电源电子开关均闭合的同时所有绕组电子开关闭合，采用有限元软件仿真获得电机运行于P 8(n p8,T p8)时的逆变器输出电流。
综上所述，电机根据子区域特点，分为8个工作模式：
1、当电机运行于第一区低速负载区时，第一电源电子开关与第二电源电子开关均闭合，在第一绕组电子开关到第N/2电源电子开关中任意闭合K个开关，且在第(N/2+1)绕组电子开关到第N个电源电子开关中任意闭合K个开关。两个逆变器输出同频同幅值不同相位的电流，两个部分的电机单元1工作在同一运行点，转矩波形波谷补偿，大幅度降低此子区域内电机的转矩脉动，提高转矩质量。
2、当电机运行于第二区低速过载区时，所有的电源电子开关与绕组电子开关均闭合。两个逆变器输出同频同幅值同相位的电流，两个部分的电机单元1工作在同一运行点，电机单元1输出的转矩代数叠加，有效提高电机峰值转矩，增强电机转矩输出能力。
3、当电机运行于第三区高速低载区时，第一电源电子开关与第二电源电子开关有且仅有一个闭合的同时所有绕组电子开关闭合。电机单元1升功率运行，有效提高了电机工作效率与整机运行效率。
4、当电机运行于第四区高速高效区时，第一电源电子开关与第二电源电子开关有且仅有一个闭合的同时所有绕组电子开关闭合。N/2个单元电机正常运行，此时电机运行效率较高。
5、当电机运行于第五区高速负载区时，所有电源电子开关均闭合的同时所有绕组电子开关闭合，所有电机单元1升功率运行，有效提高了电机工作效率与电机系统运行效率。
6、当电机运行于第六区双倍高效区时，所有电源电子开关均闭合的同时所有绕组电子开关闭合，所有电机单元1正常运行，电机高效运行。
7、当电机运行于第七区双倍高效区时，所有电源电子开关均闭合的同时所有绕组电子开关闭合。两个逆变器输出电流，两个部分的电机单元1工作在不同运行点，输出转矩代数叠加，大幅度提高电机运行效率。
8、当电机运行于第八区高速弱磁区时，所有电源电子开关均闭合的同时所有绕组电子开关闭合，所有电机单元1正常运行，电机高速运行。
上文所列出的一系列的详细说明仅是针对本发明的可行性实施方式的具体说明，它们并非用以限制本发明的保护范围，凡未脱离本发明技术所创的等效方式或变更均应包含在本发明的保护范围之内。

Claims (17)

  1. 一种多极少槽单元化永磁轮毂电机，其特征是：其由沿径向截面圆周方向均匀分布的N个相同的电机单元(1)组成，每个电机单元(1)包括1/N的外转子(2)、1/N的内定子(3)以及1/N的集中式绕组(4)，内定子(3)同轴心套在外转子(2)内部，内定子(3)上绕有集中式绕组(4)，每个电机单元(1)中的集中式绕组(4)均为三相对称且分布相同；外转子(2)具有一个转子铁芯(2.3)，2a个永磁磁钢组(2.4)沿圆周方向均匀分布在转子铁芯(2.3)上，每个永磁磁钢组(2.4)均由第一、第二矩形永磁磁钢(2.4.1、2.4.2)和弧形永磁磁钢(2.4.3)组成，第一、第二矩形永磁磁钢(2.4.1、2.4.2)结构相同且径向截面均呈矩形，内外斜向为矩形长度方向，呈开口朝向气隙侧的V型布置于弧形永磁磁钢(2.4.3)外侧，相对于弧形永磁磁钢(2.4.3)沿直径方向的中心线对称；第一、第二矩形永磁磁钢(2.4.1、2.4.2)充磁方向垂直于自身长度方向，弧形永磁磁钢(2.4.3)充磁方向和中心线方向一致，属于同一个永磁磁钢组(2.4)的第一、第二矩形永磁磁钢(2.4.1、2.4.2)及弧形永磁体(2.4.3)充磁方向同时指向或背离气隙，相邻两个永磁磁钢组(2.4)充磁方向相反；转子极对数P r、定子槽数N s、电机相数m、电机槽距角τ以及N需同时满足：P r＞N s、P r＝Na、N s＝mNb、
    Figure PCTCN2022128230-appb-100001
    cτ＝d*2π、
    Figure PCTCN2022128230-appb-100002
    N＝2i，i,a,b,c,d,e均是正整数。
  2. 根据权利要求1所述的一种多极少槽单元化永磁轮毂电机，其特征是：每个所述的1/N的外转子(2)，沿轴向上分为相同的M个转子段，M个转子段沿同一旋转方向依次地旋转一个机械错位角布置，20mm≤l ef/M≤120mm，l ef为电机轴向长度。
  3. 根据权利要求2所述的一种多极少槽单元化永磁轮毂电机，其特征是：所述的一个机械错位角的确定方法是：
    步骤1)：先赋值初始机械错位角为0；
    步骤2)：仿真电机输出的转矩波形，计算出初始转矩脉动；
    步骤3)：对所述的转矩波形傅里叶分解，求取最高幅值谐波分量的谐波次数k，计算出过渡机械错位角
    Figure PCTCN2022128230-appb-100003
    再仿真过渡机械错位角α 1时的转矩波形，计算出过渡转矩脉动；
    步骤4)：比较所述的过渡转矩脉动和所述的初始转矩脉动，若过渡转矩脉动小于初始转矩脉动，则过渡机械错位角α 1即是转子段旋转的一个机械错位角，反之，则将过渡机械错位角α 1赋值到初始机械错位角后循环步骤2)-3)。
  4. 根据权利要求1所述的一种多极少槽单元化永磁轮毂电机，其特征是：每个弧形永磁磁钢(2.4.3)在径向截面上均由一条弧形永磁磁钢外长边(2.4.3.1)，一条弧形永磁磁钢内长边(2.4.3.2)以及两条弧形永磁磁钢短边(2.4.3.3)围成，弧形永磁磁钢外长边(2.4.3.1)和弧形永磁磁钢内长边(2.4.3.2)的弧形中心与外转子(2)中心相同，弧形永磁磁钢短边(2.4.3.3)和外转子(2)的直径方向一致，弧形永磁磁钢内长边(2.4.3.2)为正弦曲线f 1(θ 1)＝f 1maxsin(θ 1)，θ 1∈[π,2π]，f 1max为幅值，当θ 1为3π/2时，点f 1(3π/2)位于外转子(2)内表面上。
  5. 根据权利要求4所述的一种多极少槽单元化永磁轮毂电机，其特征是：每个第一、第二矩形永磁磁钢(2.4.1、2.4.2)在靠近气隙侧的端部设有内磁障(2.5)，远离气隙侧的端部设有外磁障(2.6)，每个弧形永磁磁钢(2.4)的切向两端各设有一个成为气隙一部分的虚拟槽(2.7)。
  6. 根据权利要求5所述的一种多极少槽单元化永磁轮毂电机，其特征是：所述的内磁障(2.5)径向截面为五边形，五边形的第一边为第一、第二矩形永磁磁钢(2.4.1、2.4.2)靠近气隙侧长边的延长边，第二边为与外转子(2)同轴心的圆弧边，第三边位于外转子(2)半径上，第四边平行于第一边且位于第一边外侧，第一边与第四边间的距离小于第一、第二矩形永磁磁钢(2.4.1、2.4.2)的宽度，第五边与第一、第二矩形永磁磁钢(2.4.1、2.4.2)靠近气隙侧的短边重合。
  7. 根据权利要求5所述的一种多极少槽单元化永磁轮毂电机，其特征是：所述的外磁障(2.6)径向截面是五边形，五边形的第一边为第一、第二矩形永磁磁钢(2.4.1、2.4.2)靠近气隙侧长边的延长线，第二边位于外转子(2)半径上，第三边为与外转子(2)同轴心的圆弧边，第四边平行于第一边且位于第一边外侧，第一边与第四边间的距离小于第一、第二矩形永磁磁钢(2.4.1、2.4.2)的宽度，第五边与第一、第二矩形永磁磁钢(2.4.1、2.4.2)远离气隙侧的短边重合。
  8. 根据权利要求6所述的一种多极少槽单元化永磁轮毂电机，其特征是：所述的虚拟槽(2.7)在径向截面上是一段正弦曲线f 2(θ 2)＝f 2maxsin(θ 2)，θ 2为[π/2,π]，f 2max为幅值，θ 2为π/2时，点f 2(π/2)是弧形永磁磁钢内长边(2.4.3.2)与弧形永磁磁钢短边(2.4.3.3)的交点，θ 2为π时，点f 2(π)是经过内磁障(2.5)的第二边与第三边的交点的一条半径与外转子(2)内表面的交点。
  9. 根据权利要求1所述的一种多极少槽单元化永磁轮毂电机，其特征是：所述的第一、第二矩形永磁磁钢(2.4.1、2.4.2)形成V型夹角β pm满足：40°≤β pm≤65°，长边长度w pm与 宽边长度h pm满足：2≤w pm/h pm≤4。
  10. 根据权利要求1所述的一种多极少槽单元化永磁轮毂电机，其特征是：所述的弧形永磁磁钢2.4.3沿径向上的的最小宽度h pmin与最大宽度h pmax满足：1.5≤h pmax/h pmin≤2。
  11. 一种如权利要求1所述的多极少槽单元化永磁轮毂电机的协同控制系统，其特征是：包括一个电池、两个控制模块和N个绕组电子开关，一个绕组电子开关控制一个电机单元(1)中集中式绕组(4)的通断，每个控制模块均由一个电源电子开关，一个DSP控制器和一个逆变器依次串联组成，两个电源电子开关的输入端分别连接电池的输出端，每个逆变器的输出端分别连接N/2个绕组电子开关，集中式绕组(4)输出端经整流器连接电池。
  12. 一种如权利要求11所述的协同控制系统的控制方法，其特征是包括以下步骤：
    步骤1)：闭合两个电源电子开关与N个绕组电子开关，以横坐标为电机转速，纵坐标为电机输出转矩，仿真获得电机的外特性曲线g；再断开其中一个电源电子开关，仿真获得电机的外特性曲线f；以外特性曲线f上最高转矩对应的最高转速作为临界转速n b，当电机运行点的转速n p≤n b，为恒转矩区；
    步骤2)：将所述的恒转矩区分成第一区和第二区，当运行点的转矩T p≤T b，为第一区，当T p＞T b，为第二区，T b为临界转矩T b，是一个电机单元(1)峰值转矩T x的N-2倍；第二区的运行点转矩的最大值不超过外特性曲线g对应的转矩；
    步骤3)：当运行点在第一区，两个电源电子开关均闭合，与每个控制模块连接的N/2电机单元(1)中至少闭合
    Figure PCTCN2022128230-appb-100004
    个绕组电子开关，T p1为第一区运行点的转矩；两个逆变器输出同幅值不同相位的电流；当运行点在第二区，两个电源电子开关和N个绕组电子开关均闭合。
  13. 根据权利要求12所述的控制方法，其特征是：步骤3)中，仿真计算K个绕组电子开关时电机输出转矩波形和两个逆变器输出的过渡电流，傅里叶分解所述的输出转矩波形，求取主要谐波分量的谐波次数r，计算出电流错位角
    Figure PCTCN2022128230-appb-100005
    则两个逆变器输出的三相电流幅值I max＝1.05 I max0，I max0为过渡电流幅值，第一个逆变器输出的三相电流相位超前过渡电流相位β/2，第二个逆变器输出的三相电流相位滞后过渡电流相位β/2。
  14. 一种如权利要求11所述的协同控制系统的控制方法，其特征是包括以下步骤：
    步骤(Ⅰ)：闭合两个电源电子开关与N个绕组电子开关，以横坐标为电机转速，纵坐标为电机输出转矩，仿真获得电机的外特性曲线g；再断开其中一个电源电子开关，仿真获得电机的外特性曲线f；以外特性曲线f上最高转矩对应的最高转速作为临界转速n b，当 电机运行点转速n p＞n b，为恒功率区；
    步骤(Ⅱ)：将所述的恒功率区分成第三区至第八区，当运行点的效率η p≥η b，则为第四区，η b为单一控制模块运行时电机的边界效率；由横坐标轴、外特性曲线g与经过第四区速度最高点E的一条垂直于横坐标的直线j所围的区域为第八区；转矩小于第四区速度最低点D的转矩且转速小于点D的转速的区域为区域S 31，由横坐标、经过点D的一条垂直于横坐标的直线k、第四区点D、E下半周边界线、直线j所围成的区域为区域S 32，S 31与S 32的并集为第三区；转速与第四区转速相同且转矩是第四区转矩的两倍的区域是第六区，第六区的速度最高点G在直线j上，速度最低点F在直线k上；转矩小于点F的转矩、高于点D的转矩且转速小于点F转速的区域为区域S 51，由直线k、直线j、第四区点D、E上半周边界、第六区点F、G下半周边界所围成的区域为区域S 52，S 51与S 52的并集为第五区；恒功率区中的剩余区域为第七区；
    步骤(Ⅲ)：当运行点在第三区，两个电源电子开关有且仅有一个闭合，N个绕组电子开关均闭合，且采用不改变电机转速的同时提高转矩的升功率运行；当运行点在第四区，两个电源电子开关有且仅有一个闭合，N个绕组电子开关均闭合；当运行点在第五区，两个电源电子开关均闭合，N个绕组电子开关均闭合且升功率运行；当运行点在第六区，两个电源电子开关均闭合，N个绕组电子开关均闭合；当运行点在第七区，两个电源电子开关均闭合，N个绕组电子开关均闭合，两个逆变器输出的电流不同；当运行点在第八区，两个电源电子开关均闭合，N个绕组电子开关均闭合。
  15. 根据权利要求14所述的控制方法，其特征是：步骤(Ⅲ)中第三区的升功率运行是：与第三区运行点P 3(n p3,T p3)转速相同的过渡点P 3’(n p3,T p3’)，当满足条件
    Figure PCTCN2022128230-appb-100006
    时形成集合S P3，集合S P3中属于第三区或第四区且转矩大于或等于T P3的运行点为升功率运行点H 3(n H3,T H3)，仿真获得电机运行于升功率运行点H 3(n H3,T H3)时的逆变器输出的三相电流，将多余能量回馈至电池；第五区的升功率运行与第三区的升功率运行雷同；n p3、T p3和η p3分别为运行点P 3(n p3,T p3)的转速、转矩和效率，T p3’和η p3’分别为过渡点的转矩和效率，η pg为发电效率。
  16. 根据权利要求14所述的控制方法，其特征是：步骤(Ⅲ)中，在第七区，与第七区运行点转速相等有两个过渡点，两个过渡点转矩之和等于第七区运行点转矩，计算出所述的两个过渡点的系统消耗总功率之和，当有两个运行点的系统消耗总功率之和等于所述的两个过渡点系统消耗总功率之和的最小值时，则所述的两个运行点分别由对应的两个控 制模块控制运行。
  17. 根据权利要求16所述的控制方法，其特征是：N个绕组电子开关闭合且仅有第一个电源电子开关闭合，仿真获得第一个逆变器的输出电流，控制两个运行点中的第一个运行点；N个绕组电子开关闭合且仅有第二个电源电子开关闭合，仿真获得第二逆变器的输出电流，控制两个运行点中的第二个运行点。
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