WO2023149119A1 - High frequency circuit and communication device - Google Patents

High frequency circuit and communication device Download PDF

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Publication number
WO2023149119A1
WO2023149119A1 PCT/JP2022/047399 JP2022047399W WO2023149119A1 WO 2023149119 A1 WO2023149119 A1 WO 2023149119A1 JP 2022047399 W JP2022047399 W JP 2022047399W WO 2023149119 A1 WO2023149119 A1 WO 2023149119A1
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Prior art keywords
terminal
filter
band
frequency circuit
impedance
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PCT/JP2022/047399
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French (fr)
Japanese (ja)
Inventor
▲琢▼真 黒▲柳▼
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株式会社村田製作所
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Publication of WO2023149119A1 publication Critical patent/WO2023149119A1/en

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    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
    • H03H9/00—Networks comprising electromechanical or electro-acoustic devices; Electromechanical resonators
    • H03H9/15—Constructional features of resonators consisting of piezoelectric or electrostrictive material
    • H03H9/17—Constructional features of resonators consisting of piezoelectric or electrostrictive material having a single resonator
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
    • H03H9/00—Networks comprising electromechanical or electro-acoustic devices; Electromechanical resonators
    • H03H9/70—Multiple-port networks for connecting several sources or loads, working on different frequencies or frequency bands, to a common load or source
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
    • H03H9/00—Networks comprising electromechanical or electro-acoustic devices; Electromechanical resonators
    • H03H9/70—Multiple-port networks for connecting several sources or loads, working on different frequencies or frequency bands, to a common load or source
    • H03H9/72—Networks using surface acoustic waves

Definitions

  • the present invention relates to high frequency circuits and communication devices.
  • Patent Document 1 discloses a high-frequency front-end circuit (high-frequency circuit) capable of simultaneous communication of signals in a plurality of frequency bands.
  • the high-frequency circuit includes a switch connected to the antenna, a transmit filter connected to the switch, a receive filter, a transmit filter, between the switch and the transmit filter, between the switch and the receive filter, and between the switch and the receive filter. and a phase adjustment circuit positioned at least one between the transmission filters.
  • the phase adjustment circuit is arranged to adjust the impedance in simultaneous transmission 1 and simultaneous transmission 2, but the phase adjustment circuit is arranged in the series arm path connecting the switch and each filter. , the transmission loss of the high-frequency circuit increases.
  • a high-frequency circuit has an antenna terminal, an input terminal, an output terminal, and a passband including a transmission band of a first band for time division duplex (TDD).
  • TDD time division duplex
  • the second filter is connected between the third terminal and the output terminal; the third filter is connected to the fourth terminal; one of the first filter and the second filter is connected between the switch circuit and the input terminal;
  • a first elastic wave resonator connected between a series arm path connecting one of the output terminals and the ground, and a first inductor connected in series with the first elastic wave resonator between the series arm path and the ground. and the first elastic wave resonator is connected at a position closest to the switch circuit among the parallel arm resonators connected between the series arm path and the ground.
  • FIG. 1 is a circuit configuration diagram of a high-frequency circuit and a communication device according to an embodiment.
  • 2A is a diagram illustrating a circuit configuration example of a first filter according to the embodiment
  • FIG. 2B is a diagram illustrating a circuit configuration example of a second filter according to the embodiment
  • FIG. 3A is a plan view and a cross-sectional view schematically showing a first example of elastic wave resonators that constitute the first filter and the second filter according to the embodiment.
  • FIG. 3B is a cross-sectional view schematically showing a second example of elastic wave resonators forming the first filter and the second filter according to the embodiment.
  • FIG. 1 is a circuit configuration diagram of a high-frequency circuit and a communication device according to an embodiment.
  • 2A is a diagram illustrating a circuit configuration example of a first filter according to the embodiment
  • FIG. 2B is a diagram illustrating a circuit configuration example of a second filter according to the embodiment
  • FIG. 3A is a plan view
  • FIG. 3C is a cross-sectional view schematically showing a third example of elastic wave resonators forming the first filter and the second filter according to the embodiment.
  • FIG. 4 is a circuit configuration diagram of a high-frequency circuit according to a comparative example, and a Smith chart showing the impedance of each filter viewed from a switch selection terminal.
  • FIG. 5 is a circuit configuration diagram of a high-frequency circuit according to the embodiment, and a Smith chart showing the impedance of each filter viewed from a switch selection terminal.
  • FIG. 4 is a circuit configuration diagram of a high-frequency circuit according to a comparative example, and a Smith chart showing the impedance of each filter viewed from a switch selection terminal.
  • FIG. 5 is a circuit configuration diagram of a high-frequency circuit according to the embodiment, and a Smith chart showing the impedance of each filter viewed from a switch selection terminal.
  • 6A is a circuit configuration diagram of a high-frequency circuit according to the embodiment, and a Smith chart showing impedances of a first filter and a third filter commonly connected from a switch common terminal and an antenna terminal
  • 6B is a circuit configuration diagram of the high-frequency circuit according to the embodiment, and a Smith chart showing the impedance of the second filter and the third filter commonly connected from the switch common terminal and the antenna terminal
  • FIG. 7 is a circuit configuration diagram of a high-frequency circuit according to Modification 1, and a Smith chart showing the impedance of each filter viewed from a switch selection terminal.
  • 8A is a circuit configuration diagram of a high-frequency circuit according to Modification 1, and a Smith chart showing impedance when viewing a first filter and a third filter commonly connected from an antenna terminal.
  • 8B is a circuit configuration diagram of a high-frequency circuit according to Modification 1, and a Smith chart showing impedance when viewing a second filter and a third filter commonly connected from an antenna terminal;
  • FIG. 9 is a circuit configuration diagram of a high-frequency circuit according to Modification 2.
  • FIG. 10 shows a Smith chart showing the impedance of the first filter viewed from each point on the transmission path and the impedance of the second filter viewed from each point on the reception path in the high-frequency circuit according to Modification 2. Smith chart.
  • connection means not only direct connection with connection terminals and/or wiring conductors, but also electrical connection via other circuit elements. Also, “connected between A and B” and “connected between A and B” mean being connected to A and B on a path connecting A and B.
  • path refers to a transmission line composed of a wire through which a high-frequency signal propagates, an electrode directly connected to the wire, and a terminal directly connected to the wire or the electrode.
  • the component A is arranged in series with the path B” means that both the signal input terminal and the signal output terminal of the component A are connected to the wiring, electrodes, or terminals that constitute the path B. means that there is
  • the passband of a filter is defined as the frequency band between two frequencies that are 3 dB greater than the minimum value of insertion loss within the passband.
  • FIG. 1 is a circuit configuration diagram of a high frequency circuit 1 and a communication device 4 according to an embodiment.
  • the communication device 4 includes a high frequency circuit 1, an antenna 2, and an RF signal processing circuit (RFIC) 3.
  • RFIC RF signal processing circuit
  • the high frequency circuit 1 transmits high frequency signals between the antenna 2 and the RFIC 3 .
  • a detailed circuit configuration of the high-frequency circuit 1 will be described later.
  • the antenna 2 is connected to an antenna connection terminal 100 of the high-frequency circuit 1, transmits a high-frequency signal (hereinafter referred to as a transmission signal) output from the high-frequency circuit 1, and receives a high-frequency signal (hereinafter referred to as a received signal) from the outside. ) is received and output to the high-frequency circuit 1 .
  • a transmission signal a high-frequency signal
  • a received signal a high-frequency signal
  • the RFIC 3 is an example of a signal processing circuit that processes high frequency signals. Specifically, the RFIC 3 performs signal processing such as down-conversion on the received signal input via the receiving path of the high-frequency circuit 1, and converts the received signal generated by the signal processing into a baseband signal processing circuit (BBIC, not shown). Further, the RFIC 3 performs signal processing such as up-conversion on the signal input from the BBIC, and outputs the transmission signal generated by the signal processing to the transmission path of the high frequency circuit 1 .
  • the RFIC 3 also has a control section that controls the switches and amplification elements of the high-frequency circuit 1 . A part or all of the functions of the RFIC 3 as a control unit may be implemented outside the RFIC 3, for example, in the BBIC or the high-frequency circuit 1.
  • the RFIC 3 controls the connection of the switch circuit 30 included in the high-frequency circuit 1 based on which band (frequency band) the signal is to be transmitted and which of the transmission signal and the reception signal is to be transmitted. It has a function as a department.
  • the antenna 2 is not an essential component in the communication device 4 according to the present embodiment.
  • the high frequency circuit 1 includes filters 11, 12 and 20, a switch circuit 30, an inductor 90, a power amplifier 41, a low noise amplifier 42, an antenna connection terminal 100, and an input terminal 110. , an output terminal 120 and an input/output terminal 130 .
  • the antenna connection terminal 100 is an example of an antenna terminal and is connected to the antenna 2.
  • the input terminal 110 is a terminal for receiving a band A transmission signal from the RFIC 3 .
  • the output terminal 120 is a terminal for outputting the reception signal of band A to the RFIC 3 .
  • the input/output terminal 130 is a terminal that receives a band B transmission signal from the RFIC 3 or outputs a band B reception signal to the RFIC 3 .
  • the filter 11 is an example of a first filter, and is a transmission filter having a passband including the transmission band of band A for time division duplex (TDD).
  • Filter 11 has one or more elastic wave resonators.
  • the filter 12 is an example of a second filter, and is a filter for reception having a passband including the reception band of band A for TDD.
  • Filter 12 has one or more elastic wave resonators.
  • the filter 20 is an example of a third filter, and has a passband including at least part of band B different from band A.
  • Filter 20 has one or more elastic wave resonators.
  • Band B may be a frequency division duplex (FDD) band or a TDD band.
  • the switch circuit 30 has a terminal 30a (first terminal), a terminal 30b (second terminal), a terminal 30c (third terminal) and a terminal 30d (fourth terminal).
  • the switch circuit 30 can (1) simultaneously connect the terminals 30a and 30b and connect the terminals 30a and 30d, and (2) connect the terminals 30a and 30c and connect the terminals 30a and 30d. can be executed simultaneously.
  • the switch circuit 30 has, for example, terminals 30a, 30b, and 30c, and a first SPDT (Single Pole Double Throw) switch that switches between the connection between the terminals 30a and 30b and the connection between the terminals 30a and 30c. and a SPST (Single Pole Single Throw) type second switch that has terminals 30a and 30d and switches connection and disconnection between the terminals 30a and 30d.
  • the first switch may perform a state in which it is not connected to any of the terminals 30a and 30b and 30c.
  • the switch circuit 30 has terminals 30a, 30d, and 30e, and switches connection/disconnection between the terminals 30a and 30d, and switches connection/disconnection between the terminals 30a and 30e. and terminals 30b, 30c, and 30f, the terminal 30f is connected to the terminal 30e, and the fourth SPDT type switching the connection between the terminal 30f and the terminal 30b and the connection between the terminal 30f and the terminal 30c. It may be configured with a switch.
  • the switch circuit 30 has a control terminal to which a signal from the control section of the RFIC 3 is input, and a control circuit for controlling switching of each switch based on the control signal is provided inside the switch circuit 30.
  • the power amplifier 41 is connected between the input terminal 110 and the filter 11 and amplifies the band A transmission signal.
  • Low noise amplifier 42 is connected between filter 12 and output terminal 120 and amplifies the received band A signal.
  • the inductor 90 is an example of a second inductor, and is connected between the antenna connection path connecting the antenna connection terminal 100 and the terminal 30a and the ground.
  • the antenna connection terminal 100 is connected to the terminal 30a.
  • Filter 11 is connected between terminal 30b and input terminal 110 .
  • Filter 12 is connected between terminal 30 c and output terminal 120 .
  • Filter 20 is connected between terminal 30 d and input/output terminal 130 .
  • the high-frequency circuit 1 does not have to include the power amplifier 41, the low-noise amplifier 42, and the inductor 90.
  • the high-frequency circuit 1 can simultaneously transmit a transmission signal of band A and a signal of band B, and can simultaneously transmit a reception signal of band A and a signal of band B. is.
  • terminals 30a and 30b are connected, and terminals 30a and 30d are connected.
  • terminals 30a and 30c are connected, and the terminals 30a and 30d are connected.
  • band A and band B are used for communication systems constructed using radio access technology (RAT), such as standardization organizations (e.g., 3GPP (registered trademark), IEEE (Institute of Electrical and Electronics Engineers ) etc.).
  • RAT radio access technology
  • standardization organizations e.g., 3GPP (registered trademark), IEEE (Institute of Electrical and Electronics Engineers ) etc.
  • 4G (4th Generation)-LTE (Long Term Evolution) system Long Term Evolution) system
  • WLAN Wireless Local Area Network
  • band A for example, 4G-LTE band 41 or 5G-NR band n41 (2496-2690 MHz) is applied.
  • band B for example, 4G-LTE band 40 or 5G-NR band n40 (2300-2400 MHz), 4G-LTE band 1 or 5G-NR band n1 (uplink operating band: 1920-1980 MHz , downlink operating band: 2110-2170 MHz) and either band 32 of 4G-LTE or band n32 of 5G-NR (downlink operating band: 1452-1496 MHz).
  • the uplink operating band means the frequency range designated for the uplink among the above bands.
  • the downlink operating band means the frequency range designated for the downlink among the above bands.
  • the band B is located on the lower frequency side than the band A.
  • FIG. 2A is a diagram showing a circuit configuration example of the filter 11 according to the embodiment.
  • FIG. 2B is a diagram showing a circuit configuration example of the filter 12 according to the embodiment.
  • the filter 11 according to the present embodiment has, for example, the circuit configuration of the elastic wave filter shown in FIG. 2A.
  • the filter 11 includes, for example, series arm resonators 101, 102, 103, 104, 105, 106 and 107; and output terminals 111 and 112 .
  • Each of the series arm resonators 101-107 and the parallel arm resonators 151-156 is an elastic wave resonator.
  • the input/output terminal 112 is connected to the terminal 30 b and the input/output terminal 111 is connected to the output terminal of the power amplifier 41 .
  • Series arm resonators 101 to 107 are arranged on a first series arm path connecting input/output terminals 111 and 112 .
  • each of the parallel arm resonators 151 to 156 is connected between each connection point of the series arm resonators 101 to 107 and the ground.
  • the filter 11 constitutes a ladder-type bandpass filter.
  • the filter 11 may be a longitudinal coupling filter in which a plurality of IDT (InterDigital Transducer) electrodes are arranged on a substrate having piezoelectricity in the elastic wave propagation direction.
  • IDT InterDigital Transducer
  • the filter 12 includes, for example, series arm resonators 201, 202, 203, 204 and 205; parallel arm resonators 251, 252, 253, 254 and 255; an inductor 250; 121 and 122.
  • Each of the series arm resonators 201-205 and the parallel arm resonators 251-255 is an elastic wave resonator.
  • the input/output terminal 121 is connected to the terminal 30c, and the input/output terminal 122 is connected to the input terminal of the low noise amplifier 42.
  • Series arm resonators 201 to 205 are arranged on a second series arm path connecting input/output terminals 121 and 122 . Also, each of the parallel arm resonators 251 to 255 is connected between each connection point of the series arm resonators 201 to 205 and the input/output terminal 122 and the ground.
  • Parallel arm resonator 251 is an example of a first elastic wave resonator, and among parallel arm resonators 251 to 255 connected between a second series arm path connecting input/output terminals 121 and 122 and the ground, It is connected closest to the switch circuit 30 .
  • the inductor 250 is an example of a first inductor, is arranged between the second series arm path and the ground, and is connected in series with the parallel arm resonator 251 .
  • inductor 250 is connected between parallel arm resonator 251 and the ground, but inductor 250 may be connected between the second series arm path and parallel arm resonator 251 .
  • the filter 12 constitutes a ladder-type bandpass filter.
  • the number of series arm resonators and parallel arm resonators is arbitrary.
  • the filter 12 may have at least the parallel arm resonator 251 and the inductor 250.
  • the parallel arm resonator 251, the inductor 250, and the plurality of IDT electrodes are used for elastic wave propagation on a piezoelectric substrate. and a tandem-coupled filter section arranged in a direction.
  • the impedance of the own band (band A) of the filter 12 can be adjusted without shifting the impedance of the other band (band A).
  • the phase width of the impedance of B) can be adjusted.
  • the phase adjustment circuit arranged between the filters 11 and 12 and the switch circuit 30 is eliminated by the inductor 250, and the transmission signal of the band A is reduced. It is possible to simultaneously transmit the transmission signal or reception signal of band A and the signal of band B while reducing the difference in impedance matching that occurs when switching between the transmission and reception signals. In other words, since the number of phase adjustment circuits can be reduced, it is possible to reduce the transmission loss when simultaneously transmitting the high-frequency signal of band A and the high-frequency signal of band B for TDD.
  • the filter 12 for passing the received signal has the first acoustic wave resonator and the first inductor, but the filter 11 has the first acoustic wave resonator and the first inductor.
  • both filters 11 and 12 may have a first acoustic wave resonator and a first inductor.
  • one of the filters 11 and 12 may have the first inductor and the other of the filters 11 and 12 may not have the first inductor.
  • the first inductor has the function of phase-adjusting the matching band impedance of one of the filters 11 and 12 with the matching band impedance of the other of the filters 11 and 12 . Therefore, it is sufficient if the first inductor is placed in only one of filters 11 and 12 . According to this, it is possible to reduce the transmission loss in the case of simultaneously transmitting the high frequency signal of band A and the high frequency signal of band B for TDD with a simplified circuit configuration.
  • FIG. 3A is a plan view and a cross-sectional view schematically showing a first example of elastic wave resonators forming filters 11 and 12 according to the embodiment.
  • the figure illustrates the basic structure of the elastic wave resonators forming the filters 11 and 12 .
  • the elastic wave resonator 60 shown in FIG. 3A is for explaining a typical structure of an elastic wave resonator, and the number and length of the electrode fingers constituting the electrodes are Not limited.
  • the elastic wave resonator 60 is composed of a piezoelectric substrate 50 and comb electrodes 60a and 60b.
  • a pair of comb electrodes 60a and 60b facing each other are formed on the substrate 50.
  • the comb-shaped electrode 60a is composed of a plurality of parallel electrode fingers 61a and busbar electrodes 62a connecting the plurality of electrode fingers 61a.
  • the comb-shaped electrode 60b is composed of a plurality of parallel electrode fingers 61b and a busbar electrode 62b connecting the plurality of electrode fingers 61b.
  • the plurality of electrode fingers 61a and 61b are formed along a direction orthogonal to the elastic wave propagation direction (X-axis direction).
  • the IDT electrode 54 which is composed of a plurality of electrode fingers 61a and 61b and busbar electrodes 62a and 62b, has a laminated structure of an adhesion layer 540 and a main electrode layer 542, as shown in (b) of FIG. 3A. It's becoming
  • the adhesion layer 540 is a layer for improving adhesion between the substrate 50 and the main electrode layer 542, and is made of Ti, for example.
  • the material of the main electrode layer 542 is, for example, Al containing 1% Cu.
  • Protective layer 55 is formed to cover comb electrodes 60a and 60b.
  • the protective layer 55 is a layer for the purpose of protecting the main electrode layer 542 from the external environment, adjusting frequency temperature characteristics, and increasing moisture resistance. is.
  • the materials forming the adhesion layer 540, the main electrode layer 542 and the protective layer 55 are not limited to the materials described above.
  • the IDT electrode 54 may not have the laminated structure described above.
  • the IDT electrode 54 may be composed of, for example, metals or alloys such as Ti, Al, Cu, Pt, Au, Ag, and Pd, and may be composed of a plurality of laminates composed of the above metals or alloys. may Also, the protective layer 55 may not be formed.
  • the substrate 50 includes a high acoustic velocity supporting substrate 51, a low acoustic velocity film 52, and a piezoelectric film 53.
  • the high acoustic velocity supporting substrate 51, the low acoustic velocity film 52, and the piezoelectric film 53 are It has a structure laminated in this order.
  • the piezoelectric film 53 is, for example, a ⁇ ° Y-cut X-propagation LiTaO 3 piezoelectric single crystal or a piezoelectric ceramic (lithium tantalate single crystal cut along a plane normal to an axis rotated ⁇ ° from the Y axis with the X axis as the central axis, (or ceramics, single crystal or ceramics in which surface acoustic waves propagate in the X-axis direction). Note that the material of the piezoelectric single crystal used as the piezoelectric film 53 and the cut angle ⁇ are appropriately selected according to the required specifications of each filter.
  • the high acoustic velocity support substrate 51 is a substrate that supports the low acoustic velocity film 52 , the piezoelectric film 53 and the IDT electrodes 54 .
  • the high acoustic velocity support substrate 51 is a substrate in which the acoustic velocity of bulk waves in the high acoustic velocity support substrate 51 is faster than acoustic waves such as surface waves and boundary waves propagating through the piezoelectric film 53, and surface acoustic waves are generated. It functions so that it is confined in the portion where the piezoelectric film 53 and the low sound velocity film 52 are laminated and does not leak below the high sound velocity support substrate 51 .
  • the high acoustic velocity support substrate 51 is, for example, a silicon substrate.
  • the low sound velocity film 52 is a film in which the sound velocity of the bulk wave in the low sound velocity film 52 is lower than that of the bulk wave propagating through the piezoelectric film 53 , and is arranged between the piezoelectric film 53 and the high sound velocity support substrate 51 . be.
  • This structure and the nature of the elastic wave to concentrate its energy in a low-temperature medium suppresses leakage of the surface acoustic wave energy to the outside of the IDT electrode.
  • the low-temperature velocity film 52 is, for example, a film whose main component is silicon dioxide.
  • the laminated structure of the substrate 50 it is possible to significantly increase the Q value at the resonance frequency and anti-resonance frequency compared to the conventional structure using a single layer piezoelectric substrate. That is, since an acoustic wave resonator with a high Q value can be configured, it is possible to configure a filter with a small insertion loss using the acoustic wave resonator.
  • the high acoustic velocity support substrate 51 has a structure in which a support substrate and a high acoustic velocity film having a higher acoustic velocity than elastic waves such as surface waves and boundary waves propagating through the piezoelectric film 53 are laminated.
  • the support substrate includes piezoelectric materials such as sapphire, lithium tantalate, lithium niobate, and quartz, alumina, magnesia, silicon nitride, aluminum nitride, silicon carbide, zirconia, cordierite, mullite, steatite, and fort.
  • the high acoustic velocity film includes aluminum nitride, aluminum oxide, silicon carbide, silicon nitride, silicon oxynitride, DLC film, diamond, media containing these materials as main components, and media containing mixtures of these materials as main components. etc., various high acoustic velocity materials can be used.
  • FIG. 3B is a cross-sectional view schematically showing a second example of elastic wave resonators forming filters 11 and 12 according to the embodiment.
  • the elastic wave resonator 60 shown in FIG. 3A shows an example in which the IDT electrodes 54 are formed on the substrate 50 having the piezoelectric film 53.
  • the substrate on which the IDT electrodes 54 are formed is shown in FIG. 3B.
  • the piezoelectric single crystal substrate 57 may be a single piezoelectric layer.
  • the piezoelectric single crystal substrate 57 is composed of, for example, a piezoelectric single crystal of LiNbO 3 .
  • the acoustic wave resonator according to this example is composed of a piezoelectric single crystal substrate 57 of LiNbO 3 , an IDT electrode 54 , and a protective layer 58 formed on the piezoelectric single crystal substrate 57 and the IDT electrode 54 . .
  • the piezoelectric film 53 and the piezoelectric single crystal substrate 57 described above may be appropriately changed in laminated structure, material, cut angle, and thickness according to the required transmission characteristics of the elastic wave filter device. Even an elastic wave resonator using a LiTaO 3 piezoelectric substrate having a cut angle other than the cut angle described above can produce the same effects as the elastic wave resonator 60 using the piezoelectric film 53 described above.
  • the substrate on which the IDT electrodes 54 are formed may have a structure in which a supporting substrate, an energy trapping layer, and a piezoelectric film are laminated in this order.
  • An IDT electrode 54 is formed on the piezoelectric film.
  • the piezoelectric film is, for example, LiTaO 3 piezoelectric single crystal or piezoelectric ceramics.
  • the support substrate is the substrate that supports the piezoelectric film, the energy confinement layer, and the IDT electrodes 54 .
  • the energy confinement layer consists of one or more layers, and the velocity of the bulk acoustic wave propagating through at least one layer is greater than the velocity of the elastic wave propagating near the piezoelectric film.
  • the energy trapping layer may have a laminated structure of a low acoustic velocity layer and a high acoustic velocity layer.
  • the sound velocity layer is a film in which the sound velocity of bulk waves in the sound velocity layer is lower than the sound velocity of elastic waves propagating through the piezoelectric film.
  • the high acoustic velocity layer is a film in which the acoustic velocity of bulk waves in the high acoustic velocity layer is higher than the acoustic velocity of elastic waves propagating through the piezoelectric film.
  • the support substrate may be a high acoustic velocity layer.
  • the energy trapping layer may be an acoustic impedance layer having a configuration in which a low acoustic impedance layer with a relatively low acoustic impedance and a high acoustic impedance layer with a relatively high acoustic impedance are alternately laminated. .
  • the wavelength of the elastic wave resonator is defined by the wavelength ⁇ which is the repetition period of the plurality of electrode fingers 61a or 61b forming the IDT electrode 54 shown in (b) of FIG. 3A.
  • the electrode finger pitch is 1/2 of the wavelength ⁇
  • the line width of the electrode fingers 61a and 61b constituting the comb-shaped electrodes 60a and 60b is W
  • the distance between the adjacent electrode fingers 61a and 61b is When the space width is S, it is defined as (W+S).
  • S space width
  • the intersecting width L of the pair of comb-shaped electrodes 60a and 60b is the overlap of the electrode fingers 61a and 61b when viewed from the elastic wave propagation direction (X-axis direction). is the length of the electrode finger that
  • the electrode duty of each acoustic wave resonator is the line width occupation ratio of the plurality of electrode fingers 61a and 61b, and is the ratio of the line width to the sum of the line width and space width of the plurality of electrode fingers 61a and 61b. and is defined as W/(W+S).
  • the height of the comb electrodes 60a and 60b is h.
  • electrode parameters related to the shape of the IDT electrodes of the acoustic wave resonator such as the wavelength ⁇ , the electrode finger pitch, the crossing width L, the electrode duty, and the height h of the IDT electrodes 54, are defined as electrode parameters.
  • the electrode finger pitch of the IDT electrodes 54 is defined by the average electrode finger pitch of the IDT electrodes 54 .
  • the average electrode finger pitch of the IDT electrode 54 is defined by the total number of the electrode fingers 61a and 61b included in the IDT electrode 54 being Ni, and the electrode finger positioned at one end of the IDT electrode 54 in the elastic wave propagation direction and It is defined as Di/(Ni-1), where Di is the center-to-center distance from the positioned electrode finger.
  • FIG. 3C is a cross-sectional view schematically showing a third example of elastic wave resonators forming filters 11 and 12 according to the embodiment.
  • Bulk acoustic wave resonators are shown as acoustic wave resonators of filters 11 and 12 in FIG. 3C.
  • the bulk acoustic wave resonator has, for example, a support substrate 65, a lower electrode 66, a piezoelectric layer 67, and an upper electrode 68. , a piezoelectric layer 67, and an upper electrode 68 are laminated in this order.
  • the support substrate 65 is a substrate for supporting the lower electrode 66, the piezoelectric layer 67, and the upper electrode 68, and is, for example, a silicon substrate.
  • the support substrate 65 is provided with a cavity in a region in contact with the lower electrode 66 . This allows the piezoelectric layer 67 to vibrate freely.
  • the lower electrode 66 is an example of a first electrode and is formed on one surface of the support substrate 65 .
  • the upper electrode 68 is an example of a second electrode and is formed on one surface of the support substrate 65 .
  • the lower electrode 66 and the upper electrode 68 are made of Al containing 1% Cu, for example.
  • the piezoelectric layer 67 is formed between the lower electrode 66 and the upper electrode 68 .
  • the piezoelectric layer 67 is made of, for example, ZnO (zinc oxide), AlN (aluminum nitride), PZT (lead zirconate titanate), KN (potassium niobate), LN (lithium niobate), LT (lithium tantalate),
  • the main component is at least one of quartz and LiBO (lithium borate).
  • the bulk acoustic wave resonator having the above laminated structure induces a bulk acoustic wave in the piezoelectric layer 67 by applying electrical energy between the lower electrode 66 and the upper electrode 68 to generate resonance. It is.
  • a bulk acoustic wave generated by this bulk acoustic wave resonator propagates between the lower electrode 66 and the upper electrode 68 in a direction perpendicular to the film surface of the piezoelectric layer 67 . That is, the bulk acoustic wave resonator is a resonator that utilizes bulk acoustic waves.
  • FIG. 4 is a circuit configuration diagram of a high-frequency circuit 500 according to a comparative example, and a Smith chart showing the impedance of each filter viewed from a switch selection terminal.
  • the high frequency circuit 500 includes filters 11, 502 and 20, a switch circuit 30, an inductor 90, a power amplifier 41, a low noise amplifier 42, an antenna connection terminal 100, An input terminal 110 , an output terminal 120 and an input/output terminal 130 are provided.
  • a high frequency circuit 500 according to the comparative example differs from the high frequency circuit 1 according to the embodiment only in that a filter 502 is arranged instead of the filter 12 .
  • the high-frequency circuit 500 according to the comparative example will be described below, focusing on the points different from the high-frequency circuit 1 according to the embodiment.
  • the filter 11 has a passband including band A for TDD.
  • Filter 11 has one or more elastic wave resonators.
  • the filter 502 has a passband including band A for TDD.
  • Filter 502 has one or more elastic wave resonators.
  • Filter 502 has a configuration without inductor 250 for the circuit configuration of filter 12 shown in FIG. 2B.
  • Filter 502 includes series arm resonators 201, 202, 203, 204 and 205, parallel arm resonators 251, 252, 253, 254 and 255, and input/output terminals 121 and 122, for example.
  • Each of the series arm resonators 201-205 and the parallel arm resonators 251-255 is an elastic wave resonator. That is, in filter 502, no inductor is connected in series to parallel arm resonator 251, which is connected closest to switch circuit 30 among parallel arm resonators 251-255.
  • the filter 20 has a passband that includes at least part of band B that is different from band A.
  • Filter 20 has one or more elastic wave resonators.
  • the power amplifier 41 is connected between the input terminal 110 and the filter 11 and amplifies the band A transmission signal.
  • Low noise amplifier 42 is connected between filter 502 and output terminal 120 and amplifies the received band A signal.
  • the filter 11 is connected between the terminal 30b and the input terminal 110.
  • Filter 502 is connected between terminal 30 c and output terminal 120 .
  • Filter 20 is connected between terminal 30 d and input/output terminal 130 .
  • the high-frequency circuit 500 can simultaneously transmit a transmission signal of band A and a signal of band B, and can simultaneously transmit a reception signal of band A and a signal of band B. is.
  • FIG. 4 shows a Smith chart representing the impedance of the filter 11 viewed from the terminal 30b (S1).
  • FIG. 4(c) shows a Smith chart representing the impedance of the filter 502 viewed from the terminal 30c (S2).
  • the passband of another filter may be referred to as the other band instead of the passband (own band) of the filter itself.
  • the own band of filter 11 is band A
  • the other band of filter 11 is band B.
  • a transmission filter 11 and a reception filter 502 are arranged for transmitting the TDD band A transmission signal and the reception signal. This is because the optimum value of the output impedance of the power amplifier 41 and the optimum value of the input impedance of the low noise amplifier 42 are different, so it is necessary to differentiate the impedance characteristics of the filters connected to the power amplifier 41 and the low noise amplifier 42. due to that.
  • the impedance when looking at the filter 11 from S1 ((b) in FIG. 4) and the impedance when looking at the filter 502 from S2 ((c) in FIG. 4) are different from the impedance of the other band (band B).
  • the phases differ by 4 degrees or more.
  • the impedance of the band B when the filters 11 and 20 are viewed from the terminal 30a and the reception signal of the band A and the band B signal are transmitted at the same time the impedance of the band B when the filters 502 and 20 are viewed from the terminal 30a is greatly different.
  • FIG. 5 is a circuit configuration diagram of the high-frequency circuit 1 according to the embodiment, and a Smith chart showing the impedance of each filter viewed from the switch selection terminal.
  • the circuit configuration of the high-frequency circuit 1 is shown in (a) of FIG.
  • FIG. 5 shows a Smith chart representing the impedance of the filter 11 viewed from the terminal 30b (S1). Further, FIG. 5(c) shows a Smith chart representing the impedance of the filter 12 viewed from the terminal 30c (S2). Further, (d) of FIG. 5 shows a Smith chart representing the impedance of the filter 20 viewed from the terminal 30d (S3).
  • a transmission filter 11 and a reception filter 12 are arranged for transmission of TDD band A transmission signals and reception signals. This is because the optimum value of the output impedance of the power amplifier 41 and the optimum value of the input impedance of the low noise amplifier 42 are different, so it is necessary to differentiate the impedance characteristics of the filters connected to the power amplifier 41 and the low noise amplifier 42. due to that.
  • the impedance of the filter 11 viewed from S1 ((b) in FIG. 5)
  • the phase of the impedance of the other band (band B) is aligned with the impedance ((c) of FIG. 5).
  • an inductor connected in series with a parallel arm resonator can change the phase width of the impedance in the attenuation band without changing the impedance in the filter's own band. is.
  • inductor 250 is connected in series with parallel arm resonator 251 connected closest to switch circuit 30 (terminal 30c). It is possible to change the phase width of the impedance of the other band (band B) when the filter 12 is viewed from S2 without changing the impedance when the filter 12 is viewed from the input terminal.
  • the filter 11 is impedance-matched with the power amplifier 41 and the filter 12 is impedance-matched with the low-noise amplifier 42, the impedance phase of the band B when the filter 11 is viewed from S1 and the filter 12 from S2 It is possible to align the phase of the impedance of the band B seen from .
  • FIG. 6A is a circuit configuration diagram of the high-frequency circuit 1 according to the embodiment, and a Smith chart showing impedance when the filters 11 and 20 commonly connected from the terminal 30a and the antenna connection terminal 100 are viewed.
  • (b) of FIG. 6A shows the impedances of bands A and B when the filters 11 and 20 are viewed from the terminal 30a (S4) when the filters 11 and 20 are connected to the terminal 30a.
  • FIG. 6A (c) shows the impedances of the bands A and B when the filters 11 and 20 are viewed from the antenna connection terminal 100 when the filters 11 and 20 are connected to the terminal 30a.
  • FIG. 6B is a circuit configuration diagram of the high-frequency circuit 1 according to the embodiment, and a Smith chart showing impedance when the filters 12 and 20 commonly connected from the terminal 30a and the antenna connection terminal 100 are viewed.
  • (b) of FIG. 6B shows impedances of bands A and B when the filters 12 and 20 are viewed from the terminal 30a (S4) when the filters 12 and 20 are connected to the terminal 30a.
  • FIG. 6B (c) shows the impedances of the bands A and B when the filters 12 and 20 are viewed from the antenna connection terminal 100 when the filters 12 and 20 are connected to the terminal 30a.
  • the impedance of the own band (band A) of the filter 12 viewed from the terminal 30c is It is possible to adjust the phase width of the impedance of the other band (band B) without shifting the . Therefore, the number of phase adjustment circuits arranged between the filters 11 and 12 and the switch circuit 30 is eliminated to reduce the difference in impedance matching that occurs when switching between the transmission signal and the reception signal of the band A. , and the B-band signal can be transmitted simultaneously. In other words, since the number of phase adjustment circuits can be reduced, it is possible to reduce the transmission loss when simultaneously transmitting the high-frequency signal of band A and the high-frequency signal of band B for TDD.
  • the inductor 250 is connected in series to the parallel arm resonator 251 of the filters 11 and 12 that is connected closest to the switch circuit 30 in the filter 12 . According to this, it is possible to reduce the number of phase adjustment circuits arranged in the receiving path in which the filter 12 is arranged, so that the transmission loss of the received signal transmitted through the receiving path can be reduced. and a signal of band B are simultaneously transmitted, deterioration of reception sensitivity of band A can be further suppressed.
  • FIG. 7 is a circuit configuration diagram of a high-frequency circuit 1A according to Modification 1, and a Smith chart showing the impedance of each filter viewed from the switch selection terminal.
  • the high frequency circuit 1A includes filters 11, 12 and 20, a switch circuit 30, inductors 91, 92 and 93, a matching circuit 70, a power amplifier 41, a low noise amplifier 42 , an antenna connection terminal 100 , an input terminal 110 , an output terminal 120 and an input/output terminal 130 .
  • a high-frequency circuit 1A according to this modification differs from the high-frequency circuit 1 according to the embodiment in that inductors 91 to 93 are added, and that a matching circuit 70 is arranged instead of the inductor 90. is different.
  • the description of the same points as those of the high-frequency circuit 1 according to the embodiment will be omitted, and the points different from the high-frequency circuit 1 will be mainly described.
  • the inductor 91 is an example of a third inductor, and is connected between the first series arm path connecting the terminal 30b and the filter 11 and the ground.
  • the inductor 92 is an example of a fourth inductor, and is connected between the second series arm path connecting the terminal 30c and the filter 12 and the ground.
  • the inductor 93 is an example of a fifth inductor, and is connected between the third series arm path connecting the terminal 30d and the filter 20 and the ground.
  • the matching circuit 70 is connected between the antenna connection terminal 100 and the terminal 30a.
  • Matching circuit 70 includes inductor 94 and capacitor 80 .
  • the inductor 94 is an example of a sixth inductor, and is arranged in series in the antenna connection path connecting the antenna connection terminal 100 and the terminal 30a.
  • the capacitor 80 is connected between the antenna connection path connecting the antenna connection terminal 100 and the terminal 30a and the ground.
  • the antenna connection terminal 100 is connected to the terminal 30a via the matching circuit 70.
  • Filter 11 is connected between terminal 30b and input terminal 110 .
  • Filter 12 is connected between terminal 30 c and output terminal 120 .
  • Filter 20 is connected between terminal 30 d and input/output terminal 130 .
  • the high frequency circuit 1A need not include the power amplifier 41, the low noise amplifier 42, and the matching circuit 70.
  • the high-frequency circuit 1A can simultaneously transmit a transmission signal of band A and a signal of band B, and can simultaneously transmit a reception signal of band A and a signal of band B. is.
  • FIG. 7 shows a Smith chart representing the impedance of the filter 11 viewed from the terminal 30b (S1). Further, (c) of FIG. 7 shows a Smith chart representing the impedance of the filter 12 viewed from the terminal 30c (S2). FIG. 7(d) shows a Smith chart representing the impedance of the filter 20 viewed from the terminal 30d (S3).
  • a transmission filter 11 and a reception filter 12 are arranged for transmitting transmission signals and reception signals of band A of TDD. This is because the optimum value of the output impedance of the power amplifier 41 and the optimum value of the input impedance of the low noise amplifier 42 are different, so it is necessary to differentiate the impedance characteristics of the filters connected to the power amplifier 41 and the low noise amplifier 42. due to that.
  • the impedance ((b) in FIG. 7) when the filter 11 is viewed from S1 and the impedance when the filter 12 is viewed from S2 The impedance ((c) of FIG. 7) is in phase with the impedance of the other band (band B).
  • the filter 12 that the filter 12 has.
  • the inductor 250 is connected in series with the parallel arm resonator 251 that is connected closest to the switch circuit 30 (terminal 30c)
  • the output impedance of the filter 12 (the input of the low noise amplifier 42 It is possible to change the phase width of the impedance of the other band (band B) when the filter 12 is viewed from S2 without changing the impedance when the filter 12 is viewed from the terminal.
  • the filter 11 is impedance-matched with the power amplifier 41 and the filter 12 is impedance-matched with the low-noise amplifier 42, the impedance phase of the band B when the filter 11 is viewed from S1 and the filter 12 from S2 It is possible to align the phase of the impedance of the band B seen from .
  • the impedance of the partner band (band B) when the filter 11 is viewed from S1 the impedance of the partner band (band B) when the filter 12 is viewed from S2, and the partner band (band A) when the filter 20 is viewed from S3 Due to the arrangement of inductors 91, 92 and 93, the impedance is shifted counterclockwise on the equal conductance circle compared to the high frequency circuit 1 according to the embodiment. Therefore, the impedance of the partner band (band B) when looking at the filter 11 from S1, the impedance of the partner band (band B) when looking at the filter 12 from S2, and the partner band (band A) when looking at the filter 20 from S3 is shifted from the capacitive region to a more open region.
  • the reflection loss of band B when the filter 11 alone is viewed from the terminal 30b, the reflection loss of the band B when the filter 12 alone is viewed from the terminal 30c, and the band A reflection loss when the filter 20 is viewed from the terminal 30d can be reduced.
  • FIG. 8A is a circuit configuration diagram of a high-frequency circuit 1A according to Modification 1, and a Smith chart showing the impedance of the filters 11 and 20 commonly connected from the antenna connection terminal 100.
  • FIG. 8B is a circuit configuration diagram of the high-frequency circuit 1A according to Modification 1, and a Smith chart showing the impedance of the filters 12 and 20 commonly connected from the antenna connection terminal 100.
  • the impedance of band B when the filters 11 and 20 are viewed from the terminal 30a and the reception signal of band A are and a signal of band B are simultaneously transmitted the impedance of band B when filters 12 and 20 are viewed from terminal 30a can be substantially matched. As a result, it is possible to suppress an increase in the bit error rate in the received signal of band A when the received signal of band A and the signal of band B are simultaneously transmitted.
  • the number of phase adjustment circuits arranged between the filters 11 and 12 and the switch circuit 30 is eliminated to reduce the difference in impedance matching that occurs when switching between the transmission signal and the reception signal of the band A. , and the B-band signal can be transmitted simultaneously. In other words, since the number of phase adjustment circuits can be reduced, it is possible to reduce the transmission loss when simultaneously transmitting the high-frequency signal of band A and the high-frequency signal of band B for TDD.
  • the inductors 91 to 93 provide a reflection loss for the band B when the filter 11 alone is viewed from the terminal 30b, a reflection loss for the band B when the filter 12 alone is viewed from the terminal 30c, and a filter 20 from the terminal 30d. Since the reflection loss of the visible band A is reduced, the signal transmission loss when simultaneously transmitting the TDD band A signal and the band B signal can be further reduced.
  • FIG. 9 is a circuit configuration diagram of a high frequency circuit 1B according to Modification 2.
  • the high frequency circuit 1B includes filters 11, 12 and 20, a switch circuit 30, inductors 91, 92, 93 and 95, a power amplifier 41, a low noise amplifier 42, an antenna connection terminal 100 , an input terminal 110 , an output terminal 120 , and an input/output terminal 130 .
  • a high-frequency circuit 1B according to the present modification differs from the high-frequency circuit 1 according to the embodiment in that inductors 91 to 93 and 95 are added and inductor 90 is not added.
  • the description of the same points as the high-frequency circuit 1 according to the embodiment will be omitted, and the points different from the high-frequency circuit 1 will be mainly described.
  • the inductor 95 is an example of a seventh inductor, and is arranged in series in the first series arm path connecting the terminal 30 b and the filter 11 .
  • the inductor 91 is an example of an eighth inductor, and is connected between the first series arm path connecting the terminal 30b and the filter 11 and the ground.
  • the inductor 92 is an example of a ninth inductor, and is connected between the second series arm path connecting the terminal 30c and the filter 12 and the ground.
  • the inductor 93 is connected between the third series arm path connecting the terminal 30d and the filter 20 and the ground.
  • the high frequency circuit 1B need not include the power amplifier 41 and the low noise amplifier 42.
  • the high-frequency circuit 1B can simultaneously transmit a transmission signal of band A and a signal of band B, and can simultaneously transmit a reception signal of band A and a signal of band B. is.
  • FIG. 10 is a Smith chart showing the impedance of the filter 11 viewed from each point on the transmission path, and a Smith chart showing the impedance of the filter 12 viewed from each point on the reception path in the high-frequency circuit 1B according to Modification 2. Chart.
  • signals that can be transmitted simultaneously with band A signals include not only band B signals, but also band C, band D, band E, band F, and band G signals.
  • the filters connected to the switch circuit 30 may include filters having the band C to band G as their passbands, in addition to the filters 11, 12, and 20.
  • band A for example, band 41 of 4G-LTE or band n41 of 5G-NR is applied.
  • band B for example, band 40 of 4G-LTE or band n40 of 5G-NR is applied.
  • Band C is applied to band 1 of 4G-LTE or band n1 (downlink operating band) of 5G-NR.
  • Band D is applied to band 1 of 4G-LTE or band n1 (uplink operation band) of 5G-NR.
  • Band E is applied to band 3 of 4G-LTE or band n3 of 5G-NR (downlink operating band: 1805-1880 MHz).
  • Band F applies to 4G-LTE band 3 or 5G-NR band n3 (uplink operating band: 1710-1785 MHz).
  • Band G is applied to 4G-LTE band 32 or 5G-NR band n32 (downlink operating band).
  • the frequencies are band A, band B, band C, band D, band E, band F, and band G from the highest.
  • FIG. 10(a) shows a Smith chart representing the impedance of the filter 11 viewed from S6 (shown in FIG. 9). Since this impedance is optimally matched with the output impedance of the power amplifier 41, the impedance of band A is located in a capacitive region and a high impedance region than the reference impedance, and the concentration of the impedance of band A is high. In addition, the impedance of the other band (band B to band G) on the lower frequency side than the own band (band A) is located in the capacitive area, and the lower the frequency side, the closer to the open area.
  • FIG. 10(b) shows a Smith chart representing the impedance of the filter 11 viewed from S7 (shown in FIG. 9).
  • the inductance value of inductor 95 is L 95
  • the impedance looking at filter 11 from S7 is shifted clockwise by j ⁇ L 95 on the equal resistance circle compared to the impedance looking at filter 11 from S6. .
  • the impedance of its own band (band A) in the filter 11 is positioned near the reference impedance.
  • the amount of shift in band B located on the high frequency side becomes larger than the amount of shift in band G located on the low frequency side.
  • the addition of the inductor 95 shifts the impedance of the own band (band A) toward the reference impedance side, and the impedance of the other band (bands B to G) shifts toward the short region while increasing the phase width.
  • FIG. 10(c) shows a Smith chart representing the impedance of the filter 11 viewed from S1 (shown in FIG. 9).
  • the inductance value of the inductor 91 is L 91
  • the impedance of the filter 11 viewed from S1 is reflected from the impedance of the filter 11 viewed from S7 on the equal conductance circle by an amount corresponding to 1/j ⁇ L 91 . Shift clockwise. As a result, the shift amount of the band G located on the low frequency side becomes larger than the shift amount of the band B located on the high frequency side.
  • the addition of the inductor 91 shifts the impedance of the own band (band A) toward the reference impedance side, and the impedance of the other band (bands B to G) shifts toward the open region, while the phase width further increases. .
  • FIG. 10 shows a Smith chart representing the impedance of the filter 12 viewed from S8 (shown in FIG. 9).
  • the band A impedance is located in the capacitive region rather than the reference impedance.
  • the impedance of the other band (band B to band G) on the lower frequency side than the own band (band A) is located in the capacitive area, and the lower the frequency side, the closer to the open area.
  • the inductor 250 is arranged in the filter 12, by adjusting the inductance value L 250 of the inductor 250, it is possible to change the phase width of the impedance of the other band (bands B to G).
  • L 250 when L 250 is relatively large, the phase width of the impedance of the other band (bands B to G) is large.
  • (d') of FIG. 10 when L 250 is relatively small, the phase width of the impedance of the partner band (bands B to G) is small.
  • FIG. 10(e) shows a Smith chart representing the impedance of the filter 12 viewed from S2 (shown in FIG. 9).
  • the impedance of the filter 12 viewed from S2 is a reflection of 1/j ⁇ L 92 on the equal conductance circle compared to the impedance of the filter 12 viewed from S8. Shift clockwise.
  • the shift amount of the band G located on the low frequency side becomes larger than the shift amount of the band B located on the high frequency side.
  • the addition of the inductor 92 shifts the impedance of the own band (band A) toward the reference impedance side, and the impedance of the other band (bands B to G) shifts toward the open region while increasing the phase width.
  • the impedance of the own band (band A) of the filter 11 alone is shifted to the vicinity of the reference impedance by the inductor 95 connected in series and the inductor 91 connected in parallel, and the impedance of the own band (band A) of the filter 12 alone is changed to , is shifted to the vicinity of the reference impedance by an inductor 92 connected in parallel.
  • the inductor 250 of the filter 12 adjusts the own band (band The phase width of the impedance of the other band (bands B to G) is adjusted without shifting the impedance of A).
  • the number of inductors (phase adjustment circuits) connected in series to the filter 12 is reduced, and the difference in impedance matching that occurs when switching between the transmission signal and the reception signal of band A is reduced. Simultaneous transmission of signals or received signals and signals of bands B to G is possible.
  • the high-frequency circuit 1 includes the antenna connection terminal 100, the input terminal 110 and the output terminal 120, the filter 11 having a passband including the transmission band of band A for TDD, a filter 12 having a passband including the receive band; a filter 20 having a passband including at least a portion of band B different from band A; terminals 30a, 30b, 30c and 30d; and connection of terminals 30a and 30d, and connection of terminals 30a and 30c and connection of terminals 30a and 30d.
  • Antenna connection terminal 100 is connected to terminal 30a
  • filter 11 is connected between terminal 30b and input terminal 110
  • filter 12 is connected between terminal 30c and output terminal 120
  • filter 20 is connected to terminal 30d.
  • One of filters 11 and 12 includes a first elastic wave resonator connected between a series arm path connecting switch circuit 30 and one of input terminal 110 and output terminal 120 and ground, and the series arm path and an inductor 250 connected in series with the first acoustic wave resonator between and ground, wherein the first acoustic wave resonator is one of the parallel arm resonators connected between the series arm path and the ground is connected to the switch circuit 30 at the closest position.
  • the impedance of the other band can be adjusted without shifting the impedance of the one's own band.
  • the number of phase adjustment circuits arranged between the filters 11 and 12 and the switch circuit 30 is eliminated to reduce the difference in impedance matching that occurs when switching between the transmission signal and the reception signal of the band A. , and the B-band signal can be transmitted simultaneously.
  • the number of phase adjustment circuits can be reduced, it is possible to reduce the transmission loss when simultaneously transmitting the high-frequency signal of band A and the high-frequency signal of band B for TDD.
  • one of the filters 11 and 12 is the filter 12, and the first acoustic wave resonator is connected between a series arm path connecting the terminal 30c and the output terminal 120 and the ground.
  • the other of the filters 11 and 12 may not include the inductor 250 .
  • the high-frequency circuit 1 may further include an inductor 90 connected between the antenna connection path connecting the antenna connection terminal 100 and the terminal 30a and the ground.
  • the impedances of the bands A and B when the filters 11 and 20 are viewed from the terminal 30a and the impedances of the bands A and B when the filters 12 and 20 are viewed from the terminal 30a are both measured in the counterclockwise direction of the isoconductance circle. to match the reference impedance.
  • the high-frequency circuit 1A further includes an inductor 91 connected between a first series arm path connecting the terminal 30b and the filter 11 and the ground, and a second series arm path connecting the terminal 30c and the filter 12.
  • An inductor 92 connected between the ground, an inductor 93 connected between the third series arm path connecting the terminal 30d and the filter 20 and the ground, and an inductor 93 connected between the antenna connection terminal and the terminal 30a. and a matching circuit 70 .
  • inductors 91 to 93 provide reflection loss for band B when the filter 11 alone is viewed from the terminal 30b, reflection loss for band B when the filter 12 alone is viewed from the terminal 30c, and band A when the filter 20 is viewed from the terminal 30d. Since the reflection loss is reduced, it is possible to further reduce the signal transmission loss in the simultaneous transmission of the band A signal and the band B signal for TDD.
  • the matching circuit 70 includes an inductor 94 arranged in series in an antenna connection path connecting the antenna connection terminal 100 and the terminal 30a, an antenna connection path connecting the antenna connection terminal 100 and the terminal 30a, and a ground. and a capacitor 80 connected between.
  • the impedances of the bands A and B when the filters 11 and 20 are viewed from the terminal 30a and the impedances of the bands A and B when the filters 12 and 20 are viewed from the terminal 30a are formed into equal resistance circles by the inductor 94.
  • the clockwise shift and the clockwise shift of the equiconductance circle by the capacitor 80 allows it to be matched to the reference impedance.
  • the high-frequency circuit 1B further includes an inductor 95 arranged in series in a first series arm path connecting the terminal 30b and the filter 11, and an inductor 95 between the first series arm path connecting the terminal 30b and the inductor 95 and the ground. and an inductor 92 connected between a second series arm path connecting the terminal 30c and the filter 12 and the ground.
  • the band A impedance of filter 11 alone is shifted to the vicinity of the reference impedance by inductors 95 and 91
  • the band A impedance of filter 12 alone is shifted to the vicinity of the reference impedance by inductor 92 .
  • the inductor 250 of the filter 12 is used to adjust the impedance of the band A without shifting the impedance of the band A.
  • the phase width of the impedance of B to G can be adjusted.
  • the number of inductors (phase adjustment circuits) connected in series with the filter 12 is reduced to reduce the difference in impedance matching that occurs when switching between the transmission signal and the reception signal of band A. It is possible to simultaneously transmit the signal and the signals of the bands B to G.
  • the inductor 250 may be connected between the first acoustic wave resonator and the ground.
  • the filter 11 includes one or more series arm resonators arranged in a series arm path connecting the input terminal 110 and the terminal 30b, and one resonator connected between the series arm path and the ground. and the above parallel arm resonator.
  • the filter 12 is connected between one or more series arm resonators arranged in a series arm path connecting the output terminal 120 and the terminal 30c and between the series arm path and the ground. and one or more parallel arm resonators.
  • the terminals 30a and 30b when transmitting a band A transmission signal and a band B signal at the same time, the terminals 30a and 30b may be connected, and the terminals 30a and 30d may be connected.
  • the terminals 30a and 30c when the received signal of band A and the signal of band B are transmitted simultaneously, the terminals 30a and 30c may be connected, and the terminals 30a and 30d may be connected.
  • the high frequency circuit 1 further includes a power amplifier 41 connected between the input terminal 110 and the filter 11, and a low noise amplifier 42 connected between the filter 12 and the output terminal 120. good too.
  • the band B may be located on the lower frequency side than the band A.
  • the communication device 4 includes an RFIC 3 that processes high frequency signals, and a high frequency circuit 1 that transmits high frequency signals between the RFIC 3 and the antenna 2 .
  • the effect of the high-frequency circuit 1 can be realized in the communication device 4.
  • matching elements such as inductors and capacitors, and switch circuits may be connected between the components.
  • the inductor may include a wiring inductor that is a wiring that connects each component.
  • the present invention can be widely used in communication equipment such as mobile phones as a low-loss high-frequency circuit that can be applied to multi-band and multi-mode frequency standards.
  • RFIC radio frequency identification circuit
  • 12 RF signal processing circuit
  • 502 filter 30 switch circuit 30a, 30b, 30c, 30d, 30e, 30f terminal 41 power amplifier 42 low noise amplifier 50 substrate 51 high acoustic velocity support substrate 52 low acoustic velocity film 53 piezoelectric film 54 IDT electrode 55, 58 protective layer 57 piezoelectric single crystal substrate 60 elastic wave resonator 60a, 60b comb-shaped electrodes 61a, 61b electrode fingers 62a, 62b busbar electrode 65 support substrate 66 lower electrode 67 piezoelectric layer 68 upper electrode 70 matching circuit 80 capacitor 90, 91, 92, 93, 94, 95, 250 inductor 100 antenna connection terminal 101, 102, 103, 104, 105, 106, 107, 201, 202, 203, 204, 205 series arm resonator 110 input terminal 111, 112, 121, 122, 130 input/output terminal 120 output terminal 151

Abstract

A high frequency circuit (1) comprises: filters (11 and 12) that use a band A for TDD as a pass band; a filter (20) that uses at least a portion of a band B as a pass band; and a switch circuit (30) that can simultaneously execute the connection of terminals (30a and 30b) and the connection of terminals (30a and 30d), or simultaneously execute the connection of terminals (30a and 30c) and the connection of terminals (30a and 30d). The filter (11) is connected to the terminal (30b), the filter (12) is connected to the terminal (30c), and the filter (20) is connected to the terminal (30d). The filter (12) comprises: a first elastic wave resonator connecting the ground and a series arm path connected to the terminal (30c): and an inductor (250) connected in series with the first elastic wave resonator between the series arm path and the ground. Among parallel arm resonators, the first elastic wave resonator is connected closest to the switch circuit (30).

Description

高周波回路および通信装置High frequency circuits and communication equipment
　本発明は、高周波回路および通信装置に関する。 The present invention relates to high frequency circuits and communication devices.
　特許文献１には、複数の周波数帯域（バンド）の信号を同時通信することが可能な高周波フロントエンド回路（高周波回路）が開示されている。具体的には、高周波回路は、アンテナに接続されたスイッチと、スイッチに接続された送信フィルタと、受信フィルタと、伝送フィルタと、スイッチおよび送信フィルタの間、スイッチおよび受信フィルタの間、ならびにスイッチおよび伝送フィルタの間の少なくとも１つに配置された位相調整回路と、を備える。位相調整回路の配置により、複数のバンドの信号を同時通信する場合の通信品質の劣化を抑制することが可能となる。 Patent Document 1 discloses a high-frequency front-end circuit (high-frequency circuit) capable of simultaneous communication of signals in a plurality of frequency bands. Specifically, the high-frequency circuit includes a switch connected to the antenna, a transmit filter connected to the switch, a receive filter, a transmit filter, between the switch and the transmit filter, between the switch and the receive filter, and between the switch and the receive filter. and a phase adjustment circuit positioned at least one between the transmission filters. By arranging the phase adjustment circuit, it is possible to suppress deterioration of communication quality when simultaneously communicating signals of a plurality of bands.
国際公開第２０２０／１５３２８５号WO2020/153285
　特許文献１に開示された高周波回路において、例えば、送信フィルタを通過する信号と伝送フィルタを通過する信号との同時伝送（同時伝送１）、および、（２）受信フィルタを通過する信号と伝送フィルタを通過する信号との同時伝送（同時伝送２）、の双方を実行することが想定される。 In the high-frequency circuit disclosed in Patent Document 1, for example, simultaneous transmission of a signal passing through a transmission filter and a signal passing through a transmission filter (simultaneous transmission 1), and (2) a signal passing through a reception filter and a transmission filter Simultaneous transmission (simultaneous transmission 2) with the signal passing through the .
　しかしながら、特許文献１の高周波回路では、同時伝送１および同時伝送２におけるインピーダンスを調整すべく位相調整回路が配置されているが、スイッチと各フィルタとを結ぶ直列腕経路に配置される位相調整回路が多いほど、高周波回路の伝送損失が増加する。 However, in the high-frequency circuit of Patent Document 1, the phase adjustment circuit is arranged to adjust the impedance in simultaneous transmission 1 and simultaneous transmission 2, but the phase adjustment circuit is arranged in the series arm path connecting the switch and each filter. , the transmission loss of the high-frequency circuit increases.
　そこで、本発明は、上記課題を解決するためになされたものであって、複数のバンドの信号を同時伝送する場合の伝送損失が低減された高周波回路および通信装置を提供することを目的とする。 SUMMARY OF THE INVENTION Accordingly, it is an object of the present invention to provide a high-frequency circuit and a communication device in which transmission loss is reduced when simultaneously transmitting signals of a plurality of bands. .
　上記目的を達成するために、本発明の一態様に係る高周波回路は、アンテナ端子、入力端子および出力端子と、時分割複信（ＴＤＤ）用の第１バンドの送信帯域を含む通過帯域を有する第１フィルタと、第１バンドの受信帯域を含む通過帯域を有する第２フィルタと、第１バンドと異なる第２バンドの少なくとも一部を含む通過帯域を有する第３フィルタと、第１端子、第２端子、第３端子および第４端子を有し、第１端子および第２端子の接続と第１端子および第４端子の接続とを同時に実行可能であり、第１端子および第３端子の接続と第１端子および第４端子の接続とを同時に実行可能であるスイッチ回路と、を備え、アンテナ端子は、第１端子に接続され、第１フィルタは、第２端子と入力端子との間に接続され、第２フィルタは、第３端子と出力端子との間に接続され、第３フィルタは、第４端子に接続され、第１フィルタおよび第２フィルタの一方は、スイッチ回路と入力端子および出力端子の一方とを結ぶ直列腕経路とグランドとの間に接続された第１弾性波共振子と、直列腕経路とグランドとの間で第１弾性波共振子と直列接続された第１インダクタと、を備え、第１弾性波共振子は、直列腕経路とグランドとの間に接続された並列腕共振子のうちでスイッチ回路に最も近い位置で接続されている。 To achieve the above object, a high-frequency circuit according to one aspect of the present invention has an antenna terminal, an input terminal, an output terminal, and a passband including a transmission band of a first band for time division duplex (TDD). a first filter, a second filter having a passband including a reception band of a first band, a third filter having a passband including at least a portion of a second band different from the first band, a first terminal, a second having two terminals, a third terminal and a fourth terminal, capable of simultaneously connecting the first terminal and the second terminal and connecting the first terminal and the fourth terminal, and connecting the first terminal and the third terminal and a switch circuit capable of simultaneously connecting the first terminal and the fourth terminal, wherein the antenna terminal is connected to the first terminal, and the first filter is connected between the second terminal and the input terminal. the second filter is connected between the third terminal and the output terminal; the third filter is connected to the fourth terminal; one of the first filter and the second filter is connected between the switch circuit and the input terminal; A first elastic wave resonator connected between a series arm path connecting one of the output terminals and the ground, and a first inductor connected in series with the first elastic wave resonator between the series arm path and the ground. and the first elastic wave resonator is connected at a position closest to the switch circuit among the parallel arm resonators connected between the series arm path and the ground.
　本発明によれば、複数のバンドの信号を同時伝送する場合の伝送損失が低減された高周波回路および通信装置を提供することが可能となる。 According to the present invention, it is possible to provide a high-frequency circuit and a communication device with reduced transmission loss when simultaneously transmitting signals of multiple bands.
図１は、実施の形態に係る高周波回路および通信装置の回路構成図である。FIG. 1 is a circuit configuration diagram of a high-frequency circuit and a communication device according to an embodiment. 図２Ａは、実施の形態に係る第１フィルタの回路構成例を示す図である。2A is a diagram illustrating a circuit configuration example of a first filter according to the embodiment; FIG. 図２Ｂは、実施の形態に係る第２フィルタの回路構成例を示す図である。2B is a diagram illustrating a circuit configuration example of a second filter according to the embodiment; FIG. 図３Ａは、実施の形態に係る第１フィルタおよび第２フィルタを構成する弾性波共振子の第１例を模式的に表す平面図および断面図である。FIG. 3A is a plan view and a cross-sectional view schematically showing a first example of elastic wave resonators that constitute the first filter and the second filter according to the embodiment. 図３Ｂは、実施の形態に係る第１フィルタおよび第２フィルタを構成する弾性波共振子の第２例を模式的に表す断面図である。FIG. 3B is a cross-sectional view schematically showing a second example of elastic wave resonators forming the first filter and the second filter according to the embodiment. 図３Ｃは、実施の形態に係る第１フィルタおよび第２フィルタを構成する弾性波共振子の第３例を模式的に表す断面図である。FIG. 3C is a cross-sectional view schematically showing a third example of elastic wave resonators forming the first filter and the second filter according to the embodiment. 図４は、比較例に係る高周波回路の回路構成図、およびスイッチ選択端子から各フィルタを見たインピーダンスを示すスミスチャートである。FIG. 4 is a circuit configuration diagram of a high-frequency circuit according to a comparative example, and a Smith chart showing the impedance of each filter viewed from a switch selection terminal. 図５は、実施の形態に係る高周波回路の回路構成図、およびスイッチ選択端子から各フィルタを見たインピーダンスを示すスミスチャートである。FIG. 5 is a circuit configuration diagram of a high-frequency circuit according to the embodiment, and a Smith chart showing the impedance of each filter viewed from a switch selection terminal. 図６Ａは、実施の形態に係る高周波回路の回路構成図、およびスイッチ共通端子およびアンテナ端子から共通接続された第１フィルタおよび第３フィルタを見たインピーダンスを示すスミスチャートである。FIG. 6A is a circuit configuration diagram of a high-frequency circuit according to the embodiment, and a Smith chart showing impedances of a first filter and a third filter commonly connected from a switch common terminal and an antenna terminal; 図６Ｂは、実施の形態に係る高周波回路の回路構成図、およびスイッチ共通端子およびアンテナ端子から共通接続された第２フィルタおよび第３フィルタを見たインピーダンスを示すスミスチャートである。6B is a circuit configuration diagram of the high-frequency circuit according to the embodiment, and a Smith chart showing the impedance of the second filter and the third filter commonly connected from the switch common terminal and the antenna terminal; FIG. 図７は、変形例１に係る高周波回路の回路構成図、およびスイッチ選択端子から各フィルタを見たインピーダンスを示すスミスチャートである。FIG. 7 is a circuit configuration diagram of a high-frequency circuit according to Modification 1, and a Smith chart showing the impedance of each filter viewed from a switch selection terminal. 図８Ａは、変形例１に係る高周波回路の回路構成図、およびアンテナ端子から共通接続された第１フィルタおよび第３フィルタを見たインピーダンスを示すスミスチャートである。FIG. 8A is a circuit configuration diagram of a high-frequency circuit according to Modification 1, and a Smith chart showing impedance when viewing a first filter and a third filter commonly connected from an antenna terminal. 図８Ｂは、変形例１に係る高周波回路の回路構成図、およびアンテナ端子から共通接続された第２フィルタおよび第３フィルタを見たインピーダンスを示すスミスチャートである。8B is a circuit configuration diagram of a high-frequency circuit according to Modification 1, and a Smith chart showing impedance when viewing a second filter and a third filter commonly connected from an antenna terminal; FIG. 図９は、変形例２に係る高周波回路の回路構成図である。FIG. 9 is a circuit configuration diagram of a high-frequency circuit according to Modification 2. As shown in FIG. 図１０は、変形例２に係る高周波回路において、送信経路上の各点から第１フィルタを見たインピーダンスを示すスミスチャート、および、受信経路上の各点から第２フィルタを見たインピーダンスを示すスミスチャートである。FIG. 10 shows a Smith chart showing the impedance of the first filter viewed from each point on the transmission path and the impedance of the second filter viewed from each point on the reception path in the high-frequency circuit according to Modification 2. Smith chart.
　以下、本発明の実施の形態について、実施例、変形例および図面を用いて詳細に説明する。なお、以下で説明する実施例および変形例は、いずれも包括的または具体的な例を示すものである。以下の実施例および変形例で示される数値、形状、材料、構成要素、構成要素の配置および接続形態などは、一例であり、本発明を限定する主旨ではない。以下の実施例および変形例における構成要素のうち、独立請求項に記載されていない構成要素については、任意の構成要素として説明される。また、図面に示される構成要素の大きさまたは大きさの比は、必ずしも厳密ではない。 Hereinafter, embodiments of the present invention will be described in detail using examples, modifications, and drawings. It should be noted that the embodiments and modifications described below are all comprehensive or specific examples. Numerical values, shapes, materials, constituent elements, arrangement of constituent elements, connection forms, and the like shown in the following examples and modifications are examples, and are not intended to limit the present invention. Among components in the following examples and modifications, components not described in independent claims will be described as optional components. Also, the sizes or size ratios of components shown in the drawings are not necessarily exact.
　本開示において、「接続される」とは、接続端子および／または配線導体で直接接続される場合だけでなく、他の回路素子を介して電気的に接続される場合も含むことを意味する。また、「ＡとＢとの間に接続される」、「ＡおよびＢの間に接続される」とは、ＡおよびＢを結ぶ経路上でＡおよびＢと接続されることを意味する。 In the present disclosure, "connected" means not only direct connection with connection terminals and/or wiring conductors, but also electrical connection via other circuit elements. Also, "connected between A and B" and "connected between A and B" mean being connected to A and B on a path connecting A and B.
　また、本開示において、「経路」とは、高周波信号が伝搬する配線、当該配線に直接接続された電極、および当該配線または当該電極に直接接続された端子等で構成された伝送線路であることを意味する。 In addition, in the present disclosure, the term “path” refers to a transmission line composed of a wire through which a high-frequency signal propagates, an electrode directly connected to the wire, and a terminal directly connected to the wire or the electrode. means
　また、本開示において、「部品Ａが経路Ｂに直列配置される」とは、部品Ａの信号入力端および信号出力端の双方が、経路Ｂを構成する配線、電極、または端子に接続されていることを意味する。 In addition, in the present disclosure, “the component A is arranged in series with the path B” means that both the signal input terminal and the signal output terminal of the component A are connected to the wiring, electrodes, or terminals that constitute the path B. means that there is
　また、本開示において、フィルタの通過帯域は、当該通過帯域内における挿入損失の最小値から３ｄＢ大きい２つの周波数間の周波数帯域と定義される。 Also, in this disclosure, the passband of a filter is defined as the frequency band between two frequencies that are 3 dB greater than the minimum value of insertion loss within the passband.
　（実施の形態）
　［１　高周波回路１および通信装置４の回路構成］
　本実施の形態に係る高周波回路１および通信装置４の回路構成について、図１を参照しながら説明する。図１は、実施の形態に係る高周波回路１および通信装置４の回路構成図である。 (Embodiment)
[1 Circuit configuration of high-frequency circuit 1 and communication device 4]
Circuit configurations of a high-frequency circuit 1 and a communication device 4 according to the present embodiment will be described with reference to FIG. FIG. 1 is a circuit configuration diagram of a high frequency circuit 1 and a communication device 4 according to an embodiment.
　［１．１　通信装置４の回路構成］
　まず、通信装置４の回路構成について説明する。図１に示すように、本実施の形態に係る通信装置４は、高周波回路１と、アンテナ２と、ＲＦ信号処理回路（ＲＦＩＣ）３と、を備える。 [1.1 Circuit Configuration of Communication Device 4]
First, the circuit configuration of the communication device 4 will be described. As shown in FIG. 1, the communication device 4 according to the present embodiment includes a high frequency circuit 1, an antenna 2, and an RF signal processing circuit (RFIC) 3.
　高周波回路１は、アンテナ２とＲＦＩＣ３との間で高周波信号を伝送する。高周波回路１の詳細な回路構成については後述する。 The high frequency circuit 1 transmits high frequency signals between the antenna 2 and the RFIC 3 . A detailed circuit configuration of the high-frequency circuit 1 will be described later.
　アンテナ２は、高周波回路１のアンテナ接続端子１００に接続され、高周波回路１から出力された高周波信号（以降、送信信号と記す）を送信し、また、外部から高周波信号（以降、受信信号と記す）を受信して高周波回路１へ出力する。 The antenna 2 is connected to an antenna connection terminal 100 of the high-frequency circuit 1, transmits a high-frequency signal (hereinafter referred to as a transmission signal) output from the high-frequency circuit 1, and receives a high-frequency signal (hereinafter referred to as a received signal) from the outside. ) is received and output to the high-frequency circuit 1 .
　ＲＦＩＣ３は、高周波信号を処理する信号処理回路の一例である。具体的には、ＲＦＩＣ３は、高周波回路１の受信経路を介して入力された受信信号をダウンコンバート等により信号処理し、当該信号処理して生成された受信信号をベースバンド信号処理回路（ＢＢＩＣ、図示せず）へ出力する。また、ＲＦＩＣ３は、ＢＢＩＣから入力された信号をアップコンバート等により信号処理し、当該信号処理して生成された送信信号を、高周波回路１の送信経路に出力する。また、ＲＦＩＣ３は、高周波回路１が有するスイッチおよび増幅素子等を制御する制御部を有する。なお、ＲＦＩＣ３の制御部としての機能の一部または全部は、ＲＦＩＣ３の外部に実装されてもよく、例えば、ＢＢＩＣまたは高周波回路１に実装されてもよい。 The RFIC 3 is an example of a signal processing circuit that processes high frequency signals. Specifically, the RFIC 3 performs signal processing such as down-conversion on the received signal input via the receiving path of the high-frequency circuit 1, and converts the received signal generated by the signal processing into a baseband signal processing circuit (BBIC, not shown). Further, the RFIC 3 performs signal processing such as up-conversion on the signal input from the BBIC, and outputs the transmission signal generated by the signal processing to the transmission path of the high frequency circuit 1 . The RFIC 3 also has a control section that controls the switches and amplification elements of the high-frequency circuit 1 . A part or all of the functions of the RFIC 3 as a control unit may be implemented outside the RFIC 3, for example, in the BBIC or the high-frequency circuit 1. FIG.
　また、ＲＦＩＣ３は、どのバンド（周波数帯域）の信号を伝送するか、および、送信信号および受信信号のいずれを伝送するか、に基づいて、高周波回路１が有するスイッチ回路３０の接続を制御する制御部としての機能を有する。 Further, the RFIC 3 controls the connection of the switch circuit 30 included in the high-frequency circuit 1 based on which band (frequency band) the signal is to be transmitted and which of the transmission signal and the reception signal is to be transmitted. It has a function as a department.
　なお、本実施の形態に係る通信装置４において、アンテナ２は、必須の構成要素ではない。 Note that the antenna 2 is not an essential component in the communication device 4 according to the present embodiment.
　［１．２　高周波回路１の回路構成］
　次に、高周波回路１の回路構成について説明する。図１に示すように、高周波回路１は、フィルタ１１、１２および２０と、スイッチ回路３０と、インダクタ９０と、電力増幅器４１と、低雑音増幅器４２と、アンテナ接続端子１００と、入力端子１１０と、出力端子１２０と、入出力端子１３０と、を備える。 [1.2 Circuit Configuration of High Frequency Circuit 1]
Next, the circuit configuration of the high frequency circuit 1 will be described. As shown in FIG. 1, the high frequency circuit 1 includes filters 11, 12 and 20, a switch circuit 30, an inductor 90, a power amplifier 41, a low noise amplifier 42, an antenna connection terminal 100, and an input terminal 110. , an output terminal 120 and an input/output terminal 130 .
　アンテナ接続端子１００は、アンテナ端子の一例であり、アンテナ２に接続される。入力端子１１０は、ＲＦＩＣ３からバンドＡの送信信号を受ける端子である。出力端子１２０は、バンドＡの受信信号をＲＦＩＣ３へ出力する端子である。入出力端子１３０は、ＲＦＩＣ３からバンドＢの送信信号を受ける、または、バンドＢの受信信号をＲＦＩＣ３へ出力する端子である。 The antenna connection terminal 100 is an example of an antenna terminal and is connected to the antenna 2. The input terminal 110 is a terminal for receiving a band A transmission signal from the RFIC 3 . The output terminal 120 is a terminal for outputting the reception signal of band A to the RFIC 3 . The input/output terminal 130 is a terminal that receives a band B transmission signal from the RFIC 3 or outputs a band B reception signal to the RFIC 3 .
　フィルタ１１は、第１フィルタの一例であり、時分割複信（ＴＤＤ）用のバンドＡの送信帯域を含む通過帯域を有する送信用のフィルタである。フィルタ１１は、１以上の弾性波共振子を有する。 The filter 11 is an example of a first filter, and is a transmission filter having a passband including the transmission band of band A for time division duplex (TDD). Filter 11 has one or more elastic wave resonators.
　フィルタ１２は、第２フィルタの一例であり、ＴＤＤ用のバンドＡの受信帯域を含む通過帯域を有する受信用のフィルタである。フィルタ１２は、１以上の弾性波共振子を有する。 The filter 12 is an example of a second filter, and is a filter for reception having a passband including the reception band of band A for TDD. Filter 12 has one or more elastic wave resonators.
　フィルタ２０は、第３フィルタの一例であり、バンドＡと異なるバンドＢの少なくとも一部を含む通過帯域を有する。フィルタ２０は、１以上の弾性波共振子を有する。なお、バンドＢは、周波数分割複信（ＦＤＤ）用のバンドであってもよく、また、ＴＤＤ用のバンドであってもよい。 The filter 20 is an example of a third filter, and has a passband including at least part of band B different from band A. Filter 20 has one or more elastic wave resonators. Band B may be a frequency division duplex (FDD) band or a TDD band.
　スイッチ回路３０は、端子３０ａ（第１端子）、端子３０ｂ（第２端子）、端子３０ｃ（第３端子）および端子３０ｄ（第４端子）を有する。スイッチ回路３０は、（１）端子３０ａおよび３０ｂの接続と端子３０ａおよび端子３０ｄの接続とを同時に実行可能であり、また（２）端子３０ａおよび端子３０ｃの接続と端子３０ａおよび端子３０ｄの接続とを同時に実行可能である。 The switch circuit 30 has a terminal 30a (first terminal), a terminal 30b (second terminal), a terminal 30c (third terminal) and a terminal 30d (fourth terminal). The switch circuit 30 can (1) simultaneously connect the terminals 30a and 30b and connect the terminals 30a and 30d, and (2) connect the terminals 30a and 30c and connect the terminals 30a and 30d. can be executed simultaneously.
　スイッチ回路３０は、例えば、端子３０ａ、３０ｂおよび３０ｃを有し、端子３０ａと端子３０ｂとの接続、および、端子３０ａと端子３０ｃとの接続を切り替えるＳＰＤＴ(Single Pole Double Throw)型の第１スイッチと、端子３０ａおよび３０ｄを有し、端子３０ａと端子３０ｄとの接続および非接続を切り替えるＳＰＳＴ(Single Pole Single Throw)型の第２スイッチとで構成されていてもよい。なお、第１スイッチは、端子３０ａと端子３０ｂおよび３０ｃのいずれとも接続しない状態を実行してもよい。 The switch circuit 30 has, for example, terminals 30a, 30b, and 30c, and a first SPDT (Single Pole Double Throw) switch that switches between the connection between the terminals 30a and 30b and the connection between the terminals 30a and 30c. and a SPST (Single Pole Single Throw) type second switch that has terminals 30a and 30d and switches connection and disconnection between the terminals 30a and 30d. Note that the first switch may perform a state in which it is not connected to any of the terminals 30a and 30b and 30c.
　また、スイッチ回路３０は、端子３０ａ、３０ｄおよび３０ｅを有し、端子３０ａと端子３０ｄとの接続および非接続を切り替え、端子３０ａと端子３０ｅとの接続および非接続を切り替えるＳＰＤＴ型の第３スイッチと、端子３０ｂ、３０ｃおよび３０ｆを有し、端子３０ｆが端子３０ｅに接続されており、端子３０ｆと端子３０ｂとの接続および端子３０ｆと端子３０ｃとの接続との接続を切り替えるＳＰＤＴ型の第４スイッチとで構成されていてもよい。 The switch circuit 30 has terminals 30a, 30d, and 30e, and switches connection/disconnection between the terminals 30a and 30d, and switches connection/disconnection between the terminals 30a and 30e. and terminals 30b, 30c, and 30f, the terminal 30f is connected to the terminal 30e, and the fourth SPDT type switching the connection between the terminal 30f and the terminal 30b and the connection between the terminal 30f and the terminal 30c. It may be configured with a switch.
　なお、スイッチ回路３０は、ＲＦＩＣ３の制御部からの信号が入力される制御端子を備え、スイッチ回路３０の内部にはその制御信号に基づいて各スイッチの切り替えを制御するための制御回路を備えていてもよい。 The switch circuit 30 has a control terminal to which a signal from the control section of the RFIC 3 is input, and a control circuit for controlling switching of each switch based on the control signal is provided inside the switch circuit 30. may
　電力増幅器４１は、入力端子１１０とフィルタ１１との間に接続されており、バンドＡの送信信号を増幅する。低雑音増幅器４２は、フィルタ１２と出力端子１２０との間に接続されており、バンドＡの受信信号を増幅する。 The power amplifier 41 is connected between the input terminal 110 and the filter 11 and amplifies the band A transmission signal. Low noise amplifier 42 is connected between filter 12 and output terminal 120 and amplifies the received band A signal.
　インダクタ９０は、第２インダクタの一例であり、アンテナ接続端子１００と端子３０ａとを結ぶアンテナ接続経路とグランドとの間に接続されている。 The inductor 90 is an example of a second inductor, and is connected between the antenna connection path connecting the antenna connection terminal 100 and the terminal 30a and the ground.
　アンテナ接続端子１００は端子３０ａに接続されている。フィルタ１１は端子３０ｂと入力端子１１０との間に接続されている。フィルタ１２は端子３０ｃと出力端子１２０との間に接続されている。フィルタ２０は端子３０ｄと入出力端子１３０との間に接続されている。 The antenna connection terminal 100 is connected to the terminal 30a. Filter 11 is connected between terminal 30b and input terminal 110 . Filter 12 is connected between terminal 30 c and output terminal 120 . Filter 20 is connected between terminal 30 d and input/output terminal 130 .
　なお、本実施の形態に係る高周波回路１は、電力増幅器４１、低雑音増幅器４２、およびインダクタ９０を備えていなくてもよい。 Note that the high-frequency circuit 1 according to the present embodiment does not have to include the power amplifier 41, the low-noise amplifier 42, and the inductor 90.
　上記構成によれば、高周波回路１は、バンドＡの送信信号とバンドＢの信号とを同時伝送することが可能であり、バンドＡの受信信号とバンドＢの信号とを同時伝送することが可能である。 According to the above configuration, the high-frequency circuit 1 can simultaneously transmit a transmission signal of band A and a signal of band B, and can simultaneously transmit a reception signal of band A and a signal of band B. is.
　例えば、バンドＡの送信信号とバンドＢの信号とを同時伝送する場合、スイッチ回路３０において、端子３０ａと端子３０ｂとが接続され、かつ、端子３０ａと端子３０ｄとが接続される。また例えば、バンドＡの受信信号とバンドＢの信号とを同時伝送する場合、スイッチ回路３０において、端子３０ａと端子３０ｃとが接続され、かつ、端子３０ａと端子３０ｄとが接続される。 For example, when a transmission signal of band A and a signal of band B are simultaneously transmitted, in the switch circuit 30, terminals 30a and 30b are connected, and terminals 30a and 30d are connected. For example, when simultaneously transmitting a received signal of band A and a signal of band B, in the switch circuit 30, the terminals 30a and 30c are connected, and the terminals 30a and 30d are connected.
　なお、バンドＡおよびバンドＢは、無線アクセス技術（ＲＡＴ：Radio Access Technology）を用いて構築される通信システムのために、標準化団体など（例えば３ＧＰＰ（登録商標）、ＩＥＥＥ（Institute of Electrical and Electronics Engineers）等）によって予め定義された周波数バンドを意味する。本実施の形態では、通信システムとしては、例えば４Ｇ（4th Generation）－ＬＴＥ(Long Term Evolution)システム、５Ｇ（5th Generation）－ＮＲ（New Radio）システム、およびＷＬＡＮ（Wireless Local Area Network）システム等を用いることができるが、これらに限定されない。 In addition, band A and band B are used for communication systems constructed using radio access technology (RAT), such as standardization organizations (e.g., 3GPP (registered trademark), IEEE (Institute of Electrical and Electronics Engineers ) etc.). In this embodiment, as a communication system, for example, 4G (4th Generation)-LTE (Long Term Evolution) system, 5G (5th Generation)-NR (New Radio) system, WLAN (Wireless Local Area Network) system, etc. can be used, but is not limited to these.
　なお、バンドＡとしては、例えば、４Ｇ－ＬＴＥのバンド４１または５Ｇ－ＮＲのバンドｎ４１（２４９６－２６９０ＭＨｚ）が適用される。 As band A, for example, 4G-LTE band 41 or 5G-NR band n41 (2496-2690 MHz) is applied.
　また、バンドＢとしては、例えば、４Ｇ－ＬＴＥのバンド４０または５Ｇ－ＮＲのバンドｎ４０（２３００－２４００ＭＨｚ）、４Ｇ－ＬＴＥのバンド１または５Ｇ－ＮＲのバンドｎ１（アップリンク動作バンド：１９２０－１９８０ＭＨｚ、ダウンリンク動作バンド：２１１０－２１７０ＭＨｚ）、および４Ｇ－ＬＴＥのバンド３２または５Ｇ－ＮＲのバンドｎ３２（ダウンリンク動作バンド：１４５２－１４９６ＭＨｚ）のいずれかが適用される。 In addition, as band B, for example, 4G-LTE band 40 or 5G-NR band n40 (2300-2400 MHz), 4G-LTE band 1 or 5G-NR band n1 (uplink operating band: 1920-1980 MHz , downlink operating band: 2110-2170 MHz) and either band 32 of 4G-LTE or band n32 of 5G-NR (downlink operating band: 1452-1496 MHz).
　なお、アップリンク動作バンドとは、上記バンドのうちのアップリンク用に指定された周波数範囲を意味する。また、ダウンリンク動作バンドとは、上記バンドのうちのダウンリンク用に指定された周波数範囲を意味する。 It should be noted that the uplink operating band means the frequency range designated for the uplink among the above bands. Also, the downlink operating band means the frequency range designated for the downlink among the above bands.
　本実施の形態に係る高周波回路１において、バンドＢは、バンドＡよりも低周波側に位置している。 In the high-frequency circuit 1 according to this embodiment, the band B is located on the lower frequency side than the band A.
　［２　弾性波フィルタの構造］
　ここで、高周波回路１を構成するフィルタ１１および１２の回路構成、および、各フィルタを構成する弾性波共振子の構造について例示する。 [2 Structure of elastic wave filter]
Here, the circuit configuration of the filters 11 and 12 forming the high frequency circuit 1 and the structure of the elastic wave resonator forming each filter will be exemplified.
　図２Ａは、実施の形態に係るフィルタ１１の回路構成例を示す図である。また、図２Ｂは、実施の形態に係るフィルタ１２の回路構成例を示す図である。 FIG. 2A is a diagram showing a circuit configuration example of the filter 11 according to the embodiment. FIG. 2B is a diagram showing a circuit configuration example of the filter 12 according to the embodiment.
　本実施の形態に係るフィルタ１１は、例えば、図２Ａに示された弾性波フィルタの回路構成を有する。 The filter 11 according to the present embodiment has, for example, the circuit configuration of the elastic wave filter shown in FIG. 2A.
　フィルタ１１は、図２Ａに示すように、例えば、直列腕共振子１０１、１０２、１０３、１０４、１０５、１０６および１０７と、並列腕共振子１５１、１５２、１５３、１５４、１５５および１５６と、入出力端子１１１および１１２と、を備える。直列腕共振子１０１～１０７および並列腕共振子１５１～１５６のそれぞれは、弾性波共振子である。 The filter 11 includes, for example, series arm resonators 101, 102, 103, 104, 105, 106 and 107; and output terminals 111 and 112 . Each of the series arm resonators 101-107 and the parallel arm resonators 151-156 is an elastic wave resonator.
　入出力端子１１２は端子３０ｂに接続され、入出力端子１１１は電力増幅器４１の出力端子に接続されている。直列腕共振子１０１～１０７は、入出力端子１１１および１１２を結ぶ第１直列腕経路上に配置されている。また、並列腕共振子１５１～１５６のそれぞれは、直列腕共振子１０１～１０７の各接続点とグランドとの間に接続されている。上記接続構成により、フィルタ１１は、ラダー型のバンドパスフィルタを構成している。 The input/output terminal 112 is connected to the terminal 30 b and the input/output terminal 111 is connected to the output terminal of the power amplifier 41 . Series arm resonators 101 to 107 are arranged on a first series arm path connecting input/ output terminals 111 and 112 . Also, each of the parallel arm resonators 151 to 156 is connected between each connection point of the series arm resonators 101 to 107 and the ground. With the above connection configuration, the filter 11 constitutes a ladder-type bandpass filter.
　なお、フィルタ１１において、直列腕共振子および並列腕共振子の数は任意である。また、フィルタ１１は、複数のＩＤＴ（InterDigital Transducer）電極が圧電性を有する基板上の弾性波伝搬方向に配置された縦結合型フィルタであってもよい。 In addition, in the filter 11, the number of series arm resonators and parallel arm resonators is arbitrary. Further, the filter 11 may be a longitudinal coupling filter in which a plurality of IDT (InterDigital Transducer) electrodes are arranged on a substrate having piezoelectricity in the elastic wave propagation direction.
　フィルタ１２は、図２Ｂに示すように、例えば、直列腕共振子２０１、２０２、２０３、２０４および２０５と、並列腕共振子２５１、２５２、２５３、２５４および２５５と、インダクタ２５０と、入出力端子１２１および１２２と、を備える。直列腕共振子２０１～２０５および並列腕共振子２５１～２５５のそれぞれは、弾性波共振子である。 As shown in FIG. 2B, the filter 12 includes, for example, series arm resonators 201, 202, 203, 204 and 205; parallel arm resonators 251, 252, 253, 254 and 255; an inductor 250; 121 and 122. Each of the series arm resonators 201-205 and the parallel arm resonators 251-255 is an elastic wave resonator.
　入出力端子１２１は端子３０ｃに接続され、入出力端子１２２は低雑音増幅器４２の入力端子に接続されている。直列腕共振子２０１～２０５は、入出力端子１２１および１２２を結ぶ第２直列腕経路上に配置されている。また、並列腕共振子２５１～２５５のそれぞれは、直列腕共振子２０１～２０５および入出力端子１２２の各接続点とグランドとの間に接続されている。 The input/output terminal 121 is connected to the terminal 30c, and the input/output terminal 122 is connected to the input terminal of the low noise amplifier 42. Series arm resonators 201 to 205 are arranged on a second series arm path connecting input/ output terminals 121 and 122 . Also, each of the parallel arm resonators 251 to 255 is connected between each connection point of the series arm resonators 201 to 205 and the input/output terminal 122 and the ground.
　並列腕共振子２５１は、第１弾性波共振子の一例であり、入出力端子１２１および１２２を結ぶ第２直列腕経路とグランドとの間に接続された並列腕共振子２５１～２５５のうちで最もスイッチ回路３０に近く接続されている。 Parallel arm resonator 251 is an example of a first elastic wave resonator, and among parallel arm resonators 251 to 255 connected between a second series arm path connecting input/ output terminals 121 and 122 and the ground, It is connected closest to the switch circuit 30 .
　インダクタ２５０は、第１インダクタの一例であり、第２直列腕経路とグランドとの間に配置され、並列腕共振子２５１と直列接続されている。本実施の形態では、インダクタ２５０は、並列腕共振子２５１とグランドとの間に接続されているが、第２直列腕経路と並列腕共振子２５１との間に接続されていてもよい。 The inductor 250 is an example of a first inductor, is arranged between the second series arm path and the ground, and is connected in series with the parallel arm resonator 251 . In this embodiment, inductor 250 is connected between parallel arm resonator 251 and the ground, but inductor 250 may be connected between the second series arm path and parallel arm resonator 251 .
　上記接続構成により、フィルタ１２は、ラダー型のバンドパスフィルタを構成している。 With the above connection configuration, the filter 12 constitutes a ladder-type bandpass filter.
　なお、フィルタ１２において、直列腕共振子および並列腕共振子の数は任意である。また、フィルタ１２は、並列腕共振子２５１およびインダクタ２５０を少なくとも有していればよく、例えば、並列腕共振子２５１およびインダクタ２５０と、複数のＩＤＴ電極が圧電性を有する基板上の弾性波伝搬方向に配置された縦結合型フィルタ部と、を有していてもよい。 In addition, in the filter 12, the number of series arm resonators and parallel arm resonators is arbitrary. In addition, the filter 12 may have at least the parallel arm resonator 251 and the inductor 250. For example, the parallel arm resonator 251, the inductor 250, and the plurality of IDT electrodes are used for elastic wave propagation on a piezoelectric substrate. and a tandem-coupled filter section arranged in a direction.
　これによれば、フィルタ１２の並列腕共振子２５１に直列接続されたインダクタ２５０のインダクタンス値を調整することにより、フィルタ１２の自帯域（バンドＡ）のインピーダンスをシフトさせることなく、相手帯域（バンドＢ）のインピーダンスの位相幅を調整できる。 According to this, by adjusting the inductance value of the inductor 250 connected in series with the parallel arm resonator 251 of the filter 12, the impedance of the own band (band A) of the filter 12 can be adjusted without shifting the impedance of the other band (band A). The phase width of the impedance of B) can be adjusted.
　送信用のフィルタのインピーダンスおよび受信用のフィルタのインピーダンスの間に差異が生じると、送信用のフィルタと他のフィルタとを用いた同時伝送１、および、受信用のフィルタと他のフィルタとを用いた同時伝送２、の少なくとも一方において伝送損失の増加が懸念される。これを解消すべく、従来の高周波回路では位相調整回路が配置されているが、スイッチと各フィルタとを結ぶ直列腕経路に配置される位相調整用のリアクタンス素子が多いほど、高周波回路の伝送損失が増加する。 When a difference occurs between the impedance of the transmission filter and the impedance of the reception filter, simultaneous transmission 1 using the transmission filter and another filter, and using the reception filter and another filter are performed. There is concern about an increase in transmission loss in at least one of the two simultaneous transmissions. In order to solve this problem, a conventional high-frequency circuit has a phase adjustment circuit. increases.
　これに対して、本実施の形態に係る高周波回路１によれば、インダクタ２５０により、フィルタ１１および１２とスイッチ回路３０との間に配置される位相調整回路を削減して、バンドＡの送信信号と受信信号とを切り替える際に生じるインピーダンス整合の差異を低減しつつ、バンドＡの送信信号または受信信号とバンドＢの信号とを同時伝送することが可能となる。言い換えると、位相調整回路を削減できるので、ＴＤＤ用のバンドＡの高周波信号とバンドＢの高周波信号とを同時伝送する場合における伝送損失を低減することが可能となる。 On the other hand, according to the high-frequency circuit 1 according to the present embodiment, the phase adjustment circuit arranged between the filters 11 and 12 and the switch circuit 30 is eliminated by the inductor 250, and the transmission signal of the band A is reduced. It is possible to simultaneously transmit the transmission signal or reception signal of band A and the signal of band B while reducing the difference in impedance matching that occurs when switching between the transmission and reception signals. In other words, since the number of phase adjustment circuits can be reduced, it is possible to reduce the transmission loss when simultaneously transmitting the high-frequency signal of band A and the high-frequency signal of band B for TDD.
　なお、本実施の形態では、受信信号を通過させるフィルタ１２が第１弾性波共振子および第１インダクタを有しているが、フィルタ１１が第１弾性波共振子および第１インダクタを有していてもよい。さらに、フィルタ１１および１２の双方が、第１弾性波共振子および第１インダクタを有していてもよい。 In this embodiment, the filter 12 for passing the received signal has the first acoustic wave resonator and the first inductor, but the filter 11 has the first acoustic wave resonator and the first inductor. may Furthermore, both filters 11 and 12 may have a first acoustic wave resonator and a first inductor.
　また、フィルタ１１および１２の一方が第１インダクタを有し、フィルタ１１および１２の他方が第１インダクタを有さなくてもよい。第１インダクタは、フィルタ１１および１２の一方の相手帯域インピーダンスを、フィルタ１１および１２の他方の相手帯域インピーダンスと位相調整する機能を有する。よって、第１インダクタは、フィルタ１１および１２のいずれか一方のみに配置されていれば十分である。これによれば、簡素化された回路構成で、ＴＤＤ用のバンドＡの高周波信号とバンドＢの高周波信号とを同時伝送する場合における伝送損失を低減することが可能となる。 Also, one of the filters 11 and 12 may have the first inductor and the other of the filters 11 and 12 may not have the first inductor. The first inductor has the function of phase-adjusting the matching band impedance of one of the filters 11 and 12 with the matching band impedance of the other of the filters 11 and 12 . Therefore, it is sufficient if the first inductor is placed in only one of filters 11 and 12 . According to this, it is possible to reduce the transmission loss in the case of simultaneously transmitting the high frequency signal of band A and the high frequency signal of band B for TDD with a simplified circuit configuration.
　図３Ａは、実施の形態に係るフィルタ１１および１２を構成する弾性波共振子の第１例を模式的に表す平面図および断面図である。同図には、フィルタ１１および１２を構成する弾性波共振子の基本構造が例示されている。なお、図３Ａに示された弾性波共振子６０は、弾性波共振子の典型的な構造を説明するためのものであって、電極を構成する電極指の本数および長さなどは、これに限定されない。 FIG. 3A is a plan view and a cross-sectional view schematically showing a first example of elastic wave resonators forming filters 11 and 12 according to the embodiment. The figure illustrates the basic structure of the elastic wave resonators forming the filters 11 and 12 . Note that the elastic wave resonator 60 shown in FIG. 3A is for explaining a typical structure of an elastic wave resonator, and the number and length of the electrode fingers constituting the electrodes are Not limited.
　弾性波共振子６０は、圧電性を有する基板５０と、櫛形電極６０ａおよび６０ｂとで構成されている。 The elastic wave resonator 60 is composed of a piezoelectric substrate 50 and comb electrodes 60a and 60b.
　図３Ａの（ａ）に示すように、基板５０の上には、互いに対向する一対の櫛形電極６０ａおよび６０ｂが形成されている。櫛形電極６０ａは、互いに平行な複数の電極指６１ａと、複数の電極指６１ａを接続するバスバー電極６２ａとで構成されている。また、櫛形電極６０ｂは、互いに平行な複数の電極指６１ｂと、複数の電極指６１ｂを接続するバスバー電極６２ｂとで構成されている。複数の電極指６１ａおよび６１ｂは、弾性波伝搬方向（Ｘ軸方向）と直交する方向に沿って形成されている。 As shown in (a) of FIG. 3A, a pair of comb electrodes 60a and 60b facing each other are formed on the substrate 50. As shown in FIG. The comb-shaped electrode 60a is composed of a plurality of parallel electrode fingers 61a and busbar electrodes 62a connecting the plurality of electrode fingers 61a. The comb-shaped electrode 60b is composed of a plurality of parallel electrode fingers 61b and a busbar electrode 62b connecting the plurality of electrode fingers 61b. The plurality of electrode fingers 61a and 61b are formed along a direction orthogonal to the elastic wave propagation direction (X-axis direction).
　また、複数の電極指６１ａおよび６１ｂ、ならびに、バスバー電極６２ａおよび６２ｂで構成されるＩＤＴ電極５４は、図３Ａの（ｂ）に示すように、密着層５４０と主電極層５４２との積層構造となっている。 The IDT electrode 54, which is composed of a plurality of electrode fingers 61a and 61b and busbar electrodes 62a and 62b, has a laminated structure of an adhesion layer 540 and a main electrode layer 542, as shown in (b) of FIG. 3A. It's becoming
　密着層５４０は、基板５０と主電極層５４２との密着性を向上させるための層であり、材料として、例えば、Ｔｉが用いられる。主電極層５４２は、材料として、例えば、Ｃｕを１％含有したＡｌが用いられる。保護層５５は、櫛形電極６０ａおよび６０ｂを覆うように形成されている。保護層５５は、主電極層５４２を外部環境から保護する、周波数温度特性を調整する、および、耐湿性を高めるなどを目的とする層であり、例えば、二酸化ケイ素を主成分とする誘電体膜である。 The adhesion layer 540 is a layer for improving adhesion between the substrate 50 and the main electrode layer 542, and is made of Ti, for example. The material of the main electrode layer 542 is, for example, Al containing 1% Cu. Protective layer 55 is formed to cover comb electrodes 60a and 60b. The protective layer 55 is a layer for the purpose of protecting the main electrode layer 542 from the external environment, adjusting frequency temperature characteristics, and increasing moisture resistance. is.
　なお、密着層５４０、主電極層５４２および保護層５５を構成する材料は、上述した材料に限定されない。さらに、ＩＤＴ電極５４は、上記積層構造でなくてもよい。ＩＤＴ電極５４は、例えば、Ｔｉ、Ａｌ、Ｃｕ、Ｐｔ、Ａｕ、Ａｇ、Ｐｄなどの金属または合金から構成されてもよく、また、上記の金属または合金から構成される複数の積層体から構成されてもよい。また、保護層５５は、形成されていなくてもよい。 It should be noted that the materials forming the adhesion layer 540, the main electrode layer 542 and the protective layer 55 are not limited to the materials described above. Furthermore, the IDT electrode 54 may not have the laminated structure described above. The IDT electrode 54 may be composed of, for example, metals or alloys such as Ti, Al, Cu, Pt, Au, Ag, and Pd, and may be composed of a plurality of laminates composed of the above metals or alloys. may Also, the protective layer 55 may not be formed.
　次に、基板５０の積層構造について説明する。 Next, the laminated structure of the substrate 50 will be described.
　図３Ａの（ｃ）に示すように、基板５０は、高音速支持基板５１と、低音速膜５２と、圧電膜５３とを備え、高音速支持基板５１、低音速膜５２および圧電膜５３がこの順で積層された構造を有している。 As shown in (c) of FIG. 3A, the substrate 50 includes a high acoustic velocity supporting substrate 51, a low acoustic velocity film 52, and a piezoelectric film 53. The high acoustic velocity supporting substrate 51, the low acoustic velocity film 52, and the piezoelectric film 53 are It has a structure laminated in this order.
　圧電膜５３は、例えばθ°ＹカットＸ伝搬ＬｉＴａＯ３圧電単結晶または圧電セラミックス（Ｘ軸を中心軸としてＹ軸からθ°回転した軸を法線とする面で切断したリチウムタンタレート単結晶、またはセラミックスであって、Ｘ軸方向に弾性表面波が伝搬する単結晶またはセラミックス）からなる。なお、各フィルタの要求仕様により、圧電膜５３として使用される圧電単結晶の材料およびカット角θが適宜選択される。 The piezoelectric film 53 is, for example, a θ° Y-cut X-propagation LiTaO 3 piezoelectric single crystal or a piezoelectric ceramic (lithium tantalate single crystal cut along a plane normal to an axis rotated θ° from the Y axis with the X axis as the central axis, (or ceramics, single crystal or ceramics in which surface acoustic waves propagate in the X-axis direction). Note that the material of the piezoelectric single crystal used as the piezoelectric film 53 and the cut angle θ are appropriately selected according to the required specifications of each filter.
　高音速支持基板５１は、低音速膜５２、圧電膜５３ならびにＩＤＴ電極５４を支持する基板である。高音速支持基板５１は、さらに、圧電膜５３を伝搬する表面波および境界波などの弾性波よりも、高音速支持基板５１中のバルク波の音速が高速となる基板であり、弾性表面波を圧電膜５３および低音速膜５２が積層されている部分に閉じ込め、高音速支持基板５１より下方に漏れないように機能する。高音速支持基板５１は、例えば、シリコン基板である。 The high acoustic velocity support substrate 51 is a substrate that supports the low acoustic velocity film 52 , the piezoelectric film 53 and the IDT electrodes 54 . The high acoustic velocity support substrate 51 is a substrate in which the acoustic velocity of bulk waves in the high acoustic velocity support substrate 51 is faster than acoustic waves such as surface waves and boundary waves propagating through the piezoelectric film 53, and surface acoustic waves are generated. It functions so that it is confined in the portion where the piezoelectric film 53 and the low sound velocity film 52 are laminated and does not leak below the high sound velocity support substrate 51 . The high acoustic velocity support substrate 51 is, for example, a silicon substrate.
　低音速膜５２は、圧電膜５３を伝搬するバルク波よりも、低音速膜５２中のバルク波の音速が低速となる膜であり、圧電膜５３と高音速支持基板５１との間に配置される。この構造と、弾性波が本質的に低音速な媒質にエネルギーが集中するという性質とにより、弾性表面波エネルギーのＩＤＴ電極外への漏れが抑制される。低音速膜５２は、例えば、二酸化ケイ素を主成分とする膜である。 The low sound velocity film 52 is a film in which the sound velocity of the bulk wave in the low sound velocity film 52 is lower than that of the bulk wave propagating through the piezoelectric film 53 , and is arranged between the piezoelectric film 53 and the high sound velocity support substrate 51 . be. This structure and the nature of the elastic wave to concentrate its energy in a low-temperature medium suppresses leakage of the surface acoustic wave energy to the outside of the IDT electrode. The low-temperature velocity film 52 is, for example, a film whose main component is silicon dioxide.
　なお、基板５０の上記積層構造によれば、圧電基板を単層で使用している従来の構造と比較して、共振周波数および反共振周波数におけるＱ値を大幅に高めることが可能となる。すなわち、Ｑ値が高い弾性波共振子を構成し得るので、当該弾性波共振子を用いて、挿入損失が小さいフィルタを構成することが可能となる。 In addition, according to the laminated structure of the substrate 50, it is possible to significantly increase the Q value at the resonance frequency and anti-resonance frequency compared to the conventional structure using a single layer piezoelectric substrate. That is, since an acoustic wave resonator with a high Q value can be configured, it is possible to configure a filter with a small insertion loss using the acoustic wave resonator.
　なお、高音速支持基板５１は、支持基板と、圧電膜５３を伝搬する表面波および境界波などの弾性波よりも、伝搬するバルク波の音速が高速となる高音速膜とが積層された構造を有していてもよい。この場合、支持基板には、サファイア、リチウムタンタレート、リチウムニオベイト、および水晶等の圧電体、アルミナ、マグネシア、窒化ケイ素、窒化アルミニウム、炭化ケイ素、ジルコニア、コージライト、ムライト、ステアタイト、およびフォルステライト等の各種セラミック、ガラス等の誘電体、シリコンおよび窒化ガリウム等の半導体、ならびに樹脂基板等を用いることができる。また、高音速膜には、窒化アルミニウム、酸化アルミニウム、炭化ケイ素、窒化ケイ素、酸窒化ケイ素、ＤＬＣ膜、ダイヤモンド、これらの材料を主成分とする媒質、これらの材料の混合物を主成分とする媒質等、様々な高音速材料を用いることができる。 The high acoustic velocity support substrate 51 has a structure in which a support substrate and a high acoustic velocity film having a higher acoustic velocity than elastic waves such as surface waves and boundary waves propagating through the piezoelectric film 53 are laminated. may have In this case, the support substrate includes piezoelectric materials such as sapphire, lithium tantalate, lithium niobate, and quartz, alumina, magnesia, silicon nitride, aluminum nitride, silicon carbide, zirconia, cordierite, mullite, steatite, and fort. Various ceramics such as stellite, dielectrics such as glass, semiconductors such as silicon and gallium nitride, and resin substrates can be used. The high acoustic velocity film includes aluminum nitride, aluminum oxide, silicon carbide, silicon nitride, silicon oxynitride, DLC film, diamond, media containing these materials as main components, and media containing mixtures of these materials as main components. etc., various high acoustic velocity materials can be used.
　また、図３Ｂは、実施の形態に係るフィルタ１１および１２を構成する弾性波共振子の第２例を模式的に表す断面図である。図３Ａに示した弾性波共振子６０では、ＩＤＴ電極５４が、圧電膜５３を有する基板５０上に形成された例を示したが、当該ＩＤＴ電極５４が形成される基板は、図３Ｂに示すように、圧電体層の単層からなる圧電単結晶基板５７であってもよい。圧電単結晶基板５７は、例えば、ＬｉＮｂＯ３の圧電単結晶で構成されている。本例に係る弾性波共振子は、ＬｉＮｂＯ３の圧電単結晶基板５７と、ＩＤＴ電極５４と、圧電単結晶基板５７上およびＩＤＴ電極５４上に形成された保護層５８と、で構成されている。 Further, FIG. 3B is a cross-sectional view schematically showing a second example of elastic wave resonators forming filters 11 and 12 according to the embodiment. The elastic wave resonator 60 shown in FIG. 3A shows an example in which the IDT electrodes 54 are formed on the substrate 50 having the piezoelectric film 53. The substrate on which the IDT electrodes 54 are formed is shown in FIG. 3B. Thus, the piezoelectric single crystal substrate 57 may be a single piezoelectric layer. The piezoelectric single crystal substrate 57 is composed of, for example, a piezoelectric single crystal of LiNbO 3 . The acoustic wave resonator according to this example is composed of a piezoelectric single crystal substrate 57 of LiNbO 3 , an IDT electrode 54 , and a protective layer 58 formed on the piezoelectric single crystal substrate 57 and the IDT electrode 54 . .
　上述した圧電膜５３および圧電単結晶基板５７は、弾性波フィルタ装置の要求通過特性などに応じて、適宜、積層構造、材料、カット角、および、厚みを変更してもよい。上述したカット角以外のカット角を有するＬｉＴａＯ３圧電基板などを用いた弾性波共振子であっても、上述した圧電膜５３を用いた弾性波共振子６０と同様の効果を奏することができる。 The piezoelectric film 53 and the piezoelectric single crystal substrate 57 described above may be appropriately changed in laminated structure, material, cut angle, and thickness according to the required transmission characteristics of the elastic wave filter device. Even an elastic wave resonator using a LiTaO 3 piezoelectric substrate having a cut angle other than the cut angle described above can produce the same effects as the elastic wave resonator 60 using the piezoelectric film 53 described above.
　また、ＩＤＴ電極５４が形成される基板は、支持基板と、エネルギー閉じ込め層と、圧電膜がこの順で積層された構造を有していてもよい。圧電膜上にＩＤＴ電極５４が形成される。圧電膜は、例えば、ＬｉＴａＯ３圧電単結晶または圧電セラミックスが用いられる。支持基板は、圧電膜、エネルギー閉じ込め層、およびＩＤＴ電極５４を支持する基板である。 Also, the substrate on which the IDT electrodes 54 are formed may have a structure in which a supporting substrate, an energy trapping layer, and a piezoelectric film are laminated in this order. An IDT electrode 54 is formed on the piezoelectric film. The piezoelectric film is, for example, LiTaO 3 piezoelectric single crystal or piezoelectric ceramics. The support substrate is the substrate that supports the piezoelectric film, the energy confinement layer, and the IDT electrodes 54 .
　エネルギー閉じ込め層は１層または複数の層からなり、その少なくとも１つの層を伝搬するバルク弾性波の速度は、圧電膜近傍を伝搬する弾性波の速度よりも大きい。例えば、エネルギー閉じ込め層は、低音速層と、高音速層との積層構造となっていてもよい。低音速層は、圧電膜を伝搬する弾性波の音速よりも、低音速層中のバルク波の音速が低速となる膜である。高音速層は、圧電膜を伝搬する弾性波の音速よりも、高音速層中のバルク波の音速が高速となる膜である。なお、支持基板を高音速層としてもよい。 The energy confinement layer consists of one or more layers, and the velocity of the bulk acoustic wave propagating through at least one layer is greater than the velocity of the elastic wave propagating near the piezoelectric film. For example, the energy trapping layer may have a laminated structure of a low acoustic velocity layer and a high acoustic velocity layer. The sound velocity layer is a film in which the sound velocity of bulk waves in the sound velocity layer is lower than the sound velocity of elastic waves propagating through the piezoelectric film. The high acoustic velocity layer is a film in which the acoustic velocity of bulk waves in the high acoustic velocity layer is higher than the acoustic velocity of elastic waves propagating through the piezoelectric film. Note that the support substrate may be a high acoustic velocity layer.
　また、エネルギー閉じ込め層は、音響インピーダンスが相対的に低い低音響インピーダンス層と、音響インピーダンスが相対的に高い高音響インピーダンス層とが、交互に積層された構成を有する音響インピーダンス層であってもよい。 Also, the energy trapping layer may be an acoustic impedance layer having a configuration in which a low acoustic impedance layer with a relatively low acoustic impedance and a high acoustic impedance layer with a relatively high acoustic impedance are alternately laminated. .
　ここで、弾性波共振子６０を構成するＩＤＴ電極の電極パラメータの一例（実施例）について説明する。 Here, an example (working example) of the electrode parameters of the IDT electrodes forming the acoustic wave resonator 60 will be described.
　弾性波共振子の波長とは、図３Ａの（ｂ）に示すＩＤＴ電極５４を構成する複数の電極指６１ａまたは６１ｂの繰り返し周期である波長λで規定される。また、電極指ピッチは、波長λの１／２であり、櫛形電極６０ａおよび６０ｂをそれぞれ構成する電極指６１ａおよび６１ｂのライン幅をＷとし、隣り合う電極指６１ａと電極指６１ｂとの間のスペース幅をＳとした場合、（Ｗ＋Ｓ）で定義される。また、一対の櫛形電極６０ａおよび６０ｂの交叉幅Ｌは、図３Ａの（ａ）に示すように、電極指６１ａと電極指６１ｂとの弾性波伝搬方向（Ｘ軸方向）から見た場合の重複する電極指の長さである。また、各弾性波共振子の電極デューティーは、複数の電極指６１ａおよび６１ｂのライン幅占有率であり、複数の電極指６１ａおよび６１ｂのライン幅とスペース幅との加算値に対する当該ライン幅の割合であり、Ｗ／（Ｗ＋Ｓ）で定義される。また、櫛形電極６０ａおよび６０ｂの高さをｈとしている。以降では、波長λ、電極指ピッチ、交叉幅Ｌ、電極デューティー、ＩＤＴ電極５４の高さｈ等、弾性波共振子のＩＤＴ電極の形状に関するパラメータは、電極パラメータと定義される。 The wavelength of the elastic wave resonator is defined by the wavelength λ which is the repetition period of the plurality of electrode fingers 61a or 61b forming the IDT electrode 54 shown in (b) of FIG. 3A. The electrode finger pitch is 1/2 of the wavelength λ, the line width of the electrode fingers 61a and 61b constituting the comb-shaped electrodes 60a and 60b is W, and the distance between the adjacent electrode fingers 61a and 61b is When the space width is S, it is defined as (W+S). Moreover, as shown in (a) of FIG. 3A, the intersecting width L of the pair of comb-shaped electrodes 60a and 60b is the overlap of the electrode fingers 61a and 61b when viewed from the elastic wave propagation direction (X-axis direction). is the length of the electrode finger that The electrode duty of each acoustic wave resonator is the line width occupation ratio of the plurality of electrode fingers 61a and 61b, and is the ratio of the line width to the sum of the line width and space width of the plurality of electrode fingers 61a and 61b. and is defined as W/(W+S). Also, the height of the comb electrodes 60a and 60b is h. Hereinafter, parameters related to the shape of the IDT electrodes of the acoustic wave resonator, such as the wavelength λ, the electrode finger pitch, the crossing width L, the electrode duty, and the height h of the IDT electrodes 54, are defined as electrode parameters.
　なお、ＩＤＴ電極５４において、隣り合う電極指間の間隔が一定でない場合には、ＩＤＴ電極５４の電極指ピッチは、ＩＤＴ電極５４の平均電極指ピッチで定義される。ＩＤＴ電極５４の平均電極指ピッチは、ＩＤＴ電極５４に含まれる電極指６１ａ、６１ｂの総本数をＮｉ本とし、ＩＤＴ電極５４の、弾性波伝搬方向における一方端に位置する電極指と他方端に位置する電極指との中心間距離をＤｉとすると、Ｄｉ／（Ｎｉ－１）と定義される。 In the IDT electrodes 54 , if the intervals between adjacent electrode fingers are not constant, the electrode finger pitch of the IDT electrodes 54 is defined by the average electrode finger pitch of the IDT electrodes 54 . The average electrode finger pitch of the IDT electrode 54 is defined by the total number of the electrode fingers 61a and 61b included in the IDT electrode 54 being Ni, and the electrode finger positioned at one end of the IDT electrode 54 in the elastic wave propagation direction and It is defined as Di/(Ni-1), where Di is the center-to-center distance from the positioned electrode finger.
　また、図３Ｃは、実施の形態に係るフィルタ１１および１２を構成する弾性波共振子の第３例を模式的に表す断面図である。図３Ｃには、フィルタ１１および１２の弾性波共振子として、バルク弾性波共振子が示されている。同図に示すように、バルク弾性波共振子は、例えば、支持基板６５と、下部電極６６と、圧電体層６７と、上部電極６８と、を有しており、支持基板６５、下部電極６６、圧電体層６７、および上部電極６８がこの順で積層された構成となっている。 FIG. 3C is a cross-sectional view schematically showing a third example of elastic wave resonators forming filters 11 and 12 according to the embodiment. Bulk acoustic wave resonators are shown as acoustic wave resonators of filters 11 and 12 in FIG. 3C. As shown in the figure, the bulk acoustic wave resonator has, for example, a support substrate 65, a lower electrode 66, a piezoelectric layer 67, and an upper electrode 68. , a piezoelectric layer 67, and an upper electrode 68 are laminated in this order.
　支持基板６５は、下部電極６６、圧電体層６７、および上部電極６８を支持するための基板であり、例えば、シリコン基板である。なお、支持基板６５は、下部電極６６と接触する領域に、空洞が設けられている。これにより、圧電体層６７を自由に振動させることが可能となる。 The support substrate 65 is a substrate for supporting the lower electrode 66, the piezoelectric layer 67, and the upper electrode 68, and is, for example, a silicon substrate. The support substrate 65 is provided with a cavity in a region in contact with the lower electrode 66 . This allows the piezoelectric layer 67 to vibrate freely.
　下部電極６６は、第１電極の一例であり、支持基板６５の一方面上に形成されている。上部電極６８は、第２電極の一例であり、支持基板６５の一方面上に形成されている。下部電極６６および上部電極６８は、材料として、例えば、Ｃｕを１％含有したＡｌが用いられる。 The lower electrode 66 is an example of a first electrode and is formed on one surface of the support substrate 65 . The upper electrode 68 is an example of a second electrode and is formed on one surface of the support substrate 65 . The lower electrode 66 and the upper electrode 68 are made of Al containing 1% Cu, for example.
　圧電体層６７は、下部電極６６と上部電極６８との間に形成されている。圧電体層６７は、例えば、ＺｎＯ（酸化亜鉛）、ＡｌＮ（窒化アルミニウム）、ＰＺＴ（チタン酸ジルコン酸鉛）、ＫＮ（ニオブ酸カリウム）、ＬＮ（リチウムニオベイト）、ＬＴ（リチウムタンタレート）、水晶、およびＬｉＢＯ（ホウ酸リチウム）の少なくとも１つを主成分とする。 The piezoelectric layer 67 is formed between the lower electrode 66 and the upper electrode 68 . The piezoelectric layer 67 is made of, for example, ZnO (zinc oxide), AlN (aluminum nitride), PZT (lead zirconate titanate), KN (potassium niobate), LN (lithium niobate), LT (lithium tantalate), The main component is at least one of quartz and LiBO (lithium borate).
　上記積層構成を有するバルク弾性波共振子は、下部電極６６と上部電極６８との間に電気的なエネルギーを印加することで圧電体層６７内にてバルク弾性波を誘発して共振を発生させるものである。このバルク弾性波共振子により生成されるバルク弾性波は、下部電極６６と上部電極６８との間を、圧電体層６７の膜面に垂直な方向に伝搬する。つまり、バルク弾性波共振子は、バルク弾性波を利用した共振子である。 The bulk acoustic wave resonator having the above laminated structure induces a bulk acoustic wave in the piezoelectric layer 67 by applying electrical energy between the lower electrode 66 and the upper electrode 68 to generate resonance. It is. A bulk acoustic wave generated by this bulk acoustic wave resonator propagates between the lower electrode 66 and the upper electrode 68 in a direction perpendicular to the film surface of the piezoelectric layer 67 . That is, the bulk acoustic wave resonator is a resonator that utilizes bulk acoustic waves.
　［３　比較例に係る高周波回路５００の回路構成およびインピーダンス特性］
　次に、従来の高周波回路５００について回路構成を説明しておく。図４は、比較例に係る高周波回路５００の回路構成図、およびスイッチ選択端子から各フィルタを見たインピーダンスを示すスミスチャートである。同図の（ａ）に示すように、高周波回路５００は、フィルタ１１、５０２および２０と、スイッチ回路３０と、インダクタ９０と、電力増幅器４１と、低雑音増幅器４２と、アンテナ接続端子１００と、入力端子１１０と、出力端子１２０と、入出力端子１３０と、を備える。比較例に係る高周波回路５００は、実施の形態に係る高周波回路１と比較して、フィルタ１２に代わってフィルタ５０２が配置されている点のみが異なる。以下では、比較例に係る高周波回路５００について、実施の形態に係る高周波回路１と異なる点を中心に説明する。 [3 Circuit Configuration and Impedance Characteristics of High Frequency Circuit 500 According to Comparative Example]
Next, the circuit configuration of the conventional high frequency circuit 500 will be described. FIG. 4 is a circuit configuration diagram of a high-frequency circuit 500 according to a comparative example, and a Smith chart showing the impedance of each filter viewed from a switch selection terminal. As shown in (a) of the figure, the high frequency circuit 500 includes filters 11, 502 and 20, a switch circuit 30, an inductor 90, a power amplifier 41, a low noise amplifier 42, an antenna connection terminal 100, An input terminal 110 , an output terminal 120 and an input/output terminal 130 are provided. A high frequency circuit 500 according to the comparative example differs from the high frequency circuit 1 according to the embodiment only in that a filter 502 is arranged instead of the filter 12 . The high-frequency circuit 500 according to the comparative example will be described below, focusing on the points different from the high-frequency circuit 1 according to the embodiment.
　フィルタ１１は、ＴＤＤ用のバンドＡを含む通過帯域を有する。フィルタ１１は、１以上の弾性波共振子を有する。 The filter 11 has a passband including band A for TDD. Filter 11 has one or more elastic wave resonators.
　フィルタ５０２は、ＴＤＤ用のバンドＡを含む通過帯域を有する。フィルタ５０２は、１以上の弾性波共振子を有する。フィルタ５０２は、図２Ｂに示されたフィルタ１２の回路構成に対してインダクタ２５０が無い構成を有する。フィルタ５０２は、例えば、直列腕共振子２０１、２０２、２０３、２０４および２０５と、並列腕共振子２５１、２５２、２５３、２５４および２５５と、入出力端子１２１および１２２と、を備える。直列腕共振子２０１～２０５および並列腕共振子２５１～２５５のそれぞれは、弾性波共振子である。つまり、フィルタ５０２では、並列腕共振子２５１～２５５のうちで最もスイッチ回路３０に近く接続された並列腕共振子２５１に、インダクタが直列接続されていない。 The filter 502 has a passband including band A for TDD. Filter 502 has one or more elastic wave resonators. Filter 502 has a configuration without inductor 250 for the circuit configuration of filter 12 shown in FIG. 2B. Filter 502 includes series arm resonators 201, 202, 203, 204 and 205, parallel arm resonators 251, 252, 253, 254 and 255, and input/ output terminals 121 and 122, for example. Each of the series arm resonators 201-205 and the parallel arm resonators 251-255 is an elastic wave resonator. That is, in filter 502, no inductor is connected in series to parallel arm resonator 251, which is connected closest to switch circuit 30 among parallel arm resonators 251-255.
　フィルタ２０は、バンドＡと異なるバンドＢの少なくとも一部を含む通過帯域を有する。フィルタ２０は、１以上の弾性波共振子を有する。 The filter 20 has a passband that includes at least part of band B that is different from band A. Filter 20 has one or more elastic wave resonators.
　電力増幅器４１は、入力端子１１０とフィルタ１１との間に接続されており、バンドＡの送信信号を増幅する。低雑音増幅器４２は、フィルタ５０２と出力端子１２０との間に接続されており、バンドＡの受信信号を増幅する。 The power amplifier 41 is connected between the input terminal 110 and the filter 11 and amplifies the band A transmission signal. Low noise amplifier 42 is connected between filter 502 and output terminal 120 and amplifies the received band A signal.
　フィルタ１１は端子３０ｂと入力端子１１０との間に接続されている。フィルタ５０２は端子３０ｃと出力端子１２０との間に接続されている。フィルタ２０は端子３０ｄと入出力端子１３０との間に接続されている。 The filter 11 is connected between the terminal 30b and the input terminal 110. Filter 502 is connected between terminal 30 c and output terminal 120 . Filter 20 is connected between terminal 30 d and input/output terminal 130 .
　上記構成によれば、高周波回路５００は、バンドＡの送信信号とバンドＢの信号とを同時伝送することが可能であり、バンドＡの受信信号とバンドＢの信号とを同時伝送することが可能である。 According to the above configuration, the high-frequency circuit 500 can simultaneously transmit a transmission signal of band A and a signal of band B, and can simultaneously transmit a reception signal of band A and a signal of band B. is.
　図４の（ｂ）には、端子３０ｂ（Ｓ１）からフィルタ１１を見たインピーダンスを表すスミスチャートが示されている。また、図４の（ｃ）には、端子３０ｃ（Ｓ２）からフィルタ５０２を見たインピーダンスを表すスミスチャートが示されている。 (b) of FIG. 4 shows a Smith chart representing the impedance of the filter 11 viewed from the terminal 30b (S1). FIG. 4(c) shows a Smith chart representing the impedance of the filter 502 viewed from the terminal 30c (S2).
　なお、以下では、フィルタ自体の通過帯域（自帯域）でなく他のフィルタの通過帯域を相手帯域と記す場合がある。例えば、フィルタ１１の自帯域はバンドＡであり、フィルタ１１の相手帯域はバンドＢである。 In addition, hereinafter, the passband of another filter may be referred to as the other band instead of the passband (own band) of the filter itself. For example, the own band of filter 11 is band A, and the other band of filter 11 is band B.
　比較例に係る高周波回路５００では、ＴＤＤのバンドＡの送信信号および受信信号を伝送するにあたり、送信用のフィルタ１１と受信用のフィルタ５０２とを配置している。これは、電力増幅器４１の出力インピーダンスの最適値と低雑音増幅器４２の入力インピーダンスの最適値とが異なるため、電力増幅器４１および低雑音増幅器４２に接続されるフィルタのインピーダンス特性を異ならせる必要があることに起因する。 In the high-frequency circuit 500 according to the comparative example, a transmission filter 11 and a reception filter 502 are arranged for transmitting the TDD band A transmission signal and the reception signal. This is because the optimum value of the output impedance of the power amplifier 41 and the optimum value of the input impedance of the low noise amplifier 42 are different, so it is necessary to differentiate the impedance characteristics of the filters connected to the power amplifier 41 and the low noise amplifier 42. due to that.
　上記観点から、Ｓ１からフィルタ１１を見たインピーダンス（図４の（ｂ））と、Ｓ２からフィルタ５０２を見たインピーダンス（図４の（ｃ））とでは、相手帯域（バンドＢ）のインピーダンスの位相が４度以上異なっている。 From the above point of view, the impedance when looking at the filter 11 from S1 ((b) in FIG. 4) and the impedance when looking at the filter 502 from S2 ((c) in FIG. 4) are different from the impedance of the other band (band B). The phases differ by 4 degrees or more.
　このため、比較例に係る高周波回路５００において、バンドＡの送信信号とバンドＢの信号とを同時伝送する場合に端子３０ａからフィルタ１１および２０を見たバンドＢのインピーダンスと、バンドＡの受信信号とバンドＢの信号とを同時伝送する場合に端子３０ａからフィルタ５０２および２０を見たバンドＢのインピーダンスとは大きく異なる。これにより、例えば、バンドＡの送信信号とバンドＢの信号とを同時伝送する場合におけるバンドＢのインピーダンスを最適整合すれば、例えば、バンドＡの受信信号とバンドＢの信号とを同時伝送する場合におけるバンドＡの受信信号における符号誤り率が上昇し、受信感度が劣化してしまう。 Therefore, in the high-frequency circuit 500 according to the comparative example, when simultaneously transmitting the transmission signal of the band A and the signal of the band B, the impedance of the band B when the filters 11 and 20 are viewed from the terminal 30a and the reception signal of the band A and the band B signal are transmitted at the same time, the impedance of the band B when the filters 502 and 20 are viewed from the terminal 30a is greatly different. As a result, for example, when a transmission signal of band A and a signal of band B are simultaneously transmitted, if the impedance of band B is optimally matched, for example, when a reception signal of band A and a signal of band B are simultaneously transmitted, , the bit error rate in the received signal of band A increases, and the reception sensitivity deteriorates.
　［４　実施の形態に係る高周波回路１のインピーダンス特性］
　次に、実施の形態に係る高周波回路１のインピーダンス特性について説明する。図５は、実施の形態に係る高周波回路１の回路構成図、およびスイッチ選択端子から各フィルタを見たインピーダンスを示すスミスチャートである。同図の（ａ）には、高周波回路１の回路構成が示されている。 [4 Impedance Characteristics of High-Frequency Circuit 1 According to Embodiment]
Next, impedance characteristics of the high frequency circuit 1 according to the embodiment will be described. FIG. 5 is a circuit configuration diagram of the high-frequency circuit 1 according to the embodiment, and a Smith chart showing the impedance of each filter viewed from the switch selection terminal. The circuit configuration of the high-frequency circuit 1 is shown in (a) of FIG.
　図５の（ｂ）には、端子３０ｂ（Ｓ１）からフィルタ１１を見たインピーダンスを表すスミスチャートが示されている。また、図５の（ｃ）には、端子３０ｃ（Ｓ２）からフィルタ１２を見たインピーダンスを表すスミスチャートが示されている。また、図５の（ｄ）には、端子３０ｄ（Ｓ３）からフィルタ２０を見たインピーダンスを表すスミスチャートが示されている。 (b) of FIG. 5 shows a Smith chart representing the impedance of the filter 11 viewed from the terminal 30b (S1). Further, FIG. 5(c) shows a Smith chart representing the impedance of the filter 12 viewed from the terminal 30c (S2). Further, (d) of FIG. 5 shows a Smith chart representing the impedance of the filter 20 viewed from the terminal 30d (S3).
　実施の形態に係る高周波回路１では、ＴＤＤのバンドＡの送信信号および受信信号を伝送するにあたり、送信用のフィルタ１１と受信用のフィルタ１２とを配置している。これは、電力増幅器４１の出力インピーダンスの最適値と低雑音増幅器４２の入力インピーダンスの最適値とが異なるため、電力増幅器４１および低雑音増幅器４２に接続されるフィルタのインピーダンス特性を異ならせる必要があることに起因する。 In the high-frequency circuit 1 according to the embodiment, a transmission filter 11 and a reception filter 12 are arranged for transmission of TDD band A transmission signals and reception signals. This is because the optimum value of the output impedance of the power amplifier 41 and the optimum value of the input impedance of the low noise amplifier 42 are different, so it is necessary to differentiate the impedance characteristics of the filters connected to the power amplifier 41 and the low noise amplifier 42. due to that.
　これに対して、本実施の形態に係る高周波回路１では、比較例に係る高周波回路５００と異なり、Ｓ１からフィルタ１１を見たインピーダンス（図５の（ｂ））と、Ｓ２からフィルタ１２を見たインピーダンス（図５の（ｃ））とでは、相手帯域（バンドＢ）のインピーダンスの位相が揃っている。 On the other hand, in the high-frequency circuit 1 according to the present embodiment, unlike the high-frequency circuit 500 according to the comparative example, the impedance of the filter 11 viewed from S1 ((b) in FIG. 5) The phase of the impedance of the other band (band B) is aligned with the impedance ((c) of FIG. 5).
　これは、フィルタ１２が有するインダクタ２５０に起因するものである。１以上の弾性波共振子で構成されたフィルタにおいて、並列腕共振子に直列接続されたインダクタは、フィルタの自帯域のインピーダンスを変えることなく、減衰帯域のインピーダンスの位相幅を変化させることが可能である。 This is due to the inductor 250 that the filter 12 has. In a filter composed of one or more elastic wave resonators, an inductor connected in series with a parallel arm resonator can change the phase width of the impedance in the attenuation band without changing the impedance in the filter's own band. is.
　本実施の形態に係るフィルタ１２では、スイッチ回路３０（端子３０ｃ）に最も近く接続された並列腕共振子２５１にインダクタ２５０が直列接続されているため、フィルタ１２の出力インピーダンス（低雑音増幅器４２の入力端子からフィルタ１２を見たインピーダンス）を変化させずに、Ｓ２からフィルタ１２を見た相手帯域（バンドＢ）のインピーダンスの位相幅を変化させることが可能となる。 In filter 12 according to the present embodiment, inductor 250 is connected in series with parallel arm resonator 251 connected closest to switch circuit 30 (terminal 30c). It is possible to change the phase width of the impedance of the other band (band B) when the filter 12 is viewed from S2 without changing the impedance when the filter 12 is viewed from the input terminal.
　つまり、フィルタ１１が電力増幅器４１とインピーダンス整合し、フィルタ１２が低雑音増幅器４２とインピーダンス整合しているにも拘わらず、Ｓ１からフィルタ１１を見たバンドＢのインピーダンスの位相と、Ｓ２からフィルタ１２を見たバンドＢのインピーダンスの位相とを揃わせることが可能となる。 In other words, although the filter 11 is impedance-matched with the power amplifier 41 and the filter 12 is impedance-matched with the low-noise amplifier 42, the impedance phase of the band B when the filter 11 is viewed from S1 and the filter 12 from S2 It is possible to align the phase of the impedance of the band B seen from .
　図６Ａは、実施の形態に係る高周波回路１の回路構成図、および、端子３０ａおよびアンテナ接続端子１００から共通接続されたフィルタ１１および２０を見たインピーダンスを示すスミスチャートである。具体的には、図６Ａの（ｂ）には、フィルタ１１および２０を端子３０ａに接続した場合に端子３０ａ（Ｓ４）からフィルタ１１および２０を見たバンドＡおよびＢのインピーダンスが示されている。また、図６Ａの（ｃ）には、フィルタ１１および２０を端子３０ａに接続した場合にアンテナ接続端子１００からフィルタ１１および２０を見たバンドＡおよびＢのインピーダンスが示されている。 FIG. 6A is a circuit configuration diagram of the high-frequency circuit 1 according to the embodiment, and a Smith chart showing impedance when the filters 11 and 20 commonly connected from the terminal 30a and the antenna connection terminal 100 are viewed. Specifically, (b) of FIG. 6A shows the impedances of bands A and B when the filters 11 and 20 are viewed from the terminal 30a (S4) when the filters 11 and 20 are connected to the terminal 30a. . FIG. 6A (c) shows the impedances of the bands A and B when the filters 11 and 20 are viewed from the antenna connection terminal 100 when the filters 11 and 20 are connected to the terminal 30a.
　図６Ｂは、実施の形態に係る高周波回路１の回路構成図、および端子３０ａおよびアンテナ接続端子１００から共通接続されたフィルタ１２および２０を見たインピーダンスを示すスミスチャートである。具体的には、図６Ｂの（ｂ）には、フィルタ１２および２０を端子３０ａに接続した場合に端子３０ａ（Ｓ４）からフィルタ１２および２０を見たバンドＡおよびＢのインピーダンスが示されている。また、図６Ｂの（ｃ）には、フィルタ１２および２０を端子３０ａに接続した場合にアンテナ接続端子１００からフィルタ１２および２０を見たバンドＡおよびＢのインピーダンスが示されている。 FIG. 6B is a circuit configuration diagram of the high-frequency circuit 1 according to the embodiment, and a Smith chart showing impedance when the filters 12 and 20 commonly connected from the terminal 30a and the antenna connection terminal 100 are viewed. Specifically, (b) of FIG. 6B shows impedances of bands A and B when the filters 12 and 20 are viewed from the terminal 30a (S4) when the filters 12 and 20 are connected to the terminal 30a. . FIG. 6B (c) shows the impedances of the bands A and B when the filters 12 and 20 are viewed from the antenna connection terminal 100 when the filters 12 and 20 are connected to the terminal 30a.
　図６Ａの（ｂ）に示された端子３０ａ（Ｓ４）からフィルタ１１および２０を見たバンドＡおよびＢのインピーダンスの位置と、図６Ｂの（ｂ）に示された端子３０ａ（Ｓ４）からフィルタ１２および２０を見たバンドＡおよびＢのインピーダンスの位置とは、ほぼ一致している。 The position of the impedance of bands A and B when looking at the filters 11 and 20 from the terminal 30a (S4) shown in (b) of FIG. 6A and the filter from the terminal 30a (S4) shown in (b) of FIG. 6B The impedance positions of bands A and B, looking at 12 and 20, are nearly identical.
　これは、図５の（ｂ）および（ｃ）に示すように、Ｓ１からフィルタ１１単体を見たバンドＢのインピーダンスと、Ｓ２からフィルタ１２単体を見たバンドＢのインピーダンスとが、ほぼ一致していることに起因する。 This is because, as shown in FIGS. 5(b) and 5(c), the impedance of band B when the filter 11 alone is viewed from S1 and the impedance of band B when the filter 12 alone is viewed from S2 substantially match each other. due to the fact that
　これにより、図６Ａの（ｃ）および図６Ｂの（ｃ）に示すように、インダクタ９０により、端子３０ａ（Ｓ４）からフィルタ１１および２０を見たバンドＡおよびＢのインピーダンスと、端子３０ａ（Ｓ４）からフィルタ１２および２０を見たバンドＡおよびＢのインピーダンスとを、共に等コンダクタンス円の反時計まわりにシフトさせ、共に基準インピーダンスに整合させることが可能となる。 As a result, as shown in (c) of FIG. 6A and (c) of FIG. ), looking at filters 12 and 20, can both be shifted counterclockwise around the circle of equal conductance and both matched to the reference impedance.
　つまり、バンドＡの送信信号とバンドＢの信号とを同時伝送する場合に端子３０ａからフィルタ１１および２０を見たバンドＢのインピーダンスと、バンドＡの受信信号とバンドＢの信号とを同時伝送する場合に端子３０ａからフィルタ１２および２０を見たバンドＢのインピーダンスとを、ほぼ一致させることが可能となる。これにより、バンドＡの受信信号とバンドＢの信号とを同時伝送する場合におけるバンドＡの受信信号における符号誤り率が上昇することを抑制できる。 In other words, when simultaneously transmitting the transmission signal of band A and the signal of band B, the impedance of band B viewed from the terminal 30a of the filters 11 and 20, the received signal of band A and the signal of band B are simultaneously transmitted. In this case, it is possible to substantially match the impedance of band B when filters 12 and 20 are viewed from terminal 30a. As a result, it is possible to suppress an increase in the bit error rate in the received signal of band A when the received signal of band A and the signal of band B are simultaneously transmitted.
　実施の形態に係る高周波回路１において、フィルタ１２の並列腕共振子２５１に直列接続されたインダクタ２５０のインダクタンス値を調整することにより、端子３０ｃから見たフィルタ１２の自帯域（バンドＡ）のインピーダンスをシフトさせることなく、相手帯域（バンドＢ）のインピーダンスの位相幅を調整できる。よって、フィルタ１１および１２とスイッチ回路３０との間に配置される位相調整回路を削減して、バンドＡの送信信号と受信信号とを切り替える際に生じるインピーダンス整合の差異を低減しつつ、バンドＡの送信信号または受信信号とバンドＢの信号とを同時伝送することが可能となる。言い換えると、位相調整回路を削減できるので、ＴＤＤ用のバンドＡの高周波信号とバンドＢの高周波信号とを同時伝送する場合における伝送損失を低減することが可能となる。 In the high frequency circuit 1 according to the embodiment, by adjusting the inductance value of the inductor 250 connected in series with the parallel arm resonator 251 of the filter 12, the impedance of the own band (band A) of the filter 12 viewed from the terminal 30c is It is possible to adjust the phase width of the impedance of the other band (band B) without shifting the . Therefore, the number of phase adjustment circuits arranged between the filters 11 and 12 and the switch circuit 30 is eliminated to reduce the difference in impedance matching that occurs when switching between the transmission signal and the reception signal of the band A. , and the B-band signal can be transmitted simultaneously. In other words, since the number of phase adjustment circuits can be reduced, it is possible to reduce the transmission loss when simultaneously transmitting the high-frequency signal of band A and the high-frequency signal of band B for TDD.
　また、高周波回路１では、フィルタ１１および１２のうち、フィルタ１２においてスイッチ回路３０に最も近く接続された並列腕共振子２５１にインダクタ２５０が直列接続されている。これによれば、フィルタ１２が配置された受信経路に配置される位相調整回路を削減できるので、当該受信経路を伝送する受信信号の伝送損失を低減できる、よって、ＴＤＤ用のバンドＡの受信信号とバンドＢの信号とを同時伝送する場合におけるバンドＡの受信感度の劣化を、より抑制できる。 In addition, in the high-frequency circuit 1 , the inductor 250 is connected in series to the parallel arm resonator 251 of the filters 11 and 12 that is connected closest to the switch circuit 30 in the filter 12 . According to this, it is possible to reduce the number of phase adjustment circuits arranged in the receiving path in which the filter 12 is arranged, so that the transmission loss of the received signal transmitted through the receiving path can be reduced. and a signal of band B are simultaneously transmitted, deterioration of reception sensitivity of band A can be further suppressed.
　［５　変形例１に係る高周波回路１Ａの回路構成およびインピーダンス特性］
　図７は、変形例１に係る高周波回路１Ａの回路構成図、およびスイッチ選択端子から各フィルタを見たインピーダンスを示すスミスチャートである。同図の（ａ）に示すように、高周波回路１Ａは、フィルタ１１、１２および２０と、スイッチ回路３０と、インダクタ９１、９２および９３と、整合回路７０と、電力増幅器４１と、低雑音増幅器４２と、アンテナ接続端子１００と、入力端子１１０と、出力端子１２０と、入出力端子１３０と、を備える。本変形例に係る高周波回路１Ａは、実施の形態に係る高周波回路１と比較して、インダクタ９１～９３が付加されている点、および、インダクタ９０に代わって整合回路７０が配置されている点が異なる。以下では、本変形例に係る高周波回路１Ａについて、実施の形態に係る高周波回路１と同じ点は説明を省略し、高周波回路１と異なる点を中心に説明する。 [5 Circuit Configuration and Impedance Characteristics of High-Frequency Circuit 1A According to Modification 1]
FIG. 7 is a circuit configuration diagram of a high-frequency circuit 1A according to Modification 1, and a Smith chart showing the impedance of each filter viewed from the switch selection terminal. As shown in (a) of the figure, the high frequency circuit 1A includes filters 11, 12 and 20, a switch circuit 30, inductors 91, 92 and 93, a matching circuit 70, a power amplifier 41, a low noise amplifier 42 , an antenna connection terminal 100 , an input terminal 110 , an output terminal 120 and an input/output terminal 130 . A high-frequency circuit 1A according to this modification differs from the high-frequency circuit 1 according to the embodiment in that inductors 91 to 93 are added, and that a matching circuit 70 is arranged instead of the inductor 90. is different. In the following, regarding the high-frequency circuit 1A according to this modified example, the description of the same points as those of the high-frequency circuit 1 according to the embodiment will be omitted, and the points different from the high-frequency circuit 1 will be mainly described.
　インダクタ９１は、第３インダクタの一例であり、端子３０ｂとフィルタ１１とを結ぶ第１直列腕経路とグランドとの間に接続されている。 The inductor 91 is an example of a third inductor, and is connected between the first series arm path connecting the terminal 30b and the filter 11 and the ground.
　インダクタ９２は、第４インダクタの一例であり、端子３０ｃとフィルタ１２とを結ぶ第２直列腕経路とグランドとの間に接続されている。 The inductor 92 is an example of a fourth inductor, and is connected between the second series arm path connecting the terminal 30c and the filter 12 and the ground.
　インダクタ９３は、第５インダクタの一例であり、端子３０ｄとフィルタ２０とを結ぶ第３直列腕経路とグランドとの間に接続されている。 The inductor 93 is an example of a fifth inductor, and is connected between the third series arm path connecting the terminal 30d and the filter 20 and the ground.
　整合回路７０は、アンテナ接続端子１００と端子３０ａとの間に接続されている。整合回路７０は、インダクタ９４と、キャパシタ８０と、を備える。インダクタ９４は、第６インダクタの一例であり、アンテナ接続端子１００と端子３０ａとを結ぶアンテナ接続経路に直列配置されている。キャパシタ８０は、アンテナ接続端子１００と端子３０ａとを結ぶアンテナ接続経路とグランドとの間に接続されている。 The matching circuit 70 is connected between the antenna connection terminal 100 and the terminal 30a. Matching circuit 70 includes inductor 94 and capacitor 80 . The inductor 94 is an example of a sixth inductor, and is arranged in series in the antenna connection path connecting the antenna connection terminal 100 and the terminal 30a. The capacitor 80 is connected between the antenna connection path connecting the antenna connection terminal 100 and the terminal 30a and the ground.
　アンテナ接続端子１００は、整合回路７０を介して端子３０ａに接続されている。フィルタ１１は端子３０ｂと入力端子１１０との間に接続されている。フィルタ１２は端子３０ｃと出力端子１２０との間に接続されている。フィルタ２０は端子３０ｄと入出力端子１３０との間に接続されている。 The antenna connection terminal 100 is connected to the terminal 30a via the matching circuit 70. Filter 11 is connected between terminal 30b and input terminal 110 . Filter 12 is connected between terminal 30 c and output terminal 120 . Filter 20 is connected between terminal 30 d and input/output terminal 130 .
　なお、本変形例に係る高周波回路１Ａは、電力増幅器４１、低雑音増幅器４２、および整合回路７０を備えていなくてもよい。 It should be noted that the high frequency circuit 1A according to this modification need not include the power amplifier 41, the low noise amplifier 42, and the matching circuit 70.
　上記構成によれば、高周波回路１Ａは、バンドＡの送信信号とバンドＢの信号とを同時伝送することが可能であり、バンドＡの受信信号とバンドＢの信号とを同時伝送することが可能である。 According to the above configuration, the high-frequency circuit 1A can simultaneously transmit a transmission signal of band A and a signal of band B, and can simultaneously transmit a reception signal of band A and a signal of band B. is.
　図７の（ｂ）には、端子３０ｂ（Ｓ１）からフィルタ１１を見たインピーダンスを表すスミスチャートが示されている。また、図７の（ｃ）には、端子３０ｃ（Ｓ２）からフィルタ１２を見たインピーダンスを表すスミスチャートが示されている。また、図７の（ｄ）には、端子３０ｄ（Ｓ３）からフィルタ２０を見たインピーダンスを表すスミスチャートが示されている。 (b) of FIG. 7 shows a Smith chart representing the impedance of the filter 11 viewed from the terminal 30b (S1). Further, (c) of FIG. 7 shows a Smith chart representing the impedance of the filter 12 viewed from the terminal 30c (S2). FIG. 7(d) shows a Smith chart representing the impedance of the filter 20 viewed from the terminal 30d (S3).
　本変形例に係る高周波回路１Ａでは、ＴＤＤのバンドＡの送信信号および受信信号を伝送するにあたり、送信用のフィルタ１１と受信用のフィルタ１２とを配置している。これは、電力増幅器４１の出力インピーダンスの最適値と低雑音増幅器４２の入力インピーダンスの最適値とが異なるため、電力増幅器４１および低雑音増幅器４２に接続されるフィルタのインピーダンス特性を異ならせる必要があることに起因する。 In the high-frequency circuit 1A according to this modified example, a transmission filter 11 and a reception filter 12 are arranged for transmitting transmission signals and reception signals of band A of TDD. This is because the optimum value of the output impedance of the power amplifier 41 and the optimum value of the input impedance of the low noise amplifier 42 are different, so it is necessary to differentiate the impedance characteristics of the filters connected to the power amplifier 41 and the low noise amplifier 42. due to that.
　これに対して、本変形例に係る高周波回路１Ａでは、比較例に係る高周波回路５００と異なり、Ｓ１からフィルタ１１を見たインピーダンス（図７の（ｂ））と、Ｓ２からフィルタ１２を見たインピーダンス（図７の（ｃ））とでは、相手帯域（バンドＢ）のインピーダンスの位相が揃っている。 On the other hand, in the high-frequency circuit 1A according to this modified example, unlike the high-frequency circuit 500 according to the comparative example, the impedance ((b) in FIG. 7) when the filter 11 is viewed from S1 and the impedance when the filter 12 is viewed from S2 The impedance ((c) of FIG. 7) is in phase with the impedance of the other band (band B).
　これは、フィルタ１２が有するインダクタ２５０に起因するものである。本変形例に係るフィルタ１２では、スイッチ回路３０（端子３０ｃ）に最も近く接続された並列腕共振子２５１にインダクタ２５０が直列接続されているため、フィルタ１２の出力インピーダンス（低雑音増幅器４２の入力端子からフィルタ１２を見たインピーダンス）を変化させずに、Ｓ２からフィルタ１２を見た相手帯域（バンドＢ）のインピーダンスの位相幅を変化させることが可能となる。 This is due to the inductor 250 that the filter 12 has. In the filter 12 according to this modification, since the inductor 250 is connected in series with the parallel arm resonator 251 that is connected closest to the switch circuit 30 (terminal 30c), the output impedance of the filter 12 (the input of the low noise amplifier 42 It is possible to change the phase width of the impedance of the other band (band B) when the filter 12 is viewed from S2 without changing the impedance when the filter 12 is viewed from the terminal.
　つまり、フィルタ１１が電力増幅器４１とインピーダンス整合し、フィルタ１２が低雑音増幅器４２とインピーダンス整合しているにも拘わらず、Ｓ１からフィルタ１１を見たバンドＢのインピーダンスの位相と、Ｓ２からフィルタ１２を見たバンドＢのインピーダンスの位相とを揃わせることが可能となる。 In other words, although the filter 11 is impedance-matched with the power amplifier 41 and the filter 12 is impedance-matched with the low-noise amplifier 42, the impedance phase of the band B when the filter 11 is viewed from S1 and the filter 12 from S2 It is possible to align the phase of the impedance of the band B seen from .
　さらに、Ｓ１からフィルタ１１を見た相手帯域（バンドＢ）のインピーダンス、Ｓ２からフィルタ１２を見た相手帯域（バンドＢ）のインピーダンス、および、Ｓ３からフィルタ２０を見た相手帯域（バンドＡ）のインピーダンスは、インダクタ９１、９２および９３の配置により、実施の形態に係る高周波回路１と比較して、等コンダクタンス円上を反時計回りにシフトしている。このため、Ｓ１からフィルタ１１を見た相手帯域（バンドＢ）のインピーダンス、Ｓ２からフィルタ１２を見た相手帯域（バンドＢ）のインピーダンス、および、Ｓ３からフィルタ２０を見た相手帯域（バンドＡ）のインピーダンスは、容量性領域から、よりオープン領域へとシフトしている。 Furthermore, the impedance of the partner band (band B) when the filter 11 is viewed from S1, the impedance of the partner band (band B) when the filter 12 is viewed from S2, and the partner band (band A) when the filter 20 is viewed from S3 Due to the arrangement of inductors 91, 92 and 93, the impedance is shifted counterclockwise on the equal conductance circle compared to the high frequency circuit 1 according to the embodiment. Therefore, the impedance of the partner band (band B) when looking at the filter 11 from S1, the impedance of the partner band (band B) when looking at the filter 12 from S2, and the partner band (band A) when looking at the filter 20 from S3 is shifted from the capacitive region to a more open region.
　これにより、端子３０ｂからフィルタ１１単体を見たバンドＢの反射損失、端子３０ｃからフィルタ１２単体を見たバンドＢの反射損失、および端子３０ｄからフィルタ２０を見たバンドＡの反射損失を低減できる。 As a result, the reflection loss of band B when the filter 11 alone is viewed from the terminal 30b, the reflection loss of the band B when the filter 12 alone is viewed from the terminal 30c, and the band A reflection loss when the filter 20 is viewed from the terminal 30d can be reduced. .
　図８Ａは、変形例１に係る高周波回路１Ａの回路構成図、およびアンテナ接続端子１００から共通接続されたフィルタ１１および２０を見たインピーダンスを示すスミスチャートである。具体的には、図８Ａの（ｂ）には、フィルタ１１および２０を端子３０ａに接続した場合にアンテナ接続端子１００からフィルタ１１および２０を見たバンドＡおよびＢのインピーダンスが示されている。 FIG. 8A is a circuit configuration diagram of a high-frequency circuit 1A according to Modification 1, and a Smith chart showing the impedance of the filters 11 and 20 commonly connected from the antenna connection terminal 100. FIG. Specifically, (b) of FIG. 8A shows impedances of bands A and B when the filters 11 and 20 are viewed from the antenna connection terminal 100 when the filters 11 and 20 are connected to the terminal 30a.
　図８Ｂは、変形例１に係る高周波回路１Ａの回路構成図、およびアンテナ接続端子１００から共通接続されたフィルタ１２および２０を見たインピーダンスを示すスミスチャートである。具体的には、図８Ｂの（ｂ）には、フィルタ１２および２０を端子３０ａに接続した場合にアンテナ接続端子１００からフィルタ１２および２０を見たバンドＡおよびＢのインピーダンスが示されている。 FIG. 8B is a circuit configuration diagram of the high-frequency circuit 1A according to Modification 1, and a Smith chart showing the impedance of the filters 12 and 20 commonly connected from the antenna connection terminal 100. FIG. Specifically, (b) of FIG. 8B shows the impedances of bands A and B when the filters 12 and 20 are viewed from the antenna connection terminal 100 when the filters 12 and 20 are connected to the terminal 30a.
　図８Ａの（ｂ）に示されたアンテナ接続端子１００からフィルタ１１および２０を見たバンドＡおよびＢのインピーダンスの位置と、図８Ｂの（ｂ）に示されたアンテナ接続端子１００からフィルタ１２および２０を見たバンドＡおよびＢのインピーダンスの位置とは、ほぼ一致している。 The impedance positions of the bands A and B when the filters 11 and 20 are seen from the antenna connection terminal 100 shown in (b) of FIG. 8A, and the filter 12 and the filter 12 from the antenna connection terminal 100 shown in (b) of FIG. The positions of the impedances of bands A and B when looking at 20 are almost the same.
　これは、図７の（ｂ）および（ｃ）に示すように、Ｓ１からフィルタ１１単体を見たバンドＢのインピーダンスと、Ｓ２からフィルタ１２単体を見たバンドＢのインピーダンスとが、ほぼ一致していることに起因する。 This is because, as shown in (b) and (c) of FIG. 7, the impedance of band B when the filter 11 alone is viewed from S1 and the impedance of band B when the filter 12 alone is viewed from S2 substantially match each other. due to the fact that
　なお、図８Ａの（ｂ）に示されたアンテナ接続端子１００からフィルタ１１および２０を見たバンドＡおよびＢのインピーダンス、および、図８Ｂの（ｂ）に示されたアンテナ接続端子１００からフィルタ１２および２０を見たバンドＡおよびＢのインピーダンスは、端子３０ａからフィルタ１１および２０またはフィルタ１２および２０を見たバンドＡおよびＢのインピーダンスに対して、インダクタ９４により等レジスタンス円を時計回りにシフトさせ、キャパシタ８０により等コンダクタンス円を時計回りにシフトさせることにより、基準インピーダンスに整合させている。 Note that the impedance of the bands A and B when the filters 11 and 20 are viewed from the antenna connection terminal 100 shown in (b) of FIG. 8A, and the filter 12 from the antenna connection terminal 100 shown in (b) of FIG. 8B and 20, the impedance of bands A and B looking into filters 11 and 20 or filters 12 and 20 from terminal 30a has the equal resistance circle shifted clockwise by inductor 94. , the equiconductance circle is shifted clockwise by the capacitor 80 to match the reference impedance.
　本変形例に係る高周波回路１Ａによれば、バンドＡの送信信号とバンドＢの信号とを同時伝送する場合に端子３０ａからフィルタ１１および２０を見たバンドＢのインピーダンスと、バンドＡの受信信号とバンドＢの信号とを同時伝送する場合に端子３０ａからフィルタ１２および２０を見たバンドＢのインピーダンスとをほぼ一致させることが可能となる。これにより、バンドＡの受信信号とバンドＢの信号とを同時伝送する場合におけるバンドＡの受信信号における符号誤り率が上昇することを抑制できる。 According to the high-frequency circuit 1A according to the present modification, when the transmission signal of band A and the signal of band B are simultaneously transmitted, the impedance of band B when the filters 11 and 20 are viewed from the terminal 30a and the reception signal of band A are and a signal of band B are simultaneously transmitted, the impedance of band B when filters 12 and 20 are viewed from terminal 30a can be substantially matched. As a result, it is possible to suppress an increase in the bit error rate in the received signal of band A when the received signal of band A and the signal of band B are simultaneously transmitted.
　また、フィルタ１１および１２とスイッチ回路３０との間に配置される位相調整回路を削減して、バンドＡの送信信号と受信信号とを切り替える際に生じるインピーダンス整合の差異を低減しつつ、バンドＡの送信信号または受信信号とバンドＢの信号とを同時伝送することが可能となる。言い換えると、位相調整回路を削減できるので、ＴＤＤ用のバンドＡの高周波信号とバンドＢの高周波信号とを同時伝送する場合における伝送損失を低減することが可能となる。 In addition, the number of phase adjustment circuits arranged between the filters 11 and 12 and the switch circuit 30 is eliminated to reduce the difference in impedance matching that occurs when switching between the transmission signal and the reception signal of the band A. , and the B-band signal can be transmitted simultaneously. In other words, since the number of phase adjustment circuits can be reduced, it is possible to reduce the transmission loss when simultaneously transmitting the high-frequency signal of band A and the high-frequency signal of band B for TDD.
　さらに、高周波回路１Ａでは、インダクタ９１～９３により、端子３０ｂからフィルタ１１単体を見たバンドＢの反射損失、端子３０ｃからフィルタ１２単体を見たバンドＢの反射損失、および端子３０ｄからフィルタ２０を見たバンドＡの反射損失を低減しているので、ＴＤＤ用のバンドＡの信号とバンドＢの信号とを同時伝送する場合における信号伝送損失を、より低減できる。 Furthermore, in the high-frequency circuit 1A, the inductors 91 to 93 provide a reflection loss for the band B when the filter 11 alone is viewed from the terminal 30b, a reflection loss for the band B when the filter 12 alone is viewed from the terminal 30c, and a filter 20 from the terminal 30d. Since the reflection loss of the visible band A is reduced, the signal transmission loss when simultaneously transmitting the TDD band A signal and the band B signal can be further reduced.
　［６　変形例２に係る高周波回路１Ｂの回路構成およびインピーダンス特性］
　図９は、変形例２に係る高周波回路１Ｂの回路構成図である。同図に示すように、高周波回路１Ｂは、フィルタ１１、１２および２０と、スイッチ回路３０と、インダクタ９１、９２、９３および９５と、電力増幅器４１と、低雑音増幅器４２と、アンテナ接続端子１００と、入力端子１１０と、出力端子１２０と、入出力端子１３０と、を備える。本変形例に係る高周波回路１Ｂは、実施の形態に係る高周波回路１と比較して、インダクタ９１～９３、９５が付加されている点、および、インダクタ９０が付加されていない点が異なる。以下では、本変形例に係る高周波回路１Ｂについて、実施の形態に係る高周波回路１と同じ点は説明を省略し、高周波回路１と異なる点を中心に説明する。 [6 Circuit Configuration and Impedance Characteristics of High-Frequency Circuit 1B According to Modification 2]
FIG. 9 is a circuit configuration diagram of a high frequency circuit 1B according to Modification 2. As shown in FIG. As shown in the figure, the high frequency circuit 1B includes filters 11, 12 and 20, a switch circuit 30, inductors 91, 92, 93 and 95, a power amplifier 41, a low noise amplifier 42, an antenna connection terminal 100 , an input terminal 110 , an output terminal 120 , and an input/output terminal 130 . A high-frequency circuit 1B according to the present modification differs from the high-frequency circuit 1 according to the embodiment in that inductors 91 to 93 and 95 are added and inductor 90 is not added. In the following, regarding the high-frequency circuit 1B according to this modified example, the description of the same points as the high-frequency circuit 1 according to the embodiment will be omitted, and the points different from the high-frequency circuit 1 will be mainly described.
　インダクタ９５は、第７インダクタの一例であり、端子３０ｂとフィルタ１１とを結ぶ第１直列腕経路に直列配置されている。 The inductor 95 is an example of a seventh inductor, and is arranged in series in the first series arm path connecting the terminal 30 b and the filter 11 .
　インダクタ９１は、第８インダクタの一例であり、端子３０ｂとフィルタ１１とを結ぶ第１直列腕経路とグランドとの間に接続されている。 The inductor 91 is an example of an eighth inductor, and is connected between the first series arm path connecting the terminal 30b and the filter 11 and the ground.
　インダクタ９２は、第９インダクタの一例であり、端子３０ｃとフィルタ１２とを結ぶ第２直列腕経路とグランドとの間に接続されている。 The inductor 92 is an example of a ninth inductor, and is connected between the second series arm path connecting the terminal 30c and the filter 12 and the ground.
　インダクタ９３は、端子３０ｄとフィルタ２０とを結ぶ第３直列腕経路とグランドとの間に接続されている。 The inductor 93 is connected between the third series arm path connecting the terminal 30d and the filter 20 and the ground.
　なお、本変形例に係る高周波回路１Ｂは、電力増幅器４１および低雑音増幅器４２を備えていなくてもよい。 It should be noted that the high frequency circuit 1B according to this modification need not include the power amplifier 41 and the low noise amplifier 42.
　上記構成によれば、高周波回路１Ｂは、バンドＡの送信信号とバンドＢの信号とを同時伝送することが可能であり、バンドＡの受信信号とバンドＢの信号とを同時伝送することが可能である。 According to the above configuration, the high-frequency circuit 1B can simultaneously transmit a transmission signal of band A and a signal of band B, and can simultaneously transmit a reception signal of band A and a signal of band B. is.
　図１０は、変形例２に係る高周波回路１Ｂにおいて、送信経路上の各点からフィルタ１１を見たインピーダンスを示すスミスチャート、および、受信経路上の各点からフィルタ１２を見たインピーダンスを示すスミスチャートである。なお、本変形例では、バンドＡの信号と同時伝送可能な信号はバンドＢだけでなく、バンドＣ、バンドＤ、バンドＥ、バンドＦ、およびバンドＧの信号も含むものとしている。つまり、本変形例に係る高周波回路１Ｂでは、スイッチ回路３０に接続されるフィルタは、フィルタ１１、１２および２０の他、バンドＣ～バンドＧのそれぞれを通過帯域とするフィルタを含んでもよい。 FIG. 10 is a Smith chart showing the impedance of the filter 11 viewed from each point on the transmission path, and a Smith chart showing the impedance of the filter 12 viewed from each point on the reception path in the high-frequency circuit 1B according to Modification 2. Chart. In this modification, signals that can be transmitted simultaneously with band A signals include not only band B signals, but also band C, band D, band E, band F, and band G signals. That is, in the high-frequency circuit 1B according to this modification, the filters connected to the switch circuit 30 may include filters having the band C to band G as their passbands, in addition to the filters 11, 12, and 20.
　このとき、バンドＡは、例えば、４Ｇ－ＬＴＥのバンド４１または５Ｇ－ＮＲのバンドｎ４１が適用される。バンドＢは、例えば、４Ｇ－ＬＴＥのバンド４０または５Ｇ－ＮＲのバンドｎ４０が適用される。バンドＣは、４Ｇ－ＬＴＥのバンド１または５Ｇ－ＮＲのバンドｎ１（ダウンリンク動作バンド）が適用される。バンドＤは、４Ｇ－ＬＴＥのバンド１または５Ｇ－ＮＲのバンドｎ１（アップリンク動作バンド）が適用される。バンドＥは、４Ｇ－ＬＴＥのバンド３または５Ｇ－ＮＲのバンドｎ３（ダウンリンク動作バンド：１８０５－１８８０ＭＨｚ）が適用される。バンドＦは、４Ｇ－ＬＴＥのバンド３または５Ｇ－ＮＲのバンドｎ３（アップリンク動作バンド：１７１０－１７８５ＭＨｚ）が適用される。バンドＧは、４Ｇ－ＬＴＥのバンド３２または５Ｇ－ＮＲのバンドｎ３２（ダウンリンク動作バンド）が適用される。このとき、周波数は、高い方から、バンドＡ、バンドＢ、バンドＣ、バンドＤ、バンドＥ、バンドＦ、バンドＧとなっている。 At this time, for band A, for example, band 41 of 4G-LTE or band n41 of 5G-NR is applied. For band B, for example, band 40 of 4G-LTE or band n40 of 5G-NR is applied. Band C is applied to band 1 of 4G-LTE or band n1 (downlink operating band) of 5G-NR. Band D is applied to band 1 of 4G-LTE or band n1 (uplink operation band) of 5G-NR. Band E is applied to band 3 of 4G-LTE or band n3 of 5G-NR (downlink operating band: 1805-1880 MHz). Band F applies to 4G-LTE band 3 or 5G-NR band n3 (uplink operating band: 1710-1785 MHz). Band G is applied to 4G-LTE band 32 or 5G-NR band n32 (downlink operating band). At this time, the frequencies are band A, band B, band C, band D, band E, band F, and band G from the highest.
　図１０の（ａ）には、Ｓ６（図９に図示）からフィルタ１１を見たインピーダンスを表すスミスチャートが示されている。本インピーダンスは、電力増幅器４１の出力インピーダンスと最適整合させるため、バンドＡのインピーダンスは基準インピーダンスよりも容量性領域かつ高インピーダンス領域に位置し、バンドＡのインピーダンスの集中度は高くなる。また、自帯域（バンドＡ）よりも低周波側の相手帯域（バンドＢ～バンドＧ）のインピーダンスは、容量性領域に位置し、低周波側であるほどオープン領域に近く位置している。 FIG. 10(a) shows a Smith chart representing the impedance of the filter 11 viewed from S6 (shown in FIG. 9). Since this impedance is optimally matched with the output impedance of the power amplifier 41, the impedance of band A is located in a capacitive region and a high impedance region than the reference impedance, and the concentration of the impedance of band A is high. In addition, the impedance of the other band (band B to band G) on the lower frequency side than the own band (band A) is located in the capacitive area, and the lower the frequency side, the closer to the open area.
　図１０の（ｂ）には、Ｓ７（図９に図示）からフィルタ１１を見たインピーダンスを表すスミスチャートが示されている。インダクタ９５のインダクタンス値をＬ９５とした場合、Ｓ７からフィルタ１１を見たインピーダンスは、Ｓ６からフィルタ１１を見たインピーダンスと比較して、等レジスタンス円上をｊωＬ９５の分だけ時計回りにシフトする。これにより、フィルタ１１における自帯域（バンドＡ）のインピーダンスを基準インピーダンス近傍に位置させる。これに伴い、高周波側に位置するバンドＢのシフト量が低周波側に位置するバンドＧのシフト量よりも大きくなる。つまり、インダクタ９５の付加により自帯域（バンドＡ）のインピーダンスは基準インピーダンス側へとシフトし、相手帯域（バンドＢ～Ｇ）のインピーダンスは、ショート領域へとシフトしつつ位相幅が大きくなる。 FIG. 10(b) shows a Smith chart representing the impedance of the filter 11 viewed from S7 (shown in FIG. 9). If the inductance value of inductor 95 is L 95 , then the impedance looking at filter 11 from S7 is shifted clockwise by jωL 95 on the equal resistance circle compared to the impedance looking at filter 11 from S6. . As a result, the impedance of its own band (band A) in the filter 11 is positioned near the reference impedance. Accordingly, the amount of shift in band B located on the high frequency side becomes larger than the amount of shift in band G located on the low frequency side. In other words, the addition of the inductor 95 shifts the impedance of the own band (band A) toward the reference impedance side, and the impedance of the other band (bands B to G) shifts toward the short region while increasing the phase width.
　図１０の（ｃ）には、Ｓ１（図９に図示）からフィルタ１１を見たインピーダンスを表すスミスチャートが示されている。インダクタ９１のインダクタンス値をＬ９１とした場合、Ｓ１からフィルタ１１を見たインピーダンスは、Ｓ７からフィルタ１１を見たインピーダンスと比較して、等コンダクタンス円上を１／ｊωＬ９１に相当する分だけ反時計回りにシフトする。これにより、低周波側に位置するバンドＧのシフト量が高周波側に位置するバンドＢのシフト量よりも大きくなる。つまり、インダクタ９１の付加により自帯域（バンドＡ）のインピーダンスは基準インピーダンス側へとシフトし、相手帯域（バンドＢ～Ｇ）のインピーダンスは、オープン領域へとシフトしつつ、さらに位相幅が大きくなる。 FIG. 10(c) shows a Smith chart representing the impedance of the filter 11 viewed from S1 (shown in FIG. 9). When the inductance value of the inductor 91 is L 91 , the impedance of the filter 11 viewed from S1 is reflected from the impedance of the filter 11 viewed from S7 on the equal conductance circle by an amount corresponding to 1/jωL 91 . Shift clockwise. As a result, the shift amount of the band G located on the low frequency side becomes larger than the shift amount of the band B located on the high frequency side. In other words, the addition of the inductor 91 shifts the impedance of the own band (band A) toward the reference impedance side, and the impedance of the other band (bands B to G) shifts toward the open region, while the phase width further increases. .
　図１０の（ｄ）には、Ｓ８（図９に図示）からフィルタ１２を見たインピーダンスを表すスミスチャートが示されている。本インピーダンスは、低雑音増幅器４２の入力インピーダンスと最適整合させるため、バンドＡのインピーダンスは基準インピーダンスよりも容量性領域に位置している。また、自帯域（バンドＡ）よりも低周波側の相手帯域（バンドＢ～バンドＧ）のインピーダンスは、さらに容量性領域に位置し、低周波側であるほどオープン領域に近く位置している。 (d) of FIG. 10 shows a Smith chart representing the impedance of the filter 12 viewed from S8 (shown in FIG. 9). In order to optimally match this impedance with the input impedance of the low noise amplifier 42, the band A impedance is located in the capacitive region rather than the reference impedance. In addition, the impedance of the other band (band B to band G) on the lower frequency side than the own band (band A) is located in the capacitive area, and the lower the frequency side, the closer to the open area.
　ここで、フィルタ１２にはインダクタ２５０が配置されているため、インダクタ２５０のインダクタンス値Ｌ２５０を調整することにより、相手帯域（バンドＢ～Ｇ）のインピーダンスの位相幅を変化させることが可能である。例えば、図１０の（ｄ）に示すように、Ｌ２５０が相対的に大きい場合には相手帯域（バンドＢ～Ｇ）のインピーダンスの位相幅は大きくなる。一方、図１０の（ｄ’）に示すように、Ｌ２５０が相対的に小さい場合には相手帯域（バンドＢ～Ｇ）のインピーダンスの位相幅は小さくなる。 Here, since the inductor 250 is arranged in the filter 12, by adjusting the inductance value L 250 of the inductor 250, it is possible to change the phase width of the impedance of the other band (bands B to G). . For example, as shown in (d) of FIG. 10, when L 250 is relatively large, the phase width of the impedance of the other band (bands B to G) is large. On the other hand, as shown in (d') of FIG. 10, when L 250 is relatively small, the phase width of the impedance of the partner band (bands B to G) is small.
　図１０の（ｅ）には、Ｓ２（図９に図示）からフィルタ１２を見たインピーダンスを表すスミスチャートが示されている。インダクタ９２のインダクタンス値をＬ９２とした場合、Ｓ２からフィルタ１２を見たインピーダンスは、Ｓ８からフィルタ１２を見たインピーダンスと比較して、等コンダクタンス円上を１／ｊωＬ９２に相当する分だけ反時計回りにシフトする。これにより、低周波側に位置するバンドＧのシフト量が高周波側に位置するバンドＢのシフト量よりも大きくなる。つまり、インダクタ９２の付加により自帯域（バンドＡ）のインピーダンスは基準インピーダンス側へとシフトし、相手帯域（バンドＢ～Ｇ）のインピーダンスは、オープン領域へとシフトしつつ、位相幅が大きくなる。 FIG. 10(e) shows a Smith chart representing the impedance of the filter 12 viewed from S2 (shown in FIG. 9). When the inductance value of the inductor 92 is L 92 , the impedance of the filter 12 viewed from S2 is a reflection of 1/jωL 92 on the equal conductance circle compared to the impedance of the filter 12 viewed from S8. Shift clockwise. As a result, the shift amount of the band G located on the low frequency side becomes larger than the shift amount of the band B located on the high frequency side. In other words, the addition of the inductor 92 shifts the impedance of the own band (band A) toward the reference impedance side, and the impedance of the other band (bands B to G) shifts toward the open region while increasing the phase width.
　以上のインピーダンス整合により、Ｓ１からフィルタ１１を見たインピーダンスと、Ｓ２からフィルタ１２を見たインピーダンスとを揃えることが可能となる。 With the above impedance matching, it is possible to match the impedance of the filter 11 viewed from S1 and the impedance of the filter 12 viewed from S2.
　つまり、フィルタ１１単体の自帯域（バンドＡ）のインピーダンスを、直列接続されたインダクタ９５および並列接続されたインダクタ９１で基準インピーダンス近傍にシフトさせ、フィルタ１２単体における自帯域（バンドＡ）のインピーダンスを、並列接続されたインダクタ９２で基準インピーダンス近傍にシフトさせる。また、フィルタ１１における相手帯域（バンドＢ～Ｇ）のインピーダンスおよびフィルタ１２における相手帯域（バンドＢ～Ｇ）のインピーダンスの位置および位相幅を合わせるべく、フィルタ１２が有するインダクタ２５０により、自帯域（バンドＡ）のインピーダンスをシフトさせることなく、相手帯域（バンドＢ～Ｇ）のインピーダンスの位相幅を調整している。 That is, the impedance of the own band (band A) of the filter 11 alone is shifted to the vicinity of the reference impedance by the inductor 95 connected in series and the inductor 91 connected in parallel, and the impedance of the own band (band A) of the filter 12 alone is changed to , is shifted to the vicinity of the reference impedance by an inductor 92 connected in parallel. In addition, in order to match the position and phase width of the impedance of the partner band (bands B to G) in the filter 11 and the impedance of the partner band (bands B to G) in the filter 12, the inductor 250 of the filter 12 adjusts the own band (band The phase width of the impedance of the other band (bands B to G) is adjusted without shifting the impedance of A).
　これによれば、フィルタ１２に直列接続されるインダクタ（位相調整回路）を削減して、バンドＡの送信信号と受信信号とを切り替える際に生じるインピーダンス整合の差異を低減しつつ、バンドＡの送信信号または受信信号とバンドＢ～Ｇの信号とを同時伝送することが可能となる。 According to this, the number of inductors (phase adjustment circuits) connected in series to the filter 12 is reduced, and the difference in impedance matching that occurs when switching between the transmission signal and the reception signal of band A is reduced. Simultaneous transmission of signals or received signals and signals of bands B to G is possible.
　［７　効果など］
　以上のように、実施の形態に係る高周波回路１は、アンテナ接続端子１００、入力端子１１０および出力端子１２０と、ＴＤＤ用のバンドＡの送信帯域を含む通過帯域を有するフィルタ１１と、バンドＡの受信帯域を含む通過帯域を有するフィルタ１２と、バンドＡと異なるバンドＢの少なくとも一部を含む通過帯域を有するフィルタ２０と、端子３０ａ、３０ｂ、３０ｃおよび３０ｄを有し、端子３０ａおよび３０ｂの接続と端子３０ａおよび３０ｄの接続とを同時に実行可能であり、端子３０ａおよび３０ｃの接続と端子３０ａおよび３０ｄの接続とを同時に実行可能であるスイッチ回路３０と、を備える。アンテナ接続端子１００は端子３０ａに接続され、フィルタ１１は端子３０ｂと入力端子１１０との間に接続され、フィルタ１２は端子３０ｃと出力端子１２０との間に接続され、フィルタ２０は端子３０ｄに接続され、フィルタ１１および１２の一方は、スイッチ回路３０と入力端子１１０および出力端子１２０の一方とを結ぶ直列腕経路とグランドとの間に接続された第１弾性波共振子と、上記直列腕経路とグランドとの間で第１弾性波共振子と直列接続されたインダクタ２５０と、を備え、第１弾性波共振子は上記直列腕経路とグランドとの間に接続された並列腕共振子のうちでスイッチ回路３０に最も近い位置で接続されている。 [7 Effects, etc.]
As described above, the high-frequency circuit 1 according to the embodiment includes the antenna connection terminal 100, the input terminal 110 and the output terminal 120, the filter 11 having a passband including the transmission band of band A for TDD, a filter 12 having a passband including the receive band; a filter 20 having a passband including at least a portion of band B different from band A; terminals 30a, 30b, 30c and 30d; and connection of terminals 30a and 30d, and connection of terminals 30a and 30c and connection of terminals 30a and 30d. Antenna connection terminal 100 is connected to terminal 30a, filter 11 is connected between terminal 30b and input terminal 110, filter 12 is connected between terminal 30c and output terminal 120, and filter 20 is connected to terminal 30d. One of filters 11 and 12 includes a first elastic wave resonator connected between a series arm path connecting switch circuit 30 and one of input terminal 110 and output terminal 120 and ground, and the series arm path and an inductor 250 connected in series with the first acoustic wave resonator between and ground, wherein the first acoustic wave resonator is one of the parallel arm resonators connected between the series arm path and the ground is connected to the switch circuit 30 at the closest position.
　これによれば、フィルタ１１および１２の一方の最前段の並列腕共振子に直列接続されたインダクタ２５０のインダクタンス値を調整することにより、当該一方の自帯域のインピーダンスをシフトさせることなく、相手帯域のインピーダンスの位相幅を調整できる。よって、フィルタ１１および１２とスイッチ回路３０との間に配置される位相調整回路を削減して、バンドＡの送信信号と受信信号とを切り替える際に生じるインピーダンス整合の差異を低減しつつ、バンドＡの送信信号または受信信号とバンドＢの信号とを同時伝送することが可能となる。言い換えると、位相調整回路を削減できるので、ＴＤＤ用のバンドＡの高周波信号とバンドＢの高周波信号とを同時伝送する場合における伝送損失を低減することが可能となる。 According to this, by adjusting the inductance value of the inductor 250 connected in series to the foremost parallel arm resonator of one of the filters 11 and 12, the impedance of the other band can be adjusted without shifting the impedance of the one's own band. can adjust the phase width of the impedance of Therefore, the number of phase adjustment circuits arranged between the filters 11 and 12 and the switch circuit 30 is eliminated to reduce the difference in impedance matching that occurs when switching between the transmission signal and the reception signal of the band A. , and the B-band signal can be transmitted simultaneously. In other words, since the number of phase adjustment circuits can be reduced, it is possible to reduce the transmission loss when simultaneously transmitting the high-frequency signal of band A and the high-frequency signal of band B for TDD.
　また例えば、高周波回路１において、フィルタ１１および１２の上記一方は、フィルタ１２であり、第１弾性波共振子は、端子３０ｃと出力端子１２０とを結ぶ直列腕経路とグランドとの間に接続されていてもよい。 Further, for example, in the high-frequency circuit 1, one of the filters 11 and 12 is the filter 12, and the first acoustic wave resonator is connected between a series arm path connecting the terminal 30c and the output terminal 120 and the ground. may be
　これによれば、フィルタ１２が配置された受信経路に配置される位相調整回路を削減できるので、当該受信経路を伝送する受信信号の伝送損失を低減できる、よって、ＴＤＤ用のバンドＡの受信信号とバンドＢの信号とを同時伝送する場合におけるバンドＡの受信信号の受信感度の劣化を、より抑制できる。 According to this, it is possible to reduce the number of phase adjustment circuits arranged in the receiving path in which the filter 12 is arranged, so that the transmission loss of the received signal transmitted through the receiving path can be reduced. and a signal of band B are simultaneously transmitted, the deterioration of the receiving sensitivity of the received signal of band A can be further suppressed.
　また例えば、高周波回路１において、フィルタ１１および１２の他方は、インダクタ２５０を備えなくてもよい。 Also, for example, in the high-frequency circuit 1 , the other of the filters 11 and 12 may not include the inductor 250 .
　これによれば、簡素化された回路構成で、ＴＤＤ用のバンドＡの高周波信号とバンドＢの高周波信号とを同時伝送する場合における伝送損失を低減することが可能となる。 According to this, with a simplified circuit configuration, it is possible to reduce the transmission loss when simultaneously transmitting the high-frequency signal of band A and the high-frequency signal of band B for TDD.
　また例えば、高周波回路１は、さらに、アンテナ接続端子１００と端子３０ａとを結ぶアンテナ接続経路とグランドとの間に接続されたインダクタ９０を備えてもよい。 Further, for example, the high-frequency circuit 1 may further include an inductor 90 connected between the antenna connection path connecting the antenna connection terminal 100 and the terminal 30a and the ground.
　これによれば、端子３０ａからフィルタ１１および２０を見たバンドＡおよびＢのインピーダンスと、端子３０ａからフィルタ１２および２０を見たバンドＡおよびＢのインピーダンスとを、ともに等コンダクタンス円の反時計まわりにシフトさせ、基準インピーダンスに整合させることが可能となる。 According to this, the impedances of the bands A and B when the filters 11 and 20 are viewed from the terminal 30a and the impedances of the bands A and B when the filters 12 and 20 are viewed from the terminal 30a are both measured in the counterclockwise direction of the isoconductance circle. to match the reference impedance.
　また例えば、高周波回路１Ａは、さらに、端子３０ｂとフィルタ１１とを結ぶ第１直列腕経路とグランドとの間に接続されたインダクタ９１と、端子３０ｃとフィルタ１２とを結ぶ第２直列腕経路とグランドとの間に接続されたインダクタ９２と、端子３０ｄとフィルタ２０とを結ぶ第３直列腕経路とグランドとの間に接続されたインダクタ９３と、アンテナ接続端子と端子３０ａとの間に接続された整合回路７０と、を備えてもよい。 Further, for example, the high-frequency circuit 1A further includes an inductor 91 connected between a first series arm path connecting the terminal 30b and the filter 11 and the ground, and a second series arm path connecting the terminal 30c and the filter 12. An inductor 92 connected between the ground, an inductor 93 connected between the third series arm path connecting the terminal 30d and the filter 20 and the ground, and an inductor 93 connected between the antenna connection terminal and the terminal 30a. and a matching circuit 70 .
　これによれば、バンドＡの受信信号とバンドＢの信号とを同時伝送する場合におけるバンドＡの受信信号における符号誤り率が上昇することを抑制できる。また、フィルタ１１および１２とスイッチ回路３０との間に配置される位相調整回路を削減して、バンドＡの送信信号と受信信号とを切り替える際に生じるインピーダンス整合の差異を低減しつつ、バンドＡの送信信号または受信信号とバンドＢの信号とを同時伝送することが可能となる。さらに、インダクタ９１～９３により、端子３０ｂからフィルタ１１単体を見たバンドＢの反射損失、端子３０ｃからフィルタ１２単体を見たバンドＢの反射損失、および端子３０ｄからフィルタ２０を見たバンドＡの反射損失を低減しているので、ＴＤＤ用のバンドＡの信号とバンドＢの信号とを同時伝送する場合における信号伝送損失を、より低減できる。 According to this, it is possible to suppress the increase in the bit error rate in the received signal of band A when the received signal of band A and the signal of band B are simultaneously transmitted. In addition, the number of phase adjustment circuits arranged between the filters 11 and 12 and the switch circuit 30 is eliminated to reduce the difference in impedance matching that occurs when switching between the transmission signal and the reception signal of the band A. , and the B-band signal can be transmitted simultaneously. In addition, inductors 91 to 93 provide reflection loss for band B when the filter 11 alone is viewed from the terminal 30b, reflection loss for band B when the filter 12 alone is viewed from the terminal 30c, and band A when the filter 20 is viewed from the terminal 30d. Since the reflection loss is reduced, it is possible to further reduce the signal transmission loss in the simultaneous transmission of the band A signal and the band B signal for TDD.
　また例えば、高周波回路１Ａにおいて、整合回路７０は、アンテナ接続端子１００と端子３０ａとを結ぶアンテナ接続経路に直列配置されたインダクタ９４と、アンテナ接続端子１００と端子３０ａとを結ぶアンテナ接続経路とグランドとの間に接続されたキャパシタ８０と、を備えてもよい。 Further, for example, in the high-frequency circuit 1A, the matching circuit 70 includes an inductor 94 arranged in series in an antenna connection path connecting the antenna connection terminal 100 and the terminal 30a, an antenna connection path connecting the antenna connection terminal 100 and the terminal 30a, and a ground. and a capacitor 80 connected between.
　これによれば、端子３０ａからフィルタ１１および２０を見たバンドＡおよびＢのインピーダンスと、端子３０ａからフィルタ１２および２０を見たバンドＡおよびＢのインピーダンスとを、ともにインダクタ９４により等レジスタンス円を時計回りにシフトさせ、キャパシタ８０により等コンダクタンス円を時計回りにシフトさせることにより、基準インピーダンスに整合させることが可能となる。 According to this, the impedances of the bands A and B when the filters 11 and 20 are viewed from the terminal 30a and the impedances of the bands A and B when the filters 12 and 20 are viewed from the terminal 30a are formed into equal resistance circles by the inductor 94. The clockwise shift and the clockwise shift of the equiconductance circle by the capacitor 80 allows it to be matched to the reference impedance.
　また例えば、高周波回路１Ｂは、さらに、端子３０ｂとフィルタ１１とを結ぶ第１直列腕経路に直列配置されたインダクタ９５と、端子３０ｂとインダクタ９５とを結ぶ第１直列腕経路とグランドとの間に接続されたインダクタ９１と、端子３０ｃとフィルタ１２とを結ぶ第２直列腕経路とグランドとの間に接続されたインダクタ９２と、を備えてもよい。 Further, for example, the high-frequency circuit 1B further includes an inductor 95 arranged in series in a first series arm path connecting the terminal 30b and the filter 11, and an inductor 95 between the first series arm path connecting the terminal 30b and the inductor 95 and the ground. and an inductor 92 connected between a second series arm path connecting the terminal 30c and the filter 12 and the ground.
　これによれば、フィルタ１１単体のバンドＡのインピーダンスを、インダクタ９５および９１により基準インピーダンス近傍にシフトさせ、フィルタ１２単体におけるバンドＡのインピーダンスを、インダクタ９２により基準インピーダンス近傍にシフトさせる。また、フィルタ１１におけるバンドＢ～Ｇのインピーダンスとフィルタ１２におけるバンドＢ～Ｇのインピーダンスとの位置および位相幅を合わせるべく、フィルタ１２が有するインダクタ２５０により、バンドＡのインピーダンスをシフトさせることなく、バンドＢ～Ｇのインピーダンスの位相幅を調整できる。よって、フィルタ１２に直列接続されるインダクタ（位相調整回路）を削減して、バンドＡの送信信号と受信信号とを切り替える際に生じるインピーダンス整合の差異を低減しつつ、バンドＡの送信信号または受信信号とバンドＢ～Ｇの信号とを同時伝送することが可能となる。 According to this, the band A impedance of filter 11 alone is shifted to the vicinity of the reference impedance by inductors 95 and 91 , and the band A impedance of filter 12 alone is shifted to the vicinity of the reference impedance by inductor 92 . In addition, in order to match the positions and phase widths of the impedances of the bands B to G in the filter 11 and the impedances of the bands B to G in the filter 12, the inductor 250 of the filter 12 is used to adjust the impedance of the band A without shifting the impedance of the band A. The phase width of the impedance of B to G can be adjusted. Therefore, the number of inductors (phase adjustment circuits) connected in series with the filter 12 is reduced to reduce the difference in impedance matching that occurs when switching between the transmission signal and the reception signal of band A. It is possible to simultaneously transmit the signal and the signals of the bands B to G.
　また例えば、高周波回路１において、インダクタ２５０は、第１弾性波共振子とグランドとの間に接続されていてもよい。 Also, for example, in the high-frequency circuit 1, the inductor 250 may be connected between the first acoustic wave resonator and the ground.
　また例えば、高周波回路１において、フィルタ１１は入力端子１１０と端子３０ｂとを結ぶ直列腕経路に配置された１以上の直列腕共振子と、当該直列腕経路とグランドとの間に接続された１以上の並列腕共振子と、を備えてもよい。 Further, for example, in the high-frequency circuit 1, the filter 11 includes one or more series arm resonators arranged in a series arm path connecting the input terminal 110 and the terminal 30b, and one resonator connected between the series arm path and the ground. and the above parallel arm resonator.
　また例えば、高周波回路１において、フィルタ１２は、出力端子１２０と端子３０ｃとを結ぶ直列腕経路に配置された１以上の直列腕共振子と、当該直列腕経路とグランドとの間に接続された１以上の並列腕共振子と、を備えてもよい。 Further, for example, in the high-frequency circuit 1, the filter 12 is connected between one or more series arm resonators arranged in a series arm path connecting the output terminal 120 and the terminal 30c and between the series arm path and the ground. and one or more parallel arm resonators.
　また例えば、高周波回路１において、バンドＡの送信信号およびバンドＢの信号を同時に伝送する場合、端子３０ａと端子３０ｂとが接続され、かつ、端子３０ａと端子３０ｄとが接続されてもよい。 Further, for example, in the high-frequency circuit 1, when transmitting a band A transmission signal and a band B signal at the same time, the terminals 30a and 30b may be connected, and the terminals 30a and 30d may be connected.
　また例えば、高周波回路１において、バンドＡの受信信号およびバンドＢの信号を同時に伝送する場合、端子３０ａと端子３０ｃとが接続され、かつ、端子３０ａと端子３０ｄとが接続されてもよい。 Further, for example, in the high-frequency circuit 1, when the received signal of band A and the signal of band B are transmitted simultaneously, the terminals 30a and 30c may be connected, and the terminals 30a and 30d may be connected.
　また例えば、高周波回路１は、さらに、入力端子１１０とフィルタ１１との間に接続された電力増幅器４１と、フィルタ１２と出力端子１２０との間に接続された低雑音増幅器４２と、を備えてもよい。 Further, for example, the high frequency circuit 1 further includes a power amplifier 41 connected between the input terminal 110 and the filter 11, and a low noise amplifier 42 connected between the filter 12 and the output terminal 120. good too.
　また例えば、高周波回路１において、バンドＢはバンドＡよりも低周波側に位置してもよい。 Also, for example, in the high-frequency circuit 1, the band B may be located on the lower frequency side than the band A.
　また、実施の形態に係る通信装置４は、高周波信号を処理するＲＦＩＣ３と、ＲＦＩＣ３とアンテナ２との間で高周波信号を伝送する高周波回路１と、を備える。 Further, the communication device 4 according to the embodiment includes an RFIC 3 that processes high frequency signals, and a high frequency circuit 1 that transmits high frequency signals between the RFIC 3 and the antenna 2 .
　これによれば、高周波回路１の効果を通信装置４で実現することができる。 According to this, the effect of the high-frequency circuit 1 can be realized in the communication device 4.
　（その他の実施の形態）
　以上、本発明に係る高周波回路および通信装置について、実施の形態および変形例を挙げて説明したが、本発明は、上記実施の形態および変形例に限定されるものではない。上記実施の形態および変形例に対して本発明の主旨を逸脱しない範囲で当業者が思いつく各種変形を施して得られる変形例や、本発明に係る高周波回路および通信装置を内蔵した各種機器も本発明に含まれる。 (Other embodiments)
Although the high-frequency circuit and the communication device according to the present invention have been described above with reference to the embodiments and modifications, the present invention is not limited to the above-described embodiments and modifications. Various modifications obtained by those skilled in the art without departing from the scope of the present invention to the above embodiments and modifications, and various devices incorporating the high-frequency circuit and communication device according to the present invention are also included in the present invention. Included in the invention.
　また、例えば、上記実施の形態および変形例に係る高周波回路および通信装置において、各構成要素の間に、インダクタおよびキャパシタなどの整合素子、ならびにスイッチ回路が接続されていてもかまわない。なお、インダクタには、各構成要素間を繋ぐ配線による配線インダクタが含まれてもよい。 Also, for example, in the high-frequency circuits and communication devices according to the above embodiments and modifications, matching elements such as inductors and capacitors, and switch circuits may be connected between the components. Note that the inductor may include a wiring inductor that is a wiring that connects each component.
　本発明は、マルチバンド化およびマルチモード化された周波数規格に適用できる低損失の高周波回路として、携帯電話などの通信機器に広く利用できる。 The present invention can be widely used in communication equipment such as mobile phones as a low-loss high-frequency circuit that can be applied to multi-band and multi-mode frequency standards.
　１、１Ａ、１Ｂ、５００　　高周波回路
　２　　アンテナ
　３　　ＲＦ信号処理回路（ＲＦＩＣ）
　４　　通信装置
　１１、１２、２０、５０２　　フィルタ
　３０　　スイッチ回路
　３０ａ、３０ｂ、３０ｃ、３０ｄ、３０ｅ、３０ｆ　　端子
　４１　　電力増幅器
　４２　　低雑音増幅器
　５０　　基板
　５１　　高音速支持基板
　５２　　低音速膜
　５３　　圧電膜
　５４　　ＩＤＴ電極
　５５、５８　　保護層
　５７　　圧電単結晶基板
　６０　　弾性波共振子
　６０ａ、６０ｂ　　櫛形電極
　６１ａ、６１ｂ　　電極指
　６２ａ、６２ｂ　　バスバー電極
　６５　　支持基板
　６６　　下部電極
　６７　　圧電体層
　６８　　上部電極
　７０　　整合回路
　８０　　キャパシタ
　９０、９１、９２、９３、９４、９５、２５０　　インダクタ
　１００　　アンテナ接続端子
　１０１、１０２、１０３、１０４、１０５、１０６、１０７、２０１、２０２、２０３、２０４、２０５　　直列腕共振子
　１１０　　入力端子
　１１１、１１２、１２１、１２２、１３０　　入出力端子
　１２０　　出力端子
　１５１、１５２、１５３、１５４、１５５、１５６、２５１、２５２、２５３、２５４、２５５　　並列腕共振子
　５４０　　密着層
　５４２　　主電極層 1, 1A, 1B, 500 high frequency circuit 2 antenna 3 RF signal processing circuit (RFIC)
4 communication device 11, 12, 20, 502 filter 30 switch circuit 30a, 30b, 30c, 30d, 30e, 30f terminal 41 power amplifier 42 low noise amplifier 50 substrate 51 high acoustic velocity support substrate 52 low acoustic velocity film 53 piezoelectric film 54 IDT electrode 55, 58 protective layer 57 piezoelectric single crystal substrate 60 elastic wave resonator 60a, 60b comb-shaped electrodes 61a, 61b electrode fingers 62a, 62b busbar electrode 65 support substrate 66 lower electrode 67 piezoelectric layer 68 upper electrode 70 matching circuit 80 capacitor 90, 91, 92, 93, 94, 95, 250 inductor 100 antenna connection terminal 101, 102, 103, 104, 105, 106, 107, 201, 202, 203, 204, 205 series arm resonator 110 input terminal 111, 112, 121, 122, 130 input/output terminal 120 output terminal 151, 152, 153, 154, 155, 156, 251, 252, 253, 254, 255 parallel arm resonator 540 adhesion layer 542 main electrode layer

Claims (15)

  1. 　アンテナ端子、入力端子および出力端子と、
    　時分割複信（ＴＤＤ）用の第１バンドの送信帯域を含む通過帯域を有する第１フィルタと、
    　前記第１バンドの受信帯域を含む通過帯域を有する第２フィルタと、
    　前記第１バンドと異なる第２バンドの少なくとも一部を含む通過帯域を有する第３フィルタと、
    　第１端子、第２端子、第３端子および第４端子を有し、前記第１端子および前記第２端子の接続と前記第１端子および前記第４端子の接続とを同時に実行可能であり、前記第１端子および前記第３端子の接続と前記第１端子および前記第４端子の接続とを同時に実行可能であるスイッチ回路と、を備え、
    　前記アンテナ端子は、前記第１端子に接続され、
    　前記第１フィルタは、前記第２端子と前記入力端子との間に接続され、
    　前記第２フィルタは、前記第３端子と前記出力端子との間に接続され、
    　前記第３フィルタは、前記第４端子に接続され、
    　前記第１フィルタおよび前記第２フィルタの一方は、
    　前記スイッチ回路と前記入力端子および前記出力端子の一方とを結ぶ直列腕経路と、グランドとの間に接続された第１弾性波共振子と、
    　前記直列腕経路とグランドとの間で前記第１弾性波共振子と直列接続された第１インダクタと、を備え、
    　前記第１弾性波共振子は、前記直列腕経路とグランドとの間に接続された並列腕共振子のうちで前記スイッチ回路に最も近い位置で接続されている、
    　高周波回路。     an antenna terminal, an input terminal and an output terminal;
    a first filter having a passband including a transmission band of a first band for time division duplexing (TDD);
    a second filter having a passband including the reception band of the first band;
    a third filter having a passband that includes at least a portion of a second band different from the first band;
    having a first terminal, a second terminal, a third terminal and a fourth terminal, capable of simultaneously connecting the first terminal and the second terminal and connecting the first terminal and the fourth terminal; a switch circuit capable of simultaneously connecting the first terminal and the third terminal and connecting the first terminal and the fourth terminal;
    The antenna terminal is connected to the first terminal,
    the first filter is connected between the second terminal and the input terminal;
    the second filter is connected between the third terminal and the output terminal;
    The third filter is connected to the fourth terminal,
    one of the first filter and the second filter,
    a first acoustic wave resonator connected between a series arm path connecting the switch circuit and one of the input terminal and the output terminal and a ground;
    a first inductor connected in series with the first acoustic wave resonator between the series arm path and ground;
    The first acoustic wave resonator is connected at a position closest to the switch circuit among the parallel arm resonators connected between the series arm path and the ground,
    high frequency circuit.
  2. 　前記第１フィルタおよび前記第２フィルタの前記一方は、前記第２フィルタであり、
    　前記第１弾性波共振子は、前記第３端子と前記出力端子とを結ぶ直列腕経路と、グランドとの間に接続されている、
    　請求項１に記載の高周波回路。     the one of the first filter and the second filter is the second filter;
    The first acoustic wave resonator is connected between a series arm path connecting the third terminal and the output terminal and a ground,
    A high-frequency circuit according to claim 1.
  3. 　前記第１フィルタおよび前記第２フィルタの他方は、前記第１インダクタを備えない、
    　請求項１または２に記載の高周波回路。     the other of the first filter and the second filter does not include the first inductor;
    3. The high-frequency circuit according to claim 1 or 2.
  4. 　さらに、
    　前記アンテナ端子と前記第１端子とを結ぶアンテナ接続経路とグランドとの間に接続された第２インダクタを備える、
    　請求項１～３のいずれか１項に記載の高周波回路。     moreover,
    A second inductor connected between an antenna connection path connecting the antenna terminal and the first terminal and a ground,
    A high-frequency circuit according to any one of claims 1 to 3.
  5. 　さらに、
    　前記第２端子と前記第１フィルタとを結ぶ第１直列腕経路とグランドとの間に接続された第３インダクタと、
    　前記第３端子と前記第２フィルタとを結ぶ第２直列腕経路とグランドとの間に接続された第４インダクタと、
    　前記第４端子と前記第３フィルタとを結ぶ第３直列腕経路とグランドとの間に接続された第５インダクタと、
    　前記アンテナ端子と前記第１端子との間に接続された整合回路と、を備える、
    　請求項１～３のいずれか１項に記載の高周波回路。     moreover,
    a third inductor connected between a ground and a first series arm path connecting the second terminal and the first filter;
    a fourth inductor connected between a ground and a second series arm path connecting the third terminal and the second filter;
    a fifth inductor connected between a ground and a third series arm path connecting the fourth terminal and the third filter;
    a matching circuit connected between the antenna terminal and the first terminal;
    A high-frequency circuit according to any one of claims 1 to 3.
  6. 　前記整合回路は、
    　前記アンテナ端子と前記第１端子とを結ぶアンテナ接続経路に直列配置された第６インダクタと、
    　前記アンテナ端子と前記第１端子とを結ぶアンテナ接続経路とグランドとの間に接続されたキャパシタと、を備える、
    　請求項５に記載の高周波回路。     The matching circuit is
    a sixth inductor arranged in series in an antenna connection path connecting the antenna terminal and the first terminal;
    a capacitor connected between an antenna connection path connecting the antenna terminal and the first terminal and a ground;
    The high frequency circuit according to claim 5.
  7. 　さらに、
    　前記第２端子と前記第１フィルタとを結ぶ第１直列腕経路に直列配置された第７インダクタと、
    　前記第２端子と前記第７インダクタとを結ぶ第１直列腕経路とグランドとの間に接続された第８インダクタと、
    　前記第３端子と前記第２フィルタとを結ぶ第２直列腕経路とグランドとの間に接続された第９インダクタと、を備える、
    　請求項１～３のいずれか１項に記載の高周波回路。     moreover,
    a seventh inductor arranged in series in a first series arm path connecting the second terminal and the first filter;
    an eighth inductor connected between a ground and a first series arm path connecting the second terminal and the seventh inductor;
    a ninth inductor connected between a second series arm path connecting the third terminal and the second filter and a ground;
    A high-frequency circuit according to any one of claims 1 to 3.
  8. 　前記第１インダクタは、前記第１弾性波共振子とグランドとの間に接続されている、
    　請求項１～７のいずれか１項に記載の高周波回路。     The first inductor is connected between the first acoustic wave resonator and ground,
    A high-frequency circuit according to any one of claims 1 to 7.
  9. 　前記第１フィルタは、前記入力端子と前記第２端子とを結ぶ直列腕経路に配置された１以上の直列腕共振子と、
    　前記直列腕経路とグランドとの間に接続された１以上の並列腕共振子と、を備える、
    　請求項１～８のいずれか１項に記載の高周波回路。     the first filter includes one or more series arm resonators arranged in a series arm path connecting the input terminal and the second terminal;
    one or more parallel arm resonators connected between the series arm path and ground;
    A high-frequency circuit according to any one of claims 1 to 8.
  10. 　前記第２フィルタは、前記出力端子と前記第３端子とを結ぶ直列腕経路に配置された１以上の直列腕共振子と、
    　前記直列腕経路とグランドとの間に接続された１以上の並列腕共振子と、を備える、
    　請求項１～９のいずれか１項に記載の高周波回路。     the second filter includes one or more series arm resonators arranged in a series arm path connecting the output terminal and the third terminal;
    one or more parallel arm resonators connected between the series arm path and ground;
    A high-frequency circuit according to any one of claims 1 to 9.
  11. 　前記第１バンドの送信信号および前記第２バンドの信号を同時に伝送する場合、前記第１端子と前記第２端子とが接続され、かつ、前記第１端子と前記第４端子とが接続される、
    　請求項１～１０のいずれか１項に記載の高周波回路。     When transmitting the transmission signal of the first band and the signal of the second band at the same time, the first terminal and the second terminal are connected, and the first terminal and the fourth terminal are connected. ,
    A high-frequency circuit according to any one of claims 1 to 10.
  12. 　前記第１バンドの受信信号および前記第２バンドの信号を同時に伝送する場合、前記第１端子と前記第３端子とが接続され、かつ、前記第１端子と前記第４端子とが接続される、
    　請求項１～１０のいずれか１項に記載の高周波回路。     When simultaneously transmitting the received signal of the first band and the signal of the second band, the first terminal and the third terminal are connected, and the first terminal and the fourth terminal are connected. ,
    A high-frequency circuit according to any one of claims 1 to 10.
  13. 　さらに、
    　前記入力端子と前記第１フィルタとの間に接続された電力増幅器と、
    　前記第２フィルタと前記出力端子との間に接続された低雑音増幅器と、を備える、
    　請求項１～１２のいずれか１項に記載の高周波回路。     moreover,
    a power amplifier connected between the input terminal and the first filter;
    a low noise amplifier connected between the second filter and the output terminal;
    A high-frequency circuit according to any one of claims 1 to 12.
  14. 　前記第２バンドは、前記第１バンドよりも低周波側に位置する、
    　請求項１～１３のいずれか１項に記載の高周波回路。     The second band is located on the lower frequency side than the first band,
    A high-frequency circuit according to any one of claims 1 to 13.
  15. 　高周波信号を処理する信号処理回路と、
    　前記信号処理回路とアンテナとの間で前記高周波信号を伝送する、請求項１～１４のいずれか１項に記載の高周波回路と、を備える、
    　通信装置。     a signal processing circuit that processes high frequency signals;
    The high-frequency circuit according to any one of claims 1 to 14, which transmits the high-frequency signal between the signal processing circuit and the antenna,
    Communication device.
PCT/JP2022/047399 2022-02-01 2022-12-22 High frequency circuit and communication device WO2023149119A1 (en)

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Citations (5)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US20140295775A1 (en) * 2013-04-02 2014-10-02 Broadcom Corporation Switch arrangement
JP2014207517A (en) * 2013-04-11 2014-10-30 太陽誘電株式会社 High-frequency circuit module
WO2015041125A1 (en) * 2013-09-17 2015-03-26 株式会社村田製作所 High frequency module and communication device
WO2018061950A1 (en) * 2016-09-29 2018-04-05 株式会社村田製作所 Acoustic wave filter device, multiplexer, high-frequency front end circuit, and communication device
WO2020153285A1 (en) * 2019-01-23 2020-07-30 株式会社村田製作所 High-frequency front end circuit and communication device

Patent Citations (5)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US20140295775A1 (en) * 2013-04-02 2014-10-02 Broadcom Corporation Switch arrangement
JP2014207517A (en) * 2013-04-11 2014-10-30 太陽誘電株式会社 High-frequency circuit module
WO2015041125A1 (en) * 2013-09-17 2015-03-26 株式会社村田製作所 High frequency module and communication device
WO2018061950A1 (en) * 2016-09-29 2018-04-05 株式会社村田製作所 Acoustic wave filter device, multiplexer, high-frequency front end circuit, and communication device
WO2020153285A1 (en) * 2019-01-23 2020-07-30 株式会社村田製作所 High-frequency front end circuit and communication device

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