WO2023086401A1 - Commande de courant pour un convertisseur élévateur à double inducteur à enroulements opposés - Google Patents

Commande de courant pour un convertisseur élévateur à double inducteur à enroulements opposés Download PDF

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Publication number
WO2023086401A1
WO2023086401A1 PCT/US2022/049428 US2022049428W WO2023086401A1 WO 2023086401 A1 WO2023086401 A1 WO 2023086401A1 US 2022049428 W US2022049428 W US 2022049428W WO 2023086401 A1 WO2023086401 A1 WO 2023086401A1
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WIPO (PCT)
Prior art keywords
electrical current
winding
current
dual anti
wound inductor
Prior art date
Application number
PCT/US2022/049428
Other languages
English (en)
Inventor
Jason W. LAWRENCE
John L. Melanson
Eric J. King
Original Assignee
Cirrus Logic International Semiconductor Ltd.
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Priority claimed from US17/525,463 external-priority patent/US11695337B2/en
Application filed by Cirrus Logic International Semiconductor Ltd. filed Critical Cirrus Logic International Semiconductor Ltd.
Priority to GBGB2403548.7A priority Critical patent/GB202403548D0/en
Priority to CN202280075307.0A priority patent/CN118251829A/zh
Publication of WO2023086401A1 publication Critical patent/WO2023086401A1/fr

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Classifications

    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0064—Magnetic structures combining different functions, e.g. storage, filtering or transformation
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0003—Details of control, feedback or regulation circuits
    • H02M1/0009—Devices or circuits for detecting current in a converter
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0067—Converter structures employing plural converter units, other than for parallel operation of the units on a single load
    • H02M1/007—Plural converter units in cascade
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/40—Means for preventing magnetic saturation
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
    • H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
    • H02M3/10—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M3/145—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M3/155—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/156—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
    • H02M3/158—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load
    • H02M3/1584—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load with a plurality of power processing stages connected in parallel

Definitions

  • the present disclosure relates in general to circuits for audio devices, piezoelectric devices, haptic-feedback devices, and/or other devices, including without limitation personal audio devices such as wireless telephones and media players, and more specifically, to an augmented multi-stage boost converter that may be used in such devices.
  • Personal audio devices including wireless telephones, such as mobile/cellular telephones, cordless telephones, mp3 players, and other consumer audio devices, are in widespread use.
  • Such personal audio devices may include circuitry for driving a pair of headphones, one or more speakers, a piezoelectric transducer, a haptic feedback transducer, and/or other transducer.
  • Such circuitry often includes a driver including a power amplifier for driving a transducer output signal to the transducer.
  • a power converter may be used to provide a supply voltage to a power amplifier in order to amplify a signal driven to speakers, headphones, piezoelectric transducers, haptic feedback transducers, or other transducers.
  • a switching power converter is a type of electronic circuit that converts a source of power from one direct current (DC) voltage level to another DC voltage level.
  • switching DC-DC converters include but are not limited to a boost converter, a buck converter, a buck-boost converter, an inverting buck-boost converter, and other types of switching DC-DC converters.
  • a DC voltage such as that provided by a battery may be converted to another DC voltage used to power the power amplifier.
  • Battery-powered systems may use a boost converter to generate a power supply for an audio amplifier that is greater than a voltage of the battery.
  • a motivation for using a boost converter in a battery-powered transducer is to generate a greater signal swing at the output of a transducer amplifier than could be achieved by powering the amplifier directly from the battery.
  • boost converters often require a boost inductor external to the integrated circuit, which requires significant space.
  • inductors with a magnetic core into an integrated circuit die.
  • Advantages of an integrated inductor may include smaller total circuit area, significant reduction in height in a direction perpendicular to a surface of the integrated circuit, lower electromagnetic interference emissions, and less variation of inductor physical properties.
  • boost converter inductors for audio applications generally have inductances between 1 pH and 2 pH and saturate at between 2.5A and 4A of current.
  • a typical integrated inductor may have an inductance in the range of tens to hundreds of nanohenries with a current saturation limit at or less than 1 A.
  • a typical boost converter for audio may supply 12V into a 10W load from a 4V battery supply.
  • a standard boost converter design may draw 2.5A input current, which is well beyond the saturation point of the integrated inductor.
  • a multi-phase converter could be used to distribute the current to multiple inductors, but the small inductance causes a large current ripple that may still exceed the saturation constraint.
  • FIGURE 1A depicts a multi- wound inductor 100 with two coils 102a and 102b wrapped around a common magnetic core 104.
  • FIGURE IB depicts a cross- sectional side view of inductor 100 depicting current flow in each of coils 102a and 102b, with depicting a current I 1 flowing out of the page in a direction perpendicular to the plane of the page and with “X” depicting a current H flowing into the page in a direction perpendicular to the plane of the page.
  • Coils 102a and 102b may be wound in opposite directions such that positive current generates opposite fields in each coil.
  • a total magnetic flux ⁇ M through magnetic core 104 may equal the difference between the magnetic flux ⁇ M 1 from coil 102a and the magnetic flux ⁇ M 2 from coil 102b.
  • Magnetic fluxes ⁇ M1, ⁇ M2, in coil 102a, 102b may be proportional to currents I 1 and I2, respectively, in such coil 102a, 102b.
  • Inductor 100 may saturate when the magnetic field in magnetic core 104 exceeds a threshold, B sat .
  • the magnetic field may be proportional to the total magnetic flux ⁇ M in magnetic core 104, which may therefore be proportional to the difference in currents (e.g., I1-I2).
  • a saturation constraint for inductor 100 may be given as: where is a difference between current I 1 and current I2 that saturates inductor 100 and may typically be around 0.5 A - 1.0 A for an integrated inductor. Equation (1) above may only be valid for low to moderate levels of current.
  • FIGURE 1C illustrates a saturation profile of current I2 versus current I 1 .
  • the unsaturated region 112 is a strip along the main diagonal as described by equation (1).
  • the unsaturated region 112 shrinks in width until, at very large currents, inductor 100 is always saturated. This effect may occur because the field cancellation between coils 102a and 102b may not be perfect, especially at their respective ends.
  • some inductor designs may use extra turns of one of coils 102a, 102b to control a coupling coefficient which may further reduce the field cancellation.
  • inductor 100 may saturate even though the current difference is within its limits.
  • the condition of equation (1) may represent a necessary (but not a sufficient) condition for saturation. Instead, a sufficient condition for inductor 100 to be unsaturated is that currents I 1 and I2 must lie in unsaturated region 112 defined by points ABCDE.
  • a multi- wound inductor may extend the range of winding currents that may be used before the device is saturated. For example, if current I2 is zero, current I 1 may only extend to point E in FIGURE 1C and remain unsaturated. However, with a properly chosen value for current I2, the range of current I 1 can be extended to point D or even point C and remain unsaturated due to the field cancellation of currents I 1 and I2. This range extension can be used to help with the saturation problem of integrated boost inductors. However, the boost architecture must also be designed to take advantage of the benefits of a multi-wound inductor.
  • FIGURE 2 depicts one example of a single-stage boost converter 200 that may be used with a multi-wound inductor 100 and having a load 202.
  • Single-stage boost converter 200 may use capacitor 204 to stabilize its output voltage V out .
  • a battery 206 may supply single-stage boost converter 200 with an input voltage V in .
  • Single-stage boost converter 200 may comprise a plurality of switches 210, 212, 214, and 216, each switch having a gate G to receive a control signal to control the conductivity of such switch (e.g., to selectively open and close such switch).
  • Such control signals may comprise pulse-width modulation control signals labeled P 1 and P 2 in FIGURE 2, along with their respectively logical complements, signals labeled and in FIGURE 2.
  • Switches 210 and 212 may toggle top coil 102a of inductor 100 between a charging state in which coil 102a is coupled between battery 206 and ground and a transfer state wherein coil 102a is coupled between power supply 206 and load 202.
  • switches 214 and 216 may toggle bottom coil 102b of inductor 100 between a charging state in which coil 102b is coupled between battery 206 and ground and a transfer state wherein coil 102b is coupled between power supply 206 and load 202.
  • the boost voltage ratio, Vout/Vin may be related to the pulse- width modulation duty cycle D of control signals P 1 and P 2 with an equation that is very similar to that of a standard boost converter: assuming no inductor or switching losses.
  • FIGURE 3A depicts a circuit simulation of currents I 1 and I2 for single-stage boost converter 200 over one pulse-width modulation cycle, with an output voltage V out of 12 V, an output power of 10 W, and an input voltage Vm of 4 V, which may represent standard nominal operation conditions for a boost converter in an audio application.
  • the simulation results as depicted in FIGURE 3 A also model resistive losses in switches 210, 212, 214, and 216 and inductor 100.
  • FIGURE 3B depicts current difference I 1 - I 2 and saturation level for inductor 100.
  • FIGURE 3C depicts currents I 2 versus I 1 on a plot along with the saturation boundary l sat also plotted in FIGURE 3C, showing that although current difference I 1 - E remained below saturation level in FIGURE 3B, their individual amplitudes exceeded saturation boundary I sat in FIGURE 3C. Accordingly, single-stage boost converter 200 may not be useful for a desired application.
  • FIGURE 4 depicts one example of a two-stage boost converter 400 that may be used with multi-wound inductor 100 and having a load 202.
  • Each stage 401a, 401b of two- stage boost converter 400 may be identical to single-stage boost converter 200 shown in FIGURE 2, and stages 401a, 401b may be coupled in series.
  • One disadvantage to two- stage boost converter 400 is that it requires two capacitors, 204 and 205, to stabilize the output of each stage 401 compared to the single capacitor 204 required for single-stage converter 200. Both capacitors 204 and 205 may be large and may contribute significantly to the total circuit area.
  • the boosted output of first stage 401a supplies the input voltage to second stage 401b. Therefore, the total boost ratio of both stages 401 is the product of the boost ratio of each stage 401a, 401b. Because both stages 401a, 401b may operate with identical duty cycles, the total boost ratio of two-stage boost converter 400 may be given as: assuming no inductor or switching losses. Comparing equation (3) with equation (2), two- stage boost converter 400 may require a lower duty cycle than single-stage boost converter 200 to achieve the same boost voltage ratio. For example, to boost from 4V to 12V, single- stage boost converter 200 may require a duty cycle of 0.67 versus 0.42 for the two-stage boost converter 400. A lower duty cycle may decrease the magnitude of the current ripple, which should help prevent saturation.
  • FIGURE 5 A depicts a circuit simulation of currents I I-STAGE1 , I 2-STAGE1 , I I-STAGE2 , and I 2-STAGE2 for two-stage boost converter 400 over one pulse-width modulation cycle.
  • FIGURE 5A depicts current difference I I-STAGE1 - I 2-STAGE1 , current difference I I-STAGE2 - I 2-STAGE2 , and saturation level for inductors 100.
  • FIGURE 5C depicts currents I 2-STAGE1 versus I I-STAGE1 and currents I 2-STAGE2 versus I 1 .
  • neither single-stage boost converter 200 nor two-stage boost converter 400 may satisfy the saturation constraints of inductor 100 for desired applications.
  • one or more disadvantages and problems associated with existing inductor-based power converters may be reduced or eliminated.
  • a system may include a power converter comprising at least one stage having a dual anti-wound inductor having a first winding and a second winding constructed such that its windings generate opposing magnetic fields in its magnetic core and constructed such that a coupling coefficient between the first winding and the second winding is less than approximately 0.95 and a current control subsystem for controlling an electrical current through the dual anti-wound inductor, the current control subsystem configured to minimize a magnitude of a magnetizing electrical current of the dual anti-wound inductor to prevent core saturation of the dual anti- wound inductor and regulate an amount of output electrical current delivered by the power converter to the load in accordance with a reference input signal.
  • a method may be provided for controlling an electrical current through a dual anti-wound inductor integral to a power converter, the dual anti-wound inductor having a first winding and a second winding constructed such that its windings generate opposing magnetic fields in its magnetic core and constructed such that a coupling coefficient between the first winding and the second winding is less than approximately 0.95.
  • the method may include minimizing a magnitude of a magnetizing electrical current of the dual anti-wound inductor to prevent core saturation of the dual anti-wound inductor and regulating an amount of output electrical current delivered by the power converter to the load in accordance with a reference input signal.
  • a system may be provided for controlling an electrical current through a dual anti-wound inductor integral to a power converter, the dual anti-wound inductor having a first winding and a second winding constructed such that its windings generate opposing magnetic fields in its magnetic core and constructed such that a coupling coefficient between the first winding and the second winding is less than approximately 0.95.
  • the system may include an input for receiving a reference input signal and a current control subsystem configured to minimize a magnitude of a magnetizing electrical current of the dual anti-wound inductor to prevent core saturation of the dual anti- wound inductor and regulate an amount of output electrical current delivered by the power converter to the load in accordance with the reference input signal.
  • FIGURES 1A and IB depict a multi-wound integrated inductor, in accordance with embodiments of the present disclosure
  • FIGURE 1C illustrates a saturation profile of currents within the multi-wound integrated inductor shown in FIGURES 1A and IB, in accordance with embodiments of the present disclosure
  • FIGURE 2 illustrates a single-stage boost converter using a multi- wound integrated inductor, in accordance with embodiments of the present disclosure
  • FIGURE 3A depicts a circuit simulation of currents for the multi-wound integrated inductor of the single-stage boost converter shown in FIGURE 2 over one pulse-width modulation cycle, in accordance with embodiments of the present disclosure
  • FIGURE 3B depicts a circuit simulation of a current difference and a current saturation level for the multi-wound integrated inductor of the single-stage boost converter shown in FIGURE 2, in accordance with embodiments of the present disclosure
  • FIGURE 3C illustrates a saturation profile of currents within the multi-wound integrated inductor of the single-stage boost converter shown in FIGURE 2, in accordance with embodiments of the present disclosure
  • FIGURE 4 illustrates a two-stage boost converter with each stage using a multi- wound integrated inductor, in accordance with embodiments of the present disclosure
  • FIGURE 5A depicts a circuit simulation of currents for the multi-wound integrated inductors of the two-stage boost converter shown in FIGURE 4 over one pulse-width modulation cycle, in accordance with embodiments of the present disclosure
  • FIGURE 5B depicts a circuit simulation of a current difference and a current saturation level for the multi-wound integrated inductors of the two-stage boost converter shown in FIGURE 4, in accordance with embodiments of the present disclosure
  • FIGURE 5C illustrates a saturation profile of currents within the multi-wound integrated inductors of the two-stage boost converter shown in FIGURE 4, in accordance with embodiments of the present disclosure
  • FIGURE 6 illustrates selected components of an example personal mobile device, in accordance with embodiments of the present disclosure
  • FIGURE 7 illustrates a block diagram of selected components of an example integrated circuit of a personal mobile device for driving a transducer, in accordance with embodiments of the present disclosure
  • FIGURE 8 illustrates a block and circuit diagram of selected components of an example switched mode amplifier, in accordance with embodiments of the present disclosure
  • FIGURE 9 illustrates selected components of an augmented two-stage boost converter with each stage using a multi-wound integrated inductor, in accordance with embodiments of the present disclosure
  • FIGURES 10A and 10B depict equivalent circuit diagrams showing connectivity of selected components of the augmented two-stage boost converter of FIGURE 9 based on the values of switch control signals for the augmented two-stage boost converter, in accordance with embodiments of the present disclosure
  • FIGURES 11 A- 11C depict a circuit simulation of currents for the multi- wound integrated inductors of the augmented two-stage boost converter shown in FIGURE 9 over one pulse-width modulation cycle, in accordance with embodiments of the present disclosure
  • FIGURE 12A depicts an example model for modeling effects of disturbance in generation of pulse-width modulation control signals, in accordance with embodiments of the present disclosure
  • FIGURE 12B depicts an ideal pulse- width modulated control signal and the ideal pulse-width modulated control signal affected by a disturbance, in accordance with embodiments of the present disclosure
  • FIGURES 13A and 13B depict a simulation of an example step disturbance in the generation of pulse-width modulated control signals using the disturbance model of FIGURE 12A, in accordance with embodiments of the present disclosure
  • FIGURE 14A depicts magnetizing currents of inductors resulting from the response of the augmented two-stage boost converter shown in FIGURE 9 to the step disturbance depicted in FIGURES 13 A and 13B along with a magnetizing current saturation limit, in accordance with embodiments of the present disclosure
  • FIGURE 14B illustrates a saturation profile of currents within the multi-wound integrated inductors of the augmented two-stage boost converter shown in FIGURE 9 during a cycle of operation prior to a disturbance in generation of a pulse-width modulated control signal, in accordance with embodiments of the present disclosure;
  • FIGURE 14C illustrates a saturation profile of currents within the multi-wound integrated inductors of the augmented two-stage boost converter shown in FIGURE 9 during a cycle of operation after a disturbance in generation of a pulse-width modulated control signal, in accordance with embodiments of the present disclosure
  • FIGURE 15 illustrates an example voltage control loop that may be used in connection with a boost converter, in accordance with embodiments of the present disclosure
  • FIGURE 16A illustrates selected components of an example current control scheme that may be used in connection with a boost converter, in accordance with embodiments of the present disclosure
  • FIGURE 16B illustrates selected components of an example control subsystem that may be used in connection with a boost converter and the current control scheme depicted in FIGURE 16A, in accordance with embodiments of the present disclosure
  • FIGURE 17 A illustrates selected components, including a cycle average calculator, for performing calculation of cycle averages of inductor coil currents, in accordance with embodiments of the present disclosure
  • FIGURE 17B illustrates selected components of a system for using a triangle carrier signal of a triangle modulator to generate pulse-width modulated control signals and using such triangle carrier signal to trigger a midpoint sampler configured to sample current values of the midpoint of the “ON” time of a pulse- width modulated control signals, in accordance with embodiments of the present disclosure
  • FIGURE 17C illustrates a selected portion of a boost converter and selected components of a midpoint sampler, in accordance with embodiments of the present disclosure
  • FIGURES 18A and 18B depict a simulation of an example step disturbance in the generation of pulse- width modulated control signals, in accordance with embodiments of the present disclosure
  • FIGURE 18C depicts example waveforms for cycle average currents of inductor currents in response to the step disturbance of FIGURE 18 A, in accordance with embodiments of the present disclosure
  • FIGURES 19A-19C depict various waveforms for a three-cycle simulation of coordination of a midpoint sampler with a triangle modulator and the sampling process of FIGURE 17C, in accordance with embodiments of the present disclosure
  • FIGURES 20A and 20B depict waveforms showing an effect of offset caused by resistive losses and imbalance between boost converter stages, in accordance with embodiments of the present disclosure
  • FIGURES 21A and 21B illustrate selected components, including a forward transform block and a reverse transform block, for coordinate transformation, in accordance with embodiments of the present disclosure
  • FIGURE 21C illustrates a saturation profile of currents within a multi-wound integrated inductor depicting transformed coordinate axes, in accordance with embodiments of the present disclosure
  • FIGURES 22A-22D depict a simulation of an example step disturbance in the generation of pulse-width modulated control signals and an application of a forward transform to inductor coil currents, in accordance with embodiments of the present disclosure
  • FIGURES 23A-23C depict selected components of an observer for use in a control subsystem for a boost converter, in accordance with embodiments of the present disclosure
  • FIGURES 24A-24D depict simulated actual values of pulse-width modulated control signal disturbances and coil currents and estimated versions of such parameters as estimated by the Kalman filter shown in FIGURE 23B, in accordance with embodiments of the present disclosure
  • FIGURE 25A illustrates selected components of a control subsystem for a boost converter implementing two independent control loops for magnetizing and battery modes of the control subsystem, in accordance with embodiments of the present disclosure.
  • FIGURE 25B depicts an example implementation of a state-space model for a control block of the control subsystem shown in FIGURE 25A, in accordance with embodiments of the present disclosure.
  • FIGURE 6 illustrates an example personal mobile device 1, in accordance with embodiments of the present disclosure.
  • FIGURE 6 depicts personal mobile device 1 having a speaker 7.
  • Speaker 7 is merely an example, and it is understood that personal mobile device 1 may be used in connection with a variety of transducers including magnetic coil loudspeakers, piezo speakers, haptic feedback transducers, and others.
  • personal mobile device 1 may be coupled to a headset 3 in the form of a pair of earbud speakers 8 A and 8B.
  • Headset 3 depicted in FIGURE 6 is merely an example, and it is understood that personal mobile device 1 may be used in connection with a variety of audio transducers, including without limitation, headphones, earbuds, in-ear earphones, and external speakers.
  • a plug 4 may provide for connection of headset 3 to an electrical terminal of personal mobile device 1.
  • Personal mobile device 1 may provide a display to a user and receive user input using a touch screen 2, or alternatively, a standard liquid crystal display (LCD) may be combined with various buttons, sliders, and/or dials disposed on the face and/or sides of personal mobile device 1.
  • personal mobile device 1 may include an integrated circuit (IC) 9 for generating an analog signal for transmission to speaker 7, headset 3, and/or another transducer.
  • IC integrated circuit
  • FIGURE 7 illustrates a block diagram of selected components of an example IC 9 of a personal mobile device for driving a transducer, in accordance with embodiments of the present disclosure.
  • a microcontroller core 18 may supply a digital input signal DIG_IN to a digital-to- analog converter (DAC) 14, which may convert the digital input signal to an analog input signal VIN.
  • DAC 14 may supply analog signal Vi to an amplifier 16 which may amplify or attenuate analog input signal Vi to provide a differential audio output signal Vo, which may operate a speaker, a headphone transducer, a piezoelectric transducer, a haptic feedback transducer, a line level signal output, and/or other suitable output.
  • DAC digital-to- analog converter
  • DAC 14 may be an integral component of amplifier 16.
  • a power supply 10 may provide the power supply rail inputs of amplifier 16.
  • power supply 10 may comprise a switched-mode power converter, as described in greater detail below.
  • FIGURES 6 and 7 contemplate that IC 9 resides in a personal mobile device, systems and methods described herein may also be applied to electrical and electronic systems and devices other than a personal mobile device, including transducer systems for use in a computing device larger than a personal mobile device, an automobile, a building, or other structure.
  • FIGURE 8 illustrates a block and circuit diagram of selected components of an example switched mode amplifier 20, in accordance with embodiments of the present disclosure.
  • switched mode amplifier 20 may implement all or a portion of amplifier 16 described with respect to FIGURE 7.
  • switched mode amplifier 20 may comprise a loop filter 22, a controller 24, and a power converter 26.
  • Loop filter 22 may comprise any system, device, or apparatus configured to receive an input signal (e.g., audio input signal VIN or a derivative thereof) and a feedback signal (e.g., audio output signal Vo, a derivative thereof, or other signal indicative of audio output signal Vo) and based on such input signal and feedback signal, generate a controller input signal to be communicated to controller 24.
  • a controller input signal may comprise a signal indicative of an integrated error between the input signal and the feedback signal.
  • such controller input signal may comprise a signal indicative of a target current signal to be driven as an output current I OUT or a target voltage signal to be driven as an output voltage V o to a load coupled to the output terminals of second control loop 28.
  • Controller 24 may comprise any system, device, or apparatus configured to, based on an input signal (e.g., input signal INPUT), output signal Vo, and/or other characteristics of switched mode amplifier 20, control switching of switches integral to power converter 26 in order to transfer electrical energy from a power supply V SUPPLY to the load of switched-mode amplifier 20 in accordance with the input signal.
  • an input signal e.g., input signal INPUT
  • output signal Vo e.g., output signal Vo
  • V SUPPLY e.g., V SUPPLY
  • Power converter 26 may comprise any system, device, or apparatus configured to receive at its input a voltage V SUPPLY (e.g., provided by power supply 10), and generate at its output output voltage Vo.
  • voltage V SUPPLY may be received via input terminals of power converter 26 including a positive input terminal and a negative input terminal which may be coupled to a ground voltage.
  • power converter 26 may comprise a power inductor and a plurality of switches that are controlled by control signals received from controller 24 in order to convert voltage V SUPPLY to voltage Vo, such that audio output signal V o is a function of the input signal to loop filter 22.
  • FIGURE 9 depicts selected components of an example augmented two-stage boost converter 900 that may be used with multi-wound inductors 100 and having a load 202, in accordance with embodiments of the present disclosure.
  • augmented two-stage boost converter 900 may be used to implement all or a portion of power supply 10 depicted in FIGURE 7.
  • augmented two- stage boost converter 900 may be used to implement all or a portion of power converter 26 depicted in FIGURE 8.
  • Augmented two-stage boost converter 900 shown in FIGURE 9 may be similar in many respects to two-stage boost converter 400 depicted in FIGURE 4, and thus, only differences between augmented two-stage boost converter 900 and two-stage boost converter 400 may be discussed below.
  • first stage 901a of augmented two-stage boost converter 900 may be similar to first stage 401a of two-stage boost converter 400
  • augmented second stage 901b of augmented two-stage boost converter 900 may include additional switches 910, 912, 914, and 916 and capacitor 905 (in lieu of capacitor 405) arranged as shown in FIGURE 9 and controlled by control signals P 2 , and as shown in FIGURE 9.
  • inductors 100a, 100b of each of stages 901a and 901b are dual, anti- wound inductors comprising a plurality of coils including coils 102a and 102b and wound in such a manner that a magnetic field in a core 104 produced by coils 102a and 102b cancels when currents through coils 102a and 102b are positive.
  • FIGURES 10A and 10B depict equivalent circuit diagrams showing connectivity of selected components of augmented two-stage boost converter 900 based on the values of switch control signals for augmented two-stage boost converter 900, in accordance with embodiments of the present disclosure.
  • FIGURE 10A depicts connectivity of top coils 102a of each of inductors 100a and 100b when control signal P 1 is asserted (and control signal is deasserted)
  • FIGURE 10B depicts connectivity of top coils 102a of each of inductors 100a and 100b when control signal P 1 is deasserted (and control signal is asserted).
  • FIGURES 10A and 10B neglect all resistive switch losses.
  • top coils 102a of inductors 100a and 100b are in parallel to the power supply of battery 206 and ground.
  • the configuration shown in FIGURE 10A is a charging phase of augmented two-stage boost converter 900 in which energy is stored in top coils 102a.
  • FIGURE 10B when control signal P 1 is deasserted (and control signal is asserted), top coils 102a of inductors 100a and 100b are in series to the power supply of battery 206 and ground.
  • the configuration shown in FIGURE 10B is a transfer phase of augmented two-stage boost converter 900 in which energy is transferred from top coils 102a to capacitor 204 and load 202.
  • the unique behavior of charging coils 102a from the two stages in parallel and transferring stored energy from coils 102a in series may be an advantage of this architecture.
  • the bottom coils 102b of inductors 100a and 100b may be controlled in a similar manner.
  • Equation 4 shows that the boost action of each stage 901a, 901b combines additively, in contrast to two-stage boost converter 400 in which the boost action of each stage 401a, 401b combines multiplicatively.
  • augmented two- stage boost converter 900 may require a smaller duty cycle than single-stage boost converter 200 in order to achieve the same boost ratio (though to a lesser extent than two- stage converter 400) which may minimize current ripple.
  • control signal P 1 of augmented two-stage boost converter 900 transitions from asserted to deasserted (and control signal transitions from deasserted to asserted)
  • currents I I-STAGE1 and I I-STAGE2 in coils 102a may not be exactly equal.
  • This unequal current may occur because when control signal P 1 is asserted, the conduction path resistance for coils 102a of inductors 100a and 100b may be different (e.g., inductor 100b may have an extra switch in its conduction path when control signal P 1 is asserted that can add switch resistance).
  • inductor 100b may have an extra switch in its conduction path when control signal P 1 is asserted that can add switch resistance.
  • switch 912 may couple capacitor 905 between the common electrical node of coils 102a (when control signal P 1 is deasserted) and ground, providing an alternative path to any such excess current.
  • switch 916 may be used for a similar purpose for which switch 912 is used.
  • capacitor 905 may be much smaller with minimal impact to total circuit area. In fact, in some instances such capacitor could have a sufficiently small capacitance that capacitor 905 may be formed within the integrated circuit of augmented two-stage power converter 900.
  • a natural consequence of the architecture of augmented two-stage power converter 900 is that capacitor 905 may balance current between first stage 901a and augmented second stage 901b.
  • FIGURES 11 A- 11C depict a circuit simulation of currents for the multi- wound integrated inductors of the augmented two-stage boost converter 900 shown in FIGURE 9 over one pulse-width modulation cycle, in accordance with embodiments of the present disclosure.
  • FIGURE 11A depicts example control signals P 1 and P 2 and
  • FIGURE 11B depicts currents of coils 102a and 102b of inductors 100a and 100b.
  • control signal P 1 is asserted and control signal P 2 is deasserted or control signal P 1 is deasserted and control signal P 2 is asserted
  • at least of a pair of coils 102 is coupled in series to load 202 as shown in FIGURE 10B.
  • energy may be transferred from the magnetic fields of inductors 100A and 100B at the same time energy may be simultaneously stored in the magnetic field.
  • energy may be transferred from one coil 102 of an inductor 100 to the other coil 102 of the inductor.
  • Such transformer action may keep excessive energy from building up in magnetic core 104, thereby potentially preventing early saturation.
  • Augmented two-stage power converter 900 may prevent current saturation because it may minimize the total magnetic field in magnetic core 104, thereby minimizing the amount of magnetic energy stored in magnetic core 104.
  • the total magnetic field in magnetic core 104 may be proportional to magnetization current, I mag , which (for each inductor 100) may be defined as:
  • magnetization current I mag When magnetization current I mag is greater than or equal to magnetization current saturation limit magnetic core 104 may saturate.
  • FIGURE 11C depicts magnetizing currents Imagi and I mag2 for inductors 100a and 100b, respectively.
  • control signals P 1 and P 2 are both asserted, currents in coils 102 are both increasing because both are coupled in parallel between power supply and ground, as previously shown in FIGURE 10A.
  • the magnetizing currents Imagi and I mag2 stay relatively flat because the flux generated by each coil 102 is changing at equal rates, and thus the difference remains constant. This flatness of currents may, in effect, create a “flat- top” to the magnetizing current waveforms as shown in FIGURE 11C that prevents the magnetizing currents I mag1 and I mag2 from saturating.
  • the advantage of the two-stage, augmented boost architecture depicted in FIGURE 9 is that it may reduce peak currents compared to the single-stage architecture depicted in FIGURE 2 and the two- stage architecture depicted in FIGURE 4 and may minimize energy stored in the core of inductors 100, thus minimizing a likelihood of core saturation. Because of these advantages, the current control methodology described below contemplates use of the two-stage, augmented boost architecture depicted in FIGURE 9. However, the concepts, methods, and systems discussed below could be extended to the single-stage architecture depicted in FIGURE 2, the two-stage architecture depicted in FIGURE 4, or any other boost architectures, including without limitation those disclosed in U.S. Pat. App. Ser. No. 16/692,072 filed November 22, 2019, which is incorporated by reference herein in its entirety.
  • the output of augmented two-stage boost converter 900 may be controlled by the duty cycles of control signals P 1 and P 2 . These duty cycles may be time-varying signals that must be carefully chosen to prevent core saturation due to disturbances and regulate the amount of output current.
  • FIGURES 11A-11C discussed above depict idealized pulse-width modulation and current waveforms during steady-state operation.
  • augmented two- stage boost converter 900 may experience some deviations from these idealized waveforms. If the pulse-width modulated control signals P 1 and P 2 are generated by a digital process, then clock jitter, quantization effects, and mismatch in the digital components may cause the actual pulse-width modulated control signals P 1 and P 2 to vary from what is commanded.
  • non-idealities in analog circuitry of augmented two- stage boost converter 900 such as switching time of transistor switches, gate drive effects (e.g., switch non-overlap time), and thermal effects, may cause additional deviations in the response of augmented two-stage boost converter 900.
  • FIGURE 12A depicts an example model for modeling effects of disturbance in generation of pulse-width modulated control signals P 1 and P 2
  • FIGURE 12B depicts an ideal pulse-width modulated control signal (e.g., control signal P 1 ) and the ideal pulse-width modulated control signal affected by a disturbance, in accordance with embodiments of the present disclosure.
  • PWM pulse- width modulation
  • FIGURE 12A two separate pulse- width modulation (PWM) generators 1202 may use two independent duty cycles D 1 and D 2 to generate control signals P 1 , P 2 , and for augmented two-stage boost converter 900.
  • Each PWM generator 1202 may have an independent disturbance source that causes a perturbation in its generated PWM waveform, as shown in FIGURE 12B. This disturbance may move one or more edges of the PWM waveform, effectively changing its duty cycle.
  • FIGURES 13A and 13B depict a simulation of an example step disturbance using the disturbance model of FIGURE 12A.
  • FIGURE 13B shows a simulation of currents through coils 102a and 102b of inductors 100 responsive to the step disturbance.
  • FIGURE 14 A depicts magnetizing currents I mag1 and I mag2 for inductors 100a and 100b, respectively, resulting from the response of augmented two-stage boost converter 900 to the step disturbance depicted in FIGURES 13A and 13B, along with saturation level in accordance with embodiments of the present disclosure.
  • FIGURE 14B illustrates a saturation profile of currents within the multi-wound integrated inductors of augmented two-stage boost converter 900 during a cycle of operation prior to a disturbance in generation of a pulse-width modulated control signal, in accordance with embodiments of the present disclosure.
  • FIGURE 14C illustrates a saturation profile of currents within the multi-wound integrated inductors of augmented two-stage boost converter 900 during a cycle of operation after a disturbance in generation of a pulse-width modulated control signal, in accordance with embodiments of the present disclosure.
  • currents of inductors 100 may remain within the saturation boundary I sat .
  • one or more currents of inductors 100 may exceed the saturation boundary l sat .
  • a 1% disturbance in control signals P 1; P 2 , and P 2 may be sufficient to cause a failure of augmented two-stage boost converter 900 due to inductor core saturation.
  • Larger disturbances in either or both of control signals P 1 , P 2 may cause augmented two-stage boost converter 900 to exceed its limits by an even greater extent.
  • augmented two-stage boost converter 900 may be very sensitive to disturbance generation of control signals P 1 , P 2 , and Such sensitivity may be largely due to the small inductance values of integrated inductors 100.
  • control signals P 15 P 2 and may be controlled to regulate currents and prevent inductor saturation from disturbances.
  • control signals P 1; P 2 may be controlled to deliver power to load 202 of augmented two-stage boost converter 900 and maintain a regulated output boost voltage V out .
  • output boost voltage V out may vary by three times or more to meet the requirements of a class H or class G/H amplifier.
  • Loading on the output of augmented two- stage boost converter 900 may vary over a wide range as well. Audio content may have a high crest factor and may vary rapidly from silence (0 Watts) to full scale ( ⁇ 10 Watts) in tens of microseconds.
  • impedance of load 202 may vary by two times or more across operating frequencies and temperatures. Thus, it may be desirable that augmented two-stage boost converter 900 be capable of responding to rapidly changing output voltage commands and loading conditions.
  • FIGURE 15 illustrates selected components of an example voltage control loop 1500 that may be used in connection with a boost converter 1506 (e.g., which may be implemented with augmented two-stage boost converter 900 or any other suitable boost converter), in accordance with embodiments of the present disclosure.
  • a compensator 1502 may receive a reference voltage V rej which may represent the desired regulated output boost voltage V out , and which may be time-varying.
  • Compensator 1502 may compare reference voltage V ref and sensed output boost voltage V out and generate a control signal CTRL based on the comparison and a control algorithm.
  • control signal CTRL may drive an inner current control loop implemented by a current controller 1504.
  • Such current control loop may regulate an amount of average current delivered to capacitor 204 and load 202, thereby regulating output boost voltage V out .
  • current controller 1504 may need to simultaneously prevent inductor saturation.
  • a boost converter (e.g., augmented two-stage boost converter 900 or other boost converter 1506) using a multi- wound inductor 100 have a current controller 1504 that meets two requirements: (a) current controller 1504 regulates currents in inductors 100 to prevent saturation from disturbances; and (b) current controller 1504 regulates an output current driven to the load of boost converter 1506 to meet the requirements of audio systems with time varying boost voltages and loading.
  • FIGURE 16A illustrates selected components of an example current control scheme that may be used in connection with boost converter 1506, in accordance with embodiments of the present disclosure.
  • current controller 1504 may include a measurement block 1602 configured to receive a sensed inductor current I (e.g., a current / 1 or I 2 of an inductor 100) and perform a calculation on sensed inductor current I to generate a signal that is indicative of inductor coil currents l 1 or I 2 .
  • Current controller 1504 may have two external signals: (a) saturation control signal SATCTRL which may regulate current saturation; and (b) a current control signal ICTRL which controls a target output current delivered to load 202.
  • a control block 1604 may receive an output of measurement block 1602, saturation control signal SATCTRL, and current control signal ICTRL to generate control signals for switches of boost converter 1506.
  • FIGURE 16B illustrates selected components of an example control subsystem 1600 that may be used in connection with boost converter 1506 and the current control scheme depicted in FIGURE 16A, in accordance with embodiments of the present disclosure.
  • compensator 1502 may be used to generate current control signal ICTRL based on a comparison of reference voltage V ref and sensed output boost voltage V out , which may allow compensator 1502 to regulate an amount of current delivered from boost converter 1506 and thus regulate output boost voltage V out .
  • saturation control signal SATCTRL may be permanently set to zero to minimize energy stored in the cores of inductors 100 and prevent saturation due to disturbances.
  • Control block 1604 may receive an output of measurement block 1602 (indicative of measured current inductor coil currents / 1 or / 2 ), saturation control signal SATCTRL, and current control signal ICTRL to generate control signals P 1 . P 2 , and for switches of boost converter 1506.
  • control subsystem 1600 The discussion below outlines several example embodiments, grouped into four sections related to either control or measurement of circuit parameters, for implementing control subsystem 1600.
  • the measurement process of measurement block 1602 may comprise calculating the cycle averages of inductor coil currents l 1 and I 2 of coils 102a and 102b of inductors 100 in a manner that is coordinated or linked to the PWM generation process.
  • FIGURE 17A illustrates selected components, including a cycle average calculator 1704, for performing calculation of cycle averages of inductor coil currents / 1 and I 2 , in accordance with embodiments of the present disclosure.
  • FIGURE 18B shows a simulation of currents through coils 102a and 102b of a dual- wound inductor 100 responsive to the step disturbance.
  • FIGURE 18C depicts a plot of the average winding currents l 1 and I 2 of coils 102a and 102b. Calculating an average of currents l 1 and I 2 may remove the effect of the current ripple within a PWM cycle and may be a much clearer indicator of the current dynamic behavior. The use of average currents may therefore simplify current control by current controller 1504.
  • FIGURES 18B and 18C depict only currents I 1 and I 2 of coils 102a and 102b of first-stage inductor 100a of augmented two-stage power converter 900 and omit currents for second-stage inductor 100b.
  • currents l 1 and I 2 for inductor 100a may be similar to that of 100b, and current controller 1504 may take advantage of this similarity by only measuring currents of one of inductors 100 and controlling all currents based on such measurement.
  • Cycle average calculator 1704 may calculate average cycle values for currents l 1 and I 2 in any suitable manner, including directly by sampling multiple points and summing, implementing a circuit that automatically integrates currents l 1 and I 2 over a cycle, or using a triangle carrier signal of a triangle modulator 1702 to generate pulse-width modulated control signals P 1 , P 2 , and as is known in the art, and use such triangle carrier signal to trigger a midpoint sampler 1706 configured to sample values of currents l 1 and I 2 at the midpoint of the “ON” time of a pulse- width modulated control signals P 1 . P 2 , as depicted in FIGURE 17B.
  • Midpoint sampler 1706 may provide an approximation of cycle average of currents I 1 and I 2 , as described in greater detail below.
  • Midpoint sampler 1706 may sample both currents f and I 2 simultaneously or perform an alternate sampling of currents l 1 and I 2 as shown in Figure 17C.
  • Figure 17C shows a portion of augmented two-stage boost converter 900 in which switches 210 and 214 are controlled by control signals P 1 and P 2 . respectively.
  • midpoint sampler 1706 may be implemented by sense resistor 1708 and analog-to-digital converter (ADC) 1710.
  • Sense resistor 1708 may be coupled between a ground voltage and a common node of switches 210 and 214.
  • ADC 1710 may sample a voltage across sense resistor 1708 which may be indicative of a current flowing to ground.
  • Analog-to-digital converter 1710 may be triggered by the trigger signal from triangle modulator 1702 to sample a midpoint current value.
  • Using a single ADC 1710 and sense resistor 1708 may be advantageous because they make up a smaller circuit and the smaller circuit may avoid possible mismatch if two ADCs and sense resistors were to be used.
  • FIGURES 19A-19C depict various waveforms for a three-cycle simulation of coordination of midpoint sampler 1706 with triangle modulator 1702 and the sampling process of FIGURE 17C, in accordance with embodiments of the present disclosure.
  • FIGURES 19A and 19B depict generation of pulse-width modulated control signals P 1 and P 2 from a triangle carrier wave.
  • FIGURE 19A depicts a triangle carrier wave CARRIER with a minimum of - 1 and a maximum of +1 and a period equal to a pulse-width modulation period of pulse-width modulated control signals P 1 and P 2 .
  • Reference signals refl and re/2 may be related to the desired duty cycles D 1 and D 2 as follows:
  • FIGURE 19B depicts pulse-width modulated control signals P 1 and P 2 which may be derived from reference signals ref1 and ref2 and triangle carrier wave CARRIER as follows:
  • Control signals and may be the logical complements of control signals P 1 and P 2 . respectively.
  • the alternate sampling of currents f and I 2 as shown in Figure 17C may be achieved when triangle carrier wave CARRIER equals -1 and +1, respectively.
  • the times of alternate sampling of currents I 1 and I 2 are depicted in FIGURES 19A-19C by points labeled I 1 samp and I 2 samp, respectively.
  • sampling is configured to occur at the midpoint of the “ON” time of control signals P 1 and P 2 .
  • Such midpoint sampling may be advantageous as it is maximally far away in tome from the edges of control signals P 1 and P 2 where switching transients may distort measurement.
  • midpoint sampling may ensure that current f is sensed when control signal P 1 is asserted and control signal P 2 is deasserted and that current I 2 is sensed when control signal P 2 is asserted and control signal P 1 is deasserted, regardless of the desired duty cycles D 1 and D 2 .
  • FIGURE 19C depicts simulated winding currents f and I 2 assuming lossless inductors and switches, along with the sampled current values. Due to the symmetry of the switching states and the accompanying anti-symmetry of the currents about the midpoint of the control signal “ON” time, the sampled values may represent cycle averages of inductor currents I 1 and I 2 .
  • resistive losses along with imbalance between two stages 901a and 901b of augmented two-stage boost converter 900 may distort current waveforms such that they are no longer piecewise-linear waveforms that are antisymmetric about the midpoint of the “ON” times of control signals P 1 and P 2 .
  • This distortion may cause an offset between the actual cycle-averages of inductor currents / 1 and I 2 and the values obtained from mid-point sampling.
  • FIGURES 20A and 20B show an example of this offset from a simulation with resistive losses and imbalance between stages 901a and 901b.
  • the midpoint-sampled values of inductor currents l 1 and I 2 may still be used in the current control loop described above.
  • the errors are generally small relative to the value being measured.
  • the outer voltage regulation loop may use an integrator to zero out any error in the commanded output current.
  • the estimated winding current difference from midpoint sampling also approaches zero. Therefore, if estimated coil current difference
  • the control subsystem of boost converter 1506 may implement a coordinate transformation to decouple signals in order to simplify control of boost converter 1506.
  • FIGURES 21A and 21B illustrate selected components, including a forward transform block 2102 and an inverse transform block 2104, for coordinate transformation, in accordance with embodiments of the present disclosure.
  • forward transform block 2102 may be applied to measurements for inductor currents / 1 and I 2 and control block 1604 may operate on transformed measurements l m and l b for inductor currents l 1 and I 2 to generate duty cycle control signals D m and D b which may in turn be inverse transformed by inverse transform block 2104 to generate duty cycle control signals D 1 and D 2 used to drive PWM generator 1202.
  • Forward transform block 2102 may apply the following transform to generate transformed current measurements l m and l b : wherein l m may be referred to as a magnetizing current and l b may be referred to as a battery current. Magnetizing current l m may be equal to the difference between inductor currents l 1 and I 2 and may therefore be proportional to the net magnetic field in the core of multi-wound inductor 100.
  • battery current as used herein is not in any way limited to a current sourced from a battery but may be sourced from a battery or any suitable power supply or may be a mathematical equivalent/transformative value representative of a battery current or power supply current.
  • battery current l b may be the sum of coil currents l 1 and I 2 and for a single stage converter, battery current l b may be exactly equal to an actual current flowing from the battery or power supply. For a two-stage converter, battery current l b may no longer be equal to an actual battery or power source current, but may be so termed because it is the same mathematical transformation. Because coil currents / 1 and I 2 may be the cycle average values of the winding currents, magnetizing current l m and battery current l b may also represent the cycle averages of the magnetizing and battery currents.
  • the inverse transform of transform block 2102 may be applied by inverse transform block 2104 to generate duty cycle control signals:
  • FIGURE 2 IB depicts how both transformations are implemented.
  • the saturation control signal SATCTRL and the current control signal ICTRL received by control block 1604 may be replaced by a magnetizing current reference and a battery current reference , respectively. Because of this transformation, control block 1604 may operate in magnetizing/battery coordinate space rather than a coil lOla/coil 102b coordinate space.
  • FIGURE 21C illustrates a saturation profile of currents within a multi-wound integrated inductor 100 depicting transformed coordinate axes, in accordance with embodiments of the present disclosure.
  • the plot of FIGURE 21C depicts a saturation region 110 similar to that of FIGURE 1C on a coordinate axis in which current is on the horizontal axis and current I 2 is on the vertical axis.
  • the transformation of equation (9) may be interpreted as a rotation transformation that rotates the (/ 1; / 2 ) axes by 45° to a new set of axes (I m , l b ) as shown in FIGURE 21C.
  • This rotation transformation may be useful because for most of the unsaturated region 112, the magnetizing current l m coordinate may be a direct measurement of how far away boost converter 1506 is from exceeding the magnetizing saturation limit
  • the orthogonal battery current l b coordinate is free from this constraint and may be representative of a current flowing through the boost converter 1506 to its output.
  • battery current l b may have a maximum current constraint, but the approximation in the transformed coordinate space may be useful for control over most of the operating space.
  • This example embodiment depicts how the transformation may decouple the saturation protection from output current requirements, allowing each quantity to be controlled independently.
  • Augmented two-stage boost converter 900 may be modeled using the example state-space averaging technique described in Erickson, Robert W. and Dragan Maksimociv, “Fundamentals of Power Electronics: Second
  • the system including boost converter 1506 and its control subsystem may be modeled as a second-order ordinary differential equation as follows: where lower case variables i 1; i 2 , d 1 , and d 2 represent the small signal deviations of currents and I 2 and duty cycles and D 2 from steady-state, L is the self-inductance of inductor 100, M is the mutual inductance of inductor 100, R is a resistance which is a function of a switch resistance, inductor resistance, and steady-state duty cycle, and V is a voltage that is a function of the power supply voltage, output boost voltage V out , switch resistance, inductor resistance, and steady-state current.
  • Equation (9) If the transformations of equations of (9) and (10) are applied to equation (11), the result may be: where lower case variables i m , i b , d m , and d b represent the small signal deviations of currents l m and I b and duty cycles D m and D b from steady-state, and k is a coupling coefficient defined by M/L.
  • the matrices are diagonalized, such that the magnetizing and battery modes are orthogonal.
  • the system of boost converter 1506 and its control subsystem may be decoupled into two, independent first-order modes.
  • FIGURES 22A-22D depict the results when the foregoing transformation is applied to the previous example of a 1% step disturbance in duty cycle D 1 .
  • FIGURE 22A depicts duty cycle with a disturbance occurring at 0.2 psec
  • FIGURE 22B depicts the simulated cycle- average of coil currents / 1 and I 2 for inductor 100a. Even though the disturbance occurs only on duty cycle D 1 , both currents l 1 and I 2 may be affected due to the coupled dynamics of equation (11). Transformation equation (9) may be applied to these currents l 1 and I 2 to yield magnetizing current l m in FIGURE 22C and battery current l b in FIGURE 22D.
  • the cycle-average measurements of currents l 1 and I 2 may be noisy. Additionally, if the sampling method from FIGURE 17C is implemented, information from one of the coils 101 may be missing at each sampling period. These shortcomings may degrade the performance of the control subsystem. Accordingly, the control subsystem may implement an observer 2300, as shown in FIGURE 23A. In operation, observer 2300 may receive measured current data l meas and generate an improved estimate of currents l 1 and I 2 . Observer 2300 may use a model of the control subsystem to filter out noise and fill in any missing information. The system model may require knowledge of or information related to the desired duty cycles D 1 and D 2 , as well as an indicator of which current value and I 2 is currently being measured.
  • FIGURE 23A The latter indicator signal is labeled l meas mode in FIGURE 23A.
  • Observer 2300 may be implemented using one of several methodologies including, without limitation: a Luenberger filter, a Kalman filter, and a slide mode observer.
  • FIGURE 23B shows an implementation based on a Kalman filter.
  • the Kalman filter architecture of FIGURE 23B may be used if measured current data l meas comes from the current sampling scheme shown in FIGURE 17C.
  • the Kalman filter implementation of FIGURE 23B may operate by using a model 2302 and the known model inputs of desired duty cycles D 1 and D 2 to form an estimate of currents l 1 and I 2 . These estimates may be compared by a subtractor 2304 with measured current data l meas to form error signal e.
  • Error signal e may be multiplied by Kalman gains, K. and then used to adjust the model estimates such that the error is minimized.
  • the signal l meas mode may indicate whether measured current data l meas is measuring current or current I 2 .
  • the system model may be derived by discretizing the continuous-time model equation (11) using any standard method (e.g., forward Euler, bilinear transform, Zero Order Hold (ZOH), etc.) and re-writing it in standard form:
  • a and B are 2x2 matrices and I is the sample time index.
  • This model may be extended to include the effects of noise and disturbances on generation of control signals P 1 and P 2 : where is the state vector and distl and dist2 are disturbance estimates; is the model input vector; F is a 4x4 block matrix comprising the 2x2 matrix A from equation (13), the 2x2 matrix B from equation (13), the 2x2 zero matrix, and the 2x2 identity matrix; G is a 4x2 block matrix comprising 2x2 matrix B and the 2x2 zero matrix, w is a 4x1 vector of the process noise, y is the output or I 2 ; and r is the scalar measurement noise.
  • the dependency of model 2302 on signal lmeas mode is represented in FIGURE 23B with a dashed line.
  • the Kalman filter implemented by model 2302 may be implemented in any suitable number of ways using equations (14) and (15) (e.g., in accordance with Simon, Dan, “Optimal State Estimation,” Wiley 2006, which is incorporated by reference herein in its entirety).
  • the Kalman recursion may be given by: where is the estimated state vector that contains estimated coil currents and disturbances and z i is an internal state vector (e.g., an a priori state estimate).
  • the Kalman gain, K i may be derived using any of the standard techniques (e.g., techniques disclosed in the Simon reference cited above).
  • the gain computation must account for the H i matrix that changes with time based on measured current data Imeas-
  • the Kalman gains may be calculated ahead of time over several samples. At steady state, the Kalman gains will alternate between two sets of values depending on signal l meas mode. These steady-state values may be stored and applied depending on the state of signal l meas mode , as indicated by the dashed lines in FIGURE 23B.
  • FIGURES 24A-24D depict simulation results of augmented two-stage power converter 900 in response to the 1 % step disturbance earlier described.
  • the Kalman filter of FIGURE 23B may be used to estimate the currents l 1 and I 2 as well as the disturbances on control signals P 1 and P 2 .
  • the simulation shown may use the current sampling technique disclosed in FIGURE 17C.
  • FIGURES 24A and 24B depict actual and estimated disturbances on control signals P 1 and P 2 while FIGURES 24C and 24D depict actual and estimated currents / 1 and I 2 .
  • the currents plotted may be the deviations from steady state.
  • the Kalman filter of FIGURE 23B may rapidly converge on an estimate of both the disturbances of control signals P 1 and P 2 and currents l 1 and I 2 even though the measured value of measured current data l meas only includes one of currents and I 2 on each sample.
  • FIGURE 23C depicts an alternative embodiment for observer 2300.
  • two independent observers each with a difference model 2302a and 2302b and gain Ki and K2 may be used.
  • First observer 2300a may form an estimate of current l 1 when current I 2 is being measured. Its input may be the instant measurement of current l 1 and the previous measurement of current I 2 .
  • second observer 2300b may form an estimate of current I 2 when current is being measured. Its input may be the instant measurement of current I 2 and the previous measurement of current I 1 .
  • models 2302a, 2302b and gains Ki, K2 may be fixed. Estimates for current / 1 , I 2 may be taken from either the output of first model 2302a or second model 2302b depending on the state of signal l meas mode which may control a multiplexed output 2306.
  • currents / 1 and I 2 may be controlled using two independent control loops with control blocks 1604a and 1604b that separately control the magnetizing and battery modes.
  • Control blocks 1604a and 1604b may be implemented using any standard control algorithm including without limitation proportional, proportional-integral, proportional-integral-derivative, or state-space.
  • FIGURE 25B depicts an example of a state-space control 2500 that may be used for either or both of control blocks 1604a and 1604b.
  • State-space control 2500 may implement a servo control architecture that includes an added integrator 2502 to remove steady-state error (e.g., as disclosed in Ogata, Katsuhiko, Discrete-Time Control Systems, Prentice Hall, 1995). Such control may be based on a decoupled state-space model of the system.
  • the decoupled model may be derived by applying the magnetizing/battery transform equations (9) and (10) to the discretized model of equation (13):
  • a m , a b , b m , and b b are scalar coefficients. Applying the magnetizing/battery transforms may diagonalize the matrices and decouple the magnetizing/battery modes as discussed earlier. As a result, independent state-space models may be written for each mode. In a real implementation, there may be a non-zero computational delay time for all the blocks of the algorithm. However, in this case, it is assumed there is a one-sample delay between receiving measured current data I meas and calculating the next PWM command.
  • This one-sample delay may also be included in the model for each mode as follows: where up is an internal state that represents the input during the previous sample.
  • the models given by equations (18) and (19) may each be used to construct a servo control using a method similar to the Ogata reference cited above.
  • the result may be the architecture in FIGURE 25B.
  • the block 2504 labeled “b” may calculate a gain using either the b m or b b coefficient from equation (19) or (20) depending on whether the control is for the magnetizing or battery mode.
  • the gains K 1 , K 2 , and K 3 may be calculated using any standard state-space method (e.g., pole placement, quadratic optimal control, etc.).
  • the systems and methods described herein may only function successfully when coils 102a and 102b of a dual-wound inductor 100 having a coupling coefficient k within a particular range. Specifically, a value of k satisfying 0.70 ⁇ k ⁇ 0.95 may be necessary for successful operation.
  • a value of k satisfying 0.70 ⁇ k ⁇ 0.95 may be necessary for successful operation.
  • Second, for values of k outside this range a closed loop system regulating inductor current is difficult to control, impractical to implement, or unstable. Further, it is impractical or expensive to manufacture embedded inductors with coupling coefficients outside this range.
  • Coupling coefficient k must be above a minimum value for the circuits described above to function properly.
  • coupling was necessary for the anti-wound coils 102a and 102b to cancel the magnetic field in the core and prevent saturation. Specifically, such coupling keeps the difference in winding currents below saturation level as shown in FIGURE 3B and equation (1). If coupling coefficient k is too low, field cancellation will not occur and condition (1) may be violated.
  • multi-wound inductor 100 will have too much field cancellation when the winding currents of the anti- wound inductor are both rising or falling. For example, as shown in FIGURES 11A and 11B, around times 1x10 -8 seconds and 3xl0 -8 seconds, both gate control signals are positive, and all the winding currents are simultaneously rising. As coupling coefficient k approaches 1 , the slope of the winding currents in these regions will approach infinity due to the field cancellation. This cancellation may cause high current ripple that can drive inductor 100 into the saturation region, shown in FIGURE 1C, and potentially damage the part.
  • Equation (12) A linearized model of voltage control loop 1500 was given in Equation (12) using the magnetizing and battery currents as state variables. Because the state variables are decoupled, the equations can be rearranged as two, independent, first-order ordinary differential equations:
  • this extremely large gain may lead to undesirable amplification of noise from the input (e.g., discretization noise) as well as lead to impractical demands for high-resolution control that can be costly. It can also cause large signal swings on the currents that may saturate feedback sensors.
  • Equation (22) may become large and current ib may have a very fast response.
  • Systems with fast dynamics often require fast control circuitry which can be both costly to implement and use large amounts of power, which decreases overall system efficiency.
  • the fast response time of such a plant also means that it may be more sensitive to delays in the control loop, which also makes the control system design more challenging.
  • Coupling between coils 102a and 102b may be achieved by interleaving coils 102a and 102b as shown in FIGURES 1A-1B.
  • Coupling coefficient k may be controlled by adjusting the number of turns of each coil 102a and 102b that are interleaved and the spacing between coils. If coupling coefficient k is low, it means the two coils 102 and 102b may need to be further apart, which may increase the size and cost of a device.
  • a coupling coefficient k that is too large is also impractical to manufacture. Even with all the turns of both coils 102a and 102b fully interleaved, coupling coefficient k will reach a maximum value less than 1. Finite element simulations have shown such maximum value to be around 0.95. This maximum value is because only some of the magnetic flux generated by a coil 102a/102b flows through the core and links the adjacent coil 102a/102b. This linking is especially true for the end turns where there is less core material. Further increase in coupling coefficient k may be possible by adding more core material around the windings, but such approach may become very costly and impractical to manufacture. In addition, the wiring to the coil structure, both internal to an integrated circuit chip and external to the chip, may have its own self-inductance. This factor alone may prevent a coupling coefficient k greater than 0.99.
  • references in the appended claims to an apparatus or system or a component of an apparatus or system being adapted to, arranged to, capable of, configured to, enabled to, operable to, or operative to perform a particular function encompasses that apparatus, system, or component, whether or not it or that particular function is activated, turned on, or unlocked, as long as that apparatus, system, or component is so adapted, arranged, capable, configured, enabled, operable, or operative. Accordingly, modifications, additions, or omissions may be made to the systems, apparatuses, and methods described herein without departing from the scope of the disclosure. For example, the components of the systems and apparatuses may be integrated or separated.
  • each refers to each member of a set or each member of a subset of a set.

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Dc-Dc Converters (AREA)

Abstract

Un système peut comporter un convertisseur de puissance comprenant au moins un étage présentant un double inducteur à enroulements opposés présentant un premier enroulement et un second enroulement construit de sorte que ses enroulements génèrent des champs magnétiques opposés dans son noyau magnétique et construit de sorte qu'un coefficient de couplage entre le premier enroulement et le second enroulement est inférieur à environ 0,95 et un sous-système de commande de courant destiné à commander un courant électrique à travers le double inducteur à enroulements opposés, le sous-système de commande de courant étant configuré pour réduire au minimum une amplitude d'un courant électrique de magnétisation du double inducteur à enroulements opposés pour empêcher une saturation de noyau du double inducteur à enroulements opposés et réguler une quantité de courant électrique de sortie distribué par le convertisseur de puissance à la charge en fonction d'un signal d'entrée de référence.
PCT/US2022/049428 2021-11-12 2022-11-09 Commande de courant pour un convertisseur élévateur à double inducteur à enroulements opposés WO2023086401A1 (fr)

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GBGB2403548.7A GB202403548D0 (en) 2021-11-12 2022-11-09 Current control for a boost converter with dual anti-wound inductor
CN202280075307.0A CN118251829A (zh) 2021-11-12 2022-11-09 具有双反绕电感器的升压转换器的电流控制

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Citations (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US20070090915A1 (en) * 2005-10-25 2007-04-26 Jinghai Zhou Multiphase voltage regulator having coupled inductors with reduced winding resistance
US20190149049A1 (en) * 2016-06-10 2019-05-16 Sumitomo Electric Industries, Ltd. Voltage conversion device and method of deciding leakage inductance
US20200204073A1 (en) * 2018-12-21 2020-06-25 Cirrus Logic International Semiconductor Ltd. Current control for a boost converter with dual anti-wound inductor

Patent Citations (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US20070090915A1 (en) * 2005-10-25 2007-04-26 Jinghai Zhou Multiphase voltage regulator having coupled inductors with reduced winding resistance
US20190149049A1 (en) * 2016-06-10 2019-05-16 Sumitomo Electric Industries, Ltd. Voltage conversion device and method of deciding leakage inductance
US20200204073A1 (en) * 2018-12-21 2020-06-25 Cirrus Logic International Semiconductor Ltd. Current control for a boost converter with dual anti-wound inductor

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ERICKSON, ROBERT W.DRAGAN MAKSIMOCIV: "Fundamentals of Power Electronics", 2001, SPRINGER SCIENCE+BUSINESS MEDIA
SIMON, DAN: "Optimal State Estimation", 2006, WILEY

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