WO2022110273A1 - 一种磁场调制式双凸极电机及其凸极齿分布设计方法 - Google Patents

一种磁场调制式双凸极电机及其凸极齿分布设计方法 Download PDF

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Publication number
WO2022110273A1
WO2022110273A1 PCT/CN2020/133829 CN2020133829W WO2022110273A1 WO 2022110273 A1 WO2022110273 A1 WO 2022110273A1 CN 2020133829 W CN2020133829 W CN 2020133829W WO 2022110273 A1 WO2022110273 A1 WO 2022110273A1
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Prior art keywords
stator
teeth
excitation
motor
salient pole
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PCT/CN2020/133829
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English (en)
French (fr)
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赵文祥
蒋婷婷
徐亮
吉敬华
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江苏大学
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Priority to US17/434,429 priority Critical patent/US11489429B2/en
Publication of WO2022110273A1 publication Critical patent/WO2022110273A1/zh

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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02K—DYNAMO-ELECTRIC MACHINES
    • H02K41/00—Propulsion systems in which a rigid body is moved along a path due to dynamo-electric interaction between the body and a magnetic field travelling along the path
    • H02K41/02—Linear motors; Sectional motors
    • H02K41/03—Synchronous motors; Motors moving step by step; Reluctance motors
    • H02K41/031—Synchronous motors; Motors moving step by step; Reluctance motors of the permanent magnet type
    • H02K41/033—Synchronous motors; Motors moving step by step; Reluctance motors of the permanent magnet type with armature and magnets on one member, the other member being a flux distributor
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02K—DYNAMO-ELECTRIC MACHINES
    • H02K1/00—Details of the magnetic circuit
    • H02K1/06—Details of the magnetic circuit characterised by the shape, form or construction
    • H02K1/12—Stationary parts of the magnetic circuit
    • H02K1/14—Stator cores with salient poles
    • H02K1/146—Stator cores with salient poles consisting of a generally annular yoke with salient poles
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02K—DYNAMO-ELECTRIC MACHINES
    • H02K1/00—Details of the magnetic circuit
    • H02K1/06—Details of the magnetic circuit characterised by the shape, form or construction
    • H02K1/22—Rotating parts of the magnetic circuit
    • H02K1/24—Rotor cores with salient poles ; Variable reluctance rotors
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02K—DYNAMO-ELECTRIC MACHINES
    • H02K21/00—Synchronous motors having permanent magnets; Synchronous generators having permanent magnets
    • H02K21/02—Details
    • H02K21/04—Windings on magnets for additional excitation ; Windings and magnets for additional excitation
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02K—DYNAMO-ELECTRIC MACHINES
    • H02K3/00—Details of windings
    • H02K3/46—Fastening of windings on the stator or rotor structure
    • H02K3/52—Fastening salient pole windings or connections thereto
    • H02K3/521—Fastening salient pole windings or connections thereto applicable to stators only
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02K—DYNAMO-ELECTRIC MACHINES
    • H02K2213/00—Specific aspects, not otherwise provided for and not covered by codes H02K2201/00 - H02K2211/00
    • H02K2213/03—Machines characterised by numerical values, ranges, mathematical expressions or similar information

Definitions

  • the invention relates to a magnetic field modulation type double salient pole motor and a method for designing the distribution of salient pole teeth, belonging to the field of motors.
  • the hybrid excitation double salient pole structure design with variable magnetic flux makes the motor have the performance advantages of high thrust and wide speed regulation range; two sets of windings and two sets of permanent magnets are installed on the stator, which can be used in aerospace, Transportation, wind power and other fields. .
  • Chinese Invention Patent Application No. CN201910270245.9 discloses a demagnetization and fault-tolerant double-stator electric excitation double-salient-pole motor and a method thereof.
  • a set of inner stator windings is added through the double-stator structure to make full use of the motor space; the rotor is provided with a spacer The magnetic bridge, the main magnetic flux is closed by the outer stator, the rotor and the inner stator, so that the motor can also provide enough reluctance torque in the event of a demagnetization fault, and realize the fault-tolerant operation of the motor.
  • the motor has fault tolerance, it is subject to the electric excitation motor itself.
  • a permanent magnet is introduced into the electric excitation motor, so that the motor has the high thrust density of the permanent magnet motor and has a wide speed range.
  • Chinese Invention Patent No. CN201010592364.5 discloses two types of axially excited doubly salient pole motors, both of which include at least two axially arranged doubly salient pole monolithic structures.
  • the stator and rotor iron cores form an axial magnetic circuit through the axial magnetic guide back iron; in the second type, the stator iron core, the stator axial magnetic guide back iron and the permanent magnet are all arc-shaped structures.
  • the invention has high space utilization rate, no additional air gap and high excitation efficiency. Although the invention effectively saves space, and the introduction of permanent magnets also improves the thrust density and excitation efficiency of the motor, the complex structural design also brings difficulties to production.
  • Chinese Invention Patent Application No. CN202010097798.1 discloses a doubly salient permanent magnet motor with adjustable effective permanent magnet pole number.
  • the doubly salient permanent magnet can be adjusted.
  • the number of poles of the permanent magnets of the magneto makes the motor's magnetic adjustment range wide and wide-area efficient.
  • the application of pulse current in the excitation winding also increases the difficulty of control circuit design.
  • the risk of irreversible demagnetization of permanent magnets caused by excessive pulse current also needs to be faced. The problem.
  • the purpose of the present invention is to propose a magnetic field modulation type doubly salient motor and its salient tooth distribution design method, aiming at the vacancy in the research on the salient tooth distribution of the existing doubly salient motor.
  • a The distribution design method of the salient pole teeth to find out the distribution mechanism of the salient pole teeth, so as to optimize the amplitude of the working wave and improve the thrust and magnetic adjustment capability of the motor.
  • the motor design is mainly based on the excitation winding and supplemented by permanent magnet materials.
  • a magnetic field modulation type double salient motor is designed.
  • the thrust capability of the motor is enhanced by the design of permanent magnets to enhance the anti-saturation capability of the motor.
  • the motor of the present invention is realized by adopting the following technical solutions: a magnetic field modulated doubly salient motor, comprising a stator (1) and a mover (2), the stator comprising a stator iron core, stator teeth ( 3), armature winding (6), excitation winding (5), permanent magnet 1 (81), permanent magnet 2 (82); each stator tooth (3) is split into two excitation teeth (4), each excitation The tooth (4) is split into two salient pole teeth, salient pole tooth 1 (71) and salient pole tooth 2 (72).
  • the armature winding (6) is wound on the stator tooth (3).
  • the center line of the pivot winding differs by half the stator tooth pitch, and the excitation winding (5) is wound on two adjacent excitation teeth (4) formed by the splitting of different stator teeth; permanent magnet 1 (81), permanent magnet 2 (82) ) are tangentially magnetized and placed in the excitation slot, permanent magnet 1 (81) is placed in the excitation slot, permanent magnet 2 (82) is embedded in the bottom of the excitation slot; The polarities of the magnets are opposite and the polarities of the permanent magnets in the same slot are also opposite; when the permanent magnets are excited alone, the two permanent magnets in the same slot form a closed magnetic circuit in the stator teeth; the mover is composed of multiple rotors
  • the salient poles (9) are arranged in an arrangement, and after bilateral modulation, the air-gap magnetic field contains abundant working waves.
  • tooth widths of the salient pole teeth 1 (71) and 2 (72) formed by the splitting of the same excitation tooth (4) are independent of each other and do not affect each other; the distribution of the stator salient pole teeth is optimized, and each working wave is adjusted to increase the thrust of the motor.
  • the number of teeth of the stator teeth (3) is N s
  • the widths of the salient pole teeth 1 (71) and 2 (72) are ⁇ 1 and ⁇ 2 respectively
  • the armature winding (6) and the excitation winding (5) are set
  • the slot widths are ⁇ 3 and ⁇ 5 respectively
  • the width of the excitation groove formed by the splitting of a single excitation tooth (4) is ⁇ 4
  • the number of rotor salient poles (9) is N r
  • the pole pitch of each tooth is ⁇ p
  • the relationship of each parameter can be expressed as:
  • the technical solution of the design method of the present invention includes that the total number of stator salient pole teeth is 4N s , wherein the salient pole teeth 1 (71) and the salient pole teeth 2 (72) are both 2N s , and the distribution of the salient pole teeth is the same as ⁇ 1 , ⁇ 2 , ⁇ 3 , ⁇ 4 and ⁇ 5 are related.
  • the design methods of the salient pole teeth distribution are mainly: summarizing the distribution law of the salient pole teeth; deriving the order and amplitude of the no-load air gap magnetic density in each case ; Then solve the back EMF through the magnetic density, and by comparing the size of the back EMF, the optimal way of salient tooth distribution is obtained; the details are as follows:
  • Step 1 take ⁇ 1 , ⁇ 2 , ⁇ 3 , ⁇ 4 and ⁇ 5 as the entry points, design the distribution of salient pole teeth, based on a single stator tooth, the distribution of salient pole teeth can be summarized into the following three models Model :
  • each stator tooth has a minimum unit
  • each stator tooth has two minimum units;
  • each stator tooth has four minimum units;
  • step 2 permanent magnet 1 (81) and permanent magnet 2 (82) only form a closed magnetic circuit in the stator core, and the air-gap magnetic density is generated by the excitation winding, so only the excitation winding needs to be analyzed, and the air-gap magnetic field of the excitation winding is The momentum is:
  • F fw is the magnetomotive force of the excitation winding
  • i is a positive odd number
  • ⁇ is the mechanical angle rotated by the rotor
  • N f and if are the number of series turns of each phase and the excitation current, respectively;
  • the rotor permeance is:
  • j is a positive integer
  • ⁇ is the mechanical angle rotated by the rotor
  • ⁇ 0 is the initial position angle of the rotor
  • ⁇ is the mechanical angular velocity
  • ⁇ r0 and ⁇ rj are 0 and jN r -order permeability coefficients, respectively;
  • the rotor permeability ⁇ r ( ⁇ , t) can be expressed as:
  • u 0 is the relative air permeability
  • ⁇ is the air gap length
  • ⁇ r ( ⁇ , t) is the rotor side air gap length
  • m is a positive integer
  • ⁇ r is the rotor slot width
  • R ap is the air gap radius
  • t is the corresponding time
  • Step 3 the distribution of salient pole teeth affects the stator tooth shape, and the stator permeance also changes accordingly.
  • the variable Sp represents the minimum number of cells in different situations, then the stator permeance ⁇ s ( ⁇ ) can be expressed as:
  • ⁇ s0 and ⁇ sk are 0 and kS p -order permeability coefficients, respectively;
  • stator permeance Similar to the rotor permeance, the above formula can only be analyzed qualitatively and cannot be solved quantitatively.
  • stator permeance according to the size parameters of the motor, the stator permeance can be expressed as:
  • n is a positive integer
  • ⁇ s ( ⁇ ) is the rotor side air gap length
  • b n is expressed as:
  • stator permeance From the quantitative expression form of stator permeance, the harmonic order and amplitude of stator permeance are related to the value of Sp ;
  • Step 4 the motor permeance is the combined permeance of the stator and rotor ⁇ ( ⁇ , t):
  • the field winding flux density B( ⁇ , t) is expressed as:
  • ⁇ 0 , ⁇ k , ⁇ j , and ⁇ kj are the 0-order, stator k-order, j-order, and kj-order permeability coefficients, respectively, and F i is the coefficient;
  • the magnetic density consists of five parts, the first two parts are stationary, the rest are rotating, and the orders of the two parts 4 and 5 are related to kS p , therefore, different S p makes the harmonic composition of the motor magnetic density and The corresponding amplitudes all change;
  • Step 5 the winding function N( ⁇ ) is:
  • v is a positive integer
  • Ni is the number of turns of the armature winding in series
  • phase flux linkage ⁇ p (t) is:
  • l a is the effective axial length of the motor
  • the back EMF is:
  • stator permeability, flux density, flux linkage and back EMF can be obtained under the three models.
  • influence of different distributions of salient teeth on the stator flux harmonics is analyzed from the permeability model; then, by calculating The obtained magnetic densities can be used to obtain the differences in the order and amplitude of the magnetic densities under different conditions; then, the magnitudes of the flux linkages under different distributions are compared; finally, the optimal distribution structure is obtained through the obtained back EMF.
  • the present invention designs the distribution of the salient pole teeth of the bilateral multi-tooth motor, summarizes the distribution law of the salient pole teeth, deduces the harmonic amplitude and magnitude of the air gap magnetic density, and obtains the influence of the distribution of the salient pole teeth on the working wave. ; Calculate the motor thrust with different salient tooth distributions.
  • the amplitude of the working harmonics is optimized, and the thrust of the motor is improved.
  • the mover of the present invention is only a simple salient pole structure.
  • the simple mover salient pole structure can greatly reduce the manufacturing cost and reduce the difficulty of processing;
  • the two sets of windings of the present invention are placed in different slots of the stator, and the double-layer centralized type is adopted.
  • This winding method can effectively reduce the length of the end and reduce the copper consumption of the motor.
  • the windings are placed in different slots, effectively reducing the The complexity of the winding is improved, the processability is improved, and the two sets of windings are physically independent of each other, which improves the fault tolerance of the motor;
  • the two sets of windings and the two permanent magnets of the present invention are placed on the stator, the armature winding is connected to three-phase alternating current, the excitation winding is connected to direct current, and there are no brushes and slip rings in the motor, which effectively improves the reliability of motor operation. , reducing the difficulty of regular maintenance by personnel;
  • the doubly salient hybrid excitation motor designed by the present invention with the excitation of the excitation winding as the main permanent magnet and supplemented by the permanent magnet has the advantage of wide speed regulation, and can be applied to the field of urban rail transit. In addition, it is also suitable for straight lines such as CNC machining lathes. Reciprocating sports occasions;
  • the present invention designs a doubly salient motor with the excitation winding as the main and the permanent magnet as the auxiliary. While fully exerting the excitation capacity, the amount of the permanent magnet is minimized to achieve the best cost performance; reasonable design
  • the number of pole pairs of the armature winding, the number of pole pairs of the excitation winding and the number of salient poles of the rotor enable the armature winding to fully absorb the harmonics of the excitation magnetic field, so that the motor has multiple working waves, and the thrust of the motor and the magnetic regulation ability of the excitation winding are improved;
  • FIG. 1 is a schematic structural diagram of a magnetic field modulated doubly salient motor according to an embodiment of the present invention
  • Fig. 2 is the connection schematic diagram of the armature winding of the example of the present invention.
  • Fig. 3 is the connection schematic diagram of the excitation winding of the example of the present invention.
  • FIG. 4 is a schematic diagram of the magnetic field distribution when only the permanent magnet acts alone in the example of the present invention.
  • Fig. 5 is the distribution of salient pole teeth according to the relationship between ⁇ 1 , ⁇ 2 , ⁇ 3 , ⁇ 4 and ⁇ 5 according to the embodiment of the present invention, wherein, Fig. (a) shows the salient pole teeth when the minimum unit Sp is 1 Schematic diagram of distribution; Figure ( b ) is a schematic diagram of the distribution of salient pole teeth when the minimum unit Sp is 2; Figure (c) is a schematic diagram of the distribution of salient pole teeth when the minimum unit Sp is 4;
  • Figure 6 shows the stator permeability waveform and harmonic distribution under different salient pole tooth distributions under the action of only the excitation winding alone in the embodiment of the present invention, wherein Figure (a) is the stator permeability waveform, Figure (b) is the harmonic wave distributed;
  • Fig. 7 is the air gap flux density waveform and harmonic distribution under different salient pole tooth distributions under the action of only the excitation winding alone according to the embodiment of the present invention, wherein Fig. (a) is the air gap harmonic waveform, and Fig. (b) is the harmonic wave. wave distribution;
  • FIG. 8 shows the flux linkage waveform and harmonic distribution under different salient pole tooth distributions under the action of the excitation winding alone according to the embodiment of the present invention, wherein, Figure (a) is the flux linkage waveform, and Figure (b) is the flux linkage harmonic distribution. ;
  • FIG. 9 is the back EMF waveform and harmonic distribution under different salient pole tooth distributions under the action of the excitation winding alone according to the embodiment of the present invention, wherein, Figure (a) is the back EMF waveform, and Figure (b) is the back EMF harmonic distribution. ;
  • Fig. 10 is the variation curve of the thrust of the motor with the current when there is no permanent magnet in the embodiment of the present invention, when the excitation winding and the armature winding act;
  • Fig. 11 shows the permanent magnets placed in the excitation winding according to the embodiment of the present invention, the excitation winding and the armature winding both act, and the thrust of the motor changes with the current.
  • stator 1, mover, 3, stator teeth, 4, excitation teeth, 5, excitation winding, 6, armature winding, 71, salient pole teeth 1, 72, salient pole teeth 2, 81, permanent Magnets 1, 82, permanent magnets 2, 9, mover salient poles.
  • the present invention discloses a magnetic field modulation type double salient pole motor and a method for designing salient pole teeth distribution, comprising a stator (1) and a mover (2), the stator comprising a stator iron core, an armature winding (6) ), excitation winding (5), permanent magnet 1 (81), permanent magnet 2 (82).
  • a stator tooth (3) is split into two field teeth (4), and each field tooth (4) is split into two salient pole teeth (71 and 72).
  • the armature winding (6) is wound on the stator teeth (3), the excitation winding (5) is wound around two adjacent excitation teeth (4) split by different stator teeth (3), the armature winding and the excitation
  • the connection relationship of the windings is shown in Figure 2-3.
  • the armature winding (6) is wound on the stator teeth (3).
  • the excitation winding (5) is wound around the two splits formed by the splitting of different stator teeth. adjacent excitation teeth (4).
  • the two sets of permanent magnets (81 and 82) are magnetized tangentially, the permanent magnets (81 and 82) are placed in the excitation slot, the permanent magnet 1 (81) is placed in the excitation slot, the permanent magnet 2 ( 82) Embedded in the bottom of the excitation slot.
  • the polarities of the permanent magnets placed at the same position in two adjacent slots are opposite and the polarities of the permanent magnets in the same slot are also opposite.
  • the two permanent magnets in the same slot form a closed magnetic circuit in the stator teeth. In the case of high electrical load, the permanent magnet can effectively suppress the saturation of the motor and improve the thrust capacity of the motor.
  • the motor stator has a plurality of salient pole teeth, and the two salient pole teeth formed by the splitting of the same excitation tooth are independent of each other; and the mover is only a simple salient pole, after bilateral modulation, the air gap magnetic field Contains a wealth of work waves. Optimize the distribution of stator salient teeth and adjust the amplitude of each working wave, thereby increasing the thrust of the motor.
  • the number of stator teeth is N s
  • the widths of salient pole teeth 1 (71) and 2 (72) are respectively ⁇ 1 and ⁇ 2
  • the widths of the armature windings and field winding slots are respectively ⁇ 3 and ⁇ 5
  • the width of the excitation groove formed by the splitting of a single excitation tooth is ⁇ 4
  • the number of rotor salient poles is N r
  • the pole pitch of each tooth is ⁇ p .
  • the total number of salient pole teeth is 4N s , wherein the salient pole teeth 1 (71) and the salient pole teeth 2 (72) are both 2N s .
  • the distribution of salient pole teeth is related to ⁇ 1 , ⁇ 2 , ⁇ 3 , ⁇ 4 and ⁇ 5 .
  • the main idea of the design method for the distribution of the salient pole teeth is: summarizing the distribution law of the salient pole teeth; deriving the order and amplitude of the no-load air gap magnetic density in each case; The magnitude of the electric potential can obtain the optimal way of salient tooth distribution.
  • the distribution design method of salient pole teeth is as follows:
  • Step 1 take ⁇ 1 , ⁇ 2 , ⁇ 3 , ⁇ 4 and ⁇ 5 as the entry points, design the distribution of salient pole teeth. Based on a single stator tooth, the distribution of salient pole teeth can be summarized into the following three situations:
  • each stator tooth has a minimum unit
  • each stator tooth has two minimum units;
  • each stator tooth has four minimum units.
  • step 2 the permanent magnets 1 and 2 only form a closed magnetic circuit in the stator core, and the air gap flux density is generated by the excitation winding. Therefore, only the field winding needs to be analyzed.
  • the magnetomotive force of the field winding air gap is:
  • F fw is the magnetomotive force of the excitation winding
  • i is a positive odd number
  • ⁇ is the mechanical angle rotated by the rotor
  • N f and if are the number of series turns and excitation current of each phase, respectively.
  • the rotor permeance is:
  • j is a positive integer
  • ⁇ is the mechanical angle rotated by the rotor
  • ⁇ 0 is the initial position angle of the rotor
  • ⁇ is the mechanical angular velocity
  • ⁇ r0 and ⁇ rj are 0 and jN r -order permeability coefficients, respectively.
  • the rotor permeance can be expressed as:
  • ⁇ 0 is the relative air permeability
  • m is a positive integer
  • ⁇ r is the rotor slot width
  • R ap is the air gap radius
  • Step 3 the distribution of salient pole teeth affects the stator tooth shape, and the stator permeance also changes accordingly.
  • the stator permeance can be expressed as:
  • ⁇ s0 and ⁇ sk are 0 and kS p -order permeability coefficients, respectively.
  • stator permeance Similar to the rotor permeability, the above formula can only be analyzed qualitatively and cannot be solved quantitatively.
  • stator permeance according to the motor size parameters, the stator permeance can be expressed as:
  • n is a positive integer
  • stator permeance From the quantitative expression form of stator permeance, the harmonic order and amplitude of stator permeance are related to the value of Sp .
  • Step 4 the motor permeance is composed of stator permeance and rotor permeance:
  • ⁇ 0 , ⁇ k , ⁇ j , and ⁇ kj are the 0-order, stator k-order, j-order, and kj-order permeability coefficients, respectively.
  • the magnetic density consists of five parts, the first two parts are stationary, the rest are rotating, and the orders of both parts 4 and 5 are related to kS p . Therefore, different Sp makes the harmonic composition of the motor magnetic density and the corresponding amplitude change.
  • Step 5 the winding function is:
  • v is a positive integer
  • Ni is the number of turns of the armature winding in series.
  • the flux linkage of each phase is:
  • la is the effective axial length of the motor.
  • the back EMF is:
  • stator permeability, flux density, flux linkage and back EMF can be obtained under the three models.
  • the influence of different distributions of salient pole teeth on the harmonics of stator permeability is analyzed from the permeability model; then, the differences in the order and amplitude of the magnetic density under different conditions are obtained through the obtained magnetic density; The size of the flux linkage; finally, the optimal distribution structure is obtained through the obtained back EMF.
  • FIG. 1 is a schematic structural diagram of a doubly salient linear motor according to an embodiment of the present invention.
  • the motor is a three-phase motor, including a stator (1) and a rotor (2), the stator includes a stator core, an armature winding (6), an excitation winding (5), a permanent magnet 1 (81), a permanent magnet Magnet 2 (82).
  • Each stator tooth (3) is split into two field teeth (4), and each field tooth (4) is split into two salient pole teeth (71 and 72).
  • the armature winding (6) is wound on the stator teeth (3), and the excitation winding (5) is wound around two adjacent excitation teeth (4) split by different stator teeth (3).
  • Both permanent magnet 1 and permanent magnet 2 are placed in the excitation winding slot, permanent magnet 1 is placed at the opening of the excitation slot, and permanent magnet 2 is placed at the bottom of the excitation slot. Both permanent magnets are magnetized tangentially, and the magnetization polarities of permanent magnets 1 and 2 in the same slot are opposite, forming a loop in a single stator tooth. The polarities of the permanent magnets placed at the same position in the adjacent excitation slots are opposite, and the permanent magnets in different slots are independent of each other. In the implementation row, there are 6 stator teeth N s of the motor, 12 salient pole teeth 1, 12 salient pole teeth 2, and 24 salient pole teeth in total.
  • Figure 2 shows the connection relationship of the armature winding.
  • the armature winding is connected to three-phase alternating current, and the difference between the phases is 120° electrical angle.
  • Figure 3 shows the connection relationship of the excitation winding.
  • the excitation winding is fed with direct current, and the positive and negative currents affect the direction of the thrust.
  • Figure 4 shows the magnetic field distribution when the permanent magnets are excited alone. It can be seen from the figure that the two permanent magnets in each excitation winding slot form a closed magnetic circuit in the stator teeth.
  • Fig. 5 shows the distribution of salient pole teeth according to the relationship between ⁇ 1 , ⁇ 2 , ⁇ 3 , ⁇ 4 and ⁇ 5
  • Fig. 5(a) is when the five variables are not equal to each other ( ⁇ 1 ⁇ 2 ⁇ 3 ⁇ 4 ⁇ 5 ), each stator tooth has a minimum unit
  • Figure 6 shows the stator permeability waveform and harmonic distribution under three distributions of salient pole teeth.
  • model I means that the five variables are not equal to each other, each stator tooth has only one minimum unit of salient pole tooth distribution, and the minimum number of units Sp is 1;
  • model II means that the width of the excitation winding slot and the armature winding slot are equal And the width of salient pole tooth 1 and salient pole tooth 2 are equal, each stator tooth has a salient pole tooth distribution with 2 minimum units, and the minimum unit number Sp is 2;
  • model III represents the excitation winding slot, armature winding slot and The widths of the three excitation grooves are equal, and the widths of the salient pole teeth 1 and 2 are the same.
  • Each stator tooth has 4 minimum units, and the minimum unit number Sp is 4.
  • the harmonics of model I are: 6th, 12th, 18th and 24th; the harmonics of model II are: 12th and 24th; the harmonics of model III are: 24th; the harmonics of the three models are: The times can be respectively expressed as: Sp N s .
  • Figure 7 shows the air gap flux density waveform and harmonic distribution under three distributions of salient pole teeth. It can be seen from the figure that the harmonic orders of the three models are the same, but the magnitudes of the amplitudes are different. Among them, the 2nd, 4th, 8th, 10th, 14th, 16th and 22nd orders are all working waves; for the convenience of analysis, the working waves with higher harmonic orders and less contribution are ignored. For the 2nd harmonic with the highest amplitude is model I; the 4th harmonic with the highest amplitude is model II; the 8th harmonic with the highest amplitude is model III. From the results, the distribution of salient pole teeth has a greater impact on the amplitude of harmonics.
  • Figure 8 shows the flux linkage waveform and harmonic distribution under three distributions of salient pole teeth.
  • the fundamental amplitude of model I is the highest of 0.037Wb, followed by model II is 0.036Wb, and the fundamental amplitude of model III is the lowest of 0.028Wb.
  • Model I and model II are less different, while model III is 26% lower than model I.
  • Figure 9 shows the back EMF waveforms and harmonic distributions under three distributions of salient pole teeth.
  • the situation of the back EMF is the same as that of the flux linkage.
  • the amplitude of the fundamental wave of model I is the highest at 13.0V
  • the second of model II is 12.7V
  • the amplitude of the fundamental wave of model III is the lowest of 9.9V.
  • Figure 10 shows the change curve of the thrust of the motor with the current when there is no permanent magnet, and the excitation winding and the armature winding act.
  • the thrust is from 102N to 320N;
  • the armature winding current is 10A, when the field winding current is increased from 5A to 20A, the thrust is from 172N to 553N;
  • the thrust is from 194N to 694N;
  • the armature winding current is 20A, when the field winding current is increased from 5A to 20A, the thrust is from 191N to 736N. From the results, when the armature winding current is 20A and the excitation current is 5A, the motor is saturated.
  • Figure 11 shows that the permanent magnet is placed in the excitation winding. Both the excitation winding and the armature winding act.
  • the thrust of the motor changes with the current.
  • the armature winding current is 5A, when the field winding current is increased from 5A to 20A, the thrust is from 106N to 360N; when the armature winding current is 10A, when the field winding current is increased from 5A to 20A, the thrust is from 180N to 661N;
  • the armature winding current is 15A, when the field winding current is increased from 5A to 20A, the thrust is from 210N to 866N;
  • the armature winding current is 20A, when the field winding current is increased from 5A to 20A, the thrust is from 220N to 952N. Comparing Figures 10 and 11, it can be seen that the permanent magnet design keeps the motor saturated, and the motor has a higher thrust capacity.
  • the present invention discloses a magnetic field modulated double salient pole motor and a method for designing salient pole teeth distribution.
  • the motor includes a stator and a mover, and the stator includes a stator core, two sets of windings and two sets of permanent magnets. .
  • Each stator tooth is split into two field teeth, and each field tooth is split into two salient pole teeth.
  • the two sets of windings are the armature winding and the excitation winding respectively.
  • the armature winding is wound around the stator teeth, and the excitation winding is wound around the adjacent excitation teeth which are split by different stator teeth.
  • Two sets of permanent magnets magnetized along the tangential direction are respectively placed at the notch and the bottom of the excitation slot, and the polarities of the permanent magnets placed in the same slot and the same position in the adjacent slot are opposite.
  • the structure design of the double salient poles of the present invention enables the magnetic field to modulate rich working harmonics in the air gap to improve the thrust of the motor; the permanent magnet magnetic circuit only forms a loop in the stator teeth, which solves the problem of easy saturation of the motor.
  • the salient pole tooth distribution design method of the present invention summarizes the salient pole tooth distribution mode according to the size relationship, combines the magnetic field modulation theory, and uses formulas to derive and solve the stator and rotor magnetic permeability, magnetic density, flux linkage and back electromotive force, etc., by changing the salient pole tooth distribution Adjust the harmonic contribution amplitude to improve motor thrust.
  • the patented motor of the present invention is not only suitable for linear structures, but the design idea is also applicable to rotating structures. All excitations of the motor are placed on the stator, which has great application prospects in rail transportation that requires long strokes and large thrust. .

Abstract

一种磁场调制式双凸极电机及其凸极齿分布设计方法，包括定子（1）和动子（2），定子（1）包括定子铁心、两套绕组和两套永磁体（81、82）。每个定子齿（3）分裂成两个励磁齿（4），每个励磁齿（4）分裂成两个凸极齿（71、72）。两套绕组分别为电枢绕组（6）和励磁绕组（5），电枢绕组（6）绕制于定子齿（3），励磁绕组（5）绕制在由不同定子齿（3）分裂而成的相邻励磁齿（4）。两套沿切向充磁的永磁体（81、82）分别放置在励磁槽槽口和槽底，放置于同一个槽以及相邻槽相同位置的永磁体极性相反。上述双凸极的结构设计，使得磁场能够在气隙中调制出丰富的工作谐波，提升电机推力；永磁体磁路仅在定子齿（3）中形成回路，解决了电机易饱和的问题。上述凸极齿（71、72）分布设计方法，通过改变凸极齿（71、72）分布调整谐波贡献幅值，从而提升电机推力。

Description

一种磁场调制式双凸极电机及其凸极齿分布设计方法 技术领域
本发明涉及到一种磁场调制式双凸极电机及其凸极齿分布设计方法，属于电机领域。采用可变磁通的混合励磁双凸极结构设计，使电机具有高推力、宽调速范围的性能优势；两套绕组和两套永磁体均安装于定子的结构特点，可应用于航天航空、交通运输、风力发电等领域。。
背景技术
随着开关磁阻电机的不断发展，美国学者Lipo教授等将永磁体引入开关磁阻电机提出了永磁双凸极电机。由于永磁材料偏高的价格，为了节约电机制造成本，电励磁双凸极电机在此基础上发展而来。电励磁双凸极电机的绕组均放置于定子侧，动子侧为简单的凸极结构，因而具有较好的温度管理和高速运行能力。相较于永磁双凸极电机，电励磁双凸极电机的励磁磁场可以通过改变励磁电流大小控制，控制简单，可靠性高，在航天航空、交通运输、风力发电等领域都具有广阔的应用前景。
中国发明专利申请号CN201910270245.9公开了一种失磁容错的双定子电励磁双凸极电机及其方法，通过双定子的结构增加一套内定子绕组，充分地利用电机空间；转子设有隔磁桥，主磁通经外定子、转子和内定子闭合，使电机在失磁故障时也可提供足够的磁阻转矩，实现电机的失错容错运行。虽然电机具有了容错能力，但是受制于电励磁电机本身，为了进一步提升电机的推力，将永磁体引入电励磁电机中，使电机同时具有永磁电机高得推力密度且具有宽速范围。
中国发明专利号CN201010592364.5公开了两种轴向励磁双凸极电机，两种电机均包括至少两个按轴向排列的双凸极单体结构。第一种结构定转子铁心通过轴向导磁背铁形成轴向磁路；第二种定子铁心、定子轴向导磁背铁及永磁体均为圆弧状结构。本发明空间利用率高，无附加气隙，励磁效率高。虽然该发明有效节约了空间，永磁体的引入也提升了电机的推力密度及励磁效率，但是复杂的结构设计，也给生产制造带来了难度。
中国发明专利申请号CN202010097798.1公开了一种有效永磁极数可调的双凸极永磁电机，通过在励磁绕组中施加脉冲电流改变低矫顽力永磁体的磁化方向，调节双凸极永磁电机永磁体极数，使电机调磁范围宽，广域高效。该电机虽然具有宽得调磁范围和高的推力 密度，但是，在励磁绕组中施加脉冲电流，也加重了控制电路设计难度，此外，脉冲电流过大造成永磁体不可逆退磁的风险，也是需要面临的问题。
从目前的研究来看，对于双凸极电机性能的研究，一般采用复杂的结构设计来提升电机的容错特性或推力能力，或对引入的永磁体形状和励磁绕组的电流进行研究，但却未对双凸极电机自身的凸极齿特点开展研究。
发明内容
本发明的目的是，针对现有双凸极电机凸极齿分布研究的空缺，提出了一种磁场调制式双凸极电机及其凸极齿分布设计方法，根据磁场调制的原理，公开了一种凸极齿的分布设计方法：探明凸极齿分布机理，从而优化工作波的幅值，提高电机推力和调磁能力。考虑到永磁材料价格的偏高，电机设计中以励磁绕组为主，永磁材料为辅，尽量发挥励磁绕组的性能，设计了一种磁场调制式双凸极电机，此外，为了进一步提升电机的推力能力，通过永磁体的设计，增强电机抗饱和能力。
具体地说，本发明的电机是采取以下的技术方案来实现的：一种磁场调制式双凸极电机，包括定子(1)和动子(2)，所述定子包括定子铁心、定子齿(3)、电枢绕组(6)、励磁绕组(5)、永磁体1(81)、永磁体2(82)；每个定子齿(3)分裂成两个励磁齿(4)，每个励磁齿(4)分裂成凸极齿1(71)和凸极齿2(72)这两个凸极齿；电枢绕组(6)绕制于定子齿(3)上，为了保证励磁绕组和电枢绕组中心线相差半个定子齿距，励磁绕组(5)绕制于由不同定子齿分裂所形成的两个相邻励磁齿(4)上；永磁体1(81)、永磁体2(82)均为切向充磁，且均放置于励磁槽内，永磁体1(81)放置于励磁槽口，永磁体2(82)内嵌于励磁槽底部；相邻两槽相同位置放置的永磁体极性相反且相同槽内的永磁体极性也相反；单独永磁激励时，同一个槽内的两个永磁体在定子齿内形成一个封闭的磁路；所述动子由多个转子凸极(9)排列构成，经过双边调制，气隙磁场中含有丰富的工作波。
进一步，由同一个励磁齿(4)分裂形成的凸极齿1(71)和凸极齿2(72)齿宽相互独立，互不影响；优化定子凸极齿的分布，调整每次工作波的幅值来提升电机的推力。
进一步，定子齿(3)齿数为N s，凸极齿1(71)和凸极齿2(72)的宽度分别为β 1和β 2，电枢绕组(6)和励磁绕(5)组槽口宽度分别为β 3和β 5，单个励磁齿(4)分裂后形成的励磁凹槽宽度为β 4，转子凸极(9)个数为N r，每个齿极距为τ p，各参数的关系可表示为：
β 1+β 2+β 3+β 4+β 5＝N rτ p/N s。
本发明的设计方法的技术方案包括，定子凸极齿的总数为4N s，其中，凸极齿1(71)和凸极齿2(72)均为2N s，凸极齿的分布与β 1、β 2、β 3、β 4和β 5有关，凸极齿分布的设计 方法的主要为：归纳凸极齿的分布规律；推导每种情况下空载气隙磁密的阶次和幅值；再通过磁密求解出反电势，通过对比反电势的大小，得到凸极齿分布的最优方式；具体如下：
步骤1，以β 1、β 2、β 3、β 4和β 5为切入点，对凸极齿分布进行设计，以单个定子齿为基准，凸极齿的分布可归纳为下列三种模型Model：
Model I.当β 1、β 2、β 3、β 4和β 5五个变量互不相等时，每个定子齿具有一个最小单元；
Model II.当励磁绕组槽口和电枢绕组槽口宽度相等且凸极齿1(71)和凸极齿2(72)宽度相等，即β 1≠β 4≠β 3,β 1＝β 2,β 3＝β 5，则每个定子齿具有两个最小单元；
Model III.当励磁绕组槽口、电枢绕组槽口和励磁凹槽三者宽度相等且凸极齿1(71)和凸极齿2(72)宽度相等，即β 1≠β 3,β 1＝β 2＝β 4,β 3＝β 5，则每个定子齿具有四个最小单元；
步骤2，永磁体1(81)、永磁体2(82)仅在定子铁心内形成闭合的磁路，气隙磁密由励磁绕组产生，因而只需对励磁绕组进行分析，励磁绕组气隙磁动势为：
Figure PCTCN2020133829-appb-000001
其中，F fw为励磁绕组磁动势，i为正奇数，θ为转子转过的机械角度，N f和i f分别为每相串联匝数和励磁电流；
转子磁导为：
Figure PCTCN2020133829-appb-000002
其中，j为正整数，θ为转子转过的机械角度，θ 0为转子初始位置角，ω为机械角速度，Λ r0和Λ rj分别为0和jN r阶磁导系数；
由于上述公式只能定性分析而无法定量求解，为了定量分析转子磁导，根据电机尺寸参数，转子磁导Λ r(θ,t)可表示为：
Figure PCTCN2020133829-appb-000003
其中，u 0为相对空气磁导，δ为气隙长度，δ r(θ,t)为转子侧气隙长度；
Figure PCTCN2020133829-appb-000004
其中，m为正整数，β r为转子槽口宽度，R ap为气隙半径，t为所对应的时间；
步骤3，凸极齿的分布影响定子齿形，则定子磁导也随之变化，用变量S p表示不同情况的最小单元数，则定子磁导Λ s(θ)可表示为：
Figure PCTCN2020133829-appb-000005
其中，k为正整数，Λ s0和Λ sk分别为0和kS p阶磁导系数；
与转子磁导同理，上述公式也只能定性分析而无法定量求解，为了定量分析定子磁导，根据电机尺寸参数，定子磁导可表示为：
Figure PCTCN2020133829-appb-000006
Figure PCTCN2020133829-appb-000007
其中，n为正整数，δ s(θ)为转子侧气隙长度，系数b n表示为：
Figure PCTCN2020133829-appb-000008
从定子磁导的定量表述形式来看，定子磁导谐波阶次和幅值均与S p的取值有关；
步骤4，电机磁导为定子磁导和转子磁导合成磁导Λ(θ,t)：
Figure PCTCN2020133829-appb-000009
励磁绕组磁密B(θ,t)表示为：
Figure PCTCN2020133829-appb-000010
其中，Λ 0、Λ k、Λ j、Λ kj分别为0阶、定子k阶、j次、kj次磁导系数，F i为系数；
磁密由五部分组成，前两部分为静止的，其余部分为旋转的，而4和5两个部分的阶 次均与kS p有关，因而，不同S p使得电机磁密的谐波组成及相应的幅值大小均变化；
步骤5，绕组函数N(θ)为：
Figure PCTCN2020133829-appb-000011
其中，v为正整数，N i为电枢绕组串联匝数；
每相磁链ψ p(t)为：
Figure PCTCN2020133829-appb-000012
其中，l a为电机有效轴向长度；
反电势为：
Figure PCTCN2020133829-appb-000013
通过上述公式推导，可以求得三种模型下的定子磁导、磁密、磁链和反电势，先从磁导模型分析凸极齿不同分布对定子磁导谐波的影响；然后，通过求得的磁密得出不同情况下磁密阶次和幅值的差异；之后，对比不同分布下的磁链大小；最后，通过求得的反电势得到最优的分布结构。
有益效果:
本发明采用上述设计方案后，可以具备如下有益效果：
1.本发明对双边多齿类电机凸极齿的分布进行设计，总结凸极齿的分布规律，推导气隙磁密的谐波幅值和大小，得到凸极齿的分布对工作波的影响；计算不同凸极齿分布时的电机推力。通过对凸极齿的分布设计方法的研究，优化工作谐波的幅值，提升电机的推力。
2.本发明动子仅为简单的凸极结构，对于双凸极电机而言，简单的动子凸极结构，可以大幅削减制造成本，降低加工的难度；
3.本发明两套绕组均放置于定子的不同槽内，采用双层集中式，该绕制方式可有效减小端部长度，降低电机的铜耗，绕组安放在不同的槽内，有效降低了绕线的复杂度，提高了加工的工艺性，两套绕组物理上相互独立，提高了电机的容错能力；
4.本发明的两套绕组和两永磁体均放置于定子上，电枢绕组通入三相交流电，励磁绕组通入直流电，电机内无电刷和滑环，有效提高了电机运行的可靠性，降低了人员定期维护的难度；
5.本发明所设计的以励磁绕组励磁为主永磁为辅的双凸极混合励磁电机，具有宽调速 的优势，可应用于城市轨道交通领域，此外，还适用于数控加工车床等直线往复运动场合；
6.本发明基于磁场调制原理，设计了以励磁绕组为主，永磁体为辅的双凸极电机，充分发挥励磁能力的同时，尽量减少永磁体的用量，以达到性价比最优；合理的设计电枢绕组极对数、励磁绕组极对数和转子凸极数，使电枢绕组能够充分吸收励磁磁场谐波，使得电机具有多工作波，提升电机的推力及励磁绕组的调磁能力；
附图说明
图1为本发明一个实施例的磁场调制式双凸极电机结构示意图；
图2为本发明实例电枢绕组的连接示意图；
图3为本发明实例励磁绕组的连接示意图；
图4为本发明实例仅永磁体单独作用时的磁场分布示意图；
图5为本发明实施例根据β 1、β 2、β 3、β 4和β 5的关系，凸极齿的分布情况，其中，图(a)为最小单元S p为1时，凸极齿分布示意图；图(b)为最小单元S p为2时，凸极齿分布示意图；图(c)为最小单元S p为4时，凸极齿分布示意图；
图6为本发明实施例仅励磁绕组单独作用下，不同凸极齿分布下的定子磁导示波形及谐波分布，其中，图(a)为定子磁导波形，图(b)为谐波分布；
图7为本发明实施例仅励磁绕组单独作用下，不同凸极齿分布下的气隙磁密波形和谐波分布，其中，图(a)为气隙谐波波形，图(b)为谐波分布；
图8为本发明实施例仅励磁绕组单独作用下，不同凸极齿分布下的磁链波形及谐波分布，其中，图(a)为磁链波形，图(b)为磁链谐波分布；
图9为本发明实施例仅励磁绕组单独作用下，不同凸极齿分布下的反电势波形及谐波分布，其中，图(a)为反电势波形，图(b)为反电势谐波分布；
图10为本发明实施例无永磁体，励磁绕组和电枢绕组作用时，电机的推力随电流变化曲线；
图11为本发明实施例励磁绕内放置永磁体，励磁绕组和电枢绕组均作用，电机的推力随电流变化曲线。
图中：1、定子，2、动子，3、定子齿，4、励磁齿，5、励磁绕组，6、电枢绕组，71、凸极齿1，72、凸极齿2，81、永磁体1，82、永磁体2，9、动子凸极。
具体实施方式
为了使本发明的目的、技术方案和效果更加清晰明白，下面结合附图和具体的实施例子，对本发明电机的结构特点和有益效果进行详细描述。
如图1所示，本发明公开了一种磁场调制式双凸极电机及其凸极齿分布设计方法，包括定子(1)和动子(2)，定子包括定子铁心、电枢绕组(6)、励磁绕组(5)、永磁体1(81)、永磁体2(82)。每个定子齿(3)分裂成两个励磁齿(4)，每个励磁齿(4)分裂成两个凸极齿(71和72)。电枢绕组(6)绕制于定子齿(3)上，励磁绕组(5)绕制于由不同定子齿(3)分裂而成的相邻两个励磁齿(4)，电枢绕组和励磁绕组的连接关系见图2-3。
电枢绕组(6)绕制于定子齿(3)上，为了保证励磁绕组和电枢绕组中心线相差半个定子齿距，励磁绕组(5)绕制于由不同定子齿分裂所形成的两个相邻励磁齿(4)。
其特征在于，两套永磁体(81和82)均为切向充磁，永磁体(81和82)均放置于励磁槽内，永磁体1(81)放置于励磁槽口，永磁体2(82)内嵌于励磁槽底部。相邻两槽相同位置放置的永磁体极性相反且相同槽内的永磁体极性也相反。单独永磁激励时，同一个槽内的两个永磁体在定子齿内形成一个封闭的磁路。在高电负荷情况下，永磁体可以有效抑制电机饱和，提升电机推力能力。
其特征在于，由于该电机定子具有多个凸极齿，并且由同一个励磁齿分裂形成的两个凸极齿尺寸相互独立；而动子仅为简单的凸极，经过双边调制，气隙磁场中含有丰富的工作波。优化定子凸极齿的分布，调整每次工作波的幅值，从而提升电机的推力。
其特征在于：定子齿数为N s，凸极齿1(71)和凸极齿2(72)的宽度分别为β 1和β 2，电枢绕组和励磁绕组槽口宽度分别为β 3和β 5，单个励磁齿分裂后形成的励磁凹槽宽度为β 4，转子凸极个数为N r，每个齿极距为τ p。各参数的关系可表示为：
β 1+β 2+β 3+β 4+β 5＝N rτ p/N s
其特征在于：凸极齿的总数为4N s，其中，凸极齿1(71)和凸极齿2(72)均为2N s。凸极齿的分布与β 1、β 2、β 3、β 4和β 5有关。凸极齿分布的设计方法的主要思路为：归纳凸极齿的分布规律；推导每种情况下空载气隙磁密的阶次和幅值；再通过磁密求解出反电势，通过对比反电势的大小，得到凸极齿分布的最优方式。凸极齿的分布设计方法具体如下：
步骤1，以β 1、β 2、β 3、β 4和β 5为切入点，对凸极齿分布进行设计，以单个定子齿为基准，凸极齿的分布可归纳为下列三种情况：
情况1.当五个变量互不相等时，每个定子齿具有一个最小单元；
情况2.当励磁绕组槽口和电枢绕组槽口宽度相等且凸极齿1和凸极齿2宽度相等，即(β 1≠β 4≠β 3,β 1＝β 2,β 3＝β 5)，则每个定子齿具有两个最小单元；
情况3.当励磁绕组槽口、电枢绕组槽口和励磁凹槽三者宽度相等且凸极齿1和凸极齿2宽度相等，即(β 1≠β 3,β 1＝β 2＝β 4,β 3＝β 5)，则每个定子齿具有四个最小单元。
步骤2，永磁体1和2仅在定子铁心内形成闭合的磁路，气隙磁密由励磁绕组产生。因而只需对励磁绕组进行分析。励磁绕组气隙磁动势为：
Figure PCTCN2020133829-appb-000014
其中，F fw为励磁绕组磁动势，i为正奇数，θ为转子转过的机械角度，N f和i f分别为每相串联匝数和励磁电流。
转子磁导为：
Figure PCTCN2020133829-appb-000015
其中，j为正整数，θ为转子转过的机械角度，θ 0为转子初始位置角，ω为机械角速度，Λ r0和Λ rj分别为0和jN r阶磁导系数。
由于上述公式只能定性分析而无法定量求解，为了定量分析转子磁导，根据电机尺寸参数，转子磁导可表示为：
Figure PCTCN2020133829-appb-000016
Figure PCTCN2020133829-appb-000017
其中，μ 0为相对空气磁导，m为正整数，β r为转子槽口宽度，R ap为气隙半径。
步骤3，凸极齿的分布影响定子齿形，则定子磁导也随之变化，用变量S p表示不同情况的最小单元数，则定子磁导可表示为：
Figure PCTCN2020133829-appb-000018
其中，k为正整数，Λ s0和Λ sk分别为0和kS p阶磁导系数。
与转子磁导同理，上述公式也只能定性分析而无法定量求解。为了定量分析定子磁导，根据电机尺寸参数，定子磁导可表示为：
Figure PCTCN2020133829-appb-000019
Figure PCTCN2020133829-appb-000020
其中，n为正整数，
Figure PCTCN2020133829-appb-000021
从定子磁导的定量表述形式来看，定子磁导谐波阶次和幅值均与S p的取值有关。
步骤4，电机磁导为定子磁导和转子磁导合成磁导：
Figure PCTCN2020133829-appb-000022
励磁绕组磁密表示为：
Figure PCTCN2020133829-appb-000023
其中，Λ 0、Λ k、Λ j、Λ kj分别为0阶、定子k阶、j次、kj次磁导系数。
磁密由五部分组成，前两部分为静止的，其余部分为旋转的，而4和5两个部分的阶次均与kS p有关。因而，不同S p使得电机磁密的谐波组成及相应的幅值大小均变化。
步骤5，绕组函数为：
Figure PCTCN2020133829-appb-000024
其中，v为正整数，N i为电枢绕组串联匝数。
每相磁链为：
Figure PCTCN2020133829-appb-000025
其中，l a为电机有效轴向长度。
反电势为：
Figure PCTCN2020133829-appb-000026
通过上述公式推导，可以求得三种模型下的定子磁导、磁密、磁链和反电势。先从磁导模型分析凸极齿不同分布对定子磁导谐波的影响；然后，通过求得的磁密得出不同情况下磁密阶次和幅值的差异；之后，对比不同分布下的磁链大小；最后，通过求得的反电势得到最优的分布结构。
实例
图1为本发明一个实施例的双凸极直线电机的结构示意图。如图1所示，该电机为三相电机，包括定子(1)和转子(2)，定子包括定子铁心、电枢绕组(6)、励磁绕组(5)、永磁体1(81)、永磁体2(82)。每个定子齿(3)分裂成两个励磁齿(4)，每个励磁齿(4)分裂成两个凸极齿(71和72)。电枢绕组(6)绕制于定子齿(3)上，励磁绕组(5)绕制于由不同定子齿(3)分裂而成的相邻两个励磁齿(4)。永磁体1和永磁体2均放置于励磁绕组槽内，永磁体1放置在励磁槽槽口，永磁体2放置在励磁槽槽底。两种永磁体均切向充磁，同一个槽内的永磁体1和2充磁极性相反，在单个定子齿内形成回路。相邻励磁槽的相同位置放置的永磁体极性相反，不同槽内的永磁体相互独立。实施列中电机定子齿N s为6个，凸极齿1有12个，凸极齿2有12个，凸极齿共24个。
图2为电枢绕组的连接关系，电枢绕组通入三相交流电，相与相之间相差120°电角度。
图3为励磁绕组的连接关系，励磁绕组通入直流电，电流的正负影响推力的方向。
图4为永磁体单独激励时的磁场分布，从图中可以看出，每个励磁绕组槽内的两个永磁体在定子齿内形成闭合的磁路。
图5为按照β 1、β 2、β 3、β 4和β 5的关系，凸极齿的分布情况，图5(a)为当五个变量互不相等时(β 1≠β 2≠β 3≠β 4≠β 5)，每个定子齿具有一个最小单元；图5(b)为当励磁绕组槽口和电枢绕组槽口宽度相等且凸极齿1和凸极齿2宽度相等，即(β 1≠β 4≠β 5,β 1＝β 2,β 3＝β 5)，则每个定子齿具有两个最小单元；图5(c)为当励磁绕组槽口、电枢绕组槽口和励磁凹槽三者宽度相等且凸极齿1和凸极齿2宽度相等，即(β 1≠β 5,β 1＝β 2＝β 4,β 3＝β 5)，则每个定子齿具有四个最小单元。
图6为凸极齿三种分布情况下的定子磁导波形和谐波分布。其中，model I表示五个变量互不相等，每个定子齿只有1个最小单元的凸极齿分布，最小单元数S p为1；model II表示励磁绕组槽口和电枢绕组槽口宽度相等且凸极齿1和凸极齿2宽度相等，每个定子齿具有2个最 小单元的凸极齿分布，最小单元数S p为2；model III表示励磁绕组槽口、电枢绕组槽口和励磁凹槽三者宽度相等且凸极齿1和凸极齿2宽度相等，每个定子齿具有4个最小单元，最小单元数S p为4。从图中可以看出，在24次谐波以内。model I的谐波有：6次、12次、18次和24次；model II的谐波有：12次和24次；model III的谐波有：24次；则三个模型的谐波阶次可分别表示为：S pN s。
图7为凸极齿三种分布情况下的气隙磁密波形和谐波分布。从图中可以看出，三个模型的谐波阶次相同，但幅值的大小存在差异。其中，2、4、8、10、14、16和22次均为工作波；为了方便分析，忽略谐波阶次较高、贡献较少的工作波。对于2次谐波幅值最高的为model I；4次谐波幅值最高的为model II；8次谐波幅值最高的为model III。从结果来看，凸极齿的分布对谐波的幅值具有较大的影响。
图8为凸极齿三种分布情况下的磁链波形和谐波分布。从图中可以看出，model I的基波幅值最高为0.037Wb，model II次之为0.036Wb，model III的基波幅值最低为0.028Wb。model I和model II差别较小，而model III比model I低26％。
图9为凸极齿三种分布情况下的反电势波形和谐波分布。从图中可以看出，反电势的情况和磁链相同，model I的基波幅值最高13.0V，model II次之12.7V，model III的基波幅值最低9.9V。对比可以发现，model I的凸极齿分布方式具有最好的效果。
图10为无永磁体，励磁绕组和电枢绕组作用时，电机的推力随电流变化曲线。当电枢绕组电流为5A是，励磁绕组电流从5A增加到20A时，推力从102N到320N；当电枢绕组电流为10A是，励磁绕组电流从5A增加到20A时，推力从172N到553N；当电枢绕组电流为15A是，励磁绕组电流从5A增加到20A时，推力从194N到694N；当电枢绕组电流为20A是，励磁绕组电流从5A增加到20A时，推力从191N到736N。从结果来看，当电枢绕组电流为20A，励磁电流为5A时，电机已经饱和。
图11为励磁绕内放置永磁体，励磁绕组和电枢绕组均作用，电机的推力随电流变化曲线。当电枢绕组电流为5A时，励磁绕组电流从5A增加到20A时，推力从106N到360N；当电枢绕组电流为10A时，励磁绕组电流从5A增加到20A时，推力从180N到661N；当电枢绕组电流为15A时，励磁绕组电流从5A增加到20A时，推力从210N到866N；当电枢绕组电流为20A时，励磁绕组电流从5A增加到20A时，推力从220N到952N。对比图10和11，可以看出永磁体的设计，一直了电机的饱和，电机具有更高的推力能力。
综上所述，本发明公开一种磁场调制式双凸极电机及其凸极齿分布设计方法，所述电 机包括定子和动子，所述定子包括定子铁心、两套绕组和两套永磁体。每个定子齿分裂成两个励磁齿，每个励磁齿分裂成两个凸极齿。两套绕组分别为电枢绕组和励磁绕组，电枢绕组绕制于定子齿，励磁绕组绕制在由不同定子齿分裂而成的相邻励磁齿。两套沿切向充磁的永磁体分别放置在励磁槽槽口和槽底，放置于同一个槽以及相邻槽相同位置的永磁体极性相反。本发明双凸极的结构设计，使得磁场能够在气隙中调制出丰富的工作谐波，提升电机推力；永磁体磁路仅在定子齿中形成回路，解决了电机易饱和的问题。本发明的凸极齿分布设计方法，根据尺寸关系归纳凸极齿分布方式，结合磁场调制理论，运用公式推导求解定转子磁导、磁密、磁链和反电势等，通过改变凸极齿分布调整谐波贡献幅值，从而提升电机推力。本发明专利电机除了适用于直线结构外，该设计思路同样适用于旋转结构，该电机的所有激励均放置于定子上的结构特点，在需要长行程大推力的轨道交通中具有较大的应用前景。
在本说明书的描述中，参考术语“一个实施例”、“一些实施例”、“示意性实施例”、“示例”、“具体示例”、或“一些示例”等的描述意指结合该实施例或示例描述的具体特征、结构、材料或者特点包含于本发明的至少一个实施例或示例中。在本说明书中，对上述术语的示意性表述不一定指的是相同的实施例或示例。而且，描述的具体特征、结构、材料或者特点可以在任何的一个或多个实施例或示例中以合适的方式结合。
尽管已经示出和描述了本发明的实施例，本领域的普通技术人员可以理解：在不脱离本发明的原理和宗旨的情况下可以对这些实施例进行多种变化、修改、替换和变型，本发明的范围由权利要求及其等同物限定。

Claims (4)

  1. 一种磁场调制式双凸极电机，其特征在于：包括定子(1)和动子(2)，所述定子包括定子铁心、定子齿(3)、电枢绕组(6)、励磁绕组(5)、永磁体1(81)、永磁体2(82)；每个定子齿(3)分裂成两个励磁齿(4)，每个励磁齿(4)分裂成凸极齿1(71)和凸极齿2(72)这两个凸极齿；
    电枢绕组(6)绕制于定子齿(3)上，为了保证励磁绕组和电枢绕组中心线相差半个定子齿距，励磁绕组(5)绕制于由不同定子齿分裂所形成的两个相邻励磁齿(4)上；
    永磁体1(81)、永磁体2(82)均为切向充磁，且均放置于励磁槽内，永磁体1(81)放置于励磁槽口，永磁体2(82)内嵌于励磁槽底部；相邻两槽相同位置放置的永磁体极性相反且相同槽内的永磁体极性也相反；单独永磁激励时，同一个槽内的两个永磁体在定子齿内形成一个封闭的磁路；
    所述动子由多个转子凸极(9)排列构成，经过双边调制，气隙磁场中含有丰富的工作波。
  2. 根据权利要求1所述的磁场调制式双凸极电机，其特征在于，由同一个励磁齿(4)分裂形成的凸极齿1(71)和凸极齿2(72)齿宽相互独立，互不影响；优化定子凸极齿的分布，调整每次工作波的幅值来提升电机的推力。
  3. 根据权利要求1所述的磁场调制式双凸极电机，其特征在于：定子齿(3)齿数为N s，凸极齿1(71)和凸极齿2(72)的宽度分别为β 1和β 2，电枢绕组(6)和励磁绕(5)组槽口宽度分别为β 3和β 5，单个励磁齿(4)分裂后形成的励磁凹槽宽度为β 4，转子凸极(9)个数为N r，每个齿极距为τ p，各参数的关系可表示为：
    β 1+β 2+β 3+β 4+β 5＝N rτ p/N s。
  4. 根据权利要求3所述的磁场调制式双凸极电机的凸极齿分布设计方法，其特征在于：定子凸极齿的总数为4N s，其中，凸极齿1(71)和凸极齿2(72)均为2N s，凸极齿的分布与β 1、β 2、β 3、β 4和β 5有关，凸极齿分布的设计方法的主要为：归纳凸极齿的分布规律；推导每种情况下空载气隙磁密的阶次和幅值；再通过磁密求解出反电势，通过对比反电势的大小，得到凸极齿分布的最优方式；具体如下：
    步骤1，以β 1、β 2、β 3、β 4和β 5为切入点，对凸极齿分布进行设计，以单个定子齿为基准，凸极齿的分布可归纳为下列三种模型Model：
    Model I.当β 1、β 2、β 3、β 4和β 5五个变量互不相等时，每个定子齿具有一个最小单元；
    Model II.当励磁绕组槽口和电枢绕组槽口宽度相等且凸极齿1(71)和凸极齿2(72)宽度相等，即β 1≠β 4≠β 3,β 1＝β 2,β 3＝β 5，则每个定子齿具有两个最小单元；
    Model III.当励磁绕组槽口、电枢绕组槽口和励磁凹槽三者宽度相等且凸极齿1(71)和凸极齿2(72)宽度相等，即β 1≠β 3,β 1＝β 2＝β 4,β 3＝β 5，则每个定子齿具有四个最小单元；
    步骤2，永磁体1(81)、永磁体2(82)仅在定子铁心内形成闭合的磁路，气隙磁密由励磁绕组产生，因而只需对励磁绕组进行分析，励磁绕组气隙磁动势为：
    Figure PCTCN2020133829-appb-100001
    其中，F fw为励磁绕组磁动势，i为正奇数，θ为转子转过的机械角度，N f和i f分别为每相串联匝数和励磁电流；
    转子磁导为：
    Figure PCTCN2020133829-appb-100002
    其中，j为正整数，θ为转子转过的机械角度，θ 0为转子初始位置角，ω为机械角速度，Λ r0和Λ rj分别为0和jN r阶磁导系数；
    由于上述公式只能定性分析而无法定量求解，为了定量分析转子磁导，根据电机尺寸参数，转子磁导Λ r(θ,t)可表示为：
    Figure PCTCN2020133829-appb-100003
    其中，u 0为相对空气磁导，δ为气隙长度，δ r(θ,t)为转子侧气隙长度；
    Figure PCTCN2020133829-appb-100004
    其中，m为正整数，β r为转子槽口宽度，R ap为气隙半径，t为所对应的时间；
    步骤3，凸极齿的分布影响定子齿形，则定子磁导也随之变化，用变量S p表示不同情况的最小单元数，则定子磁导Λ s(θ)可表示为：
    Figure PCTCN2020133829-appb-100005
    其中，k为正整数，Λ s0和Λ sk分别为0和kS p阶磁导系数；
    与转子磁导同理，上述公式也只能定性分析而无法定量求解，为了定量分析定子磁导， 根据电机尺寸参数，定子磁导可表示为：
    Figure PCTCN2020133829-appb-100006
    Figure PCTCN2020133829-appb-100007
    其中，n为正整数，δ s(θ)为转子侧气隙长度，系数b n表示为：
    Figure PCTCN2020133829-appb-100008
    从定子磁导的定量表述形式来看，定子磁导谐波阶次和幅值均与S p的取值有关；
    步骤4，电机磁导为定子磁导和转子磁导合成磁导Λ(θ,t)：
    Figure PCTCN2020133829-appb-100009
    励磁绕组磁密B(θ,t)表示为：
    Figure PCTCN2020133829-appb-100010
    其中，Λ 0、Λ k、Λ j、Λ kj分别为0阶、定子k阶、j次、kj次磁导系数，F i为系数；
    磁密由五部分组成，前两部分为静止的，其余部分为旋转的，而4和5两个部分的阶次均与kS p有关，因而，不同S p使得电机磁密的谐波组成及相应的幅值大小均变化；
    步骤5，绕组函数N(θ)为：
    Figure PCTCN2020133829-appb-100011
    其中，v为正整数，N i为电枢绕组串联匝数；
    每相磁链ψ p(t)为：
    Figure PCTCN2020133829-appb-100012
    其中，l a为电机有效轴向长度；
    反电势为：
    Figure PCTCN2020133829-appb-100013
    通过上述公式推导，可以求得三种模型下的定子磁导、磁密、磁链和反电势，先从磁导模型分析凸极齿不同分布对定子磁导谐波的影响；然后，通过求得的磁密得出不同情况下磁密阶次和幅值的差异；之后，对比不同分布下的磁链大小；最后，通过求得的反电势得到最优的分布结构。
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