WO2022009308A1

WO2022009308A1 - Power supply device and power supply system

Info

Publication number: WO2022009308A1
Application number: PCT/JP2020/026562
Authority: WO
Inventors: 良祐牛窪
Original assignee: Tdk株式会社
Priority date: 2020-07-07
Filing date: 2020-07-07
Publication date: 2022-01-13

Abstract

A power supply device according to one aspect of the present invention comprises: a transformer; a primary-side bridge circuit provided on the primary side of the transformer; a secondary-side bridge circuit provided on the secondary side of the transformer; and a resonant circuit provided on at least one of the primary and secondary sides of the transformer. The power supply device further comprises a control circuit that performs unidirectional or bidirectional power conversion, by driving the switch elements of the primary-side bridge circuit and the secondary-side bridge circuit by different control methods depending on the output voltage range.

Description

Power supply and power system

The present invention relates to a power supply device and a power supply system that convert DC power.

In the power supply apparatus for converting DC power, when performing power transmission in a wide output voltage range at the resonance circuit of the frequency control (LLC resonant circuit), in order to obtain a large step-up ratio, the ratio of the excitation inductor L _M and resonant inductor L _R it is necessary to design a large _{_{(= L R / L M)}} . For exciting the inductor L _M is designed from constraints of the excitation current necessary for the primary bridge to ZVS (Zero Voltage Switching), it will adjust the L _{R /} L _M in resonant inductor L _R (e.g., See Patent Document 1).

Japanese Unexamined Patent Publication No. 8-289540

By the way, when the resonance inductor _LR is increased, the applied voltage increases and the iron loss increases, so that it is necessary to increase the cross-sectional area and the number of turns of the core. Therefore, there is a problem that the volume of the resonant inductor L _R is increased. Also a resonant inductor L _R A when integrated with transformer, and the cross-sectional area of the core, it is necessary to increase the number of turns of the primary winding and the secondary winding, it is not easy to reduce the volume. Therefore, it is desirable to provide a power supply device capable of reducing the volume and a power supply system including such a power supply device.

The power supply device according to the first aspect of the present invention includes a transformer, a primary bridge circuit provided on the primary side of the transformer, a secondary bridge circuit provided on the secondary side of the transformer, and a transformer. It is provided with a resonance circuit provided on at least one of the primary side and the secondary side. This power supply further provides a control circuit that performs unidirectional or bidirectional power conversion by driving the switch elements of the primary side bridge circuit and the secondary side bridge circuit by different control methods depending on the output voltage range. I have. In both unidirectional and bidirectional power conversions, the primary side generally refers to the side connected to the commercial system, and the secondary side generally refers to the side connected to the load system. Pointing to.

The power supply system according to the second aspect of the present invention includes a power supply device that converts DC power, a stabilized DC power supply (PFC converter output, battery, etc.) connected to the primary side of the power supply device, and a power supply. It is equipped with a DC load device (various electric devices, batteries, etc.) connected to the secondary side of the device. The power supply device in this power supply system has the same configuration as the power supply device according to the first aspect.

According to the power supply device according to the first aspect of the present invention and the power supply system according to the second aspect of the present invention, the switch elements of the primary side bridge circuit and the secondary side bridge circuit are different from each other depending on the output voltage range. Since one-way or two-way power conversion is performed by driving with a control method, for example, PFM (Pulse Frequency Modulation) control is applied only to the primary side bridge circuit to apply the output voltage. Compared with the case of controlling or applying PWM (Pulse Width Modulation) control only to the secondary side bridge circuit to control the output voltage, the amount of energy stored in the resonant inductor can be reduced. As a result, the magnetic component can be miniaturized. Therefore, the volume of the power supply device can be reduced.

It is a figure which shows the circuit structure example of the DC-DC converter which concerns on 1st Embodiment of this invention. It is a figure which shows an example of the relationship between the duty ratio of the drive pulse in the DC-DC converter of FIG. 1 and the output voltage on the secondary side. It is a figure which shows an example of the drive pulse waveform in the step-down mode in the DC-DC converter of FIG. It is a figure which shows an example of the drive of the DC-DC converter in the T1 period (transmission mode) of FIG. It is a figure which shows an example of the drive of the DC-DC converter in the T2 period (reflux mode) of FIG. It is a figure which shows an example of the drive pulse waveform in the step-up mode in the DC-DC converter of FIG. It is a figure which shows an example of the drive of the DC-DC converter in the T3 period (storage mode) of FIG. It is a figure which shows an example of the drive of the DC-DC converter in the T4 period (emission mode) of FIG. It is a figure which shows an example of the relationship between the drive frequency and the output voltage of a secondary side in the case of controlling an output voltage by applying PFM control only to the primary side bridge circuit of the DC-DC converter which concerns on a comparative example. The figure which shows an example of the relationship between the duty ratio of a drive pulse and the output voltage of a secondary side in the case of controlling an output voltage by applying PWM control only to the secondary side bridge circuit of the DC-DC converter which concerns on a comparative example. Is. It is a figure which shows the circuit structure example of the DC-DC converter which concerns on the 2nd Embodiment of this invention. FIG. 11 is a diagram showing an example of the relationship between the duty ratio of the drive pulse and the peak value of the current flowing through the primary resonance circuit in the DC-DC converter of FIG. It is a figure which shows an example of the equivalent circuit of the resonance circuit of the primary side and the secondary side in the DC-DC converter of FIG. FIG. 11 is a diagram showing an example of the relationship between the drive frequency and the output voltage during charging of the DC-DC converter of FIG. 11. FIG. 11 is a diagram showing an example of the relationship between the drive frequency and the output voltage when the DC-DC converter of FIG. 11 is discharged. It is a figure which shows an example of the drive of a DC-DC converter in a transmission mode. It is a figure which shows an example of the drive of a DC-DC converter in a reflux mode. It is a figure which shows an example of the drive of a DC-DC converter in a storage mode. It is a figure which shows an example of the drive of the DC-DC converter in the emission mode. It is a figure which shows an example of the drive pulse waveform in Duty <0.5 (phase shift mode). It is a figure which shows an example of the drive pulse waveform in Duty = 0.5 (phase shift mode). It is a figure which shows an example of the drive pulse waveform in Duty> 0.5 (boosting PWM mode). It is a figure which shows an example of the drive of a DC-DC converter in Duty> 0.5 (boosting PWM mode). It is a figure which shows the circuit structure example of the DC-DC converter which concerns on 3rd Embodiment of this invention. It is a figure which shows an example of the circuit structure which generates the synchronous rectification pulse of the high side switch element on the rectification side at the time of charging. It is a figure which shows an example of the drive pulse waveform in Duty ≦ 0.5 (step-down mode). It is a figure which shows an example of the drive pulse waveform in Duty> 0.5 (boosting mode).

Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings. Each of the embodiments described below shows a preferred specific example of the present disclosure. Therefore, the numerical values, shapes, materials, components, arrangement positions of components, connection forms, and the like shown in the following embodiments are examples and are not intended to limit the present disclosure. Therefore, among the components in the following embodiments, the components not described in the independent claims indicating the highest level concept of the present disclosure are described as arbitrary components. It should be noted that each figure is a schematic view and is not necessarily exactly illustrated. Further, in each figure, the same reference numerals are given to substantially the same configurations, and duplicate explanations will be omitted or simplified. The explanation will be given in the following order.

1. 1. First Embodiment CL type one-way DC-DC converter (A one-way DC-DC converter provided with a resonance circuit by a capacitor and an inductor on the primary side will be described as an example).
2. 2. Second Embodiment CLLC type bidirectional DC-DC converter (A bidirectional DC-DC converter having a resonance circuit with a capacitor and an inductor on the primary side and the secondary side will be described as an example).
3. 3. Third Embodiment CLLC-type bidirectional DC-DC converter CLLC-type bidirectional synchronous rectification DC-DC converter

<1. First Embodiment>
[composition]
FIG. 1 shows a circuit diagram of a DC-DC converter 1 according to a first embodiment of the present invention. The DC-DC converter 1 includes one inductor L _RP and one capacitor C _RP as a resonance circuit on the primary side, and is a CL system utilizing resonance by these inductor L _RP and capacitor C _RP. It is a DC-DC converter. The DC-DC converter 1 includes a transformer 10, a primary side bridge circuit 20, a secondary side bridge circuit 30, a primary side driver 40, a secondary side driver 50, and a control circuit 60.

Transformer 10 has a primary coil _N P, 2 primary coil _{N S} and core CR. Primary coil _{N P} and the secondary coil _{N S} is wound magnetic core CR. When the number of turns of the primary coil _{N P} and _{n P,} the number of turns of the secondary coil _{N S} and _{n S,} the turns ratio of the transformer _10, n P: has a _{n S.} Inductor _{L RP} and capacitor _{C RP} is serially connected to the primary coil _{N P.} The turns ratio of the transformer 10 is set so that the reference output voltage _{(VS (std)} ) is at or near the center in the output voltage range of the DC-DC converter 1, for example, as shown in FIG. For example, with DC-DC converter 1 of the input voltage (primary voltage _{V P)} is set to 400V, when the output voltage range of the DC-DC converter 1 is set to 240V ~ 420 V, the DC-DC converter 1 The center of the output voltage range is 330V. At this time, the turns ratio (n _P : n _S ) of the transformer 10 is, for example, 5: 4.

The primary side bridge circuit 20 is a full bridge type circuit including four switch elements controlled on / off by the control circuit 60. Primary bridge circuit 20 includes two switching elements Q _B corresponding to the low side of the two switching elements of the four switching _elements, and Q _D, the two switch elements of the high-side of the four switching elements It has two corresponding switch elements Q _A and Q _C.

The secondary side bridge circuit 30 is a full bridge type circuit including two switch elements controlled on and off by the control circuit 60 and two diode elements. The secondary side bridge circuit 30 has two switch elements Q _E and Q _F _{corresponding to two low side switch elements and two diode elements D 1} and D ₂ corresponding to two high side diode elements. is doing.

The switch elements Q _A , Q _B , Q _C , Q _D , Q _E , and Q _F are composed of switch elements such as MOSFETs (Metal Oxide Semiconductor Field Effect Transistor). Switching element _{_{_{_{Q A, Q B, Q C}}}} , Q D, Q E, Q F has terminals _T _11, body diode are connected in parallel so as to be biased with respect to _{_{_{_{T 21 D QA, D QB,}}}} D QC , D _QD , D _QE , D _QF are included. Primary bridge circuit 20 includes the switching elements _{_{_{Q A, Q B, Q C}}} , the capacitor _C A connected in parallel to _{_{_{Q D, C B, C C}}} , the _{C D.} Secondary bridge circuit 30, the switch elements _Q E, _{Q F} in parallel connected capacitors _C E, has a _{C F.}

The switch elements Q _A , Q _B , Q _C , Q _D , Q _E , and Q _F may be composed of, for example, an RC (reverse conduction) -IGBT (insulated gate bipolar transistor). In this case, the FRD (fast recovery diode) built in the RC-IGBT substitutes for the body diode described above. The switch elements Q _A , Q _B , Q _C , Q _D , Q _E , and Q _F may be composed of, for example, a FET including a wide-gap semiconductor (SiC, GaN). In this case, the switch elements Q _A , Q _B , Q _C , Q _D , Q _E , and Q _F are externally attached in place of the above-mentioned body diode in order to prevent a decrease in efficiency due to the high forward voltage. Diodes may be connected in parallel.

Two switching elements _Q A, _{Q B} are connected in series with one another at a connection point _{P 1.} Two switching elements _Q C, _{Q D} are connected in series with one another at the connection point _{P 2.} Connection point _{P 1} and the connection point _{P 2} is the primary coil _{N P,} and the inductor _{L RP} and capacitor _{C RP} is connected via the series.

The diode element D ₂ and the switch element Q _F are connected in series with each other at the connection point P _3. The diode element D ₁ and the switch element Q _E are connected in series with each other at the connection point P _4. Connection point P ₄ and the connection point P ₃ is connected via the secondary coil N _S.

The DC-DC converter 1 further includes terminals T ₁₁ and T ₁₂ on the primary side and terminals T ₂₁ and T ₂₂ on the secondary side. The terminal T ₁₁ is connected to the high side of the primary side bridge circuit 20, and the terminal T ₁₂ is connected to the low side of the primary side bridge circuit 20. DC-DC converter 1, further the capacitor _{C P} between the terminal _{T 11} and the terminal _{T 12,} and a capacitor _{C S} between the terminal _{T 21} and the terminal _{T 22.} The terminal T ₂₁ is connected to the high side of the secondary side bridge circuit 30, and the terminal T ₂₂ is connected to the low side of the secondary side bridge circuit 30.

The primary side driver 40 transmits the control signal for performing phase shift control (PPS: Pulse Phase Shift) generated by the control circuit 60 to the four switch elements Q _A , Q _B , and Q included in the primary side bridge circuit 20. _C, and outputs it to the _{Q D.} Specifically, the primary driver 40, in the Duty range of 0.5 or less generated by the control circuit 60, a signal of fixed frequency and 50% within the pulse width switching element _Q A, the _{Q B} outputs for outputs switching element _Q a, a signal obtained by shifting an arbitrary phase that is determined by the Duty respect _{Q B} switching element _Q C, to _{Q D.} On the other hand, the primary side driver 40 has a fixed frequency and within 50% _{of the switch elements Q A} and Q _D and the switch elements Q _B and Q _{C in the} range where the duty generated by the control circuit 60 exceeds 0.5. Outputs a signal that turns on and off alternately with the pulse width of.

Secondary driver 50, PWM control that is generated by the control circuit 60 (PWM: Pulse Wide Modulation) 2 two switching elements _Q E contained a control signal to the secondary bridge circuit 30 which performs, and outputs the _{Q F.} Specifically, the secondary-side driver 50, in the Duty range of 0.5 or less generated by the control circuit 60, a fixed frequency and switching element signals 50 percent of the pulse width is determined by the Duty _{Q E} , Q _F is output. On the other hand, the secondary-side driver 50, in the range Duty generated by the control circuit 60 is more than 0.5, and outputs a signal having a pulse width greater than 50% at a fixed frequency switching element _Q E, the _{Q F.}

The control circuit 60 is configured by using a DSP (Digital Signal Processor), and is a switch element of the primary side bridge circuit 20 and the secondary side bridge circuit 30 via the primary side driver 40 and the secondary side driver 50. By controlling the above, power conversion from the primary side to the secondary side (one direction) is performed. The control circuit 60 detects the output voltage (secondary voltage _{V S),} generates a Duty as the output voltage (secondary voltage _{V S)} becomes the target voltage value. When the duty generated at this time is 50% or less, the control circuit 60 mainly uses the phase shift control signal output to the primary side driver 40 as the output voltage of the DC-DC converter 1 (secondary side voltage _VS). ) Is controlled. For example, the smaller the generated Duty, the smaller the phase shift amount, and a lower output voltage can be obtained. On the other hand, if the generated Duty exceeds 50%, the control circuit 60, mainly a PWM control signal to be output to the secondary-side driver 50, DC-DC converter 1 of the output voltage (secondary voltage _V S) To control. Specifically, the larger the generated Duty, the _{longer the time width during which the two switch elements Q E} and Q _F are turned on at the same time, and the larger the _{energy stored in the inductor L RP} at this time, so that a higher output voltage is obtained. Can be done.

[motion]
Next, the operation of the DC-DC converter 1 will be described.

FIG. 3 shows an example of the drive pulse waveform in the step-down mode (Duty ≦ 0.5) in the DC-DC converter 1. FIG. 4 shows an example of driving the DC-DC converter 1 in the period T1 (transmission mode) of FIG. FIG. 5 shows an example of driving the DC-DC converter 1 in the period T2 (reflux mode) of FIG. 4 and 5, together with the capacitor C _P is connected to the terminal T _11, T _12, a power supply system battery BT1 is connected as an example of the DC power supply is illustrated. Further, FIG. 4, 5, together with the capacitor C _S is connected to the terminal T _21, T _22, a power supply system battery BT2 is connected as an example of the load is illustrated. In this embodiment, the battery BT2 is assumed to be a secondary battery.

In FIG. 3, D _PPS represents the phase shift width when phase shift control is performed with respect to the primary side bridge circuit 20, and D _PWM represents the low side switch elements Q _E , Q _{F of the secondary side bridge circuit 30.} Represents the PWM width of. Further, in FIG. _{3, T SW} represents the length of one period, TAB, TCD denotes switching element QA of the primary bridge circuit 20, QB, QC, the dead time of the _{QD, T AE,} T _BF represents the delay amount to the low-side switching element _Q E of the secondary bridge circuit 30, a _{Q F} is ZVS (zero-voltage switching). D _PPS is determined by T _SW x D- (TAB + TCD), and D _PWM is _{determined by T SW} x D- (TAB + TCD) -T _AE . D _{used for deriving D PPS} and D _PWM is a duty control amount, and takes a value within the range of _{D min} to D _max.

The control circuit 60 performs a step-down operation by, for example, controlling the ratio between the length of the transmission mode period and the length of the reflux mode period. For example, as shown in FIG. 3, the control circuit 60 _{turns on the switch elements Q A} , Q _D , and Q _E, and turns off the switch elements Q _B , Q _C , and Q _F. As a result, for example, as shown in FIG. 4, a period (transmission mode) for transmitting the power conversion from the primary side to the secondary side is generated. Further, for example, as shown in FIG. 3, the control circuit 60 turns on the switch elements Q _C _{after turning off the switch elements Q D} and Q _E while keeping the switch element Q _A on. Then, for example, as shown in FIG. 5, the diode D ₂ is turned off, and a period (reflux mode) in which power is not transmitted from the primary side to the secondary side is generated.

FIG. 6 shows an example of the drive pulse waveform in the boost mode (Duty> 0.5) in the DC-DC converter 1. FIG. 7 shows an example of driving the DC-DC converter 1 in the period T3 (storage mode) of FIG. FIG. 8 shows an example of driving the DC-DC converter 1 in the period T4 (emission mode) of FIG. 7 and 8, together with the capacitor C _P is connected to the terminal T _11, T _12, a power supply system battery BT1 is connected as an example of the DC power supply is illustrated. Further, FIG. 7, 8, together with the capacitor C _S is connected to the terminal T _21, T _22, a power supply system battery BT2 is connected as an example of the load is illustrated. In this embodiment, the battery BT2 is assumed to be a secondary battery.

In FIG. 6, D _PPS represents the phase shift width when phase shift control is performed with respect to the primary side bridge circuit 20, and D _PWM represents the low side switch elements Q _E , Q _{F of the secondary side bridge circuit 30.} Represents the PWM width of. Further, in FIG. _{6, T SW} represents the length of one period, TAB, TCD denotes switching element QA of the primary bridge circuit 20, QB, QC, the dead time of the _{QD, T AE,} T _BF represents the delay amount to the low-side switching element _Q E of the secondary bridge circuit 30, a _{Q F} is ZVS (zero-voltage switching). D _PPS is determined by (1/2) × T _SW − (TAB + TCD) (= fixed value), and D _PWM is _{determined by T SW} × D − (TAB + TCD) −T _AE . D _{used for deriving D PPS} and D _PWM is a duty control amount, and takes a value within the range of _{D min} to D _max.

The control circuit 60 performs a boosting operation by, for example, controlling the ratio between the length of the period of the storage mode and the length of the period of the emission mode. For example, as shown in FIG. 6, the control circuit 60 _{turns on the switch elements Q A} , Q _D , Q _E , and Q _F and turns off the switch elements Q _B , Q _{C in the storage mode.} As a result, for example, as shown in FIG. 7, a period (storage mode) for storing energy _{in the inductor L RP is generated.} Further, for example, as shown in FIG. 6, the control circuit 60 turns off the _{switch element Q F} while keeping the switch elements Q _A , Q _D , and Q _{E on.} Then, for example, as shown in FIG. 8, a period (emission mode) in which the energy stored in the _{inductor L RP is released is generated.}

[effect]
Next, the effect of the DC-DC converter 1 according to the present embodiment will be described.

In the power supply apparatus for converting DC power, when performing power transmission in a wide output voltage range at the resonance circuit of the frequency control (LLC resonant circuit), in order to obtain a large step-up ratio, the ratio of the excitation inductor L _M and resonant inductor L _R it is necessary to design a large _{_{(= L R / L M)}} . For exciting the inductor L _M is designed from constraints of the excitation current necessary for the primary bridge to ZVS (Zero Voltage Switching), it will adjust the L _{R /} L _M in resonant inductor L _R (e.g., See Patent Document 1 above).

By the way, when the resonance inductor _LR is increased, the applied voltage increases and the iron loss increases, so that it is necessary to increase the cross-sectional area and the number of turns of the core. Therefore, there is a problem that the volume of the resonant inductor L _R is increased. Also a resonant inductor L _R A when integrated with transformer, and the cross-sectional area of the core, it is necessary to increase the number of turns of the primary winding and the secondary winding, it is not easy to reduce the volume.

On the other hand, in the present embodiment, power conversion in one direction is performed by driving the switch elements of the primary side bridge circuit 20 and the secondary side bridge circuit 30 by different control methods depending on the output voltage range. Thereby, for example, when PFM control is applied only to the primary side bridge circuit to control the output voltage (for example, see FIG. 9), or PWM control is applied only to the secondary side bridge circuit to control the output voltage. (e.g., see FIG. 10) can be the case of (hereinafter referred to as "control method according to a comparative example.") and compared, reduce the amount of energy stores in resonant inductor L _R. As a result, the magnetic component can be miniaturized. Therefore, the volume of the DC-DC converter 1 can be reduced.

Further, in the present embodiment, in the high output voltage region in the output voltage range of the DC-DC converter 1, PWM control for the secondary bridge circuit 20 is performed as the main control, and in the output voltage range of the DC-DC converter 1. In the low output voltage region, phase shift control for the primary bridge circuit 20 is performed as the main control. Thus, as compared with the control method of the comparative example, reducing the amount of energy stores in resonant inductor L _R. As a result, the magnetic component can be miniaturized. Therefore, the volume of the DC-DC converter 1 can be reduced.

_{Further, in the present embodiment, the reference output voltage (VS (std)} ) determined according to the turns ratio of the transformer 10 in the output voltage range of the DC-DC converter 1 is set to the output voltage range of the DC-DC converter 1. Set to be the center of. In the output voltage region higher than the reference output voltage ( _{VS (std)} ), the boost control is performed by performing the PWM control for the secondary side bridge circuit 30 as the main control, and the output voltage range of the DC-DC converter 1 is performed. In the output voltage region below the reference output voltage ( _{VS (std)} ) in the above, the step-down control is performed by performing the phase shift control for the primary side bridge circuit 20 as the main control. Thus, as compared with the control method of the comparative example, reducing the amount of energy stores in resonant inductor L _R. As a result, the magnetic component can be miniaturized. Therefore, the volume of the DC-DC converter 1 can be reduced.

Further, in the present embodiment, the PWM control for the secondary side bridge circuit 30 is performed as the main control in the operating region where the Duty exceeds 0.5, and the primary side bridge is performed in the operating region where the Duty is 0.5 or less. The phase shift control for the circuit 20 is performed as the main control. Thus, as compared with the control method of the comparative example, reducing the amount of energy stores in resonant inductor L _R. As a result, the magnetic component can be miniaturized. Therefore, the volume of the DC-DC converter 1 can be reduced.

Further, in the present embodiment, the switches of the primary side bridge circuit 20 and the secondary side bridge circuit 30 have a fixed drive frequency f _SW _{having a value larger than the resonance frequency f R} of the LC resonance circuit provided on the primary side. The element is controlled. As a result, the circuit current flowing through the DC-DC converter 1 at Duty = 0.5 is in continuous mode. As a result, the circuit current becomes small, so that the charging efficiency becomes higher than in the discontinuous mode.

Further, in this embodiment, the primary bridge circuit 20, a plurality of switching elements Q _A is on-off controlled by the control circuit _60, Q B, _Q _C, full-bridge, which is configured to include a Q _D It is a circuit of. Further, the secondary side bridge circuit 30 includes a plurality of low-side switch elements Q _E and Q _F controlled on and off by the control circuit 60, and a plurality of high-side diode elements D ₁ and D _2. It is a full bridge type circuit. Thus, by controlling the above switching element, as compared with the control method of the comparative example, reducing the amount of energy stores in resonant inductor L _R. As a result, the magnetic component can be miniaturized. Therefore, the volume of the DC-DC converter 1 can be reduced.

<2. Second Embodiment>
[composition]
FIG. 11 shows a circuit diagram of the DC-DC converter 2 according to the second embodiment of the present invention. The DC-DC converter 2 includes one inductor L _RP and one capacitor C _RP as the resonance circuit on the primary side, and one inductor L _RS and one capacitor as the resonance circuit on the secondary side. It is equipped with C _RS. The DC-DC converter 2 is a CLLC type DC-DC converter that utilizes resonance caused by _{these inductors L RP} and L _RS and capacitors C _RP and C _RS. The DC-DC converter 2 includes a transformer 10, a primary bridge circuit 20, a secondary bridge circuit 70, a primary driver 40, a secondary driver 80, and a control circuit 90.

Primary bridge circuit 20 includes four switching elements Q _A is on-off controlled by the control circuit _90, Q B, _Q _C, has a full bridge circuit that is configured to include a Q _D. The secondary side bridge circuit 70 is a full bridge type circuit including four switch elements controlled on / off by the control circuit 90. Secondary bridge circuit 70 includes two switching elements Q _E which corresponds to the low side of the two switching elements of the four switching _elements, and Q _F, the two switch elements of the high-side of the four switching elements It has two corresponding switch elements Q _G and Q _H.

The switch elements Q _A , Q _B , Q _C , Q _D , Q _E , Q _F , Q _G , and Q _H are composed of switch elements such as MOSFETs. Switching element _{_{_{_{Q A, Q B, Q C}}}} , Q D, Q E, Q F, Q G, Q H , the body diode _{D QA} connected in parallel so as to be biased with respect to the terminal _T _{11, T 21} , D _QB , D _QC , D _QD , D _QE , D _QF , D _QG , D _QH . Primary bridge circuit 20 includes the switching elements _{_{_{Q A, Q B, Q C}}} , the capacitor _C A connected in parallel to _{_{_{Q D, C B, C C}}} , the _{C D.} The secondary side bridge circuit 70 has _{capacitors C E} , C _F , C _G , and C _H connected in parallel to _{each switch element Q E} , Q _F , Q _G , and Q _H.

The switch elements Q _A , Q _B , Q _C , Q _D , Q _E , Q _F , Q _G , and Q _H may be composed of, for example, an RC-IGBT. In this case, the FRD built in the RC-IGBT substitutes for the body diode described above. The switch elements Q _A , Q _B , Q _C , Q _D , Q _E , Q _F , Q _G , and Q _H may be composed of, for example, FETs including a wide-gap semiconductor (SiC, GaN). In this case, the switch elements Q _A , Q _B , Q _C , Q _D , Q _E , Q _F , Q _G , and Q _H have the above-mentioned body diodes for the purpose of preventing the decrease in efficiency due to the high forward voltage. An external diode that replaces the above may be connected in parallel.

Two switching elements _Q H, _{Q F} is connected in series with each other at a connection point _{P 3.} Two switching elements _Q G, _{Q E} are connected in series with one another at a connection point _{P 4.} Connection point _{P 4} and the connection point _{P 3} is the secondary coil _{N S,} and the inductor _{L RS} and capacitor _{C RS} is connected via the series.

Primary driver 40 outputs a control signal for performing phase shift control, which is generated by the control circuit 90, the four switching elements _Q A included in the primary bridge circuit _20, Q _B, Q C, to _{Q D} .. Specifically, the primary driver 40, in the Duty range of 0.5 or less generated by the control circuit 90, a signal of fixed frequency and 50% within the pulse width switching element _Q A, the _{Q B} outputs for outputs switching element _Q a, a signal obtained by shifting an arbitrary phase that is determined by the Duty respect _{Q B} switching element _Q C, to _{Q D.} On the other hand, the primary side driver 40 has a fixed frequency and within 50% _{of the switch elements Q A} , Q _D and the switch elements Q _B , Q _{C in the} range where the duty generated by the control circuit 90 exceeds 0.5. Outputs a signal that turns on and off alternately with the pulse width of.

The secondary driver 80 outputs a control signal for PWM control generated by the control circuit 90 to the two switch elements Q _E and Q _F included in the secondary bridge circuit 70. Specifically, in the range where the Duty generated by the control circuit 90 is 0.5 or less, the secondary driver 80 outputs a signal having a fixed frequency and a pulse width of 50% or less determined by the Duty to two switch elements. Output to Q _E and Q _F. On the other hand, in the range where the Duty generated by the control circuit 60 exceeds 0.5, the secondary driver 80 outputs a signal having a pulse width exceeding 50% at a fixed frequency to the two switch elements Q _E and Q _F. do. Here, in the case of operating the secondary bridge circuit 70 as a rectifier circuit, two switch elements Q _G, does not output the control signal to the Q _H, two switching elements Q _G, the body diode or external Q _H Rectification is performed using a diode.

The control circuit 90 is configured by using a DSP, and controls the switch elements of the primary side bridge circuit 20 and the secondary side bridge circuit 70 via the primary side driver 40 and the secondary side driver 80. Power conversion is performed from the primary side to the secondary side (power row = charge mode) or from the secondary side to the primary side (regeneration = discharge mode). That is, the control circuit 90 performs bidirectional power conversion. Here, a case where power conversion is performed from the primary side to the secondary side will be described. The control circuit 90 detects the output voltage (secondary voltage _{V S),} generates a Duty as the output voltage (secondary voltage _{V S)} becomes the target voltage value. When the duty generated at this time is 50% or less, the control circuit 60 mainly uses the phase shift control signal output to the primary side driver 40 as the output voltage of the DC-DC converter 1 (secondary side voltage V). _S ) is controlled. For example, the smaller the generated Duty, the smaller the phase shift amount, and a lower output voltage can be obtained. On the other hand, when the generated duty exceeds 50%, the control circuit 60 mainly uses the PWM control signal output to the secondary side driver 80 as the output voltage of the DC-DC converter 2 (secondary side voltage _VS). ) Is controlled. Specifically the larger Duty generated, the two switching elements Q _E, the time width Q _F is turned on simultaneously increases, at this time, since the inductor L _RP, the energy stored in L _RS increases, higher The output voltage can be obtained.

FIG. 12 shows an example of the relationship between the duty ratio of the drive pulse of the DC-DC converter 2 and the peak value of the current flowing through the primary resonance circuit. As shown in FIG. 12, in the DC-DC converter 2, the duty ratio and the current peak value do not have a positive correlation. Therefore, it is difficult to apply peak current control in order to suppress the demagnetization of the transformer. _{Therefore, the DC-DC converter 2 is provided with capacitors C RP} and C _RS for the resonance circuits on the primary side and the secondary side, so that the capacitors C _RP and C _RS play a role of preventing demagnetization of the transformer 10. I am in charge.

FIG. 13 shows an example of an equivalent circuit of the resonance circuit on the primary side and the secondary side in the DC-DC converter 2. In the DC-DC converter 2, the impedance Z ₁ of the resonant circuit of the primary side _(f _R) and, of the circuit impedance conversion by the turns ratio of the transformer 10 the resonant circuit of the secondary side to the primary side in Bee Dance Z ₂ Each constant of the resonance circuit on the primary side and the secondary side is designed so that'(f _{R) becomes 0Ω as shown in the following equations (1) and (2).} As a result, L _RP = L _RS ', so that the primary side inductor L _RP and the secondary side inductor L _RS can also utilize the leakage inductance of the transformer.
Z ₁ (f _R ) = j (ω _R L _RP -1 / (ω _R C _RP )) = 0 Ω ... (1)
_{_{Z 2 '(f R) =}} j (ω R L RS' -1 / (ω R C RS ')) = 0Ω ... (2)

Figure 14 illustrates an example of the relationship between the resonant frequency f _R and the output voltage (secondary voltage V _S) at the time of the charging mode of the DC-DC converter 2. Figure 15 is a diagram showing an example of the relationship between the resonance frequency f _R and the output voltage (primary voltage V _P) in the discharge mode of the DC-DC converter 2. In FIG. 15, Vp (std) is a reference output voltage output when the Duty is 0.5. In the DC-DC converter 2, the _{primary side and the secondary side and the secondary side so that the impedances Z 1} (f _R ) and Z ₂ '(f _R ) become 0 Ω as shown in the above equations (1) and (2). If the constants of the resonance circuit side is designed, in charge mode of the DC-DC converter 2 and the discharge mode, it can be seen that the relationship between the output voltage resonance frequency f _R are identical.

[motion]
Next, the operation of the DC-DC converter 2 will be described. An example of the drive pulse waveform in the step-down mode (Duty ≦ 0.5) in the DC-DC converter 2 is the waveform shown in FIG. 3 (Duty ≦ 0.5 in the DC-DC converter 1). It is common with an example of the drive pulse waveform). Further, an example of the drive pulse waveform in the boost mode (Duty> 0.5) in the DC-DC converter 2 is the waveform shown in FIG. 6 (Duty> 0.5 in the DC-DC converter 1). It is common with an example of the drive pulse waveform).

FIG. 16 shows an example of driving the DC-DC converter 2 in the period T1 (transmission mode) of FIG. FIG. 17 shows an example of driving the DC-DC converter 2 in the period T2 (reflux mode) of FIG. 16, 17, together with the capacitor _{C P} is connected to the terminal _T _{11, T 12,} a power supply system battery BT1 is connected as an example of the DC power supply is illustrated. Further, FIG. 16, 17, together with the capacitor _{C S} is connected to the terminal _T _{21, T 22,} a power supply system battery BT2 is connected as an example of a DC load device is illustrated. In the present embodiment, both the batteries BT1 and BT2 are assumed to be secondary batteries.

The control circuit 90 performs a step-down operation by, for example, controlling the ratio between the length of the transmission mode period and the length of the reflux mode period. For example, as shown in FIGS. 3 and 16, the control circuit 90 _{turns on the switch elements Q A} , Q _D , and Q _E, and turns off the switch elements Q _B , Q _C , and Q _F. As a result, for example, as shown in FIG. 16, a period (transmission mode) for transmitting electric power from the primary side to the secondary side is generated. Further, the control circuit 90, for example, FIG. 3, as shown in FIG. 17, while to turn on the switching elements Q _A, switching element Q _D, after turning off the Q _E, to turn on the switching element Q _C .. Then, for example, as shown in FIG. 17, a reflux current flows on each of the primary side and the secondary side, and the reflux mode is set.

FIG. 18 shows an example of driving the DC-DC converter 2 in the period T3 (storage mode) of FIG. FIG. 19 shows an example of driving the DC-DC converter 2 in the period T4 (emission mode) of FIG. 18, 19, together with the capacitor _{C P} is connected to the terminal _T _{11, T 12,} battery BT1 is connected as an example of the DC power supply. Further, FIG. 18, 19, together with the capacitor _{C S} is connected to the terminal _T _{21, T 22,} the battery BT2 is connected as an example of a DC load. In the present embodiment, both the batteries BT1 and BT2 are assumed to be secondary batteries.

The control circuit 90 performs a boosting operation by, for example, controlling the ratio between the length of the period of the storage mode and the length of the period of the emission mode. For example, as shown in FIGS. 6 and 18, the control circuit 90 _{turns on the switch elements Q A} , Q _D , Q _E , and Q _F and turns off the switch elements Q _B , Q _{C in the storage mode.} .. As a result, for example, as shown in FIG. 18, it operates as a period for storing energy _{in the inductors L RP} and L _{RS (storage mode).} Further, as shown in FIGS. 6 and 19, for example, the control circuit 60 turns off the _{switch element Q F} while keeping the switch elements Q _A , Q _D , and Q _{E on.} Then, for example, as shown in FIG. 19, it operates as a period during which the energy stored in the _{inductor L RP is released (release mode).}

[effect]
Next, the effect of the DC-DC converter 2 according to the present embodiment will be described.

In the present embodiment, the switch elements of the primary side bridge circuit 20 and the secondary side bridge circuit 70 are driven by different control methods in the high output voltage region and the low output voltage region in the output voltage range of the DC-DC converter 2. By doing so, bidirectional power conversion is performed. Thereby, for example, when PFM control is applied only to the primary side bridge circuit to control the output voltage (for example, see FIG. 9), or PWM control is applied only to the secondary side bridge circuit to control the output voltage. (e.g., see FIG. 10) can be the case of (hereinafter referred to as "control method according to a comparative example.") and compared, reduce the amount of energy stores in resonant inductor L _R. As a result, the magnetic component can be miniaturized. Therefore, the volume of the DC-DC converter 2 can be reduced.

Further, in the present embodiment, in the high output voltage region in the output voltage range of the DC-DC converter 2, PWM control for the secondary bridge circuit 70 is performed as the main control, and in the output voltage range of the DC-DC converter 2. In the low output voltage region, phase shift control is performed as the main control for the primary bridge circuit 20. Thus, as compared with the control method of the comparative example, reducing the amount of energy stores in resonant inductor L _R. As a result, the magnetic component can be miniaturized. Therefore, the volume of the DC-DC converter 2 can be reduced.

Further, in the present embodiment, the reference output voltage _{(VS (std)} ) determined according to the turns ratio of the transformer 10 is set to be the center of the output voltage range of the DC-DC converter 2. Thus, as compared with the control method of the comparative example, reducing the amount of energy stores in resonant inductor L _R. As a result, the magnetic component can be miniaturized. Therefore, the volume of the DC-DC converter 2 can be reduced.

Further, in the present embodiment, the PWM control for the secondary side bridge circuit 70 is performed as the main control in the operating region where the Duty exceeds 0.5, and the primary side bridge is performed in the operating region where the Duty is 0.5 or less. The phase shift control for the circuit 20 is performed as the main control. Thus, as compared with the control method of the comparative example, reducing the amount of energy stores in resonant inductor L _R. As a result, the magnetic component can be miniaturized. Therefore, the volume of the DC-DC converter 2 can be reduced.

Further, in the present embodiment, the primary side bridge has a fixed drive frequency f _SW _{having a value larger than the resonance frequency f R of the} resonance circuit obtained by converting the LC resonance circuit on the secondary side to the primary side. The switch elements of the circuit 20 and the secondary bridge circuit 70 are controlled. As a result, the circuit current flowing through the DC-DC converter 2 at Duty = 0.5 is in continuous mode. As a result, the circuit current becomes small, so that the charging efficiency becomes higher than in the discontinuous mode.

Further, in the present embodiment, the impedance Z ₁ of the resonant circuit on the primary side of the transformer 10 _(f _R), and the impedance Z ₂ of the circuit the resonant circuit on the secondary side and impedance conversion on the primary side of the transformer 10 ' Each constant of the resonance circuit on the primary side and the secondary side is designed so that (f _{R) becomes 0Ω as shown in the above equations (1) and (2).} Thus, in charge mode of the DC-DC converter 2 and the discharge mode, it is possible to match the relationship between the resonant frequency f _R and the output voltage (secondary voltage V _S). _{Further, since L RP} = L _RS'in FIG. 13, the primary side inductor L _RP and the secondary side inductor L _RS can also utilize the leakage inductance of the transformer.

Further, in this embodiment, the primary bridge circuit 20, a plurality of switching elements Q _A is on-off controlled by the control circuit _90, Q B, _Q _C, full-bridge, which is configured to include a Q _D It is a circuit of. Further, the secondary side bridge circuit 70 is also a full bridge type circuit composed of _{a plurality of switch elements Q E} , Q _F , Q _G , and Q _{H controlled on / off by the control circuit 90.} .. Thus, by controlling the above switching element, as compared with the control method of the comparative example, reducing the amount of energy stores in resonant inductor L _R. As a result, the magnetic component can be miniaturized. Therefore, the volume of the DC-DC converter 2 can be reduced.

Further, in the present embodiment, as shown in FIG. 12, the duty ratio and _{the peak value of the primary resonance circuit current IRP} do not have a positive correlation. Therefore, it is difficult to apply peak current control in order to suppress the demagnetization of the transformer. _{Therefore, the DC-DC converter 2 is provided with capacitors C RP} and C _RS for the resonance circuits on the primary side and the secondary side, so that the capacitors C _RP and C _RS play a role of preventing demagnetization of the transformer 10. I am in charge.

In the second embodiment, consider the case of driving the switching element Q _G, Q _H. Since the pulse width requirements required for the switch elements Q _G and Q _H are the same, the description here focuses on the drive of the _{switch elements Q H.} In the Duty ≦ 0.5 (phase shift mode), for example, as shown in FIGS. 20 and 21, the control circuit 90 increases the period in which the current flows through the rectifier circuit as the Duty increases. , the width of the driving pulse to be applied to the switching element Q _H (oN width) must also be increased. However, in the Duty> 0.5 (boost PWM mode), as Duty increases, as shown by the solid line in FIG. 22, is necessary to reduce the width of the drive pulse applied to the switching element _{Q H} (ON width) be. Focusing on the rising edge of the drive pulse of the switching element Q _H (ON timing), the case of not shifting the rising position of the driving pulses of the switching element Q _H to the right (when not delayed on-timing), as indicated by the broken line switches It overlaps with the on of the element Q _F _{, and the capacitor CS} is short-circuited. Mata, in the falling edge of the drive pulse switching element Q _H Focusing on (off timing), for example, the falling position of the switch element Q _H (Not found advancing the off timing) When not shifted to the left, the switch element Q since overlaps the on the _E, as shown in FIG. 23, current is flowing back to the capacitor C _P, loss occurs. Therefore, a short circuit and such a capacitor C _S, the measures to prevent the reverse flow of current to the capacitor C _P is described.

<3. Third Embodiment>
[composition]
FIG. 24 shows a circuit diagram of the DC-DC converter 3 according to the third embodiment of the present invention. The DC-DC converter 3 includes one inductor L _RP and one capacitor C _RP as the resonance circuit on the primary side, and one inductor L _RS and one capacitor as the resonance circuit on the secondary side. It is equipped with C _RS. The DC-DC converter 3 is a CLLC type DC-DC converter that utilizes resonance caused by _{these inductors L RP} and L _RS and capacitors C _RP and C _RS. DC-DC converter 3 comprises a transformer 10, 1-side bridge circuit 20, a secondary-

side bridge circuit

70,1 primary driver 40 and a secondary-side driver 80, the current sensor _S P, _{S S} and the control circuit 100 There is.

Current sensor S _P output primary side detects the circuit current I _P flowing through the high side of the primary bridge circuit 20, and outputs a detection value corresponding to the detected circuit current I _P to the control circuit 100. Current sensor S _S of the secondary side detects the circuit current I _S flowing through the high side of the secondary bridge circuit 70, and outputs a detection value corresponding to the detected circuit current I _S to the control circuit 100. Control circuit 100, based on the primary side of the current sensor _{S P} and the secondary side of the current sensor _S detected values obtained from the _S _(I P, _{I S)} with a control signal generated within the control circuit 100 Then, a synchronous rectification signal is output to the switch element on the high side of the secondary side bridge circuit 70.

FIG. 25 shows an example of the circuit configuration of the synchronous rectification pulse generation circuit 101, which is incorporated in the control circuit 100 and is output to the switch element on the high side side of the rectification side bridge during charging. The synchronous rectification pulse generation circuit 101 includes, for example, a comparator and two AND circuits. Comparator detects the circuit current I _S of the secondary bridge circuit 70. The AND circuit synthesizes the control signal generated by the control circuit 100 and the output signal of the comparator. Thus, while synchronously rectifying the high-side switching element on the secondary side bridge, short circuit and the capacitor C _S, backflow of current to the capacitor C _P is prevented.

As an example, a method of generating a synchronous rectified signal of the switch element on the high side side of the rectified side bridge during the charging operation will be described. In Figure 25, switch when one of the AND circuit outputs a gate pulse to the switch element Q _G when a negative voltage is applied to the transformer 10, the other AND circuits which positive voltage is applied to the transformer 10 except that outputs a gate pulse to the element Q _H, for the method of generating the control signal are common, the description will focus on aND circuit connected to the gate of the switching element Q _H. The detection value ( _IS ) and the current threshold value ( _ITH ) are input to the comparator, _{and the comparison result between the detection value (IS} ) and the current threshold value ( _ITH ) is output from the comparator as a COMP signal. .. In the AND circuit, (1) a control signal input to the _{switch element Q A} _{, (2) a control signal input to the switch element Q D} , (3) a COMP signal, and (4) input to the _{switch element Q F.} a control signal, the switch element _Q F, a total of four types of signals of the inverted signal of the dead time _{T FH} and the sum signal provided as a short circuit prevention between the _{_{_{Q H (INV (Q F +}}} T FH)) is input, the output signal of the aND circuit is output to the switch element _{Q H.} Here, the switch elements Q _A and Q _D are used to determine the polarity of the voltage applied to the transformer 10, and the COMP signal is used to detect the section in which the current is flowing in the switch element. The signal (INV (Q _F + _TFH )) is used for the purpose of preventing a short circuit between the upper and lower arms.

FIG. 26 shows an example of the drive pulse waveform in Duty ≦ 0.5 (step-down mode). FIG. 27 shows an example of the drive pulse waveform when Duty> 0.5 (boost mode). Incidentally, FIG. 26, the waveform measurement of FIG. 27, with the capacitor _{C P} is connected to the terminal _T _{11, T 12,} battery BT1 is connected as an example of the DC power supply. Further, FIG. 26, the waveform measurement of FIG. 27, with the capacitor _{C S} is connected to the terminal _T _{21, T 22,} the battery BT2 is connected as an example of a DC load. At this time, both the batteries BT1 and BT2 are assumed to be secondary batteries.

During the charging operation, the synchronous rectification pulse generation circuit 101 outputs a synchronous rectification signal to the switch element on the high side side of the secondary side bridge circuit 70 according to the control by the control circuit 100. Synchronous pulse generator 101 is, for example, in the period of FIG. 26 T1 (Duty ≦ 0.5 (Buck Mode)), the switch element _{Q H,} 2-side bridge control signal for turning on the switching element _{Q H} Output to the circuit 70. At this time, synchronous rectification pulse generating circuit 101, the Duty is increased, to increase the pulse width of the control signal for turning on the switching element Q _H (ON width). Synchronous pulse generator 101 is, for example, in the period of FIG. 27 T1 (Duty> 0.5 (boost mode)), the switch element _{Q H,} 2-side bridge control signal for turning on the switching element _{Q H} Output to the circuit 70. At this time, synchronous rectification pulse generating circuit 101, the Duty is increased, narrowing the pulse width of the control signal for turning on the switching element Q _H (ON width).

[effect]
In this embodiment, a current sensor for detecting a current sensor S _P for detecting the circuit current I _P flowing through the high side of the primary bridge circuit 20, the circuit current I _S flowing through the high side of the secondary bridge circuit 70 and the S _S are provided. These current sensors _S P, _S circuit current detected by the _S _I P, based on the _{I S,} synchronous rectification on the secondary side bridge circuit 70 is performed. Thus, for example, when the Duty is greater than 0.5, the switch element _Q H, by narrowing the pulse width of the control signal for turning on the _{Q G} (ON width), short circuit and the capacitor _{C S,} it is possible to prevent the reverse flow of current to the capacitor C _P. For example, when Duty is 0.5 or less, by widening the switching element Q _H, pulse width of the control signal for turning on the Q _G (ON width) depending on the period in which current is flowing, the capacitor C _P Efficiency can be improved by preventing backflow of current to.

Although the present disclosure has been described above with reference to a plurality of embodiments, the present disclosure is not limited to each of the above embodiments and can be modified in various ways. In each of the above embodiments, an example is described in which the primary bridge is variably controlled by phase shift when Duty ≦ 0.5, but PWM control with Dmax = 0.5 may be used. The effects described in this specification are merely examples. The effects of the present disclosure are not limited to the effects described herein. The present disclosure may have effects other than those described herein.

Claims

With a transformer
The primary side bridge circuit provided on the primary side of the transformer and
The secondary side bridge circuit provided on the secondary side of the transformer and
A resonance circuit provided on at least one of the primary side and the secondary side of the transformer, and
It is provided with a control circuit that enables unidirectional or bidirectional power conversion by driving the switch elements of the primary side bridge circuit and the secondary side bridge circuit by different control methods depending on the output voltage range. Power supply.
The control circuit controls the output voltage in the high output voltage region in the output voltage range with the PWM control for the secondary bridge circuit as the main control, and in the low output voltage region in the output voltage range, the primary control circuit controls the output voltage. The power supply device according to claim 1, wherein the output voltage is controlled with the phase shift control for the side bridge circuit as the main control.
The power supply device according to claim 1, wherein the reference output voltage determined according to the turns ratio of the transformer is set at the center of the output voltage range.
The control circuit performs PWM control for the secondary bridge circuit as the main control in an operating region where the Duty exceeds 0.5, and in an operating region where the Duty is 0.5 or less, the phase with respect to the primary bridge circuit is performed. The power supply device according to claim 1, wherein shift control is performed as main control.
The first aspect of claim 1, wherein the control circuit controls the switch element with a fixed drive frequency having a value larger than the resonance frequency of the resonance circuit obtained by converting the LC resonance circuit on the secondary side to the primary side. Power supply.
The resonance circuit is configured such that the impedance of the resonance circuit on the primary side of the transformer and the impedance of the circuit obtained by impedance-converting the resonance circuit on the secondary side of the transformer to the primary side are both zero at the resonance frequency. The power supply device according to claim 1.
The primary side bridge circuit is a full bridge type circuit including a plurality of switch elements controlled on / off by the control circuit.
The secondary side bridge circuit is a full bridge type circuit including a plurality of low-side switch elements controlled on / off by the control circuit and a plurality of high-side diode elements. The power supply described in.
The primary side bridge circuit is a full bridge type circuit including a plurality of switch elements controlled on / off by the control circuit.
The power supply device according to claim 1, wherein the secondary side bridge circuit is a full bridge type circuit including a plurality of switch elements controlled on / off by the control circuit.
A resonance circuit is further provided on at least the primary side of the primary side and the secondary side of the transformer.
The power supply device according to claim 8, wherein both the resonance circuit on the primary side of the transformer and the resonance circuit on the secondary side of the transformer include an inductor and a capacitor.
The primary side current sensor that detects the circuit current pulse of the primary side bridge circuit, and
It is further equipped with a secondary side current sensor that detects the output current pulse of the secondary side bridge circuit.
The control circuit performs synchronous rectification of the secondary side bridge circuit based on the output current pulse detected by the secondary side current sensor at the time of powering.
The power supply device according to claim 1, wherein the control circuit performs synchronous rectification of the primary side bridge circuit based on the output current pulse detected by the primary side current sensor at the time of regeneration.
A power supply that converts DC power,
A first battery connected to the primary side of the power supply unit and
A second battery connected to the secondary side of the power supply unit is provided.
The power supply unit
With a transformer
The primary side bridge circuit provided on the primary side of the transformer and
The secondary side bridge circuit provided on the secondary side of the transformer and
A resonance circuit provided on at least one of the primary side and the secondary side of the transformer, and
A power supply having a control circuit that enables unidirectional or bidirectional power conversion by driving the switch elements of the primary side bridge circuit and the secondary side bridge circuit by different control methods depending on the output voltage range. system.