WO2021136046A1 - 一种光互连通信方法及系统 - Google Patents

一种光互连通信方法及系统 Download PDF

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WO2021136046A1
WO2021136046A1 PCT/CN2020/138778 CN2020138778W WO2021136046A1 WO 2021136046 A1 WO2021136046 A1 WO 2021136046A1 CN 2020138778 W CN2020138778 W CN 2020138778W WO 2021136046 A1 WO2021136046 A1 WO 2021136046A1
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optical
polarization
signal
local oscillator
oscillator light
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PCT/CN2020/138778
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English (en)
French (fr)
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冯振华
郑林
胡杰
胡烽
朱齐雄
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烽火通信科技股份有限公司
武汉飞思灵微电子技术有限公司
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Publication of WO2021136046A1 publication Critical patent/WO2021136046A1/zh

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    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B10/00—Transmission systems employing electromagnetic waves other than radio-waves, e.g. infrared, visible or ultraviolet light, or employing corpuscular radiation, e.g. quantum communication
    • H04B10/50—Transmitters
    • H04B10/516—Details of coding or modulation
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B10/00—Transmission systems employing electromagnetic waves other than radio-waves, e.g. infrared, visible or ultraviolet light, or employing corpuscular radiation, e.g. quantum communication
    • H04B10/50—Transmitters
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B10/00—Transmission systems employing electromagnetic waves other than radio-waves, e.g. infrared, visible or ultraviolet light, or employing corpuscular radiation, e.g. quantum communication
    • H04B10/50—Transmitters
    • H04B10/501—Structural aspects
    • H04B10/503—Laser transmitters
    • H04B10/505—Laser transmitters using external modulation
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B10/00—Transmission systems employing electromagnetic waves other than radio-waves, e.g. infrared, visible or ultraviolet light, or employing corpuscular radiation, e.g. quantum communication
    • H04B10/50—Transmitters
    • H04B10/501—Structural aspects
    • H04B10/503—Laser transmitters
    • H04B10/505—Laser transmitters using external modulation
    • H04B10/5055—Laser transmitters using external modulation using a pre-coder
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B10/00—Transmission systems employing electromagnetic waves other than radio-waves, e.g. infrared, visible or ultraviolet light, or employing corpuscular radiation, e.g. quantum communication
    • H04B10/50—Transmitters
    • H04B10/501—Structural aspects
    • H04B10/503—Laser transmitters
    • H04B10/505—Laser transmitters using external modulation
    • H04B10/5059—Laser transmitters using external modulation using a feed-forward signal generated by analysing the optical or electrical input
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B10/00—Transmission systems employing electromagnetic waves other than radio-waves, e.g. infrared, visible or ultraviolet light, or employing corpuscular radiation, e.g. quantum communication
    • H04B10/50—Transmitters
    • H04B10/516—Details of coding or modulation
    • H04B10/532—Polarisation modulation
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B10/00—Transmission systems employing electromagnetic waves other than radio-waves, e.g. infrared, visible or ultraviolet light, or employing corpuscular radiation, e.g. quantum communication
    • H04B10/60—Receivers
    • H04B10/61—Coherent receivers
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B10/00—Transmission systems employing electromagnetic waves other than radio-waves, e.g. infrared, visible or ultraviolet light, or employing corpuscular radiation, e.g. quantum communication
    • H04B10/60—Receivers
    • H04B10/61—Coherent receivers
    • H04B10/614—Coherent receivers comprising one or more polarization beam splitters, e.g. polarization multiplexed [PolMux] X-PSK coherent receivers, polarization diversity heterodyne coherent receivers
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B10/00—Transmission systems employing electromagnetic waves other than radio-waves, e.g. infrared, visible or ultraviolet light, or employing corpuscular radiation, e.g. quantum communication
    • H04B10/60—Receivers
    • H04B10/61—Coherent receivers
    • H04B10/615—Arrangements affecting the optical part of the receiver
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B10/00—Transmission systems employing electromagnetic waves other than radio-waves, e.g. infrared, visible or ultraviolet light, or employing corpuscular radiation, e.g. quantum communication
    • H04B10/60—Receivers
    • H04B10/61—Coherent receivers
    • H04B10/616—Details of the electronic signal processing in coherent optical receivers
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04J—MULTIPLEX COMMUNICATION
    • H04J14/00—Optical multiplex systems
    • H04J14/06—Polarisation multiplex systems
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L1/00—Arrangements for detecting or preventing errors in the information received
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L1/00—Arrangements for detecting or preventing errors in the information received
    • H04L1/004—Arrangements for detecting or preventing errors in the information received by using forward error control

Definitions

  • the invention relates to the field of high-speed optical interconnection in data centers, in particular to an optical interconnection communication method and system.
  • the network traffic load is from the long-distance backbone.
  • the network shifts to short- and medium-range metropolitan area networks and data centers.
  • data flow pressure is more obvious. It is estimated that by 2021, 94% of the business will be processed in the cloud data center, but in the data center-related network, more than 70% of the traffic will be terminated in the data center.
  • short-distance less than 2km
  • high-speed optical interconnection technology will play an important role in future big data transmission and bearing.
  • EML Electro-absorption Modulated Laser
  • PAM4 Pulse Amplitude Modulation, Level 4
  • Pulse amplitude modulation modulation technology
  • the second category is to use a simplified 3x3 coupler to replace the traditional 90° coherent mixer in order to reduce the cost of some optical components.
  • the local oscillator light and signal light are transmitted together from the originating end to the receiving end.
  • Some methods such as space division multiplexing fan-in and fan-out, or polarization separation techniques are used to separate the local oscillator light from the signal.
  • Coherent mixing is performed after optical separation, which can reduce the number and cost of lasers, and also simplify some of the receiving end DSP (Digital Signal Processing, digital signal processing) algorithm [Optics Express, vol. 21, no. 2, investigating self -homodyne coherent detection in a 19 channel space-division-multiplexed transmission link].
  • DSP Digital Signal Processing, digital signal processing
  • the traditional coherent optical communication for long-distance transmission is used for the transmitting end DSP processing flow and the receiving end DSP processing flow. Both include the sampling rate conversion part, such as the up-sampling and The re-sampling in Figure 2 is not only complicated in structure, but also consumes a lot of power because it works at a high sampling rate (greater than the baud rate).
  • the dispersion compensation at the receiving end, multi-tap adaptive equalization, frequency offset estimation, and phase recovery are computationally complex, occupying most of the power consumption of the chip.
  • this traditional DSP architecture is not directly applicable to low power consumption. Cost and low power consumption in the data center short-distance optical interconnection system.
  • the modulation format of the signal is the common polarization multiplexing phase modulation or amplitude phase modulation, such as PDM (Polarization Division Multplexed, polarization multiplexing)-PSK (Phase Shift Keying, phase shift keying), PDM-QPSK (Quadrature Phase Shift Keying) , Quaternary phase shift keying), PDM-8QAM (8-ary Quadrature Amplitude Modulation, octal quadrature amplitude phase modulation), PDM-16QAM (16-ary Quadrature Amplitude Modulation, hexadecimal quadrature amplitude phase modulation) ), PDM-64QAM (64-ary Quadrature Amplitude Modulation, 64-ary Quadrature Amplitude Phase Modulation), etc.
  • PDM Polarization Division Multplexed, polarization multiplexing
  • PDM-QPSK Quadadrature Phase Shift Keying
  • PDM-8QAM 8-ary Quadrature Amplitude Modulation, o
  • the purpose of the present invention is to provide an optical interconnection communication method and system, which is suitable for multiple modulation formats and multiple rates, and can also reduce cost and power consumption.
  • an optical interconnection communication method including:
  • the binary bit stream signal to be transmitted is processed by DSP into an analog electric signal; a laser is used to output two paths of light, one of which is used as the optical carrier of the integrated dual-polarization coherent optical transmitter to modulate the analog electric signal into a complex optical field signal, The other is used as the local oscillator for time delay adjustment;
  • the complex optical field signal and the local oscillator light are coherently mixed by a polarization-independent coherent optical receiver, wherein the polarization state of the local oscillator light is controlled by feedback scrambling to prevent the polarization state of the local oscillator light from falling into X and Y directions; the mixed electric signal is processed by the DSP to restore the binary bit stream signal.
  • the digital signal processing of the binary bit stream signal to be transmitted includes:
  • the X and Y polarization states are mapped to independent constellation maps to generate two independent complex signal streams, and then perform polarization diversity precoding.
  • the two pre-coded signals After quadrature separation, time-domain pre-compensation is performed, and the compensated signal is digital-to-analog converted at the sampling rate of the baud rate to obtain the analog electrical signal.
  • the complex signal stream generated by the constellation map mapping is [X 1 , Y 1 ] T
  • the precoding symbol stream is [X 2 , Y 2 ] T
  • the digital signal processing of the mixed electric signal includes:
  • the mixed electrical signal is analog-to-digital converted into a digital signal and subjected to anti-aliasing filtering.
  • the filtered signal is subjected to feedforward equalization and clock synchronization, and then performs single-tap adaptive equalization to achieve polarization demultiplexing, and then performs de-precoding, Demapping and FEC decoding to obtain the binary bit stream signal.
  • the two signals obtained by the polarization demultiplexing are [A 1 , B 1 ] T
  • [A 1 , B 1 ] T is subjected to depolarization diversity precoding processing using a matrix to obtain [A 2 , B 2 ] T
  • the feedback scrambling to control the polarization state of the local oscillator light includes:
  • the polarization-independent coherent optical receiver divides the received local oscillation light into two branches with polarization states perpendicular to each other.
  • the local oscillation light of each branch is divided into two according to different power ratios, and each branch is extracted.
  • the optical power of the local oscillator is the one with the smaller optical power, and the extracted power difference of the two local oscillators is obtained, and the power difference is converted into the photocurrent amplitude.
  • the absolute value of the photocurrent amplitude is greater than the preset value At the threshold value, the local oscillator light is disturbed by a control signal proportional to the magnitude of the absolute value.
  • the applicable modulation format of the optical interconnection communication method includes QPSK, 8QAM, 16QAM, 32QAM and 64QAM;
  • the applicable information rate of the optical interconnection communication method includes 100G, 200G, 400G, 600G, and 800G.
  • an optical interconnection communication system including:
  • the first optical splitter is configured to receive the continuous light and divide it into two paths, one path provides an optical carrier, and the other path serves as a local oscillator;
  • the sending DSP chip is used to convert the binary bit stream signal to be transmitted into an analog electric signal after being processed by the DSP;
  • Integrated dual-polarization coherent optical transmitter for receiving the optical carrier and modulating the analog electrical signal into a complex optical field signal
  • the first optical fiber channel is used to transmit the complex optical field signal output by the integrated dual-polarization coherent optical transmitter
  • Coarse adjustable delay line used to roughly adjust the delay of the local oscillator optical transmission line to match the length of the complex optical field signal transmission link;
  • the second optical fiber channel is used to transmit the local oscillator light adjusted by the coarsely adjustable delay line;
  • Polarization-independent coherent optical receiver for receiving the complex optical field signal and the local oscillator light respectively transmitted by the first optical fiber channel and the second optical fiber channel and performing coherent mixing;
  • the polarization-independent coherent optical receiver Including a polarization scrambler, which is used to control the polarization state of the local oscillator light through feedback scrambling to prevent the polarization state of the local oscillator light from falling into the X and Y directions;
  • the receiving DSP chip is used to receive the mixed electric signal, and restore the binary bit stream signal through DSP processing.
  • the laser is a DFB laser; the splitting ratio of the first optical splitter 2 is 7:3; the coarse and adjustable delay line is realized by a single-mode fiber; the first fiber channel and the second fiber channel are both ordinary single-mode fiber .
  • the polarization-independent coherent optical receiver includes:
  • the first polarization beam splitter is used to divide the complex optical field signal into two branches whose polarization states are perpendicular to each other;
  • the finely adjustable delay line is used to precisely adjust the transmission delay and optical path difference of the local oscillator light relative to the signal light to ensure that the signal light and the local oscillator light meet the coherence length;
  • the second polarization beam splitter is used to divide the local oscillator light into two branches whose polarization states are perpendicular to each other;
  • the polarization scrambler which receives the local oscillator light adjusted by the finely adjustable delay line, controls the polarization state of the local oscillator light input to the second polarization beam splitter, and avoids that the local oscillator light entering the second polarization beam splitter is exactly Horizontal or vertical direction;
  • the second optical splitter and the third optical splitter are used to split the two local oscillator lights with polarization states in different power ratios
  • the fifth balanced detector is used to receive the local oscillator light with a smaller power split by the second optical splitter and the third optical splitter respectively, obtain the power difference, and convert it into a photocurrent amplitude output;
  • a controller used for converting the photocurrent amplitude into a disturbance control signal for the polarization state of the local oscillator light, and controlling the scrambler;
  • the mixing gain module is used to coherently mix the two complex optical field signals output by the first polarization beam splitter with the higher power local oscillator light from the second optical splitter and the third optical splitter, and then Obtain the real and imaginary parts of the X and Y polarizations respectively, perform amplification and realize automatic gain control.
  • the mixing gain module includes:
  • the first 90° mixer is used to coherently mix a complex optical field signal output by the first polarization beam splitter with a local oscillator light with greater power split by the second optical splitter;
  • the second 90° mixer is used to coherently mix another complex optical field signal output by the first polarization beam splitter with a local oscillator light with a higher power split by the third optical splitter;
  • the first balanced detector and the second balanced detector are used to convert the optical signal output by the first 90° mixer into an electrical signal to obtain the real part and the imaginary part of the X polarization respectively;
  • the third balanced detector and the fourth balanced detector are used to convert the optical signal output by the second 90° mixer into an electrical signal to obtain the real part and the imaginary part of the Y polarization respectively;
  • the transimpedance amplifier is used to amplify the real and imaginary parts of the X polarization and the real and imaginary parts of the Y polarization and realize automatic gain control.
  • the ratio of the second beam splitter and the third beam splitter are the same, both being 95:5;
  • the output of a control electrical signal proportional to the magnitude of the absolute value acts on the scrambler, and the threshold is the first Five balance detectors detect 90% of the maximum photocurrent.
  • the originating DSP chip includes:
  • the FEC encoding module is used to perform FEC encoding on the binary bit stream signal to be transmitted;
  • the constellation mapping module is used to perform independent constellation mapping for the two polarization states of X and Y according to the preset modulation format, and generate two independent complex signal streams [X 1 ,Y 1 ] T ;
  • the precoding module is configured to perform polarization hierarchical precoding on the complex signal stream [X 1 , Y 1 ] T , and output two precoding symbol streams [X 2 , Y 2 ] T , the precoding rule is:
  • [X 2 ,Y 2 ] T H ⁇ [X 1 ,Y 1 ] T , Is the precoding matrix, which satisfies ad-bc ⁇ 0.
  • the pre-compensation module is used to perform time-domain pre-compensation on the two pre-coded signals after being orthogonally separated;
  • the DAC module is used to perform digital-to-analog conversion on the compensated signal at the sampling rate at the baud rate to obtain the analog electrical signal.
  • the receiving DSP chip includes:
  • ADC module which is used to convert the mixed electric signal into a digital signal
  • Two low-pass filters perform anti-aliasing filtering on the X polarization signal and Y polarization signal respectively
  • Two feedforward equalizers each corresponding to a low-pass filter, perform feedforward equalization on the filtered signal
  • the adaptive equalization module performs adaptive equalization on the signal after clock recovery, realizes polarization demultiplexing, and obtains two equalized signals [A 1 , B 1 ] T.
  • the de-precoding module uses a matrix to perform de-polarization diversity pre-coding processing on [A 1 , B 1 ] T to obtain [A 2 , B 2 ] T.
  • Two constellation diagram mapping modules respectively de-map the signals that have completed depolarization diversity precoding
  • Two FEC decoding modules respectively perform FEC decoding on the demapped binary bits to restore the binary bit stream signal.
  • the pre-compensation module of the originating DSP chip uses a real-number domain finite impulse response filter to perform time-domain pre-compensation, and the number of taps of the finite impulse response filter is less than or equal to 3;
  • the adaptive equalization module of the receiving DSP chip adopts a single-tap 2x2 complex butterfly filter to perform adaptive equalization
  • Both the DAC module and the ADC module work at a sampling rate of 1 times the baud rate.
  • the transmitter is provided with a laser.
  • the self-coherent technology is adopted to avoid the use of expensive narrow linewidth lasers at the receiver, and only one laser is used at the transmitter to reduce the cost of the short-distance optical interconnection system.
  • the transmitter uses DFB (Distributed Feedback laser) lasers. Its line width is generally larger, on the order of MHz, and the price is cheaper than ECL (External Cavity Laser), which is generally a narrow line. For wide lasers, the line width is of the order of KHz.
  • ECL External Cavity Laser
  • the line width is of the order of KHz.
  • current commercial coherent optical communication systems all require ECL lasers of around 100 kHz, so the use of DFB lasers further reduces system costs compared to other lasers.
  • the polarization state of the local oscillator light is controlled based on the combination of power balance detection and the polarizer to avoid the risk of demodulation failure when the local oscillator light is exactly X or Y polarization, realizes polarization-independent coherent reception, and solves the self-coherent system Difficulties in the control of the polarization state of the local oscillator.
  • Coarse adjustable delay line and fine adjustable delay line are used on the optical path to adjust the optical path difference between the local oscillator light and the signal light to ensure the matching of the coherence length, thereby avoiding frequency offset estimation and phase recovery at the receiving end, and the maximum Simplify the receiving DSP algorithm and circuit to a great extent.
  • the baud rate sampling rate of the DAC and ADC conversion is adopted, which greatly reduces the power consumption of the coherent DSP chip; on the other hand, a set of minimalist transceiver DSP signal processing procedures are proposed to simplify the DSP The complexity of the chip saves power consumption and area of the chip.
  • polarization diversity precoding in the transmitting DSP chip to artificially introduce correlation between the two polarization signals, it is expected to improve the tolerance to polarization-dependent loss, and to a certain extent solve the performance loss of low-cost devices in coherent systems problem.
  • FFE Forward Feedback Equalizer
  • ISI Inter Symbol Interference
  • different high-level modulation formats can be applied, such as QPSK, 8QAM, 16QAM, 32QAM, 64QAM, etc., and it is also suitable for multiple information rates, such as 100G, 200G, 400G, 600G, 800G, etc.
  • the final beneficial effect achieved by the present invention is that it can realize short-distance optical interconnection with low cost, high speed and low power consumption, which is very suitable for the internal optical interconnection scenarios of data centers that are sensitive to cost and power consumption.
  • Figure 1 is a schematic diagram of the originating DSP processing flow of the traditional coherent optical communication used for long-distance transmission;
  • Fig. 2 is a schematic diagram of the receiving end DSP processing flow diagram of the traditional coherent optical communication used for long-distance transmission;
  • FIG. 3 is a schematic diagram of the structure of a single-tap adaptive equalizer according to an embodiment of the present invention.
  • FIG. 4 is a schematic diagram of an optical interconnection communication system according to an embodiment of the present invention.
  • Figure 5 is a schematic diagram of the realization of a polarization-independent coherent optical receiver
  • Fig. 6 is a schematic diagram of an originating DSP chip according to an embodiment of the present invention.
  • Fig. 7 is a schematic diagram of a receiving DSP chip according to an embodiment of the present invention.
  • FIG. 8 is a flow chart of polarization control of local oscillator light according to an embodiment of the present invention.
  • Figure 9 is a QPSK constellation diagram according to an embodiment of the present invention.
  • FIG. 10 is a constellation diagram of QAM according to Embodiment 8 of the present invention.
  • 51-FEC encoding module 52-constellation map mapping module, 53-pre-encoding module, 54-pre-compensation module, 55-DAC module;
  • 700-mixing gain module 701-first polarization beam splitter, 702-first 90°mixer, 703-first balanced detector, 704-second balanced detector, 705-transimpedance amplifier, 706- The second 90° mixer, 707-third balanced detector, 708-fourth balanced detector, 709-second beam splitter, 710-third beam splitter, 711-second polarization beam splitter, 712-th Five balanced detectors, 713-controller, 714- scrambler, 715-precisely adjustable delay line;
  • 81-ADC module 82-low pass filter, 83-feedforward equalizer, 84-clock recovery module, 85-adaptive equalization module, 86-de-precoding module, 87-constellation diagram mapping module, 88-FEC decoding Module.
  • An embodiment of an optical interconnection communication method which specifically includes:
  • the binary bit stream signal to be transmitted is processed by the DSP into an analog electric signal.
  • a laser is used to provide a DC optical carrier for optical interconnection communication, and it also serves as a local oscillator for coherent detection.
  • a laser is used to output two channels of light, one of which is used as the optical carrier of the integrated dual-polarization coherent optical transmitter to modulate the analog electrical signal obtained by DSP processing into a complex optical field signal; the other is used as the local oscillator for time delay adjustment.
  • a coarsely adjustable delay line can be used to roughly adjust the delay of the local oscillator optical transmission line to match the length of the complex optical field signal transmission link to ensure that the complex optical field signal and the local oscillator are near the coherence length.
  • the above-mentioned complex optical field signal and the local oscillator light are received by the polarization-independent coherent optical receiver, and the coherent mixing is performed, and the modulated signal is moved to the baseband to obtain the baseband electrical signal.
  • the polarization state of the local oscillator light is controlled by feedback scrambling to prevent the polarization state of the local oscillator light from falling into the X and Y directions, thereby causing the loss of data in one of the polarization states, and ensuring the correct demodulation of the polarization multiplexed data.
  • the mixed electric signal is processed by DSP to recover the binary bit stream signal.
  • four electrical signals are obtained after mixing, which are two electrical signals of X polarization and Y polarization, and their respective in phase and quadrature.
  • the above-mentioned control of the polarization state of the local oscillator light by feedback scrambling specifically includes: the polarization-independent coherent optical receiver divides the received local oscillator light into two branches whose polarization states are perpendicular to each other, and the local oscillator light of each branch is in accordance with Different power ratios are divided into two paths, extract the path with the smaller optical power of each branch's local oscillator, and obtain the power difference of the two extracted local oscillators, and convert the power difference into the photocurrent amplitude
  • the absolute value of the photocurrent amplitude is greater than the preset threshold, the local oscillator light is disturbed by a control signal proportional to the magnitude of the absolute value.
  • an embodiment of DSP processing in the sending direction and the receiving direction is further provided.
  • the digital signal processing of the binary bit stream signal to be transmitted specifically includes the following steps:
  • the FEC-encoded data is subjected to independent constellation mapping for the X and Y polarization states (ie, the horizontal and vertical polarization states) to generate two independent complex signal streams [X 1 ,Y 1 ] T.
  • the complex signal stream [X 1 ,Y 1 ] T is then pre-encoded with polarization diversity, and two pre-encoded symbol streams [X 2 , Y 2 ] T are output; the pre-coding rules are:
  • [X 2 ,Y 2 ] T H ⁇ [X 1 ,Y 1 ] T , where, Is a precoding matrix and satisfies ad-bc ⁇ 0.
  • the two pre-coded signals are first orthogonally separated to obtain the corresponding four signals, which represent the real and imaginary parts of the X and Y polarization states, respectively, and then respectively pass through a real number domain FIR (Finite Impulse Response, finite impulse response). Excitation response filter) performs time-domain pre-compensation to adjust the relative time delay between signals and compensate the bandwidth of some optoelectronic devices.
  • FIR Finite Impulse Response, finite impulse response
  • Excitation response filter performs time-domain pre-compensation to adjust the relative time delay between signals and compensate the bandwidth of some optoelectronic devices.
  • the number of FIR taps is not more than three.
  • the four pre-compensated signals are sent to the DAC, and digital-to-analog conversion is performed at the sampling rate at the baud rate to obtain the electric signal to be modulated.
  • the digital signal processing of the mixed electrical signal specifically includes the following steps:
  • the four-channel electrical signals output by the polarization-independent coherent receiver are subjected to analog-to-digital conversion at the baud rate sampling rate to obtain four digital signals.
  • S202 Perform digital anti-aliasing filtering processing on the four channels of digital signals by using a low-pass filter to remove signal spectrum aliasing that may be caused by single sampling.
  • S203 Perform feedforward equalization on the filtered signal to compensate for the influence of inter-symbol interference (ISI) caused by bandwidth limitation and short-distance fiber dispersion, and also to compensate for the time difference between channels;
  • ISI inter-symbol interference
  • Xin, Yin, Xin' and Yin' respectively represent the input and output signal vectors on the two polarization states of the adaptive equalizer X and Y
  • Wxx, Wxy, Wyx and Wyy are butterfly filters, respectively
  • the four sets of tap coefficients of the filter respectively represent the influence of X polarization input on X polarization output, the effect of Y polarization input on X polarization output, the effect of X polarization input on Y polarization output, and the effect of Y polarization input on Y polarization output.
  • single-tap adaptive equalization is performed to realize polarization demultiplexing, and the two equalized signals are [A 1 , B 1 ] T.
  • a single-tap adaptive filter is used to perform adaptive equalization on the clock-recovered signal, such as a single-tap 2x2 complex butterfly filter.
  • the number of taps used in feedforward equalization in S205 is not more than 10, and the tap adaptive update criterion can be constant modulus algorithm (CMA), multi-mode algorithm (MMA), cascaded multi-mode algorithm (CMMA) Or the least mean square error algorithm (LMS).
  • CMA constant modulus algorithm
  • MMA multi-mode algorithm
  • CMMA cascaded multi-mode algorithm
  • LMS least mean square error algorithm
  • the update period of the tap coefficient of the single-tap adaptive filter is not less than every 10 symbol periods.
  • the present invention also provides an embodiment of an optical interconnection communication system, which can implement at least one of the above-mentioned practical methods.
  • the system includes a transmitting end and a receiving end.
  • the transmitting end includes a laser 1, a first optical splitter 2, a coarsely adjustable delay line 3, an integrated dual-polarization coherent optical transmitter 4, a transmitting DSP chip 5, a first optical fiber channel 6 and a second optical fiber Channel 9;
  • the receiving end includes a polarization-independent coherent optical receiver 7 and a receiving end DSP chip 8; the transmitting end and the receiving end are transmitted through two optical fiber channels.
  • the output of the laser 1 is connected to the first optical splitter 2, and the output of the first optical splitter 2 is respectively connected to the integrated dual-polarization coherent optical transmitter 4 and the coarse tunable delay line 3, and the integrated dual-polarization coherent optical transmitter 4 is also connected to the transmitting end
  • the DSP chip 5 is connected to the first fiber channel 6.
  • the other port of the coarse adjustable delay line 3 is connected to the second optical fiber channel 9.
  • the other ends of the first fiber channel 6 and the second fiber channel 9 are respectively connected to the signal light and local oscillator optical interfaces of the polarization-independent coherent optical receiver 7; the output electrical interface of the polarization-independent coherent optical receiver 7 is connected to the receiving end DSP chip .
  • the laser 1 is used to output continuous light, provide a DC optical carrier for the optical interconnection system, and also serves as a local oscillator for coherent detection.
  • the line width of the laser 1 is not less than 10 MHz.
  • the laser 1 preferably adopts a DFB (Distributed Feedback laser, distributed feedback laser).
  • the first optical splitter 2 is used to divide the output of the laser 1 into two channels, and at the same time control the distribution ratio of the optical carrier and the optical power of the local oscillator.
  • the light splitting ratio of the first beam splitter 2 is greater than 6:4 and less than 8:2.
  • the transmitting DSP chip 5 is used for encoding, constellation map mapping and appropriate pre-compensation of the binary bit stream signal to be transmitted into an analog electrical signal, which is used to drive the integrated dual-polarization coherent optical transmitter 4.
  • the DAC used in the sending DSP chip 5 works at a sampling rate of 1 times the baud rate to reduce system power consumption.
  • the integrated dual-polarization coherent optical transmitter 4 is used to modulate the analog electrical signal into a polarization multiplexed complex optical field signal, that is, signal light, to complete the conversion of the signal to be sent from the electrical domain to the optical field.
  • the first optical fiber channel 6 serves as a low-loss transmission medium for transmitting modulated signal light.
  • the coarse adjustable delay line 3 is used to roughly adjust the delay of the local oscillator optical transmission line to match the length of the optical signal transmission link and ensure that the signal light and the local oscillator light are near the coherence length.
  • the second optical fiber channel 9 is used as a low-loss transmission medium for transmitting the local oscillator optical signal adjusted by the coarsely adjustable delay line 3.
  • the lengths of the first optical fiber channel 6 and the second optical fiber channel 9 are approximately equal, and do not exceed 5 kilometers.
  • the polarization-independent coherent optical receiver 7 is used to coherently mix the local oscillator light and the signal light, move the modulated signal to the baseband, and obtain the baseband electrical signal. Further, the polarization-independent coherent optical receiver 7 realizes feedback scrambling, thereby automatically adapting the input local oscillator light in any polarization state.
  • the receiving end DSP chip 8 is used to convert the received baseband electrical signal into a digital signal, and perform certain digital signal processing, such as damage equalization and compensation, polarization demultiplexing, demodulation and decoding, etc., and finally restore the transmitting binary data stream .
  • the ADC of the receiving DSP chip 8 also works at a sampling rate of 1 times the baud rate to reduce system power consumption.
  • an embodiment of the polarization-independent coherent optical receiver 7 in the above-mentioned embodiment is provided, and its internal structure specifically includes a first polarization beam splitter 701, a second beam splitter 709, a third beam splitter 710, and a first polarization beam splitter 701.
  • the mixing gain module 700 consists of a first 90° mixer 702, a first balanced detector 703, a second balanced detector 704, a transimpedance amplifier 705, a second 90° mixer 706, a third balanced detector 707 and The fourth balanced detector 708 is constituted.
  • the input of the first polarization beam splitter 701 is used as the signal light input port of the polarization-independent coherent optical receiver 7, and one end of the finely adjustable delay line 715 is used as the local oscillator optical input port of the polarization-independent coherent optical receiver 7, and a transimpedance amplifier
  • the output of 705 is used as the electrical signal output port of the polarization-independent coherent optical receiver 7.
  • the two outputs of the first polarization beam splitter 701 are respectively connected to one input port of the first 90° mixer 702 and the second 90° mixer 706; one of the second beam splitter 709 and the second beam splitter 710
  • the output ports are respectively connected to the other input ports of the first 90° mixer 702 and the second 90° mixer 706.
  • the four outputs of the first 90° mixer 702 are connected to the input ports of the first balanced detector 703 and the second balanced detector 704, respectively.
  • the four outputs of the second 90° mixer 706 are connected to the input ports of the third balanced detector 707 and the fourth balanced detector 708, respectively.
  • the outputs of the first balanced detector 703, the second balanced detector 704, the third balanced detector 707, and the fourth balanced detector 708 are connected to the transimpedance amplifier 705.
  • the input of the second polarization beam splitter 711 is connected to the output of the scrambler 714, and its two outputs are respectively connected to the input of the second beam splitter 709 and the third beam splitter 710.
  • the other output ports of the second beam splitter 709 and the third beam splitter 710 are connected to the input of the fifth balanced detector 712.
  • the output of the fifth balance detector 712 is connected to the input of the controller 713, the output of the controller 713 is connected to the control port of the scrambler 714, and the input of the scrambler 714 is connected to the output of the finely adjustable delay line 715.
  • the finely adjustable delay line 715 is used to precisely adjust the transmission delay and optical path difference of the two optical paths of the local oscillator light and the signal light in a small range to ensure that the signal light and the local oscillator light meet the coherence length.
  • the first polarization beam splitter 701 is used to divide the complex optical field signal (ie, signal light) into two branches whose polarization states are perpendicular to each other.
  • the second polarization beam splitter 711 is used to divide the local oscillator light into two branches with polarization states perpendicular to each other.
  • the polarization scrambler 714 is used to control the polarization state of the local oscillator light input to the second polarization beam splitter 711 to prevent the polarization state of the local oscillator light entering the second polarization beam splitter 711 from falling into the X and Y directions, that is, to avoid the local oscillator light Exactly horizontal or vertical.
  • the second beam splitter 709 and the third beam splitter 710 are used to split the two local oscillator lights with polarization states in different power ratios, and the path with the smaller power is sent to the fifth balanced detector 712.
  • the fifth balanced detector 712 is used to receive the local oscillator light with a smaller power split by the second beam splitter and the third beam splitter, extract the power difference of the local oscillator after polarization beam splitting, and convert it into a photocurrent amplitude output .
  • the controller 713 is used to convert the amplitude of the photocurrent to the disturbance control signal of the polarization state of the local oscillator light, and to control the polarizer 714 in a certain manner to realize polarization disturbance.
  • the frequency mixing gain module 700 is used for cohering the two complex optical field signals output by the first polarization beam splitter 701 with the higher power local oscillator light split by the second beam splitter 709 and the third beam splitter 711 respectively. The mixing then obtains the real and imaginary parts of the X and Y polarizations respectively, and then amplifies and realizes automatic gain control. Further, the specific functions of its internal structure are as follows:
  • the first 90° mixer 702 is used to coherently mix a complex optical field signal output by the first polarization beam splitter 701 and a local oscillator light with a larger power split by the second optical splitter 709.
  • the second 90° mixer 706 is used to coherently mix another complex optical field signal output by the first polarization beam splitter 701 with a local oscillator light with a larger power split by the third optical splitter 710.
  • the first balanced detector 703 and the second balanced detector 704 are used to convert the optical signal output by the first 90° mixer 702 into an electrical signal to obtain the real part and the imaginary part of the X polarization, respectively.
  • the third balanced detector 707 and the fourth balanced detector 708 are used to convert the optical signal output by the second 90° mixer 706 into an electrical signal to obtain the real part and the imaginary part of the Y polarization, respectively.
  • the transimpedance amplifier 705 is used to amplify the real and imaginary parts of the X polarization and the real and imaginary parts of the Y polarization, and realize automatic gain control.
  • the aforementioned polarization-independent coherent optical receiver 7 realizes feedback scrambling by detecting the differential power signals of the second optical splitter 709 and the third optical splitter 710, thereby automatically adapting the input local oscillator light of any polarization state.
  • the process of the controller 713 controlling the scrambler 714 in a certain manner specifically includes:
  • the fifth balanced detector 712 detects the optical power difference of the local oscillator light with the lower output power of the third beam splitter 709 and the fourth beam splitter 710, and converts it into a photocurrent amplitude output.
  • the controller 713 judges whether the absolute value is greater than a preset threshold, and if so, it proceeds to S304; if not, it returns to S301.
  • the threshold value is 90% of the maximum photocurrent detected by the fifth balance detector 712.
  • the controller 713 outputs a control electrical signal proportional to the above-mentioned absolute value, and sends it to the scrambler 714.
  • the polarization scrambler 714 perturbs the input polarization state of the local oscillator light according to the magnitude of the received control electrical signal.
  • the laser 1 is a DFB laser with a nominal line width of 10 MHz and an output power of 16 dBm.
  • the light splitting ratio of the first beam splitter 2 is 7:3.
  • the coarse adjustable delay line 3 is realized by using a single-mode fiber of 1 to 5 meters.
  • Both the first optical fiber channel 6 and the second optical fiber channel 9 are ordinary single-mode optical fibers with a length of 2 kilometers.
  • the ratio of the second beam splitter 709 and the third beam splitter 710 is the same, both being 95:5.
  • the originating DSP chip 5 specifically includes an FEC encoding module 51, a constellation map mapping module 52, a pre-encoding module 53, a pre-compensation module 54 and a DAC module 55.
  • the FEC encoding module 51 is used to perform FEC encoding on the binary bit stream signal to be transmitted.
  • the constellation mapping module 52 is configured to perform independent constellation mapping for the two polarization states of X and Y respectively according to a preset modulation format, and generate two independent complex signal streams [X 1 , Y 1 ] T.
  • the precoding module 53 is used to perform polarization hierarchical precoding on the complex signal stream [X 1 , Y 1 ] T , and output two precoding symbol streams [X 2 , Y 2 ] T.
  • the precoding rule is:
  • [X 2 ,Y 2 ] T H ⁇ [X 1 ,Y 1 ] T , Is the precoding matrix, which satisfies ad-bc ⁇ 0.
  • the pre-compensation module 54 is used to orthogonally separate the two pre-coded signals to obtain four corresponding signals, which represent the real and imaginary parts of the X and Y polarization states, and then perform time-domain pre-compensation to adjust The relative time delay between the signals compensates for the bandwidth of some optoelectronic devices.
  • the pre-compensation module 54 uses a real-number domain finite impulse response filter (FIR) to perform time-domain pre-compensation; to ensure low complexity and low power consumption, the number of FIR taps is not more than three.
  • FIR real-number domain finite impulse response filter
  • the DAC module 55 is used to perform digital-to-analog conversion on the compensated signal at a sampling rate of 1 times the baud rate to obtain an analog electrical signal.
  • the receiving end DSP chip 8 includes an ADC module 81, two low-pass filters 82, two feedforward equalizers 83, two clock recovery modules 84, an adaptive equalization module 85, and a deprecoding module 86. , Two constellation diagram mapping modules 87 and two FEC decoding modules 88.
  • the ADC module 81 is used to convert the four-channel electrical signals output by the polarization-independent coherent receiver 7 into four-channel digital signals at a sampling rate of 1 times the baud rate.
  • the two low-pass filters 82 are used to perform anti-aliasing filtering processing on the X-polarized signal and the Y-polarized signal, respectively, to remove signal spectrum aliasing that may be caused by single sampling.
  • Two feedforward equalizers 83 each corresponding to a low-pass filter 82, perform feedforward equalization on the filtered signal; to compensate for the influence of inter-symbol interference (ISI) caused by system bandwidth limitation and short-distance fiber dispersion, It can also compensate for the time difference between channels.
  • ISI inter-symbol interference
  • Two clock recovery modules 84 respectively recover the optimal sampling clock and phase of each feedforward equalized signal; the clock synchronization algorithm needs to work at a single sampling rate.
  • the adaptive equalization module 85 is used to perform adaptive equalization on the clock-recovered signal, realize polarization demultiplexing, and obtain two equalized signals [A 1 , B 1 ] T.
  • Two constellation diagram mapping modules 87 respectively de-map the signals that have completed de-mapping and encoding.
  • Two FEC decoding modules 88 respectively perform FEC decoding on the demapped binary bits to restore the binary bit stream signal.
  • the baud rate is 32 Gbaud
  • the adopted modulation format is polarization multiplexing QPSK, that is, a PDM-QPSK signal.
  • the constellation diagram and bit mapping mode are shown in FIG. 9.
  • the polarization diversity precoding matrix is The number of taps used by the feedforward equalizer 83 is 7, and the adaptive equalization module 85 uses a single-tap adaptive filter.
  • the adaptive update of the taps is quasi-cascaded multi-mode algorithm (CMMA), and the update period of tap coefficients is equal to every 32 symbols. cycle.
  • the baud rate is 42 Gbaud
  • the adopted modulation format is polarization multiplexing 8QAM, that is, PDM-8QAM signal.
  • the constellation diagram and bit mapping mode are shown in Fig. 10.
  • Figure 10 shows a special 8QAM, which is a subset of the 16QAM constellation points, which is different from the regular square or ring 8QAM because it has the largest minimum Euclidean distance, and thus has better BER performance under AWGN.
  • the mapping relationship between bits and symbols specifies the uniquely determined correspondence method from binary bits to complex signal symbols.
  • the polarization diversity precoding matrix is The number of taps used in feedforward equalization is 9, and the tap adaptive update of the single-tap adaptive filter is quasi-multimodal algorithm (MMA), and the update period of the tap coefficients is equal to every 16 symbol periods.
  • MMA quasi-multimodal algorithm

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Abstract

一种光互连通信方法及系统，涉及数据中心内的高速光互连领域，包括：将待传送的二进制比特流信号经DSP处理后变为模拟电信号；采用激光器输出两路光，一路作为集成双偏相干光发射机的光载波将所述模拟电信号调制为复光场信号，另一路作为本振光进行时延调整；通过偏振无关相干光接收机对所述复光场信号和所述本振光进行相干混频，其中，通过反馈扰偏控制所述本振光的偏振态，避免本振光的偏振态落入X和Y方向；混频后的电信号通过DSP处理恢复所述二进制比特流信号。本发明适用于多种调制格式以及多种速率，还可以降低成本和功耗。

Description

一种光互连通信方法及系统 技术领域
本发明涉及数据中心内的高速光互连领域，具体涉及一种光互连通信方法及系统。
背景技术
随着HDTV(High Definition Television，高清视频)、VR(Virtual Reality，虚拟现实)、远程会议、移动互联以及云计算等新型、带宽密集型网络应用的出现和普及，网络流量每年以超过22％的复合年增长率保持快速增长。光纤通信网络作为信息承载、传输和交换的基石，也将面临着巨大的压力。特别是在5G即将商用的趋势下，DCN(Data Center Network,数据中心网络)和CDN(Content Delivery Network，内容分发网络)的新建，悄然改变着网络中大数据流量的分布。以在线直播、视频传输、文件共享为主的大数据业务占据互联网流量的主体，它们主要是通过DCN和CDN承载和分发，大多数场景下不需要经过长途网传输，因而网络流量负荷从长途骨干网向中短距城域网和数据中心转移。研究表明2016年全球数据中心相关的IP流量已经高达6.8ZB(1ZB＝10 9TB)，预计到2021年将增长到20.6ZB，增长速率约为5年翻2倍。特别是在云计算数据中心中，数据流量压力更加明显。预计到2021年，94％的业务都将在云数据中心中处理，不过在数据中心相关的网络中，超过70％的流量都被终结在数据中心之内。很明显，短距离(小于2km)、高速光互连技术将在未来大数据传输和承载方面发挥重要作用。
面对数据中心内光互连的带宽升级需求，容量、功耗和成本是三个重要的考虑因素，国际主流标准组织对数据中心用的光模块进行了相应的规范。目前，基于短波长的VCSEL(Vertical Cavity Surface Emitting Laser，垂直腔面发射激光器)和多模光纤，利用多芯MPO(Multi Push On，多点推送)连接器实现10发10收，最终完成100G左右的光互连，但其传输距离通常被限制在100米左右。对于大型数据中心内，连接场景需要覆盖5km。此时通常需要升级成DML(Directly Modulated Laser，直调激光器)和单模光纤结合的方案，采用OOK(On-Off Keying，二进制启闭键控)调制方案，如MSA(Multi Source Agreement，多源协议)100G CWDM4(Coarse Wavelength Division Multiplexing，粗波分复用)标准的光模可支持2km、100G传输，但单向需要4套收发设备。对于下一代400G、2km左右的互连需求，则需要在单模光纤的基础上，采用EML(Electro-absorption Modulated Laser，电吸收调制激光器)替代DML结合高阶PAM4(4 Pulse Amplitude Modulation，4级脉冲幅度调制)调制技术，波长数增加到8个，成本和功耗显著增加。对于后续进一步扩容升级，这种IMDD(Intensity Modulation Direct Detection，强度调制直接探测)技术，将显得捉襟见肘，模块尺寸、功耗和成本都将成为挑战。
相较于传统的IMDD，数字相干光通信由于具有更好的灵敏度，更高的频谱效率，更强的损伤补偿能力，已经成为未来长距光通信的主流技术，并且已经大规模商用部署。但是现有的长距100G/200G相干DWDM(Dense Wavelength Division Multiplexing，密集型光波复用)系统并不能直接用于短距数据中心内的光互连，因为数据中心内的端口数量庞多，对收发机模块的成本极其敏感。同时，长距模块的功耗和尺寸都较大，进一步限制了其在数据中心高密度光互连中的 应用。那么如何降低现有相干光通信系统的成本、功耗是相干技术用于短距数据中心光互连场景面临的首要问题。
目前，低成本的相干光通信技术主要有三类。其一为试图用纯模拟的光电信号处理方式来避免使用高速数模/模数转换器(DAC/ADC)及复杂的DSP ASIC(Digital Signal Processing Application Specific Integrated Circuit，数字信号处理芯片)，以期降低成本和功耗。不过，这种完全不需要DSP的方案在性能上会有较大的劣化，因为完全不能补偿器件带宽限制，也难以适用于更高阶的调制格式，同时低复杂度、无串扰的偏振解复用是其走向应用的限制因素之一。[Journal of Lightwave technology,Vol.35,no.21,Design of Low-Power DSP-Free Coherent Receivers for Data Center Links]。第二类为采用简化的3x3耦合器来取代传统的90°相干混频器以期降低部分光器件的成本。[Journal of Lightwave technology,Vol.36,no.16,Comparison of low complexity coherent receivers for UDWDM-PONs]。还有一类则是采用自相干的方法，本振光和信号光一起从发端传输到收端，采用某种方法如空分复用扇入扇出，或偏振分离等技术将本振光与信号光分离后再进行相干混频，这样可以降低激光器的数量和成本，同时还可以简化部分收端的DSP(Digital Signal Processing，数字信号处理)算法[Optics Express,vol.21,no.2,investigating self-homodyne coherent detection in a 19 channel space-division-multiplexed transmission link]。考虑到产业化及模块性能一致性，不难发现，基于DSP的低成本、低功耗相干光互连仍是业界最为期待的方案。但是上述第二类和第三类低成本相干系统中，由于涉及高速率的DA(数模)和AD(模数)采样，因而其功耗较高，难以满足数据中心可插拔光模块的功耗要求。降低系统采样率、进一步简化相干DSP算法及架构是降低功耗可能的方向，也是数据中心高速相干光互连技术急待解决的难题。
如图1和图2所示，分别为传统的用于长距传输的相干光通信的发端DSP处理流程和收端DSP处理流程，都包含有采样率变换的部分，如图1的上采样以及图2中的重采样等，不仅结构复杂，而且由于工作在高倍采样率(大于波特率)条件下因而功耗较大。特别是收端的色散补偿、多抽头的自适应均衡、频偏估计和相位恢复等功能的计算复杂度较大，占据芯片功耗的大部分，显然这种传统的DSP架构并不能直接适用于低成本和低功耗的数据中心短距光互连系统中。信号的调制格式是常见的偏振复用的相位调制或幅度相位调制，如PDM(Polarization Division Multplexed，偏振复用)-PSK(Phase Shift Keying，相移键控)、PDM-QPSK(Quadrature Phase Shift Keying，四进制相移键控)、PDM-8QAM(8-ary Quadrature Amplitude Modulation，8进制正交幅度相位调制)、PDM-16QAM(16-ary Quadrature Amplitude Modulation，16进制正交幅度相位调制)、PDM-64QAM(64-ary Quadrature Amplitude Modulation，64进制正交幅度相位调制)等。
发明内容
针对现有技术中存在的缺陷，本发明的目的在于提供一种光互连通信方法及系统，适用于多种调制格式以及多种速率，还可以降低成本和功耗。
为达到以上目的，一方面，采取一种光互连通信方法，包括：
将待传送的二进制比特流信号经DSP处理后变为模拟电信号；采用激光器输出两路光，一路作为集成双偏相干光发射机的光载波将所述模拟电信号调制为复光场信号，另一路作为本振光进行时延调整；
通过偏振无关相干光接收机对所述复光场信号和所述本振光进行相干混频，其中，通过反馈扰偏控制所述本振光的偏振态，避免本振光的偏振态落入X和Y方向；混频后的电信号通过DSP处理恢复 所述二进制比特流信号。
优选的，所述待传送的二进制比特流信号经数字信号处理包括：
待传送的二进制比特流信号进行FEC编码后，对X、Y两个偏振态进行独立的星座图映射，生成两路独立的复数信号流，再进行偏振分集预编码，预编码后的两路信号经正交分离后进行时域预补偿，补偿后的信号以波特率为采样率进行数模转换，得到所述模拟电信号。
优选的，所述星座图映射生成的复数信号流为[X 1,Y 1] T，所述预编码后的符号流为[X 2,Y 2] T，预编码的规则为[X 2,Y 2] T＝H·[X 1,Y 1] T，
Figure PCTCN2020138778-appb-000001
为预编码矩阵，且满足ad-bc≠0。
优选的，混频后的电信号通过数字信号处理包括：
混频后的电信号模数转换为数字信号并进行抗混叠滤波，滤波后的信号进行前馈均衡和时钟同步，再进行单抽头自适应均衡实现偏振解复用，然后进行解预编码、解映射和FEC解码，得到所述二进制比特流信号。
优选的，所述偏振解复用得到的两路信号为[A 1,B 1] T，利用一个矩阵对[A 1,B 1] T进行解偏振分集预编码处理得到[A 2,B 2] T，所述解偏振分集预编码的规则为：[A 2,B 2] T＝H‘·[A 1,B 1] T，其中H‘为预编码矩阵H的逆矩阵，
Figure PCTCN2020138778-appb-000002
优选的，所述反馈扰偏控制所述本振光的偏振态包括：
所述偏振无关相干光接收机将收到的本振光分为两个偏振态互相垂直的支路，每个支路的本振光均按照不同功率比例分为两路，提取每个支路本振光分出的光功率较小的一路，并获得提取的两路本振光的功率差，并将所述功率差转换为光电流幅度，当所述光电流幅度的绝对值大于预设阈值时，通过一个与所述绝对值大小成正比的控制 信号，对本振光进行扰动。
优选的，所述光互连通信方法适用调制格式包括QPSK、8QAM、16QAM、32QAM以及64QAM；
所述光互连通信方法适用信息速率包括100G、200G、400G、600G以及800G。
另一方面，提供一种光互连通信系统，包括：
激光器，用于输出连续光；
第一分光器，用于接收所述连续光并分为两路，一路提供光载波，一路作为本振光；
发端DSP芯片，用于将待传送的二进制比特流信号经DSP处理后变为模拟电信号；
集成双偏相干光发射机，用于接收所述光载波，将所述模拟电信号调制为复光场信号；
第一光纤通道，用于传输所述集成双偏相干光发射机输出的复光场信号；
粗可调延时线，用于粗调所述本振光传输线路的时延，以匹配复光场信号传输链路长度；
第二光纤通道，用于传输所述粗可调延时线调整后的本振光；
偏振无关相干光接收机，用于接收分别由第一光纤通道和第二光纤通道传输后的所述复光场信号和所述本振光并进行相干混频；所述偏振无关相干光接收机包括扰偏器，用于通过反馈扰偏控制所述本振光的偏振态，避免本振光的偏振态落入X和Y方向；
收端DSP芯片，用于接收混频后的电信号，并通过DSP处理恢复所述二进制比特流信号。
优选的，所述激光器为DFB激光器；第一分光器2的分光比例 为7：3；粗可调延时线采用单模光纤实现；第一光纤信道和第二光纤信道均为普通单模光纤。
优选的，所述偏振无关相干光接收机包括：
第一偏振分束器，用于将所述复光场信号分为两个偏振态互相垂直的支路；
精可调延时线，用于精确调整本振光相对于信号光的传输时延和光程差，确保信号光和本振光满足相干长度以内；
第二偏振分束器，用于将所述本振光分为两个偏振态互相垂直的支路；
扰偏器，其接收精可调延时线调整后的本振光，控制所述本振光输入到第二偏振分束器的偏振态，避免进入第二偏振分光器的本振光恰好为水平或垂直方向；
第二分光器和第三分光器，用于将两个具有偏振态的本振光进行不同功率比例地分路；
第五平衡探测器，用于接收第二分光器和第三分光器分别分出的功率较小的一路本振光，获得功率差，并转换为光电流幅度输出；
控制器，用于所述光电流幅度转换为本振光偏振态的扰动控制信号，控制所述扰偏器；
混频增益模块，用于将第一偏振分束器输出的两路复光场信号分别与第二分光器、第三分光器分出的功率较大的一路本振光进行相干混频，再分别得到X、Y偏振的实部和虚部，进行放大并实现自动增益控制。
优选的，所述混频增益模块包括：
第一90°混频器，用于将第一偏振分束器输出的一路复光场信号与第二分光器分出的功率较大的一路本振光进行相干混频；
第二90°混频器，用于将第一偏振分束器输出的另一路复光场信号与第三分光器分出的功率较大的一路本振光进行相干混频；
第一平衡探测器和第二平衡探测器，用于将第一90°混频器输出的光信号转为电信号，分别得到X偏振的实部和虚部；
第三平衡探测器和第四平衡探测器，用于将第二90°混频器输出的光信号转为电信号，分别得到Y偏振的实部和虚部；
跨阻放大器，用于将所述X偏振的实部和虚部、所述Y偏振的实部和虚部进行放大并实现自动增益控制。
优选的，所述第二分光器和第三分光器的比例相同，均为95：5；
所述第五平衡探测器输出的光电流幅度的绝对值大于预设阈值时，所述输出一个与所述绝对值大小成正比的控制电信号作用于所述扰偏器，所述阈值为第五平衡探测器所探测到最大光电流的90％。
优选的，所述发端DSP芯片包括：
FEC编码模块，用于将待传送的二进制比特流信号进行FEC编码；
星座图映射模块，用于按照预设的调制格式，分别针对X、Y两个偏振态进行独立的星座图映射，生成两路独立的复数信号流[X 1,Y 1] T；
预编码模块，用于将所述复数信号流[X 1,Y 1] T进行偏振分级预编码，输出两路预编码符号流[X 2,Y 2] T，预编码的规则为：
[X 2,Y 2] T＝H·[X 1,Y 1] T，
Figure PCTCN2020138778-appb-000003
为预编码矩阵，满足ad-bc≠0。
预补偿模块，用于对预编码后的两路信号经正交分离后进行时域预补偿；
DAC模块，用于对补偿后的信号以波特率为采样率进行数模转 换，得到所述模拟电信号。
优选的，所述收端DSP芯片包括：
ADC模块，其用于将混频后的电信号转换为数字信号；
两个低通滤波器，分别对X偏振信号和Y偏振信号分别进行抗混叠滤波处理，
两个前馈均衡器，各对应一个低通滤波器，对滤波后的信号进行前馈均衡；
两个时钟恢复模块，分别将每个前馈均衡后的信号恢复最佳采样时钟和相位；
自适应均衡模块，对时钟恢复后的信号进行自适应均衡，实现偏振解复用，得到均衡后的两路信号[A 1,B 1] T。
解预编码模块，利用一个矩阵对[A 1,B 1] T进行解偏振分集预编码处理得到[A 2,B 2] T，所述解偏振分集预编码的规则为：[A 2,B 2] T＝H‘·[A 1,B 1] T，其中H‘为预编码矩阵H的逆矩阵；
两个星座图解映射模块，分别对完成解偏振分集预编码的信号解映射；
两个FEC解码模块，分别对解映射后的二进制比特进行FEC解码，恢复所述二进制比特流信号。
优选的，所述发端DSP芯片的预补偿模块采用实数域有限冲激响应滤波器进行时域预补偿，有限冲激响应滤波器的抽头个数小于等于3个；
所述收端DSP芯片的自适应均衡模块采用单抽头2x2复数蝶形滤波器进行自适应均衡；
所述DAC模块和ADC模块均工作在1倍波特率采样速率下。
上述技术方案中具有如下有益效果：
本发明中仅有发端设有激光器，相对于现有技术，采用自相干技术来避免收端采用昂贵的窄线宽激光器，只用发端一个激光器，降低短距光互连系统的成本。
进一步的，发端采用DFB(Distributed Feedback laser,分布反馈激光器)激光器，它的线宽一般比较大，是MHz量级，并且价格较ECL(External Cavity Laser，外腔激光器)便宜，ECL一般是窄线宽激光器，线宽是KHz量级。而目前商用相干光通信系统都需要100kHz左右的ECL激光器，因此采用DFB激光器相对于其他激光器，进一步降低了系统成本。
采用基于功率平衡探测和扰偏器相结合的方式对本振光的偏振态进行控制，避免本振光恰好为X或Y偏振时解调失效的风险，实现偏振无关的相干接收，解决自相干系统中本振光偏振态控制的难点。
在光路上采用粗可调延时线和精可调延时线来调节本振光和信号光的光程差，保证相干长度的匹配，从而避免在收端进行频偏估计和相位恢复，最大程度地精简收端的DSP算法和电路。
在DSP方面，一方面采用波特率采样速率的DAC和ADC转换，较大程度地降低了相干DSP芯片的功耗；另一方面通过提出一套极简的收发端DSP信号处理流程，简化DSP的复杂度，节省芯片功耗和面积。
在发端DSP芯片中利用偏振分集预编码，在两个偏振态的信号之间人为地引入相关性，有望提高对偏振相关损耗的容忍度，一定程度上解决低成本器件用于相干系统中性能损失问题。
在收端DSP芯片中，利用FFE(Forward Feedback Equalizer，前馈均衡器)来补偿光纤色散、器件带宽限制引入的ISI(Inter Symbol Interference，码间干扰)问题，然后再利用单抽头的蝶形滤波器来实 现偏振解复用，再进行解预编码，恢复X，Y两个偏振态上的符号。整个DSP算法不仅相比于传统的相干DSP算法得到了极大地简化，而且所有的算法子模块都工作在单倍采样率下，因而DSP芯片的功耗有望得到大幅降低。
并且，基于发端的DSP处理，可以适用不同的高阶调制格式，如QPSK、8QAM、16QAM、32QAM、64QAM等，同时适用于多信息种速率，如100G、200G、400G、600G、800G等。
综上所述，本发明最终实现的有益效果是可以实现低成本、高速率、低功耗的短距光互连，非常适用于对成本功耗敏感的数据中心内部光互连场景。
附图说明
图1为传统用于长距传输的相干光通信的发端DSP处理流程示意图；
图2为传统用于长距传输的相干光通信的收端DSP处理流程示意图；
图3为本发明实施例单抽头自适应均衡器结构示意图；
图4为本发明实施例光互连通信系统的示意图；
图5为偏振无关相干光接收机的实现原理图；
图6为本发明实施例发端DSP芯片示意图；
图7为本发明实施例收端DSP芯片示意图；
图8为本发明实施例本振光偏振控制流程图；
图9为本发明实施例QPSK星座图；
图10为本发明实施例8QAM的星座图。
附图标记：
1-激光器、2-第一分光器、3-粗可调延时线、4-集成双偏相干光 发射机、5-发端DSP芯片、6-第一光纤信道、7-偏振无关相干光接收机、8-收端DSP芯片、9-第二光纤信道；
51-FEC编码模块、52-星座图映射模块、53-预编码模块、54预补偿模块、55-DAC模块；
700-混频增益模块、701-第一偏振分束器、702-第一90°混频器、703-第一平衡探测器、704-第二平衡探测器、705-跨阻放大器、706-第二90°混频器、707-第三平衡探测器、708-第四平衡探测器、709-第二分光器、710-第三分光器、711-第二偏振分束器、712-第五平衡探测器、713-控制器、714-扰偏器、715-精可调延时线；
81-ADC模块、82-低通滤波器、83-前馈均衡器、84-时钟恢复模块、85-自适应均衡模块、86-解预编码模块、87-星座图解映射模块、88-FEC解码模块。
具体实施方式
为了使本发明的目的、技术方案及优点更加清楚明白，以下结合附图及实施例，对本发明进行进一步详细说明。应当理解，此处所描述的具体实施例仅用以解释本发明，并不用于限定本发明。此外，下面所描述的本发明各个实施方式中所涉及到的技术特征只要彼此之间未构成冲突就可以相互组合。
提供一种光互连通信方法的实施例，具体包括：
在发送方向，将待传送的二进制比特流信号经DSP处理后变为模拟电信号。通过一个激光器为光互连通信提供直流光载波，同时也作为相干探测的本振光。具体的，采用激光器输出两路光，其中一路作为集成双偏相干光发射机的光载波，将DSP处理得到的模拟电信号调制为复光场信号；另一路作为本振光进行时延调整。其中，可以通过粗可调延时线，粗略调整本振光传输线路的时延，以匹配复光 场信号传输链路长度，保证复光场信号和本振光在相干长度附近。
在接收方向，通过偏振无关相干光接收机接收上述复光场信号和本振光，进行相干混频，将调制信号搬移到基带，得到基带电信号。其中，通过反馈扰偏控制本振光的偏振态，避免本振光的偏振态落入X和Y方向，进而造成其中一个偏振态上数据的丢失，保证偏振复用数据的正确解调。混频后的电信号通过DSP处理恢复二进制比特流信号。其中，混频后得到四路电信号，分别为X偏振和Y偏振两路电信号，以及各自的in phase和quadrature。
上述通过反馈扰偏控制本振光的偏振态具体包括：偏振无关相干光接收机将收到的本振光分为两个偏振态互相垂直的支路，每个支路的本振光均按照不同功率比例再分为两路，提取每个支路本振光分出的光功率较小的一路，并获得提取的两路本振光的功率差，并将该功率差转换为光电流幅度，当光电流幅度的绝对值大于预设阈值时，通过一个与所述绝对值大小成正比的控制信号，对本振光进行扰动。
基于上述实施例，进一步提供一种发送方向和接收方向上DSP处理的实施例。上述待传送的二进制比特流信号经数字信号处理，具体包括步骤：
S101.将待传送的二进制比特流信号进行FEC(Forward Error Correction，前向纠错)编码。
S102.按照预先选定的调制格式，经过FEC编码后的数据分别针对X、Y两个偏振态(即水平和垂直两个偏振态)进行独立的星座图映射，生成两路独立的复数信号流[X 1,Y 1] T。
S103.复数信号流[X 1,Y 1] T再进行偏振分集预编码，输出两路预编码符号流[X 2,Y 2] T；预编码的规则为：
[X 2,Y 2] T＝H·[X 1,Y 1] T，其中，
Figure PCTCN2020138778-appb-000004
为预编码矩阵，且满 足ad-bc≠0。
S104.预编码后的两路信号先进行正交分离，得到对应的四路信号，分别代表X和Y偏振态的实部和虚部，然后分别经过一个实数域FIR(Finite Impulse Response，有限冲激响应滤波器)进行时域预补偿，以调整信号之间的相对时延并补偿部分光电器件的带宽。
优选的，为确保复杂度和功耗低，FIR的抽头个数不多于3个。
S105.经过预补偿后的四路信号送入DAC中，以波特率为采样率进行数模转换，得到待调制的电信号。
在接收方向，混频后的电信号通过数字信号处理具体包括如下步骤：
S201.偏振无关相干接收机输出的四路电信号以波特率为采样速率进行模数转换后，得到四路数字信号。
S202.利用低通滤波器对四路数字信号进行数字抗混叠滤波处理，以去除由于单倍采样可能导致的信号频谱混叠。
S203.对滤波后的信号进行前馈均衡，以补偿由于带宽受限及短距光纤色散导致的码间干扰(ISI)的影响，也可以补偿通道间的时间差；
S204.对经过前馈均衡后的信号进行时钟同步，恢复最佳采样时钟和相位，时钟同步算法需要工作在单倍采样速率下。
S205.如图3所示，Xin、Yin、Xin’和Yin’分别表示自适应均衡器X、Y两个偏振态上的输入和输出信号矢量，Wxx、Wxy、Wyx和Wyy分别为蝶形滤波器的四组抽头系数，分别表示X偏振输入对X偏振输出的影响，Y偏振输入对X偏振输出的影响，X偏振输入对Y偏振输出的影响，Y偏振输入对Y偏振输出的影响。按照一定的抽头自适应更新准则和更新周期，进行单抽头自适应均衡实现偏振解 复用，得到均衡后的两路信号为[A 1,B 1] T。
优选的，采用单抽头自适应滤波器对时钟恢复后的信号进行自适应均衡，如单抽头2x2复数蝶形滤波器。
S206.利用一个矩阵对[A 1,B 1] T进行解偏振分集预编码处理，得到[A 2,B 2] T，解偏振分集预编码的规则为：[A 2,B 2] T＝H‘·[A 1,B 1] T，其中H‘为预编码矩阵H的逆矩阵，
Figure PCTCN2020138778-appb-000005
S207.根据预先选定的调制格式，完成解预编码后信号的解映射，将恢复的符号变成二进制比特。
S208.对解映射后的二进制比特进行FEC解码，恢复发端的二进制数据流。
优选的，S205中前馈均衡时所用的抽头数量不大于10个，抽头自适应更新准则可以为恒模算法(CMA)，也可以为多模算法(MMA)、级联多模算法(CMMA)或最小均方误差算法(LMS)。单抽头自适应滤波器的抽头系数更新周期不小于每10个符号周期。
如图4所示，本发明还提出一种光互连通信系统的实施例，可以实现上述实方法中的至少一个。该系统包括发端和收端，发端包括激光器1、第一分光器2、粗可调延时线3、集成双偏相干光发射机4、发端DSP芯片5、第一光纤信道6和第二光纤信道9；收端包括偏振无关相干光接收机7和收端DSP芯片8；发端和收端之间通过两个光纤通道进行传输。
激光器1的输出与第一分光器2相连，第一分光器2的输出分别与集成双偏振相干光发射机4和粗可调延时线3相连，集成双偏相干光发射机4还与发端DSP芯片5和第一光纤信道6相连。粗可调延时线3的另一端口与第二光纤信道9相连。第一光纤信道6和第二光纤信道9的另一端分别连接到偏振无关相干光接收机7的信号光和本 振光接口；偏振无关相干光接收机7的输出电接口与收端DSP芯片相连。
激光器1用于输出连续光，为光互连系统提供直流光载波，同时也作为相干探测的本振光。优选的，激光器1线宽不小于10MHz。激光器1优选采用DFB(Distributed Feedback laser,分布反馈激光器)。
第一分光器2用于将激光器1的输出分为两路，同时控制光载波和本振光功率的分配比例。优选的，第一分光器2的分光比例大于6：4，小于8：2。
发端DSP芯片5用于将待传送的二进制比特流信号进行编码、星座图映射和适当的预补偿后转变成模拟电信号，用于驱动集成双偏相干光发射机4。发端DSP芯片5中采用的DAC工作在1倍波特率采样速率下，以降低系统功耗。
集成双偏相干光发射机4用于将模拟电信号调制为偏振复用的复光场信号，即信号光，完成待发送信号从电域到光场的变换。
第一光纤信道6作为低损耗的传输媒介，用于传输调制后的信号光。
粗可调延时线3用于粗略调整本振光传输线路的时延，以匹配光信号传输链路长度，保证信号光和本振光在相干长度附近。
第二光纤信道9作为低损耗的传输媒介，用于传输粗可调延时线3调整后的本振光信号。
进一步的，第一光纤信道6和第二光纤信道9的长度大致相等，并且不超过5公里。
偏振无关相干光接收机7用于将本振光与信号光进行相干混频，将调制信号搬移到基带，得到基带电信号。进一步地，所述偏振无关相干光接收机7实现反馈扰偏，从而自动适配任意偏振态输入的本振 光。
收端DSP芯片8用于将接收到的基带电信号转换成数字信号后，进行一定的数字信号处理，如损伤均衡和补偿、偏振解复用、解调解码等，最终还原发端的二进制数据流。收端DSP芯片8的ADC也工作在1倍波特率采样速率下，以降低系统功耗。
如图5所示，提供一种上述实施例中偏振无关相干光接收机7的实施例，其内部结构具体包括第一偏振分束器701、第二分光器709、第三分光器710、第二偏振分束器711、第五平衡探测器712、控制器713、扰偏器714、精可调延时线715和一个混频增益模块700。混频增益模块700由第一90°混频器702、第一平衡探测器703、第二平衡探测器704、跨阻放大器705、第二90°混频器706、第三平衡探测器707和第四平衡探测器708构成。
第一偏振分束器701的输入作为偏振无关相干光接收机7的信号光输入端口，精可调延时线715的一端作为偏振无关相干光接收机7的本振光输入端口，跨阻放大器705的输出作为偏振无关相干光接收机7的电信号输出端口。
第一偏振分束器701的两个输出分别连接至第一90°混频器702和第二90°混频器706的一个输入端口；第二分光器709和第二分光器710的其中一个输出端口，分别连接到第一90°混频器702和第二90°混频器706的另一个输入端口。第一90°混频器702的四个输出分别连接到第一平衡探测器703和第二平衡探测器704的输入端口。第二90°混频器706的四个输出分别连接到第三平衡探测器707和第四平衡探测器708的输入端口。第一平衡探测器703、第二平衡探测器704、第三平衡探测器707和第四平衡探测器708的输出连接到跨阻放大器705。
第二偏振分束器711的输入与扰偏器714的输出相连，其两个输出分别与第二分光器709和第三分光器710的输入相连。第二分光器709及第三分光器710的另一个输出端口与第五平衡探测器712的输入相连。第五平衡探测器712的输出与控制器713的输入相连，控制器713的输出与扰偏器714的控制端口相连，扰偏器714的输入端与精可调延时线715的输出相连。
精可调延时线715用于精确小范围调整本振光和信号光两个光路的传输时延和光程差，确保信号光和本振光满足相干长度以内。
第一偏振分束器701用于将复光场信号(即信号光)分为两个偏振态互相垂直的支路。
第二偏振分束器711用于将本振光分为两个偏振态互相垂直的支路。
扰偏器714用于控制本振光输入到第二偏振分束器711的偏振态，避免进入第二偏振分光器711的本振光的偏振态落入X和Y方向，即避免本振光恰好为水平或垂直方向。
第二分光器709和第三分光器710用于将两个具有偏振态的本振光进行不同功率比例地分路，功率较小的那路送给第五平衡探测器712。
第五平衡探测器712用于接收第二分光器和第三分光器分别分出的功率较小的一路本振光，提取偏振分束后的本振光功率差，并转换为光电流幅度输出。
控制器713用于将光电流幅度转换为本振光偏振态的扰动控制信号，并且按照一定的方式控制扰偏器714，实现偏振扰动。
混频增益模块700，用于将第一偏振分束器701输出的两路复光场信号分别与第二分光器709、第三分光器711分出的功率较大的一 路本振光进行相干混频再分别得到X、Y偏振的实部和虚部，进行放大并实现自动增益控制。进一步的，其内部结构具体功能如下：
第一90°混频器702用于将第一偏振分束器701输出的一路复光场信号与第二分光器709分出的功率较大的一路本振光进行相干混频。
第二90°混频器706用于将第一偏振分束器701输出的另一路复光场信号与第三分光器710分出的功率较大的一路本振光进行相干混频。
第一平衡探测器703和第二平衡探测器704，用于将第一90°混频器702输出的光信号转为电信号，分别得到X偏振的实部和虚部。
第三平衡探测器707和第四平衡探测器708，用于将第二90°混频706器输出的光信号转为电信号，分别得到Y偏振的实部和虚部。
跨阻放大器705用于将上述X偏振的实部和虚部、上述Y偏振的实部和虚部均进行放大，并实现自动增益控制。
上述偏振无关相干光接收机7通过检测第二分光器709和第三分光器710的差分功率信号实现反馈扰偏，从而自动适配任意偏振态输入的本振光。
如图6所示，上述控制器713按照一定的方式控制扰偏器714的过程具体包括：
S301.第五平衡探测器712检测第三分光器709和第四分光器710输出功率较小的本振光的光功率差，并转换为光电流幅度输出。
S302.对第五平衡探测器712输出的光电流幅度取绝对值。
S303.控制器713判断绝对值是否大于预先设置的阈值，若是，进入S304；若否，转回S301。优选的，阈值为第五平衡探测器712所探测到最大光电流的90％。
S304.控制器713输出一个与上述绝对值大小成正比的控制电信号，送给扰偏器714。
S305.扰偏器714根据接收到的控制电信号的大小，对输入的本振光偏振态进行扰动。
上述光互连通信系统中，优选的，激光器1为DFB激光器，标称线宽为10MHz，出光功率16dBm。第一分光器2的分光比例为7：3。粗可调延时线3采用1至5米的单模光纤来实现。第一光纤信道6和第二光纤信道9均为普通单模光纤，长度为2公里。所述第二分光器709和第三分光器710的比例相同，均为95：5。
如图7所示，提供一个光互连通信系统中发端DSP芯片5的实施例。发端DSP芯片5具体包括FEC编码模块51、星座图映射模块52、预编码模块53、预补偿模块54和DAC模块55。
FEC编码模块51用于将待传送的二进制比特流信号进行FEC编码。
星座图映射模块52用于按照预设的调制格式，分别针对X、Y两个偏振态进行独立的星座图映射，生成两路独立的复数信号流[X 1,Y 1] T。
预编码模块53用于将复数信号流[X 1,Y 1] T进行偏振分级预编码，输出两路预编码符号流[X 2,Y 2] T，预编码的规则为：
[X 2,Y 2] T＝H·[X 1,Y 1] T，
Figure PCTCN2020138778-appb-000006
为预编码矩阵，满足ad-bc≠0。
预补偿模块54用于对预编码后的两路信号经正交分离后，得到对应的四路信号，分别代表X，Y偏振态的实部和虚部，再进行时域预补偿，以调整信号之间的相对时延并补偿部分光电器件的带宽。优选的，预补偿模块54采用实数域有限冲激响应滤波器(FIR)进行时域预补偿；为确保复杂度和功耗低，FIR的抽头个数不多于3个。
DAC模块55用于对补偿后的信号以1倍波特率为采样速率进行数模转换，得到模拟电信号。
如图8所示，收端DSP芯片8包括ADC模块81、两个低通滤波器82、两个前馈均衡器83、两个时钟恢复模块84、自适应均衡模块85、解预编码模块86、两个星座图解映射模块87以及两个FEC解码模块88。
ADC模块81用于将偏振无关相干接收机7输出的四路电信号，以1倍波特率为采样速率转换为四路数字信号。
两个低通滤波器82，用于对X偏振信号和Y偏振信号分别进行抗混叠滤波处理，以去除由于单倍采样可能导致的信号频谱混叠。
两个前馈均衡器83，各对应一个低通滤波器82，对滤波后的信号进行前馈均衡；以补偿由于系统带宽受限及短距光纤色散导致的码间干扰(ISI)的影响，也可以补偿通道间的时间差。
两个时钟恢复模块84，分别将每个前馈均衡后的信号恢复最佳采样时钟和相位；时钟同步算法需要工作在单倍采样速率下。
自适应均衡模块85，用于对时钟恢复后的信号进行自适应均衡，实现偏振解复用，得到均衡后的两路信号[A 1,B 1] T。
解预编码模块86，利用一个矩阵对[A 1,B 1] T进行解偏振分集预编码处理得到[A 2,B 2] T，解解偏振分集预编码的规则为：[A 2,B 2] T＝H‘·[A 1,B 1] T，其中H‘为预编码矩阵H的逆矩阵。
两个星座图解映射模块87，分别对完成解映射编码的信号解映射。
两个FEC解码模块88，分别对解映射后的二进制比特进行FEC解码，恢复二进制比特流信号。
基于上述系统，提供一个DSP处理的使用实施例。本实施例中， 波特率为32Gbaud，采用的调制格式为偏振复用QPSK，即PDM-QPSK信号，其星座图和比特映射方式如图9所示。偏振分集预编码矩阵为
Figure PCTCN2020138778-appb-000007
前馈均衡器83所用的抽头数量为7个，自适应均衡模块85采用单抽头自适应滤波器，抽头自适应更新准为级联多模算法(CMMA)，抽头系数更新周期等于每32个符号周期。
基于上述系统，提供另一个DSP处理的使用实施例。本实施例中，波特率为42Gbaud，所采用的调制格式为偏振复用8QAM，即PDM-8QAM信号，其星座图和比特映射方式见图10。图10表示一种特殊8QAM，它是16QAM星座点的一个子集，其区别于常的方形或者环形的8QAM，因为它的最小欧氏距离最大，因而有更好的AWGN下的BER性能。比特到符号的映射关系就规定了二进制比特到复信号码元的唯一确定的对应方法。偏振分集预编码矩阵为
Figure PCTCN2020138778-appb-000008
Figure PCTCN2020138778-appb-000009
前馈均衡时所用的抽头数量为9个，单抽头自适应滤波器的抽头自适应更新准为多模算法(MMA)，抽头系数更新周期等于每16个符号周期。
本发明不局限于上述实施方式，对于本技术领域的普通技术人员来说，在不脱离本发明原理的前提下，还可以做出若干改进和润饰，这些改进和润饰也视为本发明的保护范围之内。本说明书中未作详细描述的内容属于本领域专业技术人员公知的现有技术。

Claims (15)

  1. 一种光互连通信方法，其特征在于，包括：
    将待传送的二进制比特流信号经DSP处理后变为模拟电信号；采用激光器输出两路光，一路作为集成双偏相干光发射机的光载波将所述模拟电信号调制为复光场信号，另一路作为本振光进行时延调整；
    通过偏振无关相干光接收机对所述复光场信号和所述本振光进行相干混频，其中，通过反馈扰偏控制所述本振光的偏振态，避免本振光的偏振态落入X和Y方向；混频后的电信号通过DSP处理恢复所述二进制比特流信号。
  2. 如权利要求1所述的光互连通信方法，其特征在于，所述待传送的二进制比特流信号经数字信号处理包括：
    待传送的二进制比特流信号进行FEC编码后，对X、Y两个偏振态进行独立的星座图映射，生成两路独立的复数信号流，再进行偏振分集预编码，预编码后的两路信号经正交分离后进行时域预补偿，补偿后的信号以波特率为采样率进行数模转换，得到所述模拟电信号。
  3. 如权利要求2所述的光互连通信方法，其特征在于：所述星座图映射生成的复数信号流为[X 1,Y 1] T，所述预编码后的符号流为[X 2,Y 2] T，预编码的规则为[X 2,Y 2] T＝H·[X 1,Y 1] T，
    Figure PCTCN2020138778-appb-100001
    为预编码矩阵，且满足ad-bc≠0。
  4. 如权利要求3所述的光互连通信方法，其特征在于，混频后的电信号通过数字信号处理包括：
    混频后的电信号模数转换为数字信号并进行抗混叠滤波，滤波后的信号进行前馈均衡和时钟同步，再进行单抽头自适应均衡实现偏振解复用，然后进行解预编码、解映射和FEC解码，得到所述二进制比特流信号。
  5. 如权利要求4所述的光互连通信方法，其特征在于：所述偏振解复用得到的两路信号为[A 1,B 1] T，利用一个矩阵对[A 1,B 1] T进行解偏振分集预编码处理得到[A 2,B 2] T，所述解偏振分集预编码的规则为：[A 2,B 2] T＝H‘·[A 1,B 1] T，其中H‘为预编码矩阵H的逆矩阵，
    Figure PCTCN2020138778-appb-100002
  6. 如权利要求1所述的光互连通信方法，其特征在于，所述反馈扰偏控制所述本振光的偏振态包括：
    所述偏振无关相干光接收机将收到的本振光分为两个偏振态互相垂直的支路，每个支路的本振光均按照不同功率比例分为两路，提取每个支路本振光分出的光功率较小的一路，并获得提取的两路本振光的功率差，并将所述功率差转换为光电流幅度，当所述光电流幅度的绝对值大于预设阈值时，通过一个与所述绝对值大小成正比的控制信号，对本振光进行扰动。
  7. 如权利要求1-6任一项所述的光互连通信方法，其特征在于：
    所述光互连通信方法适用调制格式包括QPSK、8QAM、16QAM、32QAM以及64QAM；
    所述光互连通信方法适用信息速率包括100G、200G、400G、600G以及800G。
  8. 一种光互连通信系统，其特征在于，包括：
    激光器，用于输出连续光；
    第一分光器，用于接收所述连续光并分为两路，一路提供光载波，一路作为本振光；
    发端DSP芯片，用于将待传送的二进制比特流信号经DSP处理后变为模拟电信号；
    集成双偏相干光发射机，用于接收所述光载波，将所述模拟电信 号调制为复光场信号；
    第一光纤通道，用于传输所述集成双偏相干光发射机输出的复光场信号；
    粗可调延时线，用于粗调所述本振光传输线路的时延，以匹配复光场信号传输链路长度；
    第二光纤通道，用于传输所述粗可调延时线调整后的本振光；
    偏振无关相干光接收机，用于接收分别由第一光纤通道和第二光纤通道传输后的所述复光场信号和所述本振光并进行相干混频；所述偏振无关相干光接收机包括扰偏器，用于通过反馈扰偏控制所述本振光的偏振态，避免本振光的偏振态落入X和Y方向；
    收端DSP芯片，用于接收混频后的电信号，并通过DSP处理恢复所述二进制比特流信号。
  9. 如权利要求8所述的光互连通信系统，其特征在于：所述激光器为DFB激光器；第一分光器2的分光比例为7：3；粗可调延时线采用单模光纤实现；第一光纤信道和第二光纤信道均为普通单模光纤。
  10. 如权利要求8所述的光互连通信系统，其特征在于，所述偏振无关相干光接收机包括：
    第一偏振分束器，用于将所述复光场信号分为两个偏振态互相垂直的支路；
    精可调延时线，用于精确调整本振光相对于信号光的传输时延和光程差，确保信号光和本振光满足相干长度以内；
    第二偏振分束器，用于将所述本振光分为两个偏振态互相垂直的支路；
    扰偏器，其接收精可调延时线调整后的本振光，控制所述本振光 输入到第二偏振分束器的偏振态，避免进入第二偏振分光器的本振光恰好为水平或垂直方向；
    第二分光器和第三分光器，用于将两个具有偏振态的本振光进行不同功率比例地分路；
    第五平衡探测器，用于接收第二分光器和第三分光器分别分出的功率较小的一路本振光，获得功率差，并转换为光电流幅度输出；
    控制器，用于所述光电流幅度转换为本振光偏振态的扰动控制信号，控制所述扰偏器；
    混频增益模块，用于将第一偏振分束器输出的两路复光场信号分别与第二分光器、第三分光器分出的功率较大的一路本振光进行相干混频，再分别得到X、Y偏振的实部和虚部，进行放大并实现自动增益控制。
  11. 如权利要求10所述的光互连通信系统，其特征在于，所述混频增益模块包括：
    第一90°混频器，用于将第一偏振分束器输出的一路复光场信号与第二分光器分出的功率较大的一路本振光进行相干混频；
    第二90°混频器，用于将第一偏振分束器输出的另一路复光场信号与第三分光器分出的功率较大的一路本振光进行相干混频；
    第一平衡探测器和第二平衡探测器，用于将第一90°混频器输出的光信号转为电信号，分别得到X偏振的实部和虚部；
    第三平衡探测器和第四平衡探测器，用于将第二90°混频器输出的光信号转为电信号，分别得到Y偏振的实部和虚部；
    跨阻放大器，用于将所述X偏振的实部和虚部、所述Y偏振的实部和虚部进行放大并实现自动增益控制。
  12. 如权利要求10所述的光互连通信系统，其特征在于：
    所述第二分光器和第三分光器的比例相同，均为95：5；
    所述第五平衡探测器输出的光电流幅度的绝对值大于预设阈值时，所述输出一个与所述绝对值大小成正比的控制电信号作用于所述扰偏器，所述阈值为第五平衡探测器所探测到最大光电流的90％。
  13. 如权利要求8所述的光互连通信系统，其特征在于，所述发端DSP芯片包括：
    FEC编码模块，用于将待传送的二进制比特流信号进行FEC编码；
    星座图映射模块，用于按照预设的调制格式，分别针对X、Y两个偏振态进行独立的星座图映射，生成两路独立的复数信号流[X 1,Y 1] T；
    预编码模块，用于将所述复数信号流[X 1,Y 1] T进行偏振分级预编码，输出两路预编码符号流[X 2,Y 2] T，预编码的规则为：
    [X 2,Y 2] T＝H·[X 1,Y 1] T，
    Figure PCTCN2020138778-appb-100003
    为预编码矩阵，满足ad-bc≠0。
    预补偿模块，用于对预编码后的两路信号经正交分离后进行时域预补偿；
    DAC模块，用于对补偿后的信号以波特率为采样率进行数模转换，得到所述模拟电信号。
  14. 如权利要求13所述的光互连通信系统，其特征在于，所述收端DSP芯片包括：
    ADC模块，其用于将混频后的电信号转换为数字信号；
    两个低通滤波器，分别对X偏振信号和Y偏振信号分别进行抗混叠滤波处理，
    两个前馈均衡器，各对应一个低通滤波器，对滤波后的信号进行 前馈均衡；
    两个时钟恢复模块，分别将每个前馈均衡后的信号恢复最佳采样时钟和相位；
    自适应均衡模块，对时钟恢复后的信号进行自适应均衡，实现偏振解复用，得到均衡后的两路信号[A 1,B 1] T。
    解预编码模块，利用一个矩阵对[A 1,B 1] T进行解偏振分集预编码处理得到[A 2,B 2] T，所述解偏振分集预编码的规则为：[A 2,B 2] T＝H‘·[A 1,B 1] T，其中H‘为预编码矩阵H的逆矩阵；
    两个星座图解映射模块，分别对完成解偏振分集预编码的信号解映射；
    两个FEC解码模块，分别对解映射后的二进制比特进行FEC解码，恢复所述二进制比特流信号。
  15. 如权利要求14述的光互连通信系统，其特征在于：
    所述发端DSP芯片的预补偿模块采用实数域有限冲激响应滤波器进行时域预补偿，有限冲激响应滤波器的抽头个数小于等于3个；
    所述收端DSP芯片的自适应均衡模块采用单抽头2x2复数蝶形滤波器进行自适应均衡；
    所述DAC模块和ADC模块均工作在1倍波特率采样速率下。
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