WO2021082476A1 - 永磁同步电机的控制方法 - Google Patents
永磁同步电机的控制方法 Download PDFInfo
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- WO2021082476A1 WO2021082476A1 PCT/CN2020/097632 CN2020097632W WO2021082476A1 WO 2021082476 A1 WO2021082476 A1 WO 2021082476A1 CN 2020097632 W CN2020097632 W CN 2020097632W WO 2021082476 A1 WO2021082476 A1 WO 2021082476A1
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- stator
- temperature
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- permanent magnet
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/0003—Control strategies in general, e.g. linear type, e.g. P, PI, PID, using robust control
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
- H02P21/141—Flux estimation
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/22—Current control, e.g. using a current control loop
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P25/00—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details
- H02P25/02—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details characterised by the kind of motor
- H02P25/022—Synchronous motors
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P2207/00—Indexing scheme relating to controlling arrangements characterised by the type of motor
- H02P2207/05—Synchronous machines, e.g. with permanent magnets or DC excitation
Definitions
- the invention relates to a control method of a motor, in particular to a control method of a permanent magnet synchronous motor.
- Permanent magnet synchronous motors have been widely used in the field of rail transit because of their high efficiency, energy saving, and high power density.
- the most important performance requirement is to generate accurate torque and achieve high-efficiency control in various environments.
- One of the most important factors affecting the torque accuracy and high-efficiency control of permanent magnet motors is the motor The parameter changes during the operation of the motor, which leads to a mismatch between the motor control algorithm and the controlled motor.
- the changes of motor parameters are mainly caused by the iron core magnetic saturation effect caused by the changes of motor operating temperature and stator current, which results in the changes of stator inductance L d , L q , stator resistance R s and permanent magnet flux linkage ⁇ f in the motor parameters.
- the traditional vector control method is widely used in permanent magnet synchronous motor control, but it does not consider the temperature on the motor parameters R s , ⁇ f , and the iron core saturation effect on L d , L q , the torque accuracy of the motor output And the operating efficiency of the motor will be greatly affected.
- the invention solves the problem that the torque accuracy of the motor output and the motor operating efficiency are affected due to the inaccurate motor parameters used in the existing permanent magnet synchronous motor control method, and proposes a permanent magnet synchronous motor control method.
- the control method The motor parameters can be determined in real time and accurately, thereby reducing the influence of inaccurate motor parameters on motor control.
- a robust decoupling control method for permanent magnet synchronous motors is provided, which realizes the decoupling control of the motor and enhances the robustness of the motor control system.
- the present invention is realized by adopting the following technical scheme: an optimized control method for a permanent magnet synchronous motor.
- the control block diagram includes a resolver module, a temperature sensor module, a stator resistance calculation module, a Clark transformation module, a Park transformation module, and a stator inductance calculation look-up table module.
- the resolver is installed on the permanent magnet synchronous motor, and the rotor position ⁇ of the permanent magnet synchronous motor is measured by the resolver, and the rotor position ⁇ is differentiated to obtain the rotation speed w e of the permanent magnet synchronous motor;
- a temperature sensor is embedded in the stator of the motor, and the real-time stator temperature T of the motor is obtained from the temperature sensor;
- the motor temperature change causes the motor stator resistance R s to change, test and draw the motor stator temperature resistance comparison table, obtain the stator real-time temperature value T through the temperature sensor, and obtain the stator resistance R s by querying the motor stator temperature resistance comparison table (T);
- stator currents i ⁇ and i ⁇ undergo Park transformation to obtain the currents i d and i q in the dq stationary coordinate system;
- the input of the permanent magnet flux calculation module is: current i d , i q , stator resistance R s (T), stator inductance L d (T) and L q (T), the last calculation period (or called the last beat) the stator voltage u 'd, u' q, the previous calculation cycle (upper or a beat) calculated by the permanent magnet flux linkage ⁇ f '(T r), the rotation speed w e.
- the calculation method of the output ⁇ f (T r ) of the permanent magnet flux calculation module is as follows:
- K p_ ⁇ f and K i_ ⁇ f are control parameters, Respectively represent the estimated value of the current, t is the acquisition time, T r represents the actual operating temperature of the permanent magnet;
- K p_ ⁇ f and K i_ ⁇ f The process of debugging to obtain K p_ ⁇ f and K i_ ⁇ f is as follows: Given a small parameter of K i_ ⁇ f (for example, 0.001), first adjust the parameter K p_ ⁇ f to make the output ⁇ f (T r ) in a state of constant amplitude oscillation, and then The parameter K i_ ⁇ f is adjusted to make the output ⁇ f (T r ) converge.
- the parameters K p_ ⁇ f and K i_ ⁇ f obtained at this time are control parameters.
- the input of the torque calculation module is current i d , i q , stator resistance R s (T), stator inductance L d (T) and L q (T), speed w e , ⁇ f (T r );
- the output of the torque calculation module-electromagnetic torque T eb is obtained according to the following calculation formula:
- T ex1 1.5n p i q [(L d (T)-L q (T))i d ]
- n p is the number of pole pairs of the motor.
- T ex2 1.5n p i q [(L d (T 0 )-L q (T 0 ))i d ]
- L d (T 0 ) and L q (T 0 ) are the values of stator inductance under rated conditions.
- T ex T ex1 -T ex2
- T est 1.5n p i q [ ⁇ f (T r )+(L d (T 0 )-L q (T 0 ))i d ]
- T * e1 , T eb , stator inductance L d (T), stator inductance L q (T), ⁇ f (T r ) and n p are the inputs of the current calculation module;
- the given torque T * e1 passes through the calculation module to obtain the given current value I * and the current angle ⁇ (as shown in Figure 3);
- the given torque T * e1 passes through the calculation module to obtain the given current value I * and the calculation process of the current angle ⁇ as follows:
- the relationship between the torque and current of the control algorithm is expressed as:
- the given torque command Change to the standard unit value t en format, and then pass the formula Solve to get the current
- the unit value of i dn can be calculated
- the given current can be calculated
- T * e1 and T eb are passed through the PI regulator, and the output is ⁇ , which is the compensation value for the given current angle.
- the input of the given current generating module is the given current value I * and the current angle ⁇ 1 ;
- the input parameters of the robust decoupling controller are current i d * , i q * , i d , i q , stator resistance R s (T), stator inductance L d (T) and L q (T), permanent magnet magnetism Chain ⁇ f (T r ) and speed w e , the output parameters of which are stator voltage u d , u q ;
- ⁇ x is the control parameter, and the control parameter ⁇ x changes with different modulation strategies.
- the control parameter ⁇ x is proportional to the switching frequency, which is expressed as follows:
- ⁇ b is the reference value of the control parameter, which is selected between 0.1 and 1.0, and f kx is the switching frequency of the inverter;
- ⁇ d and ⁇ q are the key parts of the robust controller, and the calculation formula is as follows:
- ⁇ is the control parameter, and the control parameter ⁇ is selected by trial and error, which can be 80;
- i d1 and i q1 are intermediate variables in the algorithm process, and the calculation formulas of i d1 and i q1 are as follows:
- ⁇ 'd, ⁇ ' q is the previous calculation cycle (or on a beat) calculated variable
- the input of the PWM modulation module is the stator voltage u d , u q , the DC bus voltage u dc , the speed w e and the angle ⁇ ; the output of the PWM modulation module is 6 PWM waves to drive the three-phase inverter bridge module to work.
- the present invention obtains accurate motor parameters stator resistance R s (T), stator inductance L d (T) and L q (T) through online look-up table of motor stator temperature T, current amplitude I s and current vector angle ⁇ ,
- the flux linkage observation model is used to calculate the flux linkage value ⁇ f (T r ) in real time, eliminating the need for rotor temperature detection equipment, improving the accuracy of motor control and decoupling; and using the torque closed loop output results to redistribute Given the stator current, the permanent magnet synchronous motor is kept running on a better trajectory, and the heat and loss of the motor are reduced.
- Figure 1 is a control block diagram of the control method of the present invention
- Figure 2 is a flow chart of the stator inductance calculation look-up table module
- Figure 3 is a control block diagram of the current calculation module
- Figure 4 is a control block diagram of the robust decoupling controller
- Figure 5 is a schematic diagram of the segmented modulation algorithm.
- Permanent magnet synchronous motor control method its control block diagram (as shown in Figure 1) includes resolver module 1, temperature sensor module 2, stator resistance calculation module 3, Clark transformation module 4, Park transformation module 5, stator inductance calculation look-up table Module 6, permanent magnet flux calculation module 7, torque calculation module 8, current calculation module 9, given current generation module 10, robust decoupling controller module 11, PWM modulation module 12, three-phase inverter bridge module 13 ;
- the resolver is installed on the permanent magnet synchronous motor, and the rotor position ⁇ of the permanent magnet synchronous motor is measured by the resolver, and the rotor position ⁇ is differentiated to obtain the rotation speed w e of the permanent magnet synchronous motor;
- a temperature sensor is embedded in the stator of the motor, and the real-time stator temperature T of the motor is obtained from the temperature sensor;
- the motor temperature change causes the motor stator resistance R s to change, test and draw the motor stator temperature resistance comparison table, obtain the stator real-time temperature value T through the temperature sensor, and obtain the stator resistance R s by querying the motor stator temperature resistance comparison table (T);
- stator currents i ⁇ and i ⁇ undergo Park transformation to obtain the currents i d and i q in the dq stationary coordinate system;
- stator inductance L d and L q The change of the stator current will cause the magnetic saturation effect of the stator core. With the change of the d and q axis current, the stator inductance L d and L q will change. At the same time, the stator temperature T of the motor will also affect the stator inductance. In order to obtain more accurate stator inductance parameters L d and L q , a look-up table method is used to obtain stator inductances L d and L q ;
- stator current and stator current I S ⁇ S phase were low-pass filtered, and the obtained value I SLPF ⁇ SLPF filtered.
- the table corresponding to each temperature point reflects the changes in stator inductance L d and L q with I S and ⁇ S (or I SLPF and ⁇ SLPF ); in specific implementation, the temperature range is [-30°C, 160 °C] interval, and each temperature value of an integer multiple of ten is used as the temperature point.
- the process of looking up the table is as follows: when the stator temperature T collected in real time is not equal to the temperature value of any temperature point, select the two closest temperature points T x , the stator inductance L d of T x +10 to the stator temperature T.
- L d (T) and L q (T) are the current operating conditions (current temperature T, current I S and ⁇ S (or I SLPF and ⁇ SLPF ))
- the stator inductance of the motor is the stator inductance of the motor.
- the permanent magnet material in the permanent magnet synchronous motor rotor is greatly affected by temperature changes, and the relationship between the permanent magnet flux linkage and temperature change of the motor can be expressed as
- ⁇ f (T 0) is a permanent magnet flux of the rated conditions
- ⁇ f (T r) is a permanent magnet flux at the actual motor operating temperature
- T 0 taken as 20 °C
- T r is the actual permanent magnet Operating temperature
- ⁇ is the temperature coefficient of remanence.
- the flux linkage value ⁇ f (T r ) is obtained in real time using the flux linkage observation model.
- the input of the permanent magnet flux linkage calculation module is: current i d , i q, stator resistance R s (T), stator inductance L d (T) and L q (T), the previous calculation cycle (or on a beat) of the stator voltage u 'd, u' q, on a
- the calculation method of the permanent magnet flux linkage ⁇ f '(T r ) calculated by the calculation period (or the last beat), the rotation speed w e , and the output of the permanent magnet flux linkage calculation module ⁇ f (T r ) are calculated as follows:
- K p_ ⁇ f and K i_ ⁇ f are control parameters, Respectively represent the estimated value of the current, t is the acquisition time, T r represents the actual operating temperature of the permanent magnet;
- K p_ ⁇ f and K i_ ⁇ f The process of debugging to obtain K p_ ⁇ f and K i_ ⁇ f is as follows: Given a small parameter of K i_ ⁇ f (for example, 0.001), first adjust the parameter K p_ ⁇ f to make the output ⁇ f (T r ) in a state of constant amplitude oscillation, and then Adjust the parameter K i_ ⁇ f to make the output ⁇ f (T r ) converge.
- the parameters K p_ ⁇ f and K i_ ⁇ f obtained at this time are control parameters;
- the input of the torque calculation module is current i d , i q , stator resistance R s (T), stator inductance L d (T) and L q (T), speed w e , ⁇ f (T r );
- the output of the torque calculation module-electromagnetic torque T eb is obtained according to the following calculation formula:
- T ex1 1.5n p i q [(L d (T)-L q (T))i d ]
- n p is the number of pole pairs of the motor
- T ex2 1.5n p i q [(L d (T 0 )-L q (T 0 ))i d ]
- L d (T 0 ) and L q (T 0 ) are the values of stator inductance under rated conditions
- T ex T ex1 -T ex2
- T est 1.5n p i q [ ⁇ f (T r )+(L d (T 0 )-L q (T 0 ))i d ]
- T * e1 , T eb , stator inductance L d (T) and L q (T), ⁇ f (T r ) and n p are the inputs of the current calculation module;
- the given torque T * e1 passes through the calculation module to obtain the given current value I * and the current angle ⁇ (as shown in Figure 3);
- the given torque T * e1 passes through the calculation module to obtain the given current value I * and the calculation process of the current angle ⁇ as follows:
- the relationship between the torque and current of the control algorithm is expressed as:
- the given torque Change to the standard unit value t en format, and then pass the formula Solve to get the current
- the unit value of i dn can be calculated
- the given current can be calculated
- T * e1 and T eb are passed through the PI regulator, and the output is ⁇ , which is the compensation value for the given current angle.
- the input of the given current generating module is the given current value I * and the current angle ⁇ 1 ;
- the robust decoupling controller realizes the decoupling control of the permanent magnet synchronous motor, and on the other hand, it improves the anti-interference function of the motor control.
- Its input parameters are current i d * , i q * , i d , i q , stator resistance R s (T), stator inductance L d (T) and L q (T), permanent magnet flux ⁇ f (T r ) and speed w e , the output parameters of which are stator voltage u d , u q ;
- the robust decoupling controller is shown in Figure 4.
- ⁇ x is the control parameter, and the control parameter ⁇ x changes with different modulation strategies.
- the control parameter ⁇ x is proportional to the switching frequency, which is expressed as follows:
- ⁇ b is the reference value of the control parameter, which can be selected between 0.1 and 1.0 (for example, 0.1, 0.2, 0.3, 0.4, 0.5, 0.7, 0.9, 1.0; 0.5 is preferred), and f kx is the switch of the inverter frequency;
- ⁇ d and ⁇ q are the key parts of the robust controller, and the calculation formula is as follows:
- ⁇ is the control parameter
- the control parameter ⁇ is selected by trial and error method, which can be 80.
- i d1 and i q1 are intermediate variables in the algorithm process, and the calculation formulas of i d1 and i q1 are as follows:
- ⁇ 'd, ⁇ ' q is the previous calculation cycle (or on a beat) calculated variable
- the input of the PWM modulation module is the stator voltage u d , u q , the DC bus voltage u dc , the speed w e and the angle ⁇ .
- the output of the PWM modulation module is 6 PWM waves, which drive the three-phase inverter bridge module to work.
- Segment modulation is divided into asynchronous modulation and synchronous modulation. Synchronous modulation is limited by the switching frequency and can be divided into multiple segments, and finally enters into square wave modulation. Under square wave modulation, the voltage utilization rate is high and the harmonics are small.
- the available modulation algorithms include SPWM modulation, specific sub-harmonic elimination PWM (SHEPWM) modulation, etc.
- SHEPWM specific sub-harmonic elimination PWM
- the control method proposed in the present invention can be implemented in a DSP chip.
- the permanent magnet synchronous motor control is implemented by a two-level interrupt.
- the motor control algorithm is run in the first-level interrupt (ie steps 1-11), and the design control cycle is 250us; another
- the motor PWM modulation algorithm is run in the level interrupt, and the interrupt period is related to the current modulation strategy and the motor operating frequency.
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Abstract
本发明涉及交流电机的控制方法,具体为永磁同步电机的控制方法。解决现有永磁同步电机的控制方法中因使用的电机参数不准确造成电机输出的转矩精度和电机运行效率被影响的问题。该控制方法通过电机定子温度T,使用在线查表得到定子电阻R s(T);通过电机定子温度T、电流幅值I s、电流矢量角β,使用在线查表得到定子电感L d(T)和L q(T);使用磁链观测模型实时计算出磁链值Ψ f(T r),从而提高了电机控制和解耦的准确性;利用转矩闭环的输出结果重新分配了给定的定子电流,维持了永磁同步电机按照较优的控制轨迹运行,降低了电机的发热和损耗。实现了电机的解耦控制、增强了电机控制系统的鲁棒性。
Description
本发明涉及电机的控制方法,具体为永磁同步电机的控制方法。
永磁同步电机以其具有高效、节能、高功率密度等优势在轨道交通领域得到了广泛的应用。对于永磁同步电机而言,最重要的性能要求是在各种环境下产生准确转矩并实现高效率的控制,而影响永磁电机转矩精度和高效率的控制的一个最重要因素是电机参数在电机运行过程中的变化,这导致了电机控制算法与被控电机的不匹配。电机参数的变化主要由电机工作温度和定子电流变化引起的铁芯磁饱和效应引起,从而造成电机参数中的定子电感L
d、L
q、定子电阻R
s和永磁体磁链Ψ
f的变化。传统的矢量控制方法在永磁同步电机控制中被广泛使用,但其没有考虑温度对电机参数R
s、Ψ
f,以及铁芯饱和效应对L
d、L
q的影响,电机输出的转矩精度和电机运行效率将很大程度上被影响。
发明内容
本发明解决现有永磁同步电机的控制方法中因使用的电机参数不准确造成电机输出的转矩精度和电机运行效率被影响的问题,提出一种永磁同步电机的控制方法,该控制方法可以实时而准确的确定电机参数,从而降低电机参数不准确对电机控制的影响。同时,在准确的确定电机参数的基础上,提供一种永磁同步电机鲁棒解耦控制方法,实现了电机的解耦控制、增强了电机控制系统的鲁棒性。
本发明是采用如下技术方案实现的:永磁同步电机的优化控制方法,其控制框图包括旋转变压器模块、温度传感器模块、定子电阻计算模块、Clark变换模块、Park变换模块、定子电感计算查表模块、永磁体磁链计算模块、转矩计算模块、电流计算模块、给定电流生成模块、鲁棒解耦控制器模块、PWM调制模块、三相逆变桥模块;
1)旋转变压器模块
旋转变压器安装在永磁同步电机上,通过旋转变压器测量得到永磁同步电机的转子位置θ,转子位置θ经过微分,得到永磁同步电机的转速w
e;
2)温度传感器模块
电机定子中埋有温度传感器,由温度传感器得到电机的实时定子温度T;
3)定子电阻计算模块
电机温度变化导致电机定子电阻R
s的变化,测试并绘制电机定子的温度阻值对照表,通过温度传感器获取定子实时温度值T,并通过查询电机定子的温度阻值对照表得到定子电阻R
s(T);
4)Clark变换模块
采集两相定子电流i
a、i
b,经过Clark变换得到定子电流i
α、i
β;
5)Park变换模块
定子电流i
α、i
β经过Park变换得到d-q静止坐标系下的电流i
d、i
q;
6)定子电感计算查表模块
定子电感L
d和L
q与电机定子温度T、定子电流的幅值I
S和定子电流的相位β
S三个变量相关;
先制作在一定温度区间内的不同温度点下,定子电感L
d、L
q随I
S和β
S变化的表格;
随后用实时采集到的定子温度T、定子电流的幅值I
S和定子电流的相位β
S,通过查询表格,得到定子电感L
d(T)、L
q(T);
7)永磁体磁链计算模块
永磁体磁链计算模块的输入为:电流i
d、i
q,定子电阻R
s(T),定子电感L
d(T)和L
q(T),上一计算周期(或称上一拍)的定子电压u’
d,u’
q,上一计算周期(或称上一拍)计算得到的永磁体磁链Ψ
f’(T
r),转速w
e。永磁体磁链计算模块的输出Ψ
f(T
r)计算方式如下:
调试得到K
p_Ψf和K
i_Ψf的过程如下:给定K
i_Ψf一个较小的参数(例如0.001),先调节参数K
p_Ψf使输出得到的Ψ
f(T
r)处于等幅震荡的状态,这时再调节参数K
i_Ψf,使输出Ψ
f(T
r)收敛,这时得到的参数K
p_Ψf和K
i_Ψf是控制参数。
8)转矩计算模块
转矩计算模块的输入为电流i
d、i
q,定子电阻R
s(T),定子电感L
d(T)和L
q(T),转速w
e,Ψ
f(T
r);
转矩计算模块的输出——电磁转矩T
eb按如下计算公式得到:
T
ex1=1.5n
pi
q[(L
d(T)-L
q(T))i
d]
式中,n
p是电机的极对数。
T
ex2=1.5n
pi
q[(L
d(T
0)-L
q(T
0))i
d]
式中,L
d(T
0)和L
q(T
0)分别是在额定工况下的定子电感的值。
T
ex=T
ex1-T
ex2
T
est=1.5n
pi
q[ψ
f(T
r)+(L
d(T
0)-L
q(T
0))i
d]
T
eb=T
ex+T
est
9)电流计算模块
目标转矩T
*
e经过限幅和斜坡处理后,得到给定转矩T
*
e1;
T
*
e1、T
eb、定子电感L
d(T)、定子电感L
q(T)、Ψ
f(T
r)和n
p为电流计算模块的输入;
给定转矩T
*
e1经过计算模块,得到给定电流值I
*和电流角度β(如图3所示);
给定转矩T
*
e1经过计算模块,得到给定电流值I
*和电流角度β的计算过程如下:
通过运算得到标幺值基值t
eb和i
bx,其中i
bx是电流的标幺值基值,通过i
bx=ψ
f(T
r)/(L
q(T)-L
d(T))计算得到;t
eb是转矩的标幺值基值,可以通过t
eb=n
pψ
f(T
r)i
bx计算得到;标幺值基值随着电机参数的变化而变化。
在标幺值的形式下,控制算法的转矩和电流的关系表示为:
通过将给定转矩指令
变为标幺值t
en的格式,再通过公式
求解得到电流
的标幺值i
dn,最后再通过式
可计算得到给定电流
电流
的标幺值i
qn可通过公式t
en=i
qn(1-i
dn)计算得到,此时t
en和i
dn是已知量,再通过式
可计算得到给定电流
T
*
e1与T
eb的差经过PI调节器,输出为Δβ,Δβ是对给定电流角度的补偿值,通过辨识实时转矩与给定转矩进行对比,对给定电流角度进行校正;
最终得到电流角度β
1通过如下公式计算得到:
β
1=β+Δβ
10)给定电流生成模块
给定电流生成模块的输入为给定电流值I
*和电流角度β
1;
计算过程如下:
11)鲁棒解耦控制器
鲁棒解耦控制器的输入参数有电流i
d
*、i
q
*、i
d、i
q、定子电阻R
s(T)、定子电感L
d(T)和L
q(T)、永磁体磁链Ψ
f(T
r)和转速w
e,其输出参数为定子电压u
d、u
q;
控制算法中间变量u
d1、u
q1计算过程如下:
式中,β
x是控制参数,控制参数β
x随着调制策略的不同而变化,控制参数β
x与开关频率成一定的比例,其表示如下:
β
x=β
b×f
kx
式中,β
b是控制参数基准值,在0.1-1.0之间选择,f
kx是逆变器的开关频率;
通过以上的控制算法计算得到u
d1、u
q1,实现了永磁同步电机的解耦控制;
控制算法中间变量u
d2、u
q2计算过程如下:
式中,δ
d、δ
q是鲁棒控制器的关键部分,其计算公式如下:
式中,λ为控制参数,控制参数λ采用试凑法选取,可取80;i
d1、i
q1为算法过程中的中间变量,i
d1、i
q1的计算公式如下:
其中,δ’
d、δ’
q是上一计算周期(或称上一拍)计算得到的变量;
电压u
d、u
q的计算过程如下:
这里为了增强控制系统的稳定性,实现控制解耦,在dq轴上分别加入项R
S(T)×k
R×i
d和R
S(T)×k
R×i
q+Ψ
f(T
r)w
e,K
R为阻尼系数,取为0.9;
12)PWM调制模块
PWM调制模块的输入为定子电压u
d、u
q,直流母线电压u
dc,转速w
e和角 度θ;PWM调制模块的输出为6路PWM波,驱动三相逆变桥模块工作。
本发明带来的有益效果:
(1)本发明通过电机定子温度T、电流幅值I
s、电流矢量角β在线查表得到准确的电机参数定子电阻R
s(T),定子电感L
d(T)和L
q(T),使用磁链观测模型实时计算出磁链值Ψ
f(T
r),省去了对转子温度的检测设备,提高了电机控制和解耦的准确性;且利用转矩闭环的输出结果重新分配了给定的定子电流,保持永磁同步电机在较优的轨迹运行,降低了电机的发热和损耗。
(2)通过在电流环中使用鲁棒解耦控制方法,提高了系统控制的抗干扰性能,并实现了永磁同步电机的解耦控制。
图1为本发明所述控制方法的控制框图;
图2为定子电感计算查表模块的流程框图;
图3为电流计算模块控制框图;
图4为鲁棒解耦控制器的控制框图;
图5为分段调制算法示意图。
永磁同步电机的控制方法,其控制框图(如图1所示)包括旋转变压器模块1、温度传感器模块2、定子电阻计算模块3、Clark变换模块4、Park变换模块5、定子电感计算查表模块6、永磁体磁链计算模块7、转矩计算模块8、电流计算模块9、给定电流生成模块10、鲁棒解耦控制器模块11、PWM调制模块12、三相逆变桥模块13;
1)旋转变压器模块
旋转变压器安装在永磁同步电机上,通过旋转变压器测量得到永磁同步电机的转子位置θ,转子位置θ经过微分,得到永磁同步电机的转速w
e;
2)温度传感器模块
电机定子中埋有温度传感器,由温度传感器得到电机的实时定子温度T;
3)定子电阻计算模块
电机温度变化导致电机定子电阻R
s的变化,测试并绘制电机定子的温度阻值对照表,通过温度传感器获取定子实时温度值T,并通过查询电机定子的温度阻 值对照表得到定子电阻R
s(T);
4)Clark变换模块
采集两相定子电流i
a、i
b,经过Clark变换得到定子电流i
α、i
β;
5)Park变换模块
定子电流i
α、i
β经过Park变换得到d-q静止坐标系下的电流i
d、i
q;
6)定子电感计算查表模块
定子电流的变化会引起定子铁芯的磁饱和效应,随着d、q轴电流的变化,定子电感L
d、L
q都会发生变化,同时电机的定子温度T也会对定子电感产生影响。为了得到较为准确的定子电感参数L
d和L
q,采用查表法来得到定子电感L
d和L
q;
定子电感L
d和L
q与电机定子温度T、定子电流的幅值I
S和定子电流的相位β
S三个变量相关(如图2所示);
其中:
考虑到i
q为0时,程序计算时会出现问题,因此给分母加一个特别小的数字k
α,k
α可以等于0.0000001;
进一步地,定子电流的幅值I
S和定子电流的相位β
S分别经过低通滤波,得到滤波后的值I
SLPF和β
SLPF。
先制作在一定温度区间内的不同温度点下,定子电感L
d、L
q随I
S和β
S(或I
SLPF和β
SLPF)变化的表格(表格数量与温度区间内所设温度点的数量一致,与每个温度点对应的表格反映定子电感L
d、L
q随I
S和β
S(或I
SLPF和β
SLPF)变化的内容);具体实施时,温度区间为[-30℃,160℃]区间,并以每个十的整数倍的温度值作为温度点,这样,在[-30℃,160℃]区间,有二十个温度点:-30℃、-20℃、-10℃、0℃、10℃、20℃、……、150℃、160℃;具体的制作表格的过程为:测试的永磁同步电机和对拖电机同轴或通过齿轮箱连接,将对拖电机运行在额定转速以下,在转速稳定且定子温度在某一温度点(以-20℃为例)的情况下,给定永磁同步电机不同的电流指令I
S,I
S>0和β
S,90°<β
S<180°(或I
SLPF,I
SLPF>0和β
SLPF,90°<β
SLPF<180°)(电流i
*
d、i
*
q),用高频注入法测得与不同的I
S和β
S(或 I
SLPF和β
SLPF)对应的电机定子电感L
d、L
q;改变定子温度,从而获得多组不同温度点下的定子电感L
d、L
q随I
S和β
S(或I
SLPF和β
SLPF)变化的表格。
随后用实时采集到的定子温度T、定子电流的幅值I
S和定子电流的相位β
S(或I
SLPF和β
SLPF),通过查询表格,得到实时的定子电感L
d(T)、L
q(T);查表的过程如下:实时采集的定子温度T不等于任何温度点的温度值时,选择离定子温度T最近的两个温度点T
x、T
x+10的定子电感L
d、L
q随I
S和β
S(或I
SLPF和β
SLPF)变化的表格作为查表使用的第一表格和第二表格(如,定子温度T为23℃,选择温度点T
x=20℃和T
x+10=30℃的表格作为第一和第二表格);实时采集的定子温度T等于某温度点的温度值时,选择与定子温度T相等的温度点T
x的定子电感L
d、L
q随I
S和β
S(或I
SLPF和β
SLPF)变化的表格,和该温度点增温侧相邻的温度点T
x+10的定子电感L
d、L
q随I
S和β
S(或I
SLPF和β
SLPF)变化的表格,作为查表使用的第一表格和第二表格;根据采集的定子电流的幅值I
S和定子电流的相位β
S(或I
SLPF和β
SLPF),在第一表格里查到L
d1(T)和L
q1(T),在第二表格里查到L
d2(T)和L
q2(T);按如下公式得到L
d(T)和L
q(T):
L
d(T)和L
q(T)为当前工况下(当前温度T,当前I
S和β
S(或I
SLPF和β
SLPF))
的电机定子电感值。
7)永磁体磁链计算模块
永磁同步电机转子中永磁材料受温度变化影响大,电机永磁体磁链与温度的变化关系可以表示为
其中:Ψ
f(T
0)是额定工况下的永磁体磁链,Ψ
f(T
r)是电机实际运行温度下的永磁体磁链,T
0可取为20℃,T
r是永磁体实际工作温度,α是剩磁的温度系数。
由于电机的永磁体安装在电机的转子上,无法得到其实际工作温度,因此采用磁链观测模型实时得到磁链值Ψ
f(T
r),永磁体磁链计算模块的输入为:电流i
d、i
q,定子电阻R
s(T),定子电感L
d(T)和L
q(T),上一计算周期(或称上一拍)的 定子电压u’
d,u’
q,上一计算周期(或称上一拍)计算得到的永磁体磁链Ψ
f’(T
r),转速w
e,永磁体磁链计算模块的输出Ψ
f(T
r)计算方式如下:
调试得到K
p_Ψf和K
i_Ψf的过程如下:给定K
i_Ψf一个较小的参数(例如0.001),先调节参数K
p_Ψf使输出得到的Ψ
f(T
r)处于等幅震荡的状态,这时再调节参数K
i_Ψf,使输出Ψ
f(T
r)收敛,这时得到的参数K
p_Ψf和K
i_Ψf是控制参数;
8)转矩计算模块
转矩计算模块的输入为电流i
d、i
q,定子电阻R
s(T),定子电感L
d(T)和L
q(T),转速w
e,Ψ
f(T
r);
转矩计算模块的输出——电磁转矩T
eb按如下计算公式得到:
T
ex1=1.5n
pi
q[(L
d(T)-L
q(T))i
d]
式中,n
p是电机的极对数;
T
ex2=1.5n
pi
q[(L
d(T
0)-L
q(T
0))i
d]
式中,L
d(T
0)和L
q(T
0)分别是在额定工况下的定子电感的值;
T
ex=T
ex1-T
ex2
T
est=1.5n
pi
q[ψ
f(T
r)+(L
d(T
0)-L
q(T
0))i
d]
T
eb=T
ex+T
est
9)电流计算模块
目标转矩T
*
e经过限幅和斜坡处理后,得到给定转矩T
*
e1;
T
*
e1、T
eb、定子电感L
d(T)和L
q(T)、Ψ
f(T
r)和n
p为电流计算模块的输入;
给定转矩T
*
e1经过计算模块,得到给定电流值I
*和电流角度β(如图3所示);
给定转矩T
*
e1经过计算模块,得到给定电流值I
*和电流角度β的计算过程如下:
通过运算得到标幺值基值t
eb和i
bx,其中i
bx是电流的标幺值基值,通过i
bx=ψ
f(T
r)/(L
q(T)-L
d(T))计算得到;t
eb是转矩的标幺值基值,可以通过t
eb=n
pψ
f(T
r)i
bx计算得到;标幺值基值随着电机参数的变化而变化。
在标幺值的形式下,控制算法的转矩和电流的关系表示为:
通过将给定转矩
变为标幺值t
en的格式,再通过公式
求解得到电流
的标幺值i
dn,最后再通过式
可计算得到给定电流
电流
的标幺值i
qn可通过公式t
en=i
qn(1-i
dn)计算得到,此时t
en和i
dn是已知量,再通过式
可计算得到给定电流
T
*
e1与T
eb的差经过PI调节器,输出为Δβ,Δβ是对给定电流角度的补偿值,通过辨识实时转矩与给定转矩进行对比,对给定电流角度进行校正;
最终得到电流角度β
1通过如下公式计算得到:
β
1=β+Δβ
10)给定电流生成模块
给定电流生成模块的输入为给定电流值I
*和电流角度β
1;
计算过程如下:
11)鲁棒解耦控制器
鲁棒解耦控制器一方面实现永磁同步电机的解耦控制,另一方面提高电机控制的抗干扰功能,其输入参数有电流i
d
*、i
q
*、i
d、i
q、定子电阻R
s(T)、定子电感L
d(T)和L
q(T)、永磁体磁链Ψ
f(T
r)和转速w
e,其输出参数有定子电压u
d、u
q;
鲁棒解耦控制器如图4所示。
控制算法中间变量u
d1、u
q1计算过程如下:
式中,β
x是控制参数,控制参数β
x随着调制策略的不同而变化,控制参数β
x与开关频率成一定的比例,其表示如下:
β
x=β
b×f
kx
式中,β
b是控制参数基准值,在0.1-1.0之间选择(如,0.1、0.2、0.3、0.4、0.5、0.7、0.9、1.0;其中优选0.5),f
kx是逆变器的开关频率;
通过以上的控制算法计算得到u
d1、u
q1,实现了永磁同步电机的解耦控制;
控制算法中间变量u
d2、u
q2计算过程如下:
式中,δ
d、δ
q是鲁棒控制器的关键部分,其计算公式如下:
式中,λ为控制参数,控制参数λ采用试凑法选取,可取80。i
d1、i
q1为算法过程中的中间变量,i
d1、i
q1的计算公式如下:
其中,δ’
d、δ’
q是上一计算周期(或称上一拍)计算得到的变量;
电压u
d、u
q的计算过程如下:
这里为了增强控制系统的稳定性,实现控制解耦,在dq轴上分别加入项R
S(T)×k
R×i
d和R
S(T)×k
R×i
q+Ψ
f(T
r)w
e,K
R为阻尼系数,可以取为0.9;
12)PWM调制模块
PWM调制模块的输入为定子电压u
d、u
q,直流母线电压u
dc,转速w
e和角度θ。PWM调制模块的输出为6路PWM波,驱动三相逆变桥模块工作。
因散热等条件的制约,大功率永磁同步电机的调制算法受到开关频率的限制,一般采用多种调制方式相结合的分段调制策略。分段调制策略的示意图如图5所示。
分段调制分为异步调制和同步调制,同步调制受到开关频率的限制可分为多段,最终进入方波调制。方波调制下,电压利用率高、谐波小。同步分段调制算法中,可采用的调制算法有SPWM调制、特定次谐波消除PWM(SHEPWM)调制等,各种调制算法有其优缺点和适用范围。
本发明中提出的控制方法,可以在DSP芯片中实现,永磁同步电机控制由两级中断实现,一级中断中运行电机控制算法(即步骤1-11),设计控制周期为250us;另一级中断中运行电机PWM调制算法,其中断周期与当前的调制策略与电机运行频率相关。
Claims (10)
- 永磁同步电机的控制方法,其特征在于,其控制框图包括旋转变压器模块(1)、温度传感器模块(2)、定子电阻计算模块(3)、Clark变换模块(4)、Park变换模块(5)、定子电感计算查表模块(6)、永磁体磁链计算模块(7)、转矩计算模块(8)、电流计算模块(9)、给定电流生成模块(10)、鲁棒解耦控制器模块(11)、PWM调制模块(12)、三相逆变桥模块(13);1)旋转变压器模块旋转变压器安装在永磁同步电机上,通过旋转变压器测量得到永磁同步电机的转子位置θ,转子位置θ经过微分,得到永磁同步电机的转速w e;2)温度传感器模块电机定子中埋有温度传感器,由温度传感器得到电机的实时定子温度T;3)定子电阻计算模块电机温度变化导致电机定子电阻R s的变化,测试并绘制电机定子的温度阻值对照表,通过温度传感器获取定子实时温度值T,并通过查询电机定子的温度阻值对照表得到定子电阻R s(T);4)Clark变换模块采集两相定子电流i a、i b,经过Clark变换得到定子电流i α、i β;5)Park变换模块定子电流i α、i β经过Park变换得到d-q静止坐标系下的电流i d、i q;6)定子电感计算查表模块定子电感L d和L q与电机定子温度T、定子电流的幅值I S和定子电流的相位β S三个变量相关;先制作在一定温度区间内的不同温度点下,定子电感L d、L q随I S和β S变化的表格;随后用实时采集到的定子温度T、定子电流的幅值I S和定子电流的相位β S,通过查询表格,得到定子电感L d(T)、L q(T);7)永磁体磁链计算模块永磁体磁链计算模块的输入为:电流i d、i q,定子电阻R s(T),定子电感L d(T)和L q(T),上一计算周期的定子电压u’ d,u’ q,上一计算周期计算得到的永磁体磁 链Ψ’ f(T r),转速w e;永磁体磁链计算模块的输出Ψ f(T r)计算方式如下:调试得到K p_Ψf和K i_Ψf的过程如下:给定K i_Ψf一个较小的参数(例如0.001),先调节参数K p_Ψf使输出得到的Ψ f(T r)处于等幅震荡的状态,这时再调节参数K i_Ψf,使输出Ψ f(T r)收敛,这时得到的参数K p_Ψf和K i_Ψf是控制参数;8)转矩计算模块转矩计算模块的输入为电流i d、i q,定子电阻R s(T),定子电感L d(T)和L q(T),转速w e,Ψ f(T r);转矩计算模块的输出——电磁转矩T eb按如下计算公式得到:T ex1=1.5n pi q[(L d(T)-L q(T))i d]式中,n p是电机的极对数;T ex2=1.5n pi q[(L d(T 0)-L q(T 0))i d]式中,L d(T 0)和L q(T 0)分别是在额定工况下的定子电感的值;T ex=T ex1-T ex2T est=1.5n pi q[ψ f(T r)+(L d(T 0)-L q(T 0))i d]T eb=T ex+T est9)电流计算模块目标转矩T * e经过限幅和斜坡处理后,得到给定转矩T * e1;T * e1、T eb、定子电感L d(T)、定子电感L q(T)、Ψ f(T r)和n p为电流计算模块的输入;给定转矩T * e1经过计算模块,得到给定电流值I *和电流角度β;T * e1与T eb的差经过PI调节器,输出为Δβ;最终得到电流角度β 1通过如下公式计算得到:β 1=β+Δβ10)给定电流生成模块给定电流生成模块的输入为给定电流值I *和电流角度β 1;计算过程如下:11)鲁棒解耦控制器鲁棒解耦控制器的输入参数有电流i d *、i q *、i d、i q、定子电阻R s(T)、定子电感L d(T)和L q(T)、永磁体磁链Ψ f(T r)和转速w e,其输出参数为定子电压u d、u q;控制算法中间变量u d1、u q1计算过程如下:式中,β x是控制参数,其表示如下:β x=β b×f kx式中,β b是控制参数基准值,在0.1-1.0之间选择,f kx是逆变器的开关频率;控制算法中间变量u d2、u q2计算过程如下:式中,δ d、δ q的计算公式如下:式中,λ为控制参数,控制参数λ采用试凑法选取;i d1、i q1为算法过程中的 中间变量,i d1、i q1的计算公式如下:其中,δ’ d、δ’ q是上一计算周期(或称上一拍)计算得到的变量;电压u d、u q的计算过程如下:K R为阻尼系数;12)PWM调制模块PWM调制模块的输入为定子电压u d、u q,直流母线电压u dc,转速w e和角度θ;PWM调制模块的输出为6路PWM波,驱动三相逆变桥模块工作。
- 根据权利要求2所述的永磁同步电机的控制方法,其特征在于,在6)定子电感计算查表模块中:定子电流的幅值I S和定子电流的相位β S分别经过低通滤波,得到滤波后的值I SLPF和β SLPF。
- 根据权利要求2所述的永磁同步电机的控制方法,其特征在于,在6)定子电感计算查表模块中:温度区间为[-30℃,160℃]区间,并以每个十的整数倍的温度值作为温度点,这样,在[-30℃,160℃]区间,有二十个温度点:-30℃、-20℃、-10℃、0℃、10℃、20℃、……、150℃、160℃。
- 根据权利要求3所述的永磁同步电机的控制方法,其特征在于,在6)定子电感计算查表模块中:温度区间为[-30℃,160℃]区间,并以每个十的整数倍的温度值作为温度点,这样,在[-30℃,160℃]区间,有二十个温度点:-30℃、 -20℃、-10℃、0℃、10℃、20℃、……、150℃、160℃。
- 根据权利要求4所述的永磁同步电机的控制方法,其特征在于,制作表格的过程为:测试的永磁同步电机和对拖电机同轴或通过齿轮箱连接,将对拖电机运行在额定转速以下,在转速稳定且定子温度在某一温度点的情况下,给定永磁同步电机不同的I S,I S>0和β S,90°<β S<180°,用高频注入法测得与不同的I S和β S对应的电机定子电感L d、L q;改变定子温度,从而获得多组不同温度点下的定子电感L d、L q随I S和β S变化的表格。
- 根据权利要求5所述的永磁同步电机的控制方法,其特征在于,制作表格的过程为:测试的永磁同步电机和对拖电机同轴或通过齿轮箱连接,将对拖电机运行在额定转速以下,在转速稳定且定子温度在某一温度点的情况下,给定永磁同步电机不同的I SLPF,I SLPF>0和β SLPF,90°<β SLPF<180°,用高频注入法测得与不同的I SLPF和β SLPF对应的电机定子电感L d、L q;改变定子温度,从而获得多组不同温度点下的定子电感L d、L q随I SLPF和β SLPF变化的表格。
- 根据权利要求6所述的永磁同步电机的控制方法,其特征在于,在6)定子电感计算查表模块中:查表的过程如下:实时采集的定子温度T不等于任何温度点的温度值时,选择离定子温度T最近的两个温度点T x、T x+10的定子电感L d、L q随I S和β S变化的表格作为查表使用的第一表格和第二表格;实时采集的定子温度T等于某温度点的温度值时,选择与定子温度T相等的温度点T x的定子电感L d、L q随I S和β S变化的表格,和该温度点增温侧相邻的温度点T x+10的定子电感L d、L q随I S和β S变化的表格,作为查表使用的第一表格和第二表格;根据采集的定子电流的幅值I S和定子电流的相位β S,在第一表格里查到L d1(T)和L q1(T),在第二表格里查到L d2(T)和L q2(T);按如下公式得到L d(T)和L q(T):
- 根据权利要求7所述的永磁同步电机的控制方法,其特征在于,在6)定子电感计算查表模块中:查表的过程如下:实时采集的定子温度T不等于任何温度点的温度值时,选择离定子温度T最近的两个温度点T x、T x+10的定子电感L d、 L q随I SLPF和β SLPF变化的表格作为查表使用的第一表格和第二表格;实时采集的定子温度T等于某温度点的温度值时,选择与定子温度T相等的温度点T x的定子电感L d、L q随I SLPF和β SLPF变化的表格,和该温度点增温侧相邻的温度点T x+10的定子电感L d、L q随I SLPF和β SLPF变化的表格,作为查表使用的第一表格和第二表格;根据采集的I SLPF和β SLPF,在第一表格里查到L d1(T)和L q1(T),在第二表格里查到L d2(T)和L q2(T);按如下公式得到L d(T)和L q(T):
- 根据权利要求1-9任一个所述的永磁同步电机的控制方法,其特征在于,9)电流计算模块中,给定转矩T * e1经过计算模块,得到给定电流值I *和电流角度β的计算过程如下:通过运算得到标幺值基值t eb和i bx,其中i bx是电流的标幺值基值,通过i bx=ψ f(T r)/(L q(T)-L d(T))计算得到;t eb是转矩的标幺值基值,可以通过t eb=n pψ f(T r)i bx计算得到;在标幺值的形式下,控制算法的转矩和电流的关系表示为: 通过将给定转矩 变为标幺值t en的格式,再通过公式 求解得到电流 的标幺值i dn,最后再通过式 计算得到给定电流 电流 的标幺值i qn通过公式t en=i qn(1-i dn)计算得到,此时t en和i dn是已知量,再通过式 计算得到给定电流
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CN105811836A (zh) * | 2016-05-30 | 2016-07-27 | 中车永济电机有限公司 | 一种大功率表面式永磁同步电机优化控制方法 |
CN106655941A (zh) * | 2017-01-24 | 2017-05-10 | 广州汽车集团股份有限公司 | 一种内嵌式永磁同步电机的参数估计方法及装置 |
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Cited By (2)
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