WO2020248468A1

WO2020248468A1 - Ripple current generation circuit

Info

Publication number: WO2020248468A1
Application number: PCT/CN2019/112994
Authority: WO
Inventors: 贾宇锋; 蔡家齐; 聂燊
Original assignee: 广州金升阳科技有限公司
Priority date: 2019-06-11
Filing date: 2019-10-24
Publication date: 2020-12-17
Also published as: CN110221645B; CN110221645A

Abstract

A ripple current generation circuit, for use in reducing the loss of a charging switch while achieving discharge energy recovery of an electrolytic capacitor and the low frequency pulsating current charging, and making it possible for the average aging voltage of the electrolytic capacitor to follow the input voltage without being affected by the discharge current. The ripple current generation circuit comprises an input voltage, a switch circuit, an inductor, and a capacitor. The switch circuit, the inductor, and the capacitor are connected in series to form a series circuit. Both ends of the series circuit are connected to both ends of an input power supply Vin. A resonant cycle of the inductor and the capacitor is between four-thirds and four times the turn-on time of the switch circuit. When the power supply Vin charges the capacitor, it is unnecessary to provide a voltage higher than the voltage required for the aging of the capacitor, so that it is possible to charge without the switching loss.

Description

A ripple current generating circuit

Technical field

The invention relates to a ripple current generating circuit, in particular to a generating circuit for testing low-frequency charging ripple current of an electrolytic capacitor.

Background technique

At present, there are many applications of switching power supplies in electronic system power supply systems. For occasions where the input power is below 75W and power factor (PF, Power Factor, also known as power factor) is not required, the fly-back switching power supply is reliable due to its simple circuit topology, wide input voltage range, and few components. It has relatively high performance and has a wide range of applications. The flyback switching power supply is referred to as the flyback switching power supply, and the common topology is shown in Figure 1. The prototype of this figure comes from page 60 of "Switching Power Supply Power Converter Topology and Design" written by Dr. Zhang Xingzhu. ISBN978-7-5083-9015-4. It consists of a rectifier bridge 101, a filter circuit 200, and a basic flyback topology unit circuit 300. The practical circuit is also equipped with EMI (Electromagnetic Interference) protection circuits before the rectifier bridge to ensure the electromagnetic compatibility of the flyback switching power supply. Claim.

The filter circuit 200 in FIG. 1 is generally composed of an electrolytic capacitor CL. When designing a switching power supply, the life of the electrolytic capacitor is often faced, and its life is generally composed of withstand voltage, equivalent series resistance (ESR, is the abbreviation of Equivalent Series Resistance), ripple current (Ripplecurrent), loss angle (tgδ) ) And other factors, especially the maximum ripple current, also known as the maximum allowable ripple current, or rated ripple current (IRAC). It is defined as: the maximum effective value of AC ripple current that the capacitor can withstand under the highest operating temperature condition. And the specified ripple current is a sine wave with a standard frequency (usually 100Hz-120Hz).

Electrolytic capacitors have their special ripple current in actual use. When charging, it is the charging current generated when the alternating current reaches close to the peak voltage. This is fully explained in paragraph 0008 of the manual of the authorized announcement number CN102594175B. The frequency of the charging current is twice the frequency of alternating current, which is a low-frequency pulsating current. When discharging, it is the high-frequency ripple current, which is basically the excitation current of the power level of the flyback switching power supply. In the current discontinuous mode, the waveform is a triangle wave. Figure 10-9(b) on page 162 of the above-mentioned "Switching Power Supply Power Converter Topology and Design" is shown. Since it is a well-known technology, it is not shown graphically here.

In summary, when an electrolytic capacitor is used as an input rectifier filter capacitor in a flyback switching power supply, its ripple current is: charging is charging with low-frequency pulsating current, and discharging is discharging with high-frequency ripple current.

In order to design the life of the flyback switching power supply, it is necessary to design the life of the electrolytic capacitor, which requires the life test of the electrolytic capacitor by simulating the actual ripple current in the application of the electrolytic capacitor. For the ripple test of electrolytic capacitors, the patent document of the Patent Authorization Announcement No. CN105242737A gives a ripple current generation method and circuit, as shown in Figure 2, including DC source, inductor, transformer, and capacitor (the capacitor under test) , Diodes and control and drive circuits. The patent document also provides another embodiment, as shown in FIG. 3. Including DC source, inductor, transformer, capacitor (capacitor under test), diode and control and drive circuit, which can realize the charging and discharging function of electrolytic capacitor, and has the characteristics of low cost, low energy consumption, simple wiring and small size. However, the circuit shown in Fig. 2 can only realize the lifetime verification of electrolytic capacitor high-frequency charging and discharging, and the circuit shown in Fig. 3 can only realize the lifetime verification of electrolytic capacitor DC current charging and high-frequency current discharging. In fact, the operating current characteristics of the electrolytic capacitor in the switching converter are: low-frequency pulsating current charging and high-frequency pulsating current discharging. Therefore, the method and circuit proposed by the patent do not meet the requirement of low-frequency pulsating current charging.

In view of the requirement of low-frequency pulsating current charging, the patent scheme of patent application number 201811177216.X is shown in Figure 4. The basic principle of this scheme is described in detail in the first embodiment of the patent application number 201811177216.X, and the excerpt is as follows:

(1) The driving signal Vg2 of the control and driving circuit controls the power tube Q2 at twice the frequency of the power frequency voltage and with an on-time of less than 1ms to realize the charging of the measured capacitor C1. The low-frequency pulsating current charging circuit works in intermittent conduction. In the on mode, the expression of the charging current is ic(t)=(Vin-Vc)/L1*t, where Vin is the DC source voltage and Vc is the voltage of the capacitor under test; when the power tube Q2 is turned off, the inductor L1 The current continues to flow through the diode D2 until the charging current drops to 0. The charging current expression in the freewheeling phase is ic(t)=Ipk-(Vc)/L1*t, where Ipk is the peak current of the inductor L1, Vc is the terminal voltage of the capacitor under test. In this way, the low-frequency charging current ripple is obtained to simulate the charging characteristics of the electrolytic capacitor by the bridge rectifier circuit in practical applications. Among them, the inductor L1 can also avoid the impact of the larger charging current in the circuit. D2 provides a freewheeling loop for inductor L1;

(2) The drive signal Vg1 of the control and drive circuit controls the power tube Q1 with high frequency (such as 65KHz) to realize the discharge of the electrolytic capacitor and obtain the high-frequency discharge current to simulate the energy of the power conversion topology in practical applications. In the transfer process, the energy released by the electrolytic capacitor is fed back to the input power terminal through the flyback transformer to realize the energy feedback function.

The circuit test simulation result is shown in Figure 5 (the black shadow in the figure is caused by the dense waveform). The simulation waveform symbol description in the figure: Vds is the drain-source voltage of the power tube Q1, Vc is the electrolytic capacitor terminal voltage, and Ip is the transformer The primary winding current, Iin is the input current, Ic is the current on the measured capacitor, and Is is the current on the transformer secondary diode D1. By observing the current on the measured capacitor:

[t0, t1] stage: the input power is charged by the main circuit of low-frequency pulsating current to charge the capacitor under test, the current Ic on the capacitor under test rises rapidly, and the terminal voltage Vc of the capacitor under test rises rapidly, and it remains until t1. Ic and voltage Vc reach the maximum; the capacitor under test is also undergoing high-frequency discharge at this stage.

[t1, t2] stage: the electrolytic capacitor continues to maintain high-frequency discharge until t2.

Figure 6 and Figure 7 are respectively expanded diagrams of the simulated waveforms of charging and discharging the capacitor under test. The circuit is in the cyclical work of the above process.

According to the above principle description, the expression of the charging current when Q2 is turned on is ic(t)=((Vin-Vc)/L1)*t. It can be seen that the charging current continues to increase during the Q2 turn-on process, and the increase in speed is caused by the inductor The voltage difference between the two ends Vin-Vc and the inductance L1 are determined. It means that when Q2 is turned off, it is turned off with a large current, and there will be a large switching loss. When the current of the inductor L1 continues to flow through the diode D2 and is cut off, due to the reverse recovery time of D2, a certain switching loss will also occur. Therefore, when the charging current of the electrolytic capacitor increases, the switching loss of the charging circuit of this scheme will increase significantly.

Since the discharge energy of the capacitor comes from the charging energy, if the discharge current is larger, the charging current is required to be larger. As shown in Figure 6, the actual charging current in the freewheeling phase is ic(t)=Ipk-((Vc)/L1)*t. Because Vc is very large, ic(t) is reduced to zero in a very short time. The charging current after cycle averaging is extremely small. Therefore, the average charging current of the electrolytic capacitor is basically equal to the average value of the charging current during the full cycle when Q2 is turned on. Charging current ic(t)=((Vin-Vc)/L1)*t when Q2 is turned on. The charging time t is the turn-on time of Q2 in the excerpt and is generally fixed at 1ms. The aging voltage Vc of the electrolytic capacitor needs to be set according to the aging requirements, and is generally the rated voltage of the electrolytic capacitor. In order to ensure that ic(t) can change with the change of the discharge current, the inductor L1 or the power supply voltage Vin is required to change with the change of ic(t). Because the inductance L1 is difficult to change in real time after the production is completed, when the electrolytic capacitor is aging under different currents, the power supply voltage needs to change with the aging current.

In summary, for the low-frequency pulsating charging current of the electrolytic capacitor, the existing charging scheme has the problem of large charging switch loss and the power supply voltage needs to be changed with the adjustment of the test current.

Summary of the invention

In view of the problems existing in the existing electrolytic capacitor aging scheme, the present invention provides a ripple current generating circuit, which can realize the energy recovery of electrolytic capacitor discharge and low-frequency pulsating current charging while reducing the loss of the charging switch and realizing the average aging voltage of the electrolytic capacitor to follow The power supply voltage is not affected by the test current.

The technical scheme of the present invention is as follows:

A ripple current generating circuit, which is characterized in that it includes an input power supply, a switching circuit, an inductor, and a capacitor; the switching circuit, the inductor and the capacitor are connected in series to form a series circuit, and both ends of the series circuit are connected to the input power supply At both ends of Vin, the resonant period of the inductor and the capacitor is between four-thirds and four times the turn-on time of the switch circuit.

Preferably, the ripple current generating circuit further includes a unidirectional conduction circuit, the unidirectional conduction circuit is connected between any two components of the series circuit, and the connection direction of the unidirectional conduction circuit must be such that Current flows from the positive input power source Vin to the negative input power source Vin, preventing current from flowing from the negative input power source Vin to the positive input power source Vin.

Preferably, the unidirectional conduction circuit includes a first diode.

Preferably, in the ripple current generating circuit described in the above solution, the switch circuit includes a control and drive circuit and a MOS tube, the drain of the MOS tube is used as one end of the switching circuit, and the source of the MOS tube is used as the switch circuit. At the other end, the gate of the MOS tube is connected to the control and drive circuit, which controls the on and off of the MOS tube.

Preferably, the ripple current generating circuit further includes a second diode, and the anode of the second diode is connected to one end of the capacitor and the inductance series circuit. The cathode of the second diode is connected to the other end of the capacitor and inductance series circuit.

The beneficial effect of the present invention is that the power supply Vin does not need to provide a voltage higher than the voltage required for capacitor aging when charging the capacitor. The switching loss of the charging circuit is less than that of the existing solution. If the resonance period of the inductor and the capacitor is twice the on-time of the switching circuit, it can be charged without switching loss.

Description of the drawings

Figure 1 is a flyback switching power supply topology in the prior art;

Figure 2 is one of the ripple current generating circuits in the prior art;

Figure 3 is the second ripple current generating circuit in the prior art;

Figure 4 is a low-frequency pulsating current charging circuit in the prior art;

Figure 5 is a test simulation waveform diagram of a low-frequency pulsating current charging circuit in the prior art;

Figure 6 is one of the expanded simulation waveforms of charging and discharging the measured capacitor;

Figure 7 is the second expanded view of the simulation waveform of the capacitor under test charging and discharging;

Fig. 8 is a circuit schematic diagram of the first embodiment of the present invention;

Figure 9 is a waveform diagram of the first embodiment of the present invention;

Fig. 10 is a schematic circuit diagram of the second embodiment of the present invention;

Figure 11 is a waveform diagram of the second embodiment of the present invention;

Figure 12 is a voltage waveform of Vout and a simulated current waveform of inductor L1 in a state;

Figure 13 shows the voltage waveform of Vout and the simulated current waveform of inductor L1 in another state.

Detailed ways

Example one

The circuit principle diagram of the first embodiment of the present invention is shown in FIG. 8, which includes a switch circuit K2, an inductor L1, a capacitor C1, and an input power source Vin. R1 is the load of the aged capacitor C1. The switch circuit K2, the inductor L1, and the capacitor C1 form a series circuit, and the series position of the switch circuit K2, the inductor L1 and the capacitor C1 can be interchanged without affecting the working principle of the series circuit and the beneficial effects of the present invention. The two ends of the series circuit are connected to both ends of the input power source Vin, and the load is connected to both ends of C1. In this embodiment, one end of K2 is used as one end of the series circuit, the other end of K2 is connected to one end of L1, the other end of L1 is connected to one end of C1, and the other end of C1 is used as the other end of the series circuit.

The basic working process is: when the switch circuit K2 is connected, the voltage of the power supply Vin is higher than the voltage of the capacitor C1, and the power supply Vin charges the capacitor C1 through the inductor L1 and starts resonance. When the inductor and capacitor resonate to a quarter of a cycle, the voltage of C1 rises to the same voltage as Vin, and the current that inductor L1 charges capacitor C1 is the largest. When the inductor and the capacitor resonate to a half cycle, the inductor current is just completely charged to the capacitor C1, and the voltage Vc1 of the capacitor C1 is higher than the power supply voltage Vin. And at this time, Vc1-Vin is equal to the value of Vin-Vc1 when K2 is just turned on. At this time, since the capacitor voltage is higher than the power supply voltage, the capacitor C1 starts to discharge the power supply in reverse. When the inductor and the capacitor resonate to three quarters of the cycle, the voltage of the capacitor C1 drops to equal to the voltage of the power supply Vin, and the charging current of the inductor L1 to the power supply becomes the maximum.

The resonance waveform is shown in Figure 9. The resonant period of the inductor and the capacitor is between four-thirds and four times the turn-on time of the switch circuit, and the turn-on time of the corresponding switch circuit is between one-quarter and three-quarters of the resonant period of the inductor and the capacitor. If the resonance period of the inductor and the capacitor is four-thirds of the turn-on time of the switch circuit, the turn-off time of the switch circuit is t3 in FIG. 9. If the resonant period of the inductor and capacitor is twice the turn-on time of the switch circuit, the turn-off time of the switch circuit is t2 in Fig. 9. If the resonance period of the inductor and the capacitor is 4 times the turn-on time of the switch circuit, the turn-off time of the switch circuit is t1 in FIG. 9. It can be seen from FIG. 9 that when the switch circuit is turned off between t1 and t3, the voltage Vc1 of the capacitor is not less than the voltage of the power supply Vin. Therefore, this solution can charge the capacitor voltage Vc1 to the voltage of the power supply Vin every time. The power supply Vin only needs to be consistent with the capacitor aging voltage Vc1, and does not need to change with the change of the test current.

When the switching circuit is turned off between t1-t2, the current when the switching circuit is turned off is less than the maximum charging current. Since the peak current of the triangular wave in the same area is greater than the peak current of the sine wave, the switching loss of this solution is smaller than that of the existing solution, and the objective of the present invention can be achieved.

When the switching circuit is turned off at t2, there is no current in the switching circuit at the moment of turning off, so there is no switching loss. Therefore, the switching loss when the switching circuit is turned off is smaller than the existing solution, and the objective of the present invention can be achieved.

When the switching circuit is turned off between t2-t3, the current when the switching circuit is turned off is less than the maximum charging current. The absolute value of the peak value of the positive and negative current of the sine wave is the same. Since the peak current of the triangular wave in the same area is greater than the peak current of the sine wave, the switching loss of this solution is smaller than that of the existing solution, and the objective of the present invention can be achieved.

Example two

The second embodiment of the present invention is shown in FIG. 10. This embodiment is an improvement on the basis of the first embodiment. A one-way conduction circuit is added to the series circuit. The one-way conduction circuit can be connected between any two components of the switch circuit K2, the inductor L1 and the capacitor C1. The connection direction of the one-way conduction circuit must enable the current to flow from the input voltage Vin The positive flow to the negative input power source Vin prevents current from flowing from the negative input power source Vin to the positive input power source Vin. In this embodiment, one of the series circuits is selected, and the unidirectional conduction circuit is connected between L1 and C1. The other series connection modes of the series circuit and the connection of the unidirectional conduction circuit between any two other components are the same as the working principle of this embodiment and can achieve the same technical effect.

In this embodiment, the unidirectional conduction circuit is the diode D3, but it is not limited to a diode, and may also be other unidirectional conduction components.

In this embodiment, the switch circuit K2 includes a MOS transistor Q2 and a control and drive circuit 103. One end of the control and drive circuit outputs a control signal Vg2 to connect to the gate of Q2 to control the on and off of Q2; the control and drive circuit The other end is connected to the input power Vin negative; the drain and source of Q2 are respectively used as the two ends of the switch circuit. The MOS tube Q2 can be an NMOS tube or a PMOS tube, which does not affect the working principle of the series circuit.

Preferably, a diode D2 can be added, and the anode of D2 is connected to one end of the series circuit of the capacitor C1 and the inductor L1. The cathode of the second diode is connected to the other end of the series circuit of the capacitor C1 and the inductor L1.

The effect of increasing D2 is: when K2 is disconnected before the inductor L1 and the capacitor C1 resonate to half of the resonant period, the inductor L1 still has a resonant current. At this time, the diodes D2, L1 and C1 form a current loop, so that the inductor L1 can continue to charge the capacitor C1, thereby improving the charging efficiency. The effect of adding D3 is: after the inductor L1 and the capacitor C1 resonate to one-half of the resonant period, that is, after resonating to t2 in Figure 9, the resonant current starts to be negative, that is, the current flows from the input power source Vin to the input power source. Vin is right. D3 prevents the negative resonant current from flowing, that is, L1 and C1 stop resonating at t2.

The switching period of Q2 is Tq2=10mS, and the turn-on time of the MOS tube is Ton=1mS, that is, the turn-on duty cycle is 10%. The specification of the aging capacitor is 100uF/450V electrolytic capacitor. According to the resonance period of L1 and C1, which is between four-thirds to four times of the on-time of the switch circuit, the range of the inductance L1 can be calculated. The formula for calculating the resonant period of L1 and C1 is:

When T=4/3Ton=1.33mS, L=0.448mH.

When T=2Ton=2mS, L=1mH.

When T=4Ton=4mS, L=4mH.

The selected value range for calculating the inductance L1 is 0.448mH-4mH.

When the inductance is 1mH, the input Vin=450V, the voltage waveform of Vout and the current waveform of the inductor L1 are shown in Fig. 11. It can be seen from the simulation waveform that the aging voltage Vout of the electrolytic capacitor fluctuates around 450V, and the average value is equal to Vin. At the same time, when the switch Q2 is turned off, the current flowing through Q2 is zero.

The selected value of the inductance is 4mH, the input Vin=450V, the voltage waveform of Vout and the current waveform of the inductor L1 are output as shown in Figure 12. It can be seen from the simulation waveform that the aging voltage Vout of the electrolytic capacitor can reach 450V at the highest point, and the average value is less than Vin. At the same time, when the switching tube Q2 is turned off, the current flowing through Q2 is at the maximum value, so Q2 has switching loss.

The selected value of the inductance is 0.448mH, the input Vin=450V, the voltage waveform of Vout and the current waveform of the inductor L1 are output as shown in Figure 13. It can be seen from the simulation waveform that the aging voltage Vout of the electrolytic capacitor fluctuates around 450V, and the average value is equal to Vin. At the same time, when the switch Q2 is turned off, the current flowing through Q2 is zero. However, the charging current waveform of the inductor L1 only maintains 0.75mS. From the resonant period calculation formula, it can be known that as the inductance L1 decreases, the current charging time of L1 will deviate more and more obviously from the charging time index of 1mS.

According to the above simulation results, in the technical solution of the present invention, when the inductor-capacitor resonance period T=4/3Ton-2Ton, the average voltage of the electrolytic capacitor is equal to the power supply voltage Vin, but the charging time is between 0.75mS-1mS. It is basically satisfied that the charging time is within the error range to realize that the electrolytic capacitor voltage follows the input voltage, and the switch tube is turned on and off with zero current.

When the inductance capacitor resonance period T=2Ton-4Ton, the charging time is 1mS, but the average voltage of the electrolytic capacitor is lower than the power supply voltage Vin, and the highest voltage of the electrolytic capacitor can be guaranteed not to be lower than the power supply voltage Vin. Basically meet the requirement of electrolytic capacitor voltage to follow the input voltage.

The above are only the preferred embodiments of the present invention. It should be noted that the above preferred embodiments should not be regarded as limiting the present invention, and the protection scope of the present invention should be subject to the scope defined by the claims. For those of ordinary skill in the art, without departing from the spirit and scope of the present invention, several improvements and modifications can be made, and these improvements and modifications should also be regarded as the protection scope of the present invention.

Claims

A ripple current generating circuit, characterized in that it includes an input power supply, a switching circuit, an inductor, and a capacitor; the switching circuit, the inductor and the capacitor are connected in series to form a series circuit, and both ends of the series circuit are connected to both ends of the input power supply, The resonant period of the inductor and the capacitor is between four-thirds and four times the turn-on time of the switch circuit.
The ripple current generating circuit according to claim 1, characterized in that it further comprises a unidirectional conduction circuit, the unidirectional conduction circuit is connected between any two components of the series circuit, the unidirectional conduction circuit The connection direction should enable the current to flow from the positive input power supply to the negative input power supply, and prevent the current from flowing from the negative input power supply to the positive input power supply.
3. The ripple current generating circuit according to claim 2, wherein the unidirectional conduction circuit comprises a first diode.
The ripple current generating circuit according to any one of claims 1 to 3, wherein the switch circuit includes a control and drive circuit, a MOS tube, the drain of the MOS tube is used as one end of the switching circuit, and the source of the MOS tube The pole is the other end of the switch circuit, the gate of the MOS tube is connected to the control and drive circuit, and the control and drive circuit controls the on and off of the MOS tube.
The ripple current generating circuit according to claim 4, further comprising a second diode, the anode of the second diode is connected to one end of the capacitor and the inductance series circuit, and the cathode of the second diode is connected to At the other end of the capacitor and inductor series circuit.