WO2020222889A1 - Calage passif à grande efficacité - Google Patents

Calage passif à grande efficacité Download PDF

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Publication number
WO2020222889A1
WO2020222889A1 PCT/US2020/017672 US2020017672W WO2020222889A1 WO 2020222889 A1 WO2020222889 A1 WO 2020222889A1 US 2020017672 W US2020017672 W US 2020017672W WO 2020222889 A1 WO2020222889 A1 WO 2020222889A1
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WO
WIPO (PCT)
Prior art keywords
circuit
rectifier
clamp
capacitor
series
Prior art date
Application number
PCT/US2020/017672
Other languages
English (en)
Inventor
Ionel Jitaru
Original Assignee
Rompower Technology Holdings, Llc
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Priority claimed from US16/503,432 external-priority patent/US10574148B1/en
Priority claimed from US16/775,967 external-priority patent/US10972014B2/en
Application filed by Rompower Technology Holdings, Llc filed Critical Rompower Technology Holdings, Llc
Priority to EP20798538.3A priority Critical patent/EP3963707A4/fr
Priority to CN202080047002.XA priority patent/CN114245961A/zh
Publication of WO2020222889A1 publication Critical patent/WO2020222889A1/fr

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Classifications

    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/33569—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements
    • H02M3/33576—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements having at least one active switching element at the secondary side of an isolation transformer
    • H02M3/33592—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements having at least one active switching element at the secondary side of an isolation transformer having a synchronous rectifier circuit or a synchronous freewheeling circuit at the secondary side of an isolation transformer
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0048—Circuits or arrangements for reducing losses
    • H02M1/0054—Transistor switching losses
    • H02M1/0058—Transistor switching losses by employing soft switching techniques, i.e. commutation of transistors when applied voltage is zero or when current flow is zero
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/32—Means for protecting converters other than automatic disconnection
    • H02M1/34—Snubber circuits
    • H02M1/346—Passive non-dissipative snubbers
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03K—PULSE TECHNIQUE
    • H03K17/00—Electronic switching or gating, i.e. not by contact-making and –breaking
    • H03K17/08—Modifications for protecting switching circuit against overcurrent or overvoltage
    • H03K17/081—Modifications for protecting switching circuit against overcurrent or overvoltage without feedback from the output circuit to the control circuit
    • H03K17/0814—Modifications for protecting switching circuit against overcurrent or overvoltage without feedback from the output circuit to the control circuit by measures taken in the output circuit
    • H03K17/08142—Modifications for protecting switching circuit against overcurrent or overvoltage without feedback from the output circuit to the control circuit by measures taken in the output circuit in field-effect transistor switches
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0048—Circuits or arrangements for reducing losses
    • H02M1/0051—Diode reverse recovery losses
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/32—Means for protecting converters other than automatic disconnection
    • H02M1/34—Snubber circuits
    • H02M1/342—Active non-dissipative snubbers
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/32—Means for protecting converters other than automatic disconnection
    • H02M1/34—Snubber circuits
    • H02M1/344—Active dissipative snubbers
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

Definitions

  • the present invention relates generally to electronic devices, and more particularly to power converters.
  • Flyback topology is, ideally, one of the most used circuit topologies in the field of power conversion, especially in lower-to-medium power applications (such as AC -DC adapters, for example).
  • the reason for such a high level of adoption of flyback topology is its simplicity and low cost of implementation, as well as the fact that the so-configured electrical circuitry can operate efficiently over a very large range of input voltages.
  • the circuits formatted according to flyback topology are used to operate after an output from a simple bridge rectifier, while the alternating-current input voltage ranges from 90V ac to 264Vac (conventionally, a rectifier is an electrical device that converts alternating current to direct current, which flows in only one direction).
  • the flyback converter has to be able to operate efficiently with a DC input voltage ranging from 127Vdc to 375Vdc, which is a range in which the ratio of the upper input voltage limit to the lower voltage limit is almost three to one.
  • the new standards for power delivery require that the adapters provide a voltage output ranging from 5 V to 20V (with the ratio of the upper voltage limit to the lower voltage limit of four to one, as far as the output voltage is concerned).
  • forward-derived topologies such as, for example, half-bridge topology, two-transistor forward topology, full bridge topology, and others
  • the efficiency of the AC-DC adapters has been increased from about 70% to about 89-90% (in recent products such as the Apple 30W adapter, for example). This has been mostly due to the significant progress in semiconductor industry and a better understanding of magnetic technology.
  • the flyback topology possesses several drawbacks that limit its efficiency of operation.
  • the flyback-topology circuitry operates in a discontinuous mode. In a discontinuous mode of operation, the magnetizing current is first built up from zero to a peak conducting, and after the main switch turns off, the magnetizing current flows into the secondary side winding and transfers the energy to the output capacitor until the value of the magnetizing current decreases to zero.
  • This portion of the operation cycle is followed by a second period of time, referred as“dead time”, when no energy is stored in the transformer or transferred to the secondary.
  • “dead time” is reduced to the transition time (which is the time interval during which the voltage across the main switch decays from the level it had when the magnetizing current flowed into the secondary winding to its lowest level, which occurs in the beginning of the“dead time”)
  • this mode of operation is referred as a critical conduction mode of operation.
  • a circuit has primary and secondary sides and includes a flyback power converter including an input voltage source and a transformer having primary and secondary windings on the primary and secondary sides, respectively.
  • a main switch is in series with the primary winding on the primary side.
  • a passive clamp circuit is across the main switch, the passive clamp circuit including a clamp diode, a clamp capacitor, and an auxiliary circuit having first and second rectifiers in series with each other and in series with an electronic component configured to store electromagnetic energy.
  • the electronic component has first and second terminals.
  • a cathode of the first rectifier is connected with the passive clamp circuit, and an anode of the first rectifier is connected to the second terminal of electronic component.
  • an anode of the second rectifier is connected with the cathode of the first rectifier, and a cathode of the second rectifier is connected with the first terminal of the electronic component.
  • the auxiliary circuit is configured to reduce an RMS current through the clamp capacitor from a first current value to a second current value which is at least thirty percent lower than the first current value, and to reduce a charge through the clamp capacitor from a first charge value to a second charge value which is at least thirty percent lower than the first charge value.
  • the clamp diode is formed by several diodes in parallel.
  • the first rectifier is in series with a resistor.
  • the first rectifier is in series with a resistor.
  • the electronic component includes a voltage source, the first terminal is a positive terminal of the voltage source, and the second terminal is a negative terminal of the voltage source.
  • the electronic component includes a voltage source, and the voltage source provides a bias voltage to the flyback power converter.
  • the electronic component includes a bias voltage source for the flyback power converter having a bias winding in the transformer, the bias winding coupled with the primary winding of the transformer and having first and second terminals, a bias synchronous rectifier having a gate and a drain, wherein the drawn of the bias synchronous rectifier is connected to the first terminal of the bias winding, and a bias capacitor connected to the second terminal of the bias winding and to the source of the bias synchronous rectifier.
  • a resistor is connected between the positive terminal of the input voltage source and a junction between the clamp capacitor and clamp diode.
  • a method of operating a circuit having primary and secondary sides includes providing a flyback power converter having an input voltage source and a transformer having primary and secondary windings on the primary and second sides, respectively. The method includes providing a main switch in series with the primary winding on the primary side, providing a synchronous rectifier in series with the secondary winding on the secondary side, providing a passive clamp circuit across the main switch, the passive clamp circuit including a clamp diode and a clamp capacitor in series with the clamp diode.
  • the circuit is characterized by an electrical charge, injected into the clamp capacitor after the main switch is turned off, the electrical charge having a first charge value.
  • the method further includes electrically connecting the circuit with an auxiliary circuit in series with the clamp capacitor, wherein the auxiliary circuit includes an auxiliary energy storage, first and second rectifiers in series with each other, and an electronic component configured to store electromagnetic energy in series with the first and second rectifiers.
  • a cathode of the first rectifier is connected with the passive clamp circuit, and an anode of the first rectifier is connected to the second terminal of electronic component.
  • An anode of the second rectifier is connected with the cathode of the first rectifier, and a cathode of the second rectifier is connected with the first terminal of the electronic component.
  • the method further includes directing a current, flowing through a leakage inductance reflected in the primary side of the transformer, to flow through the clamp capacitor and through the second rectifier toward the auxiliary energy storage, so as to change the first charge value, thereby imparting a second charge value to flow through the first rectifier during a reverse recovery time of the clamp diode, so as to balance the electrical charge in the clamp capacitor.
  • the step of electrically connecting further includes connecting the auxiliary energy storage between a ground and the cathode of the second rectifier.
  • the step of electrically connecting further includes connecting the anode of the first rectifier to the ground via a resistor.
  • a resistor is connected between a positive terminal of the input voltage source and a junction between the clamp capacitor and clamp diode.
  • a circuit having primary and secondary sides includes a flyback power converter including an input voltage source having positive and negative terminals, a transformer having primary and secondary windings on the primary and secondary sides, respectively.
  • a main is in series with the primary winding on the primary side.
  • a parasitic capacitor is across the main switch, and a synchronous rectifier is in series with the secondary winding on the secondary side.
  • a passive clamp circuit is across the main switch, the passive clamp circuit includes a clamp diode, a clamp capacitor in series with the clamp diode, a resistor connected between the positive terminal of the input voltage source and a junction between the clamp capacitor and clamp diode.
  • First and second rectifiers are in series with each other and in series with an electronic component configured to store electromagnetic energy.
  • a cathode of the first rectifier is connected with the passive clamp circuit, and an anode of the first rectifier is connected to a second terminal of electronic component.
  • An anode of the second rectifier is connected with the cathode of the first rectifier, and a cathode of the second rectifier is connected with a first terminal of the electronic component.
  • a resistor is in series with the first rectifier.
  • a current injection circuit includes a current injection winding in the transformer having first and second terminals, a current injection switch is connected to the first terminal of the current injection winding.
  • a current injection diode has a cathode connected to the second terminal of the current injection winding, and an anode connected to the electronic component configured to store electromagnetic energy.
  • An energy source collects energy of a leakage inductance of the transformer via the passive clamp circuit and second rectifier.
  • the current injection winding is configured to inject the energy as a pulse of current into the transformer via the current injection winding, so as to discharge the parasitic capacitor to create a zero voltage switching condition for the main switch.
  • a resonant capacitor is between the second terminal of the current injection winding and the current injection switch at a terminal of current injection switch which is not connected to the current injection winding.
  • the circuit includes a bias diode having an anode, connected to the second terminal of the electronic component, and a cathode, connected to a bias voltage.
  • the clamp diode includes at least two diodes in parallel.
  • the current injection switch turns on when a voltage across the main switch reaches its lowest amplitude, and the main switch turns on when the voltage across the main switch reaches a predetermined value. In an embodiment, the predetermined value is zero.
  • a circuit has primary and secondary sides and includes an input voltage source connected to a primary winding of a transformer, the transformer having additional windings, and a leakage inductance between the primary winding and the additional windings.
  • the circuit further includes a main switch in series with the primary winding, wherein a magnetizing current of the transformer has a low impedance path to further circulate after the main switch turns off.
  • the circuit includes a passive clamp circuit having of a diode and a capacitor in series, wherein the passive clamp circuit is connected to an energy extraction circuit having first and second rectifiers connected in series with each other and in series with an electronic component configured to store electromagnetic energy.
  • the cathode of the first rectifier is connected with the passive clamp circuit, and an anode of the first rectifier is connected to a second terminal of electronic component.
  • An anode of the second rectifier is connected with the cathode of the first rectifier, and a cathode of the second rectifier is connected with a first terminal of the electronic component.
  • the circuit includes a current injection winding in the transformer having first and second terminals, a current injection switch connected to the first terminal of the current injection winding, and a current injection diode, with a cathode connected to the second terminal of the current injection winding, and an anode connected to the electronic component configured to store electromagnetic energy.
  • the circuit further includes an energy source to collect energy of a leakage inductance of the transformer via the passive clamp circuit and second rectifier.
  • the current injection winding is configured to inject the energy as a pulse of current into the transformer via the current injection winding, so as to discharge the parasitic capacitor to create a zero voltage switching condition for the main switch.
  • the circuit further includes a resistor in series with the first rectifier.
  • the circuit further includes a resistor connected between a positive terminal of the input voltage source and a junction between the capacitor and diode.
  • a DC-DC converter in another embodiment, includes an input voltage source in series with a parasitic capacitance, wherein a voltage of the input voltage source changes sufficiently abruptly to cause resonant ringing, wherein the input voltage source is connected to a rectifier means which is connected to an output circuit.
  • the converter includes a passive clamp circuit across the rectifier means, wherein the passive clamp circuit includes a clamp diode, a clamp capacitor, and an auxiliary circuit, said auxiliary circuit including first and second rectifiers in series with each other and in series with an electronic component configured to store electromagnetic energy, the electronic component having first and second terminals.
  • a cathode of the first rectifier is connected with the passive clamp circuit, and an anode of the first rectifier is connected to the second terminal of the electronic component.
  • An anode of the second rectifier is connected with the cathode of the first rectifier, and a cathode of the second rectifier is connected with the first terminal of the electronic component. Directing a current, flowing through a leakage inductance, to flow through the clamp capacitor and through the second rectifier toward the auxiliary energy storage, causes some of the leakage inductance to transfer to an auxiliary energy storage and damps the resonant ringing.
  • the anode of the first rectifier is connected to a ground via a resistor.
  • the electronic component includes a voltage source which provides a bias voltage to the DC-DC converter.
  • this disclosure presents several electronic-circuitry configurations that address the limitations conventionally associated with flyback topology.
  • the proposed solutions increase the efficiency of flyback-topology-utilizing power converters above about 94 %, decrease the level of dissipated heat and, as a result, produce a much higher power density (for example, above 27W/in ⁇ '3).
  • FIG. 1 is a schematic of a prior art power converter
  • FIG. 2 illustrates waveforms of the converter of FIG. 1 ;
  • FIG. 3A is a schematic of a flyback converter
  • FIG. 3B illustrates waveforms of the converter of FIG. 3 A
  • FIG. 4A is a schematic of a resonant circuit
  • FIG. 4B illustrates equations characterizing behavior of the resonant circuit of FIG. 4A
  • FIG. 5A illustrates reverse recovery characteristics of an exemplary diode
  • FIG. 5B illustrates an exemplary diode an equation for negative voltage
  • FIG. 6 is a schematic of an RDC snubber
  • FIG. 7 illustrates waveforms for the RDC snubber of FIG. 6
  • FIG. 8 illustrates voltage across a main switch of a flyback converter, showing ringing
  • FIG. 9 illustrates voltage across a main switch of a flyback converter using a snubber, showing damping of ringing
  • FIG. 10 is a schematic of a flyback converter with an active clamp
  • FIG. 11 illustrates waveforms of the flyback converter of FIG. 10
  • FIG. 12 is a schematic of a flyback converter with an active clamp
  • FIG. 13 illustrates waveforms of the flyback converter of FIG. 13
  • FIG. 14 is a schematic of a flyback converter with a passive clamp
  • FIG. 15 illustrates waveforms of the flyback converter of FIG. 14
  • FIGS. 16-18 are schematic of a different flyback converters, each with a passive clamp;
  • FIG. 19 illustrates waveforms of the flyback converter of FIG. 17;
  • FIG. 20 illustrates waveforms across a main switch of the flyback converter of FIG. 17 at different input voltages
  • FIG. 21 illustrates waveforms of voltage across the main switch, and current through the passive clamp, of the flyback converter of FIG. 17;
  • FIG. 22 is a schematic of an exemplary application of a flyback converter
  • FIG. 23 illustrates waveforms of the flyback converter of FIG. 22
  • FIG. 24A is a schematic of a prior art implementation of an RC snubber
  • FIG. 24B illustrates a voltage across a diode in the RC snubber of FIG. 24A
  • FIG. 25A is a schematic of a prior art implementation of an RC snubber
  • FIG. 25B illustrates voltages in elements of the RC snubber of FIG. 25 A
  • FIG. 26A is a schematic of an implementation of an RC snubber.
  • FIG. 26B illustrates voltages in elements of the RC snubber of FIG. 26A.
  • FIG. 1 illustrates a simplified schematic of electronic circuitry of a prior art power converter 100 configured according to a flyback topology using a passive clamp.
  • the flyback converter 100 is formed by a transformer (Tr, 16) that has a primary winding 12 (with corresponding inductance LI and N1 turns in the corresponding coil) on the primary side, and a secondary winding 14 (with corresponding inductance L2 and N2 turns in the corresponding coil).
  • the flyback converter 100 has a primary or main switch (Ml, 18) controlled by a control voltage signal (VcMl, 22) on the primary side.
  • the flyback converter also includes a parasitic capacitance (Ceq, 20) that represents the total parasitic capacitance reflected across the primary switch and is disposed, on the primary side, between a terminal of the primary winding (LI, 12) and the ground 34.
  • the source of the input voltage labelled as Vin or reference character 10 is connected to another terminal of the primary winding 12.
  • the converter also includes a synchronous rectifier (SR, 28) on the secondary side that is controlled by a control voltage signal (VcSR, 30), and an output capacitor (Co, 26) disposed between the ground 34 and the terminal of the secondary winding (L2, 14).
  • the output voltage signal (Vo, 24) can be read across the capacitor (Co, 26).
  • any of the primary winding(s) and secondary winding(s) are discussed as possessing a corresponding inductance.
  • the terms“main switch” and“primary switch” may be used interchangeably.
  • FIG. 2 illustrates the plots representing the key waveforms of a flyback converter 100 operating in discontinuous mode. These key waveforms are presented in order from top to bottom and include: 1) the control signal VcMl for the main switch Ml; 2) the current IM1 through the main switch; 3) the voltage across the main switch VM1; and 4) the voltage VSR across the synchronous rectifier 28.
  • VcMl for the main switch Ml
  • IM1 current IM1 through the main switch
  • VM1 the voltage across the main switch VM1
  • VSR voltage across the synchronous rectifier 28.
  • such first energy is dissipated as is the second energy contained in the ringing portion 44 of the VM1 signal across the main switch during the dead time period.
  • the ringing portion 44 contains the second energy at frequencies that are lower than frequencies corresponding to the ringing 52 and the first energy.
  • the energy contained in the leakage inductance at full load is about 6.8uJ.
  • the second energy corresponding to the lower frequency ringing 44 across the main switch after the energy is fully delivered to the secondary side, has a lower value.
  • Ceq parasitic capacitance
  • 20V output voltage such second energy is about 1.8uJ.
  • the energy contained in the parasitic capacitance (Ceq, 20) across the primary switch is also function of the input voltage (Vin, 10).
  • the energy in Ceq is 2.58uJ
  • the energy contained in Ceq is about 18uJ.
  • Example 1 Prior Art Flyback Converter With Passive Clamp
  • FIG. 3A presents a simplified schematic of a flyback topology 102
  • FIG. 3B shows waveforms for that topology 102. It is noted here that some of the reference characters of the converter 100 are adopted to indicate similar elements. For example, both 100 and 102 have a main switch (Ml, 18).
  • the leakage inductance (Llk, 37) is shown in FIG. 3A as a discrete inductive element to help in understanding the operation of the flyback at the time when (Ml, 18) turns off.
  • the leakage inductance (Llk, 37) starts resonating with the parasitic capacitance of the main switch (Ceq, 20), as depicted in FIG. 3B. This leads to very large voltage spikes which may exceed the voltage rating of the main switch (Ml, 18), as depicted in FIG. 3B.
  • the leakage inductance (Llk, 37) the parasitic capacitor reflected across Ml (Ceq, 20), and the input voltage source (Vin, 10) form a resonant circuit with initial conditions, which initial conditions include: the current flowing through Llk at the time when Ml turns off; the voltage across the Ml; and the parasitic capacitance reflected across Ml, which is the summation of the Coss of Ml (which is the parasitic capacitance of the switch Ml itself) and the parasitic capacitance reflected from the secondary winding through the transformer and the parasitic capacitance across the primary winding.
  • FIG. 4A shows a typical resonant circuit with initial conditions.
  • the initial conditions include: Im, the current through the inductive element; Lm, the inductance of the inductive element; and Vr, the voltage across the ideal switch SW.
  • FIG. 4B shows six equations (A)-(F); these characterize the behavior of the circuit of FIG. 4A.
  • the equation (D) calculates the voltage across the switch SW and we can calculate the amplitude of the spikes across the switch SW at turn off and the frequency of ringing across the switch.
  • a solution presented here is the RDC snubber. This solution it limits the voltage spikes across the main switch and is also relatively simple and low cost.
  • FIG. 6 illustrates an RDC snubber 103, including a diode (Dl,38), a capacitor (Cl, 40), a discharge resistor (R2,42) and a snubber resistor (Rl,36).
  • the characteristics of the diode D1 play affect the functionality of this type of snubber and, more specifically, the reverse recovery characteristic of the diode Dl.
  • FIG. 5A the reverse recovery characteristic of any exemplary diode, and key parameters which describe the reverse recovery, are shown.
  • the slope of the current during the Tb portion of the reverse recovery time differentiates the diode as a soft reverse recovery wherein the dl(rec)/dt is smaller, rather than a snappy reverse recovery diode wherein dl(rec)/dt is higher.
  • a key characteristic of the diode for application as a passive snubber is the time period ta, during which there is still charge in the junction.
  • FIG. 7 presents key waveforms for the RDC snubber 103, which include: IdMl, the current through the main switch (Ml, 18); ID1, the current through the clamp diode; and Vds(Ml), the voltage across Ml.
  • the main switch Ml turns on and the current through the transformer primary winding, (LI, 12), builds up to a peak level Ipk, which is reached at time tl.
  • the main switch Ml turns off and the magnetizing current in the transformer Tr starts flowing into the secondary winding (L2,14) via the synchronized rectifier (SR, 28).
  • the leakage inductance current flows through the passive clamp via the snubber resistor (Rl, 36) and the diode (Dl, 38) into the clamp capacitor (Cl, 40), which was slightly discharged by (R2, 42) prior to time tl.
  • a resonant circuit is formed by the leakage inductance in the transformer Tr reflected in the primary and the capacitor (Cl, 40) wherein the snubber resistor (Rl, 36) acts as a dumping resistor.
  • the resonant circuit formed by the leakage inductance of the transformer and the capacitor (Cl, 40) changes. This is due to the diode Dl which becomes a high impedance device when the charge in the junction of Dl is depleted.
  • the new resonant circuit is formed now by the leakage inductance reflected in the primary of transformer Tr, and the parasitic capacitance of diode Dl.
  • the snubber resistor (Rl, 36) becomes the dumping resistor of this resonant circuit.
  • the capacitor Cl is still in the circuit, but its value is much larger than the parasitic capacitance of Dl, and as a result, its impact in the resonant circuit is negligible.
  • HF ringing The high frequency ringing (“HF ringing”) caused by the resonance between the leakage inductance reflected in the primary of the transformer (Tr, 16) and the parasitic capacitance of the diode D1 in parallel with the parasitic capacitance Ceq across the primary switch Ml creates an oscillation across the primary switch Ml, as depicted in FIG. 8, which shows the voltage across the main switch in a flyback topology with a DC input voltage of 327Vdc, a turn ratio in the transformer of 5.0 and an output voltage of 20V, and a leakage inductance in the transformer of 2. luH, and the part number used for D1 is S3N.
  • FIG. 9 illustrates the voltage across the main switch using the same transformer Tr and the same parts at the same input and output conditions with the snubber resistor of 51 Ohm.
  • the peak voltage across the main switch in the same conditions increases from 500V in FIG. 8 to 560V in FIG. 9.
  • there is a decay in efficiency because a snubber using a resistor in series with the diode is dissipative.
  • the additional voltage across the main switch is due to voltage across the snubber resistor Rl, caused by the peak current through Dl, creating the change in voltage, or AV, as depicted in FIG. 7.
  • the passive snubber solution does limit the voltage stress on the main switch and function of the reverse recovery characteristics of Dl, and some of the leakage inductance energy is recycled to the secondary by the reverse conduction through the diode due to the reverse recovery characteristic of the diode Dl.
  • the high frequency ringing across the main switch which appears after the charge in the junction of Dl is depleted, due to the oscillation between the leakage inductance of the transformer (Tr, 16) reflected in the primary, and the parasitic capacitance of the diode Dl after the charge in the diode Dl junction is depleted in parallel with the parasitic capacitance Ceq.
  • FIG. 10 illustrates the attempt of U.S. Pat. No. 5,434,768 as applied to circuitry 300 of a flyback converter.
  • the circuitry 300 adds a clamp or complementary switch M2, 50, controlled by a control voltage signal VcM2, 54.
  • the circuitry also includes a clamp capacitor Cr, 52.
  • the switch 50, control voltage signal 54, and clamp capacitor 52 together cooperate to define an active clamp connected between one terminal of the primary winding LI and the ground.
  • FIG. 11 illustrates the key waveforms of the flyback circuit 300 with the active clamp of FIG. 10. These waveforms are presented in order from top to bottom and include: 1) the control signal VcMl (for the main switch Ml); 2) the voltage across the main switch, VdsMl; the current IL1 through the primary winding (LI, 12) of the transformer Tr, 16; 3) the current ICr through the clamp circuit formed by Mosfet M2 and clamp capacitor Cr; 4) the current ISR through the synchronized rectifier (SR, 28); and 5) the control voltage signal for the clamp Mosfet (M2, 50), VcM2.
  • the main switch Ml is switched on, and the current IL1 starts to build up through the magnetizing inductance, thereby storing energy in the transformer Tr.
  • the primary switch turns off, and as a result the magnetizing current starts flowing into the secondary winding.
  • the leakage inductance reported to the primary side and the clamp capacitor 52 form a resonant circuit.
  • the current through the leakage inductance starts flowing through the clamp capacitor 52 and the resonant circuit, formed by the leakage inductance, and the clamp capacitor 52 shapes the current through the clamp circuit formed by M2 and Cr, accordingly.
  • the current ICr through the clamp capacitor Cr is characterized by ringing at the frequency(ies) determined by the resonant frequency between the leakage inductance and the clamp capacitor 52.
  • the current ICr flows through the clamp capacitor Cr, 52 decaying and substantially reaching a zero level at time t2.
  • the current through the clamp capacitor Cr becomes negative (which means that the current will be transferred to the secondary side, as depicted by the shape of the curve labeled ISR, which represents the current through the synchronous rectifier SR).
  • ISR the curve labeled current through the synchronous rectifier SR.
  • the current ICr through the clamp circuit turns positive again, and then reaches the zero level again at about time t4.
  • the current through the clamp circuit turns negative until the clamp switch M2 turns off at time t5; the energy contained in the magnetizing current adds to the existing energy contained in the resonant circuit that is formed by the primary winding (LI, 12) and the parasitic capacitance (Ceq, 20) reflected across the main switch.
  • the current through the clamp circuit is shown to have negative amplitude (If, 146), and this current increases the amplitude of ringing during the dead time of the flyback converter from 148 to 150 (as shown in the broken-line and solid-line curves of VdsMl).
  • the number of ringing cycles or undulations, as well as the polarity of the current passing through the clamp capacitor Cr, is a function of the time at which the clamp switch M2 is switched“off’ and the resonant frequency formed by leakage inductance and the clamp capacitor Cr. Accordingly, the number of such undulations and/or polarity of the current through Cr may vary.
  • the clamp circuit (formed by M2 and Cr) takes the leakage inductance energy initially by charging the clamp capacitor (Cr, 52), and some of the energy is transferred to the secondary side while some of the energy is bounced back (to the primary side) and forth before the active clamp switch turns off.
  • the negative current (If, 146) passing through the clamp capacitor 52 at time t5, when the clamp switch turns off is tailored to add to the energy in the resonant circuit formed by the inductance of the primary winding of the transformer and the parasitic capacitance (Ceq, 20) in order to increase the ringing amplitude and - by using the valley detection approach to turn on the main switch at a lower voltage level (or even at a zero voltage level) if the flyback operates in critical conduction.
  • valley detection includes identification of the valley(s) or the portion(s) of the curve VdcMl around the local minimum(s) of the ringing after time t5 and tuning the main switch“on”, at the moments corresponding to these valley(s) to reduce the main switch losses.
  • the primary side of the overall circuitry 1000 includes the primary winding LI of the transformer (Tr, 16) having N1 turns in its primary winding.
  • One terminal of the primary winding LI is directly connected to the source of the DC input voltage (Vin, 10), and another terminal of the primary winding (LI, 12) - shown as the node 1010 - is connected with the active clamp formed by the clamp or complementary switch (M2, 50) and the clamp capacitor Cr, 52.
  • the clamp switch is controlled by the source of control voltage signal VcM2, 54.
  • the secondary side of the overall circuitry 1000 includes the secondary winding L2 of the transformer (Tr, 16) that has N2 turns in its secondary winding L2 and the synchronous rectifier (SR, 28).
  • rectifier means are connected to the clamp capacitor (Cr, 52).
  • rectifier means are depicted as diodes.
  • the first rectifier (Dl, 58) has its anode connected to the clamp capacitor (Cr, 52), at the node 1020
  • the second rectifier (D2, 56) has its anode electrically connected to the input ground and its cathode to the clamp capacitor (Cr, 52) at the node 1020.
  • An energy storage component configured to store electromagnetic energy (shown here as the voltage source (VB, 60)), is further added between the ground and the cathode of the first rectifier.
  • This electronic circuitry addition to the active clamp portion of the circuit 1000 is designated with the reference character 1030.
  • the sub circuit designated as 1030 is also identified herein as an“energy extraction circuit”.
  • the key waveforms representing the operation of the circuit 1000 are schematically depicted in FIG. 13. These include, from top to bottom: 1) the control signal (VcMl, 22) of the main switch Ml; 2) the voltage VdsMl across the main switch Ml; 3) the current IL1 through the primary winding LI of the transformer (Tr, 16); 4) the current ICr through the clamp capacitor Cr; 5) the current ISR through the synchronous rectifier (SR, 28); and 6) the control signal (VcM2,54) for the clamp switch M2.
  • the main switch Ml is configured to conduct, and the magnetizing current builds up the transformer Tr.
  • the main switch Ml turns off (shown as VcMl reaching zero) and the magnetizing current starts flowing towards the secondary winding (L2, 14) and through the synchronous rectifier (SR, 28).
  • the current flowing through the leakage inductance of the primary side starts flowing though the clamp circuit formed by (M2, 50) and the clamp capacitor (Cr, 52), and then through the first rectifier (Dl, 58) towards the auxiliary energy storage, depicted as voltage source (VB, 60).
  • the current from the leakage inductance here is directed towards the voltage source VB, 60.
  • This directionality completely changes the mode(s) of the operation of the clamp circuit: for example, a comparison of the shape of ICr representing the current passing through the clamp capacitor (Cr, 52) of FIG. 11 with that of FIG. 13 immediately illustrates the advantageous nature of operation of the embodiment 1000.
  • the ringing portion associated with the resonance between the leakage inductance and the clamp capacitor as depicted in FIG. 11 (see the period from times t2 to t4) substantially disappears, and the time interval between times tl and t2 (during which the current though the clamp capacitor reaches the zero level) substantially shortens.
  • the charge injected into the clamp capacitor Cr (and substantially corresponding to the area under the ICR curve between the times tl and t2) is significantly decreased in the case of operation of the circuitry 1000 as compared to that of the circuitry whose operation is described in FIG. 11.
  • the electrical charge extracted from the clamp capacitor between times t2 and t3 is decreased as well.
  • Injecting (delivering) the current through the diode D1 into a voltage source (auxiliary storage of EM energy) VB causes the transfer of energy to the voltage source VB and produces a dumping effect, substantially reducing the ringing in the active clamp portion of the circuitry 1000.
  • a significant portion of the leakage inductance energy is transferred to the voltage source (VB, 64).
  • the time interval between times tl and t2 is a function of the voltage across VB.
  • the time interval between times tl to t2 decreases, comparatively.
  • the voltage across VB is reflected across the voltage across the main switch, as depicted in the Vds Ml of FIG. 12 (VB is the “overshoot” portion of the VdsMl curve, which is characteristically not present in corresponding VdsMl curve of FIG. 11).
  • a reduction of the duration of the time interval from tl to t2 leads to a considerable decrease of the RMS current through clamp circuit formed by M2, 50 and Cr, 52.
  • the RMS current ICr through the clamp circuit is about 0.313A when the structure and operation shown in FIGS. 10 and 11 is used.
  • the advantageous impact of the circuitry portion 1030 is even stronger.
  • the RMS current through the clamp circuit would be about 0.527 A.
  • the energy extraction circuit 1030 as shown in FIG. 12 is used according to the principle of FIG. 13, then for a VB of about 10V, the RMS current through the clamp circuit is reduced to about 0.249A (that is, by about 52%).
  • the time interval measured between the times tl and t2 in FIG. 11 is about 1.12us.
  • the time interval tl to t2 is reduced to about 0.644us for Vb of 10V.
  • circuit 1030 improves the operation of the flyback converter with an active clamp by substantially reducing the RMS current passing through the clamp capacitor.
  • Example 3 has many advantages in comparison with the solution presented in Example 2 and Example 1. However, it requires an active clamp, which uses another switching element and a floating drive, which increases the cost of the solution. For applications such as AC-DC adapters, cost and simplicity are paramount.
  • the embodiment presented here processes the leakage inductance energy which is non-dissipative, and the energy from the leakage inductance is recycled back to the secondary and some of it is used for other purposes. This helps eliminate the ringing and spikes depicted in Example 1, and also does so without dissipation, as in Example 3.
  • Example 3 the presence of the energy extraction circuit 1030 reduces the charge injected in the clamp capacitor Cr during the time interval from tl to t2.
  • the charge which is extracted from Cr during the time interval between t2 and t3 will also decrease.
  • the active clamp switch (M2, 50) of FIG. 12 can be replaced by a diode (Del, 86) of FIG. 14, and the basic mode of operation does not change.
  • the energy extraction circuit 2030 of FIG. 14 allows a full active clamp function while using a diode and employing the reverse recovery characteristic of the diode.
  • the interval between times t2 and t3 (in FIG. 13) is then reduced by the use of the energy extraction circuit to become comparable with the reverse recovery time of the diode Del from FIG. 14.
  • power conversion engineers have looked for a solution using a passive clamp which would emulate the function of an active clamp, wherein the leakage inductance energy is fully recycled.
  • the charge injected and further extracted from the clamp capacitor were too much to be able to make this concept work. In other words, attempts at solving this problem had been made without a successful solution in the prior art.
  • FIG. 14 presents a passive clamp circuit 2000.
  • the primary side of the overall circuit 2000 includes the primary winding LI of the transformer (Tr, 16) which has N1 turns in its primary winding.
  • One terminal of the primary winding LI is directly connected to the source of the DC input voltage (Vin, 10) and another terminal of (LI, 12) - shown as the node 2010 - is connected to the diode Del, 86 in series with the clamp capacitor (Cr, 43) and further connected in series with an energy extraction circuit 2030.
  • the energy extraction circuit 2030 is an electronic component formed by several elements: two rectifiers (or rectifier means, which in this example are depicted as diodes), the first rectifier (Dl, 58) has its anode connected to the clamp capacitor (Cr, 43) at the node 2020, the second rectifier (D2, 59) has its anode electrically connected to a resistor (Rs,70) which is further connected to the input ground, and also has its cathode connected to the clamp capacitor (Cr, 52) at the node 2020.
  • the energy extraction circuit further includes an energy storage (shown here as the voltage source (VB, 60)) disposed between the ground and the cathode of the first rectifier (Dl,58).
  • This energy extraction circuit 2030 thus is an electronic component also known as a sub-circuit; it contains a group of electronic devices which perform a certain, specific function.
  • the secondary side of the overall circuitry 2000 includes the secondary winding L2 of the transformer (Tr, 16) that has N2 turns in its secondary winding and the synchronous rectifier (SR, 28).
  • Key waveforms representing the operation of the circuit 2000 are schematically depicted in FIG. 15. These key waveforms are presented in order from top to bottom and include: l) the control signal (VcMl, 22) of the main switch Ml; 2) the voltage (VdsMl) across the main switch Ml ; 3) the current IL1 through the primary winding LI of the transformer (Tr, 16); 4) the current ICr through the clamp capacitor (Cr, 43); and 5) the current ISR through the synchronous rectifier (SR, 28).
  • the main switch Ml in operation of the circuitry 2000, in the period between times tO and tl, the main switch Ml is configured to conduct and the magnetizing current builds up the transformer Tr. At time tl, the main switch Ml turns off (shown as VcMl reaching zero) and the magnetizing current starts flowing towards the secondary winding (L2, 14) and through the synchronous rectifier (SR, 28). The current flowing through the leakage inductance of the primary side starts flowing though the passive clamp circuit formed by the diode (Del, 86) and the clamp capacitor (Cr, 43), and then through the first rectifier (Dl, 58) toward the auxiliary energy storage, depicted as voltage source (VB, 60). Unlike in Example 1 showing the traditional or prior art passive clamp, the current from the leakage inductance here is directed towards the voltage source VB, 60.
  • FIG. 15 The key waveforms of FIG. 15 are similar to those presented in FIG. 13. However, in FIG. 15 there is not a control signal for the active clamp switch VcM2.
  • the interval between times t2 to t3 should be smaller than the reverse recovery time trr of the diode (Del, 86).
  • the reverse recovery characteristics of the diode (Del, 86) have to be larger than the time interval between t2 to t3, in all the operating conditions of the flyback converter.
  • the leakage inductance of the transformer (Tr, 16) be under a certain value for a given power level, and a given turns ratio of the tranformer ( Tr, 16) and a given value for (VB,60).
  • Injecting, or delivering, the current through Dl,58 into voltage source VB, 60 causes a transfer of energy to the voltage source VB, 60 and produces a dumping effect, substantially reducing the ringing in the active clamp portion of the circuitry.
  • a significant portion of the leakage inductance energy is transferred to the voltage source VB, 60.
  • the interval between times tl and t2 is a function of the voltage across VB.
  • the energy from the leakage inductance (which is harvested/injected in the voltage source VB) is used to provide some or all of the bias power in the flyback converter circuitry.
  • One such embodiment configured to implement this practical situation is presented in the circuit 2300 of FIG. 16.
  • a bias winding (L3, 166) with N3 turns in the corresponding coil
  • a synchronous rectifier (SRB, 68), and a bias capacitor (CB1, 72) are included and form an electronic sub-circuit 2310 of the electronic circuitry 2300.
  • the synchronous rectifier SRB, 68 in the one embodiment can be configured as a simple diode. Based on such arrangements, in cases where the current injected in the bias portion 2310 is larger than the current required by the bias portion, the extra energy is transferred to the secondary side via the syncronous rectifier SRB, 68.
  • Example 5 Using the Leakage Inductance Energy Via a High-Efficiency Rompower Passive Clamp to Obtain Zero-Voltage Switching Conditions for the Main Switch of the Circuit
  • the harvested energy of the leakage inductance may be re-used to reduce (and even eliminate) one of the largest losses in the flyback type converter, specifically, the switching losses associated with the hard discharge of the parasitic capacitance (Ceq, 20) when the main switch (Ml, 28) is turned on.
  • the circuit 2500 of FIG. 17 achieves this goal: this circuit 2500 harvests, in operation as illustrated schematically in FIG. 17, the leakage inductance energy and uses such energy to discharge the parasitic capacitance of the capacitor Ceq across the main switch Ml in order to create zero voltage switching conditions for the main switch Ml.
  • the embodiment of the circuit 2500 is configured so that the capacitor (Crc, 72) is charged with the harvested leakage inductance energy, and, in operation, the energy from the capacitor (Crc, 72), is used to inject a pulse of current into the transformer (Tr, 16) via the auxiliary (current injection winding 78 with the purpose of discharging the parasitic capacitance (Ceg, 20).
  • the current injection circuit is formed with the use of a current injection winding (Linj, 78), a current injection switch (Minj, 80) such as a Mosfet, controlled by a current injection control signal (VcMinj, 82), and an energy source (represented by charged capacitor Crc).
  • an optional smaller capacitor (Cinj, 76) is also added as shown.
  • the capacitance of the capacitor (Cinj, 76) is preferably at least ten times lower than the capacitance of the capacitor (Crc).
  • the optional capacitor (Cinj, 76) is charged in forward mode via the winding 78, during the period of conduction of the main switch (Ml, 18).
  • the energy in this capacitor (Cinj, 76) is preferably much smaller than the energy coming from the capacitor (Crc, 72).
  • the use of the capacitor Cinj is to shape the current through the current injection switch (Minj, 80) to become negative before the moment of time when the (VcMinj, 80) turns off.
  • the capacitor Cinj adds energy into the current injection circuit.
  • electronic circuitry has primary and secondary sides and includes: (a) a flyback power converter that includes an input voltage source, a transformer having a primary winding on the primary side and a secondary winding on the secondary side, respectively, a main switch in series with the primary winding on the primary side, a parasitic capacitor across the main switch, and a synchronous rectifier in series with the secondary winding on the secondary side; (b) a passive clamp circuit across the main switch, wherein the passive clamp circuit contains a clamp diode and the clamp capacitor in series with the clamp diode; and (c) a current injection circuit including a current injection winding, a current injection switch connected to a first terminal of the current injection circuit, and a source of energy between a second terminal of the current injection circuit and the found.
  • a flyback power converter that includes an input voltage source, a transformer having a primary winding on the primary side and a secondary winding on the secondary side, respectively, a main switch in series with the primary winding on the primary side, a parasitic capacitor across
  • the current injection circuit is configured, in operation of the electronic circuitry, to collect energy of a leakage inductance of the electronic circuitry and to inject this energy in a form of a pulse of current into the transformer via the current injection winding to discharge the parasitic capacitor to create a zero voltage switching condition for the main switch.
  • the above electronic circuitry may additionally include an auxiliary circuit that contains two rectifiers connected such that a cathode of a first of the two rectifiers is directly electrically connected with a cathode of a passive clamp circuit at a second node, an anode of a second of the two rectifiers is directly electrically connected with the cathode of the first of the two rectifiers at the second node, and a control switch is connected between the second node and the ground.
  • the embodiment of the electronic circuitry may further include a control electronic circuitry configured to generate a control signal governing an operation of the control switch and electrically connected to each of the main switch, the bias synchronous rectifier, and the control switch.
  • the combination of the flyback converter with the passive clamp is characterized by a first value of RMS current through the clamp capacitor.
  • the auxiliary circuit is configured to reduce the RMS current through the clamp capacitor from a first value to a second value, such that the RMS current passing through the clamp capacitor of the electronic circuit has the second value, which is at least 40% lower than the first value.
  • the clamp diode may be configured: i) to be turned on at a moment of time after the main switch is turned off; and ii) to be turned off at the moment of time prior to the moment of time at which current passing through the secondary winding reaches a zero level.
  • the source of energy is configured as a first capacitor
  • the current injection circuit additionally includes an injection capacitor in parallel with the first capacitor, to shape, in operation of the electronic circuitry, a current through the current injection switch to become negative before a moment of time when the current injection switch turns off.
  • a capacitance of the first capacitor is a least then times higher than a capacitance of the injection capacitor.
  • Key waveforms of the circuit 2500 of FIG. 17 are presented in FIG. 19 in order from top to bottom and include: 1) the control signal (VcMl, 22) for the main switch (Ml, 18); 2) the voltage (VdsMl) across the main switch (Ml, 18); 3) the magnetizing current Im(Tr) through the transformer (Tr, 16); 4) the voltage VCinj across the capacitor Cinj; and 5) the voltage VCrc across (Crc,72); 6) ID1, the current through the rectifier (Dl, 58), 7) the current through the switch (Minj, 80) and the current coming from (Crc,72); and 8) the control signal, VcMinj, for the current injection switch.
  • the mains switch Ml turns off when the magnetizing current IM(Tr) is at its peak.
  • the current through the leakage inductance continues to flow via (Del, 86) and the clamp capacitor Cr, 43) and further through the rectifier D 1,58 to charge the capacitor (Crc, 72).
  • the voltage across the capacitor (Crc, 72) increases between the times tl and t2 (as shown in FIG. 19), with the shape of the waveform VCrc affected by the current passing through Dl being injected in the capacitor Crc.
  • the switch Minj is turned on by the control signal (VcMinj, 82).
  • the capacitor (Cinj, 76) starts to discharge and the current through the switch Minj increases.
  • the leakage inductance of the transformer forms a resonant circuit with the capacitor (Ceq, 20), and the current through Minj is substantially sinusoidal.
  • the voltage across (Cinj, 76) decays until it reaches the voltage level across Ccr at the time t5.
  • the current injection is provided by the energy contained in the capacitor Crc (in parallel with Cinj, in this example).
  • the capacitance of Cinj be much smaller than the capacitance of the Crc (one-tenth in value or even smaller), in which case, after time t5, most of the energy is delivered from the capacitor Crc.
  • the current injection current (that is, the current through Minj) reaches zero.
  • the current through Minj becomes negative, because it flows into the Cinj capacitor charging it between the times t6 and t7, when both the main switch Ml and the current injection switch Minj are on.
  • the current injection switch Minj is turned off by the control signal VcMinj.
  • the current injection circuit 2700 depicted in FIG. 18 is also operational without the capacitor Cinj.
  • the role of the capacitor Cinj is not essential to the performance of this circuit 2700, because most of the energy is delivered from the capacitor Crc, energy which comes from the leakage inductance energy of the transformer (Tr, 20).
  • the amplitude of the current injection is a self- adjusting function of the voltage across the main switch when Minj is turned on.
  • the current injection circuit the voltage across Crc does not change significantly.
  • the current injection amplitude is a function of the voltage across the main switch Ml at the time when the current injection turns on. For low voltage across the main switch, the amplitude of the current injection is small. If the voltage across the main switch is high when the current injection tuns on, then the amplitude of the current injection will be high.
  • the amplitude of the current injection will be small, and if the turn“on” of Minj occurs at ahigher input voltage across Ml, the amplitude of the current injection will increase.
  • the voltage across Ml decays as a result of the discharge of parasitic capacitance (Ceq, 20) by the current injection through Minj reflected into the primary winding 12. At t6, the voltage across Ml is zero, which creates zero voltage switching conditions for Ml.
  • the leakage inductance energy is used to discharge the parasitic capacitance (Ceq, 20) and create zero voltage switching conditions for Ml.
  • FIG. 18 shows the clamp diode formed by several diodes in parallel (Del, Dc2, ... Den), which brings several additional benefits.
  • One benefit is the distribution of the leakage inductance current through each diode which reduces the dl/dt through each diode and reduces the amplitude of the forward voltage across the diode due to the forward reverse recovery characteristics of the diodes.
  • the charge in the junctions of all the diodes is increased, which allows better operation of the high-efficiency passive clamp in marginal conditions when the leakage inductance in the transformer is higher or when the amplitude of the magnetizing current is higher.
  • Such method includes the steps of: (a) turning the main switch off at time tl, when a magnetizing current of the transformer is at a peak reached during a period of conduction of the main switch; (b) between times tl and t2, charging the source of energy (Crc,72), with a current from a leakage inductance of the electronic circuitry that has passed through the passive clamp and through the second of the two rectifiers, to increase a voltage across the source of energy; and (c) transferring energy contained in the magnetizing current to the secondary side via the synchronous rectifier to store the energy in an output capacitor disposed between the ground and a terminal of the secondary winding, so as to harvest energy of leakage inductance of the electronic circuitry and the use the energy to discharge the parasitic capacitor to crease zero-voltage switching conditions for the main switch.
  • the method additionally includes the step of, at time t3 (after t2), having a substantially constant voltage across the main switch changed after time t2 to an oscillating voltage and then turning the current injection switch on at the valley at time t4.
  • the method may also include the following steps: after the current injection switch has been turned on, discharging the source of energy to shape a current passing through the current injection switch to be substantially sinusoidal, and switching off the current injection switch at a moment when the current passing through the current injection switch is negative or substantially zero.
  • the switching off may include switching off the current injection switch after a moment when the current passing through the current injection switch became negative.
  • FIG. 20 depicts two experimental waveforms of the voltage across the main switch of the circuit 2500 of FIG. 17.
  • the plot at the top of the page illustrates the voltage across the main switch (Ml, 318) when the input voltage Vin, 10 is 115 volts AC.
  • the plot at the bottom of the page illustrates the voltage across the main switch (Ml, 318) when the input voltage Vin, 10 is 230 volts AC.
  • use of the passive clamp eliminates spikes and ringing across the main switch (Ml, 138) when the main switch is turned off.
  • FIG. 21 depicts an experimental waveform of the voltage VdsMl across the main switch (Ml, 318) of the circuit 2500 of FIG. 17 when the input voltage is 115 volts AC.
  • FIG. 21 depicts an experimental waveform of the current IDcl through the passive clamp of the circuit 2500, also when the input voltage is 115 volts AC.
  • FIG. 22 presents an example within a forward topology in which the reset of the transformer is done through an auxiliary winding. It uses only one power switch in the primary such as the flyback topology.
  • the reset of the transformer is done with an auxiliary winding.
  • the number of turns in the auxiliary winding (LR, 302) is equal to the number of turns in the primary of the transformer, (LI, 312) and the voltage across the main switch (Ml, 318) at turn off reaches an amplitude of 2*Vin.
  • the voltage across (Ml, Ml) can be higher than 2*Vin and, as a result, the maximum duty cycle can be higher than 50%.
  • FIG. 23 depicts the key waveforms of this topology. These include, in order from top to bottom: 1) the control signal (VcMl, 322) for the main switch (Ml, 318); 2) the voltage (VdsMl) across the main switch (Ml, 318); 3) the current IL1 through the primary winding of the transformer (Tr, 316); 4) the current ICr through the clamp capacitor (Cr, 343); and 5) the current ILR through the reset winding (LR, 302).
  • the reverse charge flowing through the diode (Del, 386) is equal with the forward charge injected in the clamp capacitor (Cr, 343).
  • the reverse charge is smaller than reverse recovery charge of (Del, 386). If the above condition is met, then the simple diode (Del, 386) operates as an active clamp and the energy from the leakage inductance of the transformer (Tr, 316) is partially transferred to the secondary winding and further via (Lo,337).
  • the voltage VdsMl spikes across the main switch when the main switch turns off. This occurs when the embodiments disclosed herein are not used. With those embodiments, the leakage inductance energy is partially recycled and some of it is used to charge (Crc,372) to be further used to energize the current injection circuit and obtain zero voltage switching conditions across the main switch, when the main switch is turned on.
  • Another application of this embodiment is for the active forward with active clamp topology wherein the sub circuit formed by the clamp switch connected in series with the clamp capacitor is not placed across the main switch or across the primary winding but is instead connected across an auxiliary winding.
  • This active clamp forward topology contains a main switch and a clamp switch which is complementary to the main switch and a clamp capacitor in series with the clamp switch.
  • the sub circuit formed by the clamp switch in series with the clamp capacitor is generally connected across the main switch, or it may be connected across the primary winding. The clamp switch is on when the main switch is off and the clamp switch is off when the main switch is off.
  • the advantage of this topology is that the reset of the transformer occurs during the entire off time of the main switch and, as a result, the voltage across the main switch is minimized for a given duty cycle of operation.
  • the flux through the transformer is symmetrical to zero which leads to better utilization of the magnetic core of the transformer.
  • One disadvantage of this topology for high voltage input application is that it requires another high voltage switch which is the clamp switch. Further, a high voltage floating drive is required to control the active clamp switch while the main switch is driven from the ground level.
  • One solution to avoid this complexity and cost is to place the clamp switch in series with the clamp capacitor across an auxiliary winding. In such case, by choosing the right number of turns in the auxiliary winding, the clamp switch can have a lower voltage rating. In addition, the clamp switch can be driven from the ground level.
  • a passive clamp such as the one presented in FIG.14 can be used across the main switch.
  • the passive clamp would be formed by Del, Cr, Dl, D2, Rs and a voltage source VB which can be the bias power of the converter.
  • the diode Dl can be connected to the current injection circuit, as depicted in FIG.22.
  • the active clamp forward topology can be implemented in a very simple way to eliminate voltage stress associated with the spikes and glitches across the main switch due to the leakage inductance in between the primary winding and the auxiliary winding.
  • the embodiments disclosed herein can apply to any circuit containing an input voltage source connected to a transformer primary winding, where the transformer contains other additional windings and there is a leakage inductance between the primary winding and the additional winding(s).
  • the circuit would further contain a main switch in series with the transformer primary winding, wherein the magnetizing current of the transformer has a low impedance path to further circulate after the main switch turns off.
  • the circuit would also include a passive clamp formed from a diode and a capacitor in series, where the passive clamp is further connected to an energy extraction circuit that includes two additional rectifiers connected in series with one another and in series with an electronic component configured to store electromagnetic energy, and wherein a cathode of the first of the two additional rectifiers is directly electrically connected with the passive clamp circuit, and wherein an anode of the second of the two additional rectifiers is directly electrically connected with the cathode of the first of the two additional rectifiers, and wherein a cathode of the second of the two additional rectifiers is directly electrically connected with a first terminal of the electronic component, and wherein a second terminal of the electronic component is electrically connected with the anode of the first of the two additional rectifiers.
  • This concept applies to any other topology with leakage inductance in the transformer.
  • the magnetizing current has a low impedance path to circulate after the main switch turns off; this is the secondary winding of the transformer.
  • the low impedance path for the magnetizing current after the main switch turns off is through the rest winding.
  • Example 7 Applications as a Snubber
  • the embodiments described herein can be applied to many other applications, such as snubber circuits in DC-DC converters.
  • abrupt voltage changes can result in transient ringing in the primary and circuit of a transformer.
  • the leakage inductance of the transformer may interact resonantly with the reverse recovery characteristics of the diode(s) or with the junction capacitance of the diodes and other parasitic capacitance in the circuit. That will lead to large spikes and ringing across the diodes. That may require larger voltage diodes for which some of the key characteristics will be sacrificed, such as the voltage drop.
  • FIG. 24A illustrates the most common prior art solution to suppress the voltage spikes across the rectifiers, namely, using a typical RC snubber.
  • the RC snubber is formed by a capacitor (Cs,413) and a resistor (Rs,415).
  • FIG. 24B presents the voltage across the diode (DB,416) without an RC snubber as the waveform 450, and the voltage across (DB,416) with an RC snubber as the waveform 452.
  • the problems with the RC snubber are that the first spike does not attenuate as much and that this snubber technique is dissipative, negatively impacting the efficiency of the DC-DC converter.
  • active snubbers were introduced.
  • FIG. 25A This snubber technology is presented in FIG. 25A.
  • a controlled switch (SW,432) is introduced in series with an RC snubber network formed by (Cx,434) and (Rx,436).
  • the current path to the capacitor (Cs,434) is interrupted by the switch (SW, 432) before the capacitor (Cx, 434) discharges and thereafter at each such voltage change at each cycle, the capacitor (Cx,434) is no longer charged from its totally discharged state but nevertheless damps the ringing.
  • This snubber technique is more efficient and also attenuates the first spike more effectively.
  • FIG. 25B presents several key waveforms associated with this snubbing technique.
  • the voltage Vcs across (Cx, 434) is depicted by waveform 456.
  • the voltage across the snubber capacitor (Cx, 434) is not discharged to zero at each cycle, as occurs in the standard RC snubber of FIG. 24A. Though this snubbing technique is more effective in the attenuation of the first spike and also very efficient, it is relatively complex because it requires a control signal, 430, to timely control the switch (SW, 432).
  • the embodiments can be used very effectively in a passive snubber as depicted in FIG. 26A.
  • the passive snubber shown there is a passive clamp circuit known as a Rompower Passive Clamp or Rompower Passive Snubber.
  • the passive snubber of FIG. 26A includes a diode (Del, 470) with a large reverse recovery time and soft reverse recovery characteristics, and a capacitor (Cr, 472) in series with an auxiliary or energy extraction circuit 480.
  • the energy extraction circuit 480 is an electronic component including two diodes (Dl, 474) and (D2, 476) which are in series with the resistor (Ry, 479) and a voltage source (VB, 478) placed in the cathode of (Dl, 474).
  • FIG. 26B presents some of the key waveforms of the clamp circuit of FIG. 26A.
  • the voltage in (A) rises rapidly, caused by the increase of the voltage of (Vin, 410).
  • the current I(Dcl) flows through (Del, 470), the capacitor (Cr, 472) and the diode (Dl, 474), thereby injecting current in the voltage source (VB,478).
  • a portion of the energy from the leakage inductance (Llk, 412) is delivered to the output, to (Lo,418), via reverse recovery of (Del, 470) (Lo,418) which decreases the current flowing from the input voltage source (Vin, 410) via (DA, 414).
  • the damping of the ringing across (DB,416) is due to the energy extraction circuit 480, which takes the ringing energy and transfers it into the auxiliary energy storage (VB,478).
  • VB can be the bias voltage of the DC-DC converter which incorporates the circuit from FIG. 26A.
  • This snubber technique is a lossless snubber technique because the damping effect is not through power dissipation but through energy extraction, wherein the ringing energy is used for other power needs in power conversion such as bias power.
  • diode Del in FIG. 26A can be eliminated and replaced by a short.
  • the energy from the leakage inductance contained in Llk, 412 is transferred to the voltage source VB via the clamp capacitor Cr and the diode Dl. Cr is then directly connected to the node (A) at the cathode of DA and DB.
  • This snubber technique works by extracting the energy from the leakage inductance, Llk, 412, and transferring it to the voltage source VB.
  • the clamp capacitor Cr will be recharged when the voltage in the node A swings to a lower voltage during the conduction of the diode DB.
  • the elimination of the diode Del simplifies the circuit and eliminates restrictions associated with the reverse recovery of Del, 470.
  • the snubber circuit formed by just Cr, Dl, D2, Ry and VB and without the diode Del is another embodiment.
  • the elimination of the ringing and spikes in the node A is done by extracting the energy from the leakage inductance Llk, 412 and transferring this energy to the voltage source VB.
  • VB in this case can be the bias supply or can supply the energy for the current injection circuit or any other applications.
  • Extracting the energy form the parasitic elements such as the energy from the leakage inductance and using it for other purposes such as the bias supply is a very efficient way of eliminating ringing and spikes across the switch elements.
  • the diodes DA and DB in many applications are substituted by Mosfets which will work as synchronous rectifiers, and the snubber circuit described herein will work in the same way.
  • the specified characteristic or quality descriptor means“mostly”, “mainly”, “considerably”,“by and large”, “essentially”,“to great or significant extent”, “largely but not necessarily wholly the same” such as to reasonable denote language of approximation and describe the specified characteristic or descriptor so that its scope would be understood by a person of skill in the art.
  • the terms“approximately”, “substantially”, and“about”, when used in reference to a numerical value represent a range of plus or minus 20% with respect to the specified value, more preferably plus or minus 10%, even more preferably plus or minus 5%, most preferably plus or minus 2% with respect to the specified value.
  • two values being“substantially equal” to one another implies that the difference between the two values may be within the range of +/-20% of the value itself, preferably within the +/-10% range of the value itself, more preferably within the range of +1-5% of the value itself, and even more preferably within the range of +/-2% or less of the value itself.
  • the term substantially equivalent is used in the same fashion.

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Dc-Dc Converters (AREA)

Abstract

Circuit ayant des côtés primaire et secondaire comprenant un convertisseur indirect ayant une source de tension d'entrée (Vin, 410 (26)) et un transformateur ayant des enroulements primaire et secondaire. Un commutateur principal est en série avec l'enroulement primaire. Un circuit de calage passif comprend une diode de calage (Dcl, 470), un condensateur de calage (Cr, 472) et un circuit auxiliaire (480) comprenant des premier (D1, 474) et second (D2, 476) redresseurs en série l'un avec l'autre et avec un composant électronique (VB) conçu pour stocker de l'énergie électromagnétique. Le composant électronique (VB) comprend des première et seconde bornes. Une cathode du premier redresseur (D1, 474) est connectée au circuit de calage passif, et une anode du premier redresseur (D1, 474) est connectée à la seconde borne du composant électronique. Une anode du second redresseur (D2, 476) est connectée à la cathode du premier redresseur (D1, 474), et une cathode du second redresseur (D2, 476) est connectée à la première borne du composant électronique (VB).
PCT/US2020/017672 2019-05-01 2020-02-11 Calage passif à grande efficacité WO2020222889A1 (fr)

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CN202080047002.XA CN114245961A (zh) 2019-05-01 2020-02-11 高效无源箝位

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US201962841694P 2019-05-01 2019-05-01
US62/841,694 2019-05-01
US16/503,432 2019-07-03
US16/503,432 US10574148B1 (en) 2019-07-03 2019-07-03 Self-adjusting current injection technology
US16/775,967 US10972014B2 (en) 2017-10-12 2020-01-29 High efficiency passive clamp
US16/775,967 2020-01-29

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