WO2020100275A1

WO2020100275A1 - Radar device and signal processing method

Info

Publication number: WO2020100275A1
Application number: PCT/JP2018/042432
Authority: WO
Inventors: 聡影目; 照幸原
Original assignee: 三菱電機株式会社
Priority date: 2018-11-16
Filing date: 2018-11-16
Publication date: 2020-05-22
Also published as: JP6664554B1; JPWO2020100275A1

Abstract

This radar device (1) comprises a pulse width control unit (14), signal generation circuit (10), transmission and reception unit (11), reception circuit (13), correlation processing unit (41), and domain conversion unit (44). The pulse width control unit (14) sets a plurality of pulse widths so as to form a distribution of pulse widths that increase or decrease in steps within a set range greater than or equal to a standard pulse width. The signal generation circuit (10) continuously generates a plurality of transmission pulse signals respectively having the plurality of pulse widths. When the transmission and reception unit (11) receives a plurality of reflected wave signals from an external space, the reception circuit (13) samples each of the plurality of reflected wave signals and generates a plurality of reception signals. The correlation processing unit (41) carries out correlation processing on each of the plurality of reception signals and generates a plurality of pulse compression signals. The domain conversion unit (44) generates a plurality of frequency domain signals by carrying out domain conversion processing on the plurality of pulse compression signals.

Description

Radar device and signal processing method

The present invention relates to radar technology for detecting a target such as a moving object.

A general pulse Doppler radar continuously transmits a plurality of pulse waves at a pulse repetition period (Pulse Repetition Interval, PRI), and then receives a plurality of reflected waves corresponding to the plurality of pulse waves from a target. It is possible to generate a plurality of received signals and estimate target information such as a target relative speed based on the plurality of received signals. It is possible to improve the signal-to-noise ratio (Signal-to-Noise Ratio, SNR) by coherently integrating the plurality of received signals in the pulse hit direction. Such a pulse Doppler radar is disclosed, for example, in Patent Document 1 (JP-A-6-294864).

JP-A-6-294864 (see, for example, FIG. 1)

However, the above-mentioned pulse Doppler radar has a problem that the improvement effect of the signal-to-noise ratio by coherent integration is limited. For example, when a state called a transmission blind occurs, a pulsed Doppler radar causes a loss of a reflected wave from a target. In this case, even if the coherent integration of the received signal is executed, the effect of improving the signal-to-noise ratio is limited. Here, the transmission blind refers to a state in which the reflected wave from the target cannot be received during the transmission time of each pulse wave. In order to detect a target at a relatively long distance, the pulse width of each pulse wave may be widened to increase the average transmission power. However, if the pulse width is wide, transmission blinds are likely to occur.

In view of the above, an object of the present invention is to provide a radar device and a signal processing method capable of improving the signal-to-noise ratio (SNR) while suppressing the influence of the transmission blind.

A radar device according to an aspect of the present invention includes a pulse width control unit that sets a plurality of pulse widths that form a pulse width distribution that gradually increases or decreases within a setting range equal to or greater than a predetermined reference pulse width, and A signal generation circuit for continuously generating a plurality of transmission pulse signals each having a plurality of pulse widths and subjected to intra-pulse modulation, and for transmitting the plurality of transmission pulse signals to an external space, the external space From the transmitting / receiving unit that receives a plurality of reflected wave signals respectively corresponding to the plurality of transmitted pulse signals, and a plurality of received signals that respectively correspond to the plurality of transmitted pulse signals by sampling each of the plurality of reflected wave signals. A receiving circuit that generates a signal, a correlation processing unit that generates a plurality of pulse compression signals by performing a correlation process using a reference signal for each of the plurality of reception signals, and a plurality of pulse compression signals On the other hand, a domain conversion unit that generates a plurality of frequency domain signals by performing a domain conversion process from the time domain to the frequency domain, and a target detection unit that detects a target candidate based on the plurality of frequency domain signals. ..

According to one aspect of the present invention, a plurality of transmission pulses that have been subjected to intra-pulse modulation using a plurality of pulse widths that form a pulse width distribution that gradually increases or decreases within a setting range that is equal to or larger than the reference pulse width. Since the signals are continuously generated, it is possible to generate a high SNR pulse compression signal while suppressing the influence of the transmission blind. As a result, since a high SNR frequency domain signal is generated, it is possible to provide a radar device with improved target detection performance.

It is a block diagram which shows schematic structure of the radar apparatus of Embodiment 1 which concerns on this invention. FIG. 3 is a block diagram schematically showing a configuration example of a signal generation circuit of the first embodiment. 3 is a block diagram schematically showing a configuration example of a receiving circuit according to the first embodiment. FIG. 3 is a block diagram showing an example of a hardware configuration that implements the functions of a pulse width control unit and a radar signal processing circuit according to the first embodiment. FIG. 3 is a flowchart schematically showing an operation procedure of the radar device according to the first embodiment. 6A and 6B are diagrams schematically showing an example of a transmitted wave and an example of a reflected wave corresponding to the transmitted wave. 7A and 7B are diagrams schematically showing another example of the transmitted wave and an example of the reflected wave corresponding to the transmitted wave. FIG. 8A and FIG. 8B are diagrams schematically showing an example of a transmission wave composed of a transmission pulse signal group forming a pulse width distribution that gradually increases and an example of a reflected wave corresponding to the transmission wave. 9A and 9B are diagrams schematically showing an example of a transmission wave composed of a transmission pulse signal group forming a pulse width distribution that gradually decreases and an example of a reflected wave corresponding to the transmission wave. 10A and 10B are diagrams schematically showing an example of a transmission wave composed of a transmission pulse signal group that forms a pulse width distribution that gradually increases and then decreases, and an example of a reflected wave corresponding to the transmission wave. Is. 11A and 11B are graphs schematically showing an example of power distribution of a pulse compression signal. 12 is a graph schematically showing a signal-to-noise ratio (SNR) obtained from the power distributions shown in FIGS. 11A and 11B. It is a graph which shows roughly the example of the power distribution of a pulse compression signal. It is a graph which shows roughly SNR obtained from the electric power distribution shown to FIG. 11A and FIG. It is a graph which shows the example of a frequency domain signal. It is a block diagram which shows schematic structure of the radar apparatus of Embodiment 2 which concerns on this invention. 17A and 17B are graphs showing frequency domain signals obtained by Fourier transforming the reference signal. It is a graph which shows the relationship between SNR and a distance of a frequency domain signal.

Hereinafter, various embodiments according to the present invention will be described in detail with reference to the drawings. In addition, the components denoted by the same reference numerals throughout the drawings have the same configuration and the same function.

Embodiment 1.
1 is a block diagram showing a schematic configuration of a radar device 1 according to a first embodiment of the present invention. As shown in FIG. 1, the radar device 1 has a plurality of transmission pulse signals Tx (h, t) each having a plurality of pulse widths T _p (h) at a predetermined pulse repetition period (Pulse Repetition Interval, PRI). ), And the plurality of transmission pulse signals Tx (h, t), which are output to the antenna (antenna) 12, and then correspond to the plurality of transmission pulse signals Tx (h, t), respectively. A transmitting / receiving unit 11 that receives a plurality of reflected wave signals Rx (h, t), and a plurality of received analog signals W ₀ (h, t) by performing analog signal processing on the plurality of reflected wave signals Rx (h, t). And a receiving circuit 13 for converting the plurality of receiving analog signals W ₀ (h, t) into a plurality of receiving digital signals (receiving video signals) V ₀ (h, m), and the plurality of receiving digital signals. A radar signal processing circuit 31 that detects a target candidate by performing digital signal processing on V ₀ (h, m) and a display device 60 that displays the detection result are provided.

Here, the pulse width T _p (h), the transmission pulse signal Tx (h, t), the reflected wave signal Rx (h, t), the reception analog signal W ₀ (h, t), and the reception digital signal V ₀ (h, t). In m), the variable t represents time, the variable h is an integer in the range of 0 to H-1 representing the pulse hit number, and H is the pulse hit number. Hereinafter, the pulse hit number h will be referred to as “hit number h”. The variable m in the received digital signal V ₀ (h, m) is an integer in the range of 0 to M−1 that represents the sampling number, and M is the number of sampling points in the pulse repetition period.

The antenna 12 can radiate a transmission wave Tw corresponding to the transmission pulse signals Tx (0, t) to Tx (H-1, t) to the external space, and then returns from the target Tgt in the external space. The reflected wave Rw is received. The transmission / reception unit 11 outputs the reflected wave signals Rx (0, t) to Rx (H-1, t) corresponding to the reception output of the antenna 12 to the reception circuit 13. As the frequency band used by the radar device 1, for example, a frequency band such as a millimeter wave band or a microwave band can be used.

Further, as shown in FIG. 1, the radar device 1 includes a pulse width control unit 14 that sets a plurality of pulse widths T _p (h) used in the signal generation circuit 10 and the radar signal processing circuit 31. As will be described later, the pulse width control unit 14 forms a plurality of pulse widths T _p (that forms at least one pulse width distribution that increases or decreases stepwise within a preset reference pulse width T ₀ or larger setting range). It has a function of setting 0) to T _p (H-1). The pulse width control unit 14 is a component different from the signal generation circuit 10, but is not limited to this. There may be an embodiment in which the pulse width control unit 14 is incorporated in the signal generation circuit 10 or the radar signal processing circuit 31.

FIG. 2 is a block diagram schematically showing a configuration example of the signal generation circuit 10 according to the first embodiment. As shown in FIG. 2, the signal generation circuit 10 includes a local oscillator 20, a pulse generator 21, an intrapulse modulator 22, and an output unit 23. The local oscillator 20 generates a local oscillation signal _L 0 usable frequency band (t), and outputs the local oscillation signal _L 0 (t) to the pulse generator 21 and the reception circuit 13.

Specifically, the local oscillator 20 has a constant transmission frequency f ₀ within a certain observation period (the period from time t = 0 to time t = T _obs ) as shown in the following equation (1). A local oscillation signal L ₀ (t) can be generated.

Here, t is the time, A _L is the amplitude of the local oscillation signal L ₀ (t), φ ₀ is the initial phase of the local oscillation signal L ₀ (t), _Tobs is the upper limit of the observation period, and j is the imaginary unit. .

Next, the pulse generator 21 shown in FIG. 2 operates in synchronization with the pulse repetition period T _pri, and modulates the local oscillation signal L ₀ (t) to set the pulse set by the pulse width control unit 14. It is possible to generate pulse signals L _pls (0, t) to L _pls (H-1, t) having widths T _p (0) to T _p (H-1), respectively. For example, the pulse generator 21 can generate a plurality of pulse signals L _pls (h, t) (h = 0, 1, ..., H−1) represented by the following equation (2).

In Expression (2), A _L is the amplitude of the pulse signal L _pls (h, t), and Ω [h] is a set of times t that satisfy the following Expression (3).

Next, the intra-pulse modulator 22 subjects each of the plurality of pulse signals to intra-pulse modulation to generate a plurality of intra-pulse modulation signals as transmission pulse signals Tx (h, t). Examples of intra-pulse modulation include known frequency modulation such as chirp modulation. The output unit 23 outputs the transmission pulse signals Tx (h, t) to the transmission / reception unit 11. At this time, the output unit 23 may perform processing such as amplification on the transmission pulse signal Tx (h, t). Specifically, the intra-pulse modulator 22 first uses the modulation bandwidth B ₀ according to the following equation (4) to modulate the pulse signal L _pls (h, t) with a modulation control signal L _chp. (H, t) can be generated.

Further, the intra-pulse modulator 22 has the intra-pulse modulation signal that is frequency-modulated using the modulation control signal L _chp (h, t), that is, the transmission pulse signal Tx (h, t) can be generated.

The antenna 12 can radiate a plurality of transmission pulse signals Tx (h, t) to the external space as transmission waves Tw, and then receive the reflected waves Rw returning from the target Tgt in the external space. The transmitter / receiver 11 can output the reflected wave signal Rx (h, t) as expressed by the following equation (6).

In the formula (6), _{A R} is the amplitude of the reflected wave signal reflected by the target Tgt Rx (h, t), R 0 is the initial target relative distance, v is the target relative speed, tau is localized in the 1 pulse At a certain time, c is the speed of light. Further, Λ [h] is a set of times t that satisfy the following expression (7).

Next, the configuration of the receiving circuit 13 will be described. FIG. 3 is a block diagram schematically showing a configuration example of the receiving circuit 13. As shown in FIG. 3, the reception circuit 13 includes a down converter (mixer) 24, a bandpass filter 25, an amplifier 26, a phase detector 27, and an A / D converter 28.

The down converter 24 shown in FIG. 3 converts the reflected wave signal Rx (h, t) into an analog signal in a lower frequency band (for example, an intermediate frequency band). The bandpass filter 25 filters the analog signal and outputs a filtered signal. The amplifier 26 amplifies the filter signal and outputs an amplified signal. Then, the phase detector 27 phase-detects the amplified signal and generates a detection signal including an in-phase component and a quadrature component as a reception analog signal W ₀ (h, t). The following expression (8) is an expression representing the received analog signal W ₀ (h, m).

Here, A _V indicates the amplitude of the received analog signal W ₀ (h, t), and the upper right subscript “*” indicates complex conjugate. The local oscillation signal L ₀ ^* (t) is a complex conjugate of the local oscillation signal L ₀ (t).

The A / D converter 28 samples the reception analog signal W ₀ (h, t) at a sampling interval Δt corresponding to a predetermined sampling frequency f _s , so that the reception digital signal as expressed by the following equation (9) is obtained. A signal (received video signal) V ₀ (h, m) can be generated.

In Equation (9), A is the amplitude of the received digital signal V ₀ (h, m), m is an integer in the range of 0 to M−1 that represents the sampling number, and Ψ [h] is It is a set of sampling numbers m that satisfy the conditional expression of the following expression (10).

The radar signal processing circuit 31 can perform digital signal processing on the received digital signal V ₀ (h, m) to detect a target candidate. As shown in FIG. 1, the radar signal processing circuit 31 includes a correlation processing unit 41, a region conversion unit 44, and a target detection unit 50. The correlation processing unit 41 performs a correlation process using the reference signal and the pulse width T _p (h) on the received digital signal V ₀ (h, m) to obtain the pulse compression signal F _{V · Ex} (h, m). m) is generated. Details of the correlation processing will be described later. The domain transforming unit 44 performs domain transform processing from the time domain to the frequency domain in the pulse hit direction on the pulse compression signal F _{V · Ex} (h, m) to _{generate the} frequency domain signal f _d (h _fft , m). ) Is generated. As the domain transform processing, discrete Fourier transform based on a predetermined algorithm such as a fast Fourier transform (Fast Fourier Transform, FFT) algorithm or a chirp z transform (Chirp Z-Transform, CZT) algorithm may be executed.

The target detection unit 50 includes a target candidate detection unit 51 that detects a target candidate based on the frequency domain signal f _d (h _fft , m), and a target candidate information calculation unit 52 that calculates target information regarding the detected target candidate. And have.

All or a part of the hardware configuration of the pulse width control unit 14 and the radar signal processing circuit 31 described above is an LSI (Large Scale Integrated) such as an ASIC (Application Specific Integrated Circuit) or an FPGA (Field-Programmable Gate Array). It should be realized.

FIG. 4 is a block diagram showing a hardware configuration example for realizing the functions of the pulse width control unit 14 and the radar signal processing circuit 31. The signal processing circuit 70 shown in FIG. 4 is configured to include a processor 71 configured by an LSI, an input / output interface 74, a memory 72, a storage device 73, and a signal path 75. The signal path 75 is a bus for connecting the processor 71, the input / output interface 74, the memory 72, the storage device 73, and the signal path 75 to each other. The processor 71 is connected to the display unit 60 and the receiving circuit 13 via the input / output interface 74.

The memory 72 is, for example, a program memory that stores various program codes to be executed by the processor 71 in order to realize the functions of the pulse width control unit 14 and the radar signal processing circuit 31, and when the processor 71 executes digital signal processing. And a temporary storage memory in which data used in the digital signal processing is expanded. As the memory 72, a plurality of semiconductor memories such as a ROM (Read Only Memory) and an SDRAM (Synchronous Dynamic Random Access Memory) may be used.

The processor 71 can access the storage device 73. The storage device 73 is used to store various data such as setting data to be used by the processor 71 and signal data generated by the processor 71. As the storage device 73, for example, a volatile memory such as SDRAM, a HDD (Hard Disk Drive), or an SSD (Solid State Drive) can be used. It should be noted that the storage device 73 can also store data to be stored in the memory 72.

In the example of FIG. 4, the signal processing circuit 70 is realized by using the single processor 71, but it is not limited to this. The functions of the pulse width control unit 14 and the radar signal processing circuit 31 may be realized by using a plurality of processors that operate in cooperation with each other. Furthermore, any of the functions of the pulse width control unit 14 and the radar signal processing circuit 31 may be realized by dedicated hardware.

Next, the configuration and operation of the radar device 1 will be described in detail with reference to FIG. FIG. 5 is a flowchart schematically showing an operation procedure of the radar device 1 according to the first embodiment.

Before the transmission of the transmission wave Tw, the pulse width control unit 14 forms the H pulse widths T _p (0) to T that form a pulse width distribution that gradually increases or decreases within a set range of the reference pulse width T ₀ or more. T _p (H-1) is set (step ST11). The pulse width distribution may be a linear distribution or a non-linear distribution. For example, when the inequality of T ₀ ≦ T _p (i) <T _p (i + 1) holds for any integer i within the range of 0 to H−2, the pulse width T _p (0) ˜T _p (H−1) form a stepwise increasing pulse width distribution. When the inequality of T _p (i)> T _p (i + 1) ≧ T ₀ is satisfied, the pulse widths T _p (0) to T _p (H−1) have a pulse width distribution that gradually decreases. Form.

As the setting range of the pulse width distribution, for example, a range from the lower limit value T ₀ to the upper limit value 2T ₀ can be used, but the setting range is not limited to this. As the values of the pulse widths T _p (0) to T _p (H-1), set values stored in advance in a memory (not shown) may be used, or according to an arithmetic expression incorporated in advance. It may be calculated. The pulse widths T _p (0) to T _p (H-1) are given to the signal generation circuit 10 and the correlation processing unit 41.

Next, the signal generation circuit 10, the transmission / reception unit 11, the antenna 12, and the reception circuit 13 execute transmission / reception processing (step ST12). Specifically, the signal generation circuit 10 transmits the intra-pulse modulation signals having the pulse widths T _p (0) to T _p (H−1) set by the pulse width control unit 14 to the transmission pulse signal Tx (0, t) to Tx (H-1, t) are continuously generated. When the transmission pulse signals Tx (0, t) to Tx (H-1, t) are input from the signal generation circuit 10 via the transmission / reception unit 11, the antenna 12 receives the transmission pulse signals Tx (0, t) to Tx. The transmitted wave Tw corresponding to (H-1, t) is radiated to the external space. When the antenna 12 receives the reflected wave Rw returned from the target Tgt in the external space, the transmission / reception unit 11 causes the reflected wave signals Rx (0, t) to Rx (H-1, t) corresponding to the reception output of the antenna 12. ) Is output to the receiving circuit 13. The receiving circuit 13 receives the reflected wave signals Rx (0, t) to Rx (H-1, t) as received digital signals (received video signals) V ₀ (0, m) to V ₀ (H-1, m), respectively. Convert to.

In the example of FIG. 5, after the pulse widths T _p (0) to T _p (H-1) are set in step ST11, the transmission / reception process is executed in step ST12. Alternatively, the pulse width control unit 14 may sequentially set the pulse widths T _p (0) to T _p (H-1) in parallel with the transmission / reception processing in step ST12.

After the execution of step ST12, when the received digital signal V ₀ (h, m) is input, the correlation processing unit 41 outputs the reference signal Ex (h, m) to the received digital signal V ₀ (h, m). By executing the correlation processing used, the pulse compression signal F _{V · Ex} (h, m) is generated (step ST13 in FIG. 5). Specifically, the correlation processing unit 41 executes the correlation calculation between the reference signal Ex (h, m) and the received digital signal V ₀ (h, m) to obtain the pulse compression signal F _{V · Ex} (h , M) can be generated. As the reference signal Ex (h, m), a reference signal having a modulation component B ₀ / (2T _p (h)) of the modulation control signal L _chp (h, t) as shown in the following equation (11) is used. It is possible.

In Expression (11), A _E is the amplitude of the reference signal Ex (h, m). As shown in Expression (11), the reference signal Ex (h, m) is a function whose variable is the product mΔt of the sampling interval Δt and the sampling number m. The width (effective data length or effective data range) of the reference signal Ex (h, m) is limited by the pulse width T _p (h). That is, the value of the reference signal Ex (h, m) is zero outside the range of 0 ≦ mΔt ≦ T _p (h).

For example, the correlation processing unit 41 can execute the correlation calculation by executing the convolution calculation in the time domain as shown in the following Expression (12).

Here, M _p (h) is the number of sampling points within the pulse regarding the pulse number (hit number) h. Note that a correlation calculation based on a known convolution calculation in the frequency domain may be executed instead of the correlation calculation shown in Expression (12).

Next, the domain transforming unit 44 performs domain transforming processing from the time domain to the frequency domain in the pulse hit direction on the pulse compression signal F _{V · Ex} (h, m) to _generate the frequency domain signal f _d (h _fft , m) is generated (step ST14 in FIG. 5). As the domain conversion processing, discrete Fourier transform based on a predetermined algorithm such as FFT algorithm may be executed. The discrete Fourier transform is expressed by the following equation (13).

Here, h _fft is the sampling number in the frequency domain, and H is the number of discrete Fourier transform points.

By transforming the equation (13) using the above equation, the following equation (14) is obtained.

Here, A _h is the amplitude of the frequency domain signal f _d (h _fft , m).

By rearranging the equation (14), the following equation (15) can be obtained.

The right side of equation (15) consists of the product of three terms. If the magnitude of the value of the third term in the product of the right side is maximized, high integration efficiency can be obtained in the discrete Fourier transform. The condition that the magnitude of the value of the third term becomes almost maximum is as shown in the following expression (16).

If the sampling number h _fft satisfying the condition of Expression (16) is represented by h _{fft, peak} , the sampling number h _{fft, peak} is expressed as shown in the following Expression (17).

Therefore, high integration efficiency is obtained for the sampling number h _{fft, peak} in the frequency domain. The frequency range based on the pulse repetition period T _pri can be calculated based on the velocity value v _amb in the following equation (18).

By the way, in the present embodiment, the transmission pulse signal Tx (0, 0, 0, 1) is generated using the pulse widths T _p (0) to T _p (H-1) that form a pulse width distribution that gradually increases or decreases within the set range. Since t) to Tx (H-1, t) are continuously generated, the influence of the transmission blind (a state where the reflected wave from the target Tgt cannot be received during the transmission time of each pulse wave) is suppressed. At the same time, it becomes possible to generate the pulse compression signal F _{V · Ex} (h, m) having a high SNR. Therefore, the high SNR frequency domain signal f _d (h _fft , m) can be generated based on the high SNR pulse compression signal F _{V · Ex} (h, m). This point will be described below.

6A and 6B are diagrams schematically showing an example of a transmission wave Tw0 composed of a transmission pulse signal group having the same pulse width and an example of a reflected wave Rw0 corresponding to the transmission wave Tw0. In the example of FIG. 6A, the pulse widths T _tp of the transmission pulse signals for hit numbers 0 to H-1 are all the reference pulse width T ₀ . At this time, as shown in FIG. 6B, it is assumed that the spread width T _rp of the received pulse signal in the reflected wave Rw0 is the width T ₁ that does not cause the reception blind for the hit numbers 0 to H-1.

On the other hand, FIGS. 7A and 7B are diagrams schematically showing an example of a transmission wave Tw1 including transmission pulse signal groups having the same pulse width and an example of a reflected wave Rw1 corresponding to the transmission wave Tw1. is there. In the example of FIG. 7A, the pulse widths T _tp of the transmission pulse signals for hit numbers 0 to H-1 are all twice the reference pulse width T ₀ . At this time, as shown in FIG. 7B, since the spread width T _rp of the received pulse signal in the reflected wave Rw1 exceeds the width T ₁ , it is assumed that the reception blind occurs. In this case, the hatched portion ΔLs1 of the reflected wave Rw1 becomes the loss portion. In the present embodiment, the reason why the occurrence of such a loss portion ΔLs1 can be suppressed will be described below.

8A and 8B show an example of a transmission wave Tw2 including a transmission pulse signal group forming a pulse width distribution that increases stepwise with respect to hit numbers 0 to H-1, and a reflected wave Rw2 corresponding to the transmission wave Tw2. It is a figure which shows an example and roughly. In the example of FIG. 8A, the pulse width distribution is formed such that the pulse width T _tp gradually increases from the lower limit value T ₀ to the upper limit value 2T ₀ . In the example of FIG. 8B, the hatched portion ΔLs2 of the reflected wave Rw2 is the loss portion due to the reception blind.

9A and 9B show an example of a transmission wave Tw3 including a transmission pulse signal group that forms a pulse width distribution that gradually decreases for hit numbers 0 to H-1, and a reflected wave corresponding to the transmission wave Tw3. It is a figure which shows the example of Rw3 roughly. In the example of FIG. 9A, the pulse width distribution is formed such that the pulse width T _tp gradually decreases from the upper limit value 2T ₀ to the lower limit value T ₀ . In the example of FIG. 9B, the hatched portion ΔLs3 of the reflected wave Rw3 is the loss portion due to the reception blind.

Further, FIGS. 10A and 10B correspond to an example of a transmission wave Tw4 including a transmission pulse signal group that forms a pulse width distribution that gradually increases and then decreases for hit numbers 0 to H-1, and the transmission wave Tw4. It is a figure which shows roughly the example of the reflected wave Rw4 which performs. In the example of FIG. 10A, the pulse width distribution has a symmetrical distribution with respect to the hit number h = (H−1) / 2. In the example of FIG. 10B, the hatched portion ΔLs4 of the reflected wave Rw4 is the loss portion due to the reception blind.

When a pulse width distribution that increases or decreases stepwise within the set range of the reference pulse width T ₀ or more is formed in this manner, the correlation function Ex of the above equation (11) is greater than that when the pulse width is constant. The width (effective data length or effective data range) of (h, m) can be expanded. This makes it possible to generate the pulse compression signal F _{V · Ex} (h, m) having a high SNR. Further, since the pulse width distribution is formed so as to increase or decrease in steps, it is possible to suppress the influence of the transmission blind. As shown in FIG. 7A, when all the pulse widths are set to twice the reference pulse width T ₀ , not only the loss due to the transmission blind increases but also the duty ratio between the pulse transmission time and the reception time becomes high. Therefore, it may happen that the cooling period of the circuit for generating the transmission pulse signal cannot be sufficiently secured. When such a situation occurs, there is a problem that it becomes difficult to control the circuit due to heat generation. In the present embodiment, since the pulse width distribution that increases or decreases stepwise is formed, it is possible to sufficiently secure the cooling period of the signal generation circuit 10.

FIG. 11A is a graph schematically showing an example of the power distribution of a pulse compression signal obtained when the pulse width is constant. The power distribution shown in FIG. 11A forms peaks of almost constant height for hit numbers 0 to H-1. On the other hand, FIG. 11B schematically shows an example of the power distribution of the pulse compression signal F _{V · Ex} (h, m) obtained when forming the pulse width distribution that gradually increases as shown in FIG. 8A. It is a graph shown in. Since the power distribution shown in FIG. 11B forms a higher peak as the hit number h increases, the SNR is improved compared to the case of FIG. 11A. FIG. 12 is a graph schematically showing SNR η ₀ obtained from the power distribution shown in FIG. 11A and SNR η ₂ obtained from the power distribution shown in FIG. 11B. In FIG. 12, the horizontal axis represents the hit number and the vertical axis represents the signal-to-noise ratio (SNR). As shown in FIG. 12, the SNR η ₂ increases as the hit number h increases, so that it can be seen that the SNR is improved.

More specifically, for example, the evaluation value eta _out of SNRη _{_0, PC,} _0, it can be expressed by the following equation (19), the evaluation value of _{_{SNRη 2 Η out, PC (h}} ) , the following equation (20 ) Can be represented.

In Expressions (19) and (20), A is the amplitude of the received digital signal (received video signal) V ₀ (h, m), M ₀ is the number of sampling points based on the reference pulse width T ₀ , and M _p (h) Is the number of sampling points based on the pulse width T _p (h), and σ _nis ² is the variance of noise.

Evaluation value eta _out of SNRη _{_0,} a _{PC, 0,} the following relationship (21) is established between the evaluation value of _{_{SNRη 2 Η out, PC (h}} ).

In order to form the pulse width distribution that gradually increases or decreases as described above, the number of sampling points M ₀ based on the reference pulse width T ₀ shown in the following equation (22) and the following equation (23) a pulse width in the hit number h _T p (h) the number of sampling points based on _M p (h) that, it is desirable to maintain the relation of the following equation (24) between the sampling points M of the pulse repetition period _{T pri} ..

In the present embodiment, a pulse width distribution that increases or decreases stepwise is formed, but it is not necessary to change the sampling frequency f _s, and it is not necessary to change the modulation bandwidth B ₀ . It is possible to obtain the radar device 1 with improved target detection performance without significantly modifying the configuration of the radar device.

When forming the pulse width distribution that increases stepwise, the pulse width control unit 14 can set the pulse width T _p (h) as shown in the following Expression (25), for example.

Here, ΔT _p is a pulse width change rate. According to equation (25), a pulse width distribution that gradually increases from the lower limit value T ₀ of the set range is obtained.

Further, when forming a symmetrical pulse width distribution, the pulse width control unit 14 uses a window function such as a triangular window, a Hanning window, or a Hamming window to make the pulse width control symmetrical. The pulse width T _p (h) that forms the pulse width distribution can be set. This makes it possible to reduce the side lobe in the Doppler frequency direction without degrading the resolution while suppressing the loss of the main lobe. The following expression (26) is an expression representing the pulse width T _p (h) when the triangular window is used.

When the window function is represented by w (h), the pulse width T _p (h) is represented by the following equation (27).

When a Hamming window is used as the window function w (h), the window function w (h) is expressed by the following equation (28).

FIG. 13 is a graph schematically showing an example of the power distribution of the pulse compression signal F _{V · Ex} (h, m) obtained when the formula (26) is used. The power distribution shown in FIG. 13 forms a high peak when the hit number h is (H-1) / 2, and forms a symmetrical distribution in the pulse hit direction. Therefore, the SNR is improved as compared with the case of FIG. 11A. FIG. 14 is a graph schematically showing SNR η ₀ obtained from the power distribution shown in FIG. 11A and SNR η ₄ obtained from the power distribution shown in FIG. 13. In FIG. 14, the horizontal axis represents the hit number and the vertical axis represents the signal-to-noise ratio (SNR). As shown in FIG. 14, it can be seen that the SNR is improved.

Further, the side lobe level in the Doppler frequency direction can be suppressed by setting the pulse width T _p (h) using the window function. FIG. 15 is a graph showing an example of the frequency domain signal. In FIG. 15, the horizontal axis represents the Doppler frequency, the vertical axis represents the power, and f _D represents the target Doppler frequency. Further, the solid line represents the power in the case of a pulse width distribution that increases stepwise, the broken line represents the power in the case of a symmetrical pulse width distribution, and the dashed-dotted line represents the power in the case of a constant pulse width. ing. As shown in FIG. 15, in the case of a pulse width distribution that gradually increases, the SNR is improved and the resolution is not deteriorated. Further, in the case of the symmetrical pulse width distribution, the SNR is improved, the resolution is not deteriorated, and the level of the first side lobes appearing on both sides of the main lobe is suppressed.

The pulse width control unit 14 sets the pulse width T _p (h) (in the case of the formula (25) so that the SNR of the frequency domain signal f _d (h _fft , m) exceeds a predetermined required value η _{out, rq.} It is desirable to set the pulse width change rate ΔT _p ). Specifically, for example, the pulse width control unit 14 may set the pulse width T _p (h) so that the SNR of the frequency domain signal f _d (h _fft , m) satisfies the following equation (29). desirable.

Here, H _out is the evaluation value of the SNR of the frequency domain signal f _d (h _fft , m), A is the amplitude of the received digital signal (received video signal) V ₀ (h, m), and σ _nis ² is the variance of noise.

The SNR evaluation value η _{out, 0} when the amplitude A is constant and the number of sampling points M _p (h) is constant (= M ₀ ) is derived from the equation (29) as shown in the following equation (30). It

The pulse width control unit 14 sets the pulse width T _p (h) so as to satisfy the following conditional expression (31), thereby improving the SNR of the frequency domain signal f _d (h _fft , m) to achieve the target. The radar device 1 with improved detection performance can be provided.

As described above, when all the pulse widths are set to twice the reference pulse width T ₀ , not only the loss due to the transmission blind increases but also the pulse transmission time (pulse width) T _p and the reception time (pulse repetition) Since the duty ratio Dty with respect to the cycle) T _pri becomes too high, a situation may occur in which the cooling period of the circuit that generates the transmission pulse signal cannot be sufficiently secured, and it becomes difficult to control the circuit. The duty ratio Dty is given by the following equation (32).

In the present embodiment, since the pulse width distribution is formed so as to increase or decrease in steps, it is possible to reduce the loss due to the transmission blind as represented by the following inequality (33).

Here, L _2T0 indicates a loss when all pulse widths are set to twice the reference pulse width T ₀ , and L _Tp indicates a loss when a pulse width distribution that gradually increases or decreases is formed. It shows the loss.

After the generation of the frequency domain signal f _d (h _fft , m) (step ST14 of FIG. 5), the target candidate detection unit 51 determines the frequency domain signal f _d (h _fft , m) based on the signal strength of the frequency domain signal f _d (h _fft , m). A target candidate is detected (step ST15). Specifically, for example, the target candidate detection unit 51 may detect the target candidate using a known CA-CFAR (Cell Average-Constant False Alarm Rate) process. For example, in the CA-CFAR processing, since the maximum detection probability can be obtained so that the false alarm probability Pfa becomes a constant value, the false detection can be controlled, and the noise in the frequency domain signal can be minimized. f _d _{(h fft,} m) can be detected target candidates based on the signal strength.

The target candidate detecting unit 51 determines the target candidate number ntg assigned to the detected single or plural target candidates, the sampling number m = m _ntg corresponding to the target candidate number ntg, and the frequency region corresponding to the target candidate number ntg. it is possible to output a sampling number _{_h fft =} _h _fft, and _ntg target candidate information calculation section 52. For convenience of explanation, it is assumed that the target candidate number ntg is an integer within the range of 1 to Ntg.

Next, the target candidate information calculation unit 52 calculates target information such as a relative distance and a relative speed regarding the target candidate, and outputs data indicating the target information to the display device 60 (step ST16 in FIG. 5). Specifically, for example, the target candidate information calculation unit 52 calculates the relative distance R _{0, ntg} of the ntg-th target candidate based on the target candidate number ntg and the sampling number m _ntg according to the following equation (34). be able to.

Further, the target candidate information calculation unit 52 can calculate the relative speed V _{0, ntg} of the ntg-th target candidate according to the following equation (35).

In Expression (35), Δv _fft is a relative velocity sampling interval as shown in the following Expression (36).

The target candidate information calculation unit 52 can output a combination of the target candidate number ntg, the relative distance R _{0, ntg,} and the relative speed V _{0, ntg to} the display 60 as target information. The display device 60 can display the target information on the screen.

As described above, in the first embodiment, the signal generation circuit 10 forms a plurality of pulse widths T _p (0) that form a pulse width distribution that gradually increases or decreases within the set range of the reference pulse width T ₀ or more. ) To T _p (H-1) are used to continuously generate intra-pulse modulated transmission pulse signals Tx (0, t) to Tx (H-1, t), the correlation processing unit 41 Can generate the high SNR pulse compression signal F _{V · Ex} (h, m) while suppressing the influence of the transmission blind. As a result, the domain conversion unit 44 can generate the high SNR frequency domain signal f _d (h _fft , m). Therefore, it is possible to provide the radar device 1 with improved target detection performance.

Moreover, since a pulse width distribution that increases or decreases in stages is formed, a sufficient cooling period for the signal generation circuit 10 can be secured. As a result, the stability of the operation of the signal generation circuit 10 can be realized, and the radar device 1 that maintains heat resistance can be provided.

Further, since the pulse width control unit 14 can set the pulse width T _p (h) that forms a symmetrical pulse width distribution in the pulse hit direction by using the window function, the loss of the main lobe is suppressed. At the same time, the side lobe can be suppressed while maintaining the resolution in the Doppler frequency direction. As a result, the separation performance from a target having a low electric power that is close to the target is improved, so that it is possible to provide the radar device 1 having an improved target separation performance. The window function may be selected or designed to form a desired sidelobe level distribution. By using such a window function and setting the pulse width T _p (h) that forms a symmetrical pulse width distribution in the pulse hit direction, it is possible to effectively suppress the side lobe level. It should be noted that a pulse width T _p (h) that forms a symmetrical pulse width distribution in the pulse hit direction is set by using a means other than the window function (for example, a filter function) to form a desired side lobe level distribution. By doing so, the side lobe level may be suppressed.

Embodiment 2.
Next, a second embodiment according to the present invention will be described. FIG. 16 is a block diagram showing a schematic configuration of the radar device 2 according to the second embodiment of the present invention. As shown in FIG. 16, the radar device 2 of this embodiment includes a signal generation circuit 10, a transmission / reception unit 11, a reception circuit 13, a radar signal processing circuit 32, and a display 60. The radar signal processing circuit 32 of the present embodiment has the configuration of the radar signal processing circuit 31 of the first embodiment except that the correlation processing unit 41 of the first embodiment is replaced by the correlation processing unit 42 of FIG. The configuration is the same as that of.

When the received digital signal V ₀ (h, m) is input, the correlation processing unit 42 of the present embodiment receives the reference signal Exa (h, m) and the pulse with respect to the received digital signal V ₀ (h, m). The pulse compression signal F _{V · Ex} (h, m) is generated by performing the correlation processing using the width T _p (h). The processing content of the correlation processing unit 42 is the same as the processing content of the correlation processing unit 41 of the first embodiment, except that the reference signal used for the correlation processing is different. The correlation process can be executed by a convolution operation in the time domain or a convolution operation in the frequency domain. Specifically, the correlation processing unit 42 can use the reference signal Exa (h, m) as shown in the following Expression (37).

Here, m _b is a parameter corresponding to a distance bin that is not affected by the transmission blind. The reference signal Exa (h, m) gives a valid value when the two conditions of “0 ≦ mΔt ≦ T _p (h)” and “m _b + m <M” are simultaneously satisfied, and otherwise (“ A zero value is given to "otherwise"). Therefore, the width (effective data length or the effective data range) of the reference signal Ex (h, m) is limited by the pulse width _T p (h) and the parameter _{m b.}

FIG. 17A shows a frequency domain signal Fx (h, m _f ) (m _f = 0 to M−1) obtained by Fourier transforming the reference signal Ex (h, m) of the first embodiment with respect to the sampling number m. FIG. 17B is a graph, and FIG. 17B is a frequency domain signal Fxa (h, m _f ) (m _f = 0 to M−M obtained by Fourier transforming the reference signal Exa (h, m) of the second embodiment with respect to the sampling number m. It is a graph showing 1). In FIGS. 17A and 17B, the axis labeled “First Time Frequency” corresponds to the sampling number m _f in the frequency domain.

The width (effective data length or effective data range) of the reference signal Ex (h, m) in the first embodiment corresponds to the band range Ru indicated by hatching as shown in FIG. 17A. In the correlation processing in the first embodiment, frequency domain signal components fx (0), ..., fx (h), ..., fx (H-1) are used. On the other hand, in the width (effective data length or effective data range) of the reference signal Exa (h, m) in the second embodiment, as shown in FIG. 17B, the band range Rn indicated by hatching is excluded. In addition, it corresponds to the band range Ru indicated by hatching. In the correlation processing in the second embodiment, frequency domain signal components fy (0), ..., Fy (h), ..., Fy (H-1) are used. The band range Rn is a range in which only noise due to the influence of the transmission blind exists. The band range Rn can be excluded by adjusting the parameter m _b .

FIG. 18 is a graph showing the relationship between the signal-to-noise ratio (SNR) of the frequency domain signal f _d (h _fft , m) and the distance. In FIG. 18, R _{p, 0} represents the size of the distance affected by the transmission blind due to the formation of the pulse width distribution that increases in stages. Further, H _out is an evaluation value of SNR, η _{out, rq} is a required value, R _pri is a distance corresponding to a pulse repetition period, and R _na is a maximum value of a pulse width that is not affected by the transmission blind. Further, the solid line shows the SNR obtained in the case of the second embodiment, the broken line shows the SNR obtained in the case of the first embodiment (incrementally increasing pulse width distribution), and the dashed-dotted line shows the constant pulse width. The SNRs obtained in each case are shown. It can be seen that a better SNR can be obtained in the case of the second embodiment than in the case of the first embodiment.

As described above, in the second embodiment, the width (effective data length or effective data range) of the reference signal Exa (h, m) used for the correlation processing depends on the pulse width T _p (h) and the parameter _mb . Since it is limited, it is possible to avoid SNR deterioration due to the influence of the transmission blind. Even if the pulse width distribution that increases or decreases stepwise as described above is formed, it is possible to provide the radar device 2 with good target detection performance.

The hardware configuration of all or part of the pulse width control unit 14 and the radar signal processing circuit 32 according to the second embodiment may be realized by an LSI such as an ASIC or FPGA. Similar to the case of the first embodiment, the hardware configuration of all or part of the pulse width control unit 14 and the radar signal processing circuit 32 of the second embodiment is realized by the signal processing circuit 70 shown in FIG. Good.

Although the first and second embodiments according to the present invention have been described above with reference to the drawings, the first and second embodiments are examples of the present invention, and various embodiments other than the first and second embodiments are described. It is possible. Within the scope of the present invention, it is possible to freely combine the first and second embodiments, modify any constituent element of each embodiment, or omit any constituent element of each embodiment.

The radar device and the signal processing method according to the present invention can be used in a radar system that detects a relative position and a relative speed of a target. Further, the radar device according to the present invention can be used in a state of being installed on the ground or in a state of being mounted on a moving body such as an aircraft, an artificial satellite, a vehicle or a ship.

1, 2 radar device, 10 signal generation circuit, 11 transmission / reception unit, 12 antenna, 13 reception circuit, 14 pulse width control unit, 20 local oscillator, 21 pulse generator, 22 pulse modulator, 23 output unit, 24 down converter , 25 bandpass filter, 26 amplifier, 27 phase detector, 28 A / D converter, 31, 32 radar signal processing circuit, 41, 42 correlation processing unit, 44 area conversion unit, 50 target detection unit, 51 target candidate detection unit , 52 target candidate information calculation unit, 60 display device, 70 signal processing circuit, 71 processor, 72 memory, 73 storage device, 74 input / output interface, 75 signal path, Tgt target, Tw transmitted wave, Rw reflected wave.

Claims

A pulse width control unit that sets a plurality of pulse widths that form a pulse width distribution that gradually increases or decreases within a setting range equal to or greater than a predetermined reference pulse width,
A signal generation circuit for continuously generating a plurality of transmission pulse signals each having the plurality of pulse widths and respectively subjected to intra-pulse modulation;
A transmitting / receiving unit that transmits the plurality of transmission pulse signals to an external space and receives a plurality of reflected wave signals respectively corresponding to the plurality of transmission pulse signals from the external space,
A receiving circuit that generates a plurality of received signals respectively corresponding to the plurality of transmission pulse signals by sampling each of the plurality of reflected wave signals,
A correlation processing unit that generates a plurality of pulse compression signals by performing a correlation process using a reference signal for each of the plurality of received signals,
A domain conversion unit that generates a plurality of frequency domain signals by performing a domain conversion process from the time domain to the frequency domain on the plurality of pulse compression signals,
A radar apparatus comprising: a target detection unit that detects a target candidate based on the plurality of frequency domain signals.
The radar device according to claim 1, wherein the pulse width distribution is a distribution that gradually increases from a lower limit value of the setting range to an upper limit value of the setting range.
The radar device according to claim 1, wherein the pulse width distribution is a distribution that gradually decreases from an upper limit value of the setting range to a lower limit value of the setting range.
The radar device according to claim 1, wherein the pulse width distribution has a symmetrical distribution in a pulse hit direction.
The radar device according to claim 4, wherein the pulse width distribution is calculated using a window function having a symmetrical distribution.
The radar device according to any one of claims 1 to 5, wherein the correlation processing unit limits the effective data range of the reference signal by the pulse width and executes the correlation processing. A radar device characterized by the above.
The radar device according to any one of claims 1 to 5, wherein the correlation processing unit limits an effective data range of the reference signal by a parameter corresponding to a distance bin and the pulse width. And performing the correlation process.
The radar device according to any one of claims 1 to 7, wherein the intra-pulse modulation is frequency modulation.
The radar device according to any one of claims 1 to 8, wherein the area conversion unit executes a discrete Fourier transform based on a predetermined algorithm as the area conversion processing. Radar device.
The radar device according to claim 9, wherein the predetermined algorithm is a fast Fourier transform algorithm.
A signal generation circuit that continuously generates a plurality of transmission pulse signals and a plurality of transmission pulse signals that are sent to an external space and that receives a plurality of reflected wave signals that correspond to the plurality of transmission pulse signals, respectively. Signal processing performed by a radar device including a transmitting / receiving unit and a receiving circuit that generates a plurality of received signals respectively corresponding to the plurality of transmission pulse signals by sampling each of the plurality of reflected wave signals. Method,
A step of setting a plurality of pulse widths that form a pulse width distribution that increases or decreases stepwise with time within a preset range of a predetermined reference pulse width or more, respectively, as the pulse width of the plurality of transmission pulse signals,
Generating a plurality of pulse-compressed signals by performing a correlation process using a reference signal for each of the plurality of received signals,
Generating a plurality of frequency domain signals by performing a domain conversion process from the time domain to the frequency domain on the plurality of pulse compression signals,
A step of detecting a target candidate based on the plurality of frequency domain signals.