WO2020093463A1

WO2020093463A1 - Megawatt-level direct-drive permanent magnet electric drive system for electric locomotive

Info

Publication number: WO2020093463A1
Application number: PCT/CN2018/116996
Authority: WO
Inventors: 王彬; 詹哲军; 张瑞峰; 张巧娟; 张吉斌; 梁海刚; 牛剑博; 杨高兴; 路瑶; 苏鹏程
Original assignee: 中车永济电机有限公司
Priority date: 2018-11-08
Filing date: 2018-11-22
Publication date: 2020-05-14
Also published as: CN111162715B; CN111162715A

Abstract

Provided is a megawatt-level direct-drive permanent magnet electric drive system for electric locomotive, comprising: a first precharging module, a second precharging module, a first four-quadrant rectifier, a second four-quadrant rectifier, a first chopping module, a second chopping module, an intermediate DC circuit, a first inverter module, a second inverter module, a third inverter module and an auxiliary converter. The system can control the direct-drive permanent magnet synchronous motor in electric locomotive using high-power direct-drive permanent magnet synchronous motor, filling the gap in the application of high-power direct-drive permanent magnet synchronous motor in electric locomotive.

Description

Megawatt direct drive permanent magnet electric drive system for electric locomotive

Technical field

The invention relates to the technical field of train control, in particular to a megawatt direct drive permanent magnet electric drive system for electric locomotives.

Background technique

The traction converter of the electric locomotive is an important part of the electric locomotive. It is used to convert the electric energy of the traction power supply network into the electric energy supplied to the traction motor to achieve the purpose of controlling the speed of the traction motor and manipulating the speed of the locomotive. The design of the main circuit of the traction converter is one of the main factors of the working performance of the traction converter, which directly affects the weight, size, efficiency and related technical and economic indicators of the electric locomotive.

The electric locomotive in the prior art generally adopts the driving mode of AC asynchronous motor plus gear box. In order to improve the efficiency of the electric locomotive and reduce the loss, the present invention uses a high-power direct-drive permanent magnet synchronous motor to be applied to the electric locomotive. On the one hand, the high-power direct-drive permanent magnet synchronous motor makes full use of the advantages of high efficiency, low loss, high power density and large starting torque of the permanent magnet synchronous motor. On the one hand, the gear box is removed, and the permanent magnet synchronous motor is directly driven. Combined with the locomotive wheel pair, it reduces the quality and the loss caused by the gear box, and further improves the overall efficiency of the electric locomotive.

The current traction converters and existing control methods in electric locomotives are not designed for high-power direct-drive permanent magnet synchronous motors, so there is no electric drive system that can be directly applied to use high-power direct-drive permanent magnet synchronous motors. Electric motor in electric locomotive. How to design a megawatt direct-drive permanent magnet electric drive system for electric locomotives in electric locomotives using high-power direct-drive permanent magnet synchronous motors is a technical problem that needs to be solved urgently.

Summary of the invention

The invention provides a megawatt direct drive permanent magnet electric drive system for electric locomotives, which controls the high power direct drive permanent magnet synchronous motor in the electric locomotive using a high power direct drive permanent magnet synchronous motor, filling the high power direct The application of permanent magnet synchronous motors in electric locomotives is blank.

The invention provides a megawatt direct drive permanent magnet electric drive system for electric locomotives, which is used to control an electric locomotive using a high-power direct drive permanent magnet synchronous motor. The electric locomotive includes three high-power direct drive permanent magnet synchronous motors; The megawatt direct drive permanent magnet electric drive system for electric locomotive includes: a first precharge module, a second precharge module, a first four-quadrant rectifier, a second four-quadrant rectifier, a first chopper module, a second chopper Wave module, intermediate DC link, first inverter module, second inverter module, third inverter module and auxiliary converter, the first four-quadrant rectifier and the second four-quadrant rectifier pass through the first precharge module and the first The two pre-charging modules are connected to the main transformer of the electric locomotive. The first four-quadrant rectifier and the second four-quadrant rectifier are respectively connected to the intermediate DC circuit through the first chopper module and the second chopper module, and the intermediate DC circuit is connected to the first inverter module , The second inverter module, the third inverter module and the auxiliary converter;

The first pre-charging module includes a first charging capacitor, a first pre-charging contactor and a first main working contactor, and the second precharging module includes a second charging capacitor, a second precharging contactor and a second main working contact Converter, the first four-quadrant rectifier and the second four-quadrant rectifier each include eight switch tubes, the first chopper module includes the first switch tube, the first current sensor, the first reverse diode and the first chopper resistor, the second The chopping module includes a second switch tube, a second current sensor, a second reverse diode and a second chopping resistor. The intermediate DC loop includes a first DC-side support capacitor, a second DC-side support capacitor, and a slow discharge The resistance, DC bus voltage sensor and ground detection module, the first inverter module, the second inverter module and the third inverter module all include a three-phase inverter circuit composed of six switch tubes;

The megawatt direct drive permanent magnet electric drive system for electric locomotives is used to: transmit the AC power of the main transformer to the first four-quadrant rectifier and the second four-quadrant rectifier through the first precharge module and the second precharge module, respectively;

After the first four-quadrant rectifier and the second four-quadrant rectifier respectively convert the alternating current transmitted by the first pre-charging module and the second pre-charging module into direct current, then output to the first chopper module and the second chopper module;

The DC power is chopped through the first chopping module and the second chopping module and then transmitted to the intermediate DC circuit;

Output the received DC power to the first inverter module, the second inverter module, the third inverter module and the auxiliary converter through the intermediate DC loop;

Convert the received DC power into three-phase AC power through the first inverter module, the second inverter module and the third inverter module, and then output to three high-power direct-drive permanent magnet synchronous motors respectively;

The auxiliary DC converter converts the received DC power into three-phase AC power and outputs it to the auxiliary load of the electric locomotive.

Optionally, in an embodiment of the present invention, the first four-quadrant rectifier and the second four-quadrant rectifier convert the alternating current of the main transformer into direct current and output to the intermediate direct current loop, including :

Sampling the AC current input to the four-quadrant rectifier to obtain the AC current within the sampling period. The AC current includes a positive half-cycle current value and a negative half-cycle current value; wherein, according to the preset sampling frequency, the input The AC current of the four-quadrant rectifier is sampled to obtain multiple sampling points, and the obtained multiple sampling points are plotted as a curve to obtain a sine or cosine curve; the preset sampling frequency is N times the IGBT on-off frequency. N≥2;

Acquiring a first sum value of the current value of the positive half cycle and a second sum value of the current value of the negative half cycle, and obtaining a current offset value according to the first sum value and the second sum value; wherein, The values of the multiple sampling points in the positive half cycle are added to obtain the first sum value P, and then the values of the multiple sampling points in the negative half cycle are added to obtain the second sum value N, the absolute value of the P value and the N value Carry out the difference calculation, the difference is Q;

Input the first difference between the current offset value and zero to the first PI controller to obtain the first output value output by the first PI controller; wherein, the DC offset value Q and zero are input to the first The PI controller, the first PI controller constitutes a control deviation according to the DC offset value Q and zero, and linearly combines the proportionality and integral of the deviation to form a control amount, controls the AC current, and eliminates the DC offset of the AC current. The control quantity is the first output value;

A pulse width modulation symbol is obtained according to the first output value and the second output value output by the PR controller, and the PR controller is used to perform static-free control of the alternating current so that the period and phase of the alternating current are The grid voltage is the same; where the AC current is input to the PR controller to ensure that the phase and cycle of the AC current are the same as the grid voltage, a stable output AC current is obtained, which is the second output value;

The on-off of the insulated gate bipolar transistor IGBT in the four-quadrant rectifier is controlled according to the pulse width modulation sign.

In a possible design, before sampling the alternating current of the input four-quadrant rectifier to obtain the alternating current in the sampling period, the method further includes:

Obtain the second difference between the DC bus voltage of the four-quadrant rectifier and the command voltage;

Input the second difference to the second PI controller, so that the third output value output by the second PI controller is multiplied by the output value of the phase-locked loop to obtain an AC current in the same phase as the grid voltage, the The phase-locked loop is used to control the period and phase of the alternating current and the period and phase of the grid voltage to be consistent.

In a possible design, the AC current input to the four-quadrant rectifier is sampled to obtain the AC current within the sampling period, including:

Sampling the alternating current input to the four-quadrant rectifier according to a preset sampling frequency to obtain a sampling current, and the preset sampling frequency is twice the on-off frequency of the IGBT;

According to the grid voltage phase determined by the phase-locked loop and the sampling current, the alternating current in the sampling period is obtained.

In a possible design, before obtaining the AC current in the sampling period according to the grid voltage phase determined by the phase-locked loop and the sampling current, the method further includes:

Filtering the sampling current through a first band-pass filter and a second band-pass filter to obtain a filtered sampling current; wherein, the first band-pass filter is used to obtain a main frequency signal of an alternating current, the The second band-pass filter is used to filter out interference harmonics.

In a possible design, before inputting the first difference between the current bias value and zero to the first PI controller and obtaining the first output value output by the first PI controller, the method further include:

Determine whether the absolute value of the first difference is greater than the absolute value of the current loop width, and the obtained judgment result is yes.

In a possible design, obtaining the pulse width modulation symbol according to the first output value and the second output value output by the PR controller includes:

Summing the first output value and the second output value to obtain a third sum value, the first output value is a current variable, and the second output value is a current value;

The pulse width modulation symbol is obtained according to the third sum value and the unipolar frequency doubling pulse modulation method.

In an embodiment of the present invention, the megawatt direct drive permanent magnet electric drive system for electric locomotives further includes: a first chopping module and a second chopping module, the first chopping module is connected to the first A four-quadrant rectifier and the intermediate DC circuit, and the second chopper module connects the second four-quadrant rectifier and the intermediate DC circuit;

The megawatt direct drive permanent magnet electric drive system for electric locomotives is also used for:

Controlling the first chopping module and the second chopping module to perform chopping processing on the DC power output by the first four-quadrant rectifier and the second four-quadrant rectifier, respectively, and then output to the intermediate DC loop;

Specifically, for any one of the first chopping module and the second chopping module, the control method further includes:

Periodic detection of the intermediate DC bus voltage, the intermediate DC bus voltage being the voltage on the DC bus on the electric locomotive;

When the detected intermediate DC bus voltage value is greater than the upper chopping threshold, the P regulator is used to adjust the intermediate DC bus voltage until the detected intermediate DC bus voltage value is less than the lower chopping threshold, the chopping The upper limit of the wave threshold is greater than the lower limit of the chopping threshold; wherein, the principle of the P regulator is to control the chopper tube to be turned on within a certain proportion of the detection cycle.

Optionally, the adjusting the intermediate DC bus voltage with a P regulator includes:

Using the P regulator to determine the chopping duty cycle within the target detection period; the target detection period includes: from the detected intermediate DC bus voltage value greater than the upper chopping threshold, to the detected intermediate DC bus voltage value Less than the chopping lower threshold between the detected detection period;

According to the chopping duty ratio, determine the opening time of the chopper tube within the target detection period;

According to the turn-on time, the turn-on or turn-off of the chopper tube is controlled so that the voltage value of the intermediate DC bus drops to less than the lower threshold of the chopper.

Optionally, the above method further includes:

When it is detected that the voltage value of the intermediate DC bus is lower than the lower chopping threshold, the chopper tube is controlled to be turned off.

Optionally, before using the P regulator to determine the chopping duty cycle within the target detection period, the method further includes:

Determine the target parameters according to the following formula;

Err = U1-chopping lower threshold

Among them, Err represents the target parameter, U1 represents the intermediate DC bus voltage value detected in the target detection period;

Correspondingly, the use of the P regulator to determine the chopping duty cycle within the target detection period includes:

Obtain the control coefficient corresponding to the P regulator;

According to the control coefficient and the target parameter, the chopping duty ratio is determined.

Optionally, the acquiring the control coefficient of the P regulator includes:

Determine the control coefficient according to the following formula;

Kp_chp = 1 / (DC bus voltage overvoltage protection threshold-chopping lower threshold)

Among them, Kp_chp represents the control coefficient.

Optionally, the determining the chopper duty cycle according to the control coefficient and the target parameter includes:

Determine the chopping duty cycle according to the following formula;

C_duty = Err * Kp_chp

Among them, C_duty represents the chopping duty ratio, Err represents the target parameter, and Kp_chp represents the control coefficient.

Optionally, before determining the opening time of the chopper tube within the target detection period according to the chopping duty cycle, the method further includes:

Perform error prevention processing on the chopping duty ratio.

Optionally, the error prevention processing of the chopping duty cycle includes:

If the value of the chopping duty ratio is greater than 1, the value of the chopping duty ratio is set to 1;

If the value of the chopping duty ratio is less than 0, the value of the chopping duty ratio is set to 0.

Optionally, in a possible implementation manner of this embodiment, the control method further includes:

Determine the speed of the high-power direct-drive permanent magnet synchronous motor to be controlled;

Determining a first control strategy according to the rotation speed and the first mapping relationship, where the first mapping relationship includes a one-to-one correspondence between at least one rotation speed range and at least one control strategy;

The expected control phase angle of the high-power direct-drive permanent magnet synchronous motor to be controlled is determined according to the first control strategy.

Optionally, the first mapping relationship includes:

Correspondence between speed below rated speed and MTPA control strategy;

Correspondence between the speed above the rated speed and the field weakening control strategy.

Optionally, the MTPA control strategy includes: determining the q-axis current reference and the d-axis current reference according to the torque current curve;

Calculating a first difference between the q-axis current reference and the q-axis actual current and a second difference between the d-axis current reference and the d-axis actual current;

The d-axis voltage reference is obtained according to the first difference value through the first PI controller, and the q-axis voltage reference is obtained based on the second difference value through the second PI controller;

Calculating the sum of the q-axis voltage reference and the q-axis feedforward voltage to obtain the actual q-axis voltage reference, and calculating the sum of the d-axis voltage reference and the d-axis feedforward voltage to obtain the actual d-axis voltage reference; wherein, The feedforward voltage can be calculated by the following feedforward decoupled closed-loop transfer function matrix:

The closed-loop transfer function of the feedforward decoupling is obtained by the following voltage calculation equation of feedforward decoupling:

Optionally, the field weakening control strategy includes: calculating, by the PI controller, the amount of d-axis current change in a given field weakening state according to the difference between the voltage limit value and the feedforward voltage amplitude;

The d-axis current reference after the field-weakening adjustment is obtained by giving the sum of the d-axis current change and the d-axis current under the given field weakening state;

Calculate the q-axis current reference after field weakening adjustment according to the d-axis current reference and the torque formula;

The PI controller obtains the work angle β according to the difference between the q-axis current setting and the q-axis actual current;

Calculate the actual q-axis voltage reference and the actual d-axis voltage reference by the following formula;

U _d = U _s cos β

U _q = U _s cos β

Among them, Us is the voltage limit value, Ud is the actual d-axis voltage given, and Uq is the actual q-axis voltage given.

Optionally, in an embodiment of this embodiment, the method further includes:

When the control strategy is switched from the MTPA control strategy to the field weakening control strategy, the voltage vector angle in the MTPA control strategy at the moment of switching is used as the initial power angle β in the field weakening control strategy;

When the control strategy is switched from the field weakening control strategy to the MTPA control strategy, the last beat power angle β in the instantaneous field weakening control strategy is passed through the formula by switching

Calculate the actual q-axis voltage setting and actual d-axis voltage setting in the MTPA control strategy.

Obtain the frequency of the modulated wave of the high-power direct-drive permanent magnet synchronous motor to be modulated;

Determining a first modulation strategy according to the range of the frequency of the modulated wave and the second mapping relationship, where the second mapping relationship includes a one-to-one correspondence between the frequency range of at least one modulated wave and at least one modulation strategy;

The PWM carrier frequency of the high-power direct-drive permanent magnet synchronous motor is determined according to the first modulation strategy.

Optionally, the second mapping relationship includes:

When the frequency of the modulated wave is in the low-speed stage, it corresponds to the asynchronous modulation strategy;

When the frequency of the modulation wave is greater than the low-speed stage and lower than the high-speed stage, it corresponds to the middle 60-degree synchronous modulation strategy;

The frequency of the modulated wave corresponds to the square wave modulation strategy at the high-speed stage.

Optionally, in a possible implementation manner of this embodiment, the method further includes:

Inject high-frequency voltage signal into the stator winding of the permanent magnet synchronous motor to be detected to obtain three-phase stator winding current;

Obtaining the d-axis target current and the q-axis target current in the expected two-phase synchronous rotating coordinate system according to the three-phase stator winding current;

The initial position angle of the rotor is obtained according to the d-axis target current and the q-axis target current, where the initial position angle is the initial position angle after compensation according to the magnetic pole polarity of the permanent magnet synchronous motor.

Further, the acquiring the initial position angle of the rotor according to the d-axis target current and the q-axis target current includes:

Acquiring the first initial position angle of the rotor according to the q-axis target current;

Obtain the rotor pole compensation angle according to the d-axis target current;

Obtain the initial position angle of the rotor according to the first initial position angle and the magnetic pole compensation angle.

Further, the obtaining the first initial position angle of the rotor according to the q-axis target current includes:

Performing low-pass filtering on the q-axis target current to obtain an error input signal;

Obtain the first initial position angle according to the error input signal.

Further, the performing low-pass filtering on the q-axis target current to obtain the error input signal includes:

Modulating the q-axis target current with a modulation signal to obtain the modulated q-axis target current;

Performing low-pass filtering on the modulated q-axis target current to obtain the error input signal.

Further, the acquiring the first initial position angle according to the error input signal includes:

Obtaining the proportional deviation and the integral deviation of the error input signal according to the input error signal;

Obtain the first initial position angle according to the linear combination of the proportional deviation and the integral deviation.

Further, the obtaining the magnetic pole compensation angle of the rotor according to the d-axis target current includes:

Injecting a plurality of voltage pulse signals with equal voltage amplitudes and different angles into the permanent magnet synchronous motor to obtain the response current of each voltage pulse signal;

Based on the plurality of response currents, the pole compensation angle of the rotor is determined.

Further, the determining the pole compensation angle of the rotor according to the plurality of response currents includes:

When the difference between the angle of the injected voltage pulse signal and the first initial position angle satisfies a preset error range, and the amplitude of the response current of the voltage pulse signal is greater than the first value, the rotor pole compensation is determined The angle is 0, and the first value is the maximum value of the amplitudes of the multiple response currents;

When the difference between the angle of the injected voltage pulse signal and the first initial position angle satisfies a preset error range, and the magnitude of the response current of the voltage pulse signal is less than the second value, the rotor pole compensation is determined The angle is π, and the second value is the minimum value of the amplitudes of the multiple response currents.

Optionally, in an embodiment of the application, the high-frequency voltage signal is:

Where U _mh is the amplitude of the high-frequency voltage signal, ω _h is the angular frequency of the high-frequency voltage signal, and t is the time to inject the high-frequency voltage signal;

The d-axis target current and the q-axis target current in the expected two-phase synchronous rotating coordinate system are obtained according to the three-phase stator winding current, and are calculated by the following formula:

Where L is the average inductance L = (L _d + L _q ) / 2, and △ L is the half-differential inductance △ L = (L _d -L _q ) / 2;

The low-pass filtering process is performed on the q-axis target current to obtain an error input signal, which is calculated by the following formula:

Among them, LPF stands for low-pass filtering; when the rotor position estimation error is small enough, the error input signal after the limit equivalent linearization is:

The first initial position angle is obtained and calculated by the following formula:

Among them, s represents Laplace operator, k _p is the coefficient of proportional term, and k _i is the coefficient of integral term.

Optionally, the megawatt direct drive permanent magnet electric drive system for electric locomotives provided in this embodiment further includes: according to a control interruption period, a modulated carrier period, and the current angular velocity of the rotor of the high power direct drive permanent magnet synchronous motor Obtaining the compensation phase angle of the rotor of the high-power direct-drive permanent magnet synchronous motor;

Obtain the current actual control phase angle according to the compensation phase angle;

Obtain the current expected control phase angle according to the current d-axis voltage given value and the current q-axis voltage given value;

According to the proportional deviation and integral deviation of the current expected control phase angle and the current actual control phase angle, the current actual control phase angle is corrected online.

Further, the obtaining the compensated phase angle of the rotor of the high-power direct-drive permanent magnet synchronous motor according to the control interruption period, the modulation carrier period, and the current angular velocity of the rotor of the high-power direct-drive permanent magnet synchronous motor includes: :

Acquiring the first sub-compensated phase angle according to the control interruption period and the current angular velocity of the rotor of the high-power direct-drive permanent magnet synchronous motor;

Acquiring the second sub-compensated phase angle according to the modulated carrier cycle and the current angular velocity of the rotor of the high-power direct-drive permanent magnet synchronous motor;

Obtaining the third sub-compensated phase angle according to the current angular velocity of the rotor of the high-power direct-drive permanent magnet synchronous motor;

According to the first sub-compensation phase angle, the second sub-compensation phase angle, and the third sub-compensation phase angle, the compensation phase angle of the high-power direct-drive permanent magnet synchronous motor is obtained.

Further, the acquiring the first sub-compensated phase angle according to the control interruption period and the current angular velocity of the rotor of the high-power direct-drive permanent magnet synchronous motor includes:

Obtaining the first phase angle delay corresponding to the first sub-compensated phase angle according to the control interruption period;

The first sub-compensated phase angle is obtained according to the first phase angle delay and the current angular velocity of the rotor of the high-power direct-drive permanent magnet synchronous motor.

Further, the obtaining the second sub-compensated phase angle according to the modulated carrier period and the current angular velocity of the rotor of the high-power direct-drive permanent magnet synchronous motor includes:

Obtaining the second phase angle delay corresponding to the modulation output according to the modulation carrier period;

According to the modulation interruption period of the modulation algorithm, obtain the third phase angle delay corresponding to the modulation calculation;

The second sub-compensated phase angle is obtained according to the second phase angle delay, the third phase angle delay, and the current angular velocity of the rotor of the high-power direct-drive permanent magnet synchronous motor.

Further, before acquiring the third sub-compensated phase angle according to the current angular velocity of the rotor of the high-power direct-drive permanent magnet synchronous motor, the method further includes:

Obtaining the stable operating angular velocity range of the high-power direct-drive permanent magnet synchronous motor according to the vector control strategy of the high-power direct-drive permanent magnet synchronous motor;

According to the given value of the d-axis current and the given value of the q-axis current, a plurality of first d-axis currents, a plurality of first q-axis currents, and each of the first d-axis currents corresponding to the stable operating angular velocity range are acquired D-axis voltage and the q-axis voltage corresponding to each of the first q-axis currents.

Further, the obtaining the third sub-compensated phase angle according to the current angular velocity of the rotor of the high-power direct-drive permanent magnet synchronous motor includes:

Acquiring the transmission error phase angle corresponding to each first angular velocity according to the d-axis voltage corresponding to each of the first d-axis currents and the q-axis voltage corresponding to each of the first q-axis currents;

The third sub-compensated phase angle is obtained according to the transmission error phase angle corresponding to each of the first angular speeds, the current angular speed of the rotor of the high-power direct-drive permanent magnet synchronous motor, and the initial position phase angle of the rotor.

Further, the obtaining the current actual control phase angle according to the compensated phase angle includes:

Obtain the current position phase angle of the rotor of the high-power direct-drive permanent magnet synchronous motor;

Acquiring the actual position phase angle of the rotor according to the current position phase angle, the initial position phase angle of the rotor, and the compensation phase angle;

The current actual control phase angle is obtained according to the actual position phase angle and the modulation phase angle of the rotor, wherein the modulation phase angle is calculated by a modulation algorithm according to the given value of the d-axis voltage and the given value of the current q-axis voltage.

Further, the online correction of the current actual control phase angle according to the proportional deviation and integral deviation of the current expected control phase angle and the current actual control phase angle includes:

Obtaining the proportional deviation and the integral deviation according to the current expected control phase angle and the current actual control phase angle;

According to the linear combination of the proportional deviation and the integral deviation, obtain the correction term of the current actual control phase angle;

Perform online correction on the current actual control phase angle according to the correction item.

Optionally, in an embodiment of the present application, the acquiring the first sub-compensation phase angle is calculated by the following formula:

θ _cmps1 = Δ _t1 · ω

Where, [omega] is the angular velocity of the current of the direct-drive permanent magnet synchronous motor rotor, a first phase angle Δ _t1 to time delay, the first delay phase angle Δ _t1 is calculated by the following equation:

Δ _t1 = A · T _ctrl ≈0.5T _ctrl .

Among them, T _ctrl is a control interruption cycle of the control algorithm;

The second sub-compensation phase angle is calculated by the following formula:

θ _cmps2 = Δ _t2 · ω

Where ω is the current angular velocity of the rotor of the direct-drive permanent magnet synchronous motor, Δ _t2 is the time delay in the PWM pulse output process, and the time delay Δ _t2 in the PWM pulse output process is calculated by the following formula:

Δ _t2 = B · T _PWM + C · T _PWM ≈0.75T _PWM

Among them, T _PWM is the PWM modulation carrier period, B is the modulation algorithm interrupt delay coefficient, and C is the PWM pulse output delay coefficient;

The current expected control phase angle is calculated by the following formula:

Among them, θ _ctrl represents the expected control phase angle,

Represents the given value of q-axis voltage,

Denote the given value of d-axis voltage;

The online correction of the current actual control phase angle is calculated by the following formula:

Among them, k _p and k _i are correction terms, θ _ctrl is the current expected phase angle, θ _PWM is the current actual phase angle, and f _Δ is the fundamental frequency compensation term;

Obtaining the stable operating angular velocity range of the direct-drive permanent magnet synchronous motor is calculated by the following formula:

Where u _d is the d-axis voltage corresponding to any first preset angular velocity, u _q is the q-axis voltage corresponding to any first preset angular velocity, R _s is the resistance of the rotor, and L _q is any first preset D-axis inductance corresponding to angular velocity, L _d is the q-axis inductance corresponding to any first preset angular velocity, i _d is the first d-axis current corresponding to the d-axis voltage, and i _q is the first q-axis current corresponding to the q-axis voltage , Ψ _f is the back-EMF of the permanent magnet flux linkage;

The phase angle θ _{Δ of the} transmission error is calculated by the following formula:

θ _Δ = tan ^-1 (u _d / u _q )

The obtained third sub-compensation phase angle θ _cmps3 is calculated by the following formula:

θ _cmps3 = k · ω.

In an embodiment of the invention, the electric locomotive further includes: at least four high-power direct-drive permanent magnet synchronous motors; the megawatt direct-drive permanent magnet electric drive system for the electric locomotive includes: a first motor, a second Motor, third motor and fourth motor;

Collect the rotor frequencies of the first motor, the second motor, the third motor and the fourth motor to obtain the real-time torque of the first motor, the first motor and the second motor are the shaft motors of the first bogie , The third motor and the fourth motor are shaft motors of a second bogie, and the first bogie is adjacent to the second bogie;

Determine the rotor frequency difference and the rotor frequency differential value of the first motor according to the collected rotor frequencies of the multiple motors;

Determine the amount of torque reduction according to the rotor frequency difference, rotor frequency differential value and real-time torque of the first motor;

The torque of the first motor is adjusted according to the torque reduction amount.

In a possible implementation manner, the method further includes:

According to the rotor frequency difference, the rotor frequency differential value and the real-time torque of the first motor, a sanding control signal is generated, and the sanding control signal is used to indicate whether to perform sanding operation.

In a possible implementation manner, the torque reduction amount is determined according to the rotor frequency difference, the rotor frequency differential value, and the real-time torque of the first motor, including:

According to the rotor frequency difference of the first motor and the preset rotor frequency difference level rules, determine the idling coasting level corresponding to the rotor frequency difference of the first motor;

The first torque reduction amount is determined according to the idling coasting level corresponding to the rotor frequency difference of the first motor and the real-time torque of the first motor;

According to the first motor rotor frequency differential value and the preset rotor frequency differential value classification rules, determine the idling coasting level corresponding to the first motor rotor frequency differential value;

The second torque reduction amount is determined according to the idling coasting level corresponding to the differential value of the rotor frequency of the first motor and the real-time torque of the first motor;

If the first torque reduction amount is greater than or equal to the second torque reduction amount, the first torque reduction amount is determined to be the torque reduction amount;

If the first torque reduction amount is smaller than the second torque reduction amount, the second torque reduction amount is determined as the torque reduction amount.

In a possible implementation manner, adjusting the torque of the first motor according to the torque reduction includes:

Reduce the torque value of the first motor from the first value to the second value within the first preset time period, and the difference between the first value and the second value is the torque reduction amount;

Keep the torque value of the first motor unchanged at the second value during the second preset time period;

Within a third preset time period, increase the torque value of the first motor from the second value to a preset percentage of the preset torque value;

Within the fourth preset time period, increase the torque value of the first motor to a preset torque value;

The recovery rate of the torque value of the first motor in the third preset time period is greater than the recovery rate of the torque value of the first motor in the fourth preset time period.

In a possible implementation, reducing the torque value of the first motor from the first value to the second value within the first preset time period includes:

During the first preset time period, the torque value of the first motor is gradually reduced according to the rate of decrease of the torque value of the first motor, and the torque value of the first motor is reduced from the first value to the second value.

In a possible implementation manner, determining the rotor frequency difference and the rotor frequency differential value of the first motor according to the collected rotor frequencies of the multiple motors includes:

Limiting filtering and low-pass filtering of the rotor frequency of multiple motors collected;

The rotor frequency difference and the rotor frequency differential value of the first motor are determined according to the rotor frequencies of the multiple motors after the limiting filtering and low-pass filtering.

In a possible implementation manner, if the locomotive is in an idle running condition, the amplitude filtering and low-pass filtering processing of the collected multiple rotor frequencies includes:

Obtain the current value of the first motor;

According to the current value of the first motor and the rotor frequency of each motor, determine the rotor frequency compensation coefficient of each motor;

Compensate the rotor frequency of each motor according to the rotor frequency compensation coefficient of each motor;

Limiting filtering and low-pass filtering are performed on the compensated rotor frequencies of multiple motors.

In an embodiment of the present invention, the main circuit further includes: a plurality of sensors; the plurality of sensors includes at least one or more of the following: input current sensor, intermediate voltage sensor, ground voltage sensor, chopper branch Current sensor, motor U-phase current sensor, motor V-phase current sensor, motor stator winding temperature sensor and motor speed sensor;

The control method further includes:

Acquiring data collected by the multiple sensors;

Judging whether at least one single item state corresponding to the multiple sensors is normal according to the data and preset conditions;

If there is an abnormal single-item state, the state of the abnormal single-item state is placed in the fault bit.

In a possible design, an input current sensor is provided on the current input terminal, wherein the single-state corresponding to the input current sensor is the input current;

Obtaining the data collected by the sensor includes:

Acquiring the first current collected by the input current sensor;

According to the data and preset conditions, determining whether at least one single item state corresponding to the sensor is normal includes:

If the duration that the first current is greater than the first preset threshold is greater than the first preset time, it is determined that the input current of the traction converter is excessive.

In a possible design, the intermediate voltage sensor and the ground voltage sensor connected in parallel with the bus capacitor, wherein the single-state corresponding to the intermediate voltage sensor is the intermediate DC bus voltage, and the single-state corresponding to the ground voltage sensor is Working status of ground voltage sensor;

Obtaining the data collected by the sensor includes:

Acquiring a first voltage collected by the intermediate voltage sensor and a second voltage collected by the ground voltage sensor;

If the duration that the first voltage is greater than the second preset threshold is greater than the second preset time, it is determined that the intermediate DC bus voltage of the traction converter is too large;

If the duration that the first voltage is less than the third preset threshold is greater than the third preset time, it is determined that the intermediate DC bus voltage of the traction converter is too small;

If the second voltage value is not within the first preset range, it is determined that the ground voltage sensor is faulty;

The method also includes:

If the first voltage is not within the second preset range, it is determined that the intermediate voltage sensor is faulty;

If the duration that the third voltage obtained by subtracting half of the first voltage from the second voltage is greater than the fourth preset threshold is greater than the fourth preset time, it is determined that the positive pole of the bus of the traction converter is grounded;

If the duration that the third voltage is less than the fifth preset threshold is greater than the fifth preset time, it is determined that the negative pole of the bus of the traction converter is grounded.

In a possible design, a chopping branch current sensor is provided on the chopping branch, wherein the single state corresponding to the chopping branch current sensor is the chopping branch current;

Obtaining the data collected by the sensor includes:

Acquiring a second current collected by the chopper branch current sensor;

If the chopper branch opens, and the duration that the second current is greater than the sixth preset threshold is greater than the sixth preset time, it is determined that the current of the chopper branch of the traction converter is excessive;

If the chopper branch is not opened, and the duration that the second current is greater than the seventh preset threshold is greater than the seventh preset time, it is determined that the chopper circuit of the traction converter is not opened but the current is detected;

If the chopper branch is opened and the second current is not detected to be greater than the eighth preset threshold within the eighth preset time, it is determined that the chopper branch of the traction converter is opened but no current is detected.

In a possible design, a motor U-phase current sensor, a motor V-phase current sensor, a motor stator winding temperature sensor and a motor speed sensor are provided at the current output terminal, wherein the single-phase state corresponding to the motor U-phase current sensor is The motor U-phase input current, the single-phase state corresponding to the motor V-phase current sensor is the motor V-phase input current, the single-phase state corresponding to the motor stator winding temperature sensor is the motor stator winding temperature, and the single-phase state corresponding to the motor speed sensor Is the motor speed;

Obtaining the data collected by the sensor includes:

Obtain the third current collected by the U-phase current sensor of the motor, obtain the fourth current collected by the V-phase current sensor of the motor, obtain the temperature collected by the temperature sensor of the stator winding of the motor, and obtain the speed sensor of the motor The first speed obtained

If the duration of the third current being greater than the ninth preset threshold is greater than the ninth preset time, it is determined that the U-phase input current of the motor is excessive;

If the duration of the fourth current being greater than the tenth preset threshold is greater than the tenth preset time, it is determined that the V-phase input current of the motor is excessive;

If the duration that the temperature is greater than the eleventh preset threshold is greater than the eleventh preset time, it is determined that the temperature of the motor stator winding is too high;

If the duration that the first speed is greater than the twelfth preset threshold is greater than the twelfth preset time, it is determined that the motor speed is too large;

If the duration obtained when the fifth current obtained by inverting the value of the third current plus the fourth current is greater than the thirteenth threshold is greater than the thirteenth preset time, it is determined that the W-phase input current of the motor is excessive.

In summary, in the megawatt direct-drive permanent magnet electric drive system for electric locomotives provided by this embodiment, the alternating current of the main transformer is passed through the "AC-DC-AC" through the four-quadrant rectifier, the intermediate DC loop and the inverter module The process is finally converted to three-phase AC power available for high-power direct-drive permanent magnet synchronous motors. Therefore, the high-power direct-drive permanent magnet synchronous motor in the electric locomotive using the high-power direct-drive permanent magnet synchronous motor is controlled by the megawatt direct-drive permanent magnet electric drive system for electric locomotives, which fills the megawatt direct-drive permanent magnet synchronous motor for electric locomotive The application of the drive permanent magnet electric drive system in electric locomotives.

BRIEF DESCRIPTION

In order to more clearly explain the embodiments of the present invention or the technical solutions in the prior art, the following will briefly introduce the drawings required in the embodiments or the description of the prior art. Obviously, the drawings in the following description are only These are some embodiments of the present invention. For those of ordinary skill in the art, without paying any creative work, other drawings can be obtained based on these drawings.

1 is a schematic structural block diagram of an embodiment of a megawatt direct-drive permanent magnet electric drive system for electric locomotives of the present invention;

2 is a schematic structural circuit diagram of an embodiment of a megawatt direct drive permanent magnet electric drive system for electric locomotives of the present invention;

3 is a schematic flow chart of an embodiment of a megawatt direct drive permanent magnet electric drive system for electric locomotives of the present invention;

4 is a partial circuit diagram of a four-quadrant rectifier provided by an embodiment of the present invention;

5 is a schematic flow chart of a method for adjusting current offset of a megawatt direct drive permanent magnet electric drive system for an electric locomotive provided by an embodiment of the present invention;

6 is a schematic flow chart of a method for adjusting a current offset of a megawatt direct drive permanent magnet electric drive system for an electric locomotive provided by an embodiment of the present invention;

7 is a schematic flowchart of a method for adjusting current offset of a megawatt direct-drive permanent magnet electric drive system for an electric locomotive provided by an embodiment of the present invention provided by this embodiment

8 is a schematic flowchart of Embodiment 1 of a chopping control method provided by the present invention;

9 is a schematic structural view of an embodiment of a megawatt direct drive permanent magnet electric drive system for electric locomotives of the present invention;

10 is a schematic flowchart of Embodiment 2 of the chopper control method provided by the present invention;

11 is another schematic flowchart of Embodiment 2 of the chopper control method provided by the present invention;

12 is a schematic flowchart of Embodiment 3 of a chopper control method provided by the present invention;

13 is a schematic flowchart of a control method for a high-power direct-drive permanent magnet synchronous motor in a megawatt direct-drive permanent magnet electric drive system for an electric locomotive provided by the present invention;

14 is a schematic structural view of a control system for a high-power direct-drive permanent magnet synchronous motor in a megawatt direct-drive permanent magnet electric drive system for an electric locomotive provided by the present invention;

15 is a schematic diagram of the system structure of the MTPA control system of the present invention;

16 is a schematic diagram of the system structure of the front-end decoupling control of the present invention;

17 is a schematic diagram of the system structure of the field weakening control of the present invention;

18 is a schematic diagram of the trajectory of MTPA control and field weakening control in the full speed range of the present invention;

19 is a schematic diagram of switching control of MTPA control and field weakening control of the present invention;

20 is a schematic flowchart of a modulation method for a high-power direct-drive permanent magnet synchronous motor in a megawatt direct-drive permanent magnet electric drive system for an electric locomotive provided by the present invention;

21 is the relationship between the modulation angle and the modulation ratio under the intermediate 60 ° modulation method provided by the present invention;

22 is a schematic diagram of a full speed range modulation strategy based on intermediate 60 ° modulation provided by the present invention;

23 is a schematic flowchart of Embodiment 1 of a method for detecting a rotor initial position angle of a permanent magnet synchronous motor provided by the present invention;

24 is a schematic diagram of the relationship between the two-phase synchronous rotating coordinate system, the two-phase stationary coordinate system and the expected two-phase synchronous rotating coordinate system provided by the present invention;

25 is a schematic flowchart of Embodiment 2 of a method for detecting a rotor initial position angle of a permanent magnet synchronous motor provided by the present invention;

26 is a schematic flowchart of Embodiment 3 of a method for detecting a rotor initial position angle of a permanent magnet synchronous motor provided by the present invention;

Figure 27 is a schematic diagram of signal changes of multiple channels during the operation of a permanent magnet synchronous motor;

Figure 28 is a schematic diagram of the response current change law;

29 is a schematic structural view of a control system for a high-power direct-drive permanent magnet synchronous motor corresponding to a megawatt direct-drive permanent magnet electric drive system for an electric locomotive provided by the present invention;

30 is a first schematic flowchart of a control method of a high-power direct-drive permanent magnet synchronous motor provided by the present invention;

31 is a second schematic flowchart of a control method of a high-power direct-drive permanent magnet synchronous motor provided by the present invention;

32 is a schematic diagram of an interruption cycle of a control algorithm provided by the present invention;

33 is a schematic diagram of the interruption cycle of the modulation algorithm provided by the present invention;

Figure 34 is a schematic diagram of a multi-mode PWM modulation strategy;

35 is a third schematic flowchart of a control method of a high-power direct-drive permanent magnet synchronous motor provided by the present invention;

Figure 36A is a schematic diagram of the theoretical coordinate system and the actual coordinate system completely coincide;

Fig. 36B is a schematic diagram of the actual coordinate system leading the theoretical coordinate system;

Figure 36C is a schematic diagram of the actual coordinate system lagging behind the theoretical coordinate system;

37 is a flowchart of an embodiment of the adhesion control method provided by the present invention;

38 is a schematic diagram of an adhesion control process provided by an embodiment of the present invention;

39 is a circuit diagram of a traction converter provided by an embodiment of the present invention;

40 is a flowchart of a fault determination method for a traction converter provided by an embodiment of the present invention;

41 is a logic judgment diagram of a protection method for a traction converter provided by an embodiment of the present invention.

detailed description

The technical solutions in the embodiments of the present invention will be described clearly and completely in conjunction with the drawings in the embodiments of the present invention. Obviously, the described embodiments are only a part of the embodiments of the present invention, but not all the embodiments. Based on the embodiments of the present invention, all other embodiments obtained by a person of ordinary skill in the art without making creative efforts fall within the protection scope of the present invention.

The terms "first", "second", "third", "fourth", etc. (if any) in the description and claims of the present invention and the above drawings are used to distinguish similar objects without using To describe a specific order or sequence. It should be understood that the data used in this way can be interchanged under appropriate circumstances, so that the embodiments of the present invention described herein can be implemented in an order other than those illustrated or described herein, for example. In addition, the terms "including" and "having" and any variations thereof are intended to cover non-exclusive inclusions, for example, processes, methods, systems, products or devices that contain a series of steps or units need not be limited to those clearly listed Those steps or units, but may include other steps or units not explicitly listed or inherent to these processes, methods, products, or equipment.

The technical solutions of the present invention will be described in detail below with specific examples. The following specific embodiments may be combined with each other, and the same or similar concepts or processes may not be repeated in some embodiments.

FIG. 1 is a schematic structural view of an embodiment of a megawatt direct drive permanent magnet electric drive system for electric locomotives of the present invention. As shown in FIG. 1, the megawatt direct drive permanent magnet electric drive system for electric locomotives provided in this embodiment includes: a first precharge module, a second precharge module, a first four-quadrant rectifier, a second four-quadrant rectifier, First chopper module, second chopper module, intermediate DC loop, first inverter module, second inverter module, third inverter module and auxiliary converter, first four-quadrant rectifier and second four-quadrant rectifier The main transformer of the electric locomotive is connected through the first pre-charging module and the second pre-charging module respectively. The DC loop is connected to the first inverter module, the second inverter module, the third inverter module and the auxiliary converter, respectively.

Specifically, the megawatt direct-drive permanent magnet electric drive system for electric locomotives provided in this embodiment can be used for electric locomotives using high-power direct-drive permanent magnet synchronous motors for controlling at least one high-power direct-drive permanent-drive permanent locomotive Magnetic synchronous motor. It should be noted that in each embodiment of the present invention, the number of high-power direct-drive permanent magnet synchronous motors in the megawatt direct-drive permanent magnet electric drive system for electric locomotives is three as an example. The tile-level direct-drive permanent magnet electric drive system can also be used to control electric locomotives with less or more than three high-power direct-drive permanent magnet synchronous motors. The principle is the same and only increases or decreases in number.

Further, FIG. 2 is a schematic structural diagram of an embodiment of a megawatt direct drive permanent magnet electric drive system for electric locomotives of the present invention. The embodiment shown in FIG. 2 provides a specific circuit design and connection method of a megawatt direct-drive permanent magnet electric drive system for electric locomotives on the basis of FIG. 1 to illustrate subsequent implementations of the present invention In the example, the control method for the megawatt direct drive permanent magnet electric drive system for electric locomotives.

Wherein, as shown in the circuit diagram shown in FIG. 2, the first pre-charging module includes a first charging capacitor, a first pre-charging contactor and a first main working contactor, and the second pre-charging module includes a second charging capacitor and a second pre-charging The charging contactor and the second main working contactor, the first four-quadrant rectifier and the second four-quadrant rectifier each include eight switch tubes, and the first chopper module includes the first switch tube, the first current sensor, and the first reverse diode And the first chopping resistor, the second chopping module includes a second switch tube, a second current sensor, a second reverse diode, and a second chopping resistor, and the intermediate DC loop includes a first DC side support capacitor connected in parallel, The second DC side support capacitor, slow discharge resistor, DC bus voltage sensor and ground detection module, the first inverter module, the second inverter module and the third inverter module all include a three-phase inverter composed of six switch tubes Circuit.

Specifically, as shown in FIG. 2, the first precharge module is used for description, and the composition and implementation principle of the second precharge module and the first precharge module are the same. The first precharge contactor AK1 is connected to the secondary winding 1 of the transformer and the first precharge resistor R1, and the first precharge resistor R1 is also connected to the output terminal of the first precharge module (connected to the input terminal of the first four-quadrant rectifier), The first main working contactor K1 is connected to the secondary winding 1 of the transformer and the output terminal of the first precharge module (connected to the input terminal of the first four-quadrant rectifier). Among them, due to the large transformer current, and in order to protect the four-quadrant rectifier from the excessive input current and damage to the switching tube, this application requires a special pre-charging module for the converter of the high-power direct-drive permanent magnet synchronous motor to prevent the transformer The excessive current is directly output to the four-quadrant rectifier. In actual use, when the converter is powered on, the first precharge contactor AK1 is closed, the first main working contactor K1 is opened, and the transformer current reaches the first four-quadrant rectifier after passing through the first precharge resistor R1, so that The current change range (di / dt) at the beginning of power-on is not too large, reducing the damage to each device. After 3-10 ms, the first main working contactor K1 is closed, the first precharge contactor AK1 is opened, and the transformer current directly reaches the first four-quadrant rectifier.

As shown in Fig. 2, the first four-quadrant rectifier and the second four-quadrant rectifier are both composed of eight switch tubes. The following uses the first four-quadrant rectifier as an example. The quadrant rectifier is the same. Among them, the first four-quadrant rectifier is composed of eight IGBT switch tubes of g1, g3, g2, g4, g5, g7, g6 and g8 in figure 1, specifically, the emitter of g1 is connected with the collector of g2, The emitter of g3 is connected to the collector of g4, the emitter of g5 is connected to the collector of g6, and the emitter of g7 is connected to the collector of g8. The emitters of g1 and g3 are connected together and connected to the first input of the first four-quadrant organizer, and the emitters of g5 and g7 are connected together and connected to the second input of the first four-quadrant rectifier, g1, The collectors of g3, g5 and g7 are connected together and connected to the first output of the first four-quadrant rectifier, and the emitters of g2, g4, g6 and g8 are connected together and connected to the second of the first four-quadrant rectifier The output is connected.

As shown in FIG. 2, the first chopping module and the second chopping module have the same implementation principle, where the first chopping module includes a chopping switch g9, a chopping current sensor A2, a reverse diode D1, and a chopping resistor R5, chopper module 2 and chopper module 1 have the same structure. The specific implementation principle of the chopper module will be described in the embodiment shown in FIG. 6 later in this application.

As shown in FIG. 2, the first inverter, the second inverter, the third inverter, and the auxiliary converter are each composed of 6 IGBTs. The following uses the first inverter as an example for description. Among them, for the first inverter, the emitter of g10 is connected to the collector of g11, the emitter of g12 is connected to the collector of g13, and the emitter of g14 is connected to the collector of g15, The collectors of g10, g12 and g14 are connected together and connected to the first input of the first inverter, and the emitters of g11, g13 and g15 are connected together and to the second input of the first inverter connection. The emitters of g10, g12, and g14 are the three-phase output terminals of the first inverter, as shown in FIG. 2, the emitter of g10 is the first output terminal of the first inverter, and the emitter of g12 is the first reverse The second output of the converter; the emitter of g14 is the third output of the first inverter.

3 is a schematic flow chart of an embodiment of a megawatt direct-drive permanent magnet electric drive system for electric locomotives of the present invention. The control method shown in FIGS. 1 and 2 is described below with reference to FIG. 3, wherein the control method of the megawatt direct drive permanent magnet electric drive system for electric locomotives includes:

S101: Transmit the AC power of the main transformer to the first four-quadrant rectifier and the second four-quadrant rectifier through the first precharge module and the second precharge module, respectively;

Specifically, the execution subject of this embodiment may be any electronic device with related control and data processing functions, such as a tablet computer, a notebook computer, a desktop computer, and a server. Alternatively, this embodiment may be further executed by the processor of the electronic device, for example, CPU, GPU, and so on.

Among them, the control method of this embodiment is used to control the main circuit shown in FIG. 1 to convert the AC power of the converter into a three-phase variable-frequency variable-voltage AC power that can be used by a high-power direct-drive permanent magnet synchronous motor. Then in S102, the first precharge module connected to the main transformer is controlled to input the AC power of the main transformer to the first four-quadrant rectifier, and the second precharge module connected to the main transformer is controlled to input the AC power of the main transformer to the second four-quadrant rectifier . The pre-charging module is used to protect the devices of the four-quadrant rectifier from being damaged by excessive current or voltage output directly from the main transformer. The input terminals of the first pre-charging module and the second pre-charging module can obtain the alternating current provided by the main transformer by connecting to the secondary traction winding of the main transformer.

S102: After the first four-quadrant rectifier and the second four-quadrant rectifier convert the alternating current transmitted by the first pre-charging module and the second pre-charging module into direct current, respectively, output to the first chopper module and the second chopper module.

Then, in S102, the first four-quadrant rectifier and the second four-quadrant rectifier can be controlled to convert the AC power of the main transformer received from the first pre-charging module and the second pre-charging module into DC power and input the first chopper module and The second chopping module. Optionally, in the same or similar main circuit alternatives of the present invention, the number of four-quadrant rectifiers is not specifically limited. For each four-quadrant rectifier arranged in parallel, each four-quadrant rectifier works independently and is used to pass the corresponding The pre-charging module receives the AC power provided by the main transformer and converts it into DC power, and outputs it to the intermediate DC loop.

S103: DC wave is chopped by the first chopping module and the second chopping module and then transmitted to the intermediate DC loop

Specifically, the DC power output by the first four-quadrant rectifier and the DC power output by the second four-quadrant rectifier are respectively chopped by controlling the first chopping module and the second chopping module, and then transmitted to the intermediate DC circuit.

S104: Output the received DC power to the first inverter module, the second inverter module, the third inverter module, and the auxiliary converter through the intermediate DC loop, respectively.

After the intermediate DC loop receives the DC power sent by the first four-quadrant rectifier and the second four-quadrant rectifier, the DC loop is controlled in S104 to direct the DC power to the first inverter module, the second inverter module, and the third Inverter module and auxiliary converter output. Among them, the first four-quadrant rectifier and the second four-quadrant rectifier share the intermediate DC circuit, and the intermediate DC circuit transmits the received multiple DC power to the first inverter module, the second inverter module, and the third inverse Transformer module and auxiliary converter output.

S105: Convert the received DC power into three-phase AC power through the first inverter module, the second inverter module, and the third inverter module, and then output to three high-power direct-drive permanent magnet synchronous motors, respectively.

Then, in S105, after receiving the DC power sent by the intermediate circuit, it is necessary to control the first inverter module, the second inverter module, the third inverter module, and the auxiliary converter to supply power to the loads connected to them respectively. Among them, the inverter module corresponds to the high-power direct-drive permanent magnet synchronous motor, and the auxiliary converter corresponds to the auxiliary load. For example, in the embodiment of the main circuit shown in FIG. 1, the electric locomotive includes three high-power direct-drive permanent magnet synchronous motors, so the main circuit also needs to be provided with three inverter modules accordingly. As shown in the connection relationship in the figure, the first inverter module is connected to a high-power direct-drive permanent magnet synchronous motor 1, and converts the received DC power into the AC power available for the high-power direct-drive permanent magnet synchronous motor 1, and outputs it to the second The inverter module is connected to the high-power direct-drive permanent magnet synchronous motor 2, and converts the received DC power into the alternating current available to the high-power direct-drive permanent magnet synchronous motor 2. The third inverter module is connected to the high-power direct-drive permanent drive The magnetic synchronous motor 3 converts the received DC power into high-power direct-drive permanent magnet synchronous motor 3 usable AC power and outputs it to it. Each inverter module drives the high-power direct-drive permanent-magnet synchronous motor through the AC power sent to the high-power direct-drive permanent-magnet synchronous motor connected to it, thereby realizing the driving of three high-power direct-drive permanent-magnet synchronous motors in the electric locomotive control.

S106: Convert the received DC power into three-phase AC power through the auxiliary converter and output it to the auxiliary load of the electric locomotive.

At the same time, in the main circuit provided in this embodiment, the auxiliary converter can also be connected to the intermediate DC circuit, and in S106, the auxiliary converter can be controlled to convert the DC power received from the intermediate DC circuit into an auxiliary load available in the electric locomotive After the AC power is supplied to the auxiliary load. Optionally, the auxiliary load described herein includes at least one or more of the following: a lighting system, a communication system, and an air conditioning system of an electric locomotive.

In summary, in the megawatt direct-drive permanent magnet electric drive system for high-power electric locomotive provided by this embodiment and its control method, the pre-charge module, four-quadrant rectifier, chopper module, intermediate DC loop and inverter module are sequentially passed , The AC power of the main transformer is finally converted into three-phase AC power available for high-power direct-drive permanent magnet synchronous motors through the "AC-DC-AC" process. In order to control the high-power direct-drive permanent magnet synchronous motor in the electric locomotive using the high-power direct-drive permanent magnet synchronous motor, it fills the converter and the type of motor of the high-power direct-drive permanent magnet synchronous motor in the electric locomotive. The blank of its control method.

Optionally, in a specific implementation of the control method of the present invention, a control method for the four-quadrant rectifier in S102 is provided to eliminate the influence of current bias during the control process of the four-quadrant rectifier.

Specifically, FIG. 4 is a partial circuit diagram of a four-quadrant rectifier provided by an embodiment of the present invention. The four-quadrant rectifier shown in FIG. 4 may be the first four-quadrant rectifier shown in FIGS. 1 and 3, or may be as shown in FIG. 1. And the second four-quadrant rectifier in Figure 3. The working mode and principle of each four-quadrant rectifier provided in this embodiment are the same, and a four-quadrant rectifier will be specifically described below. As shown in the figure, g1, g2, g3, and g4 are IGBT devices of four-quadrant rectifier, and g1, g2, g3, and g4 work together to realize the function of four-quadrant rectifier to convert AC voltage into DC voltage. However, in the prior art, when the four-quadrant rectifier is biased due to factors such as device and control, the four-quadrant rectifier will be unstable, and the IGBT device deviates from its rated working area, which will cause a large DC bias on the transformer. Based on this problem, in an embodiment of the megawatt direct drive permanent magnet electric drive system for electric locomotives of the present invention, a method for adjusting the current offset of the megawatt direct drive permanent magnet electric drive system for electric locomotives is provided in S101. The method can solve the problem of DC bias without changing the hardware structure of FIG. 1 and FIG. 3. Detailed description will be given below with reference to FIG. 5.

FIG. 5 is a schematic flowchart of a current offset adjustment method for a megawatt direct-drive permanent magnet electric drive system for an electric locomotive provided by an embodiment of the present invention. As shown in FIG. 5, the method includes:

S501. Sampling the alternating current input to the four-quadrant rectifier to obtain the alternating current in the sampling period, where the alternating current includes the current value of the positive half cycle and the current value of the negative half cycle.

Specifically, according to the preset sampling frequency, the AC current input to the four-quadrant rectifier is sampled to obtain multiple sampling points, and the obtained multiple sampling points are drawn into a curve to obtain a sine or cosine curve. The preset sampling frequency may be twice or even several times of the IGBT on-off frequency or other, as long as the complete sine or cosine curve can be sampled according to the preset sampling frequency, and the preset sampling frequency is not particularly limited here. For example, in this embodiment, the preset sampling frequency may be twice the on-off frequency of the IGBT, and then a sine or cosine curve drawn from multiple sampling points obtained according to the preset sampling frequency is divided into positive half cycles according to the phase And the negative half cycle, for example, the positive half cycle of the sine curve is 0 to π, and the negative half cycle is π to 2π, then the values of the multiple sampling points of the positive half cycle are the value of the positive half cycle of the AC current, and the number of negative half cycles The value of each sampling point is the value of the negative half cycle of the AC current.

S502. Acquire a first sum value of the current value of the positive half cycle and a second sum value of the current value of the negative half cycle, and obtain a current offset value according to the first sum value and the second sum value.

Specifically, the values of the multiple sampling points in the positive half cycle are added to obtain the first sum P, and then the values of the multiple sampling points in the negative half cycle are added to obtain the second sum N, P and N The absolute value of the value is calculated as the difference, and the resulting difference is Q. If the Q value is 0, the absolute values of the P value and the N value are also completely equal, the positive half cycle and the negative half cycle of the sine curve or cosine curve are completely symmetrical, and the AC current has no DC offset. If the Q value is not 0, the absolute value of the P value and the N value are not equal, then the positive half cycle and negative half cycle of the sine curve or cosine curve are asymmetric, the AC current has a DC offset, and the Q value is the DC offset Set value.

S503. Input the first difference between the current offset value and zero to the first PI controller, and obtain the first output value output by the first PI controller.

Specifically, the DC offset value Q and zero are input to the first PI controller. The first PI controller forms a control deviation according to the DC offset value Q and zero, and linearly combines the proportion and integral of the deviation to form a control amount. The current is controlled to eliminate the DC bias of the AC current. The controlled variable is the first output value.

S504. Obtain a pulse width modulation symbol according to the first output value and the second output value output by the PR controller. The PR controller is used for static-free control of the alternating current, so that the period of the alternating current and The phase is the same as the grid voltage.

Specifically, after inputting the AC current to the PR controller to ensure that the phase and cycle of the AC current are the same as the grid voltage, a stable output AC current is obtained, which is the second output value. Then, the first output value and the second output value are summed to obtain a third sum value. That is, the control quantity obtained by the first PI controller regulates and outputs a stable AC current, thereby suppressing the DC bias of the AC current. Then, the third sum value is modulated by a monopole frequency doubling pulse modulation method to obtain a pulse width modulation symbol.

S505. Control the turning on and off of the insulated gate bipolar transistor IGBT in the four-quadrant rectifier according to the pulse width modulation symbol.

Specifically, referring to FIG. 4, the pulse width modulation symbol is used as an input of the insulated gate bipolar transistors IGBTs g1, g2, g3, and g4 in the four-quadrant rectifier to control the turning on and off of the bipolar transistor IGBT.

Therefore, in this embodiment, a method for adjusting current offset in a megawatt direct-drive permanent magnet electric drive system for electric locomotives is provided for the four-quadrant input The AC current of the rectifier is sampled to obtain the AC current in the sampling period. The AC current includes the current value of the positive half cycle and the current value of the negative half cycle; the first sum value of the current value of the positive half cycle and the negative half cycle are obtained The second sum value of the current value, and obtain the current offset value according to the first sum value and the second sum value; input the first difference between the current offset value and zero to the first PI controller to obtain the first PI The first output value output by the controller; the pulse width modulation symbol is obtained according to the first output value and the second output value output by the PR controller, and the PR controller is used to control the AC current without static error, so that the period and phase of the AC current It is the same as the grid voltage; the on-off of the insulated gate bipolar transistor IGBT in the four-quadrant rectifier is controlled according to the pulse width modulation sign. The second output value is adjusted by the first output value output by the first PI controller to obtain a third sum value, thereby suppressing the DC bias of the alternating current, and modulating the third sum value by a unipolar frequency-doubled pulse modulation method The pulse width modulation symbol is used to control the operation of the IGBT, which prevents the IGBT device from deviating from its rated operating area, thereby effectively suppressing and eliminating the current bias on the transformer side, thereby eliminating the effect of the current bias on the control of the four-quadrant rectifier.

FIG. 6 is a schematic flowchart of a current bias adjustment method for a megawatt direct drive permanent magnet electric drive system for an electric locomotive provided by an embodiment of the present invention, and FIG. 7 is a megawatt for an electric locomotive provided by an embodiment of the present invention provided by the embodiment. Schematic diagram of the current offset adjustment method of the level direct drive permanent magnet electric drive system. As shown in FIG. 7, Udc is the DC bus voltage, the trap is mainly to filter the fluctuation value on the DC bus voltage Udc, and Udc * is the command voltage , I is the AC current input to the four-quadrant rectifier, Us is the voltage of the AC current input to the four-quadrant rectifier. With reference to FIG. 7, this embodiment describes the specific implementation process of this embodiment on the basis of the embodiment of FIG. . As shown in Figure 6, the method includes:

S601. Sampling the alternating current input to the four-quadrant rectifier according to a preset sampling frequency to obtain a sampling current, and the preset sampling frequency is twice the on-off frequency of the IGBT.

S601 provided in this embodiment is similar to S501 in the embodiment of FIG. 5, and details are not described herein again in this embodiment.

S602. Filter the sampling current by a first band-pass filter and a second band-pass filter to obtain a filtered sampling current; wherein, the first band-pass filter is used to obtain a main frequency signal of an alternating current, The second band-pass filter is used to filter out interference harmonics.

Specifically, considering the difference in the main frequency of the AC current in different regions, the passband frequency of the first bandpass filter is set between 40 Hz and 60 Hz, for example, in this embodiment, the passband frequency of the first bandpass filter 45-55 Hz, optionally, when the main frequency of the AC current is 50 Hz, the passband frequency of the first band-pass filter is set to 50 Hz, for acquiring the main frequency signal of the AC current. Similarly, in this embodiment, the switching frequency of the four-quadrant rectifier is f, that is, the on-off frequency of the IGBT is f, the pass band frequency of the second band-pass filter is 2f / (50 ± 5) Hz, and the second band The pass filter is used to filter out high-order harmonic interference. The first band-pass filter and the second band-pass filter are the filters in FIG. 5.

S603: Obtain a second difference between the DC bus voltage of the four-quadrant rectifier and the command voltage, and input the second difference to the second PI controller, so that the third output value output by the second PI controller Multiplied by the output value of the phase-locked loop, the phase-locked loop is used to obtain the grid voltage phase, thereby obtaining an alternating current with the same period and phase as the grid voltage.

Specifically, the DC bus voltage Udc and the command voltage Udc * are input to the second PI controller. According to the deviation of the DC bus voltage Udc and the command voltage Udc *, the proportional and integral of the deviation are linearly combined to form a control amount. The control amount is the third output value output by the second PI controller. Then, the third output value output by the second PI controller is multiplied by the output of the phase-locked loop to obtain an alternating current in the same phase as the grid voltage. The phase-locked loop is the PLL in FIG. 5. The phase-locked loop PLL is used to control the period and phase of the alternating current i and the period and phase of the grid voltage to be consistent. The phase of the grid voltage is calculated according to the phase controlled by the phase-locked loop. The second PI controller in S603 is the second PI in FIG. 7.

S604. Obtain an alternating current in a sampling period according to the grid voltage phase determined by the phase-locked loop and the sampling current, where the alternating current includes a current value in a positive half cycle and a current value in a negative half cycle.

Specifically, the phase of the grid voltage is calculated according to the phase controlled by the phase-locked loop PLL, the phase of the alternating current i is determined, and the phase of the sampling current is determined. According to the phase, the sampling current is divided into a positive half cycle and a negative half cycle. For example, the positive half cycle of the sine curve is 0 to π, and the negative half cycle is π to 2π, then the values of the multiple sampling points of the positive half cycle are the values of the positive half cycle of the AC current i, and the values of the multiple sampling points of the negative half cycle The value is the value of the negative half cycle of the alternating current i. S604 is the DC offset extraction calculation in FIG. 7.

S605: Acquire a first sum value of the current value of the positive half cycle and a second sum value of the current value of the negative half cycle, and obtain a current offset value according to the first sum value and the second sum value.

S605 provided in this embodiment is similar to S502 in the embodiment of FIG. 5, and S605 is also the calculation of the DC offset extraction in FIG. 7, which will not be repeated here in this embodiment.

S606. Determine whether the absolute value of the first difference is greater than the absolute value of the current loop width, and the obtained judgment result is yes.

Specifically, to avoid errors in the first difference Q caused by sampling errors, the Q value and the hysteresis loop width are calculated. The hysteresis loop width can be ± 5A or any other value as long as the first difference can be avoided There is an error in the value Q. For example, in this embodiment, the hysteresis loop width is ± 5A; the absolute value of the first difference Q is greater than 5A, and the obtained judgment result is yes, that is, the AC has a DC bias. Specifically, the first difference Q is greater than 5A, the AC current has a positive DC bias, the first difference Q is less than -5A, and the AC current has a negative DC bias.

S607: Input the first difference between the current offset value and zero to the first PI controller, and obtain the first output value output by the first PI controller.

S607 provided in this embodiment is similar to S503 in the embodiment of FIG. 5, and the first PI controller in S607 is the first PI in FIG. 7, which will not be repeated here in this embodiment.

S608: Summing the first output value and the second output value of the PR control output to obtain a third sum value, the first output value is a current variable, and the second output value is a current value; The pulse width modulation symbol is obtained according to the third sum value and the unipolar frequency doubling pulse modulation method.

S608 provided in this embodiment is similar to S504 in the embodiment of FIG. 5, and the PR controller in S608 is a PR in FIG. 7, which will not be repeated here in this embodiment.

S609. Control the turning on and off of the insulated gate bipolar transistor IGBT in the four-quadrant rectifier according to the pulse width modulation symbol.

S609 provided in this embodiment is similar to S505 in the embodiment of FIG. 5 and is also similar to the pulse modulation in FIG. 7, which will not be repeated here in this embodiment.

The method for adjusting a megawatt direct-drive permanent magnet electric drive system for an electric locomotive provided by an embodiment of the present invention samples an alternating current to obtain a sampled current, and then inputs the second difference between the DC bus voltage and the command voltage to the second PI The controller obtains a third output value output by the second PI controller, and the third output value is used to adjust the alternating current. After multiplying the third output value and the output value of the phase-locked loop, the phase of the AC current is determined according to the phase of the grid voltage calculated by the phase-locked loop, and then the phase of the sampled current is determined, and then the sampled current is divided into positive half periods and For the negative half cycle, calculate the current value of the positive half cycle and the negative half cycle, and then input the first difference between the current value of the positive half cycle and the current value of the negative half cycle to the first PI controller. The first output value output by a PI controller adjusts the second output value output by the PR controller to obtain a third sum value, thereby suppressing the DC bias of the alternating current, and modulating the third sum value with a unipolar frequency-doubled pulse Modulation mode, the pulse width modulation symbol is used to control the operation of the IGBT, which avoids the IGBT device from deviating from its rated working area, thereby effectively suppressing and eliminating the current bias on the transformer side, thereby eliminating the current bias control of the four-quadrant rectifier influences.

Further, the current bias adjustment method for the megawatt direct-drive permanent magnet electric drive system for electric locomotives provided by the embodiments of the invention improves the response speed of DC bias suppression, and uses software control algorithms to solve the DC bias, eliminating the need for The hardware circuit design solves the problem that other DC offset suppression methods are not suitable for wide-band changes of grid voltage frequency.

Optionally, in a specific implementation manner of the control method of the present invention, a control method for the intermediate DC loop in S104 is provided, which specifically relates to a method of chopper control for the intermediate DC loop to reduce the use of megabytes in electric locomotives The impact on the intermediate DC bus voltage in the tile-level direct drive permanent magnet electric drive system. The chopping control method of the intermediate DC circuit provided in this embodiment will be described below with reference to FIGS. 8 and 9.

Specifically, FIG. 8 is a schematic flowchart of Embodiment 1 of the chopping control method provided by the present invention. As shown in FIG. 8, the chopping control method provided by this embodiment includes:

S801: Perform periodic detection on the intermediate DC bus voltage, where the intermediate DC bus voltage is the voltage on the DC bus on the AC-DC-AC electric locomotive.

9 is a schematic structural view of an embodiment of a megawatt direct drive permanent magnet electric drive system for electric locomotives of the present invention. The main circuit shown in Fig. 9 is a possible connection method based on Fig. 1. The main circuit shown in FIG. 9 includes a pre-charge module 1 and a pre-charge module 2, a four-quadrant rectifier module 1 and a four-quadrant rectifier module 2, a chopper module 1 and a chopper module 2, a ground detection module, an inverter module 1, and an inverse Transformer module 2 and inverter module 3, and auxiliary modules.

The pre-charging module 1 includes a pre-charging resistor R1, a pre-charging contactor AK1 and a main working contactor K1, and the pre-charging module 2 and the pre-charging module 1 have the same structure. The four-quadrant rectifier module 1 is composed of eight switch tubes g1, g3, g2, g4, g5, g7, g6 and g8. The four-quadrant rectifier module 2 and the four-quadrant rectifier module 1 have the same structure. The chopper module 1 includes a chopper switch g9, a chopper current sensor A2, a reverse diode D1, and a chopper resistor R5. The chopper module 2 and the chopper module 1 have the same structure. The grounding detection module includes resistors R3 and R4, and the resistance value of R3 is equal to R4. The resistors R3 and R4 are connected in series at both ends of the DC loop to form a grounding resistance detection loop. The inverter module 1 includes a three-phase inverter circuit composed of six switch tubes g10, g11, g12, g13, g14, and g15. The inverter module 2, the inverter module 3, and the inverter module 1 have the same structure. K2 is a motor isolation contactor, M is a direct-drive permanent magnet motor, C1 and C3 are DC-side supporting capacitors, R2 is a slow discharge resistor, and U1 is a DC bus voltage sensor. The auxiliary module includes a three-phase inverter circuit composed of six switch tubes, g16, g17, g18, g19, g20 and g21, and an auxiliary filter cabinet. In the topology diagram of the main circuit shown in FIG. 9, the intermediate DC bus voltage mentioned in this embodiment refers to the voltage measured by U1.

S802. When the detected intermediate DC bus voltage value is greater than the upper chopping threshold, the P regulator is used to adjust the intermediate DC bus voltage; until the detected intermediate DC bus voltage value is less than the lower chopping threshold, the The upper chopping threshold is greater than the lower chopping threshold.

Among them, the principle of the P regulator is to control the chopper tube to be in an open state within a certain time proportion of the detection cycle. The specific time ratio is related to the detected intermediate DC bus voltage value. The larger the detected intermediate DC bus voltage value, the greater the time ratio.

Because the voltage value of the intermediate DC bus decreases from greater than the upper chopping threshold to less than the lower chopping threshold, the chopper tube is not always in the open state. Compared with the prior art, the intermediate DC bus is reduced. The impact of voltage.

It should be noted that after the P regulator is used to adjust the intermediate DC bus voltage, when it is detected that the intermediate DC bus voltage value is less than the lower chopping threshold, the chopper tube is directly controlled to be turned off.

The chopping control method provided in this embodiment is applied to AC-DC-AC electric drive locomotives to periodically detect the intermediate DC bus voltage. When the detected intermediate DC bus voltage value is greater than the upper chopping threshold, the P regulator is used to The intermediate DC bus voltage is adjusted; until the detected value of the intermediate DC bus voltage is less than the lower chopping threshold, the impact on the intermediate DC bus voltage is reduced.

10 is a schematic flowchart of Embodiment 2 of the chopper control method provided by the present invention. This embodiment is a further description of the achievable manner of S802 in the above embodiment. As shown in FIG. 10, S802 includes:

S1001: Using the P regulator, determine the chopping duty cycle within the target detection period.

Wherein, the target detection period includes: the detected detection period between the detected intermediate DC bus voltage value being greater than the upper chopping threshold and the detected intermediate DC bus voltage value being less than the lower chopping threshold.

For example, assuming that the detection period is 1min, if the voltage value of the intermediate DC bus detected in the current detection period (1min) is greater than the upper chopping threshold, then the P regulator will be used to adjust the intermediate DC bus voltage. If the middle DC bus voltage value is less than the lower chopping threshold in the fifth detection period from the current detection period, the current 1min, the second 1min, the third 1min, and the fourth 1min are the target detection period.

Among them, the chopping duty ratio refers to: the ratio of the time that the chopper tube is turned on to the detection period within one detection period.

Optionally, referring to FIG. 11, the above achievable way of determining the chopping duty cycle within the target detection period is:

First, determine the target parameters, specifically:

S2011, the target parameter is determined according to the following formula;

Err = U1-chopping lower threshold

Secondly, the control coefficient corresponding to the P regulator is obtained, specifically:

S2012. Determine the control coefficient according to the following formula;

Among them, Kp_chp represents the control coefficient.

Finally, according to the control coefficient and the target parameter, the chopping duty ratio is determined, specifically:

S2013. Determine the chopping duty ratio according to the following formula;

C_duty = Err * Kp_chp

Taking the topology diagram shown in FIG. 9 as an example for description: the upper chopping threshold is set to 3100V, the lower chopping threshold is set to 2900V, and the DC bus voltage overvoltage protection threshold is 3200V. The voltage measured by U1 in Figure 9 is the intermediate DC bus voltage. Assuming that the detected intermediate DC bus voltage value U1 in the current detection cycle is 3100V, since U1 is greater than the upper chopping threshold, a P regulator is used to adjust the intermediate DC bus voltage. First, the target parameter Err calculated according to S2011 is: 3100V- 2900V = 200V; secondly, the control coefficient Kp_chp calculated according to S2012 is: 1 / (3200V-2900V) ≈0.0033; finally, the chopping duty ratio calculated according to S2013 is: 200V * 0.0033 = 0.66. Then the chopping duty ratio is 0.66 in the current detection period.

S1002: Determine the turn-on time of the chopper tube in the target detection period according to the chopper duty ratio.

S1003: According to the turn-on time, control the turn-on or turn-off of the chopper tube so that the voltage value of the intermediate DC bus drops to less than the lower threshold of the chopper.

Because, the chopping duty ratio refers to: the ratio of the time that the chopper is turned on in the detection period within a detection period. Continue to use the example in S201 to explain: assuming that the detection period is 1min, on the basis of determining the chopping duty cycle in the current detection period is 0.66, the opening time of the chopper tube in the current detection period can be calculated as 1min * 0.66 = 0.66min.

Specifically, after the above-mentioned opening time is obtained, the opening time of the chopper tube in the current detection period can be controlled to be 0.66 min based on the opening time by controlling the opening or closing of the chopper tube.

The chopping control method provided in this embodiment describes a achievable way to determine the chopping duty ratio. Specifically, the target parameter Err is first determined, then the control coefficient of the P regulator is determined, and finally the target parameter and The control coefficient determines the chopping duty ratio, which provides a basis for subsequently controlling the opening time of the chopper tube according to the chopping duty ratio.

12 is a schematic flowchart of Embodiment 3 of a chopper control method provided by the present invention. On the basis of the foregoing embodiment, as shown in FIG. 12, the chopping control method provided in this embodiment further includes: performing error prevention processing on the chopping duty ratio.

Optionally, the implementation of the above error prevention processing is:

S1201: If the value of the chopping duty ratio is greater than 1, then set the value of the chopping duty ratio to 1; if the value of the chopping duty ratio is less than 0, then set the chopping duty ratio The value of the air ratio is set to 0.

Taking the topology diagram shown in FIG. 9 as an example for description: the upper chopping threshold is set to 3100V, the lower chopping threshold is set to 2900V, and the DC bus voltage overvoltage protection threshold is 3200V. The voltage measured by U1 in Figure 2 is the intermediate DC bus voltage. It is assumed that the voltage value of the intermediate DC bus detected in the current detection period is 3300V. Then the target parameter Err calculated according to S2011 is: 3300V-2900V = 400V; secondly, the control coefficient Kp_chp calculated according to S2012 is: 1 / (3200V-2900V) ≈0.0033; finally, the chopping duty ratio calculated according to S2013 is: 400V * 0.0033 = 1.32. If the calculated value of the chopping duty ratio is greater than 1, the value of the chopping duty ratio is set to 1. Similarly, when the calculated value of the chopping duty ratio is less than 0, the value of the chopping duty ratio is set to 0.

The chopping control method provided in this embodiment describes an implementable method of performing error prevention processing on the chopping duty ratio. Specifically, if the value of the chopping duty ratio is greater than 1, the chopping duty ratio is The value of the duty ratio is set to 1; if the value of the chopping duty ratio is less than 0, the value of the chopping duty ratio is set to 0. The ratio of the chopping duty cycle can be controlled in the range of 0 to 1.

Optionally, on the basis of the foregoing embodiment, an embodiment of the present invention also provides a method for controlling a high-power direct-drive permanent magnet synchronous motor in a megawatt direct-drive permanent magnet electric drive system for electric locomotive The speed-based segmented vector control strategy completes current closed-loop control to meet the requirements for high-speed operating range, high torque performance, and high efficiency according to the operating conditions of the locomotive.

Specifically, FIG. 13 is a schematic flowchart of a control method for a high-power direct-drive permanent magnet synchronous motor in a megawatt direct-drive permanent magnet electric drive system for an electric locomotive provided by the present invention, as shown in the embodiment shown in FIG. 13 include:

S1301: Determine the rotation speed of the high-power direct-drive permanent magnet synchronous motor to be controlled;

S1302: Determine a first control strategy according to a rotation speed and a first mapping relationship, where the first mapping relationship includes a one-to-one correspondence between at least one rotation speed range and at least one control strategy;

S1303: Determine the expected control phase angle of the high-power direct-drive permanent magnet synchronous motor to be controlled according to the first control strategy.

Optionally, the first mapping relationship in the foregoing embodiment includes at least: a correspondence relationship between the rated speed below and the MTPA control strategy; a correspondence relationship above the rated speed with the field weakening control strategy.

Specifically, the high-power direct-drive permanent magnet synchronous motor in this embodiment uses a speed-based segmented vector control strategy to complete the current closed-loop control. The control strategy includes: maximum torque current ratio (MTPA) control in the low-speed area and high-speed area Field weakening control. 14 is a schematic structural view of a control system for a high-power direct-drive permanent magnet synchronous motor in a megawatt direct-drive permanent magnet electric drive system for an electric locomotive provided by the present invention, and the above embodiment will be described below in conjunction with FIG. 14. As shown in Figure 14, where T_cmd is the input torque, T is the actual input torque after torque limiting, id * and iq * are the d-axis and q-axis current settings, and id and iq are the d-axis and q-axis feedback current, ud * and uq * are given by d-axis and q-axis voltage, ua, ub, uc are input phase voltage of motor a-phase, b-phase and c-phase, respectively, ia, ib are motor a-phase, b-phase Current.

For below the rated speed, MTPA control is adopted, that is, a control method that uses the reluctance torque generated by the salient pole effect of a permanent magnet synchronous motor to obtain a higher torque-current ratio. It is also called maximum torque current ratio control, and its control implementation block diagram is shown in FIG. 15, which is a schematic diagram of the system structure of the MTPA control system of the present invention. Among them, MTPA control is a control strategy adopted under non-weak magnetic field. Since the straight-axis inductance Ld of the salient pole motor is less than the cross-axis inductance Lq, the reluctance of the motor can be used when the motor is running below the rated speed. Torque to obtain a higher torque-current ratio. The key of this strategy is to set the correct current operating point, and the dynamic response of the system is realized by the optimized current inner loop control. The current current inner loop commonly has feedforward decoupling control, feedback decoupling control, and internal model decoupling control. And deviation decoupling control. Aiming at the problem that the system is under high acceleration and deceleration conditions, the d and q axis currents have serious dynamic coupling and affect the dynamic performance of the system. An optimized feedforward decoupling control strategy is used to achieve optimal control of the current inner loop. The MTPA control block diagram is shown in Figure 15. Among them, udf and uqf are the feedforward voltage of d axis and q axis respectively. Feed-forward decoupling is to add decoupling voltage terms at the output signals u _sd and u _sq of the current controller, respectively

with

So as to cancel the coupling effect between excitation and torque current. Among them, the MTPA control specifically includes the following steps: determining the q-axis current reference and the d-axis current reference according to the torque current curve; calculating the first difference between the q-axis current reference and the q-axis actual current and the d-axis current reference and The second difference value of the d-axis actual current; the d-axis voltage reference is obtained according to the first difference value through the first PI controller, and the q-axis voltage reference is obtained according to the second difference value through the second PI controller; the q-axis voltage is calculated The sum of the given and q-axis feedforward voltages gives the actual q-axis voltage reference, and the sum of the d-axis voltage reference and the d-axis feedforward voltage is calculated to get the actual d-axis voltage reference. As shown in Figure 15, firstly, the given d-axis current given id * and q-axis current given iq * are determined according to the input and torque current curve, and then the id * and d-axis actual current id are subtracted and sent to PI The controller subtracts iq * and the q-axis actual current iq and sends it to the PI controller. As shown in the figure, the two PI controllers will calculate d-axis voltage given ud and q-axis voltage given uq. Subsequently, the calculated d-axis voltage given ud is added to the d-axis feedforward voltage udf to obtain ud * as the actual output d-axis voltage given, and the calculated q-axis voltage given uq is added to the q-axis before The feed voltage uqf is given by uq * as the actual output q-axis voltage.

In particular, FIG. 16 is a schematic structural diagram of a system for front-end decoupling control of the present invention. As shown in Figure 16, assuming that the back EMF component has been cancelled, front-end decoupling control is required. Among them, according to the front-end structure control block diagram in FIG. 16, the voltage calculation equation of the front-end structure that can be written as a matrix is:

According to the above figure, it can be written as a matrix, so the feedforward decoupling voltage calculation equation is

Further, the voltage calculation equation of the front-end structure can be written in the form of matrix representation

Corresponding closed-loop transfer function matrix with feedforward decoupling can be obtained

17 is a schematic diagram of the system structure of the field weakening control of the present invention. Due to the limited capacity of the system converter, when the permanent magnet synchronous motor runs in steady state, the terminal voltage and stator current will be idle, and the voltage and current limit values cannot be exceeded. To further widen the speed regulation range, the field weakening control is adopted at the rated speed In the above, the permanent magnet synchronous motor enters the field weakening state, and the purpose of speed-up of the field weakening can be achieved by controlling the excitation current. Therefore, the control algorithm based on the above control strategy is used to calculate and obtain the current d-axis voltage given value and the current q-axis voltage given value, and further, obtain the current expected control according to the current d-axis voltage given value and the current q-axis voltage given value Phase angle. As shown in Figure 17, due to the converter capacity limitation, when the permanent magnet synchronous motor runs in steady state, the terminal voltage us and the stator current is limited, and cannot exceed the voltage and current limit values. To further widen the speed regulation range, use Field weakening control. The permanent magnet synchronous motor above the rated speed enters the field weakening state, and the purpose of speed-up of the field weakening can be achieved by controlling the excitation current; the current loop adopts the power angle control strategy. At this time, the voltage applied by the inverter on the motor is not controllable, only through The power angle β of the motor is controlled to adjust the excitation and torque of the motor. At this time, only the d-axis current of the motor is controlled. The output of the PI regulator controls the power angle to realize the control of the power angle above the fundamental frequency of the permanent magnet motor. Among them, Usmax and Ismax are voltage limit value and current limit value respectively, Δid is the change of excitation current in a given field weakening state, id_wk * and iq_wk * are given d-axis and q-axis current after field-weakening adjustment, uf is the amplitude of the feedforward voltage, and β is the power angle. Specifically, the field weakening control specifically includes the following steps: the PI controller calculates the d-axis current change amount in the given field weakening state according to the difference between the voltage limit value and the feedforward voltage amplitude; the d-axis current in the given field weakening state The sum of the amount of change and the d-axis current setting gives the d-axis current setting after the field weakening adjustment; the q-axis current setting after the field weakening adjustment is calculated according to the d-axis current setting and the torque formula; according to the q-axis through the PI controller The difference between the current setting and the q-axis actual current is the power angle β; the actual q-axis voltage setting and the actual d-axis voltage setting are calculated by the following formula;

U _d = U _s cos β

U _q = U _s cos β

Among them, Us is the voltage limit value, Ud is the actual d-axis voltage given, and Uq is the actual q-axis voltage given. Specifically, as shown in FIG. 17, in the field weakening control, it is necessary to first subtract the difference between the voltage limit value Us and the feedforward voltage amplitude uf, and obtain the excitation current under a given field weakening state through the PI controller The amount of change Δid, the sum of Δid and the d-axis current given as the d-axis current given id_wk * after the field weakening adjustment is fed into the torque formula, and the q-axis current given after the field-weakening adjustment is reversed according to the torque formula iq_wk *. Then, the difference between the q-axis current reference and the q-axis actual current iq is sent to the PI controller, and the PI controller obtains the power angle β. Finally, the actual q-axis voltage reference and the actual d-axis voltage reference are calculated according to the above formula As output. Optionally, in the embodiment shown in FIG. 17, when calculating the feedforward voltage amplitude uf, it is necessary to first obtain the feedforward Δid through the PI controller, and use the given sum of Δid and the d-axis current as id_wk *, and By inversely deducing iq_wk * according to the torque formula, and sending id_wk * and iq_wk * into the voltage equation to calculate the d-axis voltage given udf and the q-axis voltage given uqf, the formula

Calculate the feedforward voltage amplitude uf.

In addition, FIG. 18 is a schematic diagram of the trajectory of MTPA control and field weakening control in the full speed range of the present invention. In the control trajectory in the full speed range shown in FIG. 18, in the coordinate system with id and iq as coordinate axes, the OA segment is the MTPA control trajectory, and the AB and BC segments are the field weakening control trajectory; ωr1 is the rated speed, ωr2 is the highest speed. -ψf / Ld is the center of the voltage limit circle.

Further, FIG. 19 is a schematic diagram of switching control of MTPA control and field weakening control of the present invention. Figure 19 shows the block diagram of the switching between the two control strategies. Due to the smooth and reliable transition between the MTPA control strategy and the field weakening control strategy, when the output voltage of the inverter reaches the voltage limit circle, it switches to the weak In the magnetic control state, the voltage vector angle at the moment of switching is used as the initial phase angle β0 of the field weakening control; when switching from the field weakening control to the MTPA control, the voltage Usd and Usq are calculated from the power angle of the last beat. Among them, the saturation voltage is: Usat = 2 * Udc / pi.

Optionally, on the basis of the foregoing embodiment, an embodiment of the present invention also provides a modulation method for a high-power direct-drive permanent magnet synchronous motor in a megawatt direct-drive permanent magnet electric drive system for electric locomotives, by The modulation phase angle is calculated to achieve the actual control phase angle through PWM modulation.

Due to the high power of the traction converter of the high-power traction drive system, affected by the heat dissipation of the switching device and the switching loss, it needs to work at a lower switching frequency, usually not exceeding 1000 Hz. On the one hand, the highest switching frequency is generally It is about 100 Hz. On the other hand, when the output reaches the rated value, it works in the square wave mode. Therefore, in the entire speed range, the variation range of the carrier ratio is very large.

Therefore, this embodiment provides a multi-mode PWM modulation strategy, on the one hand, it can make full use of the allowable switching frequency of the inverter, and on the other hand, it can ensure a high DC voltage utilization rate after entering the field weakening control area. 20 is a schematic flowchart of a modulation method for a high-power direct-drive permanent magnet synchronous motor in a megawatt direct-drive permanent magnet electric drive system for an electric locomotive provided by the present invention; as shown in FIG. 20, the high power provided by this embodiment The control method of direct drive permanent magnet synchronous motor includes:

S2001: Obtain the frequency of the modulated wave of the high-power direct-drive permanent magnet synchronous motor to be modulated;

S2002: Determine the first modulation strategy according to the range of the frequency of the modulated wave and the second mapping relationship. The second mapping relationship includes a one-to-one correspondence between the frequency range of at least one modulated wave and the at least one modulation strategy.

S2003: Determine the PWM carrier frequency of a high-power direct-drive permanent magnet synchronous motor according to the first modulation strategy.

Wherein, optionally, the second mapping relationship at least includes: corresponding to the asynchronous modulation strategy when the frequency of the modulated wave is in the low-speed stage; corresponding to the synchronous modulation strategy of 60 degrees in the middle when the frequency of the modulated wave is greater than that in the low-speed stage; In the high-speed phase, it corresponds to the square wave modulation strategy.

Specifically, the multi-mode PWM modulation strategy is mainly composed of asynchronous SPWM modulation, synchronous SPWM modulation and square wave modulation. among them,

1. Asynchronous modulation strategy is adopted in the low-speed phase; when the asynchronous modulation has a large carrier ratio, the positive and negative half-cycle asymmetry caused by the asynchronous modulation mode has less influence, and the introduced low-order harmonics can be ignored. 2. When the speed increases, the middle 60-degree synchronous modulation strategy is adopted; as the motor frequency rises and the carrier ratio decreases, the impact of this low-order harmonic is getting larger and larger, and synchronous modulation PWM is used at this time. However, the conventional regular sampling synchronous modulation has a high content of low-order harmonics when the carrier ratio is relatively low, and the amplitude of the fundamental wave voltage obtained by sampling cannot meet the requirements of the command value, which is not conducive to entering the square wave. , So that the current has better harmonic characteristics and symmetry, and smoothly enter the square wave. 3. In the high-speed phase, square wave modulation is used; the traction inverter outputs a higher fundamental wave voltage to increase the maximum output torque of the traction motor. It will operate in the square wave working condition in the high-speed section, and the modulation method uses square wave modulation .

In this embodiment, the specific low speed and high speed in the process of acquiring the current modulation phase angle are both the angular speed of the rotor, and the specific division rule may be similar to the division rule in the prior art.

FIG. 21 is the relationship between the modulation angle and the modulation ratio under the intermediate 60 ° modulation mode provided by the present invention; FIG. 22 is a schematic diagram of the full speed range modulation strategy based on the intermediate 60 ° modulation provided by the present invention. As shown in Figure 22, the asynchronous modulation strategy is used in the low-speed phase; when the speed increases, the regular sampling synchronous modulation and the intermediate 60-degree synchronous modulation strategy with different carrier ratios are used; the high-speed phase uses square wave modulation. The switching process involved mainly includes the switching between asynchronous modulation to SVPWM synchronous modulation, the switching between synchronous modulation SVPWM and intermediate 60 ° modulation, and the internal 60 ° modulation. The main difficulty in switching is the switching between synchronous modulation SVPWM and intermediate 60 ° modulation. At a frequency division of 15, there are 15 carriers per fundamental cycle, and the phase of the fundamental wave corresponding to each carrier is 24 °, while at the mid-seventh modulation of 60 °, the phase of the fundamental wave corresponding to each carrier cycle is 20 ° . In the carrier-type PWM, you must wait until one carrier cycle ends before switching, so in order to ensure the continuity of the fundamental phase, the phase at the switching point must be a common multiple of the phase corresponding to each carrier cycle before and after switching, 20 ° and 24 ° The common multiple of is 120 °, which means that only three points can be switched in a cycle, namely 0 °, 120 °, and 240 °, and each corresponds to one of the points during the switching process. If the leakage inductance of the motor is small, it may cause a certain impact during the switching process, and the other two switching processes can achieve shockless switching. In addition, it should be noted that the abscissa in this embodiment is the frequency of the modulated wave obtained by the modulation algorithm in this embodiment. The ordinate is the PWM carrier frequency.

In particular, as shown in FIG. 21, the relationship between the modulation angle β and the modulation ratio at the middle 60 ° nineth frequency division, seventh frequency division, fifth frequency division, and third frequency division is shown. It shows that through the middle 60 ° modulation method in this embodiment, if the influence of the dead zone is not taken into account, it is possible to ensure that the actual output voltage and the reference value are completely coincident, with a very high voltage control accuracy. In addition, the characteristics of the intermediate 60 ° modulation adopted in this embodiment can be summarized as: (1) The intermediate 60 ° synchronous modulation can achieve symmetry between the three phases of the output voltage waveform when the number of pulses is not a multiple of 3, and each phase Positive and negative half cycle and 1/4 cycle symmetry, so that the motor line voltage and current only contain 6k ± 1 harmonic; (2) The switch angle under this modulation mode can be calculated online in real time, and the required calculation amount is very small . The implementation process has relatively low hardware requirements, and the pulse is relatively easy to send; (3) Through digital control, the middle 60 ° modulation can accurately output the required fundamental voltage, and the maximum output voltage under different pulse numbers does not consider the minimum pulse width Can be directly transferred to the square wave; (4) When the number of pulses in the middle 60 ° modulation is greater than 9, the current harmonics cannot be significantly improved. Different pulse numbers have consistent low-order current harmonic characteristics, resulting in low-order torque ripples with stable and relatively large ripple amplitudes under different pulse numbers and modulation ratios; (5) Intermediate 60 ° modulation The trajectories of the stator flux linkage of the motor are all hexagonal trajectories. The increase in the number of pulses only increases the number of voltage zero vectors in each sector, that is, the number of pauses of the stator flux linkage.

Optionally, in a specific implementation of the control method of the present invention, a method for detecting the initial position angle of the rotor of the high-power direct-drive permanent magnet synchronous motor in the main circuit is also provided to improve The reliability of the detection of the initial position angle of the rotor of the magnetic synchronous motor is to reduce the influence of the inaccurate detection of the initial position angle of the rotor on the performance of the vector control in the vector control of the permanent magnet synchronous motor.

Specifically, FIG. 23 is a schematic flowchart of Embodiment 1 of a method for detecting a rotor initial position angle of a high-power direct-drive permanent magnet synchronous motor provided by the present invention. The main body of the method for detecting the initial position angle of the rotor of the high-power direct-drive permanent magnet synchronous motor provided in this embodiment is the apparatus for detecting the initial position angle of the rotor of the high-power direct-drive permanent magnet synchronous motor provided by the present invention, for example, the device It is a TCU control device. As shown in FIG. 23, the method of this embodiment includes:

S2301: Inject a high-frequency voltage signal into the stator winding of the high-power direct-drive permanent magnet synchronous motor to be detected to obtain the three-phase stator winding current.

In order to make the technical solution in this embodiment more clear, first, several related coordinate systems involved in the present invention are introduced here.

Specifically, the coordinate system involved in the present invention includes a two-phase synchronous rotating coordinate system, a two-phase stationary coordinate system, and an expected two-phase synchronous coordinate system. Among them, FIG. 24 is a schematic diagram of the relationship between the two-phase synchronous rotating coordinate system, the two-phase stationary coordinate system, and the expected two-phase synchronous rotating coordinate system provided by the present invention. As shown in FIG. 1B, the αβ coordinate system is a two-phase stationary coordinate system, and the dq coordinate system is a two-phase synchronous rotating coordinate system.

The coordinate system is an expected two-phase synchronous rotating coordinate system.

Since the high-power direct-drive permanent magnet synchronous motor is in operation, there may be an error between the expected rotor position angle and the actual rotor position angle. Therefore, the estimated error of the rotor position angle is defined as

among them,

For the expected rotor position angle, θ is the actual rotor position angle, and Δθ is the rotor position angle estimation error.

Under the above expected two-phase synchronous rotating coordinate system, a high-frequency voltage signal is injected into the stator winding of a high-power direct-drive permanent magnet synchronous motor.

A possible implementation is to inject a high-frequency voltage signal as shown in the following formula into the expected two-phase synchronous rotating coordinate system:

Where U _mh is the amplitude of the high-frequency voltage signal, ω _h is the angular frequency of the high-frequency voltage signal, and t represents the time when the high-frequency voltage signal is injected.

It can be known from the above formula that the two components of the high-frequency voltage signal injected into the stator winding of the high-power direct-drive permanent magnet synchronous motor are linearly independent, and thus the inductance parameters of the high-power direct-drive permanent magnet synchronous motor can be obtained. Specifically, the inductance parameters of the high-power direct-drive permanent magnet synchronous motor can be obtained according to the mathematical model and related calculation methods of the high-power direct-drive permanent magnet synchronous motor established in the prior art.

After the high-frequency voltage signal is injected, the response current of the stator winding is obtained, and the response current is the three-phase stator winding current. In a possible implementation, the three-phase stator winding current can be obtained through a current sensor.

Among them, the three-phase stator winding current can be represented by i _a , i _b and i _c .

S2302. Obtain the d-axis target current and the q-axis target current in the expected two-phase synchronous rotating coordinate system according to the three-phase stator winding current.

It should be noted that both the d-axis target current and the q-axis target current are injected high-frequency voltage signals, and the corresponding current components are excited on the stator windings according to the structure of the high-power direct-drive permanent magnet synchronous motor and the magnetic saturation characteristics. The target current and the q-axis target current are both related to the estimation error of the rotor position angle. By performing signal processing on the d-axis target current and the q-axis target current, the initial rotor position angle can be obtained.

Therefore, according to the relationship between the expected two-phase synchronous rotating coordinate system and the two-phase stationary coordinate system, coordinate transformation is performed on the three-phase stator winding currents to obtain the d-axis target current and the q-axis target in the expected two-phase synchronous rotating coordinate system Current.

A possible implementation method is to first perform Clarke transformation on the three-phase stator winding currents i _a , i _b and i _c to obtain the α-axis current i _α and β-axis current i _{β in the} two-phase stationary coordinate system, , And then Park transform the α-axis current and β-axis current to obtain the d-axis target current

And q-axis target current

Further, the d-axis target current

And q-axis target current

As shown in the following formula:

Where L is the average inductance L = (L _d + L _q ) / 2, and ΔL is the half-differential inductance ΔL = (L _d -L _q ) / 2.

According to the above formula, the d-axis target current

And q-axis target current

Both are related to the rotor position angle estimation error Δθ.

S2303. Acquire the initial position angle of the rotor according to the d-axis target current and the q-axis target current.

Wherein, the above-mentioned initial position angle is the initial position angle compensated according to the magnetic pole polarity of the high-power direct-drive permanent magnet synchronous motor.

Specifically, according to the above formula, the q-axis target current

Contains the initial rotor position information, therefore, the q-axis target current can be signal processed to extract the initial rotor position angle.

The polarity information of the pole of the high-power direct-drive permanent magnet synchronous motor is related to the d-axis inductance. Therefore, the polarity information of the pole can be obtained according to the nonlinear magnetization characteristic of the d-axis inductance of the high-power direct-drive permanent magnet synchronous motor.

Further, the initial position angle of the rotor is compensated according to the polarity of the magnetic pole, thereby obtaining the compensated initial position angle, and the compensated initial position angle is determined as the initial position angle of the rotor.

In this embodiment, the high-frequency voltage signal is injected into the stator winding of the high-power direct-drive permanent magnet synchronous motor to be detected first to obtain the three-phase stator winding current, and then the expected two-phase synchronous rotating coordinate system is obtained according to the three-phase stator winding current D-axis target current and q-axis target current, further, the initial position angle of the rotor is obtained according to the d-axis target current and the q-axis target current, where the initial position angle is based on the pole polarity of the high-power direct-drive permanent magnet synchronous motor The initial position angle after compensation. The method provided by the present invention compensates for the initial position angle of the rotor according to the polarity of the magnetic pole by considering the influence of the magnetic pole of the high-power direct-drive permanent magnet synchronous motor. The obtained initial position angle of the rotor is more accurate and improves the initial position Reliability of angle detection. In addition, the method provided by the present invention can also obtain high-accuracy detection results under the condition that the rotor is stationary, and has a wide application range. In addition, the method provided by the present invention does not need to consider the parameters of the high-power direct-drive permanent magnet synchronous motor, and is easier to implement.

Based on the embodiment shown in FIG. 23, in some embodiments, S2303. Obtaining the initial position angle of the rotor according to the d-axis target current and the q-axis target current may be implemented in the following ways:

First, the first initial position angle of the rotor is obtained according to the q-axis target current.

A possible implementation method, when the rotor position angle estimation error Δθ is zero, the q-axis target current

Is zero, for the q-axis target current

Signal processing is performed to obtain the error input signal of the rotor position angle, and the initial position angle of the rotor is obtained according to the error input signal.

Further, the rotor pole compensation angle is obtained according to the d-axis target current.

The pole information of the high-power direct-drive permanent magnet synchronous motor is related to the d-axis inductance. Therefore, the polarity information of the magnetic pole can be obtained according to the nonlinear magnetization characteristic of the d-axis inductance of the high-power direct-drive permanent magnet synchronous motor.

Further, based on the first initial position angle and the magnetic pole compensation angle, the initial position angle of the rotor is obtained.

In this embodiment, the first initial position angle is compensated by using the magnetic pole compensation angle, and the compensated first initial position angle is determined as the initial position angle of the rotor.

Next, a specific implementation manner of acquiring the first initial position angle of the rotor according to the q-axis target current will be described.

FIG. 25 is a schematic flowchart of Embodiment 2 of a method for detecting a rotor initial position angle of a permanent magnet synchronous motor provided by the present invention. As shown in FIG. 25, obtaining the first initial position angle of the rotor according to the q-axis target current may include:

S2501: Perform low-pass filtering on the q-axis target current to obtain an error input signal.

Among them, the error input signal is an error signal related to the initial position angle of the rotor.

A possible implementation manner is to modulate the q-axis target current by using a modulation signal to obtain the modulated q-axis target current, and further, perform low-pass filtering on the modulated q-axis target current to obtain an error input signal.

Specifically, for the q-axis target current

Multiply the modulation signal 2sin (ω _h t) to obtain the modulated q-axis target current.

Among them, the modulated q-axis target current is expressed as

Further, the modulated q-axis target current is filtered by a low-pass filter to filter out the signal component of double frequency to obtain the error input signal f (Δθ), where,

Among them, LPF stands for low-pass filtering.

It can be known from the above formula that the error input signal includes the rotor position estimation error. In the process of low-pass filtering, consider the effect of filter phase delay on the extracted signal, and consider adding delay compensation during implementation to ensure that the high-frequency voltage injection phase is consistent with the estimated angle phase.

Further, when the rotor position estimation error is small enough, the error input signal after the limit equivalent linearization, namely:

S2502: Acquire the first initial position angle according to the error input signal.

In this step, the error input signal is used as the input of the PI regulator of the phase-locked loop. The PI regulator obtains the proportional deviation and integral deviation of the error input signal according to the input error signal. Further, according to the linear combination of the proportional deviation and integral deviation, the The first initial position angle.

Specifically, the first initial position angle can be obtained by the following formula:

Where s represents the Laplace operator, k _p is the coefficient of proportional term, and k _i is the coefficient of integral term;

Adjusting the proportional coefficient and integral coefficient of the PI regulator causes f (Δθ) to converge, and the output term of the PI regulator is the rotor's first initial position angle θ _first .

In this embodiment, the error input signal is obtained by modulating the q-axis target current and low-pass filtering, and further, a PI regulator is used to phase-lock and output the error input signal to obtain the first initial position angle.

Next, a specific implementation method for obtaining the rotor pole compensation angle according to the d-axis target current is introduced.

26 is a schematic flowchart of Embodiment 3 of a method for detecting a rotor initial position angle of a permanent magnet synchronous motor provided by the present invention. As shown in FIG. 26, obtaining the rotor pole compensation angle according to the d-axis target current may include:

S2601: Inject a plurality of voltage pulse signals with the same voltage amplitude and different angles into the permanent magnet synchronous motor to obtain the response current of each voltage pulse signal.

The poles of permanent magnet synchronous motors have nonlinear saturation characteristics. Specifically, a voltage pulse signal is injected into the d-axis of the permanent magnet synchronous motor. When the angle of the voltage pulse signal is closer to the N pole of the permanent magnet synchronous motor, the magnitude of the response current is larger; when the angle of the voltage pulse signal is farther away from the permanent magnet For the N pole of a synchronous motor, the smaller the magnitude of the response current. It should be noted that the d axis is the straight axis of the permanent magnet synchronous motor, and the q axis is the intersection axis of the permanent magnet synchronous motor.

Therefore, a plurality of voltage pulse signals with equal voltage amplitudes and different angles are injected into the permanent magnet synchronous motor to obtain the response current of each voltage pulse signal, thereby obtaining the variation law of the amplitude of the response current.

A possible implementation method is to inject a plurality of voltage pulse signals with a preset angle and equal amplitude into the permanent magnet synchronous motor, and to sample through the current sensor to obtain the response current of the multiple voltage pulses and further obtain the amplitude of the response current The changing law. For example, a permanent magnet synchronous motor is injected with a voltage pulse signal of equal amplitude every 5 °. It can be understood that the preset angle may also be smaller or larger, which is not limited in the present invention. It should be noted that the smaller the preset angle, the more response current data is obtained, and the accuracy of the change law of the amplitude of the response current is higher. The larger the preset angle, the response current data is obtained. The less the accuracy of the change law of the amplitude of the response current is, the more appropriate the preset angle can be selected according to the actual situation in the actual application process.

Another possible implementation method is to inject a plurality of voltage pulse signals of equal angle and equal amplitude into the permanent magnet synchronous motor, and sample through the current sensor to obtain the response current of the multiple voltage pulses and further obtain the response current The law of amplitude change.

S2602: Determine the magnetic pole compensation angle of the rotor according to multiple response currents.

Specifically, the pole compensation angle of the rotor is determined according to the magnitudes of multiple response currents.

When the difference between the angle of the injected voltage pulse signal and the first initial position angle meets the preset error range, and the amplitude of the response current of the voltage pulse signal is greater than the first value, it is determined that the rotor pole compensation angle is 0, where the first The value is the maximum value of the magnitude of multiple response currents. In other words, the d-axis direction is determined to be the magnetic pole N-pole direction.

When the difference between the angle of the injected voltage pulse signal and the first initial position angle satisfies the preset error range, and the amplitude of the response current of the voltage pulse signal is less than the second value, it is determined that the rotor pole compensation angle is π, where the second The value is the minimum value of the magnitude of multiple response currents. In other words, the d-axis direction is determined as the S-pole direction.

Accordingly, the initial position angle of the rotor is the sum of the first initial position angle and the pole compensation angle. Specifically, when the d-axis direction is determined as the N-pole direction, the initial position angle of the rotor is equal to the first initial position angle, and when the d-axis direction is determined as the S-pole direction, the initial position angle of the rotor is equal to the first initial position angle and the magnetic pole The sum of the compensation angle π.

In this embodiment, the accuracy of the identification of the magnetic pole polarity obtained based on the nonlinear saturation characteristics of the permanent magnet synchronous motor straight shaft inductance is high, and in the implementation process, it is not necessary to consider the influence of the motor parameters of the permanent magnet synchronous motor, reliability Higher and easier to implement.

Next, take a 1200kW permanent magnet synchronous motor as an example to explain some specific parameter settings during the implementation of the method of the present invention:

The inverter switching frequency is 500Hz, the motor rated power is 1200kW, the motor rated torque is 32606N.m, the rated voltage is 2150V, the rated current is 375A, the rated speed is 350r / min, the number of motor pole pairs is 7, the motor d-axis inductance Ld is 0.008771 H, the motor q-axis inductance Lq is 0.012732H.

The amplitude of the high-frequency voltage signal injected into the permanent magnet synchronous motor is 180V, the angular frequency of the high-frequency voltage signal is 200 Hz, and the inverter switching frequency is 500 Hz.

During the operation of the permanent magnet synchronous motor, the signal changes of multiple channels are collected. Among them, FIG. 27 is a schematic diagram of the signal changes of the multiple channels during the operation of the permanent magnet synchronous motor. As shown in Figure 27, the channels from top to bottom are: permanent magnet synchronous motor UV phase line voltage signal, permanent magnet synchronous motor U phase upper tube pulse signal, bus voltage signal, permanent magnet synchronous motor U phase current signal, permanent magnet Synchronous motor V-phase current signal.

Further, the method provided by the embodiment of the present invention is used to inject voltage pulse signals with the same voltage amplitude and different angles into the permanent magnet synchronous motor to obtain the response current corresponding to the voltage pulse signal. Among them, FIG. 28 is a schematic diagram of the response current variation rule. As shown in FIG. 28, when the angle of the injected voltage pulse signal is closer to the N pole of the permanent magnet synchronous motor, the magnitude of the response current is larger; when the angle of the injected voltage pulse signal The farther away from the N pole of the permanent magnet synchronous motor, the smaller the magnitude of the response current.

Further, the actual position angle of the rotor obtained by detecting the resolver is compared with the expected position angle of the rotor calculated according to the control algorithm. Through comparison of multiple sets of data, the calculation error is about ± 1.2 °, and the error is small.

Table 1

转子实际位置角Rotor actual position angle	转子预期位置角Rotor expected position angle	计算误差(弧度)Calculation error (radian)	计算误差(度)Calculation error (degree)
1.72571.7257	1.71451.7145	0.01120.0112	0.641712730.64171273
4.77374.7737	4.76944.7694	0.00430.0043	0.246371850.24637185
0.82680.8268	0.820.82	0.00680.0068	0.38961130.3896113

3.91783.9178	3.90263.9026	0.01520.0152	0.870895850.87089585
6.22646.2264	6.21876.2187	0.00770.0077	0.44117750.4411775
2.66912.6691	2.64652.6465	0.02260.0226	1.294884621.29488462
6.13296.1329	6.12156.1215	0.01140.0114	0.653171890.65317189
2.96522.9652	2.94892.9489	0.01630.0163	0.933921210.93392121
4.84284.8428	4.8414.841	0.00180.0018	0.10313240.1031324
0.08590.0859	0.08170.0817	0.00420.0042	0.240642270.24064227
0.89280.8928	0.87530.8753	0.01750.0175	1.002676141.00267614

Optionally, in a specific implementation of the control method of the present invention, a method for actually controlling the phase angle of the high-power direct-drive permanent-magnet synchronous motor in the main circuit is also provided in order to improve the high-power direct-drive permanent-magnet Synchronous motors actually control the accuracy of the phase angle.

Specifically, FIG. 29 is a schematic structural diagram of a control system of a high-power direct-drive permanent magnet synchronous motor corresponding to the control method of the high-power direct-drive permanent magnet synchronous motor provided by the present invention. As shown in FIG. The control system of the magnetic synchronous motor includes: high power direct drive permanent magnet synchronous motor, tractor, traction controller TCU, and resolver.

Among them, the control object of the control method of the high-power direct-drive permanent magnet synchronous motor provided by the present invention is the high-power direct-drive permanent magnet synchronous motor, wherein the high-power direct-drive permanent magnet synchronous motor includes a stator and a rotor.

The resolver is installed on the rotor of a high-power direct-drive permanent magnet synchronous motor, which is used to collect rotor signals and input the collected signals to the traction controller. In the present invention, the resolver is specifically used to detect the actual position of the rotor.

The dragging machine is connected with the tested high-power direct-drive permanent magnet synchronous motor, which is used to drive the high-power direct-drive permanent magnet synchronous motor.

The traction controller is connected to a high-power direct-drive permanent magnet synchronous motor and is used to control the high-power direct-drive permanent magnet synchronous motor. In the present invention, the traction controller is used to implement a speed-based segmented vector control strategy for high-power direct-drive permanent magnet synchronous motors, wherein the speed-based segmented vector control strategy will be described in detail in subsequent embodiments. Specifically, the traction controller has functions of a control algorithm and a modulation algorithm, and functions of phase angle adjustment and speed detection.

Optionally, the traction controller in the present invention includes a control algorithm unit, a modulation algorithm unit, a phase angle regulator, and a speed detector. Among them, the control algorithm unit is used to obtain the expected control phase angle; the modulation algorithm unit is used to obtain the modulated phase angle, and then the actual control phase angle is realized by PWM modulation; the phase angle regulator is used to realize the expected control phase angle and the actual control phase angle Always keep the same; the speed detector is used to obtain the angular velocity of the rotor. It should be noted that the above-mentioned control algorithm unit, modulation algorithm unit, phase angle regulator, and speed detector can be either software modules or physical modules, which are not limited by the present invention.

In the following embodiments, the control method of the high-power direct-drive permanent magnet synchronous motor provided by the present invention is implemented by using a traction controller as an executive body.

FIG. 30 is a first schematic flowchart of a control method for a high-power direct-drive permanent magnet synchronous motor provided by the present invention. The method shown in FIG. 30 is executed by a traction controller, which can be implemented by any software and / or hardware . As shown in FIG. 30, the control method of the high-power direct-drive permanent magnet synchronous motor provided by this embodiment includes:

S3001: Obtain the compensation phase angle of the rotor of the high-power direct-drive permanent magnet synchronous motor according to the control interruption period, the modulation carrier period, and the current angular velocity of the rotor of the high-power direct-drive permanent magnet synchronous motor.

The compensation phase angle of the rotor of the high-power direct-drive permanent magnet synchronous motor obtained in this embodiment is the offline compensation phase angle, that is, if the components in the control system of the high-power direct-drive permanent magnet synchronous motor are acquiring the compensated phase angle and operating normally When the setting of the parameter remains unchanged, the compensation phase angle obtained offline can be applied to the control system of the running high-power direct-drive permanent magnet synchronous motor. It is conceivable that when the settings of various components in the control system of the high-power direct-drive permanent magnet synchronous motor are changed, the new setting phase parameters can be obtained using the changed setting parameters.

Specifically, the traction controller may use a control algorithm to process the voltage signal collected by the resolver to obtain the expected phase angle. Specifically, the traction controller may control the control algorithm unit to process the voltage signal collected by the resolver to obtain Expected phase angle. Among them, the sampling period of the resolver can be the same as the control interruption period of the control algorithm.

Exemplarily, the resolver samples at time t1 and inputs the collected voltage signal to the traction controller. The control algorithm unit of the traction controller processes the voltage signal collected by the resolver at time t1, obtains the expected phase angle, and updates it at an indefinite time between the beginning of the next control interruption period and the end of the next control interruption period. That is, the expected phase angle is output to the modulation algorithm unit. In this process, the rotor is still rotating continuously, and the control algorithm interruption delay will be generated relative to the sampling time of the resolver. Further, according to the length of the interruption delay of the control algorithm and the angular velocity of the rotor, the error phase angle of the rotor in the process of the control algorithm is obtained.

Preferably, the control algorithm delay is half a control interrupt period.

The traction controller obtains the expected phase angle and uses a modulation algorithm to modulate and output the expected phase angle. Specifically, the modulation algorithm unit of the traction controller uses the modulation algorithm to modulate the expected phase angle and output PWM pulses. The modulation sampling in this embodiment has periodicity, that is, the traction controller periodically acquires the expected phase angle and performs modulation processing. Exemplarily, in this embodiment, the modulation carrier is a triangular PWM carrier, and the modulation sampling adopts an asymmetric regular sampling method, that is, sampling at the position of the symmetry axis of the vertex of each triangular PWM carrier cycle, and at the bottom of the triangular PWM carrier cycle The point symmetry axis is sampled, that is, sampled twice per modulated carrier cycle. At the beginning and at the middle of each modulated carrier cycle, the sampling of this PWM carrier cycle is performed, and the PWM command of this cycle is updated at the same time. The interruption of modulation algorithm in double sampling mode is divided into sampling, modulation calculation, PWM update and PWM output process.

Exemplarily, the traction controller obtains the expected phase angle at time t2, performs PWM modulation processing, and generates PWM pulses. After that, the PWM pulse is usually output when the carrier cycle count value is equal to the PWM comparison count value calculated by modulation. In the above process, the rotor is still rotating continuously, therefore, the modulation update delay is caused. Preferably, the modulation update delay is half a modulation carrier period;

In addition, after the PWM calculation value is updated, the continuous pulse counting method of the timer is generally used to output the PWM pulse, and the output delay will also be caused during the output. Preferably, the output delay is 1/4 modulated carrier period.

According to the modulation update delay and output delay obtained in the modulation algorithm and the current angular velocity of the rotor, the error phase angle of the rotor in the process of the modulation algorithm can be obtained.

In addition, a delay is also generated during the process of sampling and signal transmission of the rotor by the resolver, which is referred to as resolver sampling and transmission delay. Specifically, in this implementation, the error phase angle corresponding to the sampling and transmission delay of the resolver is obtained according to the multiple d-axis voltages and multiple q-axis voltages in the current angular velocity and the preset angular velocity range of the high-power direct-drive permanent magnet synchronous motor rotor .

Next, the preset angular velocity range will be described in detail.

Because the high-power direct-drive permanent magnet synchronous motor drive system in this application uses a speed-based segmented vector control strategy, the segmented vector control strategy includes maximum torque-current ratio control in the low-speed region and field weakening control in the high-speed region . Therefore, the preset angular speed range in this embodiment may be a speed range where the traction controller determines that the high-power direct-drive permanent magnet synchronous motor does not enter the field weakening control stage and operates stably. Among them, according to the traction characteristics of the high-power direct-drive permanent magnet synchronous motor, the speed corresponding to the speed point when entering the constant voltage stage, the operating speed when the voltage reaches the maximum value, is the maximum stable operating speed without entering the field weakening control stage, which is the pre Set the maximum value of the angular velocity range.

Obtain the d-axis voltage and the q-axis voltage corresponding to each preset angular velocity in the preset angular velocity range, and obtain each preset angular velocity according to the d-axis voltage and the q-axis voltage corresponding to each preset angular velocity Corresponding error phase angle, then establish a curve with the preset angular velocity as the abscissa and the error phase angle as the ordinate, and determine the slope corresponding to the curve as the error coefficient; further, according to the angular velocity of the rotor and the error coefficient corresponding to the angular velocity Obtain the error phase angle, which is the error phase angle caused by the resolver sampling and transmission delay.

Optionally, the sum of the error phase angles corresponding to the above control algorithm delay, modulation algorithm delay, and resolver acquisition and transmission delay respectively is the compensation phase angle of the rotor of the high-power direct-drive permanent magnet synchronous motor.

It should also be added that in a two-phase synchronous rotating (d, q) coordinate system, the magnetic field generated by the rotor poles corresponds to the stator magnetic field as the d-axis, and the 90-degree counterclockwise rotation is the q-axis.

S3002. Acquire the current actual control phase angle according to the compensated phase angle.

The compensated phase angle obtained in step S3001 is an offline compensated phase angle, which is applied to the running high-power direct-drive permanent magnet synchronous motor.

Therefore, the current actual control phase angle obtained in this step is the actual control phase angle after the offline correction of the rotor position angle of the high-power direct-drive permanent magnet synchronous motor is performed using the compensated phase angle obtained in step S3001.

S3003. Acquire the current expected control phase angle according to the current d-axis voltage given value and the current q-axis voltage given value.

The current voltage given value may include a current d-axis voltage given value and a current q-axis voltage given value. In this embodiment, according to the speed-based segmented vector control strategy adopted by the high-power direct-drive permanent magnet synchronous motor and the corresponding control algorithm, the current d-axis voltage given value and the current q-axis voltage given value are calculated and obtained, further, Obtain the current expected control phase angle according to the current d-axis voltage given value and the current q-axis voltage given value.

S3004. Perform online correction on the current actual control phase angle according to the proportional deviation and integral deviation of the current expected control phase angle and the current actual control phase angle.

Since the current expected control phase angle and the current actual control phase angle may be deviated by the control algorithm, the modulation algorithm, and the delay in the acquisition and transmission process of the resolver, there may be a deviation between the current expected control phase angle and the current actual control phase angle. The current actual control phase angle is corrected.

In this step, the linear combination of the proportional deviation between the current expected control phase angle and the current actual control phase angle and the integral deviation between the current expected control phase angle and the current actual control phase angle is used as the correction term to perform online correction on the current actual control phase angle .

This embodiment provides a control method for a high-power direct-drive permanent magnet synchronous motor. The method includes: obtaining a high-power direct-drive according to a control interruption period, a modulated carrier period, and a current angular velocity of a rotor of the high-power direct-drive permanent magnet synchronous motor The compensation phase angle of the rotor of the permanent magnet synchronous motor; according to the compensation phase angle, the current actual control phase angle is obtained; according to the current d-axis voltage given value and the current q-axis voltage given value, the current expected control phase angle is obtained; further, according to The proportional deviation and integral deviation of the current expected control phase angle and the current actual control phase angle are corrected online on the current actual control phase angle. The invention corrects the actual control phase angle by taking into account the time delay corresponding to the control interruption, the time delay corresponding to the carrier modulation, and the error phase angle caused by the corresponding time delay during the process of sampling and transmitting the rotor signal of the resolver, Ensure that the actual control phase angle and the expected control phase angle are always consistent, and the accuracy of the actual control phase angle is improved.

FIG. 31 is a schematic flowchart of Embodiment 2 of a control method for a high-power direct-drive permanent magnet synchronous motor provided by the present invention. As shown in FIG. 31, on the basis of the embodiment shown in FIG. 30, step S3001 may include:

S3101: Acquire the first sub-compensated phase angle according to the control interruption period and the current angular velocity of the rotor of the high-power direct-drive permanent magnet synchronous motor.

In order to make the control method provided in this embodiment more clear, here, the control interruption involved in the present application will be described in detail. 32 is a schematic diagram of control interruption of the control algorithm provided by the present invention. As shown in Fig. 32, the control interruption is divided into the processes of sampling, control calculation, and control variable update. The resolver samples the rotor signal and inputs the collected voltage signal to the traction controller at time t1. The traction controller performs control calculation on the received voltage signal, T _ctrl is a control interruption cycle of the control algorithm, the control calculation is completed at t1 + T _ctrl time, and then begins at the next control interruption cycle (time t1 + T _ctrl ) to end (Time t1 + 2T _ctrl ) At the indefinite time within this period, the control variable calculated by the control is output to the modulation algorithm unit.

In this process, the rotor is still rotating continuously, and the control algorithm interruption delay will be generated relative to the time when the control calculation is completed. In this embodiment, the first phase angle delay corresponding to the first sub-compensated phase angle is obtained according to the control interruption period of the control algorithm, where A is the control interruption delay coefficient and the value range is (0-1). Preferably, A = 0.5.

Therefore, the first phase angle delay _Δt1 can be expressed as follows:

_{_{Δ t1 = A · T ctrl ≈0.5T}} ctrl

Further, according to the first phase angle delay and the current angular velocity of the rotor of the high-power direct-drive permanent magnet synchronous motor, the first sub-compensated phase angle is obtained, and the first sub-compensated phase angle is the error phase angle corresponding to the control algorithm interrupt delay .

Specifically, the first sub-compensation phase angle θ _cmps1 can be expressed as follows:

θ _cmps1 = Δ _t1 · ω

Where ω is the current angular velocity of the rotor of a high-power direct-drive permanent magnet synchronous motor.

S3102. Acquire the second sub-compensated phase angle according to the modulated carrier cycle and the current angular velocity of the rotor of the high-power direct-drive permanent magnet synchronous motor.

Exemplarily, taking the modulation carrier in this embodiment as a triangular PWM carrier as an example for illustration, in order to improve the dynamic response of the control system of a high-power direct-drive permanent magnet synchronous motor, the modulation algorithm uses an asymmetric regular sampling method, That is, sampling at the position of the symmetrical axis of the vertex of each triangular PWM carrier cycle, and sampling at the position of the symmetrical axis of the bottom point of the triangular PWM carrier cycle, that is, sampling twice per modulated carrier cycle. At the beginning and at the middle of each modulated carrier cycle, the sampling of the PWM carrier cycle is performed, and the PWM command of this cycle is updated at the same time. The interruption of modulation algorithm in double sampling mode is divided into sampling, modulation calculation, PWM update and PWM output process.

Among them, FIG. 33 is a schematic diagram of an interruption cycle of a modulation algorithm provided by the present invention. As shown in Figure 33, the traction controller performs modulation sampling at time t, and obtains the control variables calculated by the control algorithm. Specifically, the control variable obtained by the traction controller is the expected phase angle, and the modulation algorithm calculation is completed at t + 0.5T _PWM time, and the PWM comparison count value update and the expected control phase angle sampling for the next modulation cycle are started. When the PWM carrier cycle count value is equal to the PWM comparison count value calculated by the modulation, a PWM pulse is output, and T _PWM is the PWM modulated carrier cycle.

In this process, the rotor is still rotating continuously. Compared with the moment when the modulation calculation is completed, the modulation algorithm interruption delay will be generated, which is the third phase angle delay B · T _PWM , where B is the modulation algorithm interruption延 efficient。 Extension coefficient. Optionally, B = 0.5.

After the PWM comparison calculation value is updated, the timer's continuous up and down counting method is generally used to output the PWM pulse. In this process, the PWM pulse output delay is generated. The PWM pulse output delay is C · T _PWM , which is the second phase angle Delay. Among them, C is the PWM pulse output delay coefficient, the value range is (0-0.5). Optionally, C = 0.25.

Specifically, the delay Δ _t2 in the process of modulation calculation and PWM pulse output can be shown as the following formula:

Δ _t2 = B · T _PWM + C · T _PWM ≈0.75T _PWM

Further, according to the second phase angle delay, the third phase angle delay and the current angular velocity of the rotor of the high-power direct-drive permanent magnet synchronous motor, the second sub-compensated phase angle is obtained, and the second sub-compensated phase angle is the modulation algorithm time The phase angle corresponding to the error.

Specifically, the second sub-compensation phase angle θ _cmps2 can be expressed as follows:

θ _cmps2 = Δ _t2 · ω

S3103. Acquire a third sub-compensated phase angle according to the current angular velocity of the rotor of the high-power direct-drive permanent magnet synchronous motor.

The third sub-compensation phase angle is the error phase angle corresponding to the resolver sampling and transmission delay. Obtain the d-axis voltage and q-axis voltage corresponding to each preset angular velocity in the range of stable operating angular velocity, and obtain the corresponding to each preset angular velocity according to the d-axis voltage and q-axis voltage corresponding to each preset angular velocity Error phase angle, and then establish a curve with the preset angular velocity as the abscissa and the error phase angle as the ordinate, and determine the slope corresponding to the curve as the error coefficient; further, obtain it according to the angular velocity of the rotor and the error coefficient corresponding to the angular velocity The error phase angle, which is the error phase angle caused by the resolver sampling and transmission delay.

It should also be added that in a two-phase synchronously rotating (d, q) coordinate system, the magnetic field generated by the rotor poles corresponds to the stator magnetic field as the d-axis, and counterclockwise rotation by 90 degrees is the q-axis.

S3104. Acquire the compensation phase angle of the high-power direct-drive permanent magnet synchronous motor according to the first sub-compensation phase angle, the second sub-compensation phase angle, and the third sub-compensation phase angle.

Optionally, the sum of the first compensation phase angle, the second compensation phase angle, and the third compensation phase angle is the compensation phase angle of the high-power direct-drive permanent magnet synchronous motor.

S3105. Obtain the current actual control phase angle according to the compensated phase angle.

First, the current position phase angle of the rotor of the high-power direct-drive permanent magnet synchronous motor is obtained, and then the actual position phase angle of the rotor is obtained according to the current position phase angle, the initial position phase angle of the rotor, and the compensation phase angle. Further, according to the actual rotor The position phase angle and the current modulation phase angle are used to obtain the current actual control phase angle, where the modulation phase angle is calculated by using a modulation algorithm and according to the d-axis voltage given value and the current q-axis voltage given value.

Specifically, the actual position phase angle of the rotor is obtained according to the current position phase angle of the rotor and the initial position phase angle of the rotor, and further, the above-mentioned compensated phase angle is used to modify the rotor position angle of the high-power direct-drive permanent magnet synchronous motor offline, thereby Take the corrected actual position phase angle as the rotor actual position phase angle. After that, the difference between the actual position phase angle of the rotor and the current modulation phase angle is determined as the current actual control phase angle.

In a possible implementation, the modulation algorithm unit adopts a multi-mode PWM modulation strategy. On the one hand, the allowable switching frequency of the inverter can be fully utilized, and on the other hand, a high DC voltage utilization rate can be ensured after entering the field weakening control zone. Specifically, the multi-mode PWM modulation strategy is mainly composed of asynchronous SPWM modulation, regular sampling synchronous SPWM modulation and square wave modulation.

Among them, Figure 34 is a schematic diagram of the multi-mode PWM modulation strategy. As shown in Figure 34, the asynchronous modulation strategy is used in the low speed stage; when the speed increases, the regular sampling synchronous modulation with different carrier ratios and the intermediate 60-degree synchronous modulation strategy are used; The high-speed phase uses square wave modulation. The abscissa is the frequency of the modulated wave obtained by the modulation algorithm in this embodiment. The ordinate is the PWM carrier frequency.

S3106: Obtain the current expected control phase angle according to the current d-axis voltage given value and the current q-axis voltage given value.

Specifically, the high-power direct-drive permanent magnet synchronous motor in this embodiment uses a speed-based segmented vector control strategy to complete the current closed-loop control. The control strategy includes: maximum torque current ratio (MTPA) control in the low-speed area and high-speed area Field weakening control.

Below the rated speed, MTPA control is used, that is, a control method that uses the reluctance torque generated by the salient pole effect of a permanent magnet synchronous motor to obtain a higher torque-current ratio. Due to the limited capacity of the system converter, when the permanent magnet synchronous motor runs in steady state, the terminal voltage and stator current will be idle, and the voltage and current limit values cannot be exceeded. To further widen the speed regulation range, the field weakening control is adopted at the rated speed In the above, the permanent magnet synchronous motor enters the field weakening state, and the purpose of speed-up of the field weakening can be achieved by controlling the excitation current.

Therefore, the control algorithm based on the above control strategy is used to calculate and obtain the current d-axis voltage given value and the current q-axis voltage given value, and further, obtain the current expected control according to the current d-axis voltage given value and the current q-axis voltage given value Phase angle.

Specifically, it can be calculated according to the following formula:

Among them, θ _ctrl represents the expected control phase angle,

Represents the given value of q-axis voltage,

It indicates the given value of d-axis voltage.

S3107. According to the proportional deviation and integral deviation of the current expected control phase angle and the current actual control phase angle, the current actual control phase angle is corrected online.

A possible implementation method: first, obtain the proportional deviation and integral deviation according to the current expected control phase angle and the current actual control phase angle, and then obtain the correction item of the current actual control phase angle according to the linear combination of the proportional deviation and the integral deviation, Further, this correction item is used to perform online correction on the current actual control phase angle.

Optionally, the following formula is used to obtain the correction item:

Among them, k _p and k _i are correction terms, θ _ctrl is the current expected phase angle, θ _PWM is the current actual phase angle, and f _Δ is the fundamental frequency compensation term, which is a known quantity.

After the traction controller obtains the correction items k _p and k _i , the online adjustment of the correction items enables the current actual control phase angle to track the expected control phase angle quickly and error-free, thereby realizing the online correction of the actual control phase angle.

In this step, the closed-loop PI control is adopted for the control of the phase angle, which can realize the control of the control phase angle accurately and without static error, thereby improving the control performance.

In this embodiment, by considering the control algorithm, modulation algorithm, and time delay caused by the acquisition and transmission of the resolver, and according to the proportional deviation and integral deviation of the actual control phase angle and the expected control phase angle, the current actual control phase angle Online correction is performed to make the actual control phase angle consistent with the expected control phase angle, which improves the accuracy of the actual control phase angle, reduces the probability of operating failures of high-power direct-drive permanent magnet synchronous motors, and thus improves the high-power direct drive Control performance of permanent magnet synchronous motor traction system.

FIG. 35 is a third schematic flowchart of a control method of a high-power direct-drive permanent magnet synchronous motor provided by the present invention. As shown in FIG. 35, on the basis of the embodiment implemented in FIG. 31, optionally, the following steps are included before step S3103:

S3501: According to the vector control strategy of the high-power direct-drive permanent magnet synchronous motor, obtain the stable operating angular velocity range of the high-power direct-drive permanent magnet synchronous motor.

In this embodiment, on the basis of the above-mentioned speed-based segmented vector control strategy, the stable operating angular velocity range of the high-power direct-drive permanent magnet synchronous motor is first obtained, that is, the high-power direct-drive permanent magnet synchronous motor is not weak. The speed range of the magnetic control stage and stable operation, where the speed point corresponding to the constant voltage stage is reached and the voltage reaches the maximum value, which is the highest stable operating speed without entering the field weakening control stage.

S3502. Acquire a plurality of first d-axis currents, a plurality of first q-axis currents, and each of the first d-axis currents in the range of the stable operating angular velocity according to the d-axis current given value and the q-axis current given value The d-axis voltage corresponding to the current and the q-axis voltage corresponding to each of the first q-axis currents.

A possible implementation manner is that, according to a preset angular velocity interval, when the rotor of the high-power direct-drive permanent magnet synchronous motor is within the stable operating angular velocity range, a plurality of first corresponding to the preset angular velocity interval Preset angular velocity;

When the given value of the d-axis current and the d-axis current corresponding to each first preset angular velocity meet the preset error threshold, and the given values of the q-axis current and the q-axis current corresponding to each first preset angular velocity satisfy the preset error At the threshold, the d-axis current corresponding to each first preset angular velocity is determined as the first d-axis current, and the q-axis current corresponding to each first preset angular velocity is determined as the first q-axis current;

The d-axis voltage corresponding to each first d-axis current is obtained according to each first d-axis current, and the q-axis voltage corresponding to each first q-axis current is obtained according to each first q-axis current.

In this embodiment, each first d-axis current and each first q-axis current acquired by the traction controller are the d-axis current and q-axis current of a high-power direct-drive permanent magnet synchronous motor in a steady state.

Under steady-state conditions, the differential terms of high-power direct-drive permanent magnet synchronous motors are ignored. Therefore, the steady-state equation of high-power direct-drive permanent magnet synchronous motors can be expressed as the following formula:

Where u _d is the d-axis voltage corresponding to any first preset angular velocity, u _q is the q-axis voltage corresponding to any first preset angular velocity, R _s is the resistance of the rotor, and L _q is any first preset D-axis inductance corresponding to angular velocity, L _d is the q-axis inductance corresponding to any first preset angular velocity, i _d is the first d-axis current corresponding to the d-axis voltage, and i _q is the first q-axis current corresponding to the q-axis voltage , Ψ _f is the back electromotive force of the permanent magnet flux linkage.

It can be seen from the above steady-state equation of the high-power direct-drive permanent magnet synchronous motor that when the d and q-axis currents of the high-power direct-drive permanent magnet synchronous motor are both 0, the d-axis voltage at this time is 0 and the q-axis voltage is all Generated by the back electromotive force of the permanent magnet flux linkage.

Among them, FIG. 36A is a schematic diagram in which the theoretical coordinate system and the actual coordinate system completely coincide, FIG. 36B is a schematic diagram in which the actual coordinate system leads the theoretical coordinate system, and FIG. 36C is a schematic diagram in which the actual coordinate system lags the theoretical coordinate system.

As shown in Figure 36A-36C, first define the dq coordinate system used by the control algorithm as the theoretical dq coordinate system, and the dq coordinate system used by the modulation algorithm to actually output the PWM pulse is the actual

Coordinate System. When the rotor position is accurate and ideal, the theoretical dq coordinate system and actual

The coordinate systems are completely coincident, u _d is equal to 0, u _q is equal to ωψ _f , as shown in Fig. 36A; when the rotor position is advanced, the actual

_Leading theory of the coordinate system dq The coordinate system has a certain angle θ _cmps3 , u _d is a positive value, u _q is a positive value, as shown in FIG. 36B;

The coordinate system lags behind the theoretical dq coordinate system at a certain angle θ _cmps3 , u _d is a negative value, and u _d is a positive value, as shown in FIG. 36C.

Correspondingly, step S3103 can be implemented in the following manner:

S3503. Acquire the transmission error phase angle corresponding to each first angular velocity according to the d-axis voltage corresponding to each first d-axis current and the q-axis voltage corresponding to each first q-axis current.

In this embodiment, the d-axis voltage corresponding to each first d-axis current and the q-axis voltage corresponding to each first q-axis current are used to obtain the transmission error phase angle corresponding to each first preset angular velocity. The specific phase angle θ _{Δ of} transmission error can be obtained by the following formula:

θ _Δ = tan ^-1 (u _d / u _q )

S3504. Acquire the third sub-compensated phase angle according to the transmission error phase angle corresponding to each first angular speed and the current angular speed of the rotor of the high-power direct-drive permanent magnet synchronous motor.

Taking the first preset angular velocity as the abscissa and the transmission error phase angle as the ordinate, the transmission error phase angle coefficient k can be obtained by the product of the transmission error phase angle coefficient and the current angular velocity of the rotor of the high-power direct-drive permanent magnet synchronous motor The third sub-compensated phase angle can be obtained. The specific sub-compensation phase angle θ _cmps3 can be obtained as follows:

θ _cmps3 = k · ω

In this embodiment, according to the vector control strategy of the high-power direct-drive permanent magnet synchronous motor, the stable operating angular velocity range of the high-power direct-drive permanent magnet synchronous motor is obtained, and according to the d-axis current given value and the q-axis current given value, the A plurality of first d-axis currents, a plurality of first q-axis currents, a d-axis voltage corresponding to each of the first d-axis currents, and a corresponding q-axis voltage, based on the d-axis voltage corresponding to each first d-axis current and the q-axis voltage corresponding to each first q-axis current, obtaining the transmission error phase angle corresponding to each first angular velocity, and according to each first angular velocity The corresponding transmission error phase angle and the current angular velocity of the rotor of the high-power direct-drive permanent magnet synchronous motor obtain the third sub-compensated phase angle. By pre-acquiring the transmission error phase angle corresponding to multiple first angular velocities in the stable operating speed range, and then quickly obtaining the third sub-compensated phase angle according to the current angular velocity of the rotor of the high-power direct-drive permanent magnet synchronous motor, and adopting the third sub The compensated phase angle accurately corrects the actual control phase angle online, which improves the efficiency of online correction.

Optionally, in a specific implementation of the control method of the present invention, a method for adhering control of a high-power direct-drive permanent magnet synchronous motor in the main circuit is also provided to reduce the idling and coasting degrees in a timely manner and effectively improve the adhesion The utilization rate makes the traction of the locomotive stable, reduces the abnormal load of the wheel set, and reduces the wheel scraping and peeling damage.

Wherein, when the adhesion control method provided in this embodiment is applied to an electric locomotive as shown in FIG. 1, adhesion control is performed by at least four high-power direct-drive permanent magnet synchronous motors on the electric locomotive; the at least four large The power direct-drive permanent magnet synchronous motor includes: a first motor, a second motor, a third motor, and a fourth motor.

Optionally, in a possible implementation manner of this embodiment, six high-power direct-drive permanent magnet synchronous motors are provided on the motor locomotive, and the two direct-drive permanent magnet motor locomotives as shown in the foregoing embodiments are changed The main circuit of the converter controls six high-power direct-drive permanent magnet synchronous motors respectively. The four high-power direct-drive permanent magnet synchronous motors involved in the calculation in the control method of this embodiment may be any four of the six high-power direct-drive permanent magnet synchronous motors of the electric locomotive, and the first motor and the second motor are The shaft motor of the first bogie provided on the electric locomotive, and the third motor and the fourth motor are shaft motors provided on the second bogie of the electric locomotive.

FIG. 37 is a flowchart of an embodiment of the adhesion control method provided by the present invention. The method provided in this embodiment can be applied to a direct drive permanent magnet traction system. As shown in FIG. 37, the method provided in this embodiment may include:

S3701: Collect the rotor frequencies of the first motor, the second motor, the third motor, and the fourth motor to obtain the real-time torque of the first motor. The first motor and the second motor are the axle motors of the first bogie and the third motor And the fourth motor is a shaft motor of the second bogie, and the first bogie is adjacent to the second bogie.

The four motors in this embodiment are located on adjacent bogies. The operating conditions of the locomotive can be determined according to the real-time torque of the first motor. The rotor frequencies of the first motor, the second motor, the third motor, and the fourth motor may be collected according to a preset sampling period or a preset sampling frequency.

S3702: Determine the rotor frequency difference and the rotor frequency differential value of the first motor according to the collected rotor frequencies of the multiple motors.

Optionally, in this embodiment, the smallest rotor frequency among the first motor, the second motor, the third motor, and the fourth motor is used as the rotor frequency reference. The rotor frequency difference of the first electric machine is the difference between the rotor frequency of the first electric machine and the rotor frequency reference.

Optionally, the differential value of the rotor frequency of the first motor in this embodiment may be the difference between the rotor frequency of the first motor at the current sampling time and the rotor frequency of the first motor at the previous sampling time divided by the sampling time interval.

S3703. Determine the torque reduction amount according to the rotor frequency difference, the rotor frequency differential value, and the real-time torque of the first motor.

According to the rotor frequency difference and the rotor frequency differential value of the first motor, it can be quickly and accurately determined whether the locomotive is in the idling state. Once the locomotive coasting occurs, the torque reduction amount can be determined according to the rotor frequency difference, rotor frequency differential value and real-time torque of the first motor. The torque reduction amount is used to indicate the amount of torque that the first motor needs to be unloaded.

S3704. Adjust the torque of the first motor according to the amount of torque reduction.

The torque corresponding to the torque reduction amount of the first motor is unloaded to eliminate the idling phenomenon.

The adhesion control method provided in this embodiment collects the rotor frequencies of the first motor, the second motor, the third motor, and the fourth motor on adjacent bogies, and the real-time torque of the first motor, according to the collected Rotor frequency of multiple motors, determine the rotor frequency difference and rotor frequency differential value of the first motor, determine the torque reduction amount based on the rotor frequency difference, rotor frequency differential value and real-time torque of the first motor, and reduce the torque according to the torque To adjust the torque of the first motor. The torque reduction is determined according to the rotor frequency for adhesion control, with low noise and strong resistance to external interference; according to the rotor frequency difference and rotor frequency differential value, it can quickly and accurately determine whether the locomotive is in the idling state, and reduce the idling and coasting degree in time, effectively Improve the adhesion utilization rate, make the traction of the locomotive stable, reduce the abnormal load of the wheel set, and reduce the wheel scraping and peeling damage.

Optionally, in order to further improve the adhesion utilization rate, on the basis of the foregoing embodiment, the method provided in this embodiment may further include:

According to the rotor frequency difference, the rotor frequency differential value and the real-time torque of the first motor, a sanding control signal is generated, and the sanding control signal is used to indicate whether to perform sanding operation. Sanding can increase the adhesion coefficient between the wheels and rails, and reduce the idling and sliding of the locomotive. If it is determined according to the rotor frequency difference, the rotor frequency differential value and the real-time torque of the first motor that the idling coasting level of the locomotive satisfies the preset condition, the sanding operation is performed.

Optionally, determining the amount of torque reduction according to the rotor frequency difference, rotor frequency differential value, and real-time torque of the first motor may include:

According to the rotor frequency difference of the first motor and the preset rotor frequency difference level rules, determine the idling coasting level corresponding to the rotor frequency difference of the first motor, according to the idling coasting level corresponding to the rotor frequency difference of the first motor, and the first motor The real-time torque determines the first torque reduction.

The preset rotor frequency differential level rule may include a mapping relationship between the rotor frequency difference and the idling coasting level. Different idling coasting levels correspond to different torque reduction coefficients. For example, the torque reduction coefficient corresponding to a higher idling coasting level may be set The bigger. The first torque reduction amount may be equal to the real-time torque of the first motor multiplied by the torque reduction factor corresponding to the rotor frequency difference of the first motor.

According to the rotor frequency differential value of the first motor and the preset rotor frequency differential value classification rules, determine the idling coasting level corresponding to the rotor frequency differential value of the first motor, and according to the idling coasting level corresponding to the rotor frequency differential value of the first motor, As well as the real-time torque of the first motor, the second torque reduction amount is determined.

The preset grading rules of the rotor frequency differential value can include the mapping relationship between the rotor frequency differential value and the idling coasting level. Different idling coasting levels correspond to different torque reduction coefficients. For example, the torque corresponding to the higher idling coasting level can be set The greater the reduction factor. The second torque reduction amount may be equal to the real-time torque of the first motor multiplied by the torque reduction coefficient corresponding to the rotor frequency differential value of the first motor.

If the first torque reduction amount is greater than or equal to the second torque reduction amount, the first torque reduction amount is determined as the torque reduction amount; if the first torque reduction amount is less than the second torque reduction amount, the second rotation is determined The amount of torque reduction is the amount of torque reduction. That is, the larger of the first torque reduction amount and the second torque reduction amount is selected as the torque reduction amount, and the torque of the first motor is adjusted.

On the basis of any of the above embodiments, this embodiment describes in detail the process of adjusting the torque of the first motor according to the amount of torque reduction. In this embodiment, adjusting the torque of the first motor according to the torque reduction amount may include:

During the first preset time period, the torque value of the first motor is reduced from the first value to the second value, and the difference between the first value and the second value is the torque reduction amount.

Optionally, within a first preset period of time, the torque value of the first motor is gradually reduced from the first value to the second value according to the decreasing rate of the torque value of the first motor. That is, the unloading of the torque value of the first motor is from fast to slow, which is beneficial to the search for the best sticking point and avoids a sudden drop in torque.

During the second preset time period, the torque value of the first motor is kept unchanged at the second value.

During the third preset time period, the torque value of the first motor is increased from the second value to a preset percentage of the preset torque value, for example, it can be increased to 90% of the preset torque value.

During the fourth preset time period, the torque value of the first motor is increased to the preset torque value.

The recovery rate of the torque value of the first motor in the third preset time period is greater than the recovery rate of the torque value of the first motor in the fourth preset time period. That is to say, for the recovery of the torque value of the first motor, segment recovery is adopted, and recovery is performed first and then slowly, which can effectively avoid the occurrence of idling coasting again.

The specific durations of the first preset time period, the second preset time period, the third preset time period, and the fourth preset time period in this embodiment can be set as needed, and this embodiment does not limit this. The first preset time period, the second preset time period, the third preset time period, and the fourth preset time period constitute a torque adjustment period, and adjust the torque of the first motor when idling occurs.

38 is a schematic diagram of an adhesion control process provided by an embodiment of the present invention. 38 is a schematic diagram of the process of adjusting the torque of the first motor by the adhesion control method provided by an embodiment of the present invention when idling occurs. As shown in FIG. 38, the T1, T2, T3 and T4 sub-tables represent the first preset time period, the second preset time period, the third preset time period and the fourth preset time period, and T1, T2, T3 and T4 constitutes a torque adjustment cycle. The reference frequency curve of the locomotive represents the changing trend that the rotor frequency of the first motor should follow when the locomotive is in the traction mode, and the rotor frequency curve represents the actual rotor frequency of the first motor.

Stage T1 is the stage of torque unloading. Point a is the moment when the locomotive is idling. As shown in Figure 38, once the idling is detected, the torque is quickly unloaded immediately. The unloading amount is from large to small, as shown in the figure. The torque unloading curve shown in section ab in 38 can be fitted as an inverse proportional function curve, and then continue to unload with two small slopes, as shown in section bc and cd in Figure 2, where the unloading rate of section bc is greater than that of section cd Unloading rate until the torque unloading amount is equal to the determined torque reduction amount, that is, the torque difference between point a and point d is equal to the torque reduction amount. The T2 stage is a stage where the torque is kept constant. When the torque unloading amount reaches the torque reduction amount, the locomotive does not run idle and maintains a low torque output, as shown in paragraphs d-e in FIG. 38. The T3 phase is the first recovery phase of torque. After maintaining the low torque output for a period of T2, that is, after idling disappears for a period of T2, the torque is restored to 90% of the preset torque at a preset rate, as shown in FIG. 38 As shown in paragraph ef. The T4 stage is the complete recovery stage of the torque, and the torque is restored to the preset torque, as shown in paragraph f-g in FIG. 38. The lifting rate of the f-g torque is smaller than that of the e-f torque. The preset torque may be the torque at the moment of idling, that is, the preset torque may be set equal to the torque at point a in the figure. During the torque recovery process in the T3 or T4 stage, if idling or coasting occurs again, the preset torque is immediately updated, and at the same time jump from the T3 or T4 stage to the T1 stage, according to the above logic to enter a new round of torque adjustment cycle Until the idling or sliding disappears.

In this embodiment, the torque unloading is from fast to slow, which is beneficial to the search for the best sticking point and avoids a sudden drop in torque. In the later stage of torque recovery, segment recovery is adopted, which can effectively avoid the idling again. It is understandable that the process of gliding is similar and will not be repeated here.

Optionally, determining the rotor frequency difference and the rotor frequency differential value of the first motor according to the collected rotor frequencies of the multiple motors may include:

Perform limiting filtering and low-pass filtering on the collected rotor frequencies of multiple motors, and determine the rotor frequency difference and rotor frequency differential of the first motor according to the rotor frequencies of the multiple motors after limiting filtering and low-pass filtering value. Through limiting filtering and low-pass filtering on the rotor frequency, the noise signal caused by external interference can be eliminated, the accuracy of the rotor frequency can be improved, and the accuracy of adhesion control can be improved.

Optionally, the operating conditions of the locomotive may be determined according to the real-time torque of the first electric machine, and the operating conditions of the locomotive may include idle running conditions, traction operating conditions, and braking operating conditions. For example, the first torque threshold and the second torque threshold are set, where the first torque threshold is greater than zero and the second torque threshold is less than zero. This embodiment is specific to the first torque threshold and the second torque threshold The value is not limited and can be set according to actual needs. If the real-time torque of the first motor is greater than or equal to the first torque threshold, the locomotive is in traction mode; if the real-time torque of the first motor is less than or equal to the second torque threshold, the locomotive is in braking mode; if the first If the real-time torque of the motor is greater than the second torque threshold and less than the first torque threshold, the locomotive is in an idle mode.

Optionally, if the locomotive is in an idle mode, then performing amplitude limiting filtering and low-pass filtering on the collected multiple rotor frequencies may include:

Obtain the current value of the first motor, determine the rotor frequency compensation coefficient of each motor according to the current value of the first motor and the rotor frequency of each motor, and perform the rotor frequency of each motor according to the rotor frequency compensation coefficient of each motor Compensation, limiting and low-pass filtering of the rotor frequency of multiple motors after compensation.

In this embodiment, according to the current value of the first motor and the rotor frequency of each motor, a rotor frequency compensation coefficient is determined and compensated for each motor, which improves the rotor frequency acquisition accuracy and thus the accuracy of adhesion control.

Optionally, on the basis of the foregoing embodiments, an embodiment of the present invention further provides a method for protecting a megawatt direct-drive permanent magnet electric drive system for high-power electric locomotives, where FIG. 39 is an embodiment of the present invention The circuit diagram of the provided traction converter is a detailed circuit implementation based on FIG. 1. The traction converter described here may be a megawatt for electric locomotives as shown in FIG. 1 Direct drive permanent magnet electric drive system. As shown in FIG. 39, the traction converter provided in this embodiment includes: a rectifier module, a bus capacitor, a chopper module, and an inverter module; wherein, a plurality of sensors are also provided in the traction converter. The megawatt direct-drive permanent magnet electric drive system for high-power electric locomotives provided in this embodiment is described by taking a rectifier module as an example. The rectifier module in FIG. 39 may be any four-quadrant rectifier module in FIG. 2, and, In this embodiment, an inverter module is used as an example for description. The inverter module in the figure may also be any inverter module in FIG. 2.

Specifically, the rectifier module, the bus capacitor, the chopping module, and the inverter module are connected in sequence, wherein an input current sensor TA4 is provided at the input end of the rectifier module, and an intermediate voltage sensor TV1 and a ground voltage sensor TV2 are provided in parallel with the bus capacitor The chopper module is provided with a chopper module current sensor TA3, and the output end of the inverter module is provided with a motor U-phase current sensor TA1, a motor V-phase current sensor TA2, a motor stator winding temperature sensor TMP1, and a motor rotor speed sensor SPD.

For the traction converter provided in FIG. 39, the embodiment of the present invention uses the sensors in the traction converter to detect the operating data of the circuit, thereby determining the operating state of each component in the traction converter, and judging whether there is a fault in the circuit. Introduce the method of fault determination in traction converter in detail.

40 is a flowchart of a method for determining a fault of a traction converter according to an embodiment of the present invention; as shown in FIG. 10, the method includes:

S4001: Obtain the data collected by the sensor;

Among them, the sensors are used to collect the operating data of the internal components of the traction converter in real time. The sensors can be, for example, input current sensors, intermediate voltage sensors, ground voltage sensors, chopper module current sensors, motor U-phase current sensors, and motor V-phase currents. The sensor, the motor stator winding temperature sensor, and the motor rotor speed sensor. The data collected by the corresponding sensors may be, for example, current, voltage, temperature, and speed.

S4002: Determine whether at least one single item state corresponding to the sensor is normal according to the data and preset conditions;

The preset condition refers to the condition that the operating data of each component in the circuit should not cause the traction converter circuit to fail. The specific preset condition can be a preset threshold or a preset range. The implementation of the present invention The example does not specifically limit the preset conditions. The single-item state refers to the state of a device or a component in the circuit, such as the input current, the intermediate DC bus voltage, the working state of the ground voltage sensor, the chopper module current, the U-phase input current of the motor, and the motor V Phase input current, motor stator winding temperature, motor speed.

Determine whether at least one single item corresponding to the sensor is normal according to the data collected by the sensor and the preset conditions. Specifically, whether the data collected by the comparison sensor is greater than or less than the corresponding preset threshold, or whether the data collected by the comparison sensor is If the collected data is greater than or less than the corresponding preset threshold value or the collected data exceeds the corresponding threshold value range, it is determined that the single item corresponding to the sensor collecting the data is abnormal. If the collected data meets the preset condition, it is determined that the single item corresponding to the sensor collecting the data is in a normal state.

S4003: If there is an abnormal single-item state, the state of the abnormal single-item state is placed in the fault bit.

Among them, the status bit of the single-item state refers to that in the traction converter, each single-item state has a corresponding binary bit, which is used to indicate the normal or abnormal state of the single-item state. This binary bit is the status bit. When the bit is 0, it indicates that the single item status is normal, when the status bit is 1, it indicates that the single item status is abnormal, that is, the fault bit, that is, when the single item status status bit is 1, it indicates the corresponding single item status The status bit is a fault bit.

Specifically, according to the data collected by the sensor and the preset conditions, it is determined whether the single-item state corresponding to the sensor is normal. If there is an abnormal single-item state, the abnormal single-state state is set to 1, that is, the state is located in the fault Bit. When the status bit is a fault bit, the fault information corresponding to the fault bit is reported, and the traction converter receives the fault information, thereby performing the corresponding circuit protection operation.

The fault determination method for a traction converter provided by an embodiment of the present invention acquires the data collected by the sensor; according to the data and preset conditions, it is determined whether at least one single item state corresponding to the sensor is normal; In the normal single-item state, the state of the abnormal single-item state is placed in the fault bit. According to the data collected by the sensor and the preset conditions, the operating status of each component in the circuit is determined in real time. When the traction converter fails, the operating status corresponding to each individual status can be determined according to the status bit label, so that it can be carried out quickly. The related circuit protection operation effectively reduces the failure rate of the traction converter.

The following uses a detailed embodiment to describe in detail the protection method of the traction converter provided by the embodiment of the present invention.

FIG. 41 is a logic judgment diagram of a protection method for a traction converter provided by an embodiment of the present invention; as shown in FIG. 41, the sensors in the traction converter mainly include the sensors involved in the above-mentioned FIG. 39, which will not be repeated here Among them, the fault information refers to the specific fault that may occur when the operating data of the components of the traction converter circuit does not meet the preset conditions. The fault information may be, for example, a single item is abnormal, and the fault information may also be a device in the traction converter Faults, connection faults, etc. In the program of the main control unit of the traction converter, each fault message has a corresponding binary bit, which is used to indicate whether the circuit fault corresponding to the fault message occurs or does not occur at this time. This binary bit is Status bit.

Among them, when the status bit is 0, it indicates that the fault corresponding to the fault information has not occurred, and when the status bit is 1, it indicates that the fault corresponding to the fault information has occurred in the circuit. At this time, the status bit of the fault information is the fault bit. Among them, there is a case where one sensor corresponds to one fault information, there is also a case where one sensor corresponds to multiple fault information, and there are multiple sensors corresponding to one fault information. The following describes the correspondence between sensors and fault information in detail.

1) Input current sensor TA4

The current input terminal is provided with an input current sensor TA4. First, the first current collected by the input current sensor TA4 is acquired, and secondly, it is determined whether the first current is greater than the first preset threshold, and if the duration of the first current is greater than the first preset threshold, the duration is greater than At the first preset time, it is determined that the single-item state corresponding to the input current sensor TA4 is abnormal. The specific single-item state here is that the input current of the traction converter is too large. A fault with an excessive input current is called a converter. Input overcurrent, position the input overcurrent status of the converter to the fault bit.

2) Intermediate voltage sensor TV1 and ground voltage sensor TV2

The intermediate voltage sensor TV1 and the ground voltage sensor TV2 are connected in parallel with the bus capacitor. First, the first voltage collected by the intermediate voltage sensor TV1 and the second voltage collected by the ground voltage sensor TV2 are obtained, and then the specific fault information is judged.

One of the determination logics is to determine whether the first voltage is greater than the second preset threshold, and if the duration of the first voltage greater than the second preset threshold is greater than the second preset time, it is determined that the single state corresponding to the intermediate voltage sensor TV1 is abnormal Here, the specific single-state abnormality is that the intermediate DC bus voltage of the traction converter is too high. A fault where the intermediate DC bus voltage is too large is called the intermediate bus overvoltage, and the state of the intermediate bus overvoltage is located at the fault position.

Another kind of judgment logic is to judge whether the first voltage is less than the third preset threshold, and if the duration of the first voltage being less than the third preset threshold is greater than the third preset time, it is determined that the single state corresponding to the intermediate voltage sensor TV1 is not Normal, the specific single-state condition here is abnormal is that the intermediate DC bus voltage of the traction converter is too low, and the fault that the intermediate DC bus voltage is too small is called the intermediate bus undervoltage, and the state of the intermediate bus undervoltage is located at the fault position .

Yet another judgment logic is to judge whether the second voltage is within the first preset range, if the second voltage is not within the first preset range, it is determined that the single item corresponding to the grounded voltage sensor TV2 is abnormal, the specific single item here The abnormal state is the fault of the ground voltage sensor, and the fault state of the ground voltage sensor is placed in the fault position.

In this embodiment, it can also be determined whether the first voltage is within the second preset range. If the first voltage is not within the second preset range, it is determined that the intermediate voltage sensor is faulty, and the state of the intermediate voltage sensor fault is located at the fault position. It also includes subtracting half of the first voltage from the second voltage to obtain a third voltage to determine whether the third voltage is greater than the fourth preset threshold, and if the duration of the third voltage is greater than the fourth preset threshold is greater than the fourth preset time, It is determined that the positive pole of the busbar of the traction converter is grounded, and the fault that the positive pole of the busbar is grounded is called the positive grounding of the intermediate bus, and the state of the positive grounding of the intermediate bus is located at the fault position.

Optionally, it is determined whether the third voltage is less than the fifth preset threshold, if the duration of the third voltage is less than the fifth preset threshold is greater than the fifth preset time, it is determined that the negative pole of the bus of the traction converter is grounded, and the negative pole of the bus is determined The fault of grounding is called negative grounding of the middle bus, and the state of negative grounding of the middle bus is located at the fault position.

3) Chopper module current sensor TA3

The chopping module is provided with a chopping module current sensor TA3, firstly obtains the second current collected by the chopping module current sensor TA3, and secondly judges whether the second current is greater than a sixth preset threshold, if the second current is greater than the sixth preset threshold Is longer than the sixth preset time, it is determined that the single-item state corresponding to the chopper module current sensor TA3 is abnormal. The specific single-state state here is that the current of the chopper module of the traction converter is too large, and the chopper module A fault with excessive current is called chopping overcurrent, and the state of chopping overcurrent is placed in the fault bit.

Optionally, if the main control unit does not control the chopper module to be turned on, and the chopper module is not turned on, determine whether the second current is greater than the seventh preset threshold, and if the chopper module is not turned on, the second current is greater than the first If the duration of the seven preset thresholds is greater than the seventh preset time, it is determined that the chopper module of the traction converter is not turned on but the current is detected, and the failure that the chopper module is not turned on but the current is detected is called unchopped current, Put the uncut state in the fault bit.

Further, if the chopper module is turned on, it is determined whether the second current is greater than the eighth preset threshold, and if the chopper module is turned on, the second current is not detected to be greater than the eighth preset threshold within the eighth preset time, It is determined that the chopping module of the traction converter is turned on but no current is detected, and that the chopper module is turned on but the current is not detected is called chopping and no current, and the state of chopping and no current is located at the fault position.

4) Motor U-phase current sensor TA1, motor V-phase current sensor TA2, motor stator winding temperature sensor TMP1 and motor speed sensor SPD

The current output terminal is provided with a motor U-phase current sensor TA1, a motor V-phase current sensor TA2, a motor stator winding temperature sensor TMP1, and a motor speed sensor SPD. First, the third current collected by the motor U-phase current sensor TA1 is acquired, and the motor V-phase is acquired The fourth current collected by the current sensor TA2, the temperature collected by the motor stator winding temperature sensor TMP1, and the first speed collected by the motor speed sensor SPD are collected, followed by specific fault information judgment.

One of the judgment logics is to determine whether the third current is greater than the ninth preset threshold. If the duration of the third current is greater than the ninth preset threshold is greater than the ninth preset time, then determine the single state corresponding to the motor U-phase current sensor TA1 Not normal, the specific single-item state here is abnormal, the motor U-phase input current is too large, the fault of the motor U-phase input current is too large is called inverter U-phase overcurrent, the inverter U-phase overcurrent state Located in the fault position.

Another kind of judgment logic is to judge whether the fourth current is greater than the tenth preset threshold, and if the duration of the fourth current is greater than the tenth preset threshold is greater than the tenth preset time, then determine the single state corresponding to the motor V-phase current sensor TA2 Not normal, the specific single item state here is abnormal, the motor V phase input current is too large, the fault of the motor V phase input current is too large is called inverter V phase over current, the inverter V phase over current status Located in the fault position.

Another judgment logic is to judge whether the temperature is greater than the eleventh preset threshold, if the duration of the temperature is greater than the eleventh preset threshold is greater than the eleventh preset time, it is determined that the single state corresponding to the motor stator winding temperature sensor TMP1 is not Normal, the specific single-state condition here is abnormal because the temperature of the stator winding of the motor is too high. A fault where the temperature of the stator winding of the motor is too large is called overtemperature of the traction motor, and the state of overtemperature of the traction motor is located at the fault position.

Another judgment logic is to judge whether the first speed is greater than the twelfth preset threshold, and if the duration of the first speed is greater than the twelfth preset threshold is greater than the twelfth preset time, determine the single item corresponding to the motor speed sensor SPD The state is abnormal. The specific single-state abnormality here is that the motor speed is too high. The fault of the motor speed is too large is called traction motor overspeed, and the state of traction motor overspeed is located at the fault position.

Based on the above embodiment, the value obtained by adding the third current to the fourth current may be inverted to obtain the fifth current to determine whether the fifth current is greater than the thirteenth threshold, and if the fifth current is greater than the thirteenth If the duration of the threshold is greater than the thirteenth preset time, it is determined that the W-phase input current of the motor is too large, and the fault of the W-phase input current of the motor is called the inverter W-phase overcurrent, and the inverter W-phase overcurrent The status of the flow is in the fault bit.

Further, in the precharge phase of the traction converter, it is determined whether the first voltage is less than the fourteenth preset threshold and whether the first current is greater than the fifteenth preset threshold, if the When a voltage is less than the fourteenth preset threshold and the first current is greater than the fifteenth preset threshold, it is determined that the intermediate bus of the traction converter is short-circuited, and the state of short-circuiting the intermediate bus is located at the fault position.

Optionally, if the voltage value of the fourth voltage at different times has a positive or negative range, that is, the fourth voltage is detected as a positive value at one time, and the fourth voltage is detected as a negative value at another time, and the traction changes After the blocker pulse signal, the fourth voltage of the converter becomes zero, then the four-quadrant rectifier of the traction converter is grounded. The fault of the four-quadrant rectifier grounding is called four-quadrant grounding, and the state of the four-quadrant grounding is located at the fault position. . It also includes that if the voltage value of the fourth voltage at different times has a change in the range of positive and negative, and after the traction converter blocks the pulse signal, the fourth voltage still has a change in the range of positive and negative, then determine the traction converter's The inverter is grounded, and the grounded state of the inverter is located at the fault position.

In this embodiment, the chopper module includes a timer. The timer starts timing when the chopper module starts to send pulses. When the chopper module stops sending pulses, the timer stops working, within the fifteenth preset time range, The timing data of the timer is accumulated to obtain the first time. If the first time is greater than the sixteenth preset threshold, it will cause the circuit temperature in the chopper module to be too high, and the fault is determined to be the resistance in the chopper module of the traction converter If the temperature is too high, the fault where the resistance temperature is too high is called chopping overtemperature, and the state of chopping overtemperature is located at the fault position.

Optionally, the third current effective value is subtracted from the fourth current effective value to obtain the sixth current, the third current effective value is subtracted from the fifth current effective value to obtain the seventh current, and the fourth current effective value is subtracted from the fifth The current obtains the eighth current, and determines whether the sixth current, the seventh current, and the eighth current are greater than the seventeenth preset threshold, if the sixth current is greater than the seventeenth preset threshold, or the seventh current is greater than the seventeenth preset threshold , Or the eighth current is greater than the seventeenth preset threshold, it is determined that the traction motor of the traction converter is out of phase, and the state of the traction motor is out of phase at the fault position.

On the basis of the above embodiment, it is also possible to determine the status of the traction motor not working on the premise that the traction handle is in a non-zero position. The traction handle is located in the locomotive control room, and the related operations of the traction handle are also completed in the locomotive control room. When the traction handle is at the zero position, it means that the locomotive is not performing any operation at this time, nor is it sending any signal to each component of the locomotive. The traction handle has multiple gears. When the traction handle is in a non-zero position, it indicates that the locomotive is performing an operation, such as forwarding and braking. One of the fault information is that the traction motor does not work, there is a corresponding status bit, which will be described in detail below.

In the specific implementation process, when the traction handle is not at the zero position, it is determined whether the third current is less than the eighteenth preset threshold and whether the fourth current is less than the nineteenth preset threshold, if the third current is less than the eighteenth preset If the duration of the threshold is greater than the sixteenth preset time and the duration of the fourth current is less than the nineteenth preset threshold is greater than the seventeenth preset time, it is determined that the traction motor is not working, and the traction motor is not working. Fault bit.

Further, on the basis of the foregoing embodiment, when receiving the adjacent-axis speed transmitted from the main control unit, the status bits of the speed sensor failure and the shaft-lock failure can be determined according to the adjacent-axis speed and the local-axis speed. Among them, the main control unit is the core component of the traction converter, including communication and control functions. The adjacent axis refers to an axis other than the axis where the traction converter currently performing fault judgment is located. Here, we refer to the axis where the traction converter currently undergoing fault judgment is located as the main axis, and other axes other than this axis The axis is called the adjacent axis. Specifically, there are 4 axis locomotives, 6 axis locomotives, and 8 axis locomotives. The main control unit can transmit the adjacent axis speed through the network, and then determine the corresponding fault information based on the adjacent axis speed and the local axis speed.

In the specific implementation process, receive the adjacent axis speed transmitted by the main control unit, determine the minimum value of the first speed and all adjacent axis speeds as the second speed, and determine whether the difference between the first speed and the second speed is greater than the twentieth Set a threshold, and determine whether the difference between the first speed and the maximum value of the adjacent axis speed is greater than the twenty-first preset threshold, if the difference between the first speed and the second speed is greater than the twentieth preset threshold, the duration is greater than At the eighteenth preset time, and the difference between the maximum value of the first speed and the adjacent axis speed is greater than the twenty-first preset threshold and the duration is greater than the nineteenth preset time, it is determined that the motor speed sensor is faulty, and the motor speed is The fault of the sensor fault is called the speed sensor fault, and the status of the speed sensor fault is placed in the fault bit.

On the basis of the above embodiment, when the speed sensor status position is 0, that is, not the fault position, it is determined whether the second speed is greater than the twenty-second preset threshold, and whether the first speed is less than the twenty-third preset threshold If the duration of the second speed is greater than the twenty-second preset threshold is greater than the nineteenth preset time, and the duration of the first speed is less than the twenty-third preset threshold is greater than the twentieth preset time, then determine the motor If the shaft lock fails, the failure of the motor shaft lock is called the shaft lock failure, and the state of the shaft lock failure is located at the failure position.

The method for determining the fault of the traction converter provided by the embodiment of the present invention obtains the operating data of each component in the circuit through the sensor, and determines whether the single item corresponding to the sensor is normal according to the threshold corresponding to the operating data and the operating data, and can also determine Whether the device, connection, etc. in the circuit are normal, if a single item fails, or the device, connection, etc. fails, the status corresponding to the fault is placed in the fault bit, thereby identifying the fault information in the circuit, and reporting the fault information corresponding to the fault bit For the main control unit, after receiving the fault information, the main control unit can perform circuit protection operations according to the actual situation, thereby reducing the failure rate of the traction converter.

The invention also provides a megawatt direct-drive permanent magnet electric drive system for high-power electric locomotives, which is used to supply electric locomotives using high-power direct-drive permanent magnet synchronous motors. The electric locomotive includes three high-power direct-drive permanent magnet synchronous motors Motor, converter includes: first precharge module, second precharge module, first four-quadrant rectifier, second four-quadrant rectifier, first chopping module, second chopping module, intermediate DC loop, first inverse Transformer module, second inverter module, third inverter module and auxiliary converter, the first four-quadrant rectifier and the second four-quadrant rectifier pass through the first pre-charging module and the second The charging module is connected to the main transformer of the electric locomotive, and the first four-quadrant rectifier and the second four-quadrant rectifier are respectively connected to the intermediate DC loop through the first chopper module and the second chopper module, The intermediate DC circuit is respectively connected to the first inverter module, the second inverter module, the third inverter module and the auxiliary converter;

Wherein, the first precharging module includes a first charging capacitor, a first precharging contactor and a first main working contactor, and the second precharging module includes a second charging capacitor, a second precharging contactor and a first Two main working contactors, the first four-quadrant rectifier and the second four-quadrant rectifier each include eight switch tubes, and the first chopper module includes a first switch tube, a first current sensor, and a first reverse A diode and a first chopping resistor, the second chopping module includes a second switch tube, a second current sensor, a second reverse diode and a second chopping resistor, and the intermediate DC loop includes a first parallel connected The current-side support capacitor, the second DC-side support capacitor, the slow discharge resistor, the DC bus voltage sensor and the ground detection module, the first inverter module, the second inverter module and the third inverter module all include Three-phase inverter circuit composed of six switch tubes;

The first pre-charging module and the second pre-charging module are used to transmit the alternating current of the main transformer to the first four-quadrant rectifier and the second four-quadrant rectifier, respectively;

The first four-quadrant rectifier and the second four-quadrant rectifier are used to convert the alternating current transmitted by the first pre-charging module and the second pre-charging module into direct current, and then output to the first chopping wave Module and the second chopping module;

The first chopping module and the second chopping module are used for chopping the DC power and transmitting it to the intermediate DC loop;

The intermediate DC loop uses the received DC power to output to the first inverter module, the second inverter module, the third inverter module, and the auxiliary converter, respectively;

The first inverter module, the second inverter module and the third inverter module are used to convert the received DC power into three-phase AC power and output to the three high-power direct-drive permanent magnet synchronous motors respectively;

The auxiliary converter is used to convert the received DC power into three-phase AC power and output it to the auxiliary load of the electric locomotive

The megawatt direct drive permanent magnet electric drive system for high-power electric locomotives provided by the embodiments of the present application can be used to implement the control method of the megawatt direct drive permanent magnet electric drive system for high-power electric locomotives in the foregoing corresponding embodiments, The implementation is the same as the principle and will not be repeated here.

The present invention also provides an electronic device, including: a processor coupled with a memory; the memory is used to store a computer program; the processor is used to call the computer program stored in the memory to implement the power of any of the foregoing embodiments A megawatt direct drive permanent magnet electric drive system for locomotives.

The present invention also provides a storage medium readable by an electronic device, including: a program or an instruction, when the program or the instruction runs on the electronic device, to implement any one of the foregoing embodiments of a megawatt direct-drive permanent magnet electric power locomotive Transmission system.

Persons of ordinary skill in the art may understand that all or part of the steps of the foregoing method embodiments may be completed by a program instructing relevant hardware. The aforementioned program may be stored in a computer-readable storage medium. When the program is executed, the steps including the foregoing method embodiments are executed; and the foregoing storage medium includes various media that can store program codes, such as ROM, RAM, magnetic disk, or optical disk.

Finally, it should be noted that the above embodiments are only used to illustrate the technical solution of the present invention, rather than limiting it; although the present invention has been described in detail with reference to the foregoing embodiments, those of ordinary skill in the art should understand that: The technical solutions described in the foregoing embodiments can still be modified, or some or all of the technical features can be equivalently replaced; and these modifications or replacements do not deviate from the essence of the corresponding technical solutions of the technical solutions of the embodiments of the present invention. range.

Claims

A megawatt direct drive permanent magnet electric drive system for electric locomotive is used to control the converter of electric locomotive using high power direct drive permanent magnet synchronous motor, the electric locomotive includes three high power direct drive permanent magnet synchronous motors ; Characterized by,

The megawatt direct drive permanent magnet electric drive system for electric locomotive includes: a first precharge module, a second precharge module, a first four-quadrant rectifier, a second four-quadrant rectifier, a first chopper module, a second chopper Wave module, intermediate DC loop, first inverter module, second inverter module, third inverter module and auxiliary converter, the first four-quadrant rectifier and the second four-quadrant rectifier pass through the first A pre-charging module and the second pre-charging module are connected to the main transformer of the electric locomotive, and the first four-quadrant rectifier and the second four-quadrant rectifier pass through the first chopper module and the second The chopper module is connected to the intermediate DC circuit, and the intermediate DC circuit is respectively connected to the first inverter module, the second inverter module, the third inverter module, and the auxiliary converter;

Wherein, the first precharging module includes a first charging capacitor, a first precharging contactor and a first main working contactor, and the second precharging module includes a second charging capacitor, a second precharging contactor and a first Two main working contactors, the first four-quadrant rectifier and the second four-quadrant rectifier each include eight switch tubes, and the first chopper module includes a first switch tube, a first current sensor, and a first reverse A diode and a first chopping resistor, the second chopping module includes a second switch tube, a second current sensor, a second reverse diode and a second chopping resistor, and the intermediate DC loop includes a first parallel connected The current-side support capacitor, the second DC-side support capacitor, the slow discharge resistor, the DC bus voltage sensor and the ground detection module, the first inverter module, the second inverter module and the third inverter module all include Three-phase inverter circuit composed of six switch tubes;

The megawatt direct drive permanent magnet electric drive system for electric locomotive is used for:

Transmitting the alternating current of the main transformer to the first four-quadrant rectifier and the second four-quadrant rectifier through the first pre-charging module and the second pre-charging module, respectively;

After the first four-quadrant rectifier and the second four-quadrant rectifier respectively convert the alternating current transmitted by the first pre-charging module and the second pre-charging module into direct current, and output to the first chopper module And the second chopping module;

Chopping the DC power through the first chopping module and the second chopping module, and then transmitting it to the intermediate DC loop;

Output the received DC power through the intermediate DC loop to the first inverter module, the second inverter module, the third inverter module and the auxiliary converter;

Converting the received DC power into three-phase AC power through the first inverter module, the second inverter module, and the third inverter module, and outputting them to the three high-power direct-drive permanent magnet synchronous motors, respectively;

The auxiliary DC converter converts the received DC power into three-phase AC power and outputs it to the auxiliary load of the electric locomotive.
The megawatt direct drive permanent magnet electric drive system for electric locomotives according to claim 1, characterized in that, for any one of the first four-quadrant rectifier and the second four-quadrant rectifier, the four-quadrant rectifier, Converting alternating current into direct current through the first four-quadrant rectifier and the second four-quadrant rectifier, and outputting to the first chopper module and the second chopper module, specifically including:

Sampling the AC current input to the four-quadrant rectifier to obtain the AC current within the sampling period. The AC current includes a positive half-cycle current value and a negative half-cycle current value; wherein, according to the preset sampling frequency, the input The AC current of the four-quadrant rectifier is sampled to obtain multiple sampling points, and the obtained multiple sampling points are drawn into a curve to obtain a sine or cosine curve; the preset sampling frequency is N times the IGBT on-off frequency, The N≥2;

Acquiring a first sum value of the current value of the positive half cycle and a second sum value of the current value of the negative half cycle, and obtaining a current offset value according to the first sum value and the second sum value; wherein, The values of the multiple sampling points in the positive half cycle are added to obtain the first sum value P, and then the values of the multiple sampling points in the negative half cycle are added to obtain the second sum value N, the absolute value of the P value and the N value Carry out the difference calculation, the difference is Q;

Input the first difference between the current offset value and zero to the first PI controller to obtain the first output value output by the first PI controller; wherein, the DC offset value Q and zero are input to the first PI controller, the first PI controller constitutes a control deviation according to the DC offset value Q and zero, and the proportional and integral of the deviation are linearly combined to form a control amount, which controls the AC current and eliminates the DC offset of the AC current. Is the first output value;

A pulse width modulation symbol is obtained according to the first output value and the second output value output by the PR controller, and the PR controller is used to perform static-free control of the alternating current so that the period and phase of the alternating current are The grid voltage is the same; where the AC current is input to the PR controller to ensure that the phase and cycle of the AC current are the same as the grid voltage, a stable output AC current is obtained, which is the second output value;

The on-off of the insulated gate bipolar transistor IGBT in the four-quadrant rectifier is controlled according to the pulse width modulation sign to control the four-quadrant rectifier to convert alternating current to direct current.
The megawatt direct-drive permanent magnet electric drive system for electric locomotives according to claim 2, characterized in that before the AC current of the input four-quadrant rectifier is sampled to obtain the AC current within the sampling period, include:

Obtain the second difference between the DC bus voltage of the four-quadrant rectifier and the command voltage;

Input the second difference value to the second PI controller, so that the third output value output by the second PI controller is multiplied by the output value of the phase-locked loop to obtain an alternating current in the same phase as the grid voltage, The phase-locked loop is used to control the period and phase of the alternating current to be consistent with the period and phase of the grid voltage;

The sampling of the AC current input to the four-quadrant rectifier to obtain the AC current within the sampling period includes:

Sampling the alternating current input to the four-quadrant rectifier according to a preset sampling frequency to obtain a sampling current, and the preset sampling frequency is twice the on-off frequency of the IGBT;

Obtaining the AC current in the sampling period according to the grid voltage phase determined by the phase-locked loop and the sampling current;

Before obtaining the AC current in the sampling period according to the grid voltage phase determined by the phase-locked loop and the sampling current, the method further includes:

Filtering the sampling current through a first band-pass filter and a second band-pass filter to obtain a filtered sampling current; wherein, the first band-pass filter is used to obtain a main frequency signal of an alternating current, the The second band-pass filter is used to filter out interference harmonics.
The megawatt direct drive permanent magnet electric drive system for an electric locomotive according to claim 2, wherein the first difference between the current bias value and zero is input to a first PI controller to obtain Before the first output value output by the first PI controller, it further includes:

Judging whether the absolute value of the first difference is greater than the absolute value of the current loop width, the obtained judgment result is yes;

The obtaining the pulse width modulation symbol according to the first output value and the second output value output by the PR controller includes:

Summing the first output value and the second output value to obtain a third sum value, the first output value is a current variable, and the second output value is a current value;

The pulse width modulation symbol is obtained according to the third sum value and the unipolar frequency-doubling pulse modulation method.
The megawatt direct drive permanent magnet electric drive system for electric locomotive according to claim 1, characterized in that

The megawatt direct-drive permanent magnet electric drive system for electric locomotive further includes: a first chopping module and a second chopping module, the first chopping module is connected to the first four-quadrant rectifier and the intermediate DC Loop, the second chopper module connects the second four-quadrant rectifier and the intermediate DC loop;

The megawatt direct drive permanent magnet electric drive system for electric locomotives is also used for:

DC power output from the first four-quadrant rectifier and the second four-quadrant rectifier is chopped by the first chopping module and the second chopping module, and then output to the intermediate DC loop;

Specifically, for any one of the first chopping module and the second chopping module, the megawatt direct drive permanent magnet electric drive system for electric locomotives is also used for:

Periodic detection of the intermediate DC bus voltage, the intermediate DC bus voltage being the voltage on the DC bus on the electric locomotive;

When the detected intermediate DC bus voltage value is greater than the upper chopping threshold, the P regulator is used to adjust the intermediate DC bus voltage until the detected intermediate DC bus voltage value is less than the lower chopping threshold, the chopping The upper limit of the wave threshold is greater than the lower limit of the chopping threshold; wherein, the principle of the P regulator is to control the chopper tube to be turned on within a certain proportion of the detection cycle.
The megawatt direct-drive permanent magnet electric drive system for electric locomotives according to claim 5, characterized in that the use of a P regulator to adjust the intermediate DC bus voltage includes:

Using the P regulator to determine the chopping duty cycle within the target detection period; the target detection period includes: from the detected intermediate DC bus voltage value greater than the upper chopping threshold, to the detected intermediate DC bus voltage value Less than the chopping lower threshold between the detected detection period;

According to the chopping duty ratio, determine the opening time of the chopper tube within the target detection period;

According to the turn-on time, controlling the turn-on or turn-off of the chopper tube, so that the voltage value of the intermediate DC bus drops to less than the lower threshold of the chopper;

The control method further includes:

When it is detected that the voltage value of the intermediate DC bus is lower than the lower chopping threshold, the chopper tube is controlled to be turned off.
The megawatt direct drive permanent magnet electric drive system for electric locomotives according to claim 6, characterized in that, before using the P regulator to determine the chopping duty cycle within the target detection period, it further comprises:

Determine the target parameters according to the following formula;

Err = U1-chopping lower threshold

Among them, Err represents the target parameter, U1 represents the intermediate DC bus voltage value detected in the target detection period;

Correspondingly, the use of the P regulator to determine the chopping duty cycle within the target detection period includes:

Obtain the control coefficient corresponding to the P regulator;

Determine the chopping duty cycle according to the control coefficient and the target parameter;

The acquiring the control coefficient of the P regulator includes:

Determine the control coefficient according to the following formula;

Kp_chp = 1 / (DC bus voltage overvoltage protection threshold-chopping lower threshold)

Among them, Kp_chp represents the control coefficient;

The determining the chopping duty ratio according to the control coefficient and the target parameter includes:

Determine the chopping duty cycle according to the following formula;

C_duty = Err * Kp_chp

Among them, C_duty represents the chopping duty ratio, Err represents the target parameter, and Kp_chp represents the control coefficient;

Before determining the opening time of the chopper tube within the target detection period according to the chopping duty cycle, the method further includes:

Performing error prevention processing on the chopping duty ratio;

Wherein, the error prevention processing of the chopping duty ratio includes:

If the value of the chopping duty ratio is greater than 1, the value of the chopping duty ratio is set to 1;

If the value of the chopping duty ratio is less than 0, the value of the chopping duty ratio is set to 0.
The megawatt direct drive permanent magnet electric drive system for electric locomotive according to claim 1, further comprising:

Determine the speed of the high-power direct-drive permanent magnet synchronous motor to be controlled;

Determining a first control strategy according to the rotation speed and the first mapping relationship, where the first mapping relationship includes a one-to-one correspondence between at least one rotation speed range and at least one control strategy;

The expected control phase angle of the high-power direct-drive permanent magnet synchronous motor to be controlled is determined according to the first control strategy.
The megawatt direct-drive permanent magnet electric drive system for electric locomotives according to claim 8, wherein the first mapping relationship includes: a correspondence relationship between a rotation speed below a rated rotation speed and an MTPA control strategy;

Correspondence between the speed above the rated speed and the field weakening control strategy.
The megawatt direct drive permanent magnet electric drive system for electric locomotives according to claim 9, wherein the MTPA control strategy includes:

Determine q-axis current reference and d-axis current reference according to the torque current curve;

Calculating a first difference between the q-axis current reference and the q-axis actual current and a second difference between the d-axis current reference and the d-axis actual current;

The d-axis voltage reference is obtained according to the first difference value through the first PI controller, and the q-axis voltage reference is obtained based on the second difference value through the second PI controller;

Calculating the sum of the q-axis voltage reference and the q-axis feedforward voltage to obtain the actual q-axis voltage reference, and calculating the sum of the d-axis voltage reference and the d-axis feedforward voltage to obtain the actual d-axis voltage reference; wherein, The feedforward voltage can be calculated by the following feedforward decoupled closed-loop transfer function matrix:

The closed-loop transfer function of the feedforward decoupling is obtained by the following voltage calculation equation of feedforward decoupling:
The megawatt direct drive permanent magnet electric drive system for electric locomotives according to claim 9, wherein the field weakening control strategy includes:

The PI controller calculates the d-axis current change in a given field weakening state according to the difference between the voltage limit value and the feedforward voltage amplitude;

The d-axis current reference after the field-weakening adjustment is obtained by giving the sum of the d-axis current change and the d-axis current under the given field weakening state;

Calculate the q-axis current reference after field weakening adjustment according to the d-axis current reference and torque formula;

The PI controller obtains the work angle β according to the difference between the q-axis current setting and the q-axis actual current;

Calculate the actual q-axis voltage reference and the actual d-axis voltage reference by the following formula;

U d = U s cosβ

U q = U s cosβ

Among them, Us is the voltage limit value, Ud is the actual d-axis voltage given, and Uq is the actual q-axis voltage given.
The megawatt direct drive permanent magnet electric drive system for electric locomotives according to any one of claims 9 to 11, further comprising:

When the control strategy is switched from the MTPA control strategy to the field weakening control strategy, the voltage vector angle in the MTPA control strategy at the moment of switching is used as the initial power angle β in the field weakening control strategy;

When the control strategy is switched from the field weakening control strategy to the MTPA control strategy, the last beat power angle β in the instantaneous field weakening control strategy is passed through the formula by switching
Calculate the actual q-axis voltage setting and actual d-axis voltage setting in the MTPA control strategy.
The megawatt direct drive permanent magnet electric drive system for electric locomotive according to claim 1, further comprising:

Obtain the frequency of the modulated wave of the high-power direct-drive permanent magnet synchronous motor to be modulated;

Determining a first modulation strategy according to the range of the frequency of the modulated wave and the second mapping relationship, where the second mapping relationship includes a one-to-one correspondence between the frequency range of at least one modulated wave and at least one modulation strategy;

The PWM carrier frequency of the high-power direct-drive permanent magnet synchronous motor is determined according to the first modulation strategy.
The megawatt direct drive permanent magnet electric drive system for electric locomotives according to claim 13, wherein the second mapping relationship includes:

When the frequency of the modulated wave is in the low-speed stage, it corresponds to the asynchronous modulation strategy;

When the frequency of the modulated wave is greater than the low-speed stage and lower than the high-speed stage, the corresponding synchronous modulation strategy;

The frequency of the modulated wave corresponds to the square wave modulation strategy at the high-speed stage.
The megawatt direct drive permanent magnet electric drive system for electric locomotive according to claim 1, characterized in that

Inject a high-frequency voltage signal into the stator winding of the high-power direct-drive permanent magnet synchronous motor to obtain a three-phase stator winding current;

Obtaining the d-axis target current and the q-axis target current in the expected two-phase synchronous rotating coordinate system according to the three-phase stator winding current;

Obtain the initial position angle of the rotor according to the d-axis target current and the q-axis target current, wherein the initial position angle is the initial position after compensation according to the pole polarity of the high-power direct-drive permanent magnet synchronous motor angle.
The megawatt direct-drive permanent magnet electric drive system for an electric locomotive according to claim 15, wherein the acquiring the initial position angle of the rotor according to the d-axis target current and the q-axis target current includes:

Acquiring the first initial position angle of the rotor according to the q-axis target current;

Obtain the rotor pole compensation angle according to the d-axis target current;

Acquiring the initial position angle of the rotor according to the first initial position angle and the magnetic pole compensation angle;

The obtaining the first initial position angle of the rotor according to the q-axis target current includes:

Performing low-pass filtering on the q-axis target current to obtain an error input signal;

Acquiring the first initial position angle according to the error input signal;

The process of low-pass filtering the q-axis target current to obtain an error input signal includes:

Modulating the q-axis target current with a modulation signal to obtain the modulated q-axis target current;

Performing low-pass filtering on the modulated q-axis target current to obtain the error input signal;

The obtaining the first initial position angle according to the error input signal includes:

Obtaining the proportional deviation and the integral deviation of the error input signal according to the input error signal;

Acquiring the first initial position angle according to the linear combination of the proportional deviation and the integral deviation;

The obtaining the rotor pole compensation angle according to the d-axis target current includes:

Injecting a plurality of voltage pulse signals with equal voltage amplitudes and different angles into the high-power permanent magnet synchronous motor to obtain the response current of each voltage pulse signal;

Determine the magnetic pole compensation angle of the rotor according to the plurality of response currents;

The determining the magnetic pole compensation angle of the rotor according to the plurality of response currents includes:

When the difference between the angle of the injected voltage pulse signal and the first initial position angle satisfies a preset error range, and the amplitude of the response current of the voltage pulse signal is greater than the first value, the rotor pole compensation is determined The angle is 0, and the first value is the maximum value of the amplitudes of the multiple response currents;

When the difference between the angle of the injected voltage pulse signal and the first initial position angle satisfies a preset error range, and the magnitude of the response current of the voltage pulse signal is less than the second value, the rotor pole compensation is determined The angle is π, and the second value is the minimum value of the amplitudes of the multiple response currents.
The megawatt direct-drive permanent magnet electric drive system for electric locomotives according to claim 16, wherein the high-frequency voltage signal is:

Where U mh is the amplitude of the high-frequency voltage signal, ω h is the angular frequency of the high-frequency voltage signal, and t is the time to inject the high-frequency voltage signal;

The d-axis target current and the q-axis target current in the expected two-phase synchronous rotating coordinate system are obtained according to the three-phase stator winding current, and are calculated by the following formula:

Where L is the average inductance L = (L d + L q ) / 2, and △ L is the half-differential inductance △ L = (L d -L q ) / 2;

The low-pass filtering process is performed on the q-axis target current to obtain an error input signal, which is calculated by the following formula:

Among them, LPF stands for low-pass filtering; when the rotor position estimation error is small enough, the error input signal after the limit equivalent linearization is:

The first initial position angle is obtained and calculated by the following formula:

Among them, s represents Laplace operator, k p is the coefficient of proportional term, and k i is the coefficient of integral term.
The megawatt direct drive permanent magnet electric drive system for electric locomotive according to claim 1, further comprising:

Obtaining the compensation phase angle of the rotor of the high-power direct-drive permanent magnet synchronous motor according to the control interruption period, the modulation carrier period, and the current angular velocity of the rotor of the high-power direct-drive permanent magnet synchronous motor;

Obtain the current actual control phase angle according to the compensation phase angle;

Obtain the current expected control phase angle according to the current d-axis voltage given value and the current q-axis voltage given value;

According to the proportional deviation and integral deviation of the current expected control phase angle and the current actual control phase angle, the current actual control phase angle is corrected online.
The megawatt direct-drive permanent magnet electric drive system for electric locomotives according to claim 18, characterized in that, according to the control interruption period, modulated carrier period, and the rotor of the high-power direct-drive permanent magnet synchronous motor The current angular velocity, and obtaining the compensated phase angle of the rotor of the direct drive permanent magnet synchronous motor includes:

Acquiring the first sub-compensated phase angle according to the control interruption period and the current angular velocity of the rotor of the high-power direct-drive permanent magnet synchronous motor;

Acquiring the second sub-compensated phase angle according to the modulated carrier cycle and the current angular velocity of the rotor of the high-power direct-drive permanent magnet synchronous motor;

Obtaining the third sub-compensated phase angle according to the current angular velocity of the rotor of the high-power direct-drive permanent magnet synchronous motor;

Obtaining the compensation phase angle of the high-power direct-drive permanent magnet synchronous motor according to the first sub-compensation phase angle, the second sub-compensation phase angle, and the third sub-compensation phase angle;

The obtaining the first sub-compensated phase angle according to the control interruption period and the current angular velocity of the rotor of the high-power direct-drive permanent magnet synchronous motor includes:

Obtaining the first phase angle delay corresponding to the first sub-compensated phase angle according to the control interruption period;

Acquiring the first sub-compensated phase angle according to the first phase angle delay and the current angular velocity of the rotor of the high-power direct-drive permanent magnet synchronous motor;

The obtaining the second sub-compensated phase angle according to the modulated carrier cycle and the current angular velocity of the rotor of the high-power direct-drive permanent magnet synchronous motor includes:

Obtaining the second phase angle delay corresponding to the modulation output according to the modulation carrier period;

According to the modulation interruption period of the modulation algorithm, obtain the third phase angle delay corresponding to the modulation calculation;

Acquiring the second sub-compensated phase angle according to the second phase angle delay, the third phase angle delay, and the current angular velocity of the rotor of the high-power direct-drive permanent magnet synchronous motor;

Before acquiring the third sub-compensated phase angle according to the current angular velocity of the rotor of the high-power direct-drive permanent magnet synchronous motor, the method further includes:

Obtaining the stable operating angular velocity range of the high-power direct-drive permanent magnet synchronous motor according to the vector control strategy of the high-power direct-drive permanent magnet synchronous motor;

According to the given value of the d-axis current and the given value of the q-axis current, a plurality of first d-axis currents, a plurality of first q-axis currents, and each of the first d-axis currents corresponding to the stable operating angular velocity range are acquired The d-axis voltage and the q-axis voltage corresponding to each of the first q-axis currents;

The obtaining the third sub-compensated phase angle according to the current angular velocity of the rotor of the high-power direct-drive permanent magnet synchronous motor includes:

Acquiring the transmission error phase angle corresponding to each first angular velocity according to the d-axis voltage corresponding to each of the first d-axis currents and the q-axis voltage corresponding to each of the first q-axis currents;

Acquiring the third sub-compensated phase angle according to the transmission error phase angle corresponding to each of the first angular speeds, and the current angular speed of the rotor of the high-power direct-drive permanent magnet synchronous motor;

The obtaining the current actual control phase angle according to the compensation phase angle includes:

Obtain the current position phase angle of the rotor of the high-power direct-drive permanent magnet synchronous motor;

Acquiring the actual position phase angle of the rotor according to the current position phase angle, the initial position phase angle of the rotor, and the compensation phase angle;

Obtain the current actual control phase angle according to the actual position phase angle of the rotor and the modulation phase angle, where the modulation phase angle is calculated by a modulation algorithm according to the given value of the d-axis voltage and the given value of the current q-axis voltage;

The online correction of the current actual control phase angle according to the proportional deviation and integral deviation of the current expected control phase angle and the current actual control phase angle includes:

Acquiring the proportional deviation and the integral deviation according to the current expected control phase angle and the current actual control phase angle;

According to the linear combination of the proportional deviation and the integral deviation, obtain the correction term of the current actual control phase angle;

Perform online correction on the current actual control phase angle according to the correction item.
The megawatt direct-drive permanent magnet electric drive system for electric locomotives according to claim 19, characterized in that the acquired first sub-compensated phase angle is calculated by the following formula:

θ cmps1 = Δ t1 · ω

Where, [omega] is the angular velocity of the current of the direct-drive permanent magnet synchronous motor rotor, a first phase angle Δ t1 to time delay, the first delay phase angle Δ t1 is calculated by the following equation:

Δ t1 = A · T ctrl ≈0.5T ctrl

Among them, T ctrl is a control interruption cycle of the control algorithm;

The second sub-compensation phase angle is calculated by the following formula:

θ cmps2 = Δ t2 · ω

Where ω is the current angular velocity of the rotor of the direct-drive permanent magnet synchronous motor, Δ t2 is the time delay in the PWM pulse output process, and the time delay Δ t2 in the PWM pulse output process is calculated by the following formula:

Δ t2 = B · T PWM + C · T PWM ≈0.75T PWM

Among them, T PWM is the PWM modulation carrier period, B is the modulation algorithm interrupt delay coefficient, and C is the PWM pulse output delay coefficient;

The current expected control phase angle is calculated by the following formula:

Among them, θ ctrl represents the expected control phase angle,
Represents the given value of q-axis voltage,
Denote the given value of d-axis voltage;

The online correction of the current actual control phase angle is calculated by the following formula:

Among them, k p and k i are correction terms, θ ctrl is the current expected phase angle, θ PWM is the current actual phase angle, and f Δ is the fundamental frequency compensation term;

Obtaining the stable operating angular velocity range of the direct-drive permanent magnet synchronous motor is calculated by the following formula:

Where u d is the d-axis voltage corresponding to any first preset angular velocity, u q is the q-axis voltage corresponding to any first preset angular velocity, R s is the resistance of the rotor, and L q is any first preset D-axis inductance corresponding to angular velocity, L d is the q-axis inductance corresponding to any first preset angular velocity, i d is the first d-axis current corresponding to the d-axis voltage, and i q is the first q-axis current corresponding to the q-axis voltage , Ψ f is the back-EMF of the permanent magnet flux linkage;

The phase angle θ Δ of the transmission error is calculated by the following formula:

θ Δ = tan -1 (u d / u q )

The obtained third sub-compensation phase angle θ cmps3 is calculated by the following formula:

θ cmps3 = k · ω.
The megawatt direct drive permanent magnet electric drive system for electric locomotive according to claim 1, characterized in that

The electric locomotive further includes: at least four high-power direct-drive permanent magnet synchronous motors; the at least four high-power direct-drive permanent magnet synchronous motors include: a first motor, a second motor, a third motor, and a fourth motor;

The control method further includes:

Collect the rotor frequencies of the first motor, the second motor, the third motor and the fourth motor to obtain the real-time torque of the first motor, the first motor and the second motor are the shaft motors of the first bogie , The third motor and the fourth motor are shaft motors of a second bogie, and the first bogie is adjacent to the second bogie;

Determine the rotor frequency difference and the rotor frequency differential value of the first motor according to the collected rotor frequencies of the multiple motors;

Determine the amount of torque reduction according to the rotor frequency difference, rotor frequency differential value and real-time torque of the first motor;

The torque of the first motor is adjusted according to the torque reduction amount.
The megawatt direct drive permanent magnet electric drive system for electric locomotive according to claim 21, further comprising:

Generating a sanding control signal according to the rotor frequency difference, rotor frequency differential value and real-time torque of the first motor, the sanding control signal is used to indicate whether to perform sanding operation;

The determining the amount of torque reduction according to the rotor frequency difference, rotor frequency differential value, and real-time torque of the first motor includes:

According to the rotor frequency difference of the first motor and the preset rotor frequency difference level rules, determine the idling coasting level corresponding to the rotor frequency difference of the first motor;

Determine the first torque reduction amount according to the idling coasting level corresponding to the rotor frequency difference of the first motor and the real-time torque of the first motor;

According to the rotor frequency differential value of the first motor and the preset rotor frequency differential value classification rules, determine the idling coasting level corresponding to the rotor frequency differential value of the first motor;

Determine the second torque reduction amount according to the idling coasting level corresponding to the rotor frequency differential value of the first motor and the real-time torque of the first motor;

If the first torque reduction amount is greater than or equal to the second torque reduction amount, it is determined that the first torque reduction amount is the torque reduction amount;

If the first torque reduction amount is smaller than the second torque reduction amount, it is determined that the second torque reduction amount is the torque reduction amount;

The adjusting the torque of the first motor according to the torque reduction includes:

Reduce the torque value of the first motor from the first value to the second value within a first preset time period, and the difference between the first value and the second value is the torque reduction amount ;

Keep the torque value of the first motor unchanged from the second value during the second preset time period;

Increasing the torque value of the first motor from the second value to a preset percentage of the preset torque value within a third preset time period;

Increase the torque value of the first motor to the preset torque value within a fourth preset time period;

Wherein, the recovery rate of the torque value of the first motor in the third preset time period is greater than the recovery rate of the torque value of the first motor in the fourth preset time period;

The reducing the torque value of the first motor from the first value to the second value within the first preset time period includes:

Within a first preset time period, the torque value of the first motor is gradually reduced from the first value to the second value according to the decreasing rate of the torque value of the first motor;

According to the collected rotor frequencies of the plurality of motors, determining the rotor frequency difference and the rotor frequency differential value of the first motor includes:

Limiting filtering and low-pass filtering of the rotor frequency of multiple motors collected;

Determine the rotor frequency difference and the rotor frequency differential value of the first motor according to the rotor frequencies of the plurality of motors after limit filtering and low-pass filtering;

If the locomotive is in the idle mode, the amplitude filtering and low-pass filtering of the collected rotor frequencies include:

Acquiring the current value of the first motor;

Determine the rotor frequency compensation coefficient of each motor according to the current value of the first motor and the rotor frequency of each motor;

Compensate the rotor frequency of each motor according to the rotor frequency compensation coefficient of each motor;

Limiting filtering and low-pass filtering are performed on the compensated rotor frequencies of the plurality of motors.
The megawatt direct drive permanent magnet electric drive system for electric locomotive according to claim 1, characterized in that

The megawatt direct-drive permanent magnet electric drive system for electric locomotives further includes: a plurality of sensors; the plurality of sensors includes at least one or more of the following: input current sensor, intermediate voltage sensor, ground voltage sensor, chopping Wave branch current sensor, motor U-phase current sensor, motor V-phase current sensor, motor stator winding temperature sensor and motor speed sensor;

The megawatt direct drive permanent magnet electric drive system for electric locomotives is also used to: obtain data collected by the multiple sensors;

Judging whether at least one single item state corresponding to the multiple sensors is normal according to the data and preset conditions;

If there is an abnormal single-item state, the state of the abnormal single-item state is placed in the fault bit.
The megawatt direct drive permanent magnet electric drive system for electric locomotives according to claim 23, characterized in that

An input current sensor is provided on the current input terminal, wherein the single state corresponding to the input current sensor is the input current;

Obtaining the data collected by the sensor includes:

Acquiring the first current collected by the input current sensor;

According to the data and preset conditions, determining whether at least one single item state corresponding to the sensor is normal includes:

If the duration that the first current is greater than the first preset threshold is greater than the first preset time, it is determined that the input current of the traction converter is excessive;

An intermediate voltage sensor and a ground voltage sensor connected in parallel with the bus capacitor, wherein the single state corresponding to the intermediate voltage sensor is the intermediate DC bus voltage, and the single state corresponding to the ground voltage sensor is the working state of the ground voltage sensor;

Obtaining the data collected by the sensor includes:

Acquiring a first voltage collected by the intermediate voltage sensor and a second voltage collected by the ground voltage sensor;

According to the data and preset conditions, determining whether at least one single item state corresponding to the sensor is normal includes:

If the duration that the first voltage is greater than the second preset threshold is greater than the second preset time, it is determined that the intermediate DC bus voltage of the traction converter is too large;

If the duration that the first voltage is less than the third preset threshold is greater than the third preset time, it is determined that the intermediate DC bus voltage of the traction converter is too small;

If the second voltage value is not within the first preset range, it is determined that the ground voltage sensor is faulty;

The method also includes:

If the first voltage is not within the second preset range, it is determined that the intermediate voltage sensor is faulty;

If the duration that the third voltage obtained by subtracting half of the first voltage from the second voltage is greater than the fourth preset threshold is greater than the fourth preset time, it is determined that the positive pole of the bus of the traction converter is grounded;

If the duration that the third voltage is less than the fifth preset threshold is greater than the fifth preset time, it is determined that the negative pole of the bus of the traction converter is grounded;

A chopping branch current sensor is provided on the chopping branch, wherein the single state corresponding to the chopping branch current sensor is the chopping branch current;

Obtaining the data collected by the sensor includes:

Acquiring a second current collected by the chopper branch current sensor;

According to the data and preset conditions, determining whether at least one single item state corresponding to the sensor is normal includes:

If the chopper branch is opened, and the duration that the second current is greater than the sixth preset threshold is greater than the sixth preset time, it is determined that the chopper branch current of the traction converter is excessive.
The megawatt direct drive permanent magnet electric drive system for electric locomotives according to claim 24, wherein the method further comprises:

If the chopper branch is not opened, and the duration that the second current is greater than the seventh preset threshold is greater than the seventh preset time, it is determined that the chopper circuit of the traction converter is not opened but the current is detected;

If the chopper branch is opened, and the second current is not detected to be greater than the eighth preset threshold within the eighth preset time, it is determined that the chopper branch of the traction converter is opened but no current is detected;

A motor U-phase current sensor, a motor V-phase current sensor, a motor stator winding temperature sensor and a motor speed sensor are provided at the current output end, wherein the single state corresponding to the motor U-phase current sensor is the motor U-phase input current, the The single-state corresponding to the motor V-phase current sensor is the V-phase input current of the motor, the single-state corresponding to the motor stator winding temperature sensor is the motor stator winding temperature, and the single-state corresponding to the motor speed sensor is the motor speed;

Obtaining the data collected by the sensor includes:

Obtain the third current collected by the U-phase current sensor of the motor, obtain the fourth current collected by the V-phase current sensor of the motor, obtain the temperature collected by the temperature sensor of the stator winding of the motor, and obtain the speed sensor of the motor The first speed obtained

According to the data and preset conditions, determining whether at least one single item state corresponding to the sensor is normal includes:

If the duration of the third current being greater than the ninth preset threshold is greater than the ninth preset time, it is determined that the U-phase input current of the motor is excessive;

If the duration of the fourth current being greater than the tenth preset threshold is greater than the tenth preset time, it is determined that the V-phase input current of the motor is excessive;

If the duration that the temperature is greater than the eleventh preset threshold is greater than the eleventh preset time, it is determined that the temperature of the motor stator winding is too high;

If the duration that the first speed is greater than the twelfth preset threshold is greater than the twelfth preset time, it is determined that the motor speed is too large;

The method also includes:

If the duration obtained when the fifth current obtained by inverting the value of the third current plus the fourth current is greater than the thirteenth threshold is greater than the thirteenth preset time, it is determined that the W-phase input current of the motor is excessive.