WO2020038016A1 - 短路保护的检测电路和检测方法 - Google Patents

短路保护的检测电路和检测方法 Download PDF

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Publication number
WO2020038016A1
WO2020038016A1 PCT/CN2019/084988 CN2019084988W WO2020038016A1 WO 2020038016 A1 WO2020038016 A1 WO 2020038016A1 CN 2019084988 W CN2019084988 W CN 2019084988W WO 2020038016 A1 WO2020038016 A1 WO 2020038016A1
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mos tube
circuit
short
drain
resistor
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PCT/CN2019/084988
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English (en)
French (fr)
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周阿铖
曾正球
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深圳南云微电子有限公司
广州金升阳科技有限公司
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Publication of WO2020038016A1 publication Critical patent/WO2020038016A1/zh

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    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01R—MEASURING ELECTRIC VARIABLES; MEASURING MAGNETIC VARIABLES
    • G01R31/00—Arrangements for testing electric properties; Arrangements for locating electric faults; Arrangements for electrical testing characterised by what is being tested not provided for elsewhere

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  • the invention relates to a short circuit protection detection circuit for a switching power supply, in particular to a short circuit protection detection circuit with temperature compensation, and a detection method for short circuit protection with temperature compensation.
  • the quality index of switching power supply is usually based on safety and reliability. Like other electronic equipment, short circuit is one of the most serious faults of switching power supply. Whether short circuit protection is reliable is an important factor affecting the reliability of switching power supply. When a short-circuit fault occurs, excessive current will not only shorten the withstand time of the power switching device it flows through, but also cause the switch to quickly turn off and then withstand high voltage transients, which will cause the breakdown of the switch tube. Therefore, the control chip (or control Device) must perform corresponding overcurrent detection, and then take effective protective measures.
  • the control chip 100 also detects the output short-circuit fault by comparing the voltage of FB with a set threshold.
  • the feedback pin connected to the optocoupler output or the auxiliary winding contains the output voltage information, so it can be used to determine whether the output is short-circuited.
  • some power supply structures without feedback loops such as push-pull converters and full-bridge converters, they are not controlled by the loop, and only need to control the chip to provide a complementary power tube gate signal with a certain dead time interval. Just fine.
  • the topology is realized by the control chip 100 providing two alternately driven switching signals to realize the push-pull control of the primary side.
  • the control chip 100 has integrated the switching tubes Q1 and Q2 into Internal to reduce the size of the power supply.
  • the drain voltage during the on-time of the switch tube can be used to detect the output. This voltage includes It can reflect the switch current information of whether the output is short-circuited.
  • the implementation of this solution is limited by two shortcomings. One is affected by the conduction resistance of the positive temperature coefficient of the switch.
  • the drain voltage of the switch in the on state If a fixed reference voltage is used as the short-circuit protection threshold, the drain voltage of the switch in the on state. It will be easier to reach the short-circuit threshold at high temperatures, causing the phenomenon of false triggering of short-circuit protection.
  • the second is that the drain voltage of the switch tube during the turn-off period will be very high (tens to hundreds of volts). Connect this voltage directly to the chip for internal use. The judgment of short circuit protection is easy to damage the low voltage detection circuit.
  • the purpose of the present invention is to provide a short-circuit protection detection method with temperature compensation.
  • the method can be applied to all switching power supply controllers with integrated switching tubes, and a corresponding temperature compensation circuit is designed to solve the problem in the prior art.
  • the controller requires a specific feedback loop and feedback pin. 2) Due to the conduction resistance of the positive temperature coefficient of the switch, it cannot be directly taken from the drain of the switch to reflect the problem of switching current information.
  • Another object of the present invention is to provide a short circuit protection detection circuit applying the above-mentioned short circuit protection detection method with temperature compensation.
  • the present invention provides the following technical measures to achieve: a detection method for temperature-compensated short-circuit protection for detecting output short-circuit and temperature compensation in a switching power converter, including the following steps:
  • the temperature compensation current generation step uses the two currents of the same size generated by the mirror to fall on two bipolar transistors with different current densities, and generates a voltage difference between the base and the emitter of the two bipolar transistors.
  • the voltage difference is proportional to the absolute temperature, and then converted into a positive temperature coefficient current for short circuit protection threshold compensation through a first resistor, and provided to the short circuit protection threshold value generating circuit through an output terminal of the positive temperature coefficient current generating circuit;
  • the short-circuit protection threshold generation step uses a mirror image to drop the positive temperature coefficient current in two ways onto the second and third resistors with different numbers of unit resistors. By designing the ratio of the two positive temperature coefficient currents to the two resistors, The ratio of the unit resistance is matched to generate a short-circuit protection threshold voltage that matches the temperature coefficient of the drain detection voltage of the switch NM1;
  • the detecting and comparing output step detects a drain voltage when the switching tube NM1 is turned on, compares its drain voltage with the short-circuit protection threshold voltage generated by the matching, and outputs a comparison result signal.
  • the positive temperature coefficient current generating circuit uses a mirror image to divide the detection current into two paths and land on two bipolar transistors with different numbers of unit transistors. Then, it is converted into a positive temperature coefficient current for short-circuit protection threshold compensation through the first resistor, and is provided to the short-circuit protection threshold generation circuit through the output of the positive temperature-coefficient current generation circuit; the short-circuit protection threshold generation circuit uses a mirror to convert the positive
  • the positive temperature coefficient current provided by the temperature coefficient current generating circuit is divided into two paths and falls respectively on the second resistor and the third resistor with different numbers of unit resistors.
  • the short-circuit protection threshold value that matches the temperature coefficient of the drain detection voltage of the switch is matched to generate. Voltage.
  • the resistor used to generate the short-circuit protection threshold can be a positive temperature coefficient or a negative temperature coefficient resistor, and only needs to meet the voltage generated by the combination of the resistance and the positive temperature coefficient current, and the drain of the switch at the critical point of the short-circuit protection.
  • the voltage temperature curve can be matched to the maximum.
  • it may further include a startup step, during the power-on period of the power supply, before the start point is reached, the detection circuit is disabled; when the power supply voltage reaches the start point, the subsequent circuits are enabled and the start circuit is turned off without affecting the remaining circuits normal work.
  • the present invention needs to detect is the drain voltage of the switch tube during the on phase, and during its off phase, the drain voltage is connected to the chip peripheral coil voltage, which is usually at a high level (tens to hundreds of volts) ), In order to prevent high-voltage breakdown of low-voltage devices in the chip short-circuit detection circuit at this stage, high-voltage devices need to be used for clamping and isolation between the drain voltage and the short-circuit protection comparator.
  • the invention also provides a short-circuit protection detection circuit applying the above-mentioned short-circuit protection detection method with temperature compensation, which includes a start circuit, a positive temperature coefficient current generation circuit, a short-circuit protection threshold value generation circuit, a switch tube, and a comparator. .
  • the start-up circuit In the start-up circuit, during the power-on period, before the start point is reached, it is ensured that the subsequent current mirror is turned off and the detection circuit does not work; when the power supply voltage reaches the start point, the current mirror is provided with a bias voltage and the subsequent circuit is enabled, and the branch is started Shutdown does not affect the normal operation of the remaining circuits;
  • the positive temperature coefficient current generating circuit uses two currents of the same size generated by mirror images to fall on bipolar transistors with different numbers of parallel connections to construct the base-emitter difference voltage of the two bipolar transistors. It is then converted into a positive temperature coefficient current for short-circuit protection threshold compensation through resistance;
  • the short-circuit protection threshold generation circuit drops different proportions of the above-mentioned positive temperature coefficient currents to different numbers of resistors, and designs a specific ratio of the current ratio and the number of resistors to match the temperature coefficient of the switch-source drain-source voltage. Short-circuit protection threshold voltage;
  • a high-voltage device In order to prevent the high-voltage breakdown of the low-voltage device in the chip short-circuit protection threshold generation circuit during the switch-off period, a high-voltage device needs to be used for clamping and isolation between the drain potential of the switch and the short-circuit protection comparator.
  • the switching tube is a power MOS tube integrated in a chip for switching power supply switching function.
  • the voltage of its drain contains current information flowing through the switching tube, which can be used to reflect whether the output occurs.
  • the comparator is used to compare the drain voltage of the switch with the short-circuit protection threshold voltage generated above, and output a comparison result signal, which is processed by the chip's subsequent logic to complete the short-circuit protection function.
  • the invention further provides a short circuit protection detection circuit with temperature compensation, which includes three ports of power supply, ground and the drain of the switching tube, and a starting circuit, a positive temperature coefficient current generating circuit, a short circuit protection threshold generating circuit, a switching tube and a comparison circuit. Five parts.
  • the starting circuit is composed of three PMOS tubes and a resistor.
  • the sources of the first and second PMOS transistors are both connected to the power supply terminal.
  • the gate of the first PMOS transistor is used as a bias voltage supply terminal to provide a bias voltage for subsequent current mirrors.
  • the drain of the second PMOS transistor is used as The power-on protection shutdown control terminal is used to control the detection circuit not to work during power-on; the drain of the first PMOS tube and the gate of the second PMOS tube are connected to the source of the third PMOS tube, and the third PMOS
  • the gate and drain of the tube are connected to one end of the startup resistor together, and the other end of the startup resistor is connected to the ground.
  • the positive temperature coefficient current generating circuit is composed of two PMOS tubes, two NMOS tubes, two bipolar NPN transistors, and a resistor.
  • the sources of the fourth and fifth PMOS transistors are both connected to the power supply terminal, and the gates of the fourth and fifth PMOS transistors are connected together, and the drive control terminal of the first current mirror is used to receive the bias provided by the startup circuit.
  • the drive control terminal of the first current mirror is also connected to the drain of the fifth PMOS tube and the drain of the fourth NMOS tube, and is then led out as an output terminal of the positive temperature coefficient current generating circuit for the subsequent current mirror
  • the gate of the fourth NMOS tube, the drain of the fourth PMOS tube, and the drain and gate of the third NMOS tube are connected together and led out as a positive temperature
  • the driving control terminal of the coefficient current generating circuit is configured to receive a power-on protection shutdown control signal provided by the starting circuit.
  • the source of the third NMOS tube is connected to the collector and the base of the first bipolar NPN transistor.
  • the source of the four NMOS transistor is connected to the collector of the second bipolar NPN transistor.
  • the base of the first bipolar NPN transistor is connected to the base of the second bipolar NPN transistor.
  • Emitter connected to first resistor Transmitting end, the other end of the first resistor and a first bipolar NPN transistor are connected to the ground terminal.
  • the second bipolar NPN transistor is obtained by connecting two or more bipolar NPN transistors in parallel.
  • the short-circuit protection threshold generating circuit is composed of two PMOS tubes, one high-voltage NMOS tube, two capacitors, and two resistors.
  • the sources of the sixth and seventh PMOS tubes are connected to the power source, the gates of the sixth and seventh PMOS tubes are connected together, and the output of the positive temperature coefficient current generating circuit is also connected to receive the current generated by the positive temperature coefficient current generating circuit.
  • Positive temperature coefficient current, the drain of the sixth PMOS is connected to one end of the first capacitor and one end of the second resistor to a first output node.
  • the first output node is used to provide a matching short-circuit protection threshold to the comparator.
  • the gate is connected to the power terminal, and the drain terminal of the drain detection switch of the high-voltage NMOS transistor.
  • the high-voltage NMOS transistor is a high-voltage isolation device designed to prevent the high-voltage drain of the switch from penetrating the second capacitor, the low-voltage seventh PMOS transistor, and the input device of the subsequent comparator during the off-time of the switch. It is selected according to the maximum potential of the drain terminal of the switching tube, and its gate is connected to the power supply potential, which plays a role of clamping the source potential of the high-voltage NMOS tube not higher than the power supply voltage level here.
  • the switch tube is composed of a plurality of units of NMOS tubes connected in parallel, and has the characteristics of large size, small on-resistance and large current.
  • the gate of the N switch tube receives the gate control signal generated inside the control chip, the drain of the switch tube is used as an independent terminal, and the source of the switch tube is grounded.
  • the comparator is a common comparator module based on a CMOS device design.
  • a non-inverting input terminal of the comparator is connected to a second output voltage of the short-circuit protection threshold generating circuit, and a negative-phase input terminal of the comparator is connected to the short-circuit protection threshold generating circuit.
  • the first output voltage of the circuit and the output voltage of the comparator are used as the output of the entire short-circuit detection circuit for subsequent modules.
  • the signal of the output node is also referred to as a short-circuit protection determination signal.
  • the temperature compensation design solves the short-circuit protection false triggering phenomenon under high temperature: If a fixed reference voltage is used as the short-circuit protection threshold, the conduction internal resistance of the positive temperature coefficient will affect the drain voltage of the switch tube at high temperature. Short-circuit threshold reached.
  • the invention designs a short-circuit protection threshold having the same temperature coefficient as the drain voltage of the power tube to prevent the phenomenon of short-circuit protection from being triggered by mistake at high temperatures;
  • FIG. 1 is a schematic block diagram of a circuit for implementing short-circuit protection in a secondary-side feedback system in the prior art
  • FIG. 2 is a schematic block diagram of a circuit for realizing short circuit protection in a prior art primary side feedback system
  • FIG. 3 is a circuit schematic diagram of a conventional push-pull converter
  • FIG. 4 is a circuit schematic diagram of a short-circuit protection detection circuit 101 according to an embodiment of the present invention.
  • FIG. 5 is a detailed schematic diagram of the structure of the transistor B2 of the positive temperature coefficient current generating circuit in the short-circuit protection detection circuit according to the embodiment of the present invention.
  • FIG. 6 is a graph showing a change of an index related to short circuit protection with temperature according to an embodiment of the present invention.
  • FIG. 3 is a schematic diagram of an application circuit of the push-pull controller 100 to which the short-circuit protection detection method of the present invention is applied.
  • the pin VDD of the controller 100 (hereinafter also referred to as the power supply terminal VDD) is connected to the switching power supply input voltage VIN
  • the GND terminal of the controller 100 is connected to the ground
  • the VD1 terminal is connected to the power transistor inside the controller 100
  • the drain of Q1 is connected externally to one end of winding NP1.
  • VD2 is internally connected to the drain of power tube Q2, externally connected to one end of winding NP2, and VIN to the other end of winding NP1 and winding NP2. Connect at the other end.
  • windings NS1 and NS2 are connected to the output terminal VOUT through diodes D1 and D2 respectively.
  • the other end of windings NS1 and NS2 is connected to the output negative terminal, that is, the secondary ground potential.
  • a capacitor Co is connected between output VOUT and the secondary ground potential.
  • resistance Ro resistance
  • FIG. 4 is a schematic diagram of the short-circuit protection detection circuit 101 in the controller 100 according to an embodiment of the present invention.
  • the other circuits of the controller 100 such as a low-voltage power supply VCC generating circuit and a driving signal GATE generating circuit, have a variety of circuit structures, and are not related to the present invention. They are not described below, and are not shown in FIG. 4.
  • the low-voltage power supply VCC is the working voltage of the chip obtained by stepping down the input voltage VDD of the converter, that is, the voltage source for supplying power to other sub-modules in the controller 100.
  • the low-voltage power source VCC 5V selected in the embodiment.
  • the two switching tubes Q1 and Q2 integrated in the controller 100 have the same structure in terms of electrical characteristics and size, two identical short-circuit protection detection circuits are also applied to the drains of the two switching tubes at the same time. Take one of the switching tubes Q1 as an example for detailed description.
  • the short-circuit protection detection circuit with temperature compensation includes three ports of a power supply terminal VCC, a ground terminal GND, and a drain terminal VD1 (hereinafter referred to as VD terminal) of the switch Q1, and further includes a start-up circuit 11, There are five parts: a positive temperature coefficient current generating circuit 12, a short-circuit protection threshold generating circuit 13, a switching tube unit 14, and a comparator unit 15.
  • the switching tube Q1 is an NMOS tube, and the device number of NM1 is used in FIG. 4, which may also be referred to as the switching tube NM1 as follows.
  • the starting circuit 11 is composed of three PMOS transistors PM1, PM2, and PM3, and a resistor R1.
  • the sources of the PMOS tubes PM1 and PM2 are connected to the power supply terminal VCC.
  • the gate potential V1 of PM1 is used as a control signal of the positive temperature coefficient current generating circuit 12 to provide a bias voltage for the subsequent current mirror.
  • the drain potential V2 of PM2 As another control signal of the positive temperature coefficient current generating circuit 12, it is used to control the detection circuit not to work during power-on; the drain of PM1 and the gate of PM2 are connected to the source of PM3 and the gate of PM3. After being connected to its drain, it is connected to one end of the resistor R1, and the other end of the resistor R1 is connected to the ground terminal GND.
  • the MOS tube PM2 is turned on so that the V2 potential is close to a high level VCC, so that the current mirror in the positive temperature coefficient current generating circuit 12 cannot be turned on and the detection circuit does not work; until the gate voltage of the MOS tube PM2 is VTH PM3 + V R1 and VCC The voltage difference is not enough to turn on the MOS tube PM2, that is, after the power supply voltage reaches the starting point, the subsequent state of the current mirror is turned off and the subsequent circuit of the startup circuit is enabled.
  • the gate voltage V1 of the MOS tube PM1 that is turned on is the subsequent The current mirror provides a bias voltage; at this time, the MOS tube PM2 of the startup branch is turned off, and the startup branch is turned off, which does not affect the normal operation of the remaining circuits.
  • the positive temperature coefficient current generating circuit 12 is composed of two PMOS transistors PM4 and PM5, two NMOS transistors NM3 and NM4, two bipolar NPN transistors B1 and B2, and a resistor R2. Among them, the PMOS transistors PM4 and PM5 are currents. Mirror structure and consistent size.
  • the sources of the PMOS tubes PM4 and PM5 are connected to the power supply terminal VCC, and the gates of the PMOS tubes PM4 and PM5 are connected together, and the drive control terminal as the first current mirror is used to receive the control signal V1 provided by the startup circuit 11;
  • the driving control end of a current mirror is also connected to the drain of the PMOS tube PM5 and the drain of the NMOS tube NM4, and then led out as an output terminal of a positive temperature coefficient current generating circuit for providing a subsequent current mirror with a positive temperature coefficient.
  • the positive temperature coefficient current generated by the current generation circuit; the gate of the NMOS tube NM4, the gate and drain of the NM3 are connected to the drain of the PM4 transistor P4, and the drive control terminal of the positive temperature coefficient current generation circuit is derived.
  • the source of NM3 is connected to the collector and base of bipolar NPN transistor B1, the source of NM4 is connected to the collector of bipolar NPN transistor B2, and the base of transistor B1
  • the electrode is connected to the base of the transistor B2.
  • the emitter of the transistor B2 is connected to one end of the resistor R2.
  • the other end of the resistor R2 and the emitter of the transistor B1 are connected to the ground GND.
  • FIG. 5 a detailed diagram of a positive temperature coefficient current generating circuit, wherein the transistor B2 is composed of two or more unit bipolar NPN transistors B21, B22,... B2N connected in parallel.
  • This circuit part constructs the base-emitter voltage difference of two bipolar transistors B1 and B2 working at different current densities. This voltage difference is proportional to the absolute temperature. Temperature coefficient current for short circuit protection threshold compensation. Combine the detailed circuit diagram of the temperature compensation current shown in Figure 5 to analyze the current working formula to derive the specific working formula:
  • I C I S exp (V BE / V T )
  • V T kT / q
  • V T is the thermal voltage of the transistor. Generally speaking, the corresponding thermal voltage is 26 mV at 300 K, and Is is the saturation current of the transistor.
  • the saturation current I S is proportional to Among them ⁇ is the mobility of minority carriers, and n i is the intrinsic carrier concentration of silicon. These parameters are all related to temperature. q is the electronic charge, T is the temperature, and k is a constant.
  • V BE V T ln (I C / I S )
  • the base-emitter voltage difference of the transistors B1 and B2 can be expressed as:
  • the ratio is proportional to the ratio of the number of tubes, so:
  • the positive temperature coefficient current generated by the resistance R2 for short-circuit protection threshold compensation is:
  • N is a ratio of the number of unit transistors constituting the transistor B2 to the number of unit transistors of the transistor B1.
  • R2 represents the resistance value of the resistor R2 in the positive temperature coefficient current generating circuit 12.
  • the selection value of N can be selected based on factors such as layout matching and short-circuit protection threshold setting value.
  • the recommended value of N in the circuit of the present invention is 3, that is, transistor B1 selects 1 unit transistor, and transistor B2 selects 3 unit transistors.
  • the short-circuit protection threshold generating circuit 13 is composed of two PMOS transistors PM6 and PM7, one high-voltage NMOS transistor NM2, two capacitors C1, C2, and two resistors R3 and R4.
  • the PMOS tubes PM6 and PM7 are the mirror branches of the first current mirror
  • the sources of the PMOS tubes PM6 and PM7 are connected to the power terminal VCC
  • the gates of the PMOS tubes PM6 and PM7 are connected together as the drive control terminals of the mirror branches.
  • the positive temperature coefficient current generated by the positive temperature coefficient current generating circuit 12 for short-circuit protection threshold compensation is mirrored into currents I1 and I2 through the PMOS tubes PM6 and PM7; the drain of the PMOS tube PM6 and one end of the capacitor C1 and the resistor R3 One end is connected to an output node VA, and the output node VA is used to provide a short-circuit protection threshold with temperature compensation to the negative input terminal of the comparator; the other ends of the capacitor C1 and resistor R3 are connected to the ground terminal GND, and the PMOS transistor PM7 The drain and one end of the capacitor C2 and one end of the resistor R4 are connected to an output node VB.
  • the output node VB is used to provide the detection voltage of the drain of the switch NM1 to the non-inverting input of the comparator; One end is connected to the ground GND, the other end of the resistor R4 is connected to the source of the high-voltage NMOS tube NM2, the gate of the NMOS tube NM2 is connected to the power terminal VCC, and the drain of the NMOS tube NM2 is connected to the drain of the switch NM1 for detection The voltage of the drain terminal VD of the switch NM1.
  • the NMOS tube NM2 is a high-voltage isolation device designed to prevent the high-voltage breakdown capacitor C2 of the drain of the switch NM1 and the low-voltage PMOS tube PM7 and the input device of the subsequent comparator during the off-time of the switch NM1.
  • the value is selected according to the maximum potential of the drain terminal VD of the power switch NM1, and its gate is connected to the VCC potential, which plays a role of clamping the source potential of the NM2 not higher than the power supply voltage VCC level.
  • the current generated by the above-mentioned positive temperature coefficient current generating circuit 12 is mirrored as currents I1 and I2 through the PMOS tubes PM6 and PM7, and respectively falls on the resistors R3 and R4, where the resistors R3 and R4 are composed of different numbers of unit resistors resistance.
  • the unit resistance is the root resistance.
  • the root resistance is a single resistance with a fixed aspect ratio.
  • the resistors R2, R3, and R4 in the circuit of FIG. 4 are based on different numbers of resistors connected in series and parallel.
  • the advantage of using a root resistor design is that in the process of voltage-current-voltage conversion, the resistance ratio can offset the process deviation. Assume:
  • X and Y are constant values.
  • the short-circuit protection threshold voltage that is close to the temperature coefficient of the drain-source voltage of the switch tube is matched;
  • R3 and R4 resistors can be positive temperature coefficients or negative temperature coefficient resistors, and only need to meet the resistance and positive temperature
  • the voltage generated by the combination of the coefficient current and the temperature curve of the drain voltage of the switch at the critical point of short-circuit protection can be matched to the maximum. .
  • the switching tube 14 is a MOS tube NM1 composed of a plurality of unit NMOS tubes connected in parallel, and has the characteristics of large size, small on-resistance and large current.
  • the gate of the MOS tube NM1 receives the control signal GATE generated inside the control chip.
  • the drain of the MOS tube NM1 is the drain terminal VD of the switching tube NM1.
  • the source of the MOS tube NM1 is connected to the ground terminal GND.
  • the switching tube NM1 is integrated in the chip as the main power MOS tube for the switching power switch.
  • the voltage VD of the drain contains the current I SW information flowing through the switching tube NM1, which can be used to reflect the output. Whether a short circuit has occurred.
  • the comparator 15 is a common comparator module based on a CMOS device.
  • the non-inverting input terminal is connected to the output node VB of the short-circuit protection threshold generating circuit 13 and is used to receive the detection voltage of the drain of the switch NM1.
  • the negative-phase input terminal is connected to the output node VA of the short-circuit protection threshold generating circuit 13 for receiving the short-circuit protection threshold with temperature compensation.
  • the signal of the comparator output node is also referred to as a short-circuit protection determination signal OCP_H.
  • the comparator is used to compare the drain voltage of the switch NM1 with the short-circuit protection threshold voltage generated above, and output a comparison result signal OCP_H, which is processed by the chip's subsequent logic to complete the short-circuit protection function.
  • the threshold composition of the short circuit protection is derived as follows:
  • V B -V A (I2 * R4 + V D )-(I1 * R3-V S )
  • I is the unit value of I1 or I2 current
  • R is the root resistance value of resistor R3 or R4
  • V TH_OCP V TH_OCP
  • I is a current with a positive temperature coefficient. Adjust the values of X and Y and the size of R to match the temperature coefficient of the drain-source voltage of the switch. The value is the same as the value of the drain voltage of the switch at the critical point of the short circuit. Consistent short-circuit protection threshold voltage.
  • the change curve shown in FIG. 6 is a simulation diagram of a change curve of the index related to short circuit protection with temperature according to an embodiment of the present invention.
  • the curve V TH_OCP is a curve of the short-circuit protection threshold value versus temperature obtained by the circuit design of the embodiment of the present invention
  • the curve VD_mos is a compensation object of a short-circuit protection detection circuit with temperature compensation provided by the present invention—a certain The curve of the voltage at the drain of the switching tube as a function of the temperature at the critical point of the short-circuit protection.
  • This curve is obtained by performing a temperature scan on the switching tube at a fixed gate voltage and a fixed short-circuit current; the curve ⁇ V is based on The curve of the difference between the short-circuit protection threshold value and the compensation object, that is, the voltage at the drain terminal of the switch tube, as a function of temperature, as designed by the embodiment of the invention.
  • the horizontal axis is the temperature T, and the vertical axis is the voltage V.
  • the threshold voltage V TH_OCP generated by the short-circuit protection detection circuit with temperature compensation designed by the present invention is consistent with the temperature coefficient trend of the drain voltage of the switching tube, and the In the temperature design range, V TH_OCP is also close to the drain voltage VD_mos of the switch at the critical point of short-circuit protection. From the ⁇ V curve, it can be seen that the deviation between V TH_OCP and VD_mos in the entire temperature range is not greater than 30mV. That is, it is less than 10% of the threshold center value of 292mV at typical temperature, which meets the design requirements and well follows the temperature coefficient of the drain voltage of the switch. However, the prior art does not perform temperature compensation here.
  • the short-circuit protection threshold compared with the drain of the switch is a constant value. When expressed by a change curve, the curve of the short-circuit protection threshold with temperature is a level in the entire temperature range. straight line.

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Abstract

一种短路保护的检测电路(101)，包括启动电路(11)和比较器(15)，还包括正温度系数电流生成电路(12)和短路保护阈值生成电路(13)，所述正温度系数电流生成电路(12)，利用镜像生成的两路相同大小的电流分别降落在电流密度不同的两个双极性晶体管上，产生两个双极性晶体管的基极-发射极电压的差值，该电压的差值与绝对温度成正比，然后通过电阻转化成短路保护阈值补偿的正温度系数电流；所述短路保护阈值生成电路(13)，利用镜像将两路正温度系数电流降落在单位电阻构成数量不同的两个电阻上，通过设计两路正温度系数电流的比例与两个电阻的单位电阻比例，匹配生成与开关管NM1漏极检测电压温度系数吻合的短路保护阈值电压。

Description

短路保护的检测电路和检测方法 技术领域
本发明涉及一种开关电源的短路保护检测电路，特别是一种具有温度补偿的短路保护的检测电路，以及一种具有温度补偿的短路保护的检测方法。
背景技术
评价开关电源的质量指标通常以安全性和可靠性为第一原则。同其他电子设备一样，短路是开关电源最严重的故障之一，短路保护是否可靠，是影响开关电源可靠性的重要因素。短路故障发生时，过大的电流不仅会缩短其流过的功率开关器件的承受时间，而且会使开关快速关断然后承受瞬态高压导致开关管的击穿失效，因此控制芯片(或称控制器)必须进行相应的过流检测，进而采取有效的保护措施。
为了有效的短路保护检测，前人做过了许多有效设计。如图1所示，副边反馈类的电源发生输出短路后，输出电压V OUT掉低，则可编程精密参考TL431的输出会因为输入的减小而大幅增加，使得光耦10所抽电流急剧减小，控制芯片100的反馈引脚FB的电压升高。控制芯片100通过FB电压与设定阈值的比较实现对输出短路故障的检测。如图2所示，原边反馈类的电源输出电压的变化将通过辅助绕组20和R10、R20分压电阻反馈到控制芯片100的FB引脚，因此发生输出短路后，FB引脚电压随输出Vout的下掉而降低，控制芯片100也是通过FB的电压与设定阈值的比较实现对输出短路故障的检测。针对上述两种有反馈回路的开关电源，与光耦输出或者辅助绕组连接的反馈引脚都包含了输出电压的信息，因此能够用于判定输出是否短路。但对于一些无反馈环路的电源结构，例如推挽式变换器和全桥式变换器等，它们不受环路控制，只需要控制芯片提供具有一定死区时间间隔的互补功率管栅极信号即可。
如图3所示推挽变换器的电路原理图，该拓扑由控制芯片100提供两路交替驱动的开关信号便可实现原边的推挽控制，控制芯片100已将开关管Q1、Q2集成到内部以减小该电源的体积。当输出短路时由于没有反馈引脚可以利用，而副边的短路电流会通过绕组传到原边通路，因此可通过取开关管导通期间的漏极电压来用于检测输出，该电压包含有可以体现输出是否短路的开关管电流信息。但是该方案的施行受到两个缺点的限制，一是受开关管正温度系数的导通内阻影响， 若采用固定的基准电压来做短路保护阈值，导通状态下的开关管的漏极电压在高温下会更容易到达短路阈值，引起误触发短路保护的现象；二是开关管在关断期间漏极电压会很高(几十到几百伏)，直接将该电压接至芯片内部用作短路保护的判断容易损伤低压检测电路。
发明内容
本发明目的在于提供一种具有温度补偿的短路保护的检测方法，该方法可应用于集成有开关管的所有开关电源控制器中，并设计了相应的温度补偿电路，以解决现有技术中1)要求控制器有特定反馈环路和反馈引脚，2)受限于开关管正温度系数的导通内阻影响，不能直接从开关管漏极取电压以反映开关电流信息的问题。
本发明另一个目的在于提供一种应用上述具有温度补偿的短路保护检测方法的短路保护的检测电路。
本发明提供如下技术措施来实现:一种具有温度补偿的短路保护的检测方法，用于对开关电源变换器中输出短路的检测和温度补偿，包括如下步骤：
温度补偿电流生成步骤，利用镜像生成的两路相同大小的电流分别降落在不同电流密度的两个双极性晶体管上，在两个双极性晶体管的基极-发射极上产生电压的差值，该电压差值与绝对温度成正比，然后通过第一电阻转化成用于短路保护阈值补偿的正温度系数电流，通过正温度系数电流生成电路的输出端提供给短路保护阈值生成电路；
短路保护阈值生成步骤，利用镜像将正温度系数电流分两路降落在单位电阻构成数量不同的第二电阻和第三电阻两个电阻上，通过设计两路正温度系数电流的比例与两个电阻的单位电阻构成比例，匹配生成与开关管NM1漏极检测电压温度系数吻合的短路保护阈值电压；
检测和比较输出步骤，检测开关管NM1导通时的漏极电压，将其漏极电压与所述匹配生成的短路保护阈值电压进行比较，并输出比较结果信号。
简而言之，本发明的短路保护的检测电路，所述正温度系数电流生成电路，利用镜像将检测电流分为两路并分别降落在构成的单位晶体管数量不同的两个双极性晶体管上，然后通过第一电阻转化成用于短路保护阈值补偿的正温度系数电流，通过正温度系数电流生成电路的输出端提供给短路保护阈值生成电路；所述 短路保护阈值生成电路，利用镜像将正温度系数电流生成电路提供的正温度系数电流分为两路并分别降落在单位电阻构成数量不同的第二电阻与第三电阻上，匹配生成与开关管漏极检测电压温度系数吻合的短路保护阈值电压。
其中，用于生成短路保护阈值的电阻为正温度系数或者负温度系数电阻均可，只需满足该电阻与正温度系数电流配合生成的电压，与开关管在短路保护临界点处的的漏极电压的温度曲线最大限度的吻合即可。
优选的，还可包括启动步骤，在电源上电期间，到达启动点之前，让检测电路不工作；当电源电压达到启动点后，使能后续电路，并让启动电路关断，不影响其余电路正常工作。
此外，本发明需检测的是开关管处于导通阶段的漏极电压，而在其关断阶段，漏极电压与芯片外围线圈电压连接，通常处于较高的电平(几十到几百伏)，为了防止这个阶段的高压击穿芯片短路检测电路中的低压器件，在漏极电压与短路保护比较器之间需要用高压器件做钳位及隔离。
本发明还提供一种应用上述具有温度补偿的短路保护检测方法的短路保护的检测电路，包括启动电路、正温度系数电流生成电路、短路保护阈值生成电路、开关管和比较器等五个部分组成。
所述启动电路，电源上电期间，到达启动点之前确保后续电流镜关闭，检测电路不工作；当电源电压达到启动点后，为电流镜提供偏置电压并使能后续电路，且启动支路关断，不影响其余电路正常工作；
所述正温度系数电流生成电路，利用镜像生成的两路相同大小的电流降落在不同并联个数的双极性晶体管上，构造出两个双极性晶体管的基极-发射极差值电压，然后通过电阻转化成用于短路保护阈值补偿的正温度系数电流；
所述短路保护阈值生成电路，将不同比例的上述正温度系数电流降落在不同个数的根电阻上，通过设计电流比例与电阻个数的具体比例，匹配出与开关管漏源电压温度系数接近的短路保护阈值电压；
为防止开关管关断期间的高压击穿芯片短路保护阈值生成电路中的低压器件，在开关管漏极电位与短路保护比较器之间需要用高压器件做钳位及隔离。
所述开关管，为集成在芯片之中用于开关电源开关作用的功率MOS管，当开关管导通时，其漏极的电压包含有流经开关管的电流信息，可用于反映输出是否 发生短路；
所述比较器，用于对开关管漏极电压与上述生成的短路保护阈值电压进行比较，并输出比较结果信号，该信号经过芯片后续逻辑处理，完成短路保护功能。
本发明再提供一种具有温度补偿的短路保护的检测电路，包括电源、地和开关管漏极三个端口，以及启动电路、正温度系数电流生成电路、短路保护阈值生成电路、开关管和比较器五个部分。
所述启动电路，由三个PMOS管以及电阻组成。第一和第二PMOS管的源极均接电源端，第一PMOS管的栅极引出作为偏置电压提供端，用于为后续电流镜提供偏置电压，第二PMOS管的漏极引出作为上电保护关断控制端，用于在电源上电期间，控制检测电路不工作；第一PMOS管的漏极、第二PMOS管的栅极与第三PMOS管的源极连接，第三PMOS管的栅极与漏极连接后一起接至启动电阻一端，启动电阻的另一端与地端连接。
所述正温度系数电流生成电路，由两个PMOS管、两个NMOS管、两个双极型NPN晶体管和一个电阻构成。第四和第五PMOS管的源极均接电源端，第四和第五PMOS管的栅极连接在一起，并引出作为第一电流镜的驱动控制端，用于接收启动电路提供的偏置电压；第一电流镜的驱动控制端还与第五PMOS管的漏极和第四NMOS管的漏极连接在一起，再引出作为正温度系数电流生成电路的输出端，用于向后续电流镜提供由正温度系数电流生成电路生成的正温度系数电流；第四NMOS管的栅极、第四PMOS管的漏极与第三NMOS管的漏极和栅极连接在一起，并引出作为正温度系数电流生成电路的驱动控制端，用于接收所述启动电路提供的上电保护关断控制信号，第三NMOS管的源极与第一双极型NPN晶体管的集电极和基极相连，第四NMOS管的源极与第二双极型NPN晶体管的集电极相连，第一双极型NPN晶体管的基极与第二双极型NPN晶体管的基极连接，第二双极型NPN晶体管的发射极接至第一电阻的一端，第一电阻的另一端和第一双极型NPN晶体管的发射极均接至地端。其中，第二双极型NPN晶体管是由两个以上的双极型NPN晶体管并联得到。
所述短路保护阈值生成电路，由两个PMOS管、一个高压NMOS管、两个电容和两个电阻组成。第六和第七PMOS管源极均接电源端，第六和第七PMOS管的栅极连接在一起，还连接正温度系数电流生成电路的输出端，用于接收正温度系数 电流生成电路生成的正温度系数电流，第六PMOS的漏极与第一电容的一端、第二电阻的一端连接于一个第一输出节点，第一输出节点用于提供匹配生成的短路保护阈值给比较器的反相输入端；第一电容与第二电阻的另一端均接至地端，第七PMOS的漏极与第二电容的一端、第三电阻的一端连接于一个第二输出节点，第二输出节点用于提供开关管Q1漏极的检测电压给比较器的正相输入端；第二电容的另一端接至地端，第三电阻的另一端与高压NMOS管的源极相连，高压NMOS管的栅极接电源端，高压NMOS管的漏极检测开关管的漏极端。其中高压NMOS管就是在开关管关断期间，用于防止开关管漏极高压击穿第二电容、低压的第七PMOS管以及后续比较器的输入器件所设计的高压隔离器件，其耐压值根据开关管漏端最大电位选取，其栅极接电源电位，在此处起到钳位高压NMOS管源极电位不高于电源电压电平值的作用。
所述开关管，是由若干单位NMOS管并联组成的，具有尺寸大、导通内阻小和可流经大电流的特性。N开关管的栅极接收控制芯片内部生成的栅极控制信号，开关管的漏极作为一个独立端，开关管的源极接地端。
所述比较器，是常见的基于CMOS器件设计的比较器模块，其正相输入端连接所述短路保护阈值生成电路的第二输出电压，比较器的负相输入端连接所述短路保护阈值生成电路的第一输出电压，比较器的输出电压作为整个短路检测电路的输出，供后续模块采用，在本发明中该输出节点的信号又称为短路保护判定信号。
以上对本发明的电路原理和作用等进行了分析，现将本发明的有益效果进行总结：
1)在各开关电源控制芯片中的通用性：无论开关电源的反馈引脚是何种类型或者有无反馈引脚可以利用，只要控制芯片内部集成开关管，无需特定反馈环路和反馈引脚，本发明的输出短路检测方案都可行；
2)更直接反映开关管的电流情况：短路保护的最终目的就是要保护开关管不受大电流影响而烧坏，从开关管漏极取电压的方式，较之前的方法更能直接反映开关管上的电流信息；
3)温度补偿设计解决高温下短路保护误触发的现象：若采用固定的基准电压来做短路保护阈值，受正温度系数的导通内阻影响，开关管的漏极电压在高温下会更容易到达短路阈值。本发明设计了具有与功率管漏极电压同温度系数的短路 保护阈值，来防止高温下误触发短路保护的现象；
4)巧妙应用高压管的钳位隔离：为防止开关管关断期间的高压击穿芯片短路检测电路中的低压器件，在开关管漏极电位与短路保护比较器设计高压器件做钳位及隔离。
以上为具有温度补偿的短路保护检测电路的原始技术方案，具体的工作原理和相关分析将在下文具体实施方式部分详细描述。所述的电路技术方案与方法方案相对应，各方案或技术特征的原理、作用及带来的有益效果相同，在此不再赘述。
附图说明
图1为现有技术副边反馈系统中实现短路保护的电路原理框图；
图2为现有技术原边反馈系统中实现短路保护的电路原理框图；
图3为现有推挽变换器的电路原理图；
图4为本发明实施例的短路保护检测电路101的电路原理图；
图5为本发明实施例的短路保护检测电路中正温度系数电流生成电路的晶体管B2的构成细节原理图；
图6为本发明实施例的与短路保护相关的指标随温度的变化曲线图。
具体实施方式
为了使本发明的目的、技术方案及优点更加清楚明白，以下结合附图及实施例，对本发明进一步详细说明。应当理解，此处所描述的具体实施例仅用以解释本发明，并不用于限定本发明。
实施例
图3所示为应用了本发明短路保护检测方法的推挽控制器100的应用电路示意图。如图3所示，控制器100的引脚VDD(下文又称为供电端VDD)与开关电源输入电压VIN连接，控制器100的GND端与地连接，VD1端在控制器100内部与功率管Q1的漏极连接，外部与绕组NP1的一端连接，同样的，VD2端在内部与功率管Q2的漏极连接，外部与绕组NP2的一端连接，VIN端与绕组NP1的另一端、绕组NP2的另一端连接。绕组NS1、NS2的一端分别经二极管D1、D2与输出端VOUT连接，绕组NS1、NS2的另一端与输出负端即副边地电位连接，此外输出VOUT与副边地电位之间还并联电容Co和电阻Ro。其中，绕组NP1、NP2、NS1和NS2的 匝数相等，即NP1＝NP2＝NS1＝NS2。
图4所示为本发明实施例的控制器100中的短路保护检测电路101的原理图。控制器100的其它电路，例如低压电源VCC产生电路，驱动信号GATE的产生电路等，有很多种电路结构，且与本发明无关，下文不做说明，图4也并未示出。在本发明中，低压电源VCC是由变换器输入电压VDD降压得到的芯片的工作电压，即是给控制器100内部其它子模块供电的电压源。实施例选取的低压电源VCC＝5V。由于控制器100中所集成的两个开关管Q1、Q2在电气特性和尺寸形状上都是一样的结构，两个相同的短路保护检测电路也同时施加在这两个开关管漏极，因此后续以其中一个开关管Q1为例做详细说明即可。
如图4所示，本发明的具有温度补偿的短路保护检测电路包括电源端VCC、地端GND和开关管Q1的漏极端VD1(以下简称为VD端)三个端口，还包括启动电路11、正温度系数电流生成电路12、短路保护阈值生成电路13、开关管单元14和比较器单元15五个部分。其中开关管Q1为NMOS管，图4中使用NM1的器件编号，如下也可称为开关管NM1。
启动电路11，由三个PMOS管PM1、PM2和PM3，以及电阻R1组成。PMOS管PM1和PM2的源极均接电源端VCC，PM1的栅极电位V1作为正温度系数电流生成电路12的一路控制信号，用于为后续电流镜提供偏置电压；PM2的漏极电位V2作为正温度系数电流生成电路12的另一路控制信号，用于在电源上电期间，控制检测电路不工作；PM1的漏极、PM2的栅极与PMOS管PM3的源极连接，PM3的栅极与其漏极连接后一起接至电阻R1一端，电阻R1的另一端与地端GND连接。
电源电压VCC上电期间，当VCC达到MOS管PM2和PM3导通阈值之和即VTH PM2+VTH PM3后，电阻R1上的电压V R1开始随流过的电流变大而变大，本阶段由于MOS管PM2导通使V2电位接近高电平VCC，使得正温度系数电流生成电路12中的电流镜无法开启，检测电路不工作；直到MOS管PM2的栅极电压即VTH PM3+V R1与VCC的压差不足以让MOS管PM2开启，即在电源电压达到启动点后，才解除后续电流镜的关闭状态并使能启动电路的后续电路，同时导通的MOS管PM1的栅压V1为后续电流镜提供偏置电压；此时启动支路的MOS管PM2关断，启动支路被关断，不影响其余电路正常工作。
正温度系数电流生成电路12，由两个PMOS管PM4和PM5、两个NMOS管NM3 和NM4、两个双极型NPN晶体管B1和B2及一个电阻R2构成，其中，PMOS管PM4与PM5为电流镜结构且尺寸一致。PMOS管PM4和PM5源极均接电源端VCC，PMOS管PM4和PM5的栅极连接在一起，并引出作为第一电流镜的驱动控制端，用于接收启动电路11提供的控制信号V1；第一电流镜的驱动控制端还与PMOS管PM5的漏极和NMOS管NM4的漏极连接在一起，再引出作为正温度系数电流生成电路的输出端，用于向后续电流镜提供由正温度系数电流生成电路生成的正温度系数电流；NMOS管NM4的栅极、NM3的栅极和漏极与PMOS管PM4的漏极连接在一起，并引出作为正温度系数电流生成电路的驱动控制端，用于接收启动电路11提供的控制信号V2，NM3的源极与双极型NPN晶体管B1的集电极和基极相连，NM4的源极与双极型NPN晶体管B2的集电极相连，晶体管B1的基极与晶体管B2的基极连接，晶体管B2的发射极接至电阻R2的一端，电阻R2的另一端和晶体管B1的发射极均接至地端GND。如图5所示正温度系数电流生成电路的细节图，其中，晶体管B2是由两个以上的单位双极型NPN晶体管B21、B22……B2N并联构成。
本电路部分根据工作在不同电流密度下的两个双极性晶体管B1和B2构造出它们的基极-发射极电压的差值，该电压差值与绝对温度成正比，通过电阻R2转化生成用于短路保护阈值补偿的正温度系数电流。结合图5所示的温度补偿电流的电路细节原理图分析该电流生成具体的工作公式推导：
已知对于双极性晶体管，其集电极电流I C满足：
I C＝I Sexp(V BE/V T)
其中，
V T＝kT/q
V T为晶体管的热电压，一般来说300K温度下对应热电压为26mV，Is为晶体管的饱和电流。饱和电流I S正比于
Figure PCTCN2019084988-appb-000001
其中μ为少数载流子的迁移率，n i为硅的本征载流子浓度，这些参数均与温度相关。q为电子电荷，T为温度，k为常数。
因此可以得到：
V BE＝V Tln(I C/I S)
晶体管B1与B2的基极-发射极压差可表示为：
Figure PCTCN2019084988-appb-000002
设计时MOS管PM4与PM5为电流镜结构且尺寸一致，保证流入两个晶体管集电极的电流I C1＝I C2，且晶体管B1与B2为同种类型的NPN晶体管，它们的饱和电流密度比值正比于管子个数之比，因此：
Figure PCTCN2019084988-appb-000003
通过电阻R2转化生成用于短路保护阈值补偿的正温度系数电流为：
I COMP＝V Tln N/R 2
此处N为构成晶体管B2的单位晶体管的个数与晶体管B1的的单位晶体管的个数的比值。R2代表正温度系数电流生成电路12中电阻R2的阻值。N的选值可以综合版图匹配和短路保护阈值设定值等因素来选取，本发明电路中推荐N选值为3，即晶体管B1选取1个单位晶体管，晶体管B2选取3个单位晶体管。
所述短路保护阈值生成电路13，由两个PMOS管PM6和PM7、一个高压NMOS管NM2、两个电容C1、C2和两个电阻R3、R4组成。其中，PMOS管PM6和PM7为第一电流镜的镜像支路，PMOS管PM6和PM7源极均接电源端VCC，PMOS管PM6和PM7的栅极连接在一起作为镜像支路的驱动控制端，用于接收来自启动电路11提供的控制信号V1，镜像支路的驱动控制端还连接正温度系数电流生成电路的输出端，用于接收正温度系数电流生成电路生成的正温度系数电流，以将正温度系数电流生成电路12生成的用于短路保护阈值补偿的正温度系数电流通过PMOS管PM6和PM7分两路镜像为电流I1和I2；PMOS管PM6的漏极与电容C1的一端、电阻R3的一端连接于一个输出节点VA，输出节点VA用于提供带温度补偿的短路保护阈值给比较器的负相输入端；电容C1和电阻R3的另一端均接至地端GND，PMOS 管PM7的漏极与电容C2的一端、电阻R4的一端连接于一个输出节点VB，输出节点VB用于提供开关管NM1漏极的检测电压给比较器的正相输入端；电容C2的另一端接至地端GND，电阻R4的另一端与高压NMOS管NM2的源极相连，NMOS管NM2的栅极接电源端VCC，NMOS管NM2的漏极接开关管NM1的漏极，用于检测开关管NM1的漏极端VD的电压。其中NMOS管NM2管就是在开关管NM1关断期间，用于防止开关管NM1漏极的高压击穿电容C2、低压PMOS管PM7以及后续比较器的输入器件所设计的高压隔离器件，其耐压值根据功率开关管NM1的漏极端VD的最大电位选取，其栅极接VCC电位，在此处起到钳位NM2源极电位不高于电源电压VCC电平值的作用。
将上述正温度系数电流生成电路12生成的电流通过PMOS管PM6和PM7镜像为电流I1和I2，分别降落在电阻R3和R4上，其中电阻R3和R4是由不同个数的单位电阻所构成的电阻。单位电阻即是根电阻，根电阻为宽长比固定的单个电阻，图4电路中的电阻R2、R3和R4都是基于不同个数根电阻串并联所得到的电阻。采用根电阻设计的好处在于，实现电压-电流-电压转换的过程中，电阻的比值可以抵消工艺偏差。设：
I1：I2＝1：X
R3：R4＝1：Y
X和Y均为常数值。通过设计具体的X和Y数值，匹配出与开关管漏源电压温度系数接近的短路保护阈值电压；R3和R4电阻为正温度系数或者负温度系数电阻均可，只需满足该电阻与正温度系数电流配合生成的电压，与开关管在短路保护临界点处的的漏极电压的温度曲线最大限度的吻合即可。.
所述开关管14，是由若干单位NMOS管并联组成的MOS管NM1，具有尺寸大、导通内阻小和可流经大电流的特性。MOS管NM1的栅极接收控制芯片内部生成的控制信号GATE，MOS管NM1的漏极引出即为开关管NM1的漏极端VD，MOS管NM1的源极与地端GND连接。
开关管NM1作为集成在芯片之中用于开关电源开关作用的主功率MOS管，当NM1导通时，其漏极的电压VD包含有流经开关管NM1的电流I SW信息，可用于反映输出是否发生短路。
所述比较器15，是常见的基于CMOS器件设计的比较器模块，其正相输入端 连接短路保护阈值生成电路13的输出节点VB，用于接收开关管NM1漏极的检测电压；比较器的负相输入端连接短路保护阈值生成电路13的输出节点VA，用于接收带温度补偿的短路保护阈值。在本发明中比较器输出节点的信号又称为短路保护判定信号OCP_H。
比较器用于对开关管NM1漏极电压与上述生成的短路保护阈值电压进行比较，并输出比较结果信号OCP_H，该结果信号经过芯片后续逻辑处理，完成短路保护功能。
根据电路可知，短路保护的阈值构成推导如下：
VA＝VB的时刻对应VD达到短路保护阈值发生翻转的临界点，
V B-V A＝(I2*R4+V D)-(I1*R3-V S)
其中V S为开关管NM1的源极电压，因为接地，故V S＝0，
V B-V A＝XI*YR+V D-I*R＝(XY-1)*IR+V D
其中I表示I1或I2电流的单位值，R表示电阻R3或R4的根电阻阻值，
所以令VA＝VB得到短路保护阈值V TH_OCP为：
V TH_OCP＝V D_OCP＝(XY-1)*IR
因此，设计电路时I是正温度系数的电流，调节X和Y数值以及R的大小，去配比出与开关管漏源电压温度系数一致，且数值与短路临界点上开关管漏极电压值大小一致的短路保护阈值电压。
如图6所示的变化曲线，为本发明实施例的与短路保护相关的指标随温度的变化曲线仿真图。如图所示，曲线V TH_OCP为本发明所实施例电路设计得到的短路保护阈值随温度变化的曲线；曲线VD_mos为本发明所提供的具有温度补偿的短路保护检测电路的补偿对象——某一开关管漏端电压在短路保护临界点随温度变化的曲线，本曲线是由对开关管在固定栅压下，漏极给固定大小的短路电流，再做温度扫描得到的；曲线△V为本发明实施例设计得到的短路保护阈值与补偿对象即开关管漏端电压的差值随温度变化的曲线。坐标横轴为温度T，纵轴为电压V。
由该仿真结果可知，本发明设计的具有温度补偿的短路保护检测电路所设计生成的阈值电压V TH_OCP与开关管的漏极电压的温度系数趋势一致，且在整个-40℃-到125℃的温度设计范围中，V TH_OCP在数值上也很接近处于短路保护临界点上的开关管漏极电压VD_mos，由△V曲线可以看出，V TH_OCP与VD_mos在整个温度范围内 的大小偏差不大于30mV，也就是小于了典型温度下292mV阈值中心值的10％，符合设计要求，很好地跟随了开关管漏极电压的温度系数。而现有技术未在这里做温度补偿，则与开关管漏极比较的短路保护阈值为一常量，用变化曲线图表示时，其短路保护阈值随温度变化的曲线在全温度范围内就是一条水平直线。
以上仅是本发明的优选实施例，应当指出的是，上述优选实施例不应视为对本发明的限制。按照本发明的上述内容，利用本领域的普通技术知识和惯用手段，在不脱离本发明上述基本技术思想的前提下，本发明还可以做出其它多种形式的修改、替换或变更，这些均落在本发明权利保护范围之内。

Claims (10)

  1. 一种短路保护的检测电路，包括比较器，所述比较器，对开关管的漏极电压与短路保护阈值电压进行比较，并输出比较结果信号，用以完成短路保护功能，其特征在于：还包括正温度系数电流生成电路和短路保护阈值生成电路，
    所述正温度系数电流生成电路，利用镜像生成的两路相同大小的电流分别降落在电流密度不同的两个双极性晶体管上，产生两个双极性晶体管的基极-发射极电压的差值，该电压的差值与绝对温度成正比，然后通过第一电阻转化成用于短路保护阈值补偿的正温度系数电流，通过正温度系数电流生成电路的输出端提供给短路保护阈值生成电路；
    所述短路保护阈值生成电路，利用镜像将正温度系数电流分两路降落在单位电阻构成数量不同的第二电阻和第三电阻两个电阻上，通过设计两路正温度系数电流的比例与两个电阻的单位电阻构成数量的比例，匹配生成与开关管漏极检测电压温度系数吻合的短路保护阈值电压。
  2. 根据权利要求1所述的短路保护的检测电路，其特征在于：所述正温度系数电流生成电路，由MOS管PM4、MOS管PM5、MOS管NM3、MOS管NM4、晶体管B1、晶体管B2和电阻R2构成，MOS管PM4、PM5为两个PMOS管，MOS管NM3、NM4为两个NMOS管，晶体管B1、B2为两个双极型NPN晶体管，其连接关系是，MOS管PM4和MOS管PM5的源极均接电源端VCC，MOS管PM4的栅极和MOS管PM5的栅极连接在一起，并引出作为第一电流镜的驱动控制端；第一电流镜的驱动控制端还与MOS管PM5的漏极及MOS管NM4的漏极连接在一起，再引出作为正温度系数电流生成电路的输出端；MOS管NM4的栅极、MOS管NM3的栅极和漏极与MOS管PM4的漏极连接在一起，并引出作为正温度系数电流生成电路的驱动控制端；MOS管NM3的源极与晶体管B1的集电极和基极相连，MOS管NM4的源极与晶体管B2的集电极相连，晶体管B1的基极与晶体管B2的基极连接，晶体管B2的发射极接至电阻R2的一端，电阻R2的另一端和晶体管B1的发射极均接至地端GND，其中，晶体管B2是由两个以上的单位双极型NPN晶体管并联构成。
  3. 根据权利要求2所述的短路保护的检测电路，其特征在于：所述正温度系数电流生成电路的晶体管B2，单位双极型NPN晶体管并联的数量为3个。
  4. 根据权利要求1所述的短路保护的检测电路，其特征在于：所述短路保 护阈值生成电路，由MOS管PM6、MOS管PM7、MOS管NM2、电容C1、电容C2、电阻R3和电阻R4组成，MOS管PM6、MOS管PM7为两个PMOS管，MOS管NM2为一个高压NMOS管，其连接关系是，MOS管PM6和MOS管PM7的源极均接电源端VCC，MOS管PM6和MOS管PM7的栅极连接在一起，还连接正温度系数电流生成电路的输出端；MOS管PM6的漏极与电容C1的一端、电阻R3的一端连接于一个第一输出节点，第一输出节点用于提供匹配生成的短路保护阈值给比较器的反相输入端；电容C1与电阻R3的另一端均接至地端GND，MOS管PM7的漏极与电容C2的一端、电阻R4的一端连接于一个第二输出节点，第二输出节点用于提供开关管漏极的检测电压给比较器的正相输入端；电容C2的另一端接至地端GND，电阻R4的另一端与MOS管NM2的源极相连，MOS管NM2的栅极接电源端VCC，MOS管NM2的漏极接开关管的漏极，用于检测开关管的漏极电压。
  5. 根据权利要求1所述的短路保护的检测电路，其特征在于：还包括启动电路，启动电路控制检测电路在电源电压达到启动点后才工作，且检测电路工作后启动电路关断；所述启动电路，由MOS管PM1、MOS管PM2、MOS管PM3及电阻R1组成，MOS管PM1、MOS管PM2、MOS管PM3为三个PMOS管，其连接关系是，MOS管PM1和MOS管PM2的源极均接电源端VCC，MOS管PM1的栅极引出作为偏置电压提供端，MOS管PM2的漏极引出作为上电保护关断控制端；MOS管PM1的漏极、MOS管PM2的栅极与MOS管PM3的源极连接，MOS管PM3的栅极与其漏极连接后一起接至电阻R1的一端，电阻R1的另一端与地端GND连接。
  6. 一种短路保护的检测电路，包括比较器，所述比较器，用于对开关管的漏极电压与短路保护阈值电压进行比较，并输出比较结果信号，用以完成短路保护功能，其特征在于：还包括正温度系数电流生成电路和短路保护阈值生成电路，
    所述正温度系数电流生成电路，由MOS管PM4、MOS管PM5、MOS管NM3、MOS管NM4、晶体管B1、晶体管B2和电阻R2构成，MOS管PM4、PM5为两个PMOS管，MOS管NM3、NM4为两个NMOS管，晶体管B1、B2为两个双极型NPN晶体管，其连接关系是，MOS管PM4和MOS管PM5的源极均接电源端VCC，MOS管PM4的栅极和MOS管PM5的栅极连接在一起，并引出作为第一电流镜的驱动控制端；第一电流镜的驱动控制端还与MOS管PM5的漏极及MOS管NM4的漏极连接在一起，再引出作为正温度系数电流生成电路的输出端；MOS管NM4的栅极、MOS管NM3的 栅极和漏极与MOS管PM4的漏极连接在一起，并引出作为正温度系数电流生成电路的驱动控制端；MOS管NM3的源极与晶体管B1的集电极和基极相连，MOS管NM4的源极与晶体管B2的集电极相连，晶体管B1的基极与晶体管B2的基极连接，晶体管B2的发射极接至电阻R2的一端，电阻R2的另一端和晶体管B1的发射极均接至地端GND，其中，晶体管B2是由两个以上的单位双极型NPN晶体管并联构成；
    所述短路保护阈值生成电路，由MOS管PM6、MOS管PM7、MOS管NM2、电容C1、电容C2、电阻R3和电阻R4组成，MOS管PM6、MOS管PM7为两个PMOS管，MOS管NM2为一个高压NMOS管，其连接关系是，MOS管PM6和MOS管PM7的源极均接电源端VCC，MOS管PM6和MOS管PM7的栅极连接在一起，还连接正温度系数电流生成电路的输出端；MOS管PM6的漏极与电容C1的一端、电阻R3的一端连接于一个第一输出节点，第一输出节点用于提供匹配生成的短路保护阈值给比较器的反相输入端；电容C1与电阻R3的另一端均接至地端GND，MOS管PM7的漏极与电容C2的一端、电阻R4的一端连接于一个第二输出节点，第二输出节点用于提供开关管漏极的检测电压给比较器的正相输入端；电容C2的另一端接至地端GND，电阻R4的另一端与MOS管NM2的源极相连，MOS管NM2的栅极接电源端VCC，MOS管NM2的漏极接开关管的漏极，用于检测开关管的漏极电压。
  7. 根据权利要求6所述的短路保护的检测电路，其特征在于：所述正温度系数电流生成电路的晶体管B2，单位双极型NPN晶体管并联的数量为3个。
  8. 根据权利要求6所述的短路保护的检测电路，其特征在于：还包括启动电路，所述启动电路，由MOS管PM1、MOS管PM2、MOS管PM3及电阻R1组成，MOS管PM1、MOS管PM2、MOS管PM3为三个PMOS管，其连接关系是，MOS管PM1和MOS管PM2的源极均接电源端VCC，MOS管PM1的栅极引出作为偏置电压提供端，MOS管PM2的漏极引出作为上电保护关断控制端；MOS管PM1的漏极、MOS管PM2的栅极与MOS管PM3的源极连接，MOS管PM3的栅极与其漏极连接后一起接至电阻R1的一端，电阻R1的另一端与地端GND连接。
  9. 一种短路保护的检测方法，包括如下步骤：
    温度补偿电流生成步骤，利用镜像生成的两路相同大小的电流分别降落在不同电流密度的两个双极性晶体管上，在两个双极性晶体管的基极-发射极上产生 电压的差值，该电压的差值与绝对温度成正比，然后通过第一电阻转化成用于短路保护阈值补偿的正温度系数电流，输出给短路保护阈值生成电路；
    短路保护阈值生成步骤，利用镜像将正温度系数电流分两路降落在单位电阻构成数量不同的第二电阻和第三电阻两个电阻上，通过设计两路正温度系数电流的比例与两个电阻的单位电阻构成数量的比例，匹配生成与开关管漏极与源极电压温度系数吻合的短路保护阈值电压；
    检测和比较输出步骤，检测开关管导通时的漏极电压，将其漏极电压与所述匹配生成的短路保护阈值电压进行比较，并输出比较结果信号。
  10. 根据权利要求9所述的短路保护的检测方法，其特征在于：所述温度补偿电流生成步骤中，短路保护阈值补偿的正温度系数电流遵循如下关系式：
    I COMP＝V Tln N/R 2，
    其中，V T为晶体管的热电压，N为两个晶体管的单位晶体管构成数量的比值，R 2为温度补偿电流生成步骤中第一电阻的阻值。
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