WO2019230818A1

WO2019230818A1 - Motor control device, motor control method, and motor system

Info

Publication number: WO2019230818A1
Application number: PCT/JP2019/021336
Authority: WO
Inventors: 晋衣山田; 正倫綿引
Original assignee: 日本電産株式会社
Priority date: 2018-06-01
Filing date: 2019-05-29
Publication date: 2019-12-05
Also published as: CN112204874A

Abstract

In an embodiment of the present invention, a motor control device controls a permanent magnet synchronous motor (100) which has a stator (100S) and a rotor (100R), and said motor control device comprises a processor (90) and memory (95), which stores a program (95) controlling an operation of the processor. The processor, on the basis of a speed command or a torque command, determines a q axis current in a dq axis coordinate system, which rotates in sync with the rotation of the rotor, and determines a value, which is obtained from an Nth degree harmonic current having been superimposed on a q axis current, as a q axis current command value, said Nth degree harmonic current having a phase which is inverse to a phase of an Nth degree harmonic component of cogging torque in the permanent magnet synchronous motor.

Description

Motor control device, motor control method, and motor system

The present disclosure relates to a motor control device and a motor driving method for a permanent magnet synchronous motor, and a motor system including the motor control device.

Originally, it is desirable that the torque of the permanent magnet synchronous motor has a predetermined magnitude regardless of the rotational position of the rotor. However, pulsation (ripple) can occur in torque due to various causes. Since such torque ripple causes vibration and noise, it is required to reduce the torque ripple. *

Japanese Laid-Open Patent Publication No. 2008-54386 discloses not only the torque ripple component of the sixth harmonic generated due to the structure of the motor but also the second harmonic generated due to dimensional variations in manufacturing. A technique for reducing torque ripple components is disclosed. This technique can reduce torque ripple at the frequency by adjusting the current control gain (proportional gain and integral gain) in accordance with the frequency of the harmonic to be suppressed. *

Japanese Patent Laid-Open Publication No. 2013-085439 discloses a harmonic current (6th electrical angle) in at least one of a normal d-axis current and a q-axis current (zero-order current) in order to reduce torque ripple caused by magnet torque. Hereinafter, it may be simply referred to as “sixth harmonic current”).

Japanese publication: JP 2008-54386 A Japanese publication: JP2013-85439A

In the prior art in which various gains of the current controller are set according to the torque ripple frequency, the response to the current command is delayed. In addition, superposition of the sixth-order electrical angle harmonic current has an effect on reducing the sixth-order torque ripple, but has no effect on reducing lower-order torque ripple that may be caused by dimensional variations in manufacturing. *

Embodiments of the present disclosure provide a new motor control device and a motor control method that suppress N-order harmonic components generated due to manufacturing dimensional variations. In addition, an embodiment of the present disclosure provides a motor system including the motor control device.

In an exemplary embodiment, a motor control device of the present disclosure is a motor control device that controls a permanent magnet synchronous motor having a stator and a rotor, and includes a processor and a memory that stores a program for controlling the operation of the processor. And the processor determines a q-axis current in a dq-axis coordinate system that rotates in synchronization with the rotation of the rotor based on a speed command or a torque command in accordance with a command of the program, and the permanent magnet synchronous motor A value obtained by superimposing the N-order harmonic current having a phase opposite to the phase of the N-order harmonic component (N is an integer of 2 or more) of the cogging torque on the q-axis current is determined as the q-axis current command value. Execute. *

In an exemplary embodiment, the motor control method of the present disclosure is a motor control method for controlling a permanent magnet synchronous motor having a stator and a rotor, and is synchronized with the rotation of the rotor based on a speed command or a torque command. Determining the q-axis current in the rotating dq-axis coordinate system, and determining the amplitude and phase of the N-order harmonic current having a phase opposite to the phase of the N-order harmonic component of the cogging torque in the permanent magnet synchronous motor And determining a value obtained by superimposing the Nth harmonic current on the q-axis current as a q-axis current command value. *

In an exemplary embodiment, a motor system of the present disclosure includes a surface magnet type permanent magnet synchronous motor having a split stator and a rotor, a motor driving device connected to the permanent magnet synchronous motor, and a connection to the motor driving device. A motor control device. The motor control device includes a processor and a memory that stores a program for controlling the operation of the processor. The memory further stores the phase and amplitude of the Nth harmonic component of cogging torque in the permanent magnet synchronous motor. The processor determines a q-axis current in a dq-axis coordinate system that rotates in synchronization with the rotation of the rotor based on a speed command or a torque command in accordance with a command of the program, and the memory stored in the memory Determining a current value obtained by superimposing a second harmonic current having a phase and amplitude on the q-axis current as a q-axis current command value, and an amplitude of an Nth-order harmonic component of a torque ripple in the permanent magnet synchronous motor The increase is smaller than the amplitude of the Nth harmonic component of the cogging torque.

According to the embodiment of the present disclosure, it is possible to effectively reduce the Nth harmonic component of torque ripple generated due to manufacturing variation.

FIG. 1 is a diagram schematically illustrating a configuration of a non-limiting exemplary embodiment of a motor system according to the present disclosure. FIG. 2 is a diagram illustrating a configuration example of the motor 100. FIG. 3 is a diagram illustrating an example of a hardware configuration of a motor control device in the motor system according to the present disclosure. FIG. 4 is a diagram illustrating a configuration example of the stator 100S in the present embodiment. FIG. 5 is a diagram illustrating a configuration example of the rotor 100R in the present embodiment. FIG. 6 is a diagram schematically showing deformation that occurs on the inner peripheral surface of the split stator due to manufacturing variations. FIG. 7 is a diagram schematically illustrating a configuration of another non-limiting exemplary embodiment of the motor system according to the present disclosure. FIG. 8 is a diagram schematically illustrating a configuration example of an integrated circuit device according to the present disclosure. FIG. 9 is a flowchart illustrating a procedure in the embodiment of the motor control method according to the present disclosure. FIG. 10 shows the torque amplitude and electrical angle order obtained for a motor (normal model) in which the minimum or maximum value of the stator inner diameter has a difference smaller than 0.09% from the average inner diameter. It is a graph which shows a relationship. FIG. 11 shows the torque amplitude and electrical angle order obtained for a motor (variation model) in which the minimum or maximum value of the inner diameter of the stator shows a difference of 0.09% or more from the average value of the inner diameter due to manufacturing variations. It is a graph which shows the relationship. FIG. 12 shows the torque amplitude and electrical angle order obtained when the second harmonic current is superimposed on the q-axis current so as to cancel the secondary torque ripple when the motor of FIG. 11 is operated (superposition model). It is a graph which shows the relationship. FIG. 13 is a graph showing a change in torque over time when the second harmonic current is not superimposed on the “variation model” and the “normal model”. FIG. 14 shows the time variation of the torque when the second harmonic current is not superimposed on the “variation model” and the torque when the second harmonic current is superimposed on the “variation model” (superimposition model). It is a graph which shows the time change of. FIG. 15 is a graph showing a secondary torque ripple (dotted line) when the second harmonic current is not superimposed on the q-axis current, and a secondary torque ripple (solid line) when the second harmonic current is superimposed on the q-axis current. It is.

In a permanent magnet synchronous motor including a rotor and a stator, the outer peripheral surface of the rotor faces the inner peripheral surface of the stator defined by the tip surface of the stator teeth via an air gap. The magnetic flux passing through the air gap mainly flows in the radial direction and the circumferential direction. As a result, a circumferential force (torque) and a radial force (radial force) are generated between the stator and the rotor. These forces are called “electromagnetic forces”. The interlinkage magnetic flux that generates the electromagnetic force includes the interlinkage magnetic flux generated by the permanent magnets in the rotor and the interlinkage magnetic flux formed by energizing the stator windings. Since the magnitude of the magnetic flux component (interlinkage magnetic flux) penetrating each tooth changes spatially and temporally, the electromagnetic force also changes spatially and temporally. This causes electromagnetic vibration and torque ripple. *

It is known that the vibration and noise caused by the radial force when driving the motor can be reduced by superimposing a harmonic current having an amplitude and phase satisfying a specific condition on the current flowing through the stator winding. . *

In the motor control device, the motor control method, and the motor system of the present disclosure, harmonic currents of the same order are superimposed in opposite phases in order to reduce low-order torque ripple that is generated due to manufacturing variation, not radial force. *

Before describing the embodiment of the present disclosure, main causes of torque ripple will be described below. *

Generally, the cause of torque ripple is roughly divided into the following two types. *

(1) Spatial harmonics

The number of interlinkage magnetic fluxes formed in the stator winding by the magnetic poles of the rotor changes periodically according to the rotational position of the rotor. Hereinafter, the rotational position of the rotor is referred to as “angular position” or “position” and is represented by an angle θ from the reference direction in the fixed coordinate system. The number of interlinkage magnetic fluxes formed in each stator winding when a current is passed through the stator winding also depends on an inductance that can be periodically changed according to the position θ of the rotor. Each of the magnetic pole and the inductance of the rotor may include a harmonic component (spatial harmonic component) that cannot be ignored other than the fundamental wave component having the rotor position θ as a variable. Such a spatial harmonic component causes a ripple in the torque generated when a current is passed through the stator winding.

(2) Cogging torque

Even in a state where no current flows through the stator winding (no load state), a magnetic flux due to the magnetic poles of the rotor exists in the air gap between the rotor and the stator, and magnetic energy is accumulated. Since this air gap periodically changes according to the rotor position θ due to the presence of a plurality of slots in the stator, the magnetic energy is a function of the rotor position θ. When the magnetic energy stored in the air gap in a state where no current flows through the stator winding is W (θ), a torque having a magnitude represented by δW (θ) / δθ is generated. This is referred to as “cogging torque”. The cogging torque does not depend on the presence or absence of the stator winding current (drive current). For this reason, when an electric current is passed through the stator winding and the motor is operated, a torque ripple is generated in which a component caused by the spatial harmonics and a component caused by the cogging phenomenon overlap.

Conventionally, various techniques for reducing torque ripple have been proposed. For example, the spatial harmonic component has been reduced by devising the structure of the rotor and stator or the distribution of windings. *

According to the experiments by the present inventors, even when the torque ripple can be sufficiently reduced by optimizing the structure of the rotor and the stator, the torque ripple may increase due to the manufacturing variation of the motor. In the embodiment of the present disclosure, the torque ripple caused by such manufacturing variation is reduced by superimposing the harmonic current on the winding current. *

First, the influence of the harmonic component of the winding current on the torque will be described. *

In each tooth of the stator, assuming that the magnetic flux is φ, the area of the tip of the tooth is S, the interlinkage magnetic flux is Ψ, and the number of turns of the winding is N, the magnetic flux density B in each tooth is expressed by the following Equation 1. .

The interlinkage magnetic flux ψ is expressed by a linear combination of a permanent magnet component ψ _{m produced by} the permanent magnet of the rotor and a current component ψ _{i produced by the} current flowing through the stator winding, and therefore the following formula 2 is established.

In a permanent magnet synchronous motor driven by three phases of the UVW phase, the torque T is expressed by the following Equation 3.

Here, Ψ _mU , Ψ _mV , and Ψ _mW are U, V, and W-phase linkage magnetic fluxes generated by the permanent magnets of the rotor, respectively. i _U , i _V , and i _W are winding currents of the U, V, and W phases, respectively. L _U , L _V , and L _W are self-inductances of the U, V, and W phases, respectively, and M _UV , M _VW , and M _WU are mutual inductances between the U, V, and W phases, respectively. P _n is the number of pole pairs.

Ψ _mU , Ψ _mV , and Ψ _mW are expressed by the following Equation 4 in consideration of the spatial harmonic component.

L _U , L _V , L _W and M _UV , M _VW , M _WU are expressed by the following

formulas

5 and 6, respectively, considering the spatial harmonic components.

By performing dq / UVW conversion with respect to Equation 3 described above, torque can be expressed in a dq axis coordinate system that rotates in synchronization with the rotation of the rotor. In the dq-axis coordinate system, “d” on the d-axis is an acronym for “direct”, and the d-axis faces the N pole direction of the permanent magnet of the rotor. “q” on the q-axis is an acronym for “quadrature”, and the q-axis faces a direction orthogonal to the d-axis at an electrical angle of 90 °. *

Here, for simplicity, it is assumed that the current vector in the dq axis coordinate system is a current on which no harmonic current is superimposed. That is, the current vector in the dq axis coordinate system is represented by the d axis zero order current i _d0 and the q axis zero order current i _q0 . When dq / UVW conversion is performed on the equation (3), the following equation (7) is obtained.

Here, Ψ _md0 and Ψ ′ _md6 are the amplitudes of the zero-order component and the sixth-order component of the d-axis flux linkage generated by the permanent magnet, respectively. Ψ ′ _mq6 is the amplitude of the sixth-order component of the q-axis flux linkage generated by the permanent magnet. L _d0 and L _q0 are a zero-order component of the d-axis self-inductance and a zero-order component of the q-axis self-inductance, respectively. i _d0 and i _q0 are a d-axis current and a q-axis current (zero-order current) in a state where the harmonic current is not superimposed, respectively.

[C] in Equation 7 is a matrix that defines UVW / dq conversion, and is expressed by Equation 8 below. [C] ^T, which is a transposed matrix of [C], is a matrix that defines the dq / UVW conversion, and is expressed by the following Equation 9.

In these conversions, the following

formulas

10 and 11 are established.

From Equation 7, the torque in the dq axis coordinate system is approximately represented by the sum of the zeroth order component (stationary component) and the sixth harmonic component, as shown in Equation 12 below.

Here, the zero-order component of the torque is represented by the following formula 13, and the sixth harmonic component is represented by the following formula 14.

When the harmonic current is not superimposed on the winding current, a linkage flux of a sixth harmonic component that changes in accordance with the rotor position θ is generated in the dq axis coordinate system. The sixth-order harmonic components are the first-order and fifth-order components in which the interlinkage magnetic fluxes Ψ _mU , Ψ _mV , and Ψ _mW of the U, V, and W phases generated by the permanent magnets of the rotor change according to the rotor position θ. This occurs because it is approximately expanded as the sum of the seventh-order harmonic components.

In order to reduce the sixth-order harmonic component of radial force and torque ripple, it has been proposed to superimpose the sixth-order harmonic on the d-axis current and the q-axis current as shown in Equation 15 below.

Here, i _d6 and i _q6 are the amplitudes of the superimposed sixth-order harmonic current, and θ _d6 and θ _q6 are the _phases of the sixth-order harmonic current.

The above description holds true for an ideal motor with no manufacturing variations. In an actual motor, a low-order cogging torque of secondary or tertiary can be generated due to manufacturing variations. As will be described later, particularly in the case of a motor having a divided stator, a cogging torque having a second harmonic component is likely to be generated. Further, cogging torque having a third harmonic component may be generated due to the thickness variation of the permanent magnet in the rotor. *

In the embodiment of the present disclosure, for example, a torque having a phase and an amplitude that cancels the second-order or third-order cogging torque (harmonic component) with an electrical angle generated due to manufacturing variations of such individual motors ( In order to intentionally generate an anti-phase harmonic component), a secondary or tertiary harmonic current is superimposed on the winding current. More specifically, a second harmonic component is generated in the torque T of Equation 7 by superimposing a second harmonic current, for example, on the q-axis zero-order current i _q0 shown in Equation 7. By adjusting the phase and amplitude of the superimposed second harmonic current, the harmonic component of the cogging torque can be canceled by the second harmonic component of the generated torque T.

Note that the q-axis current (q-axis zero-order current i _q0 ) and the d-axis current (d-axis zero-order current i _d0 ) are determined by a normal vector control algorithm based on the current command or the torque command.

Hereinafter, embodiments of the present disclosure will be described. *

<Example of motor system configuration>

Hereinafter, non-limiting exemplary embodiments of a motor system according to the present disclosure will be described with reference to the accompanying drawings. A more detailed description than necessary may be omitted. For example, detailed descriptions of already well-known matters and repeated descriptions for substantially the same configuration may be omitted. This is to avoid the following description from becoming unnecessarily redundant and to facilitate understanding by those skilled in the art. The inventors provide the accompanying drawings and the following description to enable those skilled in the art to fully understand the present disclosure. They are not intended to limit the claimed subject matter.

First, a basic configuration example of the motor system in the present embodiment will be described with reference to FIGS. 1 and 2. *

A motor system 1000 shown in FIG. 1 includes a permanent magnet synchronous motor 100 having a stator 100S and a rotor 100R, a motor driving device 200 connected to the permanent magnet synchronous motor 100, and a motor connected to the motor driving device 200. And a control device 300. *

Hereinafter, the permanent magnet synchronous motor 100 is simply referred to as a motor 100. As shown in FIG. 2, the stator 100S of the motor 100 includes a plurality of stator teeth 100T and windings 100W for exciting the stator teeth 100T. In FIG. 2, for the sake of simplicity, a winding 100 </ b> W provided around one stator tooth 100 </ b> T is illustrated. Actually, a winding 100W is provided around each stator tooth 100T. The rotor 100R includes a rotor core 100C and a plurality of permanent magnets 100M arranged in the circumferential direction on the outer peripheral surface of the rotor core 100C. A voltage is applied by the motor driving device 200 to the winding 100W of the stator 100S. The motor 100 in the example of FIG. 2 has an 8-pole 12-slot configuration. The configuration of the permanent magnet synchronous motor that can be used in the embodiments of the present disclosure is not limited to 8 poles and 12 slots. *

Refer to FIG. 1 again. The motor drive device 200 in FIG. 1 is a power converter having an inverter as a main circuit. The main circuit includes a plurality of power semiconductor elements (not shown in FIG. 1) as components. The motor control device 300 generates and outputs a control signal (gate signal) for switching individual power semiconductor elements in the motor driving device 200. *

The motor system 1000 includes a current sensor (not shown) that measures a current flowing through the winding 100W. The current sensor may be, for example, a current transformer (CT), but is not limited thereto. When the motor driving device 200 has one or a plurality of shunt resistors, the current flowing through the winding can be measured by measuring the voltage drop of each shunt resistor. *

In the embodiment of the present disclosure, the motor control device 300 includes a processor 90 and a memory 95 that stores a program for controlling the operation of the processor 90. The processor 90 executes the following processing in accordance with a program instruction.

(1) determining a q-axis current in a dq-axis coordinate system that rotates in synchronization with the rotation of the rotor 100R based on a speed command or a torque command;

(2) The q-axis current command is a value obtained by superimposing an N-order harmonic current having a phase opposite to the phase of the N-order harmonic component (N is an integer of 2 or more) of the cogging torque in the motor 100 on the q-axis current. To be determined as a value.

FIG. 3 shows an example of the hardware configuration of the motor control device 300. The motor control device 300 in this example includes a CPU (central processing unit) 320, a PWM circuit 330, a ROM (read only memory) 340, a RAM (random access memory) 350, and an I / F (input / output interface) connected to each other via a bus. ) 360. Other circuits or devices not shown in the figure (such as AD converters) may be additionally connected to the bus. The PWM circuit 330 gives a drive signal to the motor drive device 200. This drive signal is input to the gate terminal of the switching element in the motor driving apparatus 200, and on / off of each switching element is controlled. Programs and data that define the operation of the CPU 320 are stored in at least one of the ROM 340 and the RAM 350. Such a motor control device 300 can be realized by, for example, a 32-bit general-purpose microcontroller. Such a microcontroller can be comprised of, for example, one or more integrated circuit chips. *

Various operations performed by the motor control device 300 are defined by a program. It is also possible to change part or all of the operation of the motor control device 300 by updating part or all of the contents of the program. Such a program update may be performed using a recording medium storing the program, or may be performed by wired or wireless communication. Communication can be performed using the I / F 360 of FIG. The configuration of the motor control device 300 is not limited to the example shown in FIG. *

According to the embodiment of the present disclosure, the amplitude of the Nth harmonic component of the torque ripple in the motor 100 is smaller than the amplitude of the Nth harmonic component of the cogging torque. For example, when the amplitude of the Nth harmonic component of the cogging torque is 0.7% or more of the average torque, the amplitude of the Nth harmonic component of the torque ripple can be reduced to 0.03% or less of the average torque. . This is because the Nth order harmonic current superimposed on the q-axis current has a phase opposite to the phase of the Nth order harmonic component of the cogging torque. This is because the amplitude becomes small. In other words, an N-order harmonic current that generates a torque that cancels the N-order harmonic component of the cogging torque is superimposed on the q-axis current. The degree of this “cancellation” does not need to be complete, and it is sufficient if the amplitude of the cogging torque generated due to manufacturing variations can be reduced as compared with the value before superposition of the harmonic current. *

The memory 95 shown in FIG. 1 stores a numerical value (a set of numerical values) defined by the phase and amplitude of the Nth harmonic component of the cogging torque in the motor 100. This numerical value may be the “phase and amplitude” itself of the N-order harmonic component of the cogging torque, or may be the “antiphase and amplitude” thereof. For example, in the case of the second harmonic component, “reverse phase” corresponds to shifting the phase by 180 ° with respect to the second harmonic component of the cogging torque. When two waves having an antiphase relationship are overlapped, the amplitude of the entire wave is reduced by overlapping the concave and convex portions of the two waves. If the second harmonic component of the torque having the same or approximately the same amplitude as the second harmonic component of the cogging torque is generated in the opposite phase, the second harmonic component of the cogging torque can be greatly reduced. In order to generate such a second harmonic component of torque with an opposite phase, in the embodiment of the present disclosure, a second harmonic current having a phase shifted by 180 ° with respect to the second harmonic component of the cogging torque. Is superimposed on the q-axis current. For example, when the amplitude of the second harmonic component of the cogging torque is 0.7% or more of the average torque, if such a second harmonic current is superimposed, the amplitude of the Nth harmonic component of the torque ripple is averaged. It can be reduced to 0.03% or less of the torque. *

Note that, depending on the motor, the second harmonic component or the third harmonic component of the cogging torque resulting from manufacturing variations is sufficiently small, for example, less than 0.7% of the average torque. In such a case, the motor control according to the present disclosure may not be applied. In the case of a motor system to which the motor control according to the present disclosure is not applied, the memory 95 does not store a numerical value defined by the phase and amplitude of the Nth-order harmonic component of the cogging torque, or stores zero or other specific value. It may be memorized. *

When controlling the motor 100, the processor 90 acquires the numerical value stored in the memory 95 from the memory 95. Based on the acquired numerical value, the processor 90 determines the phase and amplitude of the Nth harmonic current superimposed on the q-axis current. *

The configuration of the motor 100 is not limited to the configuration described above. A surface magnet type motor (SPM) in which the permanent magnets are arranged on the surface of the rotor 100R may be used, or an embedded magnet type motor (IPM) in which the permanent magnets are incorporated in the rotor 100R. An example of the motor 100 to which the motor control of the present disclosure is preferably applied is a surface magnet type motor. Generally, a surface magnet type motor has a smaller torque ripple than an embedded magnet type motor, and is used for applications that strongly require a small torque ripple. For this reason, by applying the motor control technology of the present disclosure, it is possible to meet the demand. *

When the stator 100S is a split stator having a plurality of cores arranged along the circumferential direction of the rotor 100R, a second harmonic component of cogging torque is likely to occur due to manufacturing variations. In addition, the third harmonic component of cogging torque may occur due to variations in the thickness of the permanent magnet. For this reason, a typical example of the Nth harmonic component is a second or third harmonic component. The motor control in the embodiment of the present disclosure can exert the highest effect when applied to a surface magnet type motor having a split stator. *

FIG. 4 shows a configuration example of the stator 100S in the present embodiment. The stator 100S is a split stator having a plurality of cores 100Sp arranged around the central axis. In FIG. 4, one core 100Sp is described on the left side of the stator 100S for reference. A split stator is realized by connecting the cores 100Sp each having the structure shown in the figure. In the stator 100S of FIG. 4, the description of the winding 100W is omitted for simplicity. In an actual stator 100S, a winding 100W is provided around each stator tooth 100T. The inner diameter of the stator 100S is ideally a perfect circle diameter Rs. However, as will be described later, when the stator 100S is manufactured by combining a plurality of cores 100Sp, the inner peripheral surface of the stator 100S may be deformed from a perfect circle. *

FIG. 5 shows a configuration example of the rotor 100R in the present embodiment. The rotor 100R has a plurality of permanent magnets 100M arranged in the circumferential direction on the outer peripheral surface of the rotor core 100C. In this example, the thickness of the permanent magnet 100M is t, and the diameter of the rotor 100R is Rf. *

FIG. 6 is a diagram schematically showing deformation that occurs on the inner peripheral surface of the split stator due to manufacturing variations. In the illustrated example, the intervals between the six sets of stator teeth arranged symmetrically with respect to the central axis are R1, R2, R3, R4, R5, and R6, respectively. In an ideal case with no manufacturing variation, R1 = R2 = R3 = R4 = R5 = R6 holds. However, in reality, R1 to R6 can take different values due to manufacturing variations, so that a secondary cogging torque is generated, and finally the second harmonic component (secondary torque ripple) of the torque ripple is large. Become. *

According to the study by the present inventor, as a result of manufacturing variation, when the minimum value or maximum value of the inner diameter of the stator 100S has a difference of 0.09% or more from the average value of the inner diameter, a secondary torque ripple appears. For this reason, the motor control of the present embodiment can be applied to a motor in which the minimum value or the maximum value of the inner diameter of the stator 100S has a difference of 0.09% or more from the average value of the inner diameter. *

Alternatively, the secondary harmonic current may be superimposed on a motor whose secondary torque ripple amplitude is 0.7% or more of the average torque. *

<Other configuration examples of the motor system>

With reference to FIG. 7, another non-limiting exemplary embodiment of a motor system according to the present disclosure will be described. In the illustrated example, the motor system 1000 of this embodiment includes a permanent magnet synchronous motor 100 including a rotor 100R and a stator 100S, a position sensor 120 for measuring or estimating the position of the rotor 100R, and a permanent magnet synchronous motor. And a motor control device 300 for controlling 100. A typical example of the position sensor 120 is a magnetic sensor such as a Hall element or Hall IC, a rotary encoder, or a resolver. The position sensor 120 is not indispensable, and a configuration that estimates the position of the rotor 100R without a sensor may be employed.

The motor control device 300 includes a first circuit 10 that determines a zero-order current (d-axis zero-order current i _d0 and q-axis zero-order current i _q0 ) by known vector control, and a harmonic current according to the position of the rotor 100R. A second circuit 20 for determining (d-axis N-order harmonic current i _dh and q-axis N-order harmonic current i _qh ) and a third circuit 30 for determining a current command value are provided. When the motor 100 is a surface magnet type, the d-axis zero-order current i _d0 is, for example, zero. In this case, the d-axis Nth harmonic current i _dh can also be set to zero.

The third circuit 30 is obtained by superimposing a q-axis N-th harmonic current _{i qh} the d-axis 0-order current _{i d0} values obtained by superimposing the d-axis N-th harmonic current _{i dh,} and q-axis 0-order current _{i q0} value Are determined as a d-axis current command value i _d and a q-axis current command value i _q , respectively. In the illustrated example, the third circuit 30 is included in the vector control circuit 40 included in the motor control device 300. The vector control circuit 40 determines the d-axis voltage command value v _d and the q-axis voltage command value v _q based on the d-axis current command value i _d and the q-axis current command value i _q .

In addition, the motor control device 300 includes a first conversion circuit 50 that performs UVW / dq conversion, a second conversion circuit 60 that performs dq / UVW conversion, an inverter 70, and a mechanical angle (θm) indicated by the output of the position sensor 120. Is converted into an electrical angle (θe). First conversion circuit 50, d-axis voltage command value _{v d} and q-axis voltage instruction value _{v q} from the U-phase voltage command values vu, V-phase voltage command value vv, and generates a W-phase voltage command value vw to the inverter 70 Output. The inverter 70 applies a U-phase voltage u, a V-phase voltage v, and a W-phase voltage w to the U-phase winding, V-phase winding, and W-phase winding of the permanent magnet synchronous motor 100, respectively, to obtain a desired current. To the windings of each phase. A circuit that generates a PWM signal based on the voltage command values vu, vv, and vw, and a gate driver that generates a gate drive signal that switches a transistor in the inverter 70 based on the PWM signal may be provided in the previous stage of the inverter 70. . These elements are known and are not shown for simplicity.

Some or all of the components such as the first circuit 10, the second circuit 20, the third circuit 30, the vector control circuit 40, the first conversion circuit 50, and the second conversion circuit 60 may be realized by an integrated circuit device. . Such an integrated circuit device can typically be formed by one or more semiconductor components. *

An example of the integrated circuit device described above will be described with reference to FIG. The integrated circuit device 500 illustrated in FIG. 8 includes a signal processor 520 and a memory 540. The memory 540 stores a program that causes the signal processor 520 to execute the following processing.

・ Process to determine d-axis zero-order current and q-axis zero-order current

A process for determining the d-axis Nth harmonic current and the q-axis Nth harmonic current according to the position of the rotor

A value obtained by superimposing the d-axis Nth order harmonic current on the d-axis 0th order current and a value obtained by superposing the q-axis Nth order harmonic current on the q-axis 0th order current, respectively, as the d-axis current command value and the q-axis current. Processing to determine as a command value

amplitude i _dN and the phase theta _dN of the d-axis N-order harmonic current _and the amplitude i _qN and phase theta _qN q-axis N-th harmonic currents, as compared with the case without high frequency superposition, N th harmonic torque ripple It has a value that lowers the component.

The integrated circuit device 500 includes an A / D converter 560 that converts an analog signal from the position sensor 120 into a digital signal and an analog signal from a sensor (not shown) that detects a current flowing through the winding of the motor 100 as a digital signal. And an A / D converter 580 for conversion into *

The integrated circuit device 500 in this example outputs a PWM signal applied to the inverter 70. At least a part of the inverter 70 may be included in the integrated circuit device 500. Such an integrated circuit device 500 is typically realized by connecting one or more semiconductor chips to each other in one package. A part or all of the integrated circuit device 500 may be realized by writing a program specific to the present disclosure in, for example, a general-purpose microcontroller unit (MCU). *

<Example of motor control method>

With reference to FIG. 9, an embodiment of a motor control method according to the present disclosure will be described.

First, in step S1, a command value such as a current command value or a torque command value and a sensor detection value are received, and a d-axis zero-order current and a q-axis zero-order current are determined. In the determination, a table that associates various command values and sensor detection values with the command values of the d-axis zero-order current and the q-axis zero-order current may be used. Such a table can be recorded in a memory 95 or the like built in the motor control device 300. The magnitudes of the d-axis zero-order current and the q-axis zero-order current can be determined based on, for example, a current command value, a torque command value, a motor rotation speed, a motor applied voltage, and the like. The magnitudes of the d-axis zero-order current and the q-axis zero-order current may be determined based on a speed command and an actual speed output from a speed controller (not shown). *

In step S2, the d-axis Nth harmonic current and the q-axis Nth harmonic current are determined according to the position of the rotor. In this step S2, the amplitude i _dN and the phase θ _dN of the d-axis N-order harmonic current and the amplitude of the q-axis N-order harmonic current are set so that the N-th order torque ripple is smaller than when no harmonic current is superimposed. i _qN and phase θ _qN are determined. In the determination, a table that relates the position of the rotor and the amplitude and phase of the harmonic current may be used. Such a table can be recorded in a memory built in the motor control device.

The amplitude of the Nth-order harmonic current can be calculated off-line by measuring the amplitude of the cogging torque offline. Alternatively, the amplitude of the Nth harmonic current may be swept so that the current is actually passed through the motor, and the amplitude at which the torque ripple of the problem order is reduced to a predetermined level or less may be determined for each motor. Thus, the phase and amplitude of the Nth harmonic current to be superimposed are both determined before shipment of the motor system, and data defining these phase and amplitude is recorded in the memory. In step S2, read from the memory such data in motor operation, the amplitude i _dN and the phase theta _dN of the d-axis N-order harmonic _current, and determines the amplitude i _qN and phase theta _qN q-axis N-th harmonic current To do. In the case of a surface magnet type motor, the determination of the amplitude i _dN and the phase θ _dN of the d-axis N-order harmonic current may be omitted.

In step S3, a value obtained by superimposing the d-axis Nth order harmonic current on the d-axis 0th order current and a value obtained by superposing the q-axis Nth order harmonic current on the q-axis 0th order current are respectively set to the d-axis current command value and Determined as q-axis current command value. *

In step S4, the d-axis voltage command value and the q-axis voltage command value are determined based on the d-axis current command value and the q-axis current command value. *

In step S5, each UVW phase voltage command value is determined based on the d-axis voltage command value and the q-axis voltage command value. *

As a result of such motor control, the amplitude of the Nth harmonic component of the torque ripple in the motor is smaller than the amplitude of the Nth harmonic component of the cogging torque. *

In the embodiment of the present disclosure, for example, a sixth harmonic current may be additionally superimposed in order to reduce the torque ripple caused by manufacturing variation, for example, the sixth radial force. . *

<Example>

The motor control method according to the embodiment of the present disclosure was performed on the surface magnet type motor 100 having an 8-pole 12-slot configuration including the stator 100S and the rotor 100R illustrated in FIGS. 4 and 5.

FIG. 10 shows the torque amplitude and electrical angle order obtained for a motor (normal model) in which the minimum or maximum value of the stator inner diameter has a difference smaller than 0.09% from the average inner diameter. It is a graph which shows a relationship. The rotational speed was 1000 revolutions per minute. Although the sixth torque ripple is generated, the second torque ripple is negligibly small. *

FIG. 11 shows the torque amplitude and electrical angle order obtained for a motor (variation model) in which the minimum or maximum value of the inner diameter of the stator shows a difference of 0.09% or more from the average value of the inner diameter due to manufacturing variations. It is a graph which shows the relationship. The average torque and the rotation speed are the same as in the example of FIG. In this example, a secondary torque ripple larger than the sixth torque ripple was generated. *

FIG. 12 shows the relationship between the torque amplitude and the electrical angle order obtained when the secondary harmonic current is superimposed on the q-axis current so as to cancel the secondary torque ripple when the motor of FIG. 11 is operated. It is a graph to show. The amplitude of the q-axis second-order harmonic current necessary for “cancellation” was about 1/100 to 1/50 of the amplitude of the q-axis zero-order current. As a result, the amplitude of the secondary torque ripple is sufficiently smaller than the amplitude of the sixth torque ripple. Specifically, the amplitude of the secondary torque ripple was smaller than 0.6% of the average torque. *

FIG. 13 is a graph showing a change in torque over time when the second harmonic current is not superimposed on the “variation model” and the “normal model”. FIG. 14 shows the time variation of the torque when the second harmonic current is not superimposed on the “variation model” and the torque when the second harmonic current is superimposed on the “variation model” (superimposition model). It is a graph which shows the time change of. *

As can be seen from FIGS. 13 and 14, by superimposing the second harmonic current on the q-axis current, a small secondary torque ripple is realized in the “variation model” as in the “normal model”. *

FIG. 15 is a graph showing a secondary torque ripple (dotted line) when the second harmonic current is not superimposed on the q-axis current, and a secondary torque ripple (solid line) when the second harmonic current is superimposed on the q-axis current. It is. The effect of high frequency current superposition is obvious. *

In this embodiment, the second harmonic current is superimposed on the q-axis current to cancel the cogging torque generated due to the deviation of the stator inner diameter from the perfect circle. In order to cancel the cogging torque generated due to the variation in the thickness of the permanent magnet, the third harmonic current may be superimposed on the q-axis current. The phase and amplitude of the superimposed harmonic current can be determined off-line so as to sufficiently reduce the torque ripple actually generated in each motor. *

In one exemplary embodiment, a method of manufacturing a motor system according to the present disclosure includes providing a plurality of permanent magnet synchronous motors (especially a surface magnet motor having a split stator) and cogging each of the plurality of motors. Measuring the amplitude of torque and / or torque ripple, and determining the phase and amplitude of a harmonic current that reduces the amplitude of torque ripple for a motor whose amplitude exceeds a predetermined value (eg, 0.7% of average torque) And storing a numerical value (a set or table of numerical values) defined by the determined phase and amplitude in a memory of a motor controller for the motor.

The motor control device and the control method of the present disclosure, and the motor system are various permanent magnet synchronous motors that are required to reduce vibration or noise in order to reduce torque ripple of the motor by current control, and devices or systems including the permanent magnet motor Can be widely applied to.

DESCRIPTION OF SYMBOLS 10 ... 1st circuit, 20 ... 2nd circuit, 30 ... 3rd circuit, 40 ... Vector control circuit, 50 ... 1st conversion circuit, 60 ... 2nd conversion circuit, 70 ... Inverter, 90 ... Processor, 95 ... Memory, 100 ... Permanent magnet synchronous motor, 100S ... Stator, 100R ... Rotor, 120 ... Position sensor, 300 ... Motor control device, 500... Integrated circuit device, 1000... Motor system

Claims

A motor control device for controlling a permanent magnet synchronous motor having a stator and a rotor,

A processor and a memory for storing a program for controlling the operation of the processor;

The processor follows the instructions of the program,

Determining a q-axis current in a dq-axis coordinate system that rotates in synchronization with the rotation of the rotor based on a speed command or a torque command;

A value obtained by superimposing an N-order harmonic current having a phase opposite to the phase of the N-order harmonic component (N is an integer of 2 or more) of the cogging torque in the permanent magnet synchronous motor on the q-axis current. Determining as a command value,

Executing the motor control device.
The motor control apparatus according to claim 1, wherein an amplitude of an Nth harmonic component of torque ripple in the permanent magnet synchronous motor is smaller than an amplitude of an Nth harmonic component of the cogging torque.
The amplitude of the Nth harmonic component of the cogging torque is 0.7% or more of the average torque,

The motor control device according to claim 2, wherein the amplitude of the Nth harmonic component of the torque ripple is 0.03% or less of the average torque.
The memory stores numerical values defined by the phase and amplitude of the Nth-order harmonic component of cogging torque in the permanent magnet synchronous motor,

The processor obtains the numerical value stored in the memory from the memory;

The motor control apparatus in any one of Claim 1 to 3.
A motor control device according to any one of claims 1 to 4,

A motor driving device connected to the motor control device;

A permanent magnet synchronous motor connected to the motor drive device;

A motor system comprising:
The stator is a split stator having a plurality of cores arranged along the circumferential direction of the rotor,

The motor system according to claim 5, wherein N is 2 or 3.
The motor system according to claim 5 or 6, wherein the permanent magnet synchronous motor is a surface magnet type.
The motor system according to any one of claims 5 to 7, wherein a minimum value or a maximum value of the inner diameter of the stator has a difference of 0.09% or more from an average value of the inner diameter.
The motor system according to claim 5, wherein the permanent magnet synchronous motor has an 8-pole 12-slot configuration.
A motor control method for controlling a permanent magnet synchronous motor having a stator and a rotor,

Determining a q-axis current in a dq-axis coordinate system that rotates in synchronization with the rotation of the rotor based on a speed command or a torque command;

Determining the amplitude and phase of the Nth harmonic current having a phase opposite to the phase of the Nth harmonic component of the cogging torque in the permanent magnet synchronous motor;

Determining a value obtained by superimposing the Nth harmonic current on the q-axis current as a q-axis current command value;

Including a motor control method.
A surface magnet type permanent magnet synchronous motor having a split stator and a rotor;

A motor drive connected to the permanent magnet synchronous motor;

A motor control device connected to the motor drive device;

With

The motor control device

A processor and a memory for storing a program for controlling the operation of the processor;

The memory further stores numerical values defined by the phase and amplitude of the Nth harmonic component (N is an integer of 2 or more) of cogging torque in the permanent magnet synchronous motor,

The processor follows the instructions of the program,

Determining a q-axis current in a dq-axis coordinate system that rotates in synchronization with the rotation of the rotor based on a speed command or a torque command;

Determining, as a q-axis current command value, a value obtained by superimposing an Nth-order harmonic current based on the numerical value stored in the memory on the q-axis current;

Run

The motor system in which the amplitude of the Nth harmonic component of the torque ripple in the permanent magnet synchronous motor is smaller than the amplitude of the Nth harmonic component of the cogging torque.