WO2019180912A1 - Voltage switching type direct-current power supply - Google Patents

Voltage switching type direct-current power supply Download PDF

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Publication number
WO2019180912A1
WO2019180912A1 PCT/JP2018/011684 JP2018011684W WO2019180912A1 WO 2019180912 A1 WO2019180912 A1 WO 2019180912A1 JP 2018011684 W JP2018011684 W JP 2018011684W WO 2019180912 A1 WO2019180912 A1 WO 2019180912A1
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Prior art keywords
voltage
switching
parallel
power supply
series
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PCT/JP2018/011684
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French (fr)
Japanese (ja)
Inventor
田中　正一
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田中　正一
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Priority to PCT/JP2018/011684 priority Critical patent/WO2019180912A1/en
Publication of WO2019180912A1 publication Critical patent/WO2019180912A1/en

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    • B—PERFORMING OPERATIONS; TRANSPORTING
    • B60—VEHICLES IN GENERAL
    • B60L—PROPULSION OF ELECTRICALLY-PROPELLED VEHICLES; SUPPLYING ELECTRIC POWER FOR AUXILIARY EQUIPMENT OF ELECTRICALLY-PROPELLED VEHICLES; ELECTRODYNAMIC BRAKE SYSTEMS FOR VEHICLES IN GENERAL; MAGNETIC SUSPENSION OR LEVITATION FOR VEHICLES; MONITORING OPERATING VARIABLES OF ELECTRICALLY-PROPELLED VEHICLES; ELECTRIC SAFETY DEVICES FOR ELECTRICALLY-PROPELLED VEHICLES
    • B60L15/00—Methods, circuits, or devices for controlling the traction-motor speed of electrically-propelled vehicles
    • B60L15/007—Physical arrangements or structures of drive train converters specially adapted for the propulsion motors of electric vehicles
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/32—Means for protecting converters other than automatic disconnection
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/36—Means for starting or stopping converters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
    • H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
    • H02M3/10—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M3/145—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M3/155—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/5387—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
    • H02P27/04—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
    • H02P27/06—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters
    • H02P27/08—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation
    • H02P27/14—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation with three or more levels of voltage
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P3/00—Arrangements for stopping or slowing electric motors, generators, or dynamo-electric converters
    • H02P3/06—Arrangements for stopping or slowing electric motors, generators, or dynamo-electric converters for stopping or slowing an individual dynamo-electric motor or dynamo-electric converter
    • H02P3/08—Arrangements for stopping or slowing electric motors, generators, or dynamo-electric converters for stopping or slowing an individual dynamo-electric motor or dynamo-electric converter for stopping or slowing a dc motor
    • H02P3/14—Arrangements for stopping or slowing electric motors, generators, or dynamo-electric converters for stopping or slowing an individual dynamo-electric motor or dynamo-electric converter for stopping or slowing a dc motor by regenerative braking
    • B—PERFORMING OPERATIONS; TRANSPORTING
    • B60—VEHICLES IN GENERAL
    • B60L—PROPULSION OF ELECTRICALLY-PROPELLED VEHICLES; SUPPLYING ELECTRIC POWER FOR AUXILIARY EQUIPMENT OF ELECTRICALLY-PROPELLED VEHICLES; ELECTRODYNAMIC BRAKE SYSTEMS FOR VEHICLES IN GENERAL; MAGNETIC SUSPENSION OR LEVITATION FOR VEHICLES; MONITORING OPERATING VARIABLES OF ELECTRICALLY-PROPELLED VEHICLES; ELECTRIC SAFETY DEVICES FOR ELECTRICALLY-PROPELLED VEHICLES
    • B60L2210/00—Converter types
    • B60L2210/10—DC to DC converters
    • B60L2210/12—Buck converters
    • B—PERFORMING OPERATIONS; TRANSPORTING
    • B60—VEHICLES IN GENERAL
    • B60L—PROPULSION OF ELECTRICALLY-PROPELLED VEHICLES; SUPPLYING ELECTRIC POWER FOR AUXILIARY EQUIPMENT OF ELECTRICALLY-PROPELLED VEHICLES; ELECTRODYNAMIC BRAKE SYSTEMS FOR VEHICLES IN GENERAL; MAGNETIC SUSPENSION OR LEVITATION FOR VEHICLES; MONITORING OPERATING VARIABLES OF ELECTRICALLY-PROPELLED VEHICLES; ELECTRIC SAFETY DEVICES FOR ELECTRICALLY-PROPELLED VEHICLES
    • B60L2210/00—Converter types
    • B60L2210/10—DC to DC converters
    • B60L2210/14—Boost converters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P2201/00—Indexing scheme relating to controlling arrangements characterised by the converter used
    • H02P2201/09—Boost converter, i.e. DC-DC step up converter increasing the voltage between the supply and the inverter driving the motor
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P3/00—Arrangements for stopping or slowing electric motors, generators, or dynamo-electric converters
    • H02P3/06—Arrangements for stopping or slowing electric motors, generators, or dynamo-electric converters for stopping or slowing an individual dynamo-electric motor or dynamo-electric converter
    • H02P3/18—Arrangements for stopping or slowing electric motors, generators, or dynamo-electric converters for stopping or slowing an individual dynamo-electric motor or dynamo-electric converter for stopping or slowing an ac motor
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02T—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO TRANSPORTATION
    • Y02T10/00—Road transport of goods or passengers
    • Y02T10/60—Other road transportation technologies with climate change mitigation effect
    • Y02T10/64—Electric machine technologies in electromobility
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02T—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO TRANSPORTATION
    • Y02T10/00—Road transport of goods or passengers
    • Y02T10/60—Other road transportation technologies with climate change mitigation effect
    • Y02T10/72—Electric energy management in electromobility

Definitions

  • the present invention relates to a voltage switching DC power supply, and more particularly to a voltage switching DC power supply for a traction motor.
  • FIG. 1 shows a known motor driving device in which a boost chopper 1000 is arranged between a battery 101 and an inverter 102.
  • the boost chopper 1000 boosts the DC link voltage Vd applied to the smoothing capacitor 103 in the high speed region.
  • the weight and loss of the boost chopper 1000 are disadvantages of this motor drive device.
  • a known voltage-switching DC power source having a series switch that connects two batteries in series and two parallel switches that connect two batteries in parallel can avoid the switching loss of the boost chopper 1000.
  • this voltage-switching DC power supply one problem with this voltage-switching DC power supply is that the DC link voltage Vd changes suddenly. A sudden change in the DC link voltage Vd adversely affects the smoothing capacitor and the inverter.
  • Another problem is that the short circuit current flowing between the two batteries increases the battery loss when the voltages of the two batteries connected in parallel are different.
  • FIG. 2 shows an example of a voltage-switching DC power source disclosed in Patent Document 1.
  • Series relay 201 connects batteries 202 and 203 in series.
  • Two parallel relays 204 and 205 connect the batteries 202 and 203 in parallel.
  • the step-up chopper 208 shown in FIG. 2 gradually changes the DC link voltage Vd during a transition period for switching the battery connection.
  • the addition of the step-up chopper 208 increases manufacturing cost and switching loss.
  • a voltage switching DC power supply can employ a capacitor such as a lithium ion capacitor instead of a battery.
  • a capacitor such as a lithium ion capacitor instead of a battery.
  • switching between series connection and parallel connection becomes difficult due to the voltage difference between the capacitor and the battery.
  • One object of the present invention is to provide a voltage-switching DC power source suitable for driving a variable speed motor.
  • the voltage-switching DC power supply has a connection switching circuit for selecting a series connection and a parallel connection of a plurality of charge devices.
  • This connection switching circuit including an inductor in addition to at least one series switch and at least two parallel switches has a chopper mode. According to this chopper mode, in order to avoid magnetic saturation of the inductor, a series switch and / or a parallel switch composed of transistors are switched at a predetermined PWM carrier frequency. This avoids sudden changes in DC link voltage or sudden increases in charge device current.
  • the series switch is turned off when some charging device is defective.
  • the voltage-switching DC power supply can use only the remaining normal charge devices.
  • each of the two sub power supply sets includes two battery blocks and one connection switching circuit. Further, the two sub power supply sets are connected by a third connection switching circuit.
  • the voltage-switching DC power supply can generate three levels of DC link voltage. Furthermore, this voltage-switching DC power supply can execute the 2-parallel discharge mode and the 4-parallel discharge mode when one battery block becomes defective. Therefore, the reliability of the DC power supply is improved.
  • FIG. 1 is a wiring diagram showing a conventional step-up chopper type variable voltage DC power supply.
  • FIG. 2 is a wiring diagram showing a conventional voltage-switching DC power supply.
  • FIG. 3 is a wiring diagram showing the voltage-switching type DC power source of the first embodiment.
  • FIG. 4 is a wiring diagram showing the chopper operation of the connection switching circuit.
  • FIG. 5 is a wiring diagram showing a voltage-switching DC power supply having a relay box for parallel charging.
  • FIG. 6 is a schematic wiring diagram showing a boost chopper type regenerative braking operation.
  • FIG. 7 is a schematic wiring diagram showing a boost chopper type regenerative braking operation.
  • FIG. 8 is a wiring diagram showing the voltage switching type DC power source of the second embodiment.
  • FIG. 1 is a wiring diagram showing a conventional step-up chopper type variable voltage DC power supply.
  • FIG. 2 is a wiring diagram showing a conventional voltage-switching DC power supply.
  • FIG. 9 is a diagram showing a characteristic between the DC link voltage and the motor rotation speed.
  • FIG. 10 is a wiring diagram showing a voltage-switching DC power supply having a relay box for parallel charging.
  • FIG. 11 is a wiring diagram showing a modification.
  • FIG. 12 is a wiring diagram showing a voltage-switching DC power supply according to the third embodiment.
  • FIG. 13 is an equivalent circuit diagram of the voltage-switching DC power source shown in FIG.
  • FIG. 14 is a wiring diagram showing the precharge mode.
  • FIG. 15 is a wiring diagram showing the precharge mode.
  • FIG. 16 is a wiring diagram showing the engine start mode.
  • FIG. 17 is a wiring diagram showing the power generation mode.
  • FIG. 18 is a wiring diagram showing a low-torque type torque assist mode.
  • FIG. 10 is a wiring diagram showing a voltage-switching DC power supply having a relay box for parallel charging.
  • FIG. 11 is a wiring diagram showing a modification.
  • FIG. 19 is a wiring diagram showing a high torque type torque assist mode.
  • FIG. 20 is a wiring diagram showing the regenerative braking mode.
  • FIG. 21 is a wiring diagram showing the regenerative braking mode.
  • FIG. 20 is a wiring diagram showing a voltage-switching DC power supply according to the fourth embodiment.
  • FIG. 23 is a wiring diagram showing the serial mode.
  • FIG. 24 is a wiring diagram showing the parallel discharge mode.
  • FIG. 25 is a wiring diagram showing the parallel charging mode.
  • This voltage-switching DC power supply is connected to an inverter that drives a traction motor of an electric vehicle.
  • This voltage-switching DC power supply can employ a capacitor as a charging device instead of a battery.
  • This voltage-switching DC power supply can be connected to an inverter that drives another variable speed motor.
  • This voltage-switching DC power source includes batteries 1 and 2 and a connection switching circuit 10.
  • the connection switching circuit 10 includes a series transistor 3, a charging diode 4, and parallel diodes 5 and 6.
  • the connection switching circuit 10 further includes an inductor 7.
  • the voltage-switching DC power supply applies a DC link voltage Vd to the smoothing capacitor 20 and the inverter 30.
  • Inverter 30 has three legs 31, 32, and 33.
  • Inverter 30 is connected to stator coil 40 of the three-phase motor.
  • the stator coil 40 includes a U-phase coil 41, a V-phase coil 42, and a W-phase coil 43.
  • Leg 31 is connected to phase coil 41, and leg 32 is connected to phase coil 42.
  • the leg 33 is connected to the phase coil 43.
  • the negative electrode of the battery 1 is connected to the negative terminals of the smoothing capacitor 20 and the inverter 30.
  • the positive electrode of the battery 2 is connected to the positive terminals of the smoothing capacitor 20 and the inverter 30 through the inductor 7.
  • the series transistor 3 and the charging diode 4 connect the batteries 1 and 2 in series.
  • the series transistor 3 can turn off the discharge of the batteries 1 and 2.
  • a charging diode 4 connected in antiparallel to the series transistor 3 can charge the batteries 1 and 2.
  • the anode of the parallel diode 5 is connected to the negative electrode of the battery 1, and the cathode of the parallel diode 5 is connected to the negative electrode of the battery 2.
  • the anode electrode of the parallel diode 6 is connected to the positive electrode of the battery 2, and the cathode electrode of the parallel diode 6 is connected to the positive electrode of the battery 1.
  • batteries 1 and 2 each have a voltage of 320V.
  • the connection switching circuit 10 and the inverter 30 are controlled by the controller 100.
  • the precharge operation of the smoothing capacitor 20 when the key switch of the electric vehicle is turned on will be described. Since the series transistor 3 is off, the smoothing capacitor 20 is charged through the parallel diodes 5 and 6. As a result, the inrush current flowing through the smoothing capacitor 20 is halved and the power loss is 1 ⁇ 4.
  • the series transistor 3 is switched at a predetermined PWM carrier frequency. Its PWM duty ratio, which is equal to the ratio of on period / (on period + off period), is gradually increased from 0 to 1.
  • PWM duty ratio which is equal to the ratio of on period / (on period + off period)
  • the series transistor 3 is turned on, a voltage sum (about 640V) is applied to the smoothing capacitor 20 through the inductor 7, and the inductor 7 stores magnetic energy.
  • the series transistor 3 is turned off, the inductor 7 suppresses a decrease in current flowing through the inductor 7. As a result, the DC link voltage Vd gradually increases from 320V to 640V.
  • the series transistor 3 is switched at a predetermined PWM carrier frequency.
  • the PWM duty ratio is gradually reduced from 1 to 0.
  • the DC link voltage Vd gradually decreases from 640V to 320V.
  • the parallel diodes 5 and 6, the series transistor 3, and the inductor 7 serve as a known step-down chopper.
  • FIG. 4 is a schematic diagram showing the operation of the step-down chopper.
  • the circuit 300 on the left shows a state where the series transistor 3 is turned on, and the circuit 400 on the right shows a state where the series transistor 3 is turned off.
  • the inductor 7 can have a lower inductance value than the inductor of the boost chopper shown in FIG.
  • the switching frequency of the series transistor 3 can have a higher switching frequency value than that of the boost chopper shown in FIG. As a result, the switching loss of the step-down chopper increases. However, since the high frequency switching of the series transistor 3 is only in the mode switching period, this increase in switching loss can be ignored.
  • the connection switching circuit 10 has a relay box 8 and a connector 9.
  • the relay box 8 accommodates two magnet contactors 8A and 8B.
  • the positive electrode of the battery 2 is connected to the positive terminal 9 A of the connector 9, and the negative electrode of the battery 1 is connected to the negative terminal 9 B of the connector 9.
  • Terminals 9A and 9B are connected to an unillustrated external DC power supply.
  • the contactor 8 A connects the negative terminal 9 B to the negative electrode of the battery 2, and the contactor 8 B connects the positive terminal 9 A to the positive electrode of the battery 1.
  • the contactors 8A and 8B are turned on, the contactor 8A is connected in parallel with the parallel diode 5, and the contactor 8B is connected in parallel with the parallel diode 6.
  • the external DC power supply charges the batteries 1 and 2 in parallel. In the parallel discharge mode, it is possible to turn on the contactors 8A and 8B.
  • the contactor 8A is connected in parallel with the parallel diode 5, and the contactor 8B is connected in parallel with the parallel diode 6. Therefore, in the parallel discharge mode, the batteries 1 and 2 are already in the parallel discharge mode by the diodes 5 and 6 before the contactors 8A and 8B are turned on. Therefore, when the contactors 8A and 8B are turned on, the voltage difference between the batteries 1 and 2 can be sufficiently reduced. This means that when the contactors 8A and 8B are turned on, the short-circuit current flowing between the batteries 1 and 2 becomes substantially zero. Furthermore, arcing of the contactors 8A and 8B is prevented by the diodes 5 and 6 when the contactor 8A and / or the contactor 8B are turned off. This improves the life and reliability of the contactors 8A and 8B.
  • the contactor 8A In the parallel charging period of the batteries 1 and 2, when the voltage of the battery 1 exceeds a predetermined value, the contactor 8A is turned off. Similarly, when the voltage of the battery 2 exceeds a predetermined value, the contactor 8B is turned off. In the parallel discharge period of the batteries 1 and 2, when the voltage of the battery 1 becomes less than a predetermined value, the contactor 8A is turned off. Similarly, when the voltage of the battery 2 exceeds a predetermined value, the contactor 8B is turned off. Eventually, the bad battery is separated by two independent contactors 8A and 8B.
  • the contactors 8A and 8B may or may not be turned on.
  • the controller 100 instructs the inverter 30 to perform a boost chopper operation.
  • the three legs 31-33 of the inverter 30 are switched synchronously at a predetermined PWM carrier frequency.
  • Each PWM cycle period is composed of a clamp period and an output period. In the clamp period shown in FIG. 6, the lower arm transistors of the legs 31-33 are simultaneously turned on. In the output period shown in FIG. 7, the lower arm transistors of the legs 31-33 are simultaneously turned off.
  • inverter 30 When the PWM duty ratio equal to the ratio of output period / (clamp period + output period) is low, inverter 30 generates a high output voltage. Thereby, the inverter 30 which functions as a step-up chopper can generate a high charging voltage.
  • the PWM duty ratio is controlled based on the battery charging current. Since the inverter 30 is operated as a step-up chopper, the ripple rate of the charging current supplied from the inverter 30 to the batteries 1 and 2 is increased. However, this ripple rate is reduced by the inductor 7 and the smoothing capacitor 20. The loss of the batteries 1 and 2 is reduced by this ripple rate reduction. Similarly, the losses of the batteries 1 and 2 are reduced in the parallel discharge mode compared to the series discharge mode under low load conditions of the traction motor. This suppresses an increase in battery temperature. In particular, this effect is superior in older batteries with increased internal resistance.
  • a second connection switching circuit 10A and a third connection switching circuit 10B are added to the voltage-switching DC power supply of the first embodiment.
  • the battery 1 of the first embodiment is divided into two battery blocks 1A and 1B each having a rated voltage of 160V.
  • the battery 2 of the first embodiment is divided into two battery blocks 2A and 2B each having a rated voltage of 160V.
  • the inductor 7 of the first embodiment is divided into two inductors 71 and 72. Inductors 71 and 72 can have a common core.
  • Each of the added connection switching circuits 10A and 10B has substantially the same circuit configuration as that of the connection switching circuit 10.
  • the connection switching circuit 10A includes a series transistor 3A having a charging diode 4A, parallel diodes 5A and 6A, and an inductor 7.
  • the series transistor 3A connects the block 2A and the block 2B.
  • the charging diode 4A is connected in antiparallel with the series transistor 3A.
  • the parallel diode 5A connects the negative electrode of the block 2A and the negative electrode of the block 2B.
  • the parallel diode 6A connects the positive electrode of the block 2A and the positive electrode of the block 2B.
  • the connection switching circuit 10B includes a series transistor 3B having a charging diode 4B, parallel diodes 5B and 6B, and an inductor 7.
  • the serial transistor 3B connects the block 1A and the block 1B.
  • the charging diode 4B is connected in antiparallel with the series transistor 3B.
  • the parallel diode 5B connects the negative electrode of the block 1A and the negative electrode of the block 1B.
  • the parallel diode 6B connects the positive electrode of the block 1A and the positive electrode of the block 1B.
  • the controller 100 has a series discharge mode, a 2-parallel discharge mode, and a 4-parallel discharge mode.
  • the 2-parallel discharge mode is essentially the same as the parallel discharge mode of the first embodiment.
  • the series transistor 3 is turned off and the series transistors 3A and 3B are turned on.
  • the DC link voltage Vd is approximately 320V.
  • the series transistors 3, 3A and 3B are turned off.
  • the DC link voltage Vd is approximately 160V.
  • the mode switching operation from the 2-parallel discharge mode to the series discharge mode will be described.
  • the series transistors 3A and 3B are turned off.
  • the serial transistor 3 is switched at a predetermined PWM carrier frequency, and its PWM duty ratio is gradually increased from 0 to 1.
  • the DC link voltage Vd gradually increases from 320V to 640V.
  • another mode switching operation from the series discharge mode to the 2-parallel discharge mode will be described.
  • the serial transistor 3 is switched at a predetermined PWM carrier frequency, and its PWM duty ratio is gradually decreased from 1 to 0.
  • the DC link voltage Vd gradually decreases from 640V to 320V.
  • the parallel diodes 5 and 6, the series transistor 3, and the inductor 7 serve as a known step-down chopper.
  • parallel diodes 5A and 6A, the series transistor 3A, and the inductor 7 function as a known step-down chopper.
  • parallel diodes 5B and 6B, series transistor 3B, and inductor 7 also function as a known step-down chopper.
  • FIG. 9 is a diagram showing the relationship between the motor rotation speed and the DC link voltage Vd when the motor torque is the maximum value.
  • the DC link voltage Vd is, for example, 160 V in a low speed region less than 40 km / h, for example, 320 V in a medium speed region in the range of 40-80 km / h, and 640 V in a high speed region exceeding 80 km / h.
  • connection switching circuit 10 has a relay box 8 and a connector 9 as in the first embodiment.
  • the contactors 81 and 82 can be turned on in the 2-parallel discharge mode and the regenerative braking mode. Thereby, the contactor 81 is connected in parallel with the parallel diode 5, and the contactor 82 is connected in parallel with the diode 6.
  • the DC link voltage Vd is 320 V in the 2-parallel discharge mode and the regenerative braking mode. Since the regenerative braking in this embodiment is essentially the same as in the first embodiment, description thereof is omitted.
  • the contactor 81 is connected in parallel with the parallel diode 5, and the contactor 82 is connected in parallel with the parallel diode 6. Therefore, in the 2-parallel discharge mode, the batteries 1 and 2 are already in the 2-parallel discharge mode by the diodes 5 and 6 before the contactors 81 and 82 are turned on. Therefore, when the contactors 81 and 82 are turned on, the voltage difference between the batteries 1 and 2 can be sufficiently reduced. This means that when the contactors 81 and 82 are turned on, the short-circuit current flowing between the batteries 1 and 2 becomes substantially zero. Furthermore, when the contactor 81 and the contactor 82 are turned off, the arc discharge of the contactors 81 and 82 is prevented by the diodes 5 and 6. Thereby, the lifetime and reliability of the contactors 81 and 82 are improved.
  • the discharge mode in the block failure state which means the case where the voltage of one of the blocks 1A, 1B, 2A, and 2B is out of the allowable range
  • the 2-parallel discharge mode one of the batteries 1 and 2 that does not include a defective block supplies a discharge current. In other words, 50% of the battery cells can continue to discharge.
  • the serial transistor 3 is turned off and the 2-parallel charging mode is employed. Further, the serial transistor 3A, 3B connected to the defective block is turned off. Thereby, 50% of battery cells can continue charging. Eventually, according to the second embodiment using three series transistors, it is possible to select a three-stage DC link voltage Vd and to disconnect a defective battery block.
  • the 4-parallel discharge mode employed in the low speed region significantly reduces battery loss. For this reason, the rise in battery temperature can be suppressed, and the battery life can be extended particularly in a high temperature environment. This effect is particularly noticeable in old batteries with high internal resistance.
  • the charging diode 4 shown in FIG. 5 or FIG. 8 is omitted.
  • the contactors 8A and 8B are turned on during the regenerative braking period, and the inverter 30 charges the batteries 1 and 2 in parallel through the contactors 8A and 8B.
  • a power transistor can be employed in place of the contactors 8A and 8B.
  • FIG. 11 shows another variation.
  • the series transistors 3, 3A, and 3B shown in FIG. 8 each comprise a bidirectional insulated gate transistor.
  • the insulated gate transistor can have a body diode.
  • a 3-series discharge mode in which three of the four blocks 1A, 1B, 2A, and 2B shown in FIG. 8 output the DC link voltage Vd to the inverter 30 is employed.
  • the 3-series discharge mode includes a first mode and a second mode. The first mode and the second mode are preferably executed alternately at a predetermined interval.
  • the series transistors 3 and 3A are turned on, and the series transistor 3B is turned off.
  • the voltage sum of the flocks 2A, 2B, and 1A is applied to the inverter 30.
  • the series transistors 3 and 3B are turned on, and the series transistor 3A is turned off.
  • the voltage sum of the blocks 2B, 1A, and 1B is applied to the inverter 30.
  • Each of the four blocks has about 160V. Therefore, a voltage of 480 V is applied to the inverter 30.
  • series transistor 3 In this 3-series discharge mode, the series transistor 3 is turned on. Series transistors 3A and 3B are complementarily switched at a predetermined interval.
  • blocks 2B and 1A discharge faster than blocks 2A and 1B.
  • the blocks 2A and 1B discharge faster than the blocks 2B and 1A. Thereby, the voltage difference between the blocks 2B and 1A and the blocks 2A and 1B is reduced.
  • the voltage-switching DC power supply can output 480V in addition to 160V, 320V, and 640V.
  • the DC power supply includes a connection switching circuit 10C including a series transistor 3, parallel transistors 50 and 60, and an inductor 7.
  • This DC power supply has the following differences compared to the DC power supply of the first embodiment shown in FIG.
  • Parallel transistors 50 and 60 are employed in place of the parallel diodes 5 and 6.
  • Each of the serial transistor 3 and the parallel transistors 50 and 60 is a MOS transistor having an antiparallel diode.
  • a capacitor 2 is employed instead of the battery 2.
  • the battery 1 has a battery voltage Vb, and the capacitor 2 has a capacitor voltage Vc.
  • the important difference is that the inductor 7 is arranged between the capacitor 2 and the parallel transistor 60.
  • FIG. 13 shows an equivalent circuit of this DC power supply.
  • the internal resistance rc of the capacitor 2 is lower than the internal resistance rb of the battery 1.
  • the battery 1 is composed of a lithium ion battery having a rated voltage of 48V or a lead acid battery having a rated voltage of 14.4V.
  • the capacitor 2 is a lithium ion capacitor.
  • This motor with stator coil 40 is coupled to a pulley driven by the crankshaft of the internal combustion engine. The operation of this DC power source controlled by the controller 100 will be described below.
  • the precharge mode is performed when the capacitor voltage Vc is lower than the battery voltage Vb.
  • the series transistor 3 is turned off and the parallel transistor 60 is turned on.
  • the parallel transistor 50 is switched at a predetermined PWM carrier frequency.
  • the PWM duty ratio of the parallel transistor 50 is gradually increased from 0 to 1.
  • the connection switching circuit 10C including the series transistor 3, the parallel transistors 50 and 60, and the inductor 7 operates as a step-down chopper.
  • Each PWM cycle period consists of a current supply period and a freewheeling period.
  • FIG. 14 shows this current supply period during which the parallel transistor 50 is turned on.
  • the discharge current of the battery 1 flows through the parallel transistor 60, the inductor 7, the capacitor 2, and the parallel transistor 50.
  • the capacitor voltage Vc gradually approaches the battery voltage Vb.
  • FIG. 15 shows a freewheeling period in which the parallel transistor 50 is turned off.
  • the magnetic energy accumulated in the inductor 7 causes a freewheeling current to flow through the series transistor 3, the parallel transistor 60, and the inductor 7.
  • the freewheeling current flows through the antiparallel diode of the series transistor 3.
  • the parallel transistors 50 and 60 are turned off and the series transistor 3 is turned on.
  • the DC link voltage Vdc applied to the inverter 30 is approximately equal to the sum of the capacitor voltage Vc and the battery voltage Vb.
  • the series transistor 3 is turned off.
  • the magnetic energy stored in the inductor 7 causes a freewheeling current to flow through the parallel transistor 50.
  • the inductor 7 is magnetically saturated in this engine start mode.
  • Capacitor Recovery Mode Next, the capacitor recovery mode will be described with reference to FIG.
  • the motor is driven as a generator.
  • the inverter 30 as a three-phase rectifier rectifies the three-phase voltage induced in the stator coil 40.
  • the serial transistor 3 and the parallel transistor 60 are turned off, and the parallel transistor 50 is turned on. Thereby, the capacitor 2 is charged.
  • the inductor 7 is magnetically saturated in this capacitor recovery mode.
  • the power generation mode will be described with reference to FIG.
  • the parallel transistor 60 is turned on.
  • the parallel transistor 50 is turned on and the series transistor 3 is turned off.
  • the inverter 30 charges the capacitor 2 and the battery 1 connected in parallel.
  • the inverter 30 supplies power to the electric load connected to the battery 1.
  • Torque assist mode Next, a low level type torque assist mode will be described with reference to FIG. Series transistor 3 is turned off and parallel transistors 50 and 60 are turned on.
  • the controller 100 controls the inverter 30, and motor current is supplied to the stator coil 40 through the inverter 30 from the battery 1 and the capacitor 2 connected in parallel.
  • This parallel regenerative braking mode is essentially the same as the power generation mode described above.
  • the serial transistor 3 is turned off and the parallel transistors 50 and 60 are turned on.
  • Inverter 30 charges capacitor 2 and battery 1 connected in parallel.
  • Capacitor Rechargeable Regenerative Braking Mode the capacitor rechargeable regenerative braking mode will be described with reference to FIG. This mode is employed, for example, when the battery 1 has a high SOC (charge state) value.
  • the serial transistor 3 and the parallel transistor 60 are turned off, and only the parallel transistor 50 is turned on. Thereby, only the capacitor 2 absorbs the generated current of the inverter 30.
  • the capacitor voltage Vc can be higher than the battery voltage Vb.
  • the capacitor regenerative braking mode is stopped. Thereby, the braking energy is recovered satisfactorily regardless of the state of the battery 1.
  • the capacitor 2 When the capacitor voltage Vc is higher than the battery voltage Vb, the capacitor 2 is preferably discharged to the inverter 30 or the battery 1. In the discharging operation from the capacitor 2 to the battery 1, the series transistor 3 is turned off and the parallel transistor 50 is turned on. The parallel transistor 60 is switched at a predetermined PWM carrier frequency. The PWM duty ratio of the parallel transistor 60 is gradually increased from 0 to 1. Thereby, the capacitor voltage Vc becomes substantially equal to the battery voltage Vb.
  • the inverter 30 can execute the boost chopper mode. For example, when the rectified voltage of the inverter 30 decreases due to a decrease in the motor rotation speed, the boost chopper mode is executed.
  • this step-up chopper mode one of the three upper arm transistors 34-36 or the three lower arm transistors 37-39 shown in FIG. 12 is switched synchronously at a predetermined PWM duty ratio. Upper arm transistors 34-36 and lower arm transistors 37-39 are complementarily switched.
  • connection switching circuit 10c can circulate the freewheeling current of the inductor 7.
  • the effect of the third embodiment will be described. According to this voltage-switching DC power supply, it is possible to avoid the addition of an expensive battery.
  • the capacitor 2 used only for a short time can have a lower capacitance value than the battery 1.
  • Capacitor 2 with low internal resistance and long life reduces power loss for engine start, torque assist, and regenerative braking.
  • Capacitor 2 extends the life of battery 2.
  • connection switching circuit 10c has both a voltage switching function and a step-down chopper function.
  • this step-down chopper operation the capacitor voltage Vc and the battery voltage Vb are equalized.
  • engine start-up, regenerative braking, and torque assist can be performed by the capacitor 2 alone. Thereby, the lifetime of the battery 1 is further extended.
  • This DC power supply includes a connection switching circuit 10 d including a series transistor 3, parallel transistors 50 and 60, and an inductor 7. Further, the connection switching circuit 10 d has a bypass diode 73.
  • the voltage V1 of the battery 1 is approximately equal to the voltage V2 of the battery 2.
  • the main feature of this DC power supply is that the inductor 7 of the third embodiment shown in FIG. 12 is divided into two sub-inductors 71 and 72.
  • the two sub-inductors 71 and 72 have one ends connected to the inverter 30 and the smoothing capacitor 20. Each of the two subcoils 71 and 72 has the other end connected by a bypass diode 73. The anode electrode of the bypass diode 73 is connected to the sub-inductor 72, and the cathode electrode thereof is connected to the sub-inductor 71. It is also possible to employ the capacitor 2 shown in FIG. 12 instead of the battery 2 shown in FIG. The sub-inductor 71 is connected to the positive electrode of the battery 2, and the sub-inductor 72 is connected to the parallel transistor 60. Each coil of the sub-inductors 71 and 72 can be wound around a common magnetic core.
  • a fuse 81 is connected in series with the parallel transistor 50, and a fuse 82 is connected in series with the parallel transistor 60. It is also possible to connect a third fuse in series with the series transistor 3.
  • the smoothing capacitor 20 includes capacitors 21 and 22 connected in parallel.
  • the capacitor 21 is made of a lithium ion capacitor having a high capacitance value
  • the capacitor 22 is made of a film capacitor having excellent high frequency characteristics.
  • the DC power source having the connection switching circuit 10 d is connected to the smoothing capacitor 20 and the inverter 30 through system relays 91 and 92.
  • the connection switching circuit 10d has a parallel mode and a series mode.
  • the parallel mode is described with reference to FIG.
  • the serial transistor 3 is turned off and the parallel transistors 50 and 60 are turned on. Thereby, the batteries 1 and 2 are connected in parallel.
  • the serial mode is described with reference to FIG.
  • the parallel transistors 50 and 60 are turned off, and the series transistor 3 is turned on. Thereby, the batteries 1 and 2 are connected in series, and the DC link voltage Vdc equal to the voltage sum (V1 + V2) is applied to the inverter 30.
  • the PWM duty ratio of the serial transistor 3 gradually changes from 0 to 1
  • the PWM duty ratio of the parallel transistors 50 and 60 gradually changes from 1 to 0.
  • the PWM duty ratio of the serial transistor 3 gradually changes from 1 to 0, and the PWM duty ratio of the parallel transistors 50 and 60 gradually changes from 0 to 1.
  • the parallel transistors 50 and 60 are driven in a complementary manner as compared with the serial transistor 3.
  • each PWM cycle period is composed of a series period and a parallel period.
  • the series period the series transistor 3 is turned on and the parallel transistors 50 and 60 are turned off.
  • the parallel period the series transistor 3 is turned off and the parallel transistors 50 and 60 are turned on.
  • the series period includes a series discharge period and a series charge period.
  • the parallel period includes a parallel discharge period and a parallel charge period.
  • a series discharge period in which the batteries 1 and 2 are discharged will be described with reference to FIG.
  • the sub-inductor 71 stores magnetic energy.
  • the parallel discharge period will be described with reference to FIG.
  • the magnetic energy stored in the sub-inductor 71 is consumed by the freewheeling current that flows through the parallel transistor 50.
  • the parallel transistor 60 is turned on, and the sub-inductor 72 stores magnetic energy.
  • the parallel transistor 60 is turned off, and the magnetic energy stored in the sub-inductor 72 is consumed by the freewheeling current flowing through the parallel transistor 60.
  • a series charging period in which the batteries 1 and 2 are charged will be described with reference to FIG.
  • the sub-inductor 71 stores magnetic energy.
  • a freewheeling current circulates through the antiparallel diode of the series transistor 3.
  • the parallel charging period will be described with reference to FIG.
  • the parallel transistor 60 is turned off, the sub-inductors 71 and 72 store magnetic energy.
  • the parallel transistor 50 is turned off, the freewheeling current circulates through the sub-inductor 71 and the antiparallel diode of the series transistor 3.
  • the parallel transistor 60 is turned off, the freewheeling current circulates through the sub-inductor 72 and the bypass diode 73.
  • the battery voltage V1 and the battery voltage V2 are gradually equalized by the parallel mode and the parallel period of the voltage switching period.
  • the parallel transistor 50 is turned off.
  • the parallel transistor 60 is turned off. Thereby, the reliability of the DC power supply is improved.
  • inductor 7 forms a low-pass filter together with the smoothing capacitor 20, the high-frequency component of the battery current is reduced. This high frequency component of the battery current causes useless power loss in the batteries 1 and 2 and raises their temperature. Thus, inductor 7 simplifies the cooling mechanism for batteries 1 and 2 and extends the life of batteries 1 and 2.
  • the smoothing capacitor 20 has a lithium ion capacitor 21 having a high capacitance value in addition to the conventional film capacitor 22.
  • the high frequency component of the electric current which flows through the batteries 1 and 2 is reduced, and the power loss of the batteries 1 and 2 is reduced.
  • the increase in the capacity of the smoothing capacitor 20 increases the inrush current flowing through the smoothing capacitor 20.
  • the capacity increase of the smoothing capacitor 20 has a limit.
  • this inrush current is suppressed by the step-down chopper operation of the connection switching circuit having the inductor 7. Therefore, the smoothing capacitor 20 of this embodiment can include a lithium capacitor having a higher capacity than a conventional film capacitor.
  • the precharging operation of the smoothing capacitor 20 will be described.
  • the series transistor 3 is turned off, and one or both of the parallel transistors 50 and 60 are switched at a predetermined PWM carrier frequency.
  • the parallel transistors 50 and / or 60 have a predetermined PWM duty ratio.
  • the smoothing capacitor 20 can be gradually charged.
  • the connection switching circuit 10d for switching the amplitude of the DC link voltage Vdc can reduce the inrush current to the smoothing capacitor 20.
  • the loss of the smoothing capacitor 20 and the batteries 1 and 2 is reduced.

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Transportation (AREA)
  • Mechanical Engineering (AREA)
  • Dc-Dc Converters (AREA)
  • Electric Propulsion And Braking For Vehicles (AREA)

Abstract

Provided is a voltage switching type direct-current power supply which is capable of reducing loss. This direct-current power supply has a connection switching circuit which can select a serial connection or a parallel connection of two or more charge devices. The direct-current power supply applies a two-level or three-level DC link voltage to an inverter for variable speed motor driving. The connection switching circuit, which has an inductor, performs a chopper operation in order to avoid sudden changes in the DC link voltage and in order to reduce voltage differences among the plurality of charge devices. In this chopper operation, series transistors or parallel transistors of the connection switching circuit are switched at a prescribed PWM carrier frequency.

Description

電圧切替式直流電源Voltage switching DC power supply
本発明は電圧切替式直流電源に関し、特にトラクションモータ用の電圧切替式直流電源に関する。 The present invention relates to a voltage switching DC power supply, and more particularly to a voltage switching DC power supply for a traction motor.
トラクションモータ用のインバータに印加されるDCリンク電圧はモータ回転数に応じて変更されることが好適である。図１は、昇圧チョッパ１０００がバッテリ１０１とインバータ１０２との間に配置される周知のモータ駆動装置を示す。昇圧チョッパ１０００は高速領域において平滑キャパシタ１０３に印加されるDCリンク電圧Vdを昇圧する。しかし、昇圧チョッパ１０００の重量及び損失はこのモータ駆動装置の欠点となっている。 The DC link voltage applied to the inverter for the traction motor is preferably changed according to the motor rotation speed. FIG. 1 shows a known motor driving device in which a boost chopper 1000 is arranged between a battery 101 and an inverter 102. The boost chopper 1000 boosts the DC link voltage Vd applied to the smoothing capacitor 103 in the high speed region. However, the weight and loss of the boost chopper 1000 are disadvantages of this motor drive device.
２つのバッテリを直列接続する直列スイッチと、２つのバッテリを並列接続する２つの並列スイッチとをもつ公知の電圧切替式直流電源は、上記昇圧チョッパ１０００のスイッチング損失を回避することができる。しかし、この電圧切替式直流電源の１つの問題はDCリンク電圧Vdが急変することである。DCリンク電圧Vdの急変は平滑キャパシタ及びインバータに悪影響を与える。もう１つの問題は並列接続された２つのバッテリの電圧が異なる時、２つのバッテリ間を流れる短絡電流がバッテリ損失を増加させることである。 A known voltage-switching DC power source having a series switch that connects two batteries in series and two parallel switches that connect two batteries in parallel can avoid the switching loss of the boost chopper 1000. However, one problem with this voltage-switching DC power supply is that the DC link voltage Vd changes suddenly. A sudden change in the DC link voltage Vd adversely affects the smoothing capacitor and the inverter. Another problem is that the short circuit current flowing between the two batteries increases the battery loss when the voltages of the two batteries connected in parallel are different.
図２は特許文献１に開示される電圧切替式直流電源の一例を示す。直列リレー２０１はバッテリ２０２及び２０３を直列に接続する。２つの並列リレー２０４及び２０５はバッテリ２０２及び２０３を並列接続する。図２に示される昇圧チョッパ２０８はバッテリの接続を切り替えるための過渡期間においてDCリンク電圧Vdを徐々に変更する。しかし、この昇圧チョッパ２０８の追加は製造コスト及びスイッチングロスを増加させる。 FIG. 2 shows an example of a voltage-switching DC power source disclosed in Patent Document 1. Series relay 201 connects batteries 202 and 203 in series. Two parallel relays 204 and 205 connect the batteries 202 and 203 in parallel. The step-up chopper 208 shown in FIG. 2 gradually changes the DC link voltage Vd during a transition period for switching the battery connection. However, the addition of the step-up chopper 208 increases manufacturing cost and switching loss.
電圧切替式直流電源はバッテリの代わりにたとえばリチウムイオンキャパシタのようなキャパシタを採用することができる。しかし、キャパシタ及びバッテリの間の電圧差により、直列接続と並列接続の切替は困難となる。 A voltage switching DC power supply can employ a capacitor such as a lithium ion capacitor instead of a battery. However, switching between series connection and parallel connection becomes difficult due to the voltage difference between the capacitor and the battery.
特開２０１２-０６０８３８号公報JP 2012-060838 A
本発明の一つの目的は可変速モータの駆動に好適な電圧切替式直流電源を提供することである。 One object of the present invention is to provide a voltage-switching DC power source suitable for driving a variable speed motor.
本発明の１つの様相によれば、電圧切替式直流電源は、複数のチャージデバイスの直列接続及び並列接続を選択するための接続切替回路をもつ。少なくとも１つの直列スイッチ及び少なくとも２つの並列スイッチの他にインダクタを含むこの接続切替回路はチョッパモードをもつ。このチョッパモードによれば、インダクタの磁気飽和を回避するために、トランジスタからなる直列スイッチ及び／又は並列スイッチが所定のPWMキャリヤ周波数でスイッチングされる。これにより、DCリンク電圧の急変又はチャージデバイスの電流の急増が回避される。 According to one aspect of the present invention, the voltage-switching DC power supply has a connection switching circuit for selecting a series connection and a parallel connection of a plurality of charge devices. This connection switching circuit including an inductor in addition to at least one series switch and at least two parallel switches has a chopper mode. According to this chopper mode, in order to avoid magnetic saturation of the inductor, a series switch and / or a parallel switch composed of transistors are switched at a predetermined PWM carrier frequency. This avoids sudden changes in DC link voltage or sudden increases in charge device current.
１つの好適な態様において、一部のチャージデバイスが不良である時、直列スイッチがオフされる。これにより、電圧切替式直流電源は残りの正常なチャージデバイスだけを使用することができる。 In one preferred aspect, the series switch is turned off when some charging device is defective. As a result, the voltage-switching DC power supply can use only the remaining normal charge devices.
本発明のもう１つの様相によれば、２つのサブ電源セットが採用される。２つのサブ電源セットはそれぞれ、２つのバッテリブロック及び１つの接続切替回路からなる。さらに、２つのサブ電源セットは第３の接続切替回路により接続される。これにより、電圧切替式直流電源は３つのレベルのDCリンク電圧を発生することができる。さらに、この電圧切替式直流電源は、１つのバッテリブロックが不良となる場合において、２-並列放電モード及び４-並列放電モードを実行することができる。したがって、直流電源の信頼性が改善される。 According to another aspect of the present invention, two sub power supply sets are employed. Each of the two sub power supply sets includes two battery blocks and one connection switching circuit. Further, the two sub power supply sets are connected by a third connection switching circuit. As a result, the voltage-switching DC power supply can generate three levels of DC link voltage. Furthermore, this voltage-switching DC power supply can execute the 2-parallel discharge mode and the 4-parallel discharge mode when one battery block becomes defective. Therefore, the reliability of the DC power supply is improved.
図１は従来の昇圧チョッパ式可変電圧直流電源を示す配線図である。FIG. 1 is a wiring diagram showing a conventional step-up chopper type variable voltage DC power supply. 図２は従来の電圧切替式直流電源を示す配線図である。FIG. 2 is a wiring diagram showing a conventional voltage-switching DC power supply. 図３は第１実施例の電圧切替式直流電源を示す配線図である。FIG. 3 is a wiring diagram showing the voltage-switching type DC power source of the first embodiment. 図４は接続切替回路のチョッパ動作を示す配線図である。FIG. 4 is a wiring diagram showing the chopper operation of the connection switching circuit. 図５は並列充電用のリレーボックスをもつ電圧切替式直流電源を示す配線図である。FIG. 5 is a wiring diagram showing a voltage-switching DC power supply having a relay box for parallel charging. 図６は昇圧チョッパ式回生制動動作を示す模式配線図である。FIG. 6 is a schematic wiring diagram showing a boost chopper type regenerative braking operation. 図７は昇圧チョッパ式回生制動動作を示す模式配線図である。FIG. 7 is a schematic wiring diagram showing a boost chopper type regenerative braking operation. 図８は第２実施例の電圧切替式直流電源を示す配線図である。FIG. 8 is a wiring diagram showing the voltage switching type DC power source of the second embodiment. 図９はDCリンク電圧とモータ回転数との間の特性を示す図である。FIG. 9 is a diagram showing a characteristic between the DC link voltage and the motor rotation speed. 図１０は並列充電用のリレーボックスをもつ電圧切替式直流電源を示す配線図である。FIG. 10 is a wiring diagram showing a voltage-switching DC power supply having a relay box for parallel charging. 図１１は変形態様を示す配線図である。FIG. 11 is a wiring diagram showing a modification. 図１２は第３実施例の電圧切替式直流電源を示す配線図である。FIG. 12 is a wiring diagram showing a voltage-switching DC power supply according to the third embodiment. 図１３は図１２に示される電圧切替式直流電源の等価回路図である。FIG. 13 is an equivalent circuit diagram of the voltage-switching DC power source shown in FIG. 図１４は予備充電モードを示す配線図である。FIG. 14 is a wiring diagram showing the precharge mode. 図１５は予備充電モードを示す配線図である。FIG. 15 is a wiring diagram showing the precharge mode. 図１６はエンジン始動モードを示す配線図である。FIG. 16 is a wiring diagram showing the engine start mode. 図１７は発電モードを示す配線図である。FIG. 17 is a wiring diagram showing the power generation mode. 図１８は低トルクタイプのトルクアシストモードを示す配線図である。FIG. 18 is a wiring diagram showing a low-torque type torque assist mode. 図１９は高トルクタイプのトルクアシストモードを示す配線図である。FIG. 19 is a wiring diagram showing a high torque type torque assist mode. 図２０は回生制動モードを示す配線図である。FIG. 20 is a wiring diagram showing the regenerative braking mode. 図２１は回生制動モードを示す配線図である。FIG. 21 is a wiring diagram showing the regenerative braking mode. 図２０は第４実施例の電圧切替式直流電源を示す配線図である。FIG. 20 is a wiring diagram showing a voltage-switching DC power supply according to the fourth embodiment. 図２３は直列モードを示す配線図である。FIG. 23 is a wiring diagram showing the serial mode. 図２４は並列放電モードを示す配線図である。FIG. 24 is a wiring diagram showing the parallel discharge mode. 図２５は並列充電モードを示す配線図である。FIG. 25 is a wiring diagram showing the parallel charging mode.
本発明の電圧切替式直流電源の好適な実施形態が図面を参照して説明される。この電圧切替式直流電源は電気自動車のトラクションモータを駆動するインバータに接続される。この電圧切替式直流電源はバッテリの代わりにキャパシタをチャージデバイスとして採用することができる。この電圧切替式直流電源は他の可変速モータを駆動するインバータに接続されることができる。 A preferred embodiment of the voltage-switching DC power source of the present invention will be described with reference to the drawings. This voltage-switching DC power supply is connected to an inverter that drives a traction motor of an electric vehicle. This voltage-switching DC power supply can employ a capacitor as a charging device instead of a battery. This voltage-switching DC power supply can be connected to an inverter that drives another variable speed motor.
　　　　　第１実施例
電気自動車に適用される第１実施例の電圧切替式直流電源が図３-図７を参照して説明される。この電圧切替式直流電源はバッテリ１及び２及び接続切替回路１０からなる。接続切替回路１０は直列トランジスタ３、充電ダイオード４、並列ダイオード５及び６からなる。接続切替回路１０はさらにインダクタ７を含む。電圧切替式直流電源は平滑キャパシタ２０及びインバータ３０にDCリンク電圧Vdを印加する。インバータ３０は３つのレグ３１、３２、及び３３をもつ。インバータ３０は３相モータのステータコイル４０に接続される。ステータコイル４０はU相コイル４１、V相コイル４２、及びW相コイル４３からなる。レグ３１は相コイル４１に接続され、レグ３２は相コイル４２に接続され。レグ３３は相コイル４３に接続されている。 First Embodiment A voltage-switching DC power supply according to a first embodiment applied to an electric vehicle will be described with reference to FIGS. This voltage-switching DC power source includes batteries 1 and 2 and a connection switching circuit 10. The connection switching circuit 10 includes a series transistor 3, a charging diode 4, and parallel diodes 5 and 6. The connection switching circuit 10 further includes an inductor 7. The voltage-switching DC power supply applies a DC link voltage Vd to the smoothing capacitor 20 and the inverter 30. Inverter 30 has three legs 31, 32, and 33. Inverter 30 is connected to stator coil 40 of the three-phase motor. The stator coil 40 includes a U-phase coil 41, a V-phase coil 42, and a W-phase coil 43. Leg 31 is connected to phase coil 41, and leg 32 is connected to phase coil 42. The leg 33 is connected to the phase coil 43.
バッテリ１の負極は平滑キャパシタ２０及びインバータ３０の各負端子に接続されている。バッテリ２の正極はインダクタ７を通じて平滑キャパシタ２０及びインバータ３０の各正端子に接続されている。直列トランジスタ３及び充電ダイオード４はバッテリ１及び２を直列接続する。直列トランジスタ３はバッテリ１及び２の放電をオフすることができる。直列トランジスタ３に逆並列接続された充電ダイオード４はバッテリ１及び２を充電することができる。 The negative electrode of the battery 1 is connected to the negative terminals of the smoothing capacitor 20 and the inverter 30. The positive electrode of the battery 2 is connected to the positive terminals of the smoothing capacitor 20 and the inverter 30 through the inductor 7. The series transistor 3 and the charging diode 4 connect the batteries 1 and 2 in series. The series transistor 3 can turn off the discharge of the batteries 1 and 2. A charging diode 4 connected in antiparallel to the series transistor 3 can charge the batteries 1 and 2.
並列ダイオード５のアノード電極はバッテリ１の負極に接続され、並列ダイオード５のカソードはバッテリ２の負極に接続されている。並列ダイオード６のアノード電極はバッテリ２の正極に接続され、並列ダイオード６のカソード電極はバッテリ１の正極に接続されている。 The anode of the parallel diode 5 is connected to the negative electrode of the battery 1, and the cathode of the parallel diode 5 is connected to the negative electrode of the battery 2. The anode electrode of the parallel diode 6 is connected to the positive electrode of the battery 2, and the cathode electrode of the parallel diode 6 is connected to the positive electrode of the battery 1.
たとえば、バッテリ１及び２はそれぞれ３２０Vの電圧をもつ。接続切替回路１０及びインバータ３０はコントローラ１００により制御される。まず、電気自動車のキースイッチがオンされた時の平滑キャパシタ２０のプリチャージ動作が説明される。直列トランジスタ３はオフされているので、平滑キャパシタ２０は並列ダイオード５及び６を通じて充電される。これにより、平滑キャパシタ２０に流れる突入電流は半分となり、電力損失は１/４となる。 For example, batteries 1 and 2 each have a voltage of 320V. The connection switching circuit 10 and the inverter 30 are controlled by the controller 100. First, the precharge operation of the smoothing capacitor 20 when the key switch of the electric vehicle is turned on will be described. Since the series transistor 3 is off, the smoothing capacitor 20 is charged through the parallel diodes 5 and 6. As a result, the inrush current flowing through the smoothing capacitor 20 is halved and the power loss is ¼.
この電圧切替式直流電源の放電動作が説明される。直列トランジスタ３がオンされる直列放電モードにおいて、DCリンク電圧Vdはバッテリ１及び２の電圧和（＝６４０V）と等しくなる。直列トランジスタ３がオフされる並列放電モードにおいて、DCリンク電圧Vdは、バッテリ１及び２の高い方の電圧（＝約３２０V）と等しくなる。言い換えれば、２つのバッテリ１及び２の電圧差は並列ダイオード５及び６により並列放電モードにおいて自動的に解消される。 The discharge operation of this voltage-switching DC power supply will be described. In the series discharge mode in which the series transistor 3 is turned on, the DC link voltage Vd becomes equal to the voltage sum of the batteries 1 and 2 (= 640 V). In the parallel discharge mode in which the series transistor 3 is turned off, the DC link voltage Vd becomes equal to the higher voltage (= about 320 V) of the batteries 1 and 2. In other words, the voltage difference between the two batteries 1 and 2 is automatically eliminated in the parallel discharge mode by the parallel diodes 5 and 6.
次に、並列放電モードから直列放電モードへのモード切替動作が説明される。このモード切替動作において、直列トランジスタ３は所定のPWMキャリヤ周波数でスイッチングされる。オン期間/（オン期間+オフ期間）の比率に等しいそのPWMデユーティ比は０から１へ徐々に増加される。直列トランジスタ３がオンされる時、電圧和（約６４０V）がインダクタ７を通じて平滑キャパシタ２０に印加され、インダクタ７は磁気エネルギーを蓄積する。直列トランジスタ３がオフされる時、インダクタ７はインダクタ７を流れる電流の減少を抑制する。これにより、DCリンク電圧Vdは３２０Vから６４０Vへ徐々に上昇する。 Next, the mode switching operation from the parallel discharge mode to the series discharge mode will be described. In this mode switching operation, the series transistor 3 is switched at a predetermined PWM carrier frequency. Its PWM duty ratio, which is equal to the ratio of on period / (on period + off period), is gradually increased from 0 to 1. When the series transistor 3 is turned on, a voltage sum (about 640V) is applied to the smoothing capacitor 20 through the inductor 7, and the inductor 7 stores magnetic energy. When the series transistor 3 is turned off, the inductor 7 suppresses a decrease in current flowing through the inductor 7. As a result, the DC link voltage Vd gradually increases from 320V to 640V.
次に、直列放電モードから並列放電モードへのもう１つのモード切替動作が説明される。このモード切替動作において、直列トランジスタ３は所定のPWMキャリヤ周波数でスイッチングされる。そのPWMデユーティ比は１から０へ徐々に減少される。これにより、DCリンク電圧Vdは６４０Vから３２０Vへ徐々に低下する。結局、並列ダイオード５及び６、直列トランジスタ３、及びインダクタ７は、公知の降圧チョッパとして働く。 Next, another mode switching operation from the series discharge mode to the parallel discharge mode will be described. In this mode switching operation, the series transistor 3 is switched at a predetermined PWM carrier frequency. The PWM duty ratio is gradually reduced from 1 to 0. As a result, the DC link voltage Vd gradually decreases from 640V to 320V. Eventually, the parallel diodes 5 and 6, the series transistor 3, and the inductor 7 serve as a known step-down chopper.
図４は、この降圧チョッパの動作を示す模式図である。左側の回路３００は直列トランジスタ３がオンされた状態を示し、右側の回路４００は直列トランジスタ３がオフされた状態を示す。インダクタ７は、図１に示される昇圧チョッパのインダクタよりも低いインダクタンス値をもつことができる。直列トランジスタ３のスイッチング周波数は、図１に示される昇圧チョッパよりも高いスイッチング周波数値をもつことができる。その結果、降圧チョッパのスイッチング損失は増加する。しかし、直列トランジスタ３の高周波スイッチングはモード切替期間だけであるため、このスイッチング損失の増加は無視されることができる。 FIG. 4 is a schematic diagram showing the operation of the step-down chopper. The circuit 300 on the left shows a state where the series transistor 3 is turned on, and the circuit 400 on the right shows a state where the series transistor 3 is turned off. The inductor 7 can have a lower inductance value than the inductor of the boost chopper shown in FIG. The switching frequency of the series transistor 3 can have a higher switching frequency value than that of the boost chopper shown in FIG. As a result, the switching loss of the step-down chopper increases. However, since the high frequency switching of the series transistor 3 is only in the mode switching period, this increase in switching loss can be ignored.
次に、外部直流電源によるバッテリ１及び２の充電動作が図５を参照して説明される。バッテリ１及び２を並列に充電するために、この外部直流電源は約３２０Vの電圧をバッテリ１及び２に印加する。この並列充電のために、接続切替回路１０はリレーボックス８及びコネクタ９をもつ。リレーボックス８は、２つのマグネットコンタクタ８A及び８Bを収容する。バッテリ２の正極はコネクタ９の正極端子９Aに接続され、バッテリ１の負極はコネクタ９の負極端子９Bに接続されている。端子９A及び９Bは図略の図略の外部直流電源に接続される。 Next, the charging operation of the batteries 1 and 2 by the external DC power supply will be described with reference to FIG. In order to charge the batteries 1 and 2 in parallel, the external DC power supply applies a voltage of about 320V to the batteries 1 and 2. For this parallel charging, the connection switching circuit 10 has a relay box 8 and a connector 9. The relay box 8 accommodates two magnet contactors 8A and 8B. The positive electrode of the battery 2 is connected to the positive terminal 9 A of the connector 9, and the negative electrode of the battery 1 is connected to the negative terminal 9 B of the connector 9. Terminals 9A and 9B are connected to an unillustrated external DC power supply.
コンタクタ８Aは負極端子９Bをバッテリ２の負極に接続し、コンタクタ８Bは正極端子９Aをバッテリ１の正極に接続する。コンタクタ８A及び８Bがオンされる時、コンタクタ８Aは並列ダイオード５と並列に接続され、コンタクタ８Bは並列ダイオード６と並列に接続される。これにより、外部直流電源はバッテリ１及び２を並列に充電する。並列放電モードにおいて、コンタクタ８A及び８Bをオンすることは可能である。 The contactor 8 A connects the negative terminal 9 B to the negative electrode of the battery 2, and the contactor 8 B connects the positive terminal 9 A to the positive electrode of the battery 1. When the contactors 8A and 8B are turned on, the contactor 8A is connected in parallel with the parallel diode 5, and the contactor 8B is connected in parallel with the parallel diode 6. Thereby, the external DC power supply charges the batteries 1 and 2 in parallel. In the parallel discharge mode, it is possible to turn on the contactors 8A and 8B.
この実施例によれば、コンタクタ８Aが並列ダイオード５と並列接続され、コンタクタ８Bが並列ダイオード６と並列接続されている。したがって、並列放電モードにおいて、コンタクタ８A及び８Bがオンされる前に、バッテリ１及び２はダイオード５及び６により既に並列放電モードとなっている。したがって、コンタクタ８A及び８Bがオンされる時、バッテリ１及び２の間の電圧差は十分に低減されることができる。これは、コンタクタ８A及び８Bがオンされる時、バッテリ１及び２の間に流れる短絡電流がほぼゼロとなることを意味する。さらに、コンタクタ８A及び/又はコンタクタ８Bがオフされる時、コンタクタ８A及び８Bのアーク放電はダイオード５及び６により防止される。これにより、コンタクタ８A及び８Bの寿命及び信頼性が改善される。 According to this embodiment, the contactor 8A is connected in parallel with the parallel diode 5, and the contactor 8B is connected in parallel with the parallel diode 6. Therefore, in the parallel discharge mode, the batteries 1 and 2 are already in the parallel discharge mode by the diodes 5 and 6 before the contactors 8A and 8B are turned on. Therefore, when the contactors 8A and 8B are turned on, the voltage difference between the batteries 1 and 2 can be sufficiently reduced. This means that when the contactors 8A and 8B are turned on, the short-circuit current flowing between the batteries 1 and 2 becomes substantially zero. Furthermore, arcing of the contactors 8A and 8B is prevented by the diodes 5 and 6 when the contactor 8A and / or the contactor 8B are turned off. This improves the life and reliability of the contactors 8A and 8B.
バッテリ１及び２の並列充電期間において、バッテリ１の電圧が所定値を超える時、コンタクタ８Aがオフされる。同様に、バッテリ２の電圧が所定値を超える時、コンタクタ８Bがオフされる。バッテリ１及び２の並列放電期間において、バッテリ１の電圧が所定値未満となる時、コンタクタ８Aがオフされる。同様に、バッテリ２の電圧が所定値を超える時、コンタクタ８Bがオフされる。結局、不良バッテリは２つの独立コンタクタ８A及び８Bにより分離される。 In the parallel charging period of the batteries 1 and 2, when the voltage of the battery 1 exceeds a predetermined value, the contactor 8A is turned off. Similarly, when the voltage of the battery 2 exceeds a predetermined value, the contactor 8B is turned off. In the parallel discharge period of the batteries 1 and 2, when the voltage of the battery 1 becomes less than a predetermined value, the contactor 8A is turned off. Similarly, when the voltage of the battery 2 exceeds a predetermined value, the contactor 8B is turned off. Eventually, the bad battery is separated by two independent contactors 8A and 8B.
次に、この電圧切替式直流電源の回生制動期間における充電動作が図６及び図７を参照して説明される。コンタクタ８A及び８Bはオンされてもよく、オンされなくてもよい。この回生制動において、インバータ３０の発電電圧がDCリンク電圧よりも低い時、コントローラ１００はインバータ３０に昇圧チョッパ動作を指令する。この昇圧チョッパ動作において、インバータ３０の３つのレグ３１-３３は所定のPWMキャリヤ周波数で同期的にスイッチングされる。各PWMサイクル期間はそれぞれクランプ期間と出力期間とからなる。図６に示されるクランプ期間において、レグ３１-３３の下アームトランジスタが同時にオンされる。図７に示される出力期間において、レグ３１-３３の下アームトランジスタが同時にオフされる。 Next, the charging operation in the regenerative braking period of this voltage-switching DC power supply will be described with reference to FIGS. The contactors 8A and 8B may or may not be turned on. In this regenerative braking, when the generated voltage of the inverter 30 is lower than the DC link voltage, the controller 100 instructs the inverter 30 to perform a boost chopper operation. In this step-up chopper operation, the three legs 31-33 of the inverter 30 are switched synchronously at a predetermined PWM carrier frequency. Each PWM cycle period is composed of a clamp period and an output period. In the clamp period shown in FIG. 6, the lower arm transistors of the legs 31-33 are simultaneously turned on. In the output period shown in FIG. 7, the lower arm transistors of the legs 31-33 are simultaneously turned off.
出力期間/（クランプ期間+出力期間）の比率に等しいPWMデユーティ比が低い時、インバータ３０は高い出力電圧を発生する。これにより、昇圧チョッパとして働くインバータ３０は高い充電電圧を発生することができる。PWMデユーティ比はバッテリ充電電流に基づいて制御される。インバータ３０を昇圧チョッパとして動作させるため、インバータ３０からバッテリ１及び２へ供給される充電電流のリップルレートは高くなる。しかし、このリップルレートはインダクタ７及び平滑キャパシタ２０により低減される。バッテリ１及び２の損失はこのリップルレート低減により低減される。同様に、バッテリ１及び２の損失はトラクションモータの低負荷条件において並列放電モードにおいて直列放電モードよりも低減される。これは、バッテリ温度の上昇を抑制する。特に、この効果は増加された内部抵抗をもつ古いバッテリにおいて優れている。 When the PWM duty ratio equal to the ratio of output period / (clamp period + output period) is low, inverter 30 generates a high output voltage. Thereby, the inverter 30 which functions as a step-up chopper can generate a high charging voltage. The PWM duty ratio is controlled based on the battery charging current. Since the inverter 30 is operated as a step-up chopper, the ripple rate of the charging current supplied from the inverter 30 to the batteries 1 and 2 is increased. However, this ripple rate is reduced by the inductor 7 and the smoothing capacitor 20. The loss of the batteries 1 and 2 is reduced by this ripple rate reduction. Similarly, the losses of the batteries 1 and 2 are reduced in the parallel discharge mode compared to the series discharge mode under low load conditions of the traction motor. This suppresses an increase in battery temperature. In particular, this effect is superior in older batteries with increased internal resistance.
　　　　　第２実施例
電気自動車に適用される第２実施例の電圧切替式直流電源が図８を参照して説明される。この実施例によれば、第２の接続切替回路１０A及び第３の接続切替回路１０Bが第１実施例の電圧切替式直流電源に追加されている。第１実施例のバッテリ１は、それぞれ１６０Vの定格電圧をもつ２つのバッテリブロック１A及び１Bに分割されている。同様に、第１実施例のバッテリ２は、それぞれ１６０Vの定格電圧をもつ２つのバッテリブロック２A及び２Bに分割されている。第１実施例のインダクタ７は２つのインダクタ７１及び７２に分割されている。インダクタ７１及び７２は共通のコアをもつことができる。追加された接続切替回路１０A及び１０Bはそれぞれ、接続切替回路１０と本質的に同じ回路構成をもつ。 Second Embodiment A voltage switching type DC power source according to a second embodiment applied to an electric vehicle will be described with reference to FIG. According to this embodiment, a second connection switching circuit 10A and a third connection switching circuit 10B are added to the voltage-switching DC power supply of the first embodiment. The battery 1 of the first embodiment is divided into two battery blocks 1A and 1B each having a rated voltage of 160V. Similarly, the battery 2 of the first embodiment is divided into two battery blocks 2A and 2B each having a rated voltage of 160V. The inductor 7 of the first embodiment is divided into two inductors 71 and 72. Inductors 71 and 72 can have a common core. Each of the added connection switching circuits 10A and 10B has substantially the same circuit configuration as that of the connection switching circuit 10.
接続切替回路１０Aは、充電ダイオード４Aをもつ直列トランジスタ３A、並列ダイオード５A及び６A、及びインダクタ７からなる。直列トランジスタ３Aはブロック２A及びブロック２Bを接続している。充電ダイオード４Aは直列トランジスタ３Aと逆並列に接続されている。並列ダイオード５Aはブロック２Aの負極とブロック２Bの負極とを接続している。並列ダイオード６Aはブロック２Aの正極とブロック２Bの正極とを接続している。 The connection switching circuit 10A includes a series transistor 3A having a charging diode 4A, parallel diodes 5A and 6A, and an inductor 7. The series transistor 3A connects the block 2A and the block 2B. The charging diode 4A is connected in antiparallel with the series transistor 3A. The parallel diode 5A connects the negative electrode of the block 2A and the negative electrode of the block 2B. The parallel diode 6A connects the positive electrode of the block 2A and the positive electrode of the block 2B.
接続切替回路１０Bは、充電ダイオード４Bをもつ直列トランジスタ３B、並列ダイオード５B及び６B、及びインダクタ７からなる。直列トランジスタ３Bはブロック１A及びブロック１Bを接続している。充電ダイオード４Bは直列トランジスタ３Bと逆並列に接続されている。並列ダイオード５Bはブロック１Aの負極とブロック１Bの負極とを接続している。並列ダイオード６Bはブロック１Aの正極とブロック１Bの正極とを接続している。 The connection switching circuit 10B includes a series transistor 3B having a charging diode 4B, parallel diodes 5B and 6B, and an inductor 7. The serial transistor 3B connects the block 1A and the block 1B. The charging diode 4B is connected in antiparallel with the series transistor 3B. The parallel diode 5B connects the negative electrode of the block 1A and the negative electrode of the block 1B. The parallel diode 6B connects the positive electrode of the block 1A and the positive electrode of the block 1B.
この電圧切替式直流電源の放電動作が説明される。コントローラ１００は直列放電モード、２-並列放電モード、４-並列放電モードをもつ。２-並列放電モードは本質的に実施例１の並列放電モードと同じである。３つの直列トランジスタ３、３A、及び３Bがオンされる直列放電モードにおいて、DCリンク電圧Vdは４つのブロック２A、２B、１A、及び１Bの電圧和（＝約６４０V）と等しくなる。２-並列放電モードにおいて、直列トランジスタ３がオフされ、直列トランジスタ３A及び３Bはオンされる。DCリンク電圧Vdはほぼ３２０Vとなる。４-並列放電モードにおいて、直列トランジスタ３、３A及び３Bはオフされる。DCリンク電圧Vdはほぼ１６０Vとなる。 The discharge operation of this voltage-switching DC power supply will be described. The controller 100 has a series discharge mode, a 2-parallel discharge mode, and a 4-parallel discharge mode. The 2-parallel discharge mode is essentially the same as the parallel discharge mode of the first embodiment. In the series discharge mode in which the three series transistors 3, 3A, and 3B are turned on, the DC link voltage Vd is equal to the voltage sum (= about 640 V) of the four blocks 2A, 2B, 1A, and 1B. In the 2-parallel discharge mode, the series transistor 3 is turned off and the series transistors 3A and 3B are turned on. The DC link voltage Vd is approximately 320V. In the 4-parallel discharge mode, the series transistors 3, 3A and 3B are turned off. The DC link voltage Vd is approximately 160V.
次に、２-並列放電モードから直列放電モードへのモード切替動作が説明される。このモード切替動作において、直列トランジスタ３A及び３Bはオフされている。直列トランジスタ３は所定のPWMキャリヤ周波数でスイッチングされ、そのPWMデユーティ比は０から１へ徐々に増加される。これにより、DCリンク電圧Vdは３２０Vから６４０Vへ徐々に上昇する。次に、直列放電モードから２-並列放電モードへのもう１つのモード切替動作が説明される。このモード切替動作において、直列トランジスタ３は所定のPWMキャリヤ周波数でスイッチングされ、そのPWMデユーティ比は１から０へ徐々に減少される。これにより、DCリンク電圧Vdは６４０Vから３２０Vへ徐々に低下する。結局、並列ダイオード５及び６、直列トランジスタ３、及びインダクタ７は、公知の降圧チョッパとして働く。 Next, the mode switching operation from the 2-parallel discharge mode to the series discharge mode will be described. In this mode switching operation, the series transistors 3A and 3B are turned off. The serial transistor 3 is switched at a predetermined PWM carrier frequency, and its PWM duty ratio is gradually increased from 0 to 1. As a result, the DC link voltage Vd gradually increases from 320V to 640V. Next, another mode switching operation from the series discharge mode to the 2-parallel discharge mode will be described. In this mode switching operation, the serial transistor 3 is switched at a predetermined PWM carrier frequency, and its PWM duty ratio is gradually decreased from 1 to 0. As a result, the DC link voltage Vd gradually decreases from 640V to 320V. Eventually, the parallel diodes 5 and 6, the series transistor 3, and the inductor 7 serve as a known step-down chopper.
次に、４-並列放電モードから２-並列放電モードへのもう１つのモード切替動作が説明される。このモード切替動作において、直列トランジスタ３はオフされている。直列トランジスタ３A及び３Bは所定のPWMキャリヤ周波数でスイッチングされ、そのPWMデユーティ比は０から１へ徐々に増加される。これにより、DCリンク電圧Vdは１６０Vから３２０Vへ徐々に上昇する。次に、２-並列放電モードから４-並列放電モードへのもう１つのモード切替動作が説明される。このモード切替動作において、直列トランジスタ３A及び３Bは所定のPWMキャリヤ周波数でスイッチングされ、そのPWMデユーティ比は１から０へ徐々に減少される。これにより、DCリンク電圧Vdは３２０Vから１６０Vへ徐々に低下する。結局、並列ダイオード５A及び６A、直列トランジスタ３A、及びインダクタ７は公知の降圧チョッパとして働く。同様に、並列ダイオード５B及び６B、直列トランジスタ３B、及びインダクタ７も公知の降圧チョッパとして働く。 Next, another mode switching operation from the 4-parallel discharge mode to the 2-parallel discharge mode will be described. In this mode switching operation, the series transistor 3 is turned off. The series transistors 3A and 3B are switched at a predetermined PWM carrier frequency, and the PWM duty ratio is gradually increased from 0 to 1. As a result, the DC link voltage Vd gradually increases from 160V to 320V. Next, another mode switching operation from the 2-parallel discharge mode to the 4-parallel discharge mode will be described. In this mode switching operation, the series transistors 3A and 3B are switched at a predetermined PWM carrier frequency, and the PWM duty ratio is gradually reduced from 1 to 0. As a result, the DC link voltage Vd gradually decreases from 320V to 160V. Eventually, the parallel diodes 5A and 6A, the series transistor 3A, and the inductor 7 function as a known step-down chopper. Similarly, parallel diodes 5B and 6B, series transistor 3B, and inductor 7 also function as a known step-down chopper.
図９はモータトルクが最大値である場合におけるモータ回転数とDCリンク電圧Vdとの関係を示す図である。DCリンク電圧Vdは、たとえば４０km/h未満の低速領域において１６０Vとなり、たとえば４０-８０km/hの範囲の中速領域において３２０Vとなり、８０km/hを超える高速領域において６４０Vとなる。 FIG. 9 is a diagram showing the relationship between the motor rotation speed and the DC link voltage Vd when the motor torque is the maximum value. The DC link voltage Vd is, for example, 160 V in a low speed region less than 40 km / h, for example, 320 V in a medium speed region in the range of 40-80 km / h, and 640 V in a high speed region exceeding 80 km / h.
次に、外部直流電源による４つのブロック１A、１B、２A、及び２Bの充電動作が図１０を参照して説明される。直列トランジスタ３A及び３Bはオンされ、直列トランジスタ３はオフされる。これにより、直列接続されたブロック１A及び１Bは実質的にバッテリ１となる。同様に、直列接続されたブロック２A及び２Bは実質的にバッテリ２となる。バッテリ１及び２を並列に充電するために、この外部直流電源は約３２０Vの電圧をバッテリ１及び２に印加する。この並列充電のために、接続切替回路１０は第１実施例と同様にリレーボックス８及びコネクタ９をもつ。 Next, the charging operation of the four blocks 1A, 1B, 2A, and 2B by the external DC power supply will be described with reference to FIG. Series transistors 3A and 3B are turned on and series transistor 3 is turned off. Thereby, the blocks 1A and 1B connected in series substantially become the battery 1. Similarly, the blocks 2A and 2B connected in series become the battery 2 substantially. In order to charge the batteries 1 and 2 in parallel, the external DC power supply applies a voltage of about 320V to the batteries 1 and 2. For this parallel charging, the connection switching circuit 10 has a relay box 8 and a connector 9 as in the first embodiment.
コンタクタ８１及び８２は、２-並列放電モード及び回生制動モードにおいてオンされることができる。これにより、コンタクタ８１は並列ダイオード５と並列接続され、コンタクタ８２はダイオード６と並列接続される。DCリンク電圧Vdは、２-並列放電モード及び回生制動モードにおいて３２０Vとなる。この実施例における回生制動は、本質的に第１実施例と同じであるため、説明は省略される。 The contactors 81 and 82 can be turned on in the 2-parallel discharge mode and the regenerative braking mode. Thereby, the contactor 81 is connected in parallel with the parallel diode 5, and the contactor 82 is connected in parallel with the diode 6. The DC link voltage Vd is 320 V in the 2-parallel discharge mode and the regenerative braking mode. Since the regenerative braking in this embodiment is essentially the same as in the first embodiment, description thereof is omitted.
この実施例によれば、コンタクタ８１が並列ダイオード５と並列接続され、コンタクタ８２が並列ダイオード６と並列接続されている。したがって、２-並列放電モードにおいて、コンタクタ８１及び８２がオンされる前に、バッテリ１及び２はダイオード５及び６により既に２-並列放電モードとなっている。したがって、コンタクタ８１及び８２がオンされる時、バッテリ１及び２の間の電圧差は十分に低減されることができる。これは、コンタクタ８１及び８２がオンされる時、バッテリ１及び２の間に流れる短絡電流がほぼゼロとなることを意味する。さらに、コンタクタ８１及びコンタクタ８２がオフされる時、コンタクタ８１及び８２のアーク放電はダイオード５及び６により防止される。これにより、コンタクタ８１及び８２の寿命及び信頼性が改善される。 According to this embodiment, the contactor 81 is connected in parallel with the parallel diode 5, and the contactor 82 is connected in parallel with the parallel diode 6. Therefore, in the 2-parallel discharge mode, the batteries 1 and 2 are already in the 2-parallel discharge mode by the diodes 5 and 6 before the contactors 81 and 82 are turned on. Therefore, when the contactors 81 and 82 are turned on, the voltage difference between the batteries 1 and 2 can be sufficiently reduced. This means that when the contactors 81 and 82 are turned on, the short-circuit current flowing between the batteries 1 and 2 becomes substantially zero. Furthermore, when the contactor 81 and the contactor 82 are turned off, the arc discharge of the contactors 81 and 82 is prevented by the diodes 5 and 6. Thereby, the lifetime and reliability of the contactors 81 and 82 are improved.
次に、ブロック１A、１B、２A、及び２Bのうちの１つの電圧が許容範囲から離れている場合を意味するブロック不良状態における放電モードが説明される。４-並列放電モードが実行される時、不良ブロックを除く３つのブロックが放電電流を供給する。言い換えれば、７５％のバッテリセルは放電を継続することができる。２-並列放電モードにおいて、バッテリ１及び２のうち、不良ブロックを含まない方が放電電流を供給する。言い換えれば、５０％のバッテリセルは放電を継続することができる。 Next, the discharge mode in the block failure state which means the case where the voltage of one of the blocks 1A, 1B, 2A, and 2B is out of the allowable range will be described. 4-When the parallel discharge mode is executed, three blocks except the defective block supply the discharge current. In other words, 75% of the battery cells can continue to discharge. In the 2-parallel discharge mode, one of the batteries 1 and 2 that does not include a defective block supplies a discharge current. In other words, 50% of the battery cells can continue to discharge.
次に、ブロック不良状態における充電モードが説明される。直列トランジスタ３がオフされ、２-並列充電モードが採用される。さらに、直列トランジスタ３A、３Bのうち不良ブロックに接続される方がオフされる。これにより、５０％のバッテリセルは充電を継続することができる。結局、３つの直列トランジスタを使用する第２実施例によれば、３段階のDCリンク電圧Vdを選択できるとともに、不良バッテリブロックを切り離すことができる。 Next, the charging mode in the block failure state will be described. The serial transistor 3 is turned off and the 2-parallel charging mode is employed. Further, the serial transistor 3A, 3B connected to the defective block is turned off. Thereby, 50% of battery cells can continue charging. Eventually, according to the second embodiment using three series transistors, it is possible to select a three-stage DC link voltage Vd and to disconnect a defective battery block.
この実施例の利点が説明される。この実施例によれば、低速領域において採用される４-並列放電モードは、バッテリ損失を大幅に低減する。このため、バッテリ温度の上昇を抑制することができ、特に高温環境下においてバッテリ寿命を延長することができる。この効果は特に高い内部抵抗をもつ古いバッテリにおいて顕著である。 The advantages of this embodiment are described. According to this embodiment, the 4-parallel discharge mode employed in the low speed region significantly reduces battery loss. For this reason, the rise in battery temperature can be suppressed, and the battery life can be extended particularly in a high temperature environment. This effect is particularly noticeable in old batteries with high internal resistance.
第１実施例及び第２実施例の変形態様が説明される。この変形態様によれば、図５又は図８に示される充電ダイオード４が省略される。コンタクタ８A及び８Bは回生制動期間においてオンされ、インバータ３０はコンタクタ８A及び８Bを通じてバッテリ１及び２を並列充電する。コンタクタ８A及び８Bの代わりにパワートランジスタを採用することができる。図１１はもう一つの変形態様を示す。この変形態様において、図８に示される直列トランジスタ３、３A、及び３Bはそれぞれ、双方向性の絶縁ゲートトランジスタからなる。この絶縁ゲートトランジスタはボディダイオードをもつことができる。 Variations of the first and second embodiments will be described. According to this modification, the charging diode 4 shown in FIG. 5 or FIG. 8 is omitted. The contactors 8A and 8B are turned on during the regenerative braking period, and the inverter 30 charges the batteries 1 and 2 in parallel through the contactors 8A and 8B. A power transistor can be employed in place of the contactors 8A and 8B. FIG. 11 shows another variation. In this variation, the series transistors 3, 3A, and 3B shown in FIG. 8 each comprise a bidirectional insulated gate transistor. The insulated gate transistor can have a body diode.
　　　　　変形態様
この変形態様によれば、図８に示される４つのブロック１A、１B、２A、及び２Bのうちの３つのブロックがインバータ３０にDCリンク電圧Vdを出力する３-直列放電モードが採用される。この３-直列放電モードは第１モード及び第２モードからなる。第１モード及び第２モードは所定のインタバルで交互に実行されることが好適である。 Modified Mode According to this modified mode, a 3-series discharge mode in which three of the four blocks 1A, 1B, 2A, and 2B shown in FIG. 8 output the DC link voltage Vd to the inverter 30 is employed. The The 3-series discharge mode includes a first mode and a second mode. The first mode and the second mode are preferably executed alternately at a predetermined interval.
第１モードにおいて、直列トランジスタ３及び３Aがオンされ、直列トランジスタ３Bはオフされる。これにより、フロック２A、２B、及び１Aの電圧和がインバータ３０に印加される。第２モードにおいて、直列トランジスタ３及び３Bがオンされ、直列トランジスタ３Aはオフされる。これにより、ブロック２B、１A、及び１Bの電圧和がインバータ３０に印加される。４つのブロックはそれぞれ約１６０Vをもつ。したがって、４８０Vの電圧がインバータ３０に印加される。 In the first mode, the series transistors 3 and 3A are turned on, and the series transistor 3B is turned off. Thus, the voltage sum of the flocks 2A, 2B, and 1A is applied to the inverter 30. In the second mode, the series transistors 3 and 3B are turned on, and the series transistor 3A is turned off. As a result, the voltage sum of the blocks 2B, 1A, and 1B is applied to the inverter 30. Each of the four blocks has about 160V. Therefore, a voltage of 480 V is applied to the inverter 30.
この３-直列放電モードにおいて、直列トランジスタ３はオンされる。直列トランジスタ３A及び３Bは所定インタバルで相補的にスイッチングされる。 In this 3-series discharge mode, the series transistor 3 is turned on. Series transistors 3A and 3B are complementarily switched at a predetermined interval.
けれども、この３-直列放電モードによれば、ブロック２B及び１Aがブロック２A及び１Bよりもより速く放電する。しかし、２-並列放電モード及び４-並列放電モードが実行される時、ブロック２A及び１Bがブロック２B及び１Aよりも速く放電する。これにより、ブロック２B及び１Aとブロック２A及び１Bとの間の電圧差が低減される。結局、電圧切替式直流電源は、１６０V、３２０V、及び６４０Vに加えて、４８０Vを出力することができる。 However, according to this 3-series discharge mode, blocks 2B and 1A discharge faster than blocks 2A and 1B. However, when the 2-parallel discharge mode and the 4-parallel discharge mode are executed, the blocks 2A and 1B discharge faster than the blocks 2B and 1A. Thereby, the voltage difference between the blocks 2B and 1A and the blocks 2A and 1B is reduced. Eventually, the voltage-switching DC power supply can output 480V in addition to 160V, 320V, and 640V.
　　　　　第３実施例
第３実施例の電圧切替式直流電源が図１２を参照して説明される。この直流電源はマイルドハイブリッド車（MHV)により採用される。この直流電源は、直列トランジスタ３、並列トランジスタ５０及び６０、及びインダクタ７からなる接続切替回路１０Cを有する。この直流電源は図３に示される第１実施例の直流電源と比べて次の差異をもつ。並列トランジスタ５０及び６０が並列ダイオード５及び６の代わりに採用される。直列トランジスタ３並びに並列トランジスタ５０及び６０はそれぞれ、逆並列ダイオードをもつMOSトランジスタからなる。キャパシタ２がバッテリ２の代わりに採用される。バッテリ１はバッテリ電圧Vbをもち、キャパシタ２はキャパシタ電圧Vcをもつ。重要な差異は、インダクタ７は、キャパシタ２と並列トランジスタ６０との間に配置されることである。 Third Embodiment A voltage-switching DC power source according to a third embodiment will be described with reference to FIG. This DC power supply is adopted by mild hybrid vehicles (MHV). The DC power supply includes a connection switching circuit 10C including a series transistor 3, parallel transistors 50 and 60, and an inductor 7. This DC power supply has the following differences compared to the DC power supply of the first embodiment shown in FIG. Parallel transistors 50 and 60 are employed in place of the parallel diodes 5 and 6. Each of the serial transistor 3 and the parallel transistors 50 and 60 is a MOS transistor having an antiparallel diode. A capacitor 2 is employed instead of the battery 2. The battery 1 has a battery voltage Vb, and the capacitor 2 has a capacitor voltage Vc. The important difference is that the inductor 7 is arranged between the capacitor 2 and the parallel transistor 60.
図１３はこの直流電源の等価回路を示す。キャパシタ２の内部抵抗rcはバッテリ１の内部抵抗rbよりも低い。バッテリ１は定格電圧４８Vのリチウムイオンバッテリ又は定格電圧１４．４Vの鉛酸バッテリからなる。キャパシタ２はリチウムイオンキャパシタからなる。ステータコイル４０をもつこのモータは内燃エンジンのクランクシャフトにより駆動されるプーリーに結合されている。コントローラ１００により制御されるこの直流電源の動作が以下に説明される。 FIG. 13 shows an equivalent circuit of this DC power supply. The internal resistance rc of the capacitor 2 is lower than the internal resistance rb of the battery 1. The battery 1 is composed of a lithium ion battery having a rated voltage of 48V or a lead acid battery having a rated voltage of 14.4V. The capacitor 2 is a lithium ion capacitor. This motor with stator coil 40 is coupled to a pulley driven by the crankshaft of the internal combustion engine. The operation of this DC power source controlled by the controller 100 will be described below.
　　　　　予備充電モード　　　　　
最初に、予備充電モードが図１４及び図１５を参照して説明される。この予備充電モードはキャパシタ電圧Vcがバッテリ電圧Vbよりも低い時に実施される。この予備充電モードにおいて、直列トランジスタ３はオフされ、並列トランジスタ６０はオンされる。並列トランジスタ５０は所定のPWMキャリヤ周波数でスイッチングされる。並列トランジスタ５０のPWMデユーティ比は０から１へ徐々に増加される。これにより、直列トランジスタ３、並列トランジスタ５０及び６０、及びインダクタ７からなる接続切替回路１０Cは降圧チョッパとして動作する。各PWMサイクル周期は電流供給期間とフリーホィーリング期間とからなる。図１４は並列トランジスタ５０がオンされるこの電流供給期間を示す。バッテリ１の放電電流が並列トランジスタ６０、インダクタ７、キャパシタ２、及び並列トランジスタ５０を通じて流れる。これにより、キャパシタ電圧Vcは徐々にバッテリ電圧Vbに接近する。 Pre-charging mode
First, the precharge mode will be described with reference to FIGS. This precharge mode is performed when the capacitor voltage Vc is lower than the battery voltage Vb. In this precharge mode, the series transistor 3 is turned off and the parallel transistor 60 is turned on. The parallel transistor 50 is switched at a predetermined PWM carrier frequency. The PWM duty ratio of the parallel transistor 50 is gradually increased from 0 to 1. Accordingly, the connection switching circuit 10C including the series transistor 3, the parallel transistors 50 and 60, and the inductor 7 operates as a step-down chopper. Each PWM cycle period consists of a current supply period and a freewheeling period. FIG. 14 shows this current supply period during which the parallel transistor 50 is turned on. The discharge current of the battery 1 flows through the parallel transistor 60, the inductor 7, the capacitor 2, and the parallel transistor 50. Thereby, the capacitor voltage Vc gradually approaches the battery voltage Vb.
図１５は並列トランジスタ５０がオフされるフリーホィーリング期間を示す。インダクタ７に蓄積された磁気エネルギーは、直列トランジスタ３、並列トランジスタ６０、及びインダクタ７を通じてフリーホィーリング電流を流す。フリーホィーリング電流は直列トランジスタ３の逆並列ダイオードを通じて流れる。キャパシタ電圧Vcがバッテリ電圧Vbとほぼ等しくなる時、この予備充電モードは終了する。 FIG. 15 shows a freewheeling period in which the parallel transistor 50 is turned off. The magnetic energy accumulated in the inductor 7 causes a freewheeling current to flow through the series transistor 3, the parallel transistor 60, and the inductor 7. The freewheeling current flows through the antiparallel diode of the series transistor 3. When the capacitor voltage Vc becomes substantially equal to the battery voltage Vb, the precharge mode ends.
　　　　　エンジン始動モード　　　　　
次に、エンジン始動モードが図１６を参照して説明される。このエンジン始動モードによれば、並列トランジスタ５０及び６０はオフされ、直列トランジスタ３はオンされる。インバータ３０に印加されるDCリンク電圧Vdcはキャパシタ電圧Vc及びバッテリ電圧Vbの和にほぼ等しい。エンジン始動後、直列トランジスタ３はオフされる。直列トランジスタ３はオフされる時、インダクタ７に蓄積された磁気エネルギーは並列トランジスタ５０を通じてフリーホィーリング電流を流す。インダクタ７はこのエンジン始動モードにおいて磁気飽和する。 Engine start mode
Next, the engine start mode will be described with reference to FIG. According to this engine start mode, the parallel transistors 50 and 60 are turned off and the series transistor 3 is turned on. The DC link voltage Vdc applied to the inverter 30 is approximately equal to the sum of the capacitor voltage Vc and the battery voltage Vb. After starting the engine, the series transistor 3 is turned off. When the series transistor 3 is turned off, the magnetic energy stored in the inductor 7 causes a freewheeling current to flow through the parallel transistor 50. The inductor 7 is magnetically saturated in this engine start mode.
　　　　　キャパシタ回復モード
次に、キャパシタ回復モードが図１７を参照して説明される。エンジン回転数が所定値より高くなった時、モータは発電機として駆動される。３相整流器としてのインバータ３０はステータコイル４０に誘導された３相電圧を整流する。直列トランジスタ３及び並列トランジスタ６０はオフされ、並列トランジスタ５０がオンされる。これにより、キャパシタ２が充電される。インダクタ７はこのキャパシタ回復モードにおいて磁気飽和する。 Capacitor Recovery Mode Next, the capacitor recovery mode will be described with reference to FIG. When the engine speed becomes higher than a predetermined value, the motor is driven as a generator. The inverter 30 as a three-phase rectifier rectifies the three-phase voltage induced in the stator coil 40. The serial transistor 3 and the parallel transistor 60 are turned off, and the parallel transistor 50 is turned on. Thereby, the capacitor 2 is charged. The inductor 7 is magnetically saturated in this capacitor recovery mode.
　　　　　発電モード
次に、発電モードが図１７を参照して説明される。キャパシタ電圧Vcがバッテリ電圧Vbとほぼ等しくなる時、並列トランジスタ６０がオンされる。並列トランジスタ５０はオンされ、直列トランジスタ３はオフされている。これにより、インバータ３０は、並列接続されたキャパシタ２及びバッテリ１を充電する。インバータ３０はバッテリ１に接続された電気負荷に給電する。 Power Generation Mode Next, the power generation mode will be described with reference to FIG. When the capacitor voltage Vc becomes substantially equal to the battery voltage Vb, the parallel transistor 60 is turned on. The parallel transistor 50 is turned on and the series transistor 3 is turned off. Thereby, the inverter 30 charges the capacitor 2 and the battery 1 connected in parallel. The inverter 30 supplies power to the electric load connected to the battery 1.
　　　　　トルクアシストモード　　　　　
次に、低レベルタイプのトルクアシストモードが図１８を参照して説明される。直列トランジスタ３はオフされ、並列トランジスタ５０及び６０はオンされる。コントローラ１００はインバータ３０を制御し、モータ電流が、並列接続されたバッテリ１及びキャパシタ２からインバータ３０を通じてステータコイル４０に供給される。 Torque assist mode
Next, a low level type torque assist mode will be described with reference to FIG. Series transistor 3 is turned off and parallel transistors 50 and 60 are turned on. The controller 100 controls the inverter 30, and motor current is supplied to the stator coil 40 through the inverter 30 from the battery 1 and the capacitor 2 connected in parallel.
次に、高レベルタイプのトルクアシストモードが図１９を参照して説明される。強力なトルクアシストが要求される時、並列トランジスタ５０及び６０はオフされ、直列トランジスタ３がオンされる。平滑キャパシタ２０への突入電流を低減するために、直列トランジスタ３を所定のPWMキャリヤ周波数でスイッチングすることも可能である。直列トランジスタ３のPWMデユーティ比は０から１に次第に増加される。その結果、キャパシタ電圧Vcとバッテリ電圧Vbとの和に等しいDCリンク電圧Vdcがインバータ３０に印加される。 Next, a high level type torque assist mode will be described with reference to FIG. When strong torque assist is required, parallel transistors 50 and 60 are turned off and series transistor 3 is turned on. In order to reduce the inrush current to the smoothing capacitor 20, the series transistor 3 can be switched at a predetermined PWM carrier frequency. The PWM duty ratio of the series transistor 3 is gradually increased from 0 to 1. As a result, a DC link voltage Vdc equal to the sum of the capacitor voltage Vc and the battery voltage Vb is applied to the inverter 30.
　　　　　標準回生制動モード
次に、標準回生制動モードが図２０を参照して説明される。この並列回生制動モードは既述された発電モードと本質的に同じである。直列トランジスタ３はオフされ、並列トランジスタ５０及び６０がオンされる。インバータ３０は、並列接続されたキャパシタ２及びバッテリ１を充電する。 Standard Regenerative Braking Mode Next, the standard regenerative braking mode will be described with reference to FIG. This parallel regenerative braking mode is essentially the same as the power generation mode described above. The serial transistor 3 is turned off and the parallel transistors 50 and 60 are turned on. Inverter 30 charges capacitor 2 and battery 1 connected in parallel.
　　　　　キャパシタ充電式回生制動モード
次に、キャパシタ充電式回生制動モードが図２１を参照して説明される。このモードは、たとえばバッテリ１が高いSOC（充電状態）値をもつ時に採用される。直列トランジスタ３及び並列トランジスタ６０はオフされ、並列トランジスタ５０だけがオンされる。これにより、キャパシタ２だけがインバータ３０の発電電流を吸収する。キャパシタ電圧Vcはバッテリ電圧Vbより高くなることができる。キャパシタ電圧Vcが所定の高電圧値に達した時このキャパシタ回生制動モードは停止される。これにより、バッテリ１の状態にかかわらず、制動エネルギーは良好に回収される。 Capacitor Rechargeable Regenerative Braking Mode Next, the capacitor rechargeable regenerative braking mode will be described with reference to FIG. This mode is employed, for example, when the battery 1 has a high SOC (charge state) value. The serial transistor 3 and the parallel transistor 60 are turned off, and only the parallel transistor 50 is turned on. Thereby, only the capacitor 2 absorbs the generated current of the inverter 30. The capacitor voltage Vc can be higher than the battery voltage Vb. When the capacitor voltage Vc reaches a predetermined high voltage value, the capacitor regenerative braking mode is stopped. Thereby, the braking energy is recovered satisfactorily regardless of the state of the battery 1.
キャパシタ電圧Vcがバッテリ電圧Vbよりも高い時、キャパシタ２はインバータ３０又はバッテリ１に放電することが好適である。キャパシタ２からバッテリ１への放電動作において、直列トランジスタ３はオフされ、並列トランジスタ５０はオンされる。並列トランジスタ６０は所定のPWMキャリヤ周波数でスイッチングされる。並列トランジスタ６０のPWMデユーティ比は０から１に徐々に増加される。これにより、キャパシタ電圧Vcはほぼバッテリ電圧Vbと等しくなる。 When the capacitor voltage Vc is higher than the battery voltage Vb, the capacitor 2 is preferably discharged to the inverter 30 or the battery 1. In the discharging operation from the capacitor 2 to the battery 1, the series transistor 3 is turned off and the parallel transistor 50 is turned on. The parallel transistor 60 is switched at a predetermined PWM carrier frequency. The PWM duty ratio of the parallel transistor 60 is gradually increased from 0 to 1. Thereby, the capacitor voltage Vc becomes substantially equal to the battery voltage Vb.
　　　　　昇圧チョッパモード
回生制動モードにおいて、３相整流器としてのインバータ３０の整流電圧がDCリンク電圧Vdcを下回る時、インバータ３０は昇圧チョッパモードを実行することができる。たとえばモータ回転速度の低下によりインバータ３０の整流電圧が低下する時、この昇圧チョッパモードが実行される。この昇圧チョッパモードによれば、図１２に示される３つの上アームトランジスタ３４-３６又は３つの下アームトランジスタ３７-３９のどちらかが所定のPWMデユーティ比で同期的にスイッチングされる。上アームトランジスタ３４-３６及び下アームトランジスタ３７-３９は相補的にスイッチングされる。 In the boost chopper mode regenerative braking mode, when the rectified voltage of the inverter 30 as the three-phase rectifier is lower than the DC link voltage Vdc, the inverter 30 can execute the boost chopper mode. For example, when the rectified voltage of the inverter 30 decreases due to a decrease in the motor rotation speed, the boost chopper mode is executed. According to this step-up chopper mode, one of the three upper arm transistors 34-36 or the three lower arm transistors 37-39 shown in FIG. 12 is switched synchronously at a predetermined PWM duty ratio. Upper arm transistors 34-36 and lower arm transistors 37-39 are complementarily switched.
３つの下アームトランジスタ３７-３９がオンされる時、磁気エネルギーが、短絡されたステータコイル４０の３つの相コイル４１-４３に蓄積される。次に、３つの下アームトランジスタ３７-３９がオフされる時、インバータ３０は高い整流電圧を出力する。これにより、たとえモータ回転数が低下しても、回生制動モードを実行することができる。 When the three lower arm transistors 37-39 are turned on, magnetic energy is stored in the three phase coils 41-43 of the shorted stator coil 40. Next, when the three lower arm transistors 37-39 are turned off, the inverter 30 outputs a high rectified voltage. Thereby, even if the motor rotation speed decreases, the regenerative braking mode can be executed.
キャパシタ２を放電する並列トランジスタ５０がオフされる時、フリーホィーリング電流は並列トランジスタ５０の逆並列ダイオードを通じて流れる。キャパシタ２を充電する並列トランジスタ５０がオフされる時、もう１つのフリーホィーリング電流は直列トランジスタ３の逆並列ダイオードを通じて流れる。同様に、キャパシタ２を放電する直列トランジスタ３がオフされる時、フリーホィーリング電流は並列トランジスタ５０の逆並列ダイオードを通じて流れる。結局。接続切替回路１０cはインダクタ７のフリーホィーリング電流を循環することができる。 When the parallel transistor 50 that discharges the capacitor 2 is turned off, freewheeling current flows through the anti-parallel diode of the parallel transistor 50. When the parallel transistor 50 that charges the capacitor 2 is turned off, another freewheeling current flows through the anti-parallel diode of the series transistor 3. Similarly, freewheeling current flows through the anti-parallel diode of the parallel transistor 50 when the series transistor 3 discharging the capacitor 2 is turned off. After all. The connection switching circuit 10c can circulate the freewheeling current of the inductor 7.
第３実施例の効果が説明される。この電圧切替式直流電源によれば、高価なバッテリの追加を回避することができる。短時間だけ使用されるキャパシタ２はバッテリ１と比べて低い容量値をもつことができる。低い内部抵抗及び長い寿命をもつキャパシタ２は、エンジン始動、トルクアシスト、及び回生制動の電力損失を低減する。キャパシタ２はバッテリ２の寿命が延長する。 The effect of the third embodiment will be described. According to this voltage-switching DC power supply, it is possible to avoid the addition of an expensive battery. The capacitor 2 used only for a short time can have a lower capacitance value than the battery 1. Capacitor 2 with low internal resistance and long life reduces power loss for engine start, torque assist, and regenerative braking. Capacitor 2 extends the life of battery 2.
接続切替回路１０cは電圧切替機能と降圧チョッパ機能の両方をもつ。この降圧チョッパ動作において、キャパシタ電圧Vc及びバッテリ電圧Vbは均等化される。１つの変形態様において、エンジン始動、回生制動、及びトルクアシストは、キャパシタ２だけにより実行されることができる。これにより、バッテリ１の寿命はさらに延長される。 The connection switching circuit 10c has both a voltage switching function and a step-down chopper function. In this step-down chopper operation, the capacitor voltage Vc and the battery voltage Vb are equalized. In one variant, engine start-up, regenerative braking, and torque assist can be performed by the capacitor 2 alone. Thereby, the lifetime of the battery 1 is further extended.
　　　　　第４実施例
電気自動車（EV）又はハイブリッド車（HV)に適用される第４実施例の電圧切替式直流電源が図２２を参照して説明される。この直流電源は、直列トランジスタ３、並列トランジスタ５０及び６０、及びインダクタ７からなる接続切替回路１０dを有する。さらに、接続切替回路１０dはバイパスダイオード７３をもつ。バッテリ１の電圧V1はバッテリ２の電圧V２にほぼ等しい。この直流電源の主な特徴は、図１２に示される第３実施例のインダクタ７が２つのサブインダクタ７１及び７２に分割されている点にある。 Fourth Embodiment A voltage-switching DC power source according to a fourth embodiment applied to an electric vehicle (EV) or a hybrid vehicle (HV) will be described with reference to FIG. This DC power supply includes a connection switching circuit 10 d including a series transistor 3, parallel transistors 50 and 60, and an inductor 7. Further, the connection switching circuit 10 d has a bypass diode 73. The voltage V1 of the battery 1 is approximately equal to the voltage V2 of the battery 2. The main feature of this DC power supply is that the inductor 7 of the third embodiment shown in FIG. 12 is divided into two sub-inductors 71 and 72.
２つのサブインダクタ７１及び７２はインバータ３０及び平滑キャパシタ２０に接続される一端をもつ。２つのサブコイル７１及び７２はそれぞれ、バイパスダイオード７３により接続される他端をもつ。バイパスダイオード７３のアノード電極はサブインダクタ７２に接続され、そのカソード電極はサブインダクタ７１に接続されている。図２２に示されるバッテリ２の代わりに図１２に示されるキャパシタ２を採用することも可能である。サブインダクタ７１はバッテリ２の正極に接続され、サブインダクタ７２は並列トランジスタ６０に接続されている。サブインダクタ７１及び７２の各コイルは共通の磁気コアに巻かれることができる。 The two sub-inductors 71 and 72 have one ends connected to the inverter 30 and the smoothing capacitor 20. Each of the two subcoils 71 and 72 has the other end connected by a bypass diode 73. The anode electrode of the bypass diode 73 is connected to the sub-inductor 72, and the cathode electrode thereof is connected to the sub-inductor 71. It is also possible to employ the capacitor 2 shown in FIG. 12 instead of the battery 2 shown in FIG. The sub-inductor 71 is connected to the positive electrode of the battery 2, and the sub-inductor 72 is connected to the parallel transistor 60. Each coil of the sub-inductors 71 and 72 can be wound around a common magnetic core.
ヒユーズ８１が並列トランジスタ５０と直列に接続され、ヒユーズ８２が並列トランジスタ６０と直列に接続されている。第３のヒューズを直列トランジスタ３と直列に接続することも可能である。さらに、平滑キャパシタ２０は、並列に接続されたキャパシタ２１及び２２からなる。キャパシタ２１は高容量値をもつリチウムイオンキャパシタからなり、キャパシタ２２は優れた高周波特性をもつフィルムキャパシタからなる。接続切替回路１０dをもつ直流電源は、システムリレー９１及び９２を通じて平滑キャパシタ２０及びインバータ３０に接続されている。 A fuse 81 is connected in series with the parallel transistor 50, and a fuse 82 is connected in series with the parallel transistor 60. It is also possible to connect a third fuse in series with the series transistor 3. Further, the smoothing capacitor 20 includes capacitors 21 and 22 connected in parallel. The capacitor 21 is made of a lithium ion capacitor having a high capacitance value, and the capacitor 22 is made of a film capacitor having excellent high frequency characteristics. The DC power source having the connection switching circuit 10 d is connected to the smoothing capacitor 20 and the inverter 30 through system relays 91 and 92.
接続切替回路１０dは並列モード及び直列モードをもつ。並列モードが図２２を参照して説明される。直列トランジスタ３がオフされ、並列トランジスタ５０及び６０がオンされる。これにより、バッテリ１及び２は並列に接続される。直列モードが図２３を参照して説明される。並列トランジスタ５０及び６０がオフされ、直列トランジスタ３がオンされる。これにより、バッテリ１及び２は直列に接続され、電圧和（V１＋V２）に等しいDCリンク電圧Vdcがインバータ３０に印加される。 The connection switching circuit 10d has a parallel mode and a series mode. The parallel mode is described with reference to FIG. The serial transistor 3 is turned off and the parallel transistors 50 and 60 are turned on. Thereby, the batteries 1 and 2 are connected in parallel. The serial mode is described with reference to FIG. The parallel transistors 50 and 60 are turned off, and the series transistor 3 is turned on. Thereby, the batteries 1 and 2 are connected in series, and the DC link voltage Vdc equal to the voltage sum (V1 + V2) is applied to the inverter 30.
次に、並列モードと直列モードとの間の切替を徐々に実行する電圧切替モードが説明される。この電圧切替モードにおいて、直列トランジスタ３及び並列トランジスタ５０及び６０は所定のPWMキャリヤ周波数でスイッチングされる。この動作は降圧チョッパ動作と呼ばれる。 Next, a voltage switching mode in which switching between the parallel mode and the series mode is gradually performed will be described. In this voltage switching mode, the series transistor 3 and the parallel transistors 50 and 60 are switched at a predetermined PWM carrier frequency. This operation is called a step-down chopper operation.
並列モードから直列モードへの電圧切替モードにおいて、直列トランジスタ３のPWMデユーティ比は０から１に徐々に変化し、並列トランジスタ５０及び６０のPWMデユーティ比は１から０に徐々に変化する。直列モードから並列モードの電圧切替モードにおいて、直列トランジスタ３のPWMデユーティ比は１から０に徐々に変化し、並列トランジスタ５０及び６０のPWMデユーティ比は０から１に徐々に変化する。並列トランジスタ５０及び６０は直列トランジスタ３と比べて相補的に駆動される。 In the voltage switching mode from the parallel mode to the series mode, the PWM duty ratio of the serial transistor 3 gradually changes from 0 to 1, and the PWM duty ratio of the parallel transistors 50 and 60 gradually changes from 1 to 0. In the voltage switching mode from the serial mode to the parallel mode, the PWM duty ratio of the serial transistor 3 gradually changes from 1 to 0, and the PWM duty ratio of the parallel transistors 50 and 60 gradually changes from 0 to 1. The parallel transistors 50 and 60 are driven in a complementary manner as compared with the serial transistor 3.
この電圧切替モードにおいて、各PWMサイクル周期はそれぞれ、直列期間と並列期間からなる。直列期間において、直列トランジスタ３がオンされ、並列トランジスタ５０及び６０はオフされる。並列期間において、直列トランジスタ３がオフされ、並列トランジスタ５０及び６０はオンされる。直列期間は直列放電期間及び直列充電期間からなる。並列期間は並列放電期間及び並列充電期間からなる。 In this voltage switching mode, each PWM cycle period is composed of a series period and a parallel period. In the series period, the series transistor 3 is turned on and the parallel transistors 50 and 60 are turned off. In the parallel period, the series transistor 3 is turned off and the parallel transistors 50 and 60 are turned on. The series period includes a series discharge period and a series charge period. The parallel period includes a parallel discharge period and a parallel charge period.
バッテリ１及び２が放電される直列放電期間が図２３を参照して説明される。サブインダクタ７１は磁気エネルギーを蓄積する。並列放電期間が図２４を参照して説明される。サブインダクタ７１に蓄積された磁気エネルギーは、並列トランジスタ５０を通じて流れるフリーホィーリング電流により消費される。並列放電期間において、並列トランジスタ６０がオンされ、サブインダクタ７２は磁気エネルギーを蓄積する。直列放電期間において、並列トランジスタ６０がオフされ、サブインダクタ７２に蓄積された磁気エネルギーは、並列トランジスタ６０を通じて流れるフリーホィーリング電流により消費される。 A series discharge period in which the batteries 1 and 2 are discharged will be described with reference to FIG. The sub-inductor 71 stores magnetic energy. The parallel discharge period will be described with reference to FIG. The magnetic energy stored in the sub-inductor 71 is consumed by the freewheeling current that flows through the parallel transistor 50. In the parallel discharge period, the parallel transistor 60 is turned on, and the sub-inductor 72 stores magnetic energy. In the series discharge period, the parallel transistor 60 is turned off, and the magnetic energy stored in the sub-inductor 72 is consumed by the freewheeling current flowing through the parallel transistor 60.
バッテリ１及び２が充電される直列充電期間が図２３を参照して説明される。サブインダクタ７１は磁気エネルギーを蓄積する。直列トランジスタ３がオフされる時、フリーホィーリング電流が直列トランジスタ３の逆並列ダイオードを通じて循環する。並列充電期間が図２５を参照して説明される。並列トランジスタ６０がオフされる時、サブインダクタ７１及び７２は磁気エネルギーを蓄積する。並列トランジスタ５０がオフされる時、フリーホィーリング電流は、サブインダクタ７１及び直列トランジスタ３の逆並列ダイオードを通じて循環する。並列トランジスタ６０がオフされる時、フリーホィーリング電流は、サブインダクタ７２及びバイパスダイオード７３を通じて循環する。 A series charging period in which the batteries 1 and 2 are charged will be described with reference to FIG. The sub-inductor 71 stores magnetic energy. When the series transistor 3 is turned off, a freewheeling current circulates through the antiparallel diode of the series transistor 3. The parallel charging period will be described with reference to FIG. When the parallel transistor 60 is turned off, the sub-inductors 71 and 72 store magnetic energy. When the parallel transistor 50 is turned off, the freewheeling current circulates through the sub-inductor 71 and the antiparallel diode of the series transistor 3. When the parallel transistor 60 is turned off, the freewheeling current circulates through the sub-inductor 72 and the bypass diode 73.
第４実施例の効果が説明される。まず、バッテリ電圧V1及びバッテリ電圧V2は並列モード及び電圧切替期間の並列期間により徐々に均等化される。バッテリ１が不良となる時、並列トランジスタ５０がオフされる。バッテリ２が不良となる時、並列トランジスタ６０がオフされる。これにより、直流電源の信頼性が改善される。 The effect of the fourth embodiment will be described. First, the battery voltage V1 and the battery voltage V2 are gradually equalized by the parallel mode and the parallel period of the voltage switching period. When the battery 1 becomes defective, the parallel transistor 50 is turned off. When the battery 2 becomes defective, the parallel transistor 60 is turned off. Thereby, the reliability of the DC power supply is improved.
さらに、接続切替回路１０dの降圧チョッパ動作によりDCリンク電圧Vdcの急激な変化を防止することができる。インダクタ７は平滑キャパシタ２０とともにローパスフィルタを構成するため、バッテリ電流の高周波成分が低減される。このバッテリ電流の高周波成分は、バッテリ１及び２に無駄な電力損失を発生し、それらの温度を上昇させる。したがって、インダクタ７は、バッテリ１及び２のための冷却機構を簡素化し、バッテリ１及び２の寿命を延長する。 Furthermore, a sudden change in the DC link voltage Vdc can be prevented by the step-down chopper operation of the connection switching circuit 10d. Since the inductor 7 forms a low-pass filter together with the smoothing capacitor 20, the high-frequency component of the battery current is reduced. This high frequency component of the battery current causes useless power loss in the batteries 1 and 2 and raises their temperature. Thus, inductor 7 simplifies the cooling mechanism for batteries 1 and 2 and extends the life of batteries 1 and 2.
この実施例によれば、平滑キャパシタ２０が従来のフィルムキャパシタ２２の他に高容量値をもつリチウムイオンキャパシタ２１をもつ。これにより、バッテリ１及び２を流れる電流の高周波成分が低減され、バッテリ１及び２の電力損失が低減される。バッテリ１又は２が平滑キャパシタ２０に接続される時に、平滑キャパシタ２０の容量増加は平滑キャパシタ２０に流れる突入電流を増加させる。このため、従来技術において、平滑キャパシタ２０の容量増加は限界をもつ。けれども、この実施例によれば、この突入電流はインダクタ７をもつ接続切替回路の降圧チョッパ動作により抑制される。したがって、この実施例の平滑キャパシタ２０は従来のフィルムキャパシタよりも高容量のリチウムキャパシタを含むことができる。 According to this embodiment, the smoothing capacitor 20 has a lithium ion capacitor 21 having a high capacitance value in addition to the conventional film capacitor 22. Thereby, the high frequency component of the electric current which flows through the batteries 1 and 2 is reduced, and the power loss of the batteries 1 and 2 is reduced. When the battery 1 or 2 is connected to the smoothing capacitor 20, the increase in the capacity of the smoothing capacitor 20 increases the inrush current flowing through the smoothing capacitor 20. For this reason, in the prior art, the capacity increase of the smoothing capacitor 20 has a limit. However, according to this embodiment, this inrush current is suppressed by the step-down chopper operation of the connection switching circuit having the inductor 7. Therefore, the smoothing capacitor 20 of this embodiment can include a lithium capacitor having a higher capacity than a conventional film capacitor.
平滑キャパシタ２０の予備充電動作が説明される。まず直列トランジスタ３がオフされ、並列トランジスタ５０及び６０の一方又は両方が所定のPWMキャリヤ周波数でスイッチングされる。並列トランジスタ５０及び／又は６０は所定のPWMデユーティ比をもつ。これにより平滑キャパシタ２０は徐々に充電されることができる。言い換えれば、DCリンク電圧Vdcの振幅を切り替えるための接続切替回路１０dが平滑キャパシタ２０への突入電流を低減することができる。その結果、平滑キャパシタ２０及びバッテリ１及び２の損失が低減される。 The precharging operation of the smoothing capacitor 20 will be described. First, the series transistor 3 is turned off, and one or both of the parallel transistors 50 and 60 are switched at a predetermined PWM carrier frequency. The parallel transistors 50 and / or 60 have a predetermined PWM duty ratio. As a result, the smoothing capacitor 20 can be gradually charged. In other words, the connection switching circuit 10d for switching the amplitude of the DC link voltage Vdc can reduce the inrush current to the smoothing capacitor 20. As a result, the loss of the smoothing capacitor 20 and the batteries 1 and 2 is reduced.

Claims (19)

  1. 　複数のチャージデバイスの直列接続及び並列接続を切り替えるための直列スイッチ及び複数の並列スイッチに加えてインダクタを有する接続切替回路と、
    　前記スイッチの少なくとも一部を所定のPWMキャリヤ周波数でスイッチングすることにより前記インダクタの電流を低減するチョッパモードを有するコントローラと、
    　を備えることを特徴とする電圧切替式直流電源。     A connection switch circuit having an inductor in addition to a series switch and a plurality of parallel switches for switching a series connection and a parallel connection of a plurality of charge devices;
    A controller having a chopper mode for reducing current of the inductor by switching at least a part of the switch at a predetermined PWM carrier frequency;
    A voltage-switching DC power supply comprising:
  2. 　前記コントローラは、前記並列スイッチの一部及び直列スイッチをオフすることにより、不良特性をもつ前記チャージデバイスを分離する請求項１記載の電圧切替式直流電源。 The voltage-switching DC power supply according to claim 1, wherein the controller isolates the charge device having a defective characteristic by turning off a part of the parallel switch and a series switch.
  3. 　前記インバータは、並列接続されたフィルムキャパシタ及び電気二重層キャパシタからなる平滑キャパシタと並列に接続される請求項１記載の電圧切替式直流電源。 The voltage-switching DC power supply according to claim 1, wherein the inverter is connected in parallel with a smoothing capacitor comprising a film capacitor and an electric double layer capacitor connected in parallel.
  4. 　前記コントローラは、前記チャージデバイスの直列接続と並列接続との間の所定の過渡期間に前記チョッパモードを実行する請求項１記載の電圧切替式直流電源。 The voltage-switching DC power supply according to claim 1, wherein the controller executes the chopper mode during a predetermined transition period between the series connection and the parallel connection of the charge devices.
  5. 　前記コントローラは、前記前記チョッパモードを実行することにより前記チャージデバイス間の電圧差を低減する請求項１記載の電圧切替式直流電源。 The voltage-switching DC power supply according to claim 1, wherein the controller reduces the voltage difference between the charge devices by executing the chopper mode.
  6. 　前記直列スイッチは、逆並列ダイオードを有するトランジスタからなる請求項１記載の電圧切替式直流電源。 The voltage-switching DC power supply according to claim 1, wherein the series switch is composed of a transistor having an antiparallel diode.
  7. 　前記並列スイッチは、逆並列ダイオードを有するトランジスタからなる請求項１記載の電圧切替式直流電源。 The voltage-switching DC power supply according to claim 1, wherein the parallel switch comprises a transistor having an antiparallel diode.
  8. 　前記並列スイッチは、前記チャージデバイスを放電するための並列ダイオードからなる請求項１記載の電圧切替式直流電源。 The voltage-switching DC power supply according to claim 1, wherein the parallel switch comprises a parallel diode for discharging the charge device.
  9. 　前記インダクタは、前記チャージデバイス及び前記スイッチを前記インバータに接続する請求項１記載の電圧切替式直流電源。 The voltage-switching DC power supply according to claim 1, wherein the inductor connects the charge device and the switch to the inverter.
  10. 　前記インダクタは、前記チャージデバイスを前記並列スイッチに接続する請求項１記載の電圧切替式直流電源。 The voltage-switching DC power supply according to claim 1, wherein the inductor connects the charge device to the parallel switch.
  11. 　前記インダクタに接続される前記チャージデバイスは、キャパシタからなる請求項１０記載の電圧切替式直流電源。 The voltage-switching DC power supply according to claim 10, wherein the charge device connected to the inductor is a capacitor.
  12. 　前記インダクタは、前記チャージデバイスを前記インバータに接続する第１のサブインダクタと、前記並列スイッチを前記インバータに接続する第２のサブインダクタとを有する請求項１記載の電圧切替式直流電源。 2. The voltage-switching DC power supply according to claim 1, wherein the inductor includes a first sub-inductor that connects the charge device to the inverter and a second sub-inductor that connects the parallel switch to the inverter.
  13. 　前記接続切替回路は、前記並列スイッチと前記第２のサブインダクタとの間の第１の接続点に接続されるアノード電極と、前記チャージデバイスと前記第１のサブインダクタとの間の第２の接続点に接続されるカソード電極とを有するバイパスダイオードを有する請求項１２記載の電圧切替式直流電源。 The connection switching circuit includes an anode electrode connected to a first connection point between the parallel switch and the second sub-inductor, and a second electrode between the charge device and the first sub-inductor. The voltage-switching DC power supply according to claim 12, further comprising a bypass diode having a cathode electrode connected to the connection point.
  14. 　前記チャージデバイス（１）は、第２の接続切替回路（１０B）により接続される２つのブロック（１A、１B)からなり、
    　前記チャージデバイス（２）は、第３の接続切替回路（１０A）により接続される２つのブロック（２A、２B)からなり、
    　前記第２及び第３の接続切替回路（１０A、１０B）は、本質的に前記接続切替回路（１０）と本質的に同じ回路構成をもつ請求項１記載の電圧切替式直流電源。     The charge device (1) is composed of two blocks (1A, 1B) connected by a second connection switching circuit (10B).
    The charge device (2) is composed of two blocks (2A, 2B) connected by a third connection switching circuit (10A).
    The voltage-switching DC power supply according to claim 1, wherein the second and third connection switching circuits (10A, 10B) have essentially the same circuit configuration as the connection switching circuit (10).
  15. 　前記コントローラは、前記４つのブロック（１A、１B、２A、２B)を並列接続する４-並列モードをさらにもつ請求項１４記載の電圧切替式直流電源。 15. The voltage switching type DC power supply according to claim 14, wherein the controller further has a 4-parallel mode in which the four blocks (1A, 1B, 2A, 2B) are connected in parallel.
  16. 　前記コントローラは、前記４つのブロック（１A、１B、２A、２B)のうちの３つを直列接続する３-直列モードをさらにもつ請求項１４記載の電圧切替式直流電源。 15. The voltage-switching DC power supply according to claim 14, wherein the controller further has a 3-series mode in which three of the four blocks (1A, 1B, 2A, 2B) are connected in series.
  17. 　第１及び第２のチャージデバイスの直列接続及び並列接続を切り替えるための直列スイッチ及び複数の並列スイッチを有する接続切替回路と、前記接続の切替を制御するコントローラとを備える電圧切替式直流電源において、
    　前記第１のチャージデバイス（１）は、第２の接続切替回路（１０B）により接続される２つのブロック（１A、１B)からなり、
    　前記第２のチャージデバイス（２）は、第３の接続切替回路（１０A）により接続される２つのブロック（２A、２B)からなり、
    　前記第２及び第３の接続切替回路（１０A、１０B）は、本質的に前記接続切替回路（１０）と本質的に同じ回路構成をもつことを特徴とする電圧切替式直流電源。     In a voltage-switching DC power supply comprising a connection switching circuit having a series switch and a plurality of parallel switches for switching between serial connection and parallel connection of the first and second charge devices, and a controller for controlling switching of the connection,
    The first charging device (1) includes two blocks (1A, 1B) connected by a second connection switching circuit (10B).
    The second charging device (2) includes two blocks (2A, 2B) connected by a third connection switching circuit (10A),
    The voltage-switching DC power supply characterized in that the second and third connection switching circuits (10A, 10B) have essentially the same circuit configuration as the connection switching circuit (10).
  18. 　前記コントローラは、前記４つのブロックを並列に接続する４-並列モードをさらにもつ請求項１７記載の電圧切替式直流電源。 The voltage-switching DC power supply according to claim 17, wherein the controller further has a 4-parallel mode in which the four blocks are connected in parallel.
  19. 　前記コントローラは、前記４つのブロック（１A、１B、２A、２B)のうちの３つを直列接続する３-直列モードをさらにもつ請求項１７記載の電圧切替式直流電源。 The voltage-switching DC power supply according to claim 17, wherein the controller further has a 3-series mode in which three of the four blocks (1A, 1B, 2A, 2B) are connected in series.
PCT/JP2018/011684 2018-03-23 2018-03-23 Voltage switching type direct-current power supply WO2019180912A1 (en)

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