WO2019170215A1 - Dispositifs, procédés et programmes informatiques d'atténuation d'interférence dans des communications sans fil - Google Patents

Dispositifs, procédés et programmes informatiques d'atténuation d'interférence dans des communications sans fil Download PDF

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Publication number
WO2019170215A1
WO2019170215A1 PCT/EP2018/055293 EP2018055293W WO2019170215A1 WO 2019170215 A1 WO2019170215 A1 WO 2019170215A1 EP 2018055293 W EP2018055293 W EP 2018055293W WO 2019170215 A1 WO2019170215 A1 WO 2019170215A1
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WIPO (PCT)
Prior art keywords
network device
wireless communication
time domain
antennas
ofdm
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PCT/EP2018/055293
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English (en)
Inventor
Renaud-Alexandre PITAVAL
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Huawei Technologies Co., Ltd.
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Priority to PCT/EP2018/055293 priority Critical patent/WO2019170215A1/fr
Publication of WO2019170215A1 publication Critical patent/WO2019170215A1/fr

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Classifications

    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/0202—Channel estimation
    • H04L25/0212—Channel estimation of impulse response
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B7/00—Radio transmission systems, i.e. using radiation field
    • H04B7/02—Diversity systems; Multi-antenna system, i.e. transmission or reception using multiple antennas
    • H04B7/04—Diversity systems; Multi-antenna system, i.e. transmission or reception using multiple antennas using two or more spaced independent antennas
    • H04B7/0413—MIMO systems
    • H04B7/0456—Selection of precoding matrices or codebooks, e.g. using matrices antenna weighting
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B7/00—Radio transmission systems, i.e. using radiation field
    • H04B7/02—Diversity systems; Multi-antenna system, i.e. transmission or reception using multiple antennas
    • H04B7/04—Diversity systems; Multi-antenna system, i.e. transmission or reception using multiple antennas using two or more spaced independent antennas
    • H04B7/06—Diversity systems; Multi-antenna system, i.e. transmission or reception using multiple antennas using two or more spaced independent antennas at the transmitting station
    • H04B7/0613—Diversity systems; Multi-antenna system, i.e. transmission or reception using multiple antennas using two or more spaced independent antennas at the transmitting station using simultaneous transmission
    • H04B7/0615—Diversity systems; Multi-antenna system, i.e. transmission or reception using multiple antennas using two or more spaced independent antennas at the transmitting station using simultaneous transmission of weighted versions of same signal
    • H04B7/0619—Diversity systems; Multi-antenna system, i.e. transmission or reception using multiple antennas using two or more spaced independent antennas at the transmitting station using simultaneous transmission of weighted versions of same signal using feedback from receiving side
    • H04B7/0621—Feedback content
    • H04B7/0626—Channel coefficients, e.g. channel state information [CSI]
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B7/00—Radio transmission systems, i.e. using radiation field
    • H04B7/02—Diversity systems; Multi-antenna system, i.e. transmission or reception using multiple antennas
    • H04B7/12—Frequency diversity
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/0202—Channel estimation
    • H04L25/0204—Channel estimation of multiple channels
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/0202—Channel estimation
    • H04L25/022—Channel estimation of frequency response

Definitions

  • the present disclosure relates to the field of wireless communications, and more partic ularly to a network device for wireless communication, and a related method and a computer program.
  • MIMO multiple-input and multiple-output
  • devices may utilize multipath propagation of radio waves by using multiple antennas for transmitting and receiving signals. Since different propagation paths may be of different lengths, a transmitted signal may arrive to a receiving device multiple times at different instances in time. Therefore, symbols sent consecutively by a transmitting device, such as a base station, may overlap in time when they arrive to a receiving device, such as a mobile phone. This may be referred to as intersymbol interference (ISI). Furthermore, this may cause intercarrier interference (ICI), when orthogonal frequency-division multiplexing (OFDM) is used.
  • ISI intersymbol interference
  • ICI intercarrier interference
  • OFDM orthogonal frequency-division multiplexing
  • ISI and ICI may be mitigated by using a guard interval, such as a cyclic prefix (CP) with OFDM, between consecutive symbols.
  • a guard interval such as a cyclic prefix (CP) with OFDM
  • CP cyclic prefix
  • OFDM orthogonal frequency division multiple access
  • a network device for wireless communication comprises a plurality of antennas and a transceiver.
  • the transceiv er is configured to communicate signals between the antennas and at least one antenna of a second network device.
  • the network device further comprises a processor.
  • the processor is configured to acquire a plurality of time domain channel impulse response, CIR, estimates based on the signals communicated between the antennas and the at least one antenna of the second network device.
  • the processor is further configured to process the acquired plurality of time domain CIR estimates by truncating at least one time domain CIR estimate of the acquired plurality of time domain CIR estimates at a respective truncation time threshold per the time domain CIR estimate by discarding every channel tap associated with a delay larger than the respective truncation time threshold.
  • the truncation of the time domain the CIR estimate may be implemented, for example, by replacing every channel tap associated with a delay larger than the respective truncation time threshold with a zero weight or by removing every channel tap associated with a delay larger than the respective truncation time threshold.
  • any further calculation using the truncat ed time domain CIR estimates may be time-selective. Therefore, intersymbol interfer ence and intercarrier interference may be reduced in cases where the used cyclic prefix is not sufficient to absorb all unwanted interference. Since interference may be re prised, the network device may use less power in transmissions and in reception of signals, and the data rate of transmission may increase.
  • the processor is further configured to calculate an antenna weight for each antenna of the plurality of antennas based on the processed plurality of time domain CIR estimates.
  • the processor is further configured to precode a data symbol to be transmitted in an orthogonal frequency-division multi plexing, OFDM, communication between the network device for wireless communica tion and the second network device using the calculated antenna weights of the plurali ty of antennas. Since the antenna weights are calculated from the truncated time do main CIR estimate, precoding using the antenna weights is time-selective. Therefore, intersymbol interference and intercarrier interference may be reduced in cases where the used cyclic prefix is not sufficient to absorb all unwanted interference. Since inter ference may be reduced, the network device may use less power in transmissions and in reception of signals, and the data rate of transmission may increase.
  • OFDM orthogonal frequency-division multi plexing
  • the processor is further configured to combine a data symbol received from the second network device in the OFDM communication using the calculated antenna weights of the plurality of antennas. Since the antenna weights are calculated from the truncated time domain CIR estimate, com bining using the antenna weights is time- selective. Therefore, inter- symbol interference and intercarrier interference may be reduced in cases where the used cyclic prefix is not sufficient to absorb all unwanted interference. Since interference may be reduced, the network device may use less power in transmissions and in reception of signals, and the data rate of transmission may increase.
  • processor is further configured to determine each respective tmncation time threshold according to a cyclic prefix length of data symbols in the OFDM communication. Since intersymbol interference and in tercarrier interference can be highly dependent on the cyclic prefix length, such con figuration may be efficient at absorbing unwanted interference.
  • the processor is further configured to determine each respective tmncation time threshold as equal to the cyclic prefix length of the data symbols in the OFDM communication. Such configuration may tend to an interference-free transmission when the number of antennas in the plurality of antennas goes to infinity.
  • the processor is further configured to determine each respective tmncation time threshold to minimize a signal-to- interference-plus-noise ratio of a subcarrier of the OFDM communication or to mini mize an approximation of a signal-to-interference-plus-noise ratio of a subcarrier of the OFDM communication.
  • This may be especially beneficial when determining the time threshold according to the cyclic prefix may be too time-selective so that it does not collect enough channel energy through the offered multipath diversity.
  • it may be beneficial for the time threshold to be larger than the CP length in order to collect more signal energy at the cost of negligible interference.
  • the processor is further configured to determine each respective tmncation time threshold to maximize a rate of transmis sion of the OFDM communication or to maximize an approximation of a rate of trans mission of the OFDM communication. This may be especially beneficial when deter mining the time threshold according to the cyclic prefix may be too time-selective so that it does not collect enough channel energy through the offered multipath diversity. Thus, it may be beneficial for the time threshold to be larger than the CP length in or der to collect more signal energy at the cost of negligible interference.
  • the transceiver is further config ured to receive and measure pilot signals transmitted by the second network device.
  • the processor is further configured to acquire the plurality of time domain CIR esti mates based on the received and measured pilot signals.
  • Such configuration may sim plify the implementation and reduce the signaling overhead of the network, since the network device does not need to transmit the pilot signals from its plurality of anten nas.
  • the transceiver is further config ured to transmit pilot signals and receive feedback signals transmitted by the second network device in response to the pilot signals.
  • the processor is further configured to acquire the plurality of time domain CIR estimates based on the received feedback signals.
  • the second network device may measure the pilot signals to ac quire a channel estimate.
  • the second network device may then quantize the channel estimate by mapping it to an element of a codebook shared between the network device and the second network device and transmit the index of the element to the network device in the feedback signal.
  • Such configuration may be necessary to acquire channel estimation in the absence of reciprocity between the reception and transmission chan nels.
  • the pilot signals comprise OFDM signals with a longer cyclic prefix length than a cyclic prefix length of data symbols in the OFDM communication. This configuration may reduce interference of the pilot signals, which may increase accuracy of the time domain CIR estimate.
  • pilot signals comprise a single car rier signal enabling the network device to acquire the plurality of time domain CIR estimates via deconvolution.
  • Such configuration may simplify the implementation of the network device, since the time domain CIR estimate can be acquired efficiently by a direct deconvolution procedure and using pilot signals that have the same duration than data OFDM signals.
  • the processor is further configured to calculate a frequency response from the processed plurality of time domain CIR estimates using a Fourier transform.
  • the processor is further configured to perform the calculation of the antenna weights of the plurality of antennas based on the calculated frequency response.
  • the frequency response can be efficiently calculated using, for example, fast Fourier transform.
  • the antenna weights can be efficiently calculated for each OFDM subcarrier based on the value of the fre quency response at the frequency of the subcarrier.
  • the network device for wireless communication comprises one of a network node device and a client device
  • the second network device comprises one of the client device and the network node device, respectively.
  • Such configuration allows the invention to be used in communication between a base station and a client device.
  • either the base station or the client device can be configured to precode and/or combine symbols.
  • a method comprises communi cating, by a network device for wireless communication, signals between antennas of the network device for wireless communication and at least one antenna of a second network device.
  • the method further comprises acquiring, by the network device for wireless communication, a plurality of time domain channel impulse response, CIR, estimates based on the signals communicated between the antennas of the network device for wireless communication and the at least one antenna of the second network device.
  • the method further comprises processing, by the network device for wireless communication, the acquired plurality of time domain CIR estimates by tmncating at least one time domain CIR estimate of the acquired plurality of time domain CIR esti mates at a respective tmncation time threshold per the time domain CIR estimate by discarding every channel tap associated with a delay larger than the respective trunca tion time threshold.
  • the truncation of the time domain the CIR estimate may be im plemented, for example, by replacing every channel tap associated with a delay larger than the respective truncation time threshold with a zero weight or by removing every channel tap associated with a delay larger than the respective tmncation time threshold.
  • any further calculation using the tmncated time domain CIR estimates may be time-selective. Therefore, intersymbol interference and intercarrier interference may be reduced in cases where the used cyclic prefix is not sufficient to absorb all unwanted interference. Since interference may be reduced, the network device may use less power in transmissions and in reception of signals, and the data rate of transmission may increase.
  • the method further comprise calculat ing, by the processor, an antenna weight for each antenna of the plurality of antennas based on the processed plurality of time domain CIR estimates.
  • the method further comprises precoding, by the processor, a data symbol to be transmitted in an orthogo- nal frequency-division multiplexing, OFDM, communication between the network device for wireless communication and the second network device using the calculated antenna weights of the plurality of antennas. Since the antenna weights are calculated from the tmncated time domain CIR estimate, precoding using the antenna weights is time-selective. Therefore, intersymbol interference and intercarrier interference may be reduced in cases where the used cyclic prefix is not sufficient to absorb all unwanted interference. Since interference may be reduced, the network device may use less pow er in transmissions and in reception of signals, and the data rate of transmission may increase.
  • the method further comprises combining, by the processor, a data symbol received from the second network device in the OFDM communication using the calculated antenna weights of the plurality of antennas. Since the antenna weights are calculated from the truncated time domain CIR estimate, combining using the antenna weights is time-selective. Therefore, inter symbol interference and intercarrier interference may be reduced in cases where the used cyclic prefix is not sufficient to absorb all unwanted interference. Since interfer ence may be reduced, the network device may use less power in transmissions and in reception of signals, and the data rate of transmission may increase.
  • the method further comprises determining, by the processor, each respective tmncation time threshold according to a cyclic prefix length of data symbols in the OFDM communication. Since intersymbol interference and intercarrier interference can be highly dependent on the cyclic prefix length. This may be efficient at absorbing unwanted interference.
  • the method further comprises determining, by the processor, each respective tmncation time threshold as equal to the cyclic prefix length of the data symbols in the OFDM communication. This may tend to an interference-free transmission when the number of antennas in the plurality of antennas goes to infinity.
  • the method further comprises determining, by the processor, each respective tmncation time threshold to minimize a signal-to-interference-plus-noise ratio of a subcarrier of the OFDM communication or to minimize an approximation of a signal-to-interference-plus-noise ratio of a subcarri er of the OFDM communication.
  • This may be especially beneficial when determining the time threshold according to the cyclic prefix may be too time- selective so that it does not collect enough channel energy through the offered multipath diversity.
  • it may be beneficial for the time threshold to be larger than the CP length in order to collect more signal energy at the cost of negligible interference.
  • the method further comprise, determining, by the processor, each respective tmncation time threshold to maximize a rate of transmission of the OFDM communication or to maximize an approximation of a rate of transmission of the OFDM communication.
  • each respective tmncation time threshold to maximize a rate of transmission of the OFDM communication or to maximize an approximation of a rate of transmission of the OFDM communication.
  • the method further comprises receiving and measuring, by the transceiver, pilot signals transmitted by the second network device.
  • the method further comprises acquiring, by the processor, the plurali ty of time domain CIR estimates based on the received and measured pilot signals. This may simplify the implementation and reduce the power consumption of the net work device, since the network device does not need to transmit the pilot signals.
  • the method further comprises transmitting pilot signals and receiving, by the transceiver, feedback signals transmit ted by the second network device in response to the pilot signals.
  • the method further comprises acquiring, by the processor, the plurality of time domain CIR estimates based on the received feedback signals.
  • the second network device may measure the pilot signals to acquire a channel estimate.
  • the second network device may then quantize the channel estimate by mapping it to an element of a codebook shared between the network device and the second network device and transmit the index of the element to the network device in the feedback signal. This may simplify the implementation of the network device, since the network device does not need to receive or measure the pilot signals.
  • the pilot signals comprise OFDM signals with a longer cyclic prefix length than a cyclic prefix length of data symbols in the OFDM communication. This may reduce interference of the pilot sig nals, which may increase accuracy of the time domain CIR estimate.
  • pilot signals comprise a single carrier signal enabling the network device to acquire the plurality of time domain CIR estimates via deconvolution. This may simplify the implementation of the network device, since the time domain CIR estimate can be acquired efficiently by a direct de- convolution procedure.
  • the method further comprises calculating, by the processor, a frequency response from the processed plurality of time domain CIR estimates using a Fourier transform.
  • the method further comprises per forming, by the processor, the calculation of the antenna weights of the plurality of antennas based on the calculated frequency response.
  • the frequency response can be efficiently calculated using, for example, fast Fourier transform.
  • the antenna weights can be efficiently calculated for each OFDM subcarrier based on the value of the frequency response at the frequency of the subcarrier.
  • a computer program comprises program code configured to perform the method according to the second aspect, when the computer program is executed on a computer. Since the time domain CIR estimates are truncated, any further calculation using the tmncated time domain CIR estimates may be time-selective. Therefore, intersymbol interference and intercar rier interference may be reduced in cases where the used cyclic prefix is not sufficient to absorb all unwanted interference. Since interference may be reduced, the network device may use less power in transmissions and in reception of signals, and the data rate of transmission may increase.
  • Fig. 1 is a block diagram illustrating a network device for wireless communication
  • Fig. 2 is a diagram illustrating multipath propagation
  • Fig. 3 is a diagram illustrating insufficient cyclic prefix length
  • Fig. 4 is a diagram illustrating a multiple-input and multiple-output wireless communi cation system
  • Fig. 5 is diagram illustrating precoding in an orthogonal frequency-division multiplex ing system
  • Fig. 6 is a flow diagram illustrating a method
  • Fig. 7 is another flow diagram illustrating a method
  • Fig. 8 is a signaling diagram illustrating a method
  • Fig. 9 is a diagram illustrating time domain channel impulse responses before trunca tion
  • Fig. 10 is a diagram illustrating time domain channel impulse responses after trunca tion
  • Fig. 11 a diagram illustrating simulation results
  • Fig. 12 another diagram illustrating simulation results
  • FIG. 13 another diagram illustrating simulation results
  • FIG. 14 another diagram illustrating simulation results
  • FIG. 15 another diagram illustrating simulation results
  • Fig. 16 another diagram illustrating simulation results.
  • a disclosure in connection with a described method may also hold true for a corresponding device or system configured to perform the method and vice versa.
  • a corre sponding device may include a unit to perform the described method step, even if such unit is not explicitly described or illustrated in the figures.
  • a corresponding method may include a step performing the described functionality, even if such step is not explicitly described or illustrated in the figures.
  • the invention allows mitigating interference effects caused by multipath propagation in OFDM MIMO systems with insufficient CP length while still utilizing the multipath diversity provided by MIMO.
  • Fig. 2 illustrates a schematic representation of multipath propagation according to an example.
  • a base station 201 transmits a signal using, for example, an antenna array to a client device 202.
  • the signal may propagate to the cli ent device by various different paths.
  • the signal may propagate to the client device by a direct path 204, or the signal may be reflected from an object 203 and propagate to the client device 202 by a reflected path 205.
  • various other mechanisms such as scattering and diffraction, may affect the propagation of radio signals and cause multipath propagation.
  • description of radio wave propagation using such simple paths may not be possible.
  • the carrier waves propagated by these paths may be at different phases at the location of the client device 202.
  • destmctive interference may take place at the location of the client device 202, which may significantly reduce the signal power received by the client device 202.
  • the interference caused by the length difference between the two paths is constructive, the received signal power may increase.
  • multipath propagation may be utilized to provide a wireless connection between two devices, if the phase difference between different paths can be configured to be such that constructive interference takes place at the receiving device. Therefore, multipath propagation can be utilized to provide a wireless connection even when there is no line of sight between the devices.
  • multipath propagation may be utilized to transmit more than one signal to more than one device using the same channel.
  • the base station 201 may multiply data symbols to be transmitted by well-chosen complex numbers before the base station transmits the symbols 201 using multiple antennas. These numbers may be referred to as antenna weights, and they can be used to construct a precoder. The process of applying these weights to a data symbol to be transmitted may be referred to as precoding. On the other hand, these coefficients may also be applied to receive symbols, which may be referred to as combining.
  • the antenna weights behave asymptotically as a spatial filter as the number of antennas grows to infinity: only transmitted signals that are collinear with a channel path are conveyed by this path, otherwise it is totally attenuated.
  • Fig. 3 illustrates a schematic representation of orthogonal frequency-division multi plexing (OFDM) symbols according to an example.
  • OFDM orthogonal frequency-division multi plexing
  • information is transmit ted using multiple subcarriers, and each subcarrier is modulated with some modulation scheme.
  • This modulation scheme may be, for example, phase-shift keying (PSK) or quadrature amplitude modulation (QAM).
  • PSK phase-shift keying
  • QAM quadrature amplitude modulation
  • the sum of these modulated subcarriers form an OFDM symbol, and these OFDM symbols are transmitted consecutively in time.
  • a cyclic prefix, CP may be added to each symbol in order to reduce intercarrier interference (ICI) and intersymbol interference (ISI).
  • ICI intercarrier interference
  • ISI intersymbol interference
  • the CP comprises a copy of the end of the corresponding symbol
  • linear convolution of the channel and the symbol can be modelled as circular convolution. If the CP length is longer than the delay spread of the channel, the multipath channel is transformed to multiple orthogo nal flat per-subcarrier channels.
  • Fig. 3 illustrates ISI in a case where the CP length is insufficient.
  • OFDM symbols 301 may correspond, for example, to the direct path of Fig. 2, while OFDM symbols 302 may correspond to the reflected path of Fig. 2. Since the reflected path is physically longer than the direct path, OFDM symbols 302 may arrive to the client device at a later time than symbols 301. Thus, consecutive OFDM symbols may overlap in time, if the CP length is not sufficient.
  • OFDM symbol 303 overlaps in time with OFDM symbol 305 during the time period 306, because the CP length 304 is insufficient. ISI may be especially significant in systems where multipath propagation is prominent and the delay spread is large compared to the CP length.
  • Delay spread may refer to the time within which signals propagating by different paths arrive to a device. Further more, if the subcarrier spacing is large in frequency, the OFDM symbols are short in time, which may increase ISI even further, especially if the CP length is scaled down with the symbol length. ISI may be mitigated by increasing the CP length. Flowever, since the CP does not comprise any payload, increasing the CP length may reduce the data throughput.
  • Fig. 4 illustrates a schematic representation of a MIMO wireless communication sys tem.
  • a device 401 may utilize multiple antennas 402 and multipath propagation to communicate with multiple devices and/or multiple antennas at the same time.
  • Each of the reception antennas 403 may be connected to a different client device, some subset of the reception antennas 403 may be connected to a single device, or all of the reception antennas 403 may be connected to the same device.
  • the system may be referred to as multi-user MIMO.
  • the channel matrix H models the N r x N t possible connections between the N r reception antennas 403 and N t transmission antennas 402.
  • an element of H may indicate how the phase and amplitude of a complex data symbol change when the symbol is transmitted between the corresponding trans mission and reception antennas of the MIMO system.
  • the reception antennas 403 may also transmit signals and the transmission antennas 402 may also receive signals.
  • a similar model may be constructed these cases. These names only refer to the function on the antennas in the mathematical model presented above.
  • a precoder W k m( H fe ) G C NtXNs can be constmcted, where N s is the number of transmission streams and m is a func- tion that maps a channel matrix to a precoder matrix.
  • the precoder may be configured to, for example, achieve signal-to-noise ratio (SNR) minimization or data rate maximi zation.
  • SNR signal-to-noise ratio
  • a precoder has unit energy as
  • F 1.
  • Fig. 1 is a block diagram that illustrates a network device 100 for wireless communica tion.
  • the network device 100 comprises a transceiver 101 and a processor or a processing unit 102 coupled to the transceiver 101, which may be used to implement the functionalities described later in more detail.
  • the network device 100 also comprises a plurality of antennas 103 coupled to the transceiver 101.
  • the processor 102 may comprise, for example, one or more of various processing de vices, such as a co-processor, a microprocessor, a controller, a digital signal processor (DSP), a processing circuitry with or without an accompanying DSP, or various other processing devices including integrated circuits such as, for example, an application specific integrated circuit (ASIC), a field programmable gate array (FPGA), a micro controller unit (MCU), a hardware accelerator, a special-purpose computer chip, or the like.
  • various processing de vices such as a co-processor, a microprocessor, a controller, a digital signal processor (DSP), a processing circuitry with or without an accompanying DSP, or various other processing devices including integrated circuits such as, for example, an application specific integrated circuit (ASIC), a field programmable gate array (FPGA), a micro controller unit (MCU), a hardware accelerator, a special-purpose computer chip, or the like.
  • ASIC application specific integrated circuit
  • FPGA field programmable gate array
  • the network device 100 for wireless communication may further comprise a memory (not illustrated in Fig. 1) that is configured to store, for example, computer programs and the like.
  • the memory may include one or more volatile memory devices, one or more non-volatile memory devices, and/or a combination of one or more volatile memory devices and non-volatile memory devices.
  • the memory may be embodied as magnetic storage devices (such as hard disk drives, floppy disks, magnetic tapes, etc.), optical magnetic storage devices, and semiconductor memories (such as mask ROM, PROM (programmable ROM), EPROM (erasable PROM), flash ROM, RAM (random access memory), etc.).
  • the transceiver 101 is configured communicate signals between the antennas 103 and at least one antenna of a second network device.
  • the second network device may be, for example, a client device, if the network device 100 is a base station, or vice versa.
  • the base station may include e.g. a macro-eNodeB, a pico-eNodeB, a home eNodeB, a fifth-generation base station (gNB) or any such device providing an air interface for client devices to connect to the wireless network via wireless transmissions.
  • the client device may be any of various types of devices used directly by an end user entity and capable of communication in a wireless network, such as user equipment (UE). Such devices include but are not limited to smartphones, tablet computers, smart watches, lap top computers, Intemet-of- Things (IoT) devices etc.
  • IoT Intemet-of- Things
  • the signals may comprise, for example, pilot signals, also referred to as reference sig nals.
  • the network device 100 may transmit pilot signals using the plurality of antennas 103 to the at least one antenna of the second network device, and the second network device can feedback quantized channel state information to the network device. Alter natively, the second network device may transmit pilot signals to the network de vice 100.
  • the processor 102 is configured to acquire a plurality of time domain channel impulse response, CIR, estimates based on the signals communicated between the antennas 103 and the at least one antenna of the second network device.
  • the plurality of time domain CIR estimates may comprise, for example, a CIR estimate from each antenna of the plurality of antennas 103 to each antenna of the of the second network device.
  • the processor 102 is further configured to process the acquired plurality of time do main CIR estimates by truncating at least one time domain CIR estimate of the ac quired plurality of time domain CIR estimates at a respective tmncation time threshold per the time domain CIR estimate by discarding every channel tap associated with a delay larger than the respective truncation time threshold. For example, all taps in a time domain CIR estimate that correspond to a time delay larger than the time thresh old may be removed or replaced with zero weights.
  • the same truncation threshold can be used for each time domain CIR estimate of the plurality of time domain CIR esti mates, the same truncation threshold may be used for a subset of the CIR estimates, or a different tmncation threshold may be used for ach CIR estimate in the plurality of CIR estimates. Furthermore, only a subset of the CIR estimates in the plurality of time domain CIR estimates may be tmncated in the processing.
  • any further calculation using the tmncated time domain CIR estimates may be time-selective. Therefore, inter symbol interference and intercarrier interference may be reduced in cases where the used cyclic prefix is not sufficient to absorb all unwanted interference. Since interfer- ence may be reduced, the network device may use less power in transmissions and in reception of signals, and the data rate of transmission may increase.
  • the time threshold is determined according to a cyclic prefix length of the data symbols in an OFDM transmission.
  • the time threshold can be determined as equal to the cyclic prefix length of the data symbols in the OFDM transmission.
  • the OFDM transmission may become interfer ence free as the number of antennas in the plurality of antennas goes to infinity.
  • the time threshold is determined to optimize a performance criteri on of the OFDM transmission or an approximation of a performance criterion of the OFDM transmission.
  • the performance criterion can comprise a signal-to- interference-plus-noise ratio of a subcarrier of the OFDM transmission or a maximum rate of transmission of the OFDM transmission.
  • the processor 102 is further configured to calculate an antenna weight for each antenna of the plurality of antennas 103 based on the processed plurality of time domain CIR estimates.
  • the antenna weights may be used to form a precoder or to form a combiner.
  • a frequency response is calculated from the time do main CIR estimate using a Fourier transform, and the frequency response is used to calculate antenna weights for different OFDM subcarriers.
  • the processor 102 is further configured to precode a data symbol to be transmitted in an OFDM communication between the network device 100 for wireless communication and the second network device using the calculated antenna weights of the plurality of antennas 103.
  • the network device 100 may also combine data symbols received from the second network device in the OFDM trans mission using the calculated antenna weights of the plurality of antennas 103. Since the antenna weights are calculated from the truncated time domain CIR estimate, the pre coding and combining are time- selective and are only combinations of certain channel paths that fall within the desired delay range. This enables cancelling or minimizing ICI and ISI among OFDM symbols, when the CP length is insufficient to absorb all interference.
  • the invention enables improving the performance of MIMO OFDM systems whose CP length is insufficient to remove ISI and/or ICI.
  • the invention can enable a nearly inter ference-free transmission without increasing the CP length and thus without decreasing the transmission rate.
  • the invention exploits the spatial selectivity of- fered by plurality of antennas to construct a new MIMO precoding and/or combing method which is both time-delay and frequency selective.
  • the time selectivity aims at removing undesired delayed signal, while the frequency selectivity makes the precoder well-suited for OFDM waveforms.
  • the invention also exploits the fact that a short CP, even though shorter than the maximum delay spread, is beneficial to collect most of the energy of the channel contributing to the useful signal power.
  • the inven tion can be used to construct subcarrier-specific precoders according to the frequency responses of the channel. This may be beneficial, because with OFDM, data symbols are by design transmitted on the frequency-domain passing through different frequen cy-flat channels. Subcarrier-level, and thus frequency-selective, precoding enables a high precoding gain when combined with OFDM.
  • Fig. 5 illustrates a block diagram of precoding in OFDM transmission.
  • each trans mission vector x k can be fed into an inverse fast Fourier transform (IFFT) block 504_l, 504_2, 504_3.
  • IFFT inverse fast Fourier transform
  • Each IFFT block implements the OFDM for a corresponding antenna by producing a complex vector of length /V fft .
  • the sample m of OFDM symbol i can be expressed as where elements of the vector s ; [m] refer to the antennas of the plurality of anten nas 103 and N sc is the number of subcarriers used in the OFDM transmission.
  • the total signal can be expressed as where the CP of length N cp has also been taken into account.
  • each IFFT block can be converted into an analog signal, the analog sig nal can be modulated to an RF frequency. These operations can be performed, for ex ample, by the transceiver 101. Each modulated analog signal can be transmitted with the corresponding antenna of the plurality of antennas 103.
  • the received signal for 0 ⁇ k ⁇ (/V fft — 1) is
  • z[Zc] is a zero-mean additive white Gaussian noise (AWGN) with variance s .
  • AWGN additive white Gaussian noise
  • the received signal is then demodulated by FFT.
  • the demodulated symbol on the Zth subcarrier is
  • Fig. 6 shows a diagram 600 of an example method according to an example.
  • the method 600 comprises communicating, by a network device for wireless commu nication, signals between antennas of the network device for wireless communication and at least one antenna of a second network device, step 601.
  • the method further 600 comprises, acquiring, by the network device for wireless communication, a plurality of time domain channel impulse response, CIR, estimates based on the signals communi cated between the antennas of the network device for wireless communication and the at least one antenna of the second network device (801), step 601.
  • CIR time domain channel impulse response
  • the method 600 further comprises processing, by the network device for wireless communication, the acquired plurality of time domain CIR estimates by truncating at least one time domain CIR estimate of the acquired plurality of time domain CIR esti mates at a respective tmncation time threshold per the time domain CIR estimate by discarding every channel tap associated with a delay larger than the respective tmnca tion time threshold, step 602.
  • the method further comprises precoding, by the processor, a data sym bol to be transmitted in an orthogonal frequency-division multiplexing, OFDM, com munication between the network device for wireless communication and the second network device using the calculated antenna weights of the plurality of antennas, step 604.
  • OFDM orthogonal frequency-division multiplexing
  • the method 600 can be performed by the network device 100.
  • the acquiring step 601 can, for example, be performed by the transceiver 101 and the processor 102, and the tmncating step 602, the calculating step 603, and the precoding step 604 can, for ex ample, be performed by the processor 102. Further features of the method 600 directly result from the functionality of the network device 100.
  • the method 600 can be per formed by a computer program.
  • Fig. 7 shows a diagram 700 of an example method for acquiring a frequency- selective MIMO precoder according to an example.
  • the network device 100 acquires a time domain MIMO CIR estimate where H, G £ NrXNt is the channel coefficients at delay l in samples and d h is the Kron- ecker delta function.
  • the time domain MIMO CIR estimate may also be referred to as plurality of time domain CIR estimates, since the matrix H[n] comprises multiple CIR estimates.
  • the MIMO CIR estimate is truncated according to a predeter mined time threshold r tr resulting in a truncated CIR estimate where r tr ⁇ L.
  • every channel tap corresponding to a delay larger than the time threshold can be replaced with a zero-weight, removed, or ignored in following computations.
  • Such configuration may tend to an interference- free transmission when the number of antennas in the plurality of antennas 103 goes to infinity.
  • this configuration may be too time- selective so that it does not collect enough channel energy through the offered multipath diversity.
  • the time threshold is determined to optimize a per formance criterion of the OFDM transmission.
  • a criterion may be, for example, the signal-to-interference-plus-noise ratio (SINR) or the data transmission rate.
  • SINR signal-to-interference-plus-noise ratio
  • Chan nel taps escaping the CP length may contribute more to useful signal power than to interference.
  • the truncated MIMO CIR estimate is Fourier transformed, and the trans formed MIMO CIR estimate is used to evaluate frequency-domain truncated channel estimate at different subcarrier frequencies. This may also be referred to as channel frequency response. For kth subcarrier the frequency response is
  • Frequency selective MIMO precoders can then be evaluated for each subcarrier using the frequency-domain truncated channel estimate, step 704.
  • the precoder for the kth subcarrier is then G C NtXNs . In other words, steps 703 and 704
  • MRT zero-forcing
  • RZF regular- ized zero-forcing
  • D is a diagonal matrix whose diagonal elements are equal to the norm of the rows of H tr ⁇ and h is a regularization parameter.
  • RZF precoding is generalization of minimum mean square error (MMSE) precoding.
  • MMSE minimum mean square error
  • a single truncation threshold can be applied directly to the multi-user channel as described above.
  • precoder can then be computed as above.
  • a data symbol on subcarrier k to be transmitted in an OFDM symbol can be precoded according to the frequency-selective MIMO precoder MZ ⁇ step 705.
  • the network device 100 can precode a data symbol to be transmitted in an OFDM communication between the network device 100 for wireless communication and a second network device using the calculated antenna weights of the plurality of anten nas 103.
  • the method 700 can be performed by the network device 100.
  • the acquiring step 701 can, for example, be performed by the transceiver 101 and the processor 102, and the tmncating step 702, the calculating step 703, and the precoding steps 704, 705 can, for example, be performed by the processor 102. Further features of the method 700 di- rectly result from the functionality of the network device 100.
  • the method 700 can be performed by a computer program.
  • Fig. 8 illustrates a signaling diagram between the network device 100 for wireless communication and a second network device 801 according to an example.
  • the second network device 801 transmits pilot signals to the network device 100, step 802.
  • the network device 100 may have transmitted pilot signals to the second network device 801.
  • the second network device 801 may trans mit feedback signals to the network device in step 802.
  • the second net work device 801 may measure the pilot signals transmitted by the network device 100 to acquire a channel estimate in time domain or in frequency domain.
  • the second net work device 801 may then quantize the channel estimate by mapping it to an element of a codebook shared between the network device 100 and the second network device 801 and transmit the index of the element to the network device 100 in the feedback signal.
  • the pilot signals transmitted by the network device 100 or by the second network de vice 801 may have the same stmcture as OFDM symbols with CP used in data trans mission.
  • the pilots may be transmitted with a dedicated signaling with different stmcture from the data transmission signals.
  • pilot signals are transmitted in the frequency domain using OFDM with a CP length longer than the CP length used for data trans mission.
  • Each pilot symbol is used to estimate a per-subcarrier channel H k G (C N, xNt for every k and a time domain CIR estimate can be obtained by inverse Fourier trans form as
  • the CIR is estimated from a single-carrier pilot signal and us ing de-convolution procedures. Such procedure could allow having the same symbol duration for pilots and data.
  • pilot signals may not be transmitted on every subcarrier but with a certain subcarrier granularity, and channel estimations for subcarriers inside this granularity can be obtained by interpolation us ing, for example, Wiener filtering.
  • Fig. 8 illustrates the following steps in Fig. 8 and correspond to the steps of the example of Fig. 7.
  • the net work device 100 can precode data symbols and transmit the precoded symbols to the second network device 801 and combine symbols that the network device 100 has re ceived from the second network device 801, step 705’.
  • the second network device 801 can transmit and receive data symbols from the network device 100, step 803.
  • Fig. 9 and Fig. 10 illustrate time domain CIR estimate tmncation according to an ex ample.
  • time domain CIR estimates of three antennas are presented as func tions of time-delay.
  • the antennas may be, for example, part of the plurality of antennas 103.
  • Each of the CIR estimates comprises multiple channel taps 902, 903, 904, and each tap corresponds to a time-delay and a power value.
  • Each CIR estimate indicates the received channel taps, when a single tap is transmitted by a corresponding antenna.
  • the time-delay of each tap indicates the delay between transmitting the single channel tap and receiving the corresponding channel tap.
  • an antenna in of the second network device 801 may transmit an impulse
  • the channel taps 902, 903, 904 may represent the impulses received by three antennas of the network device 100 as the transmitted impulse arrives to the antennas by, for example, different propaga tion paths. It should be appreciated that this is only a theoretical description of the CIR estimate, and the CIR estimate may be acquired by different procedures, such as by using pilot signals that are not impulses.
  • Fig. 9 illustrates the corresponding three CIR estimates after truncation at the tmncation time threshold r tr . All of the channel taps corresponding to time-delays larger than z tr have been discarded. Thus the tmncated CIR estimates 902’, 903’, 904’ do not comprise any non-zero taps that correspond to time delays greater than the tmn cation time threshold r tr .
  • the time threshold r tr in Fig. 9 and Fig. 10 can be selected so that the corresponding precoder maximizes the SINR of a given subcarrier as
  • T max (SNR) arg max SINR ; (r tr ) .
  • the threshold r tr can be selected to maximize the maximum achievable rate of the transmission at a given SNR as
  • T max (SNR) arg max C(SNR; T tr ) .
  • Such computation requires the time-domain CIR, the system parameters, and the SNR. These quantities may nevertheless be quite complex to compute. However, a low- complexity asymptotic approximation of the SINR may be used for optimization. This computation is based on the power-delay profile of the channel, the system parameters, and the SNR.
  • the operational SNR needed to reach, for example, a certain symbol error rate (SER) level is shifting in the low SNR-regime by a factor of 10 log 10 N t [dB] Therefore for concrete performance optimization to be applied in the finite- antenna regime, it may be more relevant to consider the perfor- mance under a given operational SNR scaled according to the number of transmit an tennas.
  • SER symbol error rate
  • SINR j (T tr ) ® SINR”(r tr ) as N t ® ⁇ with R/s3 ⁇ 4 ® 0 such that PN t /a % SNR op is fixed, where
  • the average energy of the tap E p can be approximated from a single channel acquisi tion in the large antenna regime as
  • the asymptotic form SINRf (r tr ) is dependent of the subcarrier index /. This is be cause not all subcarriers in the IFFT are occupied, and as a result, subcarriers in the middle of the band receive more interference from neighboring subcarriers than sub carriers at the edge of the band. Thus, the SINR in the middle of the band is slightly less than the SINR at the edge of the band.
  • 2 in the SINR expression corresponds to a SNR loss compared from a drop of multipath diversity.
  • An optimized threshold can be selected to maximize an asymptotic SINR, or the as ymptotic achievable rate of the transmission as
  • a threshold can then be selected to maximize the achievable rate at a given operational SNR as
  • T max (SNR 0p ) arg max SINR°°(T tr ).
  • the optimum value r max depends of SNR op : On one hand as SNR op ® 0 the interfer ence term becomes negligible in the SINR term and r max ® , on the other hand, as SNR op ® oo the interference dominates and r max ® /V cp + 1 . Otherwise, the value T m a x(SNR op ) can be found by exhaustive search by choosing a value of r tr (which may not be unique) which maximize R(SNR op ; r tr ) or SINR” (r tr ) .
  • the optimization for a given constellation can be performed for a unique SNR value where the constellation is expected to reach, for example, a sufficiently-low error rate. Then, a unique threshold per constellation can be used.
  • Fig. 11 shows diagram 1100 illustrating simulation results for the long-term evolution (LTE) extended typical urban (ETU) channel model.
  • LTE long-term evolution
  • ETU typical urban
  • the bottom subfigure 1130 of Fig. 11 shows the asymptotic rate R°°(SNR op ; r tr ) with the proposed time-frequency selective precoding (labeled TF-precoding on the figures) as a function of the time threshold r tr and assuming an operational SNR of 25 dB.
  • the rate is a step function changing when the next channel tap is included in the precoder as r tr increases.
  • the optimization here requires only to search a maximum in a set of 4 values.
  • Fig. 12 shows a diagram 1200 illustrating the asymptotic rate R°°(SNR op ; r max ) as a function of SNR op compared to the respective asymptotic rate with conventional fre quency-selective precoding (labeled F-precoding on the figures) with normal and ex tended CP.
  • optimized time-frequency selective precoding according to the invention always performed the same or better than the conventional frequency- selective precoding, and provides notable improvements in the high-SNR region where the system is interference-limited.
  • Using an extended CP convert interference to useful power and thus improve the SINR in the high-SNR regime.
  • Optimized time-frequency selective precoding does not have an increase rate penalty from an extended CP and also remove the interference by an appropriate precoding in the high SNR regime.
  • Fig. 13 shows a diagram 1300 illustrating simulation results for new radio (NR) tapped delay line (TDL-C) channel with 2 microsecond (ps) delay scaling.
  • NR new radio
  • TDL-C tapped delay line
  • ps microsecond
  • Channel taps of this model are presented in the top subfigure 1310.
  • This channel model can be regarded as more frequency selective than the LTE ETU channel model.
  • the asymptotic rate R°°(SNR op ; r tr ) for an operational SNR of 20 dB shown on the bottom subfigure 1330 of Fig. 13 is maximized for any r max G [203, 204, ... , 229] which corresponds to precode according to the 11 first channel taps of which 6 of them are outside the CP.
  • the asymptotic SINRs, SINR”(r tr ) and SINR” 00 (T tr ) are maximized with any truncation threshold in r tr G [230,231, ... , 302] and r tr G [203, 204, ... , 229] , respectively, as is presented in the middle subfigure 1320. This shows again that max imizing the SINR on a middle subcarrier leads to same optimized truncation time threshold than maximizing the rate.
  • the optimization here requires to search a maxi mum in a set of 20 values.
  • Fig. 14 shows a diagram 1400 illustrating the optimized asymptotic rate R°°(SNR op ; r max ) as a function of SNR op compared to conventional frequency- selective precoding with normal and extended CP.
  • optimized time-frequency selective precoding according to the invention is not interference-limited in the high- SNR regime.
  • SER finite-constellation symbols
  • a block-fading Rayleigh channel is assumed with different random realization every 14 OFDM symbols.
  • the transmitter employs a MRT precoding as described above, i.e. given the CIR h[n ⁇ , the precoder on subcarrier k is w N fft. in all cases, the SER curves are obtained using only a single threshold in the time-frequency precoding.
  • Fig. 16 shows a diagram 1600 illustrating simulation results for transmission of a 16QAM with the TDL-C channel model with 2 pi s delay scaling.
  • the proposed inven tion can provide here 3dB SNR gain at 10 -3 SER compared to conventional frequen cy-selective precoding.
  • Fig. 16 shows a diagram 1600 illustrating simulation results for transmission of a 16QAM with the TDL-C channel model with 2 pi s delay scaling.
  • the functionality described herein can be performed, at least in part, by one or more computer program product components such as software components.
  • the client device 110 and/or network device 100 comprise a processor configured by the program code when executed to execute the embodiments of the operations and functionality described.
  • the functionality described herein can be performed, at least in part, by one or more hardware logic components.
  • illustrative types of hardware logic components include Field-programmable Gate Arrays (FPGAs), Pro- gram-specific Integrated Circuits (ASICs), Program- specific Standard Products (ASSPs), System-on-a-chip systems (SOCs), Complex Programmable Logic Devices (CPLDs), and Graphics Processing Units (GPUs).

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Abstract

L'invention vise à atténuer des effets d'interférence indésirables provoqués par une propagation par trajets multiples dans des systèmes de communication sans fil à entrées multiples et sorties multiples (MIMO) et à multiplexage par répartition orthogonale de la fréquence (OFMD) tout en continuant à utiliser la diversité des trajets multiples liée au système MIMO. Une estimation d'une réponse impulsionnelle de canal (CIR) tronquée est calculée à partir d'une estimation de CIR obtenue. Des pondérations d'antenne sélectives dans le temps peuvent être calculées à partir de l'estimation de CIR tronquée. Les pondérations d'antenne sélectives dans le temps peuvent servir à réduire une interférence inter-porteuses et une interférence inter-symboles dans une communication OFDM. L'invention concerne également un dispositif de réseau pour communication sans fil, un procédé et un programme informatique.
PCT/EP2018/055293 2018-03-05 2018-03-05 Dispositifs, procédés et programmes informatiques d'atténuation d'interférence dans des communications sans fil WO2019170215A1 (fr)

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Cited By (1)

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WO2021074000A1 (fr) * 2019-10-11 2021-04-22 Fraunhofer-Gesellschaft zur Förderung der angewandten Forschung e.V. Multiplexage spatial avec émetteur unique sur canaux à large bande

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US20110310870A1 (en) * 2010-06-21 2011-12-22 Qualcomm Incorporated Hybrid time and frequency domain csi feedback scheme
US20140269883A1 (en) * 2013-03-15 2014-09-18 Steven C. Thompson Non-linear time domain channel estimation in ofdm systems
US20160087706A1 (en) * 2014-09-24 2016-03-24 Mediatek Inc. Synchronization in a Beamforming System

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Publication number Priority date Publication date Assignee Title
US20110310870A1 (en) * 2010-06-21 2011-12-22 Qualcomm Incorporated Hybrid time and frequency domain csi feedback scheme
US20140269883A1 (en) * 2013-03-15 2014-09-18 Steven C. Thompson Non-linear time domain channel estimation in ofdm systems
US20160087706A1 (en) * 2014-09-24 2016-03-24 Mediatek Inc. Synchronization in a Beamforming System

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* Cited by examiner, † Cited by third party
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WO2021074000A1 (fr) * 2019-10-11 2021-04-22 Fraunhofer-Gesellschaft zur Förderung der angewandten Forschung e.V. Multiplexage spatial avec émetteur unique sur canaux à large bande

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