WO2019078127A1 - Lighting circuit and vehicle lamp tool - Google Patents

Abstract

A lighting circuit 200 supplies electric power to a light source 110. A step-down converter 220 receives an output voltage Vout1 from a step-up converter 210, and supplies a driving current IDRV to the light source 110. The step-up converter 210 operates in a burst mode where an operation period and a stop period are repeated, in accordance with the state of the step-down converter 220.

Description

Lighting circuit and vehicle lamp

The present invention relates to a lighting circuit of a light source.

Vehicle lamps are generally capable of switching between low beam and high beam. The low beam illuminates the near side with a predetermined illuminance, and a light distribution rule is defined so as not to give glare to oncoming vehicles and preceding vehicles, and is mainly used when traveling in a city area. On the other hand, the high beam illuminates a wide range and a distance ahead with relatively high illuminance, and is mainly used when traveling at high speed on a road where there are few oncoming vehicles and preceding vehicles. Therefore, although the high beam is more excellent in visibility by the driver than the low beam, there is a problem that glare is given to the driver of the vehicle existing in front of the vehicle and the pedestrian.

In recent years, ADB (Adaptive Driving Beam: light distribution variable headlamp) technology has been proposed that dynamically and adaptively controls a light distribution pattern of high beams based on the state of the surroundings of a vehicle. ADB technology reduces the glare to be given to a vehicle or pedestrian by detecting the presence or absence of a preceding vehicle in front of the vehicle, an oncoming vehicle or a pedestrian, and reducing or turning off the area corresponding to the vehicle or pedestrian. It is a thing.

FIG. 1 is a circuit diagram of a vehicular lamp. The vehicular lamp 100R includes a light source 110 and a lighting circuit 200 thereof. The lighting circuit 200 uses the battery voltage V _BAT supplied from the battery 2 as a power supply, and supplies the light source 110 with a driving current according to the target luminance.

The light source 110 includes a plurality of light emitting units 112 connected in series. Assuming that the forward voltage per stage of the light emitting unit 112 is Vf and the number of steps of the light emitting unit is N, the output voltage (load voltage) V _LOAD of the lighting circuit 200 is
V _LOAD > Vf × N
Must be satisfied.

The topology of lighting circuit 200 is selected based on the relationship between battery voltage V _BAT and output voltage V _LOAD . When N ≦ 2, the maximum value of V _LOAD is lower than the battery voltage V _BAT , so the lighting circuit 200 can be configured with a step-down converter.

On the contrary, when _NN3 , the maximum value of the output voltage V _LOAD is higher than the battery voltage V _BAT . Therefore, lighting circuit 200 must be configured with a boost converter or a combination of a boost converter and a buck converter. The lighting circuit 200R includes a boost converter 210 at a front stage, a step-down converter 220 at a rear stage, and controllers 230 and 240 thereof.

The output voltage V _OUT1 of the step-up converter 210 is stabilized at the target voltage V _{OUT1 (REF)} , and this target voltage is defined so as to satisfy V _{OUT1 (REF)} > Vf × N. Controller 230 performs constant voltage control on boost converter 210 such that output voltage V _OUT1 approaches target value V _{OUT1 (REF)} .

The downstream step-down converter 220 receives the stabilized voltage V _OUT1 as an input voltage, and supplies a drive current I _DRV to the light source 110. Controller 240 performs constant current control of step-down converter 220 such that drive current I _DRV approaches a target value.

JP, 2014-176169, A

As a result of examining the lighting circuit 200R of FIG. 1, the present inventors came to recognize the following problems.
For example, although the typical value of Vf is about 3 V, V _{OUT1 (REF)} is defined as equation (1) in consideration of the variation and temperature characteristics.
V _{OUT1 (REF)} = Vf _(MAX) × N (1)
Vf _(MAX) is the maximum value of Vf in consideration of variations, temperature characteristics, and the like. For example, if N = 12 and Vf _(MAX) = 5V, then V _{OUT1 (REF)} = 60V.

When Vf _(MAX) is increased in consideration of the margin, the step-up ratio K ₁ = V _OUT / V _IN of the step-up converter 210 is increased, and the step-down ratio K ₂ = V _OUT / V _IN of the step-down converter 220 is decreased.

This causes problems such as upsizing of parts of the step-up converter 210 and the step-down converter 220, an increase in cost, and an increase in the amount of heat generation. An increase in the amount of heat generation causes another problem of an increase in the cost (such as a huge heat sink or a cooling fan) required for heat dissipation measures.

In addition, in the ADB controlled lamp, bypass control may be performed in order to turn on / off the individual light emitting units 112 independently, and a bypass switch SW is provided in parallel with each light emitting unit 112. When a certain bypass switch SW is turned on, the current flowing to the light emitting unit 112 in parallel with it is diverted to the bypass switch SW so that it is turned off. The bypass control causes the load voltage V _LOAD across the light source 110 to dynamically fluctuate. When the number of light emitting units 112 in the lighting state is n (0 ≦ n ≦ N) at a certain time,
V _LOAD = Vf × n
It is.

Assuming that Vf _(MAX) = 5 V and N = 12, V _{OUT1 (REF)} = 60 V. The actual load voltage V _LOAD when n = 1 is Vf = 3V. Therefore, the step-down ratio K ₂ of the subsequent step-down converter 220 becomes very small, ie, K ₂ = 3/60 = 1/20, and the size of the inductor of the step-down converter increases.

The present invention has been made in view of such problems, and one of the exemplary objects of an aspect thereof is to provide a converter that can be miniaturized.

One aspect of the present invention relates to a lighting circuit for supplying power to a light source. The lighting circuit includes a boost converter, and a step-down converter that receives an output voltage of the boost converter and supplies a drive current to a light source. The boost converter operates in a burst mode (intermittent mode) in which an operation period (operation state) and a stop period (stop state) are repeated according to the state of the step-down converter.

According to this aspect, the output voltage of the upstream step-up converter is adjusted such that the relationship between the input voltage and the output voltage of the downstream step-down converter becomes appropriate, so that the range of step-down ratio of the step-down converter can be limited.

The boost converter may enter an operating period when the potential difference between the input and output of the buck converter decreases to the first threshold.

The step-up converter may enter the stop period when the potential difference between the input and output of the step-down converter reaches a second threshold higher than the first threshold.

The operation period of the boost converter may be defined by a timer. That is, after entering an operation period, when a certain time passes, it may transition to a stop period.

The lighting circuit further includes a voltage detection circuit that generates a detection signal according to the potential difference between the input and output of the step-down converter, and a hysteresis comparator that compares the detection signal with a threshold and generates a burst signal according to the comparison result. You may have. The boost converter may be controlled in response to the burst signal.

The lighting circuit compares the detection signal with a predetermined threshold value according to the comparison result, and a voltage detection circuit that generates a detection signal in which the potential difference between the input and output of the step-down converter is multiplied by a coefficient switchable by two values. And a comparator that generates a burst signal. The coefficients change in response to the burst signal, and the boost converter may be controlled in response to the burst signal.

Another aspect of the present invention relates to a vehicle lamp. The vehicular lamp includes a light source and any of the lighting circuits described above.

It is to be noted that any combination of the above-described constituent elements, and one in which the constituent elements and expressions of the present invention are mutually replaced among methods, apparatuses, systems, etc. is also effective as an aspect of the present invention.

Furthermore, the description of this item (Means for Solving the Problems) does not explain all the essential features of the present invention, and therefore, the subcombinations of these described features may also be the present invention. .

According to an aspect of the present invention, the size of the converter can be reduced.

It is a circuit diagram of a vehicle lamp. FIG. 1 is a circuit diagram of a vehicle lamp according to a first embodiment. FIG. 6 is an operation waveform diagram of a lighting circuit. FIG. 6 is an operation waveform diagram of a lighting circuit. It is a circuit diagram of a part of lighting circuit concerning one example. It is a circuit diagram of a burst controller concerning one example. FIGS. 7A to 7C are circuit diagrams showing configuration examples of the voltage detection circuit. It is a circuit diagram of a burst controller concerning one example. FIGS. 9A and 9B are circuit diagrams showing configuration examples of the burst controller. FIGS. 10A and 10B are circuit diagrams showing configuration examples of the boost converter and the converter controller. It is a circuit diagram of a part of lighting circuit concerning a 2nd embodiment. It is a circuit diagram of a burst controller concerning one example. FIG. 13 is an operation waveform diagram of the burst controller of FIG. 12; FIG. 16 is a circuit diagram of a boost converter and a converter controller according to a fourth modification.

Hereinafter, the present invention will be described based on preferred embodiments with reference to the drawings. The same or equivalent components, members, and processes shown in the drawings are denoted by the same reference numerals, and duplicating descriptions will be omitted as appropriate. In addition, the embodiments do not limit the invention and are merely examples, and all the features and combinations thereof described in the embodiments are not necessarily essential to the invention.

In the present specification, “the state in which the member A is connected to the member B” means that the members A and B are electrically connected in addition to the case where the members A and B are physically and directly connected. It also includes the case of indirect connection via other members that do not substantially affect the connection state of the connection or do not impair the function or effect provided by the connection.

Similarly, "a state where the member C is provided between the member A and the member B" means that the member A and the member C, or the member B and the member C are directly connected, and It also includes the case of indirect connection via other members that do not substantially affect the connection state of the connection or do not impair the function or effect provided by the connection.

In the present specification, reference numerals attached to electric signals such as voltage signals and current signals or circuit elements such as resistors and capacitors indicate respective voltage values, current values, or resistance values and capacitance values as necessary. It shall represent.

The vertical and horizontal axes of the waveform diagrams and time charts referred to in the present specification are scaled up and down appropriately to facilitate understanding, and each waveform shown is also simplified for ease of understanding. Or exaggerated or emphasized.

First Embodiment
FIG. 2 is a circuit diagram of the vehicular lamp 100 according to the first embodiment. The entire lamp system 1 is shown in FIG. The vehicular lamp 100 includes a light source 110 and a lighting circuit 200. The light source 110 includes a plurality of light emitting units 112_1 to 112_N connected in series. The light emitting unit 112 is, for example, an LED, and the light source 110 is also referred to as an LED bar or an LED string. The light emitting unit 112 may be an LD (laser diode) or an organic EL (Electro Luminescence) element. For example, the plurality of light emitting units 112_1 to 112_N respectively illuminate different regions on the virtual vertical screen in front of the vehicle through an optical system (not shown).

The lighting circuit 200 supplies the drive current I _DRV to the light source 110 to cause it to emit light.

As an option, the lighting circuit 200 may have a function of individually turning on and off the plurality of light emitting units 112 (bypass control). The lighting circuit 200 receives a control command _SPTN instructing a light distribution pattern from a processor (ECU: Electronic Control Unit) (not shown), and turns on and off the plurality of light emitting units 112 according to the control command _SPTN. You may control.

The lighting circuit 200 includes a boost converter 210, a step-down converter 220, a converter controller 230, a converter controller 240, a burst controller 250, and a lamp ECU 270.

The lamp ECU 270 includes a main switch 272 and a processor 274. The processor 274 is communicable with the vehicle ECU 4 and controls on / off of the main switch 272 based on control commands and information from the vehicle ECU 4 or the converter controller 230, so that an appropriate light distribution pattern can be obtained. Control 240

When main switch 272 is turned on, battery voltage V _BAT is supplied to boost converter 210. Boost converter 210, a switching operation on the basis of the control pulses _{S 1} to the converter controller 230 generates, and generates an output voltage _{V OUT1} boosts the battery voltage _{V BAT.}

The control method of converter controller 230 is not particularly limited. Converter controller 230 is in the operating state of the boost converter 210, up converter 210, to provide a greater power than the power required by the subsequent step-down converter 220 and the light source 110, which generates control pulses S _1.

For example, converter controller 230 feeds back the duty ratio of control pulse S ₁ by feedback such that output voltage V _OUT1 approaches a prescribed target value V _{OUT1 (REF)} sufficiently higher than assumed load voltage V _LOAD. You may adjust it. Or the duty ratio of the control pulse S ₁ may be fixed to a certain value.

The step-down converter 220 steps down the output voltage V _OUT1 to supply the drive current I _DRV to the light source 110. Converter controller 240 in one embodiment, the drive current _{I DRV} generates a control pulse _{S 2} so as to approach the target value _{I REF,} the buck converter 220 may be feedback-controlled (constant current control). Converter controller 240 may use known techniques.

Boost converter 210 operates in a burst mode in which an operation period and a stop period are repeated according to the state of step-down converter 220. Burst controller 250 generates burst signal S _BURST defining on / off of boost converter 210 based on the relationship between input voltage V _IN2 (= V _OUT1 ) of step-down converter 220 and output voltage V _OUT2 . Converter controller 230 switches boost converter 210 when burst signal S _BURST is on level (eg, high), and stops switching of boost converter 210 when burst signal S _BURST is off level (eg, low). .

In one embodiment, burst controller 250 controls on / off of boost converter 210 based on the potential difference ΔV (= V _IN2 −V _OUT2 ) between the input and output of step-down converter 220.

When the potential difference ΔV between the input and output of step-down converter 220 decreases to first threshold value V _TH1 , burst controller 250 sets burst signal S _BURST to the on level, and sets boost converter 210 to the operating state.

When the potential difference ΔV between the input and output of step-down converter 220 rises to second threshold value V _TH2 defined higher than first threshold value V _TH1 , burst controller 250 sets burst signal S _BURST to the off level, and boost converter 210 Is set to stop.

The above is the configuration of the lighting circuit 200. Subsequently, the operation will be described. FIG. 3 is an operation waveform diagram of the lighting circuit 200. As shown in FIG. First, in order to facilitate understanding, the case where the load voltage V _LOAD is constant will be described.

The output power of the boost converter 210 is larger than the input power of the step-down converter 220 in the subsequent stage in the on period T _ON in which the burst signal S _BURST is at a high level. Therefore, the output voltage V _OUT1 rises with time. At time _{t 1,} the potential difference ΔV reaches the second threshold value _{V TH2,} the output voltage _{V OUT1} other words reaches _V LOAD _{+ V TH2,} the burst signal _{S BURST} goes low, enters the OFF period _{T OFF.}

During the off period _{T OFF,} the control pulse _{S 1} is stopped, the switching operation of the boost converter 210 is stopped, the output voltage _{V OUT1} is lowered with time. And time _{t 2, the} the potential difference ΔV is decreased to a first threshold value _{V TH1,} the output voltage _{V OUT1} other words decreases to _V LOAD _{+ V TH1,} the burst signal _{S BURST} becomes high level, and returns to the on-period _{T ON} .

In this way, the step-up converter 210 at the front stage operates in the burst mode in which operation and stop are repeated according to the potential difference between the input and output of the step-down converter 220 at the rear stage.

Subsequently, with reference to FIG. 4, an operation in the case where the load voltage V _LOAD fluctuates will be described. FIG. 4 is an operation waveform diagram of the lighting circuit 200. As shown in FIG. Even when the load voltage V _LOAD fluctuates, the operation is similar to FIG. By the step-up operation of boost converter 210, output voltage _V.sub.OUT2 follows load voltage _V.sub.LOAD .

The above is the operation of the lighting circuit 200. Next, the advantages will be described.

According to this lighting circuit 200, the potential difference ΔV between the input and output of the step-down converter 220 can be limited to a predetermined range. That it is possible to prevent that the minimum value of the step-down ratio K ₂ of the step-down converter 220 becomes too small, it can be designed small buck converter 220.

This effect is useful even in applications where the load voltage V _LOAD is constant, but is particularly effective in applications where the load voltage V _LOAD fluctuates, for example, an application that performs dimming based on bypass control.

In the following, the first embodiment of the present invention is described not to narrow the scope but to help understand the nature and operation of the first embodiment and to explain more specific configuration examples and examples in order to clarify them. Do.

Example 1.1
FIG. 5 is a circuit diagram of a part of the lighting circuit 200 according to an embodiment. In FIG. 5, a step-down converter 220, a converter controller 240 and a burst controller 250 are shown.

Step-down converter 220 includes a converter unit 222 and a current smoothing filter 224. In this embodiment, the converter controller 240, to stabilize the coil current I _L of the converter circuit 222 by a so-called ripple control. The coil current I _L is detected by the current sense resistor R _S. Converter controller 240 turns off the switching transistor M ₁ reaches the peak threshold is detected values of the coil current I _L, the switching transistor M ₁ drops to bottom a certain threshold detected value of the coil current I _L Turn on.

The current smoothing filter 224 removes the ripple component from the coil current I _L and supplies the DC component to the light source 110 as the drive current I _DRV .

The control method of converter controller 240 is not limited to this, and constant current control using an error amplifier may be performed. In this case, current smoothing filter 224 may be omitted.

Subsequently, the configuration of the burst controller 250 will be described. The burst controller 250 includes a voltage detection circuit 252 and a hysteresis comparator 254. Voltage detection circuit 252 generates a detection signal V _S according to the potential difference ΔV (= V _IN2 −V _LOAD ) of the input and output of step-down converter 220. The hysteresis comparator 254 compares the detection signal V _S with the threshold value V _THH · V _THL that changes with two values, and generates a burst signal S _BURST according to the comparison result. The lower threshold V _THL defines the first threshold V _TH1 of FIGS. 3 and 4, and the upper threshold V _THH defines the second threshold V _TH2 of FIGS. 3 and 4. . Instead of the hysteresis comparator, two comparators may be used. Boost converter 210 at the front stage is controlled in accordance with burst signal S _BURST .

FIG. 6 is a circuit diagram of a burst controller 250 according to one embodiment. The voltage detection circuit 252 can be configured by a differential amplifier including the resistors R ₁₁ to R ₁₄ and the operational amplifier OA ₁ . Hysteresis comparator 254 may be configured with resistors R21 ~ R23 and the operational amplifier (voltage comparator) OA _2.

According to the voltage detection circuit 252 of FIG. 6, the potential difference ΔV can be detected with high accuracy using the operational amplifier.

7 (a) to 7 (c) are circuit diagrams showing configuration examples of the voltage detection circuit 252. FIG. The voltage detection circuit 252 of FIG. 7A includes resistors R ₅₁ and R ₅₂ (both having a resistance value of R), and transistors Tr ₅₁ and Tr ₅₂ . V1 represents an input voltage V _IN2 and V2 represents a load voltage V _LOAD . The transistors Tr ₅₁ and Tr ₅₂ form a current mirror circuit, and a current (V 2 −V _BE ) / R flows through the transistor Tr ₅₁ and the resistor R ₅₁ . V _BE is a voltage between the base and the emitter of the bipolar transistor and is substantially constant. This current is copied by the current mirror circuit, and the same current flows through the resistor R ₅₂ , and the voltage drop is V 2 −V _BE . Therefore, V _S = V 1 −V 2 + V _BE is obtained as the detection signal V _S , which corresponds to the difference between the two voltages V 1 and V 2.

In FIG. 7B, an emitter follower circuit including a transistor Tr ₅₃ and a resistor R ₅₃ is added to the configuration of FIG. 7A. The emitter follower circuit shifts the detection voltage V _S downward to V _BE to obtain V _S = V 1 −V 2. That is, the influence of variations and fluctuations in the voltage V _BE can be suppressed.

In FIG. 7C, voltage dividing resistors R ₅₄ and R ₅₅ are provided in place of the resistor R ₅₃ in FIG. 7B.
V _S = (V 1 −V 2) × R ₅₅ / (R ₅₄ + R ₅₅ )

According to the voltage detection circuit 252 of FIGS. 7A to 7C, although the detection accuracy is lower than that using the operational amplifier, the voltage detection range can be expanded. In particular, in an application in which the load voltage V _LOAD fluctuates over a wide range, the configuration of FIG. 6 may be difficult to adopt due to the limitation of the input range of the operational amplifier. In this case, the configurations of FIGS. 7 (a) to 7 (c) are effective.

Example 1.2
FIG. 8 is a circuit diagram of a burst controller 250 according to one embodiment. The burst controller 250 includes a voltage detection circuit 256 and a comparator 258. The voltage detection circuit 256 generates a detection signal V _S obtained by multiplying the potential difference ΔV at the input and output of the step-down converter 220 by a coefficient (gain) that can be switched by two values. The comparator 258 compares the detection signal V _S with a predetermined threshold value V _TH and generates a burst signal S _BURST according to the comparison result.

FIGS. 9A and 9B are circuit diagrams showing configuration examples of the burst controller 250. FIG. 9 (a), the voltage detection circuit 256 includes a variable voltage dividing circuit including a variable resistor _{R 40} and the fixed resistor _{R 41.} The resistance value of the variable resistor R ₄₀ changes in two values in accordance with the comparison result (S _BURST ). When the voltage drop of the variable resistor R ₄₀ is taken as the detection voltage V _S , the detection voltage V _S is proportional to the potential difference ΔV = V 1 −V 2. The comparator 258 compares the detection signal V _S with the threshold value V _TH .

FIG. 9 (b) shows a specific configuration example of the burst controller 250 of FIG. 9 (a). The resistor R ₄₀ includes resistors R ₄₂ and R ₄₃ and a transistor Tr ₄₃ . When the transistor Tr ₄₃ is off, the resistance of the resistor R ₄₀ is equal to R _42, and when the transistor Tr ₄₃ is on, the resistance of the resistor R ₄₀ is a parallel connection of the resistors R ₄₂ and R ₄₃ .

The transistor Tr ₄₁ is a voltage comparison unit. A detection voltage V _S which is a voltage drop of the resistor R ₄₀ is applied between the base and the emitter of the transistor Tr ₄₁ . The on / off state of the transistor Tr ₄₁ corresponds to the comparison result of the detection voltage V _S and the voltage between the base and the emitter. The on / off state of the transistor Tr ₄₁ is converted into a binary burst signal S _BURST by an output stage (inverter) including the transistor Tr ₄₄ . The transistor Tr ₄₂ and the resistors R ₄₅ and R ₄₆ control the on / off of the transistor Tr ₄₃ based on the on / off state of the transistor Tr ₄₁ .

In this example, although the resistor R ₄₀ is a variable resistor, the resistor R ₄₁ may be a variable resistor. Further, although the transistor Tr ₄₁ is used as the voltage comparison means, the present invention is not limited to this, and a voltage comparator including a differential amplifier may be used.

Subsequently, boost converter 210 will be described.
FIGS. 10A and 10B are circuit diagrams showing configuration examples of the boost converter 210 and the converter controller 230. FIG. A converter controller 230 may use a commercially available controller IC. A voltage obtained by adding a margin α to the maximum value V _{LOAD (MAX)} of the load voltage V _LOAD is specified as the target voltage V _{OUT1 (REF)} of the output voltage V _OUT1 , and the converter controller 230 performs feedback control of the boost converter 210 It is also good. The feedback control by converter controller 230 functions as a limiter of output voltage V _OUT1 . Converter controller 230 has a pulse-by-pulse current limiting function. Specifically, the current flowing through the switching transistor M ₂ is detected by the sense resistor R _CS. In each switching cycle, when the current detection signal based on the voltage drop of sense resistor R _CS exceeds the threshold value of the over current protection, converter controller 230 immediately brings control pulse S ₁ low.

During the operation period, as shown in FIG. 3 and FIG. 4, the output voltage V _OUT1 of the boost converter 210 is lower than the target voltage V _{OUT1 (REF) of} feedback control. Prior to the end of the on time (pulse width) required to keep the output voltage V _OUT at V _{OUT1 (REF)} , the pulse-by-pulse current limit is activated. In other words, the pulse-by-cycle current limit defines the available power of step-up converter 210, which is designed to exceed the input power of step-down converter 220.

In place of the pulse-by-pulse current limit, the maximum on-duty (maximum on-time) may be limited to define the available power during the operation period.

In the boost converter 210 in FIG. 10 (a), the control pulses _{S 1} supplied to the gate of the switching transistor _{M 2} is masked by the burst signal _{S BURST.} The logic gate 232, when the burst signal _{S BURST} indicates the operating state, is passed through the control pulses _{S 1} to the gate of the switching transistor _{M 2,} when the burst signal _{S BURST} indicates a stop state, the low gate of the switching transistor _{M 2} Fix to

In FIG. 10B, converter controller 230 is provided with an enable (EN) pin. Converter controller 230 is configured to stop the switching operation when a predetermined level (for example, low) is input to the EN pin. In this case, the burst signal S _BURST may be input to the EN pin.

Second Embodiment
FIG. 11 is a circuit diagram of a part of the lighting circuit 200 according to the second embodiment. Step-down converter 220 and burst controller 250 are shown in FIG.

In this embodiment, the operation period of boost converter 210 is defined by a timer. The burst controller 250 includes a voltage detection circuit 260, a comparator 262, and a timer 264. Voltage detection circuit 260 generates detection signal V _S according to the potential difference ΔV between the input and output of step-down converter 220. The configuration of the voltage detection circuit 260 may be the same as that described above.

The comparator 262 compares the voltage detection signal V _S with the threshold voltage V _THL . Then, it generates a trigger signal TRIG that is asserted (for example, high) when the voltage detection signal V _S falls to the threshold value V _THL . The timer 264 generates a burst signal S _BURST that has a predetermined level for a predetermined time from the assertion of the trigger signal TRIG.

FIG. 12 is a circuit diagram of a burst controller 250 according to an embodiment. The voltage detection circuit 260 includes resistors R ₉₀ and R ₉₁ , and is configured in the same manner as the voltage detection circuit 256 of FIG. 9A. Of course, the voltage detection circuit 260 may be configured in the same manner as the voltage detection circuit 252 shown in FIGS. 6 and 7A to 7C.

The comparator 262 includes transistors Tr ₉₁ and Tr ₉₂ and resistors R ₉₂ , R ₉₃ and R ₉₄ , and is configured in the same manner as the comparator 258 in FIG. 9 (b). The comparator 262 may be configured by a voltage comparator.

The timer 264 includes a transistor Tr ₉₃ , a capacitor C ₉₁ , and a transistor Tr ₉₄ . When the trigger signal TRIG becomes high level, the transistor Tr ₉₃ is turned on, the capacitor C ₉₁ is discharged, the capacitor voltage VC ₉₁ becomes zero, and the transistor Tr ₉₄ is turned on. When the trigger signal TRIG becomes low level, the transistor Tr ₉₃ is turned off, the capacitor C ₉₁ is charged through the resistors R ₉₅ and R ₉₆ , and the capacitor voltage VC ₉₁ rises. As the capacitor voltage V _C91 rises, the potential at the connection node of the resistors R ₉₅ and R ₉₆ also rises, and eventually the transistor Tr ₉₄ is turned off. The burst signal S _BURST is associated with the on / off state of the transistor Tr ₉₄ , and indicates a stop / low operation when it is high. The off period of the transistor Tr ₉₄ is considered to be the operation period of the boost converter 210. FIG. 13 is an operation waveform diagram of the burst controller 250 of FIG.

The timer 264 may be configured by a one-shot multivibrator.

The present invention has been described above based on the embodiments. It is understood by those skilled in the art that this embodiment is an exemplification, and that various modifications can be made to the combination of each component and each processing process, and such a modification is also within the scope of the present invention. is there. Hereinafter, such modifications will be described.

(Modification 1)
In one embodiment, step-down converter 220 includes a constant current driver connected in series with light source 110 to stabilize drive current I _DRV by the constant current source, and converter controller 240 connects the constant current driver and light source 110 in series. As the load, in which case the voltage across them is the load voltage V _LOAD .

(Modification 2)
In the embodiment, the burst operation of boost converter 210 is controlled based on the potential difference between the input and output of step-down converter 220, but the invention is not limited thereto. From another point of view, burst controller 250 may generate burst signal S _BURST so that the step-down ratio of step-down converter 220 does not fall below a predetermined value (within a predetermined range).

(Modification 3)
Although the embodiment has described the fluctuation of the load voltage V _LOAD due to the bypass control method, the factor of the fluctuation of the load voltage V _LOAD is not particularly limited.

(Modification 4)
FIG. 14 is a circuit diagram of a boost converter and a converter controller according to a fourth modification. Between the gate of the PWM output pin and the switching transistor _{M 2} of the converter controller 230, PMOS transistor 234 is provided. The gate of the PMOS transistor 234 is pulled up to the power supply pin through a resistor _{R 80.} The transistor 236 is provided between the gate of the switch 234 and the ground. A resistor R ₈₀ may be provided between the gate and the source (ie, the PWM pin) of the PMOS transistor 234. The drain of the PMOS transistor 234 is connected to the gate of the switching transistor _{M 2} via a voltage divider circuit including resistors _{R 81, R 82.}

When the burst signal S _BURST is high, the transistor 236 is turned on and the gate of the PMOS transistor 234 is low. In this state, when the control pulse _{S 1} is high, PMOS transistor 234 is turned on, the gate of the switching transistor _{M 2} also becomes high. When the control pulse _{S 1} is low, PMOS transistor 234 is turned off to discharge the gate capacitance of the switching transistor M2 through the body diode 238 of the PMOS transistor 234, the switching transistor _{M 2} is turned on. Thus, while the burst signal S _BURST is high, the switching transistor M ₂ can be switched according to the control pulse S ₁ .

When the burst signal S _BURST is low, transistor 236 is off and the gate of PMOS transistor 234 is pulled up through resistor R80. In this state, regardless of the high / low control pulses S ₁ of the PWM pin, the gate of the switching transistor M ₂ becomes low, the switching transistor M2 is kept turned off.

While the present invention has been described using specific terms based on the embodiments, the embodiments merely show the principles and applications of the present invention, and the embodiments are defined in the claims. Many variations and modifications of the arrangement can be made without departing from the concept of the present invention.

100: vehicle lamp, 110: light source, 112: light emitting unit, 200: lighting circuit, 210: boost converter, 220: step-down converter, 222: converter unit, 224: current smoothing filter, 230, 240: converter controller, 250: Burst controller, 252: voltage detection circuit, 254: hysteresis comparator, 256: voltage detection circuit, 258: comparator, 260: voltage detection circuit, 262: comparator, 264: timer, 270: lamp ECU, 272: main switch, 274: Processor.

The present invention relates to a lighting circuit of a light source.

Claims (8)

1. A lighting circuit for supplying power to the light source;
   A boost converter,
   A step-down converter that receives an output voltage of the step-up converter and supplies a drive current to the light source;
   Equipped with
   The lighting circuit characterized in that the boost converter operates in a burst mode in which an operation period and a stop period are repeated according to the state of the step-down converter.
2. The lighting circuit according to claim 1, wherein the step-up converter enters an operation period when the potential difference between the input and the output of the step-down converter decreases to a first threshold value.
3. The lighting circuit according to claim 2, wherein the step-up converter enters a stop period when the potential difference between the input and the output of the step-down converter reaches a second threshold value higher than the first threshold value.
4. The lighting circuit according to claim 1, wherein the operation period of the boosting converter is defined by a timer.
5. A voltage detection circuit that generates a detection signal according to the potential difference between the input and output of the step-down converter;
   A hysteresis comparator that compares the detection signal with a threshold and generates a burst signal according to the comparison result;
   The lighting circuit according to any one of claims 1 to 3, further comprising: the boost converter is controlled according to the burst signal.
6. A voltage detection circuit that generates a detection signal obtained by multiplying the potential difference between the input and output of the step-down converter by a coefficient that can be switched by two values;
   A comparator that compares the detection signal with a predetermined threshold and generates a burst signal according to the comparison result;
   The lighting circuit according to claim 1, further comprising: the coefficient changing in response to the burst signal, and the boost converter being controlled in response to the burst signal.
7. The lighting circuit according to claim 6, wherein the voltage detection circuit includes a variable voltage dividing circuit that divides a potential difference between the input and the output of the step-down converter.
8. Light source,
   A lighting circuit according to any one of claims 1 to 7;
   A vehicle lamp comprising:

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