WO2019078013A1

WO2019078013A1 - Control device

Info

Publication number: WO2019078013A1
Application number: PCT/JP2018/037158
Authority: WO
Inventors: 信太朗田中; 久保　謙二; 大内　貴之; 裕二曽部; 高橋　直也
Original assignee: 日立オートモティブシステムズ株式会社
Priority date: 2017-10-17
Filing date: 2018-10-04
Publication date: 2019-04-25
Also published as: JP6913599B2; JP2019075913A; CN111201701A; US11101739B2; DE112018004544B4; US20210135578A1; DE112018004544T5; CN111201701B

Abstract

The present invention sufficiently reduces the switching loss of a power conversion device. A DC-DC converter 100 has: a switching circuit 10 that converts inputted first direct current power into alternating current power; a transformer 20 that performs voltage conversion with respect to the alternating current power; and an output circuit 30 that converts the alternating current power into second direct current power, said alternating current power having been subjected to the voltage conversion by means of the transformer 20. A control circuit 50 that controls the DC-DC converter 100 calculates a magnetic flux density value B of the transformer 20, and controls the driving frequency of the switching circuit 10 on the basis of the magnetic flux density value B thus calculated.

Description

Control device

The present invention relates to a control device used to control a power conversion device.

BACKGROUND ART In recent years, with the background of exhaustion of fossil fuels and global environmental problems, interest in vehicles traveling using electrical energy, such as hybrid vehicles and electric vehicles, has increased and has been put to practical use. An automobile that travels using such electrical energy is equipped with a high voltage battery that supplies power to a motor for driving the wheels. Furthermore, a power conversion device is provided which steps down the output power from the high-voltage battery and supplies necessary power to low-voltage electric devices mounted on a car, such as an air conditioner, audio, various ECUs (Electronic Control Units), etc. There is also. Such a power conversion device converts input DC power into DC power of different voltages, and is also called a DC-DC converter.

In general, a DC-DC converter includes a switching circuit capable of switching operation, and performs voltage conversion of DC power by controlling on / off of the switching circuit. Specifically, the input DC power is temporarily converted into AC power using a switching circuit, and the AC power is transformed (boosted or stepped down) using a transformer. Then, the AC power after transformation is converted again into DC power using an output circuit such as a rectifier circuit. Thereby, a DC output having a voltage different from the input voltage can be obtained. The switching circuit is configured using, for example, a semiconductor switch element such as a MOSFET (Metal-Oxide-Semiconductor Field-Effect Transistor) or an IGBT (Insulated Gate Bipolar Transistor).

In a vehicle-mounted power converter, high efficiency is generally required for the purpose of effective use of natural energy and reduction of carbon dioxide. Therefore, it is important to reduce the loss during power conversion as much as possible. Here, the loss generated in the DC-DC converter includes a switching loss generated by a switching operation, a resistance loss (copper loss) generated in a transformer or a semiconductor switch element, and the like. As means for reducing the number of switching elements, for example, the following Patent Document 1 is known. The power converter disclosed in Patent Document 1 monitors the transformer current flowing into the isolation transformer, and raises the switching carrier frequency when the transformer current exceeds a current reference value set in consideration of magnetic saturation. Implement control. This reduces the switching frequency and reduces the switching loss.

JP 2008-278723 A

In the technology described in Patent Document 1, since the control of the switching carrier frequency is performed using the monitoring result of the transformer current, a control delay occurs when, for example, the change of the transformer current is abrupt, and as a result, switching Loss reduction may be inadequate.

The control device according to the present invention controls a power conversion device that converts input first direct current power into second direct current power and outputs the second direct current power, and the power conversion device includes the first direct current. A switching circuit that converts power into alternating current power, a transformer that performs voltage conversion of the alternating current power, and an output circuit that converts the alternating current power voltage-converted by the transformer into the second direct current power; The control device calculates a magnetic flux density value of the transformer, and controls a drive frequency of the switching circuit based on the calculated magnetic flux density value.

According to the present invention, the switching loss of the power converter can be sufficiently reduced.

It is a figure showing composition of a vehicle power supply concerning one embodiment of the present invention. FIG. 1 is a diagram showing a basic circuit configuration of a DC-DC converter according to a first embodiment of the present invention. It is a figure showing composition of a control circuit concerning a 1st embodiment of the present invention. It is a figure explaining operation of a comparator. It is a control flowchart of a control circuit. FIG. 7 is a diagram showing a basic circuit configuration of a DC-DC converter according to a second embodiment of the present invention. It is a figure which shows an example of the relationship between transformer temperature and magnetic flux density command value. FIG. 7 is a diagram showing a basic circuit configuration of a DC-DC converter according to a third embodiment of the present invention. It is a figure which shows an example of the relationship between an output current and magnetic flux density command value. It is a figure which shows the structure of the control circuit which concerns on the 4th Embodiment of this invention. It is a figure which shows an example of the adjustment method of PI control gain by a gain adjustment part. It is a figure showing composition of a control circuit concerning a 5th embodiment of the present invention.

Hereinafter, embodiments of a power conversion device according to the present invention will be described with reference to the drawings. In the drawings, the same elements will be denoted by the same reference symbols and redundant description will be omitted. However, the present invention is not limited to the following embodiments, and includes various modifications and applications within the technical concept of the present invention.

-First Embodiment-
(Vehicle power supply configuration)
FIG. 1 is a view showing a configuration of a vehicle power supply according to an embodiment of the present invention. As shown in FIG. 1, the vehicle power supply according to the present embodiment is mounted on a vehicle 1000, and is a power supply that performs mutual power conversion between high voltage battery V1 and low voltage battery V2 using DC-DC converter 100. It is a lineage. In the following description, the low voltage side of DC-DC converter 100, that is, the side connected to low voltage battery V2 is referred to as "L side", and is connected to the high voltage side of DC-DC converter 100, ie, high voltage battery V1. This side is called "H side".

One end of the low voltage battery V2 is connected to one end on the L side of the DC-DC converter 100, and the other end of the low voltage battery V2 is connected to the other end on the L side of the DC-DC converter 100. One end of the auxiliary device 400 such as an air conditioner is connected to one end of the L side of the DC-DC converter 100 and one end of the low voltage battery V2, and the other end of the auxiliary device 400 is the other side of the DC side of the DC-DC converter 100 It is connected to the end and the other end of the low voltage battery V2. One end of the HV device 300 is connected to one end of the DC side of the DC-DC converter 100 and one end of the high voltage battery V1, and the other end of the HV device 300 is the other end of the DC side of the DC to DC converter 100 and high voltage It is connected to the other end of the battery V1. One end of the high voltage battery V1 is connected to one end of the DC-DC converter 100 on the H side, and the other end of the high voltage battery V1 is connected to the other end of the DC-DC converter 100 on the H side.

The DC-DC converter 100, the HV system device 300 and the auxiliary device 400 are connected to the vehicle power control unit 200. Vehicle power supply control unit 200 controls the operation of each of these devices, the power transmission direction of the power exchanged between each of these devices and high voltage battery V1 and low voltage battery V2, the amount of power, and the like.

(Basic configuration of DC-DC converter 100)
FIG. 2 is a diagram showing a basic circuit configuration of the DC-DC converter 100 according to the first embodiment of the present invention. As shown in FIG. 2, the DC-DC converter 100 according to the present embodiment includes the switching circuit 10, the transformer 20, and the output circuit 30, and is connected to the control circuit 50 via the gate driver 90.

The switching circuit 10 is connected to the high voltage battery V1 via the positive electrode input terminal 1 and the negative electrode input terminal 2. The switching circuit 10 includes switch elements 11a to 14a connected in a bridge connection, and converts the DC power input from the high voltage battery V1 into high frequency AC power by switching the switch elements 11a to 14a. Output to the primary side of the transformer 20.

The transformer 20 insulates between the primary side and the secondary side, performs voltage conversion of AC power between the primary side and the secondary side, and reduces (steps up) the AC power generated by the switching circuit 10 The alternating current power is output to the output circuit 30.

The output circuit 30 is connected to the low voltage battery V 2 via the positive electrode output terminal 3 and the negative electrode output terminal 4. The output circuit 30 includes

diodes

31 and 32, and rectifies AC power voltage-converted by the transformer 20 using the

diodes

31 and 32, converts the AC power into DC power, and outputs the DC power to the low voltage battery V2. .

Control circuit 50 is provided, for example, in vehicle power supply control unit 200 of FIG. 1, and generates and outputs output signals 51 to 54 for controlling the switching operations of switch elements 11a to 14a in switching circuit 10, respectively. .

The gate driver 90 converts the output signals 51 to 54 output from the control circuit 50 into drive signals 91 to 94 for driving the switch elements 11a to 14a, respectively, and outputs the drive signals 91 to 94 to the switching circuit 10. The gate driver 90 mounts an isolation transformer 90 a and isolates between the switching circuit 10 and the control circuit 50.

Hereinafter, details of each configuration of the switching circuit 10, the transformer 20, and the output circuit 30, which the DC-DC converter 100 has, and the control circuit 50 will be described.

(Switching circuit 10)
Switching circuit 10 converts DC power input from high voltage battery V1 through positive electrode input terminal 1 and negative electrode input terminal 2 into high frequency AC power according to control of control circuit 50, and provides a primary winding of transformer 20. It has a role of supplying N1. Between the positive electrode input terminal 1 and the negative electrode input terminal 2, a voltage detector 41 and a smoothing capacitor C1 are connected in parallel with the high voltage battery V1. The voltage detector 41 detects the voltage of the DC power input to the switching circuit 10, and outputs the detected value to the control circuit 50 as the input voltage Vin.

The switching circuit 10 has a configuration in which four switch elements 11a to 14a are full-bridge connected. That is, a series circuit of two

switch elements

11a and 12a (hereinafter referred to as "first leg") between the positive electrode input terminal 1 and the negative electrode input terminal 2, and two

switch elements

13a and 14a. Series circuits (hereinafter referred to as "second legs") are connected to one another. The connection point A between the switch element 11a and the switch element 12a in the first leg is connected to one end of the primary winding N1 of the transformer 20, and the connection between the switch element 13a and the switch element 14a in the second leg The point B is connected to the other end of the primary winding N1 of the transformer 20. The switch elements 11a to 14a can be configured using any element capable of switching operation, and for example, an FET (field effect transistor) or the like is preferable.

Diodes 11b to 14b for flywheel and capacitors 11c to 14c are connected in parallel to the switch elements 11a to 14a, respectively. The diodes 11b to 14b and the capacitors 11c to 14c may be configured separately from the switch elements 11a to 14a, or may be parasitic components of the switch elements 11a to 14a. Moreover, you may use these together.

In the DC-DC converter 100 of the present embodiment, as a control method of the switching circuit 10, a phase shift control method which is a driving method capable of reducing the switching loss is used. In the phase shift control method, among the four switch elements 11a to 14a constituting the full bridge type switching circuit 10, the switch element 11a located on the upper side of the first leg and the switch element 14a located on the lower side of the second leg Is controlled according to the output voltage of the DC-DC converter 100. Similarly, the on / off phase difference between the switch element 12a below the first leg and the switch element 13a above the second leg is also controlled according to the output voltage of the DC-DC converter 100. Thus, the period in which the switch element 11a and the switch element 14a are simultaneously turned on and the period in which the switch element 12a and the switch element 13a are simultaneously turned on are adjusted according to the output voltage. Here, the power transmitted from the switching circuit 10 (primary side of the transformer 20) to the output circuit 30 (secondary side of the transformer 20) is a period during which the switch element 11a and the switch element 14a are simultaneously turned on, and It depends on the period when the element 12a and the switch element 13a are simultaneously turned on. Therefore, by controlling the phase difference as described above, the output voltage of DC-DC converter 100 can be stabilized at a desired value. In the following description, it is assumed that the period in which the switch element 11a and the switch element 14a are simultaneously turned on and the period in which the switch element 12a and the switch element 13a are simultaneously turned on have the same length. Also, the ratio of the lengths of these periods in one cycle may be referred to as a duty ratio.

(Trans 20)
The transformer 20 has a role of performing voltage conversion on the AC power generated by the switching circuit 10 and outputting the AC power after the voltage conversion to the output circuit 30. The transformer 20 includes a primary winding N1 connected to the switching circuit 10 and a secondary winding N2 connected to the output circuit 30. The transformer 20 has a center tap configuration to realize a full wave rectification circuit in combination with the output circuit 30, and the secondary winding N2 is divided into two secondary windings N2a and N2b in the middle ing. The turns ratio (N1 / N2a or N1 / N2b) between primary winding N1 and secondary windings N2a and N2b is the voltage range of input voltage Vin applied between positive electrode input terminal 1 and negative electrode input terminal 2, and It is set according to the voltage range of the output voltage Vout to be supplied between the positive electrode output terminal 3 and the negative electrode output terminal 4.

The transformer 20 has a resonant inductance L1 in series with the primary winding N1. The resonant inductance L1 and the capacitance components of the capacitors 11c to 14c respectively connected in parallel to the switch elements 11a to 14a in the switching circuit 10 form a resonant circuit that reduces the switching loss generated in the switching circuit 10. . When the value of the resonant inductance L1 in the transformer 20 is small, the value of the inductance of the resonant circuit may be increased by connecting an inductor of another element in series with the resonant inductance L1.

One end of the primary winding N1 is connected to a connection point A, which is the middle point of the first leg in the switching circuit 10, via a resonant inductance L1. Further, the other end of the primary winding N1 is connected to a connection point B which is a middle point of the second leg in the switching circuit 10. A neutral point T, which is a connection point between the secondary winding N2a and the secondary winding N2b, is connected to the output circuit 30 along with both ends of the secondary winding N2.

(Output circuit 30)
Output circuit 30 smoothes and rectifies AC power appearing in secondary windings N2a and N2b in accordance with AC power flowing through primary winding N1 of transformer 20 to convert it into DC power, and outputs positive electrode output terminal 3 and negative electrode output It has a role of outputting to the low voltage battery V2 through the terminal 4. A voltage detector 42 is connected between the positive electrode output terminal 3 and the negative electrode output terminal 4 in parallel with the low voltage battery V2. The voltage detector 42 detects the voltage of the DC power output from the output circuit 30, and outputs the detected value to the control circuit 50 as the output voltage Vout.

The output circuit 30 includes two

diodes

31 and 32 whose anodes are connected to each other at a rectifying connection point S, a smoothing coil L2 and a capacitor C2. The diode 31 is connected between one end of the secondary winding N2a of the transformer 20 and the rectification node S, and the diode 32 is connected between one end of the secondary winding N2b of the transformer 20 and the rectification node S It is done. The smoothing coil L2 is connected between the neutral point T, which is the other end of the secondary windings N2a and N2b of the transformer 20, and the positive electrode output terminal 3, and the capacitor C2 includes the positive electrode output terminal 3 and the negative electrode output terminal 4 Connected between.

In the output circuit 30 having the above-described circuit configuration, the

diodes

31 and 32 constitute a rectifier circuit that rectifies AC power output from the secondary windings N2a and N2b of the transformer 20 and converts the AC power into DC power. The smoothing coil L2 and the capacitor C2 constitute a smoothing circuit that smoothes the rectified output generated at the neutral point T. The conduction loss may be further reduced by replacing the

diodes

31 and 32 with switch elements such as FETs to perform synchronous rectification operation according to a known technique.

(Control circuit 50)
The control circuit 50 is a circuit that controls the operation of the switch elements 11a to 14a of the switching circuit 10 so that the output voltage Vout of the DC-DC converter 100 becomes a predetermined voltage target value. As shown in FIG. 2, the control circuit 50 includes a voltage control unit 60, a switching carrier frequency setting unit 70, a signal generation unit 80, and a clock unit 65.

Voltage control unit 60 calculates a duty ratio when switching elements 11 a to 14 a are switched in switching circuit 10. In DC-DC converter 100, output voltage Vout of DC power output from output circuit 30 is controlled according to the value of the duty ratio.

The switching carrier frequency setting unit 70 sets a switching carrier frequency according to a drive frequency when the switching elements 11a to 14a perform switching operation in the switching circuit 10. In DC-DC converter 100, switch elements 11a-14a are driven at a drive frequency corresponding to the switching carrier frequency.

The signal generation unit 80 generates the output signals 51 to 54 based on the duty ratio calculated by the voltage control unit 60 and the switching carrier frequency set by the switching carrier frequency setting unit 70. The output signals 51 to 54 generated by the signal generation unit 80 are output from the control circuit 50 to the gate driver 90, and are converted into drive signals 91 to 94 in the gate driver 90. The drive signals 91 to 94 are input to the gate terminals of the switch elements 11a to 14a in the switching circuit 10, and the operation timings according to the duty ratio and the switching carrier frequency when the output signals 51 to 54 are generated The elements 11a to 14a are driven respectively. Thereby, the operation of the switching circuit 10 is controlled by the control circuit 50.

The clock unit 65 internally has two counters that are counted up for each fixed clock, and controls the execution timing of the voltage control unit 60 and the switching carrier frequency setting unit 70 based on the count values of these counters. Do.

FIG. 3 is a diagram showing the configuration of the control circuit 50 according to the first embodiment of the present invention. The details of the voltage control unit 60, the switching carrier frequency setting unit 70, the signal generation unit 80, and the clock unit 65 of the control circuit 50 in the present embodiment will be described below with reference to FIG.

(Voltage control unit 60)
The voltage control unit 60 includes a subtracting unit 61, a PI control unit 62, and a duty limiting unit 63. Subtraction unit 61 calculates a difference between voltage command value Vref, which is a target value of the output voltage set in advance, and output voltage Vout detected by voltage detector 42, and outputs the difference to PI control unit 62. The PI control unit 62 performs PI operation on the difference obtained by the subtraction unit 61 to perform PI control (proportional integration control) so that the difference approaches 0, and the switch element 11 a of the switching circuit 10 A duty value D, which is the value of the duty ratio for.

Duty limiting unit 63 sets a predetermined lower limit value and an upper limit value for duty value D determined by PI control unit 62, and duty instruction value D * limited within a range from the lower limit value to the upper limit value. Ask for The duty instruction value D * obtained by the duty limiting unit 63 is output from the voltage control unit 60 to the switching carrier frequency setting unit 70, the signal generation unit 80, and the clock unit 65.

Assuming that the upper limit value and the lower limit value set by duty limiting unit 63 are set as maximum duty value Dmax and minimum duty value Dmin, duty instruction value D * output from voltage control unit 60 is expressed by the following equation (1) Meet the relationship.
Dmin D D * D Dmax (1)

(Switching carrier frequency setting unit 70)
The switching carrier frequency setting unit 70 includes a multiplying unit 71, a proportional unit 72, a subtracting unit 73, a magnetic flux density command value setting unit 74, a PI control unit 75, and a frequency limiting unit 76. Multiplication unit 71 calculates a multiplication value of duty instruction value D * output from duty limit unit 63 of voltage control unit 60 and input voltage Vin detected by voltage detector 41, and outputs the product to proportional unit 72. . The proportional unit 72 converts the multiplication value into the magnetic flux density value B of the transformer 20 by multiplying the multiplication value obtained by the multiplication unit 71 by a predetermined proportional constant. Subtraction unit 73 sets a magnetic flux density command value Bref which is a target value of the magnetic flux density preset by magnetic flux density command value setting unit 74 based on the saturation magnetic flux density of transformer 20, and a magnetic flux density value obtained by proportional unit 72. The difference with B is calculated and output to PI control unit 75. The PI control unit 75 performs PI control on the difference obtained by the subtraction unit 73 to perform PI control (proportional integration control) such that the difference approaches 0, and the driving frequency of the switching circuit 10 is set to The switching carrier frequency f according to is determined.

The frequency limiting unit 76 sets a predetermined lower limit value and an upper limit value for the switching carrier frequency f determined by the PI control unit 75, and the switching carrier frequency setting is limited within the range from the lower limit value to the upper limit value. Find the value f *. Switching carrier frequency setting value f * obtained by frequency limiting unit 76 is output from switching carrier frequency setting unit 70 to signal generation unit 80 and clock unit 65, respectively.

Assuming that the switching carrier frequency setting value f * output from the switching carrier frequency setting unit 70 has the upper limit value and the lower limit value set in the frequency limiting unit 76 as the maximum switching carrier frequency fmax and the minimum switching carrier frequency fmin, respectively. The relationship of the following equation (2) is satisfied.
fmin ≦ f * ≦ fmax (2)

The switching carrier frequency setting unit 70 outputs the switching carrier frequency setting value f * obtained as described above to the signal generation unit 80. Thereby, when the magnetic flux density value B of the transformer 20 is smaller than the magnetic flux density command value Bref, the drive frequency of the switching circuit 10 is lowered, and conversely, when the magnetic flux density value B of the transformer 20 is larger than the magnetic flux density command value Bref Can set the switching carrier frequency when the signal generation unit 80 generates the output signals 51 to 54 so as to increase the drive frequency of the switching circuit 10.

(Clock section 65)
The clock unit 65 internally has an A counter and a B counter. The A counter is a counter used to control the execution timing of the voltage control unit 60, and is counted up every fixed clock. The B counter is a counter used to control the execution timing of the switching carrier frequency setting unit 70, and, like the A counter, counts up every fixed clock.

Duty instruction value D * input from voltage control unit 60 to clock unit 65 is stored in clock unit 65 as the previous duty instruction value Da *. If the value of the A counter is less than the predetermined threshold value, the clock unit 65 outputs the stored previous duty instruction value Da * to the signal generation unit 80. When the value of the A counter exceeds the predetermined threshold value, the clock unit 65 outputs an execution command to the voltage control unit 60, and causes the voltage control unit 60 to calculate the duty instruction value D *. Thus, the new duty instruction value D * is calculated by the voltage control unit 60 and input to the signal generation unit 80, and the previous duty instruction value Da * stored in the clock unit 65 is updated.

The switching carrier frequency setting value f * input from the switching carrier frequency setting unit 70 to the clock unit 65 is stored as the previous switching carrier frequency setting value fa * in the clock unit 65. If the value of the B counter is less than the predetermined threshold value, the clock unit 65 outputs the stored previous switching carrier frequency set value fa * to the signal generation unit 80. When the value of the B counter exceeds the predetermined threshold value, the clock unit 65 outputs an execution command to the switching carrier frequency setting unit 70, and causes the switching carrier frequency setting unit 70 to calculate the switching carrier frequency setting value f *. . Thereby, the new switching carrier frequency setting value f * is calculated by the switching carrier frequency setting unit 70 and input to the signal generating unit 80, and the previous switching carrier frequency setting value fa * stored in the clock unit 65 is It will be updated.

As described above, clock unit 65 outputs the previous duty instruction value Da * or the execution command to voltage control unit 60 according to the value of A counter, and the previous setting of the switching carrier frequency according to the value of B counter. The value fa * or an execution command to the switching carrier frequency setting unit 70 is output. Thereby, the execution timing of the voltage control unit 60 and the switching carrier frequency setting unit 70 can be controlled. The same value may be set as the threshold value of the A counter and the threshold value of the B counter, or different values may be set.

(Signal generation unit 80)
The signal generation unit 80 includes an operation determination unit 81, a dead time setting unit 82, a threshold setting unit 83, an operation determination unit 84, a carrier signal generation unit 85, and a comparator 86. When voltage control unit 60 calculates duty instruction value D *, operation determination unit 81 outputs input duty instruction value D * to threshold setting unit 83, and voltage control unit 60 outputs the duty instruction value. When the duty instruction value Da * is input last time from the clock unit 65 without the calculation of D * being performed, the inputted previous duty instruction value Da * is output to the threshold value setting unit 83.

The dead time setting unit 82 outputs, to the threshold setting unit 83, a dead time set value Dd when the switch elements 11a to 14a are subjected to switching operation. Based on duty instruction value D * or duty instruction value Da * input from operation determination unit 81 and dead time set value Dd input from dead time setting unit 82, threshold setting unit 83 selects switch element 11a. The on timing thresholds 51a to 54a and the off timing thresholds 51b to 54b for respectively determining the on / off timing of the timings 14a to 14a are set and output to the comparator 86. For example, assuming that the dead time set value Dd for the first leg is Dd_12 and the dead time set value Dd for the second leg is Dd_34, the on timing threshold 51a to 54a and the off timing threshold 51b to 54b are set as follows. Ru. In the following, “Cmax” represents the maximum value of the carrier signal generated by the carrier signal generator 85.

ON timing threshold OFF timing threshold 51a: Cmax 51b: 0.5Cmax-Dd_12
52a: 0.5Cmax 52b: Cmax-Dd_12
53a: D * + Dd_34 53b: D * + 0.5 Cmax
54a: D * + 0.5Cmax + Dd_34 54b: D *

In the above, an example in which the duty instruction value D * is input to the threshold setting unit 83 is shown. However, when the duty instruction value Da * is input last time, the D * is replaced with Da *. Similarly, on timing thresholds 51a to 54a and off timing thresholds 51b to 54b can be set.

When the switching carrier frequency setting unit 70 calculates the switching carrier frequency setting value f *, the calculation determining unit 84 outputs the inputted switching carrier frequency setting value f * to the carrier signal generation unit 85, and switching is performed. When the switching carrier frequency setting value fa * is previously input from the clock unit 65 without calculation of the switching carrier frequency setting value f * in the carrier frequency setting unit 70, the previous switching carrier frequency setting value fa * input is The signal is output to the carrier signal generator 85. Carrier signal generation unit 85 generates carrier signal 55 having a frequency according to these setting values based on switching carrier frequency setting value f * or previous switching carrier frequency setting value fa * input from operation determining unit 84. , Output to the comparator 86. The carrier signal 55 generated by the carrier signal generation unit 85 is a periodic signal such as a triangular wave which continuously changes from 0 to a predetermined maximum value Cmax, and the switching carrier frequency setting value f * or the previous switching carrier frequency setting value It changes repeatedly in a cycle according to fa *.

The comparator 86 compares the carrier signal 55 input from the carrier signal generation unit 85 with the on-timing thresholds 51a to 54a and the off-timing thresholds 51b to 54b input from the threshold setting unit 83 to obtain a duty instruction value. Pulse modulation is performed according to D * or the previous duty instruction value Da *, and output signals 51 to 54 are generated. The control circuit 50 controls the on / off of the switch elements 11 a to 14 a of the switching circuit 10 by outputting the output signals 51 to 54 generated by the comparator 86 to the gate driver 90.

FIG. 4 is a diagram for explaining the operation of the comparator 86. FIG. 4A is an example of the case where the frequency of the carrier signal 55 is high (when the cycle is short), and the waveform of the carrier signal 55 is indicated by a symbol 55a. FIG. 4B is an example of the case where the frequency of the carrier signal 55 is low (when the cycle is long), and the waveform of the carrier signal 55 is indicated by reference numeral 55b. As shown in FIGS. 4A and 4B, the inclination of the carrier signal 55 corresponds to the switching carrier frequency set value f * output from the calculation determination unit 84 or the previous switching carrier frequency set value fa *. Change accordingly.

The comparator 86 compares the carrier signal 55 with the on timing thresholds 51a to 54a and the off timing thresholds 51b to 54b. As a result, as shown in FIGS. 4A and 4B, when the value of the carrier signal 55 exceeds the on timing thresholds 51a to 54a, the output signals 51 to 54 are respectively turned off (L level) Switch to on (H level). Also, when the value of the carrier signal 55 exceeds the off timing thresholds 51b to 54b, the output signals 51 to 54 are switched from on (H level) to off (L level). Specifically, for example, when the on timing thresholds 51a to 54a and the off timing thresholds 51b to 54b are set to the values as described above, the on / off of the output signals 51 to 54 are sequentially switched as follows. .

(1) The output signal 54 is turned off at the timing when the value of the carrier signal 55 reaches the off timing threshold signal 54 b.
(2) The output signal 53 is turned on at the timing when the value of the carrier signal 55 reaches the on-timing threshold signal 53a.
(3) The output signal 51 is turned off at the timing when the value of the carrier signal 55 reaches the on timing threshold signal 51 b.
(4) The output signal 52 is turned on at the timing when the value of the carrier signal 55 reaches the on-timing threshold signal 52a.
(5) The output signal 53 is turned off at the timing when the value of the carrier signal 55 reaches the off timing threshold signal 53b.
(6) The output signal 54 is turned on at the timing when the value of the carrier signal 55 reaches the on-timing threshold signal 54a.
(7) The output signal 52 is turned off at the timing when the value of the carrier signal 55 reaches the on-timing threshold signal 52b.
(8) The output signal 51 is turned on at the timing when the value of the carrier signal 55 reaches the on-timing threshold signal 51a.

As described above, the signal generation unit 80 generates the output signals 51 to 54 for setting the on / off timings of the switch elements 11a to 14a, respectively.

(Control flow)
FIG. 5 is a control flow diagram of the control circuit 50. Hereinafter, the operation of the control circuit 50 described above will be described using the control flow diagram of FIG.

In step S10, the control circuit 50 determines whether the value of the A counter of the clock unit 65 is equal to or greater than a predetermined threshold value Ta. If the value of the A counter is greater than or equal to the threshold value Ta, the process proceeds to step S20, and if less than the threshold value Ta, the process proceeds to step S40.

In step S20, the control circuit 50 causes the voltage control unit 60 to obtain the difference between the output voltage Vout detected by the voltage detector 42 and the voltage command value Vref by the subtraction unit 61, and performs PI control based on the difference. Implement at 62. Thereafter, in step S30, the duty limiting unit 63 sets the lower limit value and the upper limit value for the duty value D obtained by the PI control in step S20, and determines the duty indication value D *. After step S30, the control circuit 50 outputs the determined duty instruction value D * to the switching carrier frequency setting unit 70, the signal generation unit 80, and the clock unit 65, and the process proceeds to step S50.

In step S40, control circuit 50 outputs the previous duty instruction value Da * stored in clock unit 65 to signal generation unit 80, and the process proceeds to step S50. When the previous duty instruction value Da * is not stored in the clock unit 65, such as in the initial state immediately after activation of the control circuit 50, a predetermined initial value (for example, 0) is output as the previous duty instruction value Da *. do it.

In step S50, the control circuit 50 determines whether the value of the B counter of the clock unit 65 is equal to or greater than a predetermined threshold value Tb. If the value of the B counter is greater than or equal to the threshold Tb, the process proceeds to step S60, and if less than the threshold Tb, the process proceeds to step S100.

In step S60, the control circuit 50 causes the switching carrier frequency setting unit 70 to multiply the duty instruction value D * input from the voltage control unit 60 and the input voltage Vin detected by the voltage detector 41 by the multiplication unit 71. The proportional value is converted to the magnetic flux density value B of the transformer 20 by the proportional unit 72. Thereafter, in steps S70 to S90, PI control is performed so that the difference between the magnetic flux density value B calculated in step S60 and the magnetic flux density command value Bref approaches zero. Specifically, the difference between the magnetic flux density value B and the magnetic flux density command value Bref is determined by the subtraction unit 73, and PI control based on the difference is performed by the PI control unit 75 to determine the switching carrier frequency f. Then, setting of the lower limit value and the upper limit value is performed on the obtained switching carrier frequency f by the frequency limiting unit 76, and the switching carrier frequency setting value f * is determined. Thus, when B <Bref (S70: Yes), the switching carrier frequency setting value f * is decreased (S80), and when B ≧ Bref (S70: No), the switching carrier frequency setting value f * is increased (S70) S90). After step S80 or S90, the control circuit 50 outputs the determined switching carrier frequency set value f * to the signal generation unit 80 and the clock unit 65, and the process proceeds to step S110.

In step S100, control circuit 50 outputs the previous switching carrier frequency set value fa * stored in clock unit 65 to signal generation unit 80, and the process proceeds to step S110. When the switching carrier frequency set value fa * is not stored in the clock unit 65 last time, such as in the initial state immediately after activation of the control circuit 50, the predetermined initial value (for example, maximum switching carrier frequency fmax) is switched to the previous time. It may be output as the carrier frequency set value fa *.

In step S110, the control circuit 50 causes the signal generation unit 80 to set the duty instruction value D * obtained in step S30 or S40 or the previous duty instruction value Da * and the switching carrier frequency obtained in step S80, S90 or S100. Output signals 51 to 54 are generated based on the set value f * or the previous switching carrier frequency set value fa *. Specifically, the signal generation unit 80 outputs the duty instruction value D * or the previous duty instruction value Da * from the operation determination unit 81 to the threshold setting unit 83, and the threshold setting unit 83 responds to the dead time setting value Dd. The on timing thresholds 51a to 54a and the off timing thresholds 51b to 54b are set. Further, the switching carrier frequency set value f * or the previous switching carrier frequency set value fa * is output from the calculation determination unit 84 to the carrier signal generation unit 85, and the carrier signal generation unit 85 generates the carrier signal 55. Then, the comparator 86 compares the carrier signal 55 with the on timing thresholds 51a to 54a and the off timing thresholds 51b to 54b to generate output signals 51 to 54. After step S110, the control circuit 50 outputs the generated output signals 51 to 54 to the gate driver 90, and the process proceeds to step S120.

At step S120, control circuit 50 determines whether or not to stop DC-DC converter 100. For example, when a predetermined stop condition such as an external control stop command for the DC-DC converter 100 is input from outside, the control circuit 50 determines that the DC-DC converter 100 is to be stopped, and the control flow of FIG. Stop the operation. On the other hand, when the stop condition is not satisfied, the control circuit 50 determines that the DC-DC converter 100 is not to be stopped, returns to step S10, and repeats the above process.

According to the first embodiment of the present invention described above, the following effects can be obtained.

(1) The DC-DC converter 100, which is a power conversion device, performs voltage conversion using the switching circuit 10 for converting the input first DC power into AC power, the transformer 20 for performing voltage conversion of AC power, and the transformer 20 And an output circuit 30 for converting the AC power into a second DC power. The control circuit 50 that controls the DC-DC converter 100 calculates the magnetic flux density value B of the transformer 20, and controls the drive frequency of the switching circuit 10 based on the calculated magnetic flux density value B. As a result, the switching loss of the DC-DC converter 100 can be sufficiently reduced.

(2) The control circuit 50 calculates the duty command value D * for controlling the output voltage Vout of the output circuit 30, and based on the duty command value D * and the input voltage Vin of the switching circuit 10. Switching carrier frequency setting unit 70 which calculates magnetic flux density value B and sets the switching carrier frequency according to the driving frequency of switching circuit 10 based on calculated magnetic flux density value B, and duty instruction value D * and switching carrier frequency And a signal generator 80 for generating output signals 51 to 54 for driving the switching circuit 10 and outputting the generated output signals 51 to 54 to the switching circuit 10 via the gate driver 90. Since this is done, without detecting the current of the transformer 20, it is possible to operate the DC-DC converter 100 at an appropriate drive frequency while reliably preventing the magnetic saturation of the transformer 20, and to reduce the switching loss.

(3) When the magnetic flux density value B is smaller than a predetermined magnetic flux density command value Bref based on the saturation magnetic flux density of the transformer 20, the switching carrier frequency setting unit 70 lowers the drive frequency of the switching circuit 10 and the magnetic flux density value B Is larger than the magnetic flux density command value Bref, the switching carrier frequency is set so as to increase the drive frequency of the switching circuit 10. Since this is done, DC-DC converter 100 can be operated at an appropriate drive frequency such that magnetic flux density value B approaches magnetic flux density command value Bref.

(4) The control circuit 50 further includes a clock unit 65 that controls the execution timing of the voltage control unit 60 and the switching carrier frequency setting unit 70. Thus, the voltage control unit 60 and the switching carrier frequency setting unit 70 can be operated at appropriate timings.

-Second embodiment-
Next, a second embodiment of the present invention will be described. In this embodiment, an example in which the magnetic flux density command value Bref is changed according to the temperature of the transformer 20 in the magnetic flux density command value setting unit 74 of the switching carrier frequency setting unit 70 will be described.

FIG. 6 is a diagram showing a basic circuit configuration of a DC-DC converter 100 according to a second embodiment of the present invention. As shown in FIG. 6, the DC-DC converter 100 according to the present embodiment has been described in the first embodiment except that the temperature detector 43 for detecting the temperature of the transformer 20 is provided in the vicinity of the transformer 20. It has the same configuration as that of

In the present embodiment, the temperature detector 43 detects the temperature of the transformer 20 and outputs the detected value to the control circuit 50. The control value of the transformer temperature output from the temperature detector 43 is input to the magnetic flux density command value setting unit 74 of the switching carrier frequency setting unit 70 in the control circuit 50. The magnetic flux density command value setting unit 74 changes the magnetic flux density command value Bref to be output to the subtracting unit 73 based on the input detection value of the transformer temperature.

FIG. 7 is a diagram showing an example of the relationship between the transformer temperature and the magnetic flux density command value Bref. FIG. 7A shows an example in which the magnetic flux density command value Bref is increased at a constant rate according to the rise of the transformer temperature, and the magnetic flux density command value Bref is decreased at a constant rate according to the drop of the transformer temperature. There is. In FIG. 7B, the magnetic flux density command value Bref is continuously changed in the area where the transformer temperature is less than the predetermined value in consideration of the temperature dependency, and in the area where the transformer temperature is equal to or more than the predetermined value An example is shown where it is constant without changing. Note that the magnetic flux density command value Bref may be changed according to the transformer temperature using a relationship other than that illustrated in FIG. 7A or 7B. For example, the magnetic flux density command value Bref may be decreased according to the rise of the transformer temperature, and the magnetic flux density command value Bref may be increased according to the fall of the transformer temperature. Also, the transformer temperature and the magnetic flux density command value Bref may not be in a proportional relationship, or may not be a relationship defined by a continuous function.

According to the second embodiment of the present invention described above, the DC-DC converter 100 which is a power conversion device further includes the temperature detector 43 for detecting the temperature of the transformer 20. In the control circuit 50, the switching carrier frequency setting unit 70 changes the magnetic flux density command value Bref based on the temperature detection value of the transformer 20 by the temperature detector 43. Since this is done, in addition to the effects described in the first embodiment, the DC-DC converter 100 can be more accurately controlled.

-Third embodiment-
Next, a third embodiment of the present invention will be described. In the present embodiment, an example in which the magnetic flux density command value Bref is changed in accordance with the output current from the DC-DC converter 100 in the magnetic flux density command value setting unit 74 of the switching carrier frequency setting unit 70 will be described.

FIG. 8 is a diagram showing a basic circuit configuration of a DC-DC converter 100 according to a third embodiment of the present invention. As shown in FIG. 8, in the DC-DC converter 100 of the present embodiment, the current detector 44 for detecting the output current of the DC-DC converter 100 output from the output circuit 30 to the low voltage battery V2 has a low voltage It has the same configuration as that described in the first embodiment except that it is provided between the batteries V2.

In the present embodiment, the current detector 44 detects the output current from the output circuit 30, and outputs the detected value to the control circuit 50. In addition, although the current detector 44 is connected to the negative electrode output terminal 4 side in FIG. 8, it may be connected to the positive electrode output terminal 3 side. The detected value of the output current output from the current detector 44 is input to the magnetic flux density command value setting unit 74 of the switching carrier frequency setting unit 70 in the control circuit 50. The magnetic flux density command value setting unit 74 changes the magnetic flux density command value Bref to be output to the subtracting unit 73 based on the input detection value of the output current.

FIG. 9 is a diagram showing an example of the relationship between the output current and the magnetic flux density command value Bref. FIG. 9A shows an example in which the magnetic flux density command value Bref is increased at a constant rate according to the rise of the output current, and the magnetic flux density command value Bref is decreased at a constant rate according to the fall of the output current. There is. In FIG. 9B, the flux density command value Bref is continuously changed in the area where the output current is less than the predetermined value in consideration of the output current dependency, and in the area where the output current is equal to or more than the predetermined value, the flux density command value Bref. An example is shown in which it is made constant without changing. The magnetic flux density command value Bref may be changed according to the output current using a relationship other than that illustrated in FIGS. 9A and 9B. For example, the magnetic flux density command value Bref may be decreased according to the rise of the output current, and the magnetic flux density command value Bref may be increased according to the fall of the output current. Further, the output current and the magnetic flux density command value Bref may not be in a proportional relationship, or may not be a relationship defined by a continuous function.

According to the third embodiment of the present invention described above, the DC-DC converter 100, which is a power conversion device, further includes a current detector 44 that detects the output current from the output circuit 30. In the control circuit 50, the switching carrier frequency setting unit 70 changes the magnetic flux density command value Bref based on the detection value of the output current by the current detector 44. With this configuration, as in the second embodiment, in addition to the effects described in the first embodiment, the DC-DC converter 100 can be more accurately controlled. .

-Fourth Embodiment-
Next, a fourth embodiment of the present invention will be described. In the present embodiment, an example will be described in which gain adjustment is performed on PI control performed by the PI control unit 75 of the switching carrier frequency setting unit 70.

FIG. 10 is a diagram showing a configuration of a control circuit 50 according to a fourth embodiment of the present invention. As shown in FIG. 10, the control circuit 50 of this embodiment is the first embodiment except that the switching carrier frequency setting unit 70 further includes a gain adjustment unit 75a connected to the PI control unit 75. It has the same configuration as that described above. Below, the added gain adjustment part 75a is demonstrated.

(Gain adjustment unit 75a)
The PI control unit 75 performs PI control according to the difference between the magnetic flux density command value Bref and the magnetic flux density value B by PI calculation using a predetermined PI control gain. When the PI control gain is increased, the change of the switching carrier frequency when the magnetic flux density value B approaches the magnetic flux density command value Bref is quickened, and the responsiveness of the switching circuit 10 is improved. On the other hand, when the PI control gain is lowered, the change of the switching carrier frequency when the magnetic flux density value B approaches the magnetic flux density command value Bref is delayed, and the responsiveness of the switching circuit 10 is lowered. Therefore, in the control circuit 50 of the present embodiment, the responsiveness of the switching circuit 10 is adjusted by appropriately adjusting the PI control gain used in the PI control unit 75 in the gain adjustment unit 75a.

FIG. 11 is a diagram illustrating an example of a method of adjusting the PI control gain by the gain adjustment unit 75a. FIG. 11 (a) shows an example of adjusting the PI control gain according to the switching carrier frequency, and FIG. 11 (b) shows an example of adjusting the PI control gain according to the output current from the output circuit 30. It shows. The gain adjustment unit 75a can change the PI control gain based on the switching carrier frequency and the output current as illustrated in, for example, FIG. 11A and FIG. 11B. It is also possible to adjust the PI control gain using both the switching carrier frequency and the output current, or to adjust the PI control gain using information other than these.

According to the fourth embodiment of the present invention described above, the switching carrier frequency setting unit 70 performs PI control based on the difference between the magnetic flux density value B and the magnetic flux density command value Bref with a predetermined PI control gain. The gain control unit 75a changes the PI control gain based on at least one of the switching carrier frequency and the output current from the output circuit 30. Since this is done, in addition to the effect described in the first embodiment, there is an effect that the responsiveness of the switching circuit 10 can be appropriately adjusted while reliably preventing the magnetic saturation of the transformer 20.

-Fifth embodiment-
Next, a fifth embodiment of the present invention will be described. In this embodiment, a method of determining the upper limit value and the lower limit value of the switching carrier frequency set value f * set in the frequency limiting unit 76 of the switching carrier frequency setting unit 70 will be described.

FIG. 12 is a diagram showing a configuration of a control circuit 50 according to a fifth embodiment of the present invention. As shown in FIG. 12, the control circuit 50 of the present embodiment is the first embodiment except that the switching carrier frequency setting unit 70 further includes a limit value setting unit 76 a connected to the frequency limiting unit 76. It has the same configuration as that described in the embodiment. Hereinafter, the added limit value setting unit 76a will be described.

(Limit value setting unit 76a)
Limit value setting unit 76a determines the upper limit value and the lower limit value of switching carrier frequency setting value f * output from frequency limiting unit 76, that is, the maximum switching carrier frequency fmax and the minimum switching carrier frequency fmin in the above equation (2). These determined values are output to the frequency limiting unit 76. For example, limit value setting unit 76 a responds to a target value of magnetic flux density based on the saturation magnetic flux density of transformer 20, ie, magnetic flux density command value Bref, and the cross-sectional area of the core of transformer 20 and the number of turns of primary winding N 1. Determine the maximum switching carrier frequency fmax.

On the other hand, limit value setting unit 76a determines minimum switching carrier frequency fmin in accordance with the upper limit value of the magnetic flux density set in advance based on the saturation magnetic flux density of insulating transformer 90a in gate driver 90, for example. Specifically, the minimum switching carrier frequency fmin is determined using the following equation (3) so that the magnetic flux density of the insulating transformer 90a does not exceed the saturation magnetic flux density even when the switching carrier frequency is lowered. In Equation (3), Vdd is a voltage value input to the gate driver 90 through the insulating transformer 90a, As is a core cross-sectional area of the insulating transformer 90a, Bmax is an upper limit value of magnetic flux density of the insulating transformer 90a, and N1 is insulating It is a transformer winding number of the transformer 90a.

In addition, for example, the limit value setting unit 76a may determine the minimum switching carrier frequency fmin such that the current ripple flowing through the smoothing coil L2 in the output circuit 30 is equal to or less than a predetermined ripple current value. Specifically, even when the switching carrier frequency is lowered, the following equation (4) is used so that the current ripple in the output current from output circuit 30 flowing in smoothing coil L2 does not exceed the predetermined ripple current value. The minimum switching carrier frequency fmin is determined. In Equation (4), Vin is an input voltage value of the DC-DC converter 100 detected by the voltage detector 41, Vout is an output voltage of the DC-DC converter 100 detected by the voltage detector 42, and D * is a voltage The duty instruction value determined by the control unit 60, L2 is the inductance value of the smoothing coil L2, ΔILmax is the difference between the peak maximum ripple current value and the peak maximum ripple current value, and Nt is the turns ratio of the transformer 20.

However, the limit value setting unit 76a may determine the minimum switching carrier frequency fmin using something other than the above equation (3) or (4). As long as the magnetic flux density of isolation transformer 90a does not exceed the saturation magnetic flux density or the current ripple in the output current from output circuit 30 does not exceed a predetermined ripple current value, the minimum switching carrier frequency fmin should be determined by any method. Is possible. The minimum switching carrier frequency fmin can be determined based on both the saturation magnetic flux density of the isolation transformer 90a and the current ripple in the output current, or the minimum switching carrier frequency fmin can be determined based on other information. It is.

According to the fifth embodiment of the present invention described above, the switching circuit 10 is connected to the control circuit 50 via the gate driver 90 which is an insulated gate driver mounting the isolation transformer 90 a. In addition, switching carrier frequency setting unit 70 has frequency limiting unit 76 that limits the switching carrier frequency to a predetermined minimum switching carrier frequency fmin or more. In limit value setting unit 76a, the saturation magnetic flux density of isolation transformer 90a and The minimum switching carrier frequency fmin is determined based on at least one of the current ripple in the output current from the output circuit 30. Since this is done, in addition to the effects described in the first embodiment, not only the magnetic saturation of the transformer 20, but also the magnetic saturation of the insulating transformer 90a in the gate driver 90 and the output current of the DC-DC converter 100 It is also possible to prevent an increase in the current ripple in

In each of the embodiments of the present invention described above, the switching circuit 10, which is a voltage-type full bridge circuit configured by four switch elements 11a to 14a, and the transformer 20, which is a current-type center tap circuit, are combined. Although the present invention has been described using the example of the control circuit 50 that controls the DC-DC converter 100 according to the phase shift control method, the present invention is not limited to this. Switching circuit for converting input first DC power into AC power, transformer for performing voltage conversion of AC power, and output circuit for converting AC power subjected to voltage conversion by the transformer to second DC power The present invention can be applied to any control device that controls a power conversion device having the above-described components, and the same operation and effect as those described in the embodiments can be obtained. Moreover, each embodiment described above may be applied independently, and may be combined arbitrarily.

The embodiments and the various modifications described above are merely examples, and the present invention is not limited to these contents as long as the features of the invention are not impaired. In addition, although various embodiments and modifications have been described above, the present invention is not limited to these contents. Other embodiments considered within the scope of the technical idea of the present invention are also included within the scope of the present invention.

DESCRIPTION OF SYMBOLS 1 ... positive electrode input terminal, 2 ... negative electrode input terminal, 3 ... positive electrode output terminal, 4 ... negative electrode output terminal, 10 ... switching circuit, 11a to 14a ... switching element, 11b to 14b ... diode, 11c to 14c ... capacitor, 20 ... Transformer 30 30

output circuit

31, 32

diode

41, 42 voltage detector 43 temperature detector 44 current detector 50 control circuit 51 to 54 output signal 60 voltage controller , 61: subtraction unit, 62: PI control unit, 63: duty limitation unit, 65: clock unit, 70: switching carrier frequency setting unit, 71: multiplication unit, 72: proportional unit, 73: subtraction unit, 74: magnetic flux density Reference value setting unit 75 PI control unit 75a Gain adjustment unit 76 Frequency limiting unit 76a Limit value setting unit 80 Signal generation unit 81 Operation determination unit 82 Dead Time setting unit 83 Threshold setting unit 84 Operation determination unit 85 Carrier signal generation unit 86 Comparator 90 Gate driver 90a Insulated transformer 91 to 94 Drive signal 100 DC-DC Converter, 200: vehicle power control unit, 300: HV system equipment, 400: auxiliary equipment, 1000: vehicle, N1: primary winding, N2a, N2b: secondary winding, S: rectification connection point, T: neutral Point, V1 ... high voltage battery, V2 ... low voltage battery

Claims

A control device that controls a power conversion device that converts an input first DC power into a second DC power and outputs the second DC power.
The power conversion device includes a switching circuit that converts the first DC power into AC power, a transformer that performs voltage conversion of the AC power, and the second DC power of the AC power converted by the transformer. And an output circuit for converting into
The control apparatus which calculates the magnetic flux density value of the said transformer, and controls the drive frequency of the said switching circuit based on the calculated said magnetic flux density value.
In the control device according to claim 1,
A voltage control unit that calculates a duty indication value for controlling an output voltage of the output circuit;
A switching carrier frequency setting unit that calculates the magnetic flux density value based on the duty indication value and the input voltage of the switching circuit, and sets a switching carrier frequency according to the drive frequency based on the calculated magnetic flux density value;
A control signal generator configured to generate an output signal for driving the switching circuit based on the duty instruction value and the switching carrier frequency, and to output the generated output signal to the switching circuit.
In the control device according to claim 2,
The switching carrier frequency setting unit lowers the driving frequency when the magnetic flux density value is smaller than a predetermined magnetic flux density command value based on the saturation magnetic flux density of the transformer, and the magnetic flux density value is higher than the magnetic flux density command value. The control apparatus which sets the said switching carrier frequency so that the said driving frequency may be made high when also large.
In the control device according to claim 2 or 3,
The control device further comprising a clock unit that controls execution timings of the voltage control unit and the switching carrier frequency setting unit.
In the control device according to claim 3,
The power converter further includes a temperature detector for detecting the temperature of the transformer,
The control device, wherein the switching carrier frequency setting unit changes the magnetic flux density command value based on a temperature detection value of the transformer by the temperature detector.
In the control device according to claim 3 or 5,
The power converter further includes a current detector that detects an output current from the output circuit,
The control device, wherein the switching carrier frequency setting unit changes the magnetic flux density command value based on a detected value of the output current by the current detector.
In the control device according to claim 3, claim 5 or claim 6.
The switching carrier frequency setting unit has a PI control unit that performs proportional integral control based on a difference between the magnetic flux density value and the magnetic flux density command value with a predetermined control gain,
A control device for changing the control gain based on at least one of the switching carrier frequency and an output current from the output circuit.
In the control device according to any one of claims 2 to 7,
The switching circuit is connected to the control device via an insulated gate driver mounted with an isolation transformer,
The switching carrier frequency setting unit has a frequency limiting unit that limits the switching carrier frequency to a predetermined minimum frequency or higher.
A control device that determines the minimum frequency based on at least one of a saturation magnetic flux density of the isolation transformer and a current ripple in an output current from the output circuit.