WO2018205237A1 - 基于单频线的毫米波双频Doherty功率放大器 - Google Patents

基于单频线的毫米波双频Doherty功率放大器 Download PDF

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WO2018205237A1
WO2018205237A1 PCT/CN2017/084050 CN2017084050W WO2018205237A1 WO 2018205237 A1 WO2018205237 A1 WO 2018205237A1 CN 2017084050 W CN2017084050 W CN 2017084050W WO 2018205237 A1 WO2018205237 A1 WO 2018205237A1
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frequency
dual
power amplifier
matching network
line
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PCT/CN2017/084050
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French (fr)
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吕关胜
陈文华
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清华大学
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    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F1/00—Details of amplifiers with only discharge tubes, only semiconductor devices or only unspecified devices as amplifying elements
    • H03F1/02—Modifications of amplifiers to raise the efficiency, e.g. gliding Class A stages, use of an auxiliary oscillation

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  • the present invention relates to the field of mobile communication technologies, and in particular, to a millimeter wave dual-frequency Doherty power amplifier based on a single frequency line.
  • the millimeter wave band will be adopted.
  • WRC-15 World Radiocommunication Conference
  • ITU ITU
  • the Federal Communications Commission (FCC) has opened the 28 GHz, 37 GHz, 39 GHz, and 64-71 GHz millimeter wave bands and incorporated them into the 5G standard. It can be seen that a plurality of millimeter wave bands will be adopted in 5G communication, and the demand for dual-frequency or even multi-frequency millimeter wave power amplifiers is objectively proposed.
  • the structure of a conventional Doherty power amplifier is shown, which includes two power amplifier tubes, that is, a main power amplifier tube Main and a secondary power amplifier tube Aux.
  • the Main offset is Class AB
  • Aux is biased to Class C.
  • the main power amplifier output matching network OMN_Main and the auxiliary power amplifier output matching network OMN_Aux are respectively responsible for the output impedance matching of the two power amplifier tubes.
  • the post-matching network PMN matches the 50 ohm load to 25 ohms.
  • Offset_Main and Offset_Aux are phase shift lines with a characteristic impedance of 50 ohms and are responsible for impedance matching at back-off power.
  • the two splitter SPLIT divides the input power into two paths.
  • the main power amplifier input matching network IMN_Main and the auxiliary power amplifier input matching network IMN_Aux respectively match the input impedance of Main and Aux to 50 ohms, and compensate the main phase shift line Phaseline. The phase difference between the road and the auxiliary road.
  • the working principle of the traditional Doherty power amplifier can be divided into two areas: low power zone and high power zone.
  • the two zones are bounded by the opening of class C auxiliary power amplifier.
  • the output circuit of Aux is isolated from the output circuit of Main, and the equivalent circuit is shown in Figure 2.
  • ZSat is the load impedance that can achieve the maximum efficiency of the power amplifier tube under saturated power, and can also be obtained by the Loadpull method. Since Offset_Main and Offset_Aux have a characteristic impedance of 50 ohms, they have no effect on the matching effect of OMN_Main and OMN_Aux.
  • Main Due to the impedance pull of Aux, Main can have different load impedances in the low power zone and high power zone.
  • OMN_Main and OMN_Aux By properly designing the OMN_Main and OMN_Aux, and the phase shift lines of Offset_Main and Offset_Aux, Main can be in the low power zone and The saturation point is highly efficient. Compared with ordinary class AB amplifiers, Doherty power amplification greatly improves the efficiency under high PAPR excitation.
  • the phase shift line In order to design a dual-frequency doherty power amplifier, not only the input and output matching networks of the main and auxiliary power amplifiers are required to be dual-frequency matching networks.
  • the network also requires the phase shift line to be a dual-frequency phase shift line, which produces the required phase shift at both design frequencies while maintaining a characteristic impedance of 50 Ohm.
  • the dual-frequency line as shown in FIG. 4 is often used, and the ⁇ -type structure, the T-type structure, and the coupled line are sequentially from left to right. Although these structures can achieve the effect of dual-frequency phase shift, they are usually large in size, the line width may be unreasonable, and the bandwidth is narrow. These problems limit their application in the millimeter wave band.
  • the millimeter wave power amplifier must be realized by the integrated circuit process, and the larger dual frequency line size means a larger chip area, which greatly increases the processing cost.
  • the loss of passive components in the millimeter wave band is large. If the size of the dual frequency line is large, the loss introduced will seriously degrade the performance of the power amplifier, and the impact will be more obvious when the line width is unreasonable.
  • the integrated circuit process usually has processing errors. If the bandwidth of the dual-frequency line is too narrow, the actual dual-frequency phase shift will seriously deviate from the design value, so that the actually processed dual-frequency doherty power amplifier can not achieve dual-frequency performance. Therefore, the difficulty in realizing the millimeter-wave dual-frequency doherty power amplifier lies in the realization of the high-performance dual-frequency phase shift line. With the acceleration of the 5G communication process, this problem needs to be solved urgently.
  • the present invention aims to solve at least one of the technical problems in the related art described above to some extent.
  • the object of the present invention is to provide a millimeter-wave dual-frequency Doherty power amplifier based on a single frequency line, which realizes a dual-frequency phase shift function by using a common single-frequency transmission line, and overcomes the large size of the conventional dual-frequency transmission line in the millimeter wave frequency band.
  • the problem of high loss and narrow bandwidth can greatly improve the performance of the dual-frequency phase shift line.
  • an embodiment of the present invention provides a millimeter-wave dual-frequency Doherty power amplifier based on a single frequency line, including: a dual-frequency power splitter, a dual-frequency input matching network for a main power amplifier, and a dual-frequency input matching network for a secondary power amplifier.
  • main power amplifier tube, auxiliary power amplifier tube, main power amplifier dual-frequency output matching network, auxiliary power amplifier dual-frequency output matching network and dual-frequency post-matching network wherein the dual-frequency power splitter, main power amplifier dual-frequency input matching network, main The power amplifier tube, the main power amplifier dual-frequency output matching network and the dual-frequency post-matching network are sequentially connected; the dual-frequency power splitter, the auxiliary power amplifier dual-frequency input matching network, the auxiliary power amplifier tube, the auxiliary power amplifier dual-frequency output matching network and the dual frequency The matching network is connected in sequence; the dual-frequency power splitter is connected to the main power amplifier dual-frequency input matching network through a first single-frequency transmission line of a first preset length, and the dual-frequency output matching network of the main power amplifier passes the second preset a second single-frequency transmission line of a length is connected to the dual-frequency post-matching network, and the auxiliary power amplifier dual-frequency output matching network passes a third single-frequency transmission of a third preset length
  • the transmission line is
  • the single-frequency line-based millimeter-wave dual-frequency Doherty power amplifier according to the above-described embodiments of the present invention may further have the following additional technical features:
  • the dual-frequency power splitter distributes input power to the main power amplifier and the auxiliary power amplifier at a first design frequency f1 and a second design frequency f2, the first design frequency f1 being smaller than the second Design frequency f2.
  • a phase shift ThetaP1 corresponding to the first single frequency transmission line, a phase shift ThetaM1 corresponding to the second single frequency transmission line, and a phase shift ThetaA1 corresponding to the third single frequency transmission line are obtained.
  • the ThetaM1 has a first preset value range; when operating at the second design frequency f2, the phase shift ThetaP2 corresponding to the first single frequency transmission line, the phase shift ThetaM2 corresponding to the second single frequency transmission line, and The phase shift ThetaA2 corresponding to the third single frequency transmission line, wherein the ThetaM2 has a second preset value range.
  • ThetaM2 k*ThetaM1
  • k is the operating frequency ratio.
  • the first single frequency transmission line, the second single frequency transmission line, and the third single frequency transmission line have a characteristic impedance of 50 ohms.
  • the first design frequency f1 is 28 GHz and the second design frequency f2 is 39 GHz.
  • the main power amplifier tube and the auxiliary power amplifier tube are the same, wherein the main power amplifier is biased to class AB, the final stage drain voltage is 4V, the auxiliary power amplifier is biased to class C, and the final stage voltage is set to 4.5V.
  • the main power amplifier tube and the auxiliary power amplifier tube adopt a two-stage structure, and the transistor sizes of the driver stage and the power stage are 2 ⁇ 75 um and 4 ⁇ 75 um, respectively.
  • the dual frequency power splitter is a wide non-equal Wilkinson power splitter.
  • the dual frequency post-matching network is a two-level microstrip line structure.
  • the dual-frequency phase shift function is implemented by using a common single-frequency transmission line, which overcomes the large size, high loss, and bandwidth of the conventional dual-frequency transmission line in the millimeter wave band.
  • the narrow problem can greatly improve the performance of the dual-frequency phase-shift line and help achieve high-performance millimeter-wave dual-frequency doherty power amplifiers.
  • the performance of the ordinary transmission line is not sensitive to process errors, the reliability of the design is improved.
  • 1 is a schematic structural view of a conventional Doherty power amplifier
  • FIG. 2 is a schematic diagram of an equivalent circuit of a conventional Doherty power amplifier in a low power region
  • FIG. 3 is a schematic diagram showing the operation state of a conventional Doherty power amplifier at a saturation point in a high power region
  • FIG. 4 is a schematic structural diagram of a dual-frequency transmission line used in a conventional Doherty power amplifier
  • FIG. 5 is a structural diagram of a millimeter wave dual-frequency Doherty power amplifier based on a single frequency line according to an embodiment of the present invention. intention;
  • FIG. 6 is a schematic diagram showing the structure of a doherty power amplifier operating at a first design frequency f1 of a millimeter-wave dual-frequency Doherty power amplifier based on a single frequency line according to an embodiment of the present invention
  • FIG. 7 is a schematic diagram showing the structure of a doherty power amplifier operating at a second design frequency f2 of a millimeter-wave dual-frequency Doherty power amplifier based on a single frequency line according to an embodiment of the present invention
  • FIG. 8 is a schematic diagram of dual-frequency phase shifting with a single frequency line of a millimeter wave dual-frequency Doherty power amplifier based on a single frequency line according to an embodiment of the present invention
  • FIG. 9 is a block diagram showing the structure of a 28 GHz/39 GHz dual-frequency doherty power amplifier according to an embodiment of the present invention.
  • FIG. 10 is a schematic structural diagram of an input matching circuit of a dual-frequency power amplifier according to an embodiment of the present invention.
  • FIG. 11 is a schematic structural diagram of an inter-stage matching circuit of a dual-frequency power amplifier according to an embodiment of the present invention.
  • FIG. 12 is a schematic structural diagram of an output matching circuit of a dual-frequency power amplifier according to an embodiment of the present invention.
  • FIG. 13 is a block diagram showing the structure of a dual-frequency post-matching network PMN of 50 ohms to 25 ohms in accordance with an embodiment of the present invention.
  • connection In the description of the present invention, it should be noted that the terms “installation”, “connected”, and “connected” are to be understood broadly, and may be fixed or detachable, for example, unless otherwise explicitly defined and defined. Connected, or integrally connected; can be mechanical or electrical; can be directly connected, or indirectly connected through an intermediate medium, can be the internal communication of the two components.
  • Connected, or integrally connected can be mechanical or electrical; can be directly connected, or indirectly connected through an intermediate medium, can be the internal communication of the two components.
  • the specific meaning of the above terms in the present invention can be understood in a specific case by those skilled in the art.
  • FIG. 5 is a schematic diagram showing the structure of a millimeter wave dual frequency Doherty power amplifier based on a single frequency line according to an embodiment of the present invention.
  • the single-frequency line-based millimeter wave dual-frequency Doherty power amplifier includes: a dual-frequency power splitter 110, a main power amplifier dual-frequency input matching network 120, a secondary power amplifier dual-frequency input matching network 130, and a main power amplifier tube 140.
  • the auxiliary power amplifier 150, the main power amplifier dual frequency output matching network 160, the auxiliary power amplifier dual frequency output matching network 170 and the dual frequency rear matching network 180 is a schematic diagram showing the structure of a millimeter wave dual frequency Doherty power amplifier based on a single frequency line according to an embodiment of the present invention.
  • the single-frequency line-based millimeter wave dual-frequency Doherty power amplifier includes: a dual-frequency power splitter 110, a main power amplifier dual-frequency input matching network 120, a secondary power amplifier dual-frequency input matching network 130, and
  • the dual-frequency power splitter 110, the main power amplifier dual-frequency input matching network 120, the main power amplifier tube 140, the main power amplifier dual-frequency output matching network 160, and the dual-frequency post-matching network 180 are sequentially connected; the dual-frequency power splitter 110 and the auxiliary power amplifier The dual frequency input matching network 130, the auxiliary power amplifier tube 150, the auxiliary power amplifier dual frequency output matching network 170 and the dual frequency rear matching network 180 are sequentially connected; the dual frequency power distributor 110 passes the first preset frequency of the first single frequency transmission line 1 and The main power amplifier dual-frequency input matching network 120 is connected, and the main power amplifier dual-frequency output matching network 160 is connected to the dual-frequency post-matching network 180 through the second preset frequency second single-frequency transmission line 2, and the auxiliary power amplifier dual-frequency output matching network 170 passes the The third preset frequency transmission line 3 of three preset lengths is connected to the dual frequency post matching network 180.
  • the first single frequency transmission line 1 (such as Phaseline in FIG. 5), the second single frequency transmission line 2 (such as Offset_Main in FIG. 5), and the third single frequency transmission line 3 (as shown in FIG. 5)
  • the characteristic impedance of Offset_Aux is 50 ohms. That is to say, the Phaseline, Offset_Main, and Offset_Aux shown in FIG. 5 are ordinary single-frequency transmission lines having a characteristic impedance of 50 Ohm. In the embodiment of the present aspect, they can simultaneously be at two design frequencies by selecting an appropriate length. Implement the corresponding phase shift.
  • the dual-frequency power splitter 110 distributes the input power to the main power amplifier and the auxiliary power amplifier at the first design frequency f1 and the second design frequency f2, wherein the first design frequency f1 is smaller than the second Design frequency f2, ie f2>f1.
  • the main and auxiliary amplifiers are dual-frequency amplifiers operating at frequencies f1 and f2, which have been independently designed to achieve the best performance. At this time, the performance of the dual-frequency phase shift line directly determines the performance of the dual-frequency Doherty amplifier.
  • the phase shift ThetaP1 corresponding to the first single-frequency transmission line 1, the phase shift ThetaM1 corresponding to the second single-frequency transmission line 2, and the phase shift ThetaA1 corresponding to the third single-frequency transmission line 3 are obtained, wherein the ThetaM1 There is a first preset range of values.
  • the structure of the Doherty power amplifier implemented at the frequency f1 is as shown in FIG.
  • ThetaP1, ThetaM1, and ThetaA1 are Phaseline (ie, the first single-frequency transmission line 1), Offset_Main (ie, the second single-frequency transmission line 2), and Offset_Aux (ie, the third single-frequency transmission line 3) corresponds to the phase shift.
  • ThetaP1, ThetaM1, and ThetaA1 do not have to be a certain set of values, and the Doherty amplifiers still maintain good performance when they vary within a certain range.
  • the fallback efficiency changes little and can maintain a high value.
  • the phase shift ThetaP2 corresponding to the first single frequency transmission line 1, the phase shift ThetaM2 corresponding to the second single frequency transmission line 2, and the phase shift ThetaA2 corresponding to the third single frequency transmission line 3 are obtained, wherein the ThetaM2 There is a second preset value range.
  • ThetaP2, ThetaM2, and ThetaA2 are Phaseline (ie, the first single-frequency transmission line 1), Offset_Main (ie, the second single-frequency transmission line 2), and Offset_Aux (ie, the third single-frequency transmission line 3) corresponds to the phase shift.
  • the rollback efficiency changes little and can maintain a high value.
  • Offset_Main in a dual-band Doherty power amplifier is used to illustrate the idea of implementing dual-frequency phase shift with a single-frequency transmission line.
  • k is the operating frequency ratio, corresponding to a straight line a of the slope of k in FIG.
  • the feasible range of ThetaM1 is ThetaM11-ThetaM12 (the first preset value range), and the feasible range of ThetaM2 is ThetaM21–ThetaM22 (ie, the first preset value range), taking into account the periodicity of the transmission line, in Figure 8
  • the black shaded area is a reasonable dual-frequency phase shift zone.
  • the part of the straight line a in the black shaded area is the feasible design interval of Offset_Main. If the line a cannot intersect with the shadow area, the feasible range of ThetaM1 and ThetaM2 can be relaxed appropriately. At this time, the dual-frequency performance of the Doherty amplifier may be reduced. Properly balance the performance at f1 and f2.
  • thetaM11 can be taken as the electrical length of the Offset_Main at f1, so that there will be a minimum physical length, and the intermediate point of the intersecting region can also be taken, which is more robust.
  • the design method of Offset_Aux (ie, the third single-frequency transmission line) and Phaseline (the first single-frequency transmission line) in the dual-band Doherty power amplifier is similar to the design process of the above-mentioned Offset_Main, and will not be described here.
  • the dual-frequency line designed according to the method described above is a common transmission line with a characteristic impedance of 50 Ohm.
  • the line width is reasonable, the loss is small, and the size is much smaller than the conventional dual-frequency line, and is suitable for the millimeter wave band.
  • the performance of the ordinary transmission line is not sensitive to process errors, the reliability of the design is improved.
  • the first design frequency f1 is 28 GHz
  • the second design frequency f2 is 39 GHz
  • a dual-frequency Doherty power amplifier that can operate at 28 GHz and 39 GHz is taken as an example.
  • the dual-frequency Doherty power amplifier is based on the 0.15um pHEMT process of WIN Semiconductor, and the main power amplifier tube 140 and the auxiliary power amplifier tube 150 are the same, that is, the main auxiliary circuit uses the same dual-frequency power amplifier, wherein the main power amplifier
  • the bias is class AB, the final stage drain voltage is 4V; the auxiliary power amplifier is biased to class C, and the final stage voltage is set to 4.5V.
  • the dual-frequency power amplifier adopts a two-stage structure, that is, the main power amplifier tube 140 and the auxiliary power amplifier tube 150 adopt a two-stage structure, and the transistor sizes of the driver stage and the power stage are 2 ⁇ 75 um and 4 ⁇ 75 um, respectively.
  • the dual-frequency power splitter 110 uses a broadband non-divided Wilkinson power splitter.
  • the overall structure of the dual-frequency Doherty power amplifier is shown in Figure 9.
  • the electrical length of Offset_Main is scanned to obtain an optimum electrical length of 65 degrees, and a high back-off efficiency is maintained in the range of 40-90 degrees.
  • Scanning Offset_Aux results in an optimum electrical length of 150 and a good open circuit in the 145-165 degree range.
  • the Phaseline is scanned to obtain an optimal electrical length of 50. In the range of 30-70 degrees, the main and auxiliary circuits still have better power synthesis effects.
  • the specific design results are shown in Table 1.
  • the width of the 50 Ohm microstrip line is 70 um, and the physical length of the three-phase phase shift line and the loss at the two operating frequencies are as shown in Table 4 below. It can be seen that the three-phase phase shift line loss in this embodiment is very low, which fully embodies the advantage of realizing dual-frequency phase shift with a single frequency line. In addition, the microstrip line is easy to bend, which can further reduce the chip area.
  • the input, inter-stage, and output matching networks of the dual-frequency power amplifier are dual-frequency matching networks, and are simultaneously implemented.
  • the impedance matching at 28 GHz and 39 GHz is shown in Figures 10, 11, and 12, respectively. Since the inductance Q value of the millimeter wave band is low, the actual implementation is replaced by a microstrip line.
  • the dual-frequency post-matching network 180 (ie, PMN), for example, employs a two-stage microstrip line structure that can match a 50 Ohm load to 25 Ohm in two operating frequency bands, such as shown in FIG.
  • the dual-frequency Doherty power amplifier described in the above embodiment of the present invention has good back-off and saturation performance at 28 GHz and 39 GHz.
  • the single-frequency line-based millimeter-wave dual-frequency Doherty power amplifier implements a dual-frequency phase shift function by using a common single-frequency transmission line, and overcomes the large size and loss of the conventional dual-frequency transmission line in the millimeter wave band.
  • the problem of high bandwidth and narrow bandwidth can greatly improve the performance of dual-frequency phase-shift lines and help achieve high-performance millimeter-wave dual-band Doherty power amplifiers.
  • the performance of the ordinary transmission line is not sensitive to process errors, the reliability of the design is improved.

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Abstract

一种基于单频线的毫米波双频Doherty功率放大器，包括：双频功率分配器（110）、主功放双频输入匹配网络（120）、辅功放双频输入匹配网络（130）、主功放管（140）、辅功放管（150）、主功放双频输出匹配网络（160）、辅功放双频输出匹配网络（170）和双频后匹配网络（180），其中，双频功率分配器（110）通过第一预设长度的第一单频传输线（1）与主功放双频输入匹配网络（120）相连，主功放双频输出匹配网络（160）通过第二预设长度的第二单频传输线（2）与双频后匹配网络（180）相连，辅功放双频输出匹配网络（170）分别通过第三预设长度的第三单频传输线（3）与双频后匹配网络（180）相连。该功率放大器采用普通单频传输线实现双频相移功能，克服了传统双频传输线在毫米波频段存在的尺寸大、损耗高、带宽窄的问题，可以大大提高双频相移线的性能。

Description

基于单频线的毫米波双频Doherty功率放大器 技术领域
本发明涉及移动通信技术领域，特别涉及一种基于单频线的毫米波双频Doherty功率放大器。
背景技术
在现代通信系统中，频率资源是最为稀缺的资源之一，为了提高频谱利用率，往往使用OFDM(Orthogonal Frequency Division Multiplexing，正交频分复用技术)、CDMA(Code Division Multiple Access，码分多址技术)等现代调制方式。然而，这些现代调制方式在提高频谱利用率的同时，也带来了高峰均比PAPR(Peak to Average Power Ratio，峰值平均功率比)的问题。这种高PAPR对射频前端，尤其是射频功率放大器的设计带来了困难，尤其是对功率放大器的效率指标带来了不利的影响。传统的AB类线性功率放大器在饱和功率附近有较高的效率，而当输出功率降低时，其效率急剧下降。由于在高PAPR下功率放大器大部分时间工作在回退功率区，使得AB类功率放大器的效率远远低于起饱和效率。可以说，高PAPR是现代通信系统中使用的功率放大器必须要应对的困难之一。
另一方面，随着通信技术的发展，有越来越多的3G/4G/5G通信系统被标准化并投入运营，这些通信系统被分配在广阔的频段上。在这种背景下，当今同一个运营商同时在多个频段上运营多个不同模式的通信系统成为一种常态。双频功放由于可以工作在两个相距较远的通信频段，可以大大降低射频前端的复杂度，受到了设备制造商和运营商的广泛关注。在目前的4G基站中，双频功放已经得到实际应用，展现出巨大的优势。
由于6GHz以下通信频段已经拥挤不堪，在未来的5G通信中，为了大幅提高通信速率，毫米波频段将会被采用。国际电联(ITU)在2015年举行的世界无线电通信大会(WRC-15)上，初步拟定了全球可行频率列表，包括24.25-27.5GHz、31.8-33.4GHz、37-40.5GHz等频段。美国联邦通信委员会(FCC)开放了28GHz、37GHz、39GHz和64-71GHz毫米波频段，并将其纳入5G标准中。可见，在5G通信中将会有多个毫米波频段被采用，客观上提出了对双频甚至多频毫米波功放的需求。
综上，高PAPR和多个毫米波工作频段，将会是5G通信基站中功率放大器设计的难点和重点之一。其中为了在高PAPR环境下保持高效率，有多种技术方案可供选择，例如Doherty技术、包络跟踪技术、序列功率放大器技术等，这些技术在公知文献1(RF Power  Amplifiers for Wireless Communications,2nd Edition,Steve C.Cripps)与公知文献2(Advanced Techniques in RF Power Amplifier Design,Steve C.Cripps)中有详细的论述，在这里不再重复。在这些技术中，Doherty技术由于其结构简单，效率高，可以与传统功率放大器原位替换等优点，成为在通信基站中应用最为广泛的高效率功率放大器技术。
例如图1所示，展示了传统Doherty功率放大器的结构，其中包含两只功放管，即主功放管Main和辅功放管Aux，通常其中Main偏置为AB类，Aux则偏置为C类。在输出端，主功放输出匹配网络OMN_Main和辅功放输出匹配网络OMN_Aux分别负责两只功放管的输出阻抗匹配。后匹配网络PMN将50欧姆负载匹配为25欧姆。Offset_Main和Offset_Aux是特性阻抗为50欧姆的相移线，负责回退功率下的阻抗匹配。在输入端，二分配器SPLIT将输入功率分为两路，主功放输入匹配网络IMN_Main和辅功放输入匹配网络IMN_Aux分别将Main和Aux的输入阻抗匹配到50欧姆，并由相移线Phaseline补偿主路和辅路的相位差。
传统Doherty功率放大器的工作原理可以分为低功率区和高功率区两个区域分析，两个区域以C类辅功放管开启为界。在低功率区，辅功放管处于截止状态，Doherty结构要求Offset_Aux合理设计，使得Z2B尽可能接近开路。由于为开路，此时Z1T＝25欧姆，Doherty结构要求Offset_Main和OMN_Main结合可以将25欧姆变换为尽可能接近ZBO的阻抗Z1，其中ZBO为使得Main在回退功率下取得最大效率的负载阻抗，可以通过Loadpull方法得到。综上，由于在低功率区有Z2B为开路，因此Aux的输出电路与Main的输出电路隔离，等效电路如图2所示。
在高功率区，随着Aux的开启，由于Aux输出信号对Main输出信号的牵引作用，Z1T和Z2T逐渐发生变化，这就是所谓的有源负载牵引作用。在饱和点，这种牵引作用的效果达到最大，此时Doherty功放的工作状态如图3所示。如图3所示，在饱和点，Z1T和Z2T相等，均为50欧姆，而Doherty结构要求OMN_Main和OMN_Aux都可以将50欧姆的负载阻抗分别变换为Z1＝Z2＝ZSat。其中ZSat为使得功放管在饱和功率下可达到最大效率的负载阻抗，也可以由Loadpull方法得到。由于Offset_Main和Offset_Aux的特性阻抗为50欧姆，所以它们对OMN_Main和OMN_Aux的匹配效果没有影响。
由于Aux的阻抗牵引作用，Main在低功率区和高功率区可具有不同的负载阻抗，通过合理的设计OMN_Main和OMN_Aux，以及Offset_Main和Offset_Aux这两段相移线，可使得Main在低功率区和饱和点都具有高效率。与普通AB类功放相比，Doherty功放大大提升了高PAPR激励下的效率。
为了设计出双频doherty功放，不仅要求主辅功放的输入输出匹配网络为双频匹配网 络，还要求相移线为双频相移线，即在两个设计频率上都可以产生所需的相移，同时保持50Ohm的特性阻抗。在目前的双频doherty设计中，常采用如图4所示的双频线，从左到右依次为π型结构、T型结构、耦合线。这些结构虽然可以实现双频相移的效果，但通常尺寸较大，线宽可能会不合理，而且带宽很窄，这些问题限制了它们在毫米波频段的应用。由于很强的分布参数效应，毫米波功放必须用集成电路工艺实现，较大的双频线尺寸意味着较大的芯片面积，大大增加了加工成本。另外，在毫米波频段无源元件的损耗较大，如果双频线尺寸很大，其引入的损耗会严重降低功放的性能，当线宽不合理时影响会更明显。集成电路工艺通常存在加工误差，双频线带宽过窄会导致实际的双频相移严重偏离设计值，使得实际加工出来的双频doherty功放实现不了双频性能。因此，毫米波双频doherty功放的实现难点在于高性能双频相移线的实现，随着5G通信进程的加快，这个问题亟待解决。
发明内容
本发明旨在至少在一定程度上解决上述相关技术中的技术问题之一。
为此，本发明的目的在于提出一种基于单频线的毫米波双频Doherty功率放大器，采用普通单频传输线实现双频相移功能，克服了传统双频传输线在毫米波频段存在的尺寸大、损耗高、带宽窄的问题，可以大大提高双频相移线的性能。
为了实现上述目的，本发明的实施例提出了一种基于单频线的毫米波双频Doherty功率放大器，包括：双频功率分配器、主功放双频输入匹配网络、辅功放双频输入匹配网络、主功放管、辅功放管、主功放双频输出匹配网络、辅功放双频输出匹配网络和双频后匹配网络，其中，所述双频功率分配器、主功放双频输入匹配网络、主功放管、主功放双频输出匹配网络和双频后匹配网络依次相连；所述双频功率分配器、辅功放双频输入匹配网络、辅功放管、辅功放双频输出匹配网络和双频后匹配网络依次相连；所述双频功率分配器通过第一预设长度的第一单频传输线与所述主功放双频输入匹配网络相连，所述主功放双频输出匹配网络通过第二预设长度的第二单频传输线与所述双频后匹配网络相连，所述辅功放双频输出匹配网络通过第三预设长度的第三单频传输线与所述双频后匹配网络相连。
另外，根据本发明上述实施例的基于单频线的毫米波双频Doherty功率放大器还可以具有如下附加的技术特征：
在一些示例中，所述双频功率分配器在第一设计频率f1和第二设计频率f2处将输入功率合理分配到主功放和辅功放其中，所述第一设计频率f1小于所述第二设计频率f2。
在一些示例中，当工作在所述第一设计频率f1时，得到第一单频传输线对应的相移ThetaP1、第二单频传输线对应的相移ThetaM1和第三单频传输线对应的相移ThetaA1，其 中，所述ThetaM1在具有第一预设取值范围；当工作在所述第二设计频率f2时，得到第一单频传输线对应的相移ThetaP2、第二单频传输线对应的相移ThetaM2和第三单频传输线对应的相移ThetaA2，其中，所述ThetaM2在具有第二预设取值范围。
在一些示例中，其中，
f2/f1＝k，
ThetaM2＝k*ThetaM1，
k为工作频率比。
在一些示例中，所述第一单频传输线、第二单频传输线和第三单频传输线的特性阻抗均为50欧姆。
在一些示例中，所述第一设计频率f1为28GHz，所述第二设计频率f2为39GHz。
在一些示例中，主功放管和辅功放管相同，其中，主功放偏置为AB类，末级漏极电压为4V；辅功放偏置为C类，末级电压设置为4.5V。
在一些示例中，所述主功放管和辅功放管均采用两级结构，驱动级和功率级的晶体管尺寸分别为2X75um和4X75um。
在一些示例中，所述双频功率分配器为宽非等分威尔金森功分器。
在一些示例中，所述双频后匹配网络为两级微带线结构。
根据本发明实施例的基于单频线的毫米波双频Doherty功率放大器，采用普通单频传输线实现双频相移功能，克服了传统双频传输线在毫米波频段存在的尺寸大、损耗高、带宽窄的问题，可以大大提高双频相移线的性能，助于实现高性能的毫米波双频doherty功率放大器。另外，由于普通传输线的性能对工艺误差不敏感，因此提高了设计的可靠性。
本发明的附加方面和优点将在下面的描述中部分给出，部分将从下面的描述中变得明显，或通过本发明的实践了解到。
附图说明
本发明的上述和/或附加的方面和优点从结合下面附图对实施例的描述中将变得明显和容易理解，其中：
图1是传统Doherty功率放大器的结构示意图；
图2是传统Doherty功率放大器在低功率区的等效电路示意图；
图3是传统Doherty功率放大器在高功率区饱和点的工作状态示意图；
图4传统Doherty功率放大器采用的双频传输线的结构示意图；
图5是根据本发明一个实施例的基于单频线的毫米波双频Doherty功率放大器的结构示 意图；
图6是根据本发明一个实施例基于单频线的毫米波双频Doherty功率放大器工作在第一设计频率f1处的doherty功放结构示意图；
图7是根据本发明一个实施例基于单频线的毫米波双频Doherty功率放大器工作在第二设计频率f2处的doherty功放结构示意图；
图8是根据本发明一个实施例基于单频线的毫米波双频Doherty功率放大器用单频线实现双频相移的示意图；
图9是根据本发明的一个具体实施例的28GHz/39GHz双频doherty功率放大器结构示意图；
图10是根据本发明的一个具体实施例的双频功放的输入匹配电路结构示意图；
图11是根据本发明的一个具体实施例的双频功放的级间匹配电路结构示意图；
图12是根据本发明的一个具体实施例的双频功放的输出匹配电路结构示意图；
图13是根据本发明的一个具体实施例的50欧姆到25欧姆的双频后匹配网络PMN的结构示意图。
具体实施方式
下面详细描述本发明的实施例，所述实施例的示例在附图中示出，其中自始至终相同或类似的标号表示相同或类似的元件或具有相同或类似功能的元件。下面通过参考附图描述的实施例是示例性的，仅用于解释本发明，而不能理解为对本发明的限制。
在本发明的描述中，需要理解的是，术语“中心”、“纵向”、“横向”、“上”、“下”、“前”、“后”、“左”、“右”、“竖直”、“水平”、“顶”、“底”、“内”、“外”等指示的方位或位置关系为基于附图所示的方位或位置关系，仅是为了便于描述本发明和简化描述，而不是指示或暗示所指的装置或元件必须具有特定的方位、以特定的方位构造和操作，因此不能理解为对本发明的限制。此外，术语“第一”、“第二”仅用于描述目的，而不能理解为指示或暗示相对重要性。
在本发明的描述中，需要说明的是，除非另有明确的规定和限定，术语“安装”、“相连”、“连接”应做广义理解，例如，可以是固定连接，也可以是可拆卸连接，或一体地连接；可以是机械连接，也可以是电连接；可以是直接相连，也可以通过中间媒介间接相连，可以是两个元件内部的连通。对于本领域的普通技术人员而言，可以具体情况理解上述术语在本发明中的具体含义。
以下结合附图描述根据本发明实施例的基于单频线的毫米波双频Doherty功率放大器。
图5是根据本发明一个实施例的基于单频线的毫米波双频Doherty功率放大器结构示意图。如图5所示，该基于单频线的毫米波双频Doherty功率放大器包括：双频功率分配器110、主功放双频输入匹配网络120、辅功放双频输入匹配网络130、主功放管140、辅功放管150、主功放双频输出匹配网络160、辅功放双频输出匹配网络170和双频后匹配网络180。
其中，双频功率分配器110、主功放双频输入匹配网络120、主功放管140、主功放双频输出匹配网络160和双频后匹配网络180依次相连；双频功率分配器110、辅功放双频输入匹配网络130、辅功放管150、辅功放双频输出匹配网络170和双频后匹配网络180依次相连；双频功率分配器110通过第一预设长度的第一单频传输线1与主功放双频输入匹配网络120相连，主功放双频输出匹配网络160通过第二预设长度的第二单频传输线2与双频后匹配网络180相连，辅功放双频输出匹配网络170通过第三预设长度的第三单频传输线3与双频后匹配网络180相连。
在本发明的一个实施例中，第一单频传输线1(如图5中的Phaseline)、第二单频传输线2(如图5中的Offset_Main)和第三单频传输线3(如图5中的Offset_Aux)的特性阻抗均为50欧姆。也就是说，图5中所示的Phaseline、Offset_Main、Offset_Aux均是特性阻抗为50Ohm的普通单频传输线，在本方面的实施例中，通过选择合适的长度，它们可以同时在两个设计频率上实现相应的相移。
在本发明的一个实施例中，双频功率分配器110在第一设计频率f1和第二设计频率f2处将输入功率合理分配到主功放和辅功放，其中，第一设计频率f1小于第二设计频率f2，即f2>f1。主辅功放为工作在频率f1和f2处的双频功放，已经独立设计达到最佳性能，此时双频相移线的性能直接决定了双频Doherty功放的性能。
当工作在第一设计频率f1时，得到第一单频传输线1对应的相移ThetaP1、第二单频传输线2对应的相移ThetaM1和第三单频传输线3对应的相移ThetaA1，其中，ThetaM1在具有第一预设取值范围。具体地说，在频率f1处实现的Doherty功放的结构如图6所示，其中ThetaP1、ThetaM1和ThetaA1分别为Phaseline(即第一单频传输线1)、Offset_Main(即第二单频传输线2)和Offset_Aux(即第三单频传输线3)对应的相移。实际上，ThetaP1、ThetaM1和ThetaA1不必是一组确定的值，它们在一定范围内变化时，Doherty功放仍能保持良好的性能。在本发明的一个实施例中，设定ThetaM1在ThetaM11-ThetaM12范围(第一预设取值范围)内变化时，回退效率变化较小，都能保持较高的值。
当工作在第二设计频率f2时，得到第一单频传输线1对应的相移ThetaP2、第二单频传输线2对应的相移ThetaM2和第三单频传输线3对应的相移ThetaA2，其中，ThetaM2 在具有第二预设取值范围。具体地说，在频率f2处实现的Doherty功放的结构如图7所示，其中ThetaP2、ThetaM2和ThetaA2分别为Phaseline(即第一单频传输线1)、Offset_Main(即第二单频传输线2)和Offset_Aux(即第三单频传输线3)对应的相移。同理，可以设定ThetaM2在ThetaM21–ThetaM22范围(即第二预设取值范围)内变化时，回退效率变化较小，都能保持较高的值。
在本发明的一个具体实施例中，以双频Doherty功放中Offset_Main的实现来说明用单频传输线实现双频相移的思路。设定f2/f1＝k，则对于特定长度(第二预设长度)的单频线Offset_Main(第二单频传输线)来说，在f2和f1处相应的相移之比也为k，即ThetaM2＝k*ThetaM1，k为工作频率比，对应图8中的一条斜率为k的直线a。ThetaM1的可行区间为ThetaM11-ThetaM12(即第一预设取值范围)，ThetaM2的可行区间为ThetaM21–ThetaM22(即第一预设取值范围)，再考虑到传输线的周期性，图8中的黑色阴影区即为合理的双频相移区。直线a位于黑色阴影区的部分即为Offset_Main可行的设计区间，如果直线a无法与阴影区相交，可以适当放宽ThetaM1和ThetaM2的可行区间，此时Doherty功放的双频性能可能会有所下降，需要适当平衡f1与f2处的性能。在图8中，可以取ThetaM11作为Offset_Main在f1处的电长度，这样会有最小的物理长度，也可以取相交区域的中间点，这样会有更强的鲁棒性。双频Doherty功放中Offset_Aux(即第三单频传输线)和Phaseline(即第一单频传输线)的设计方法与上述Offset_Main的设计过程相似，此处不再赘述。
按照上述描述的方法设计出来的双频线是特性阻抗为50Ohm的普通传输线，线宽合理，损耗很小，尺寸比传统的双频线小得多，适用于毫米波频段。另外，由于普通传输线的性能对工艺误差不敏感，提高了设计的可靠性。
为了便于更好地理解本发明上述实施例的基于单频线的毫米波双频Doherty功率放大器用单频线实现双频相移的原理，以下结合附图，通过具体的实施例做进一步详细描述。
在本发明的一个具体实施例中，例如，第一设计频率f1为28GHz，第二设计频率f2为39GHz，即以一个可以工作在28GHz和39GHz的双频Doherty功率放大器为例进行描述。
具体地，在本实施例中，例如，双频Doherty功率放大器基于WIN半导体的0.15um pHEMT工艺，主功放管140和辅功放管150相同，即主辅路采用相同的双频功放，其中，主功放偏置为AB类，末级漏极电压为4V；辅功放偏置为C类，末级电压设置为4.5V。进一步地，为了实现合理的增益，双频功放采用两级结构，即主功放管140和辅功放管150均采用两级结构，驱动级和功率级的晶体管尺寸分别为2X75um和4X75um。考虑到工作频率比只有1.4，因此，双频功率分配器110采用宽带非等分威尔金森功分器。在本实施例 中，双频Doherty功率放大器的整体结构如图9所示。
具体地，在28GHz频率设计时，对Offset_Main的电长度进行扫描，得到最佳电长度为65度，在40-90度范围内均保持较高的回退效率。对Offset_Aux进行扫描，得到最佳电长度为150，在145-165度范围内均有较好的开路效果。对Phaseline进行扫描，得到最佳电长度为50，在30-70度范围内，主辅路仍然有较好的功率合成效果。具体设计结果如表1所示。
  Offset_Main Offset_Aux Phaseline
可行区间 40-90度 145-165度 30-70度
最佳值 65度 150度 50度
表1
在39GHz频率，按照同样的方法，得到如下表2所示的设计结果。
  Offset_Main Offset_Aux Phaseline
可行区间 110-140度 20-35度 80-100度
最佳值 125度 25度 95度
表2
根据单频线实现双频相移的原理，同时权衡28GHz和39GHz两个工作频率处的性能，得到如下表3所示双频线的设计结果。
  Offset_Main Offset_Aux Phaseline
28GHz 85度 147度 65度
39GHz 119度 26度 91度
表3
进一步地，在本实施例所采用的工艺中，50Ohm微带线的宽度为70um，三段相移线的物理长度及在两个工作频率处的损耗如下表4所示。可见，本实施例中的三段相移线损耗很低，充分体现了用单频线实现双频相移的优势。另外，微带线便于弯折，可以进一步降低芯片面积。
  Offset_Main Offset_Aux Phaseline
物理长度 877um 1516um 670um
损耗@28GHz 0.07dB 0.1dB 0.05dB
损耗@39GHz 0.09dB 0.16dB 0.07dB
表4
在本实施例中，双频功放的输入、级间、输出匹配网络均为双频匹配网络，同时实现 28GHz和39GHz处的阻抗匹配，分别如图10、11、12所示。由于毫米波频段的电感Q值较低，实际实现时都用微带线代替。
在本实施例中，双频后匹配网络180(即PMN)例如采用两级微带线结构，在两个工作频段内可将50Ohm负载匹配到25Ohm，例如图13所示。
经过实验仿真后得出，本发明上述实施例中描述的双频Doherty功率放大器，在28GHz和39GHz处均有很好的回退和饱和性能。
综上，根据本发明实施例的基于单频线的毫米波双频Doherty功率放大器，采用普通单频传输线实现双频相移功能，克服了传统双频传输线在毫米波频段存在的尺寸大、损耗高、带宽窄的问题，可以大大提高双频相移线的性能，助于实现高性能的毫米波双频Doherty功率放大器。另外，由于普通传输线的性能对工艺误差不敏感，因此提高了设计的可靠性。
在本说明书的描述中，参考术语“一个实施例”、“一些实施例”、“示例”、“具体示例”、或“一些示例”等的描述意指结合该实施例或示例描述的具体特征、结构、材料或者特点包含于本发明的至少一个实施例或示例中。在本说明书中，对上述术语的示意性表述不一定指的是相同的实施例或示例。而且，描述的具体特征、结构、材料或者特点可以在任何的一个或多个实施例或示例中以合适的方式结合。
尽管已经示出和描述了本发明的实施例，本领域的普通技术人员可以理解：在不脱离本发明的原理和宗旨的情况下可以对这些实施例进行多种变化、修改、替换和变型，本发明的范围由权利要求及其等同限定。

Claims (10)

  1. 一种基于单频线的毫米波双频Doherty功率放大器，其特征在于，包括：双频功率分配器、主功放双频输入匹配网络、辅功放双频输入匹配网络、主功放管、辅功放管、主功放双频输出匹配网络、辅功放双频输出匹配网络和双频后匹配网络，其中，
    所述双频功率分配器、主功放双频输入匹配网络、主功放管、主功放双频输出匹配网络和双频后匹配网络依次相连；
    所述双频功率分配器、辅功放双频输入匹配网络、辅功放管、辅功放双频输出匹配网络和双频后匹配网络依次相连；
    所述双频功率分配器通过第一预设长度的第一单频传输线与所述主功放双频输入匹配网络相连，所述主功放双频输出匹配网络通过第二预设长度的第二单频传输线与所述双频后匹配网络相连，所述辅功放双频输出匹配网络通过第三预设长度的第三单频传输线与所述双频后匹配网络相连。
  2. 根据权利要求1所述的基于单频线的毫米波双频Doherty功率放大器，其特征在于，所述双频功率分配器在第一设计频率f1和第二设计频率f2处将输入功率合理分配到主功放和辅功放，其中，所述第一设计频率f1小于所述第二设计频率f2。
  3. 根据权利要求2所述的基于单频线的毫米波双频Doherty功率放大器，其特征在于，
    当工作在所述第一设计频率f1时，得到第一单频传输线对应的相移ThetaP1、第二单频传输线对应的相移ThetaM1和第三单频传输线对应的相移ThetaA1，其中，所述ThetaM1在具有第一预设取值范围；
    当工作在所述第二设计频率f2时，得到第一单频传输线对应的相移ThetaP2、第二单频传输线对应的相移ThetaM2和第三单频传输线对应的相移ThetaA2，其中，所述ThetaM2在具有第二预设取值范围。
  4. 根据权利要求3所述的基于单频线的毫米波双频Doherty功率放大器，其特征在于，其中，
    f2/f1＝k，
    ThetaM2＝k*ThetaM1，
    k为工作频率比。
  5. 根据权利要求1-4任一项所述的基于单频线的毫米波双频Doherty功率放大器，其特征在于，
    所述第一单频传输线、第二单频传输线和第三单频传输线的特性阻抗均为50欧姆。
  6. 根据权利要求2所述的基于单频线的毫米波双频Doherty功率放大器，其特征在于，所述第一设计频率f1为28GHz，所述第二设计频率f2为39GHz。
  7. 根据权利要求6所述的基于单频线的毫米波双频Doherty功率放大器，其特征在于，主功放管和辅功放管相同，其中，
    主功放偏置为AB类，末级漏极电压为4V；
    辅功放偏置为C类，末级电压设置为4.5V。
  8. 根据权利要求7所述的基于单频线的毫米波双频Doherty功率放大器，其特征在于，
    所述主功放管和辅功放管均采用两级结构，驱动级和功率级的晶体管尺寸分别为2X75um和4X75um。
  9. 根据权利要求1所述的基于单频线的毫米波双频Doherty功率放大器，其特征在于，所述双频功率分配器为宽非等分威尔金森功分器。
  10. 根据权利要求1所述的基于单频线的毫米波双频Doherty功率放大器，其特征在于，所述双频后匹配网络为两级微带线结构。
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