WO2018146877A1 - Power supply device and method for controlling power supply device - Google Patents

Power supply device and method for controlling power supply device Download PDF

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Publication number
WO2018146877A1
WO2018146877A1 PCT/JP2017/040246 JP2017040246W WO2018146877A1 WO 2018146877 A1 WO2018146877 A1 WO 2018146877A1 JP 2017040246 W JP2017040246 W JP 2017040246W WO 2018146877 A1 WO2018146877 A1 WO 2018146877A1
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Prior art keywords
switching element
transformer
power supply
mode
current
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PCT/JP2017/040246
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French (fr)
Japanese (ja)
Inventor
圭司 田代
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住友電気工業株式会社
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Application filed by 住友電気工業株式会社 filed Critical 住友電気工業株式会社
Priority to JP2018566756A priority Critical patent/JP6930549B2/en
Publication of WO2018146877A1 publication Critical patent/WO2018146877A1/en

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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac

Definitions

  • the present invention relates to a power supply device and a control method for the power supply device.
  • This application claims priority based on Japanese Patent Application No. 2017-024055 filed on Feb. 13, 2017, and incorporates all the description content described in the above Japanese application.
  • the DC / DC converter includes an insulation type DC / DC converter (active clamp type DC / DC converter) including an active clamp circuit.
  • a series circuit of a primary winding of a transformer and a main switching element is connected to a DC power source, and an active clamp circuit including a capacitor and an auxiliary switching element is provided at both ends of the primary winding. It is connected.
  • a rectifying element in the forward direction and a rectifying element in the reverse direction are connected to the secondary winding.
  • the power supply apparatus includes a transformer, a first switching element connected in series to a primary winding of the transformer, a first capacitor connected in parallel to the first switching element, and the primary winding.
  • a power supply device comprising: a second rectifier element connected in parallel to the rectifier element; and a control unit that controls on / off of the first switching element and the second switching element.
  • the first switching element is turned on, the second switching element is turned off and the transformer is operated in a first mode, and after the first mode, the first switching element is turned on.
  • the transformer and the first capacitor are operated in a third mode to resonate, and the current flowing through the second rectifier element due to the resonance in the third mode is less than or equal to a predetermined threshold, and the control unit When the current flowing through the second rectifying element becomes equal to or less than the threshold value, the first switching element is turned on to shift to the first mode.
  • a control method for a power supply apparatus includes a transformer, a first switching element connected in series to a primary winding of the transformer, a first capacitor connected in parallel to the first switching element, A series circuit of a second switching element and a second capacitor connected in parallel to the primary winding, a first rectifier element connected in series to the secondary winding of the transformer, and the secondary winding And a second rectifying element connected in parallel to the first rectifying element, and a control unit for controlling on / off of the first switching element and the second switching element.
  • the control unit operates in a first mode in which the first switching element is turned on, the second switching element is turned off to excite the transformer, and after the first mode, the first switching element is operated.
  • the switching element is turned off, the second switching element is turned on and the transformer is operated in a second mode to reset the excitation.
  • the first switching element and the second switching element are simultaneously operated.
  • the transformer and the first capacitor are turned off to operate in a third mode, and the current flowing through the second rectifier element due to the resonance in the third mode is set to be equal to or lower than a predetermined threshold value.
  • the control unit turns on the first switching element and shifts to the first mode when the current flowing through the second rectifying element becomes equal to or less than the threshold value.
  • an object of the present disclosure is to provide a power supply apparatus and a control method for the power supply apparatus that can suppress the generation of steep counter electromotive force.
  • the power supply apparatus includes a transformer, a first switching element connected in series to a primary winding of the transformer, a first capacitor connected in parallel to the first switching element, and the primary winding.
  • a power supply device comprising: a second rectifier element connected in parallel to the rectifier element; and a control unit that controls on / off of the first switching element and the second switching element.
  • the first switching element is turned on, the second switching element is turned off and the transformer is operated in a first mode, and after the first mode, the first switching element is turned off. And operating in a second mode in which the second switching element is turned on to reset the excitation of the transformer, and after the second mode, the first switching element and the second switching element are simultaneously turned off and the second switching element is turned off.
  • the transformer and the first capacitor are operated in a third mode to resonate, and the current flowing through the second rectifier element due to the resonance in the third mode is less than or equal to a predetermined threshold, and the control unit When the current flowing through the second rectifying element becomes equal to or less than the threshold value, the first switching element is turned on to shift to the first mode.
  • a control method for a power supply apparatus includes a transformer, a first switching element connected in series to a primary winding of the transformer, a first capacitor connected in parallel to the first switching element, A series circuit of a second switching element and a second capacitor connected in parallel to the primary winding, a first rectifier element connected in series to the secondary winding of the transformer, and the secondary winding And a second rectifying element connected in parallel to the first rectifying element, and a control unit for controlling on / off of the first switching element and the second switching element.
  • the control unit operates in a first mode in which the first switching element is turned on, the second switching element is turned off to excite the transformer, and after the first mode, the first switching element is operated.
  • the switching element is turned off, the second switching element is turned on and the transformer is operated in a second mode to reset the excitation.
  • the first switching element and the second switching element are simultaneously operated.
  • the transformer and the first capacitor are turned off to operate in a third mode, and the current flowing through the second rectifier element due to the resonance in the third mode is set to be equal to or lower than a predetermined threshold value.
  • the control unit turns on the first switching element and shifts to the first mode when the current flowing through the second rectifying element becomes equal to or less than the threshold value.
  • the controller operates in the first mode in which the first switching element is turned on and the second switching element is turned off to excite the transformer.
  • a power supply voltage on the input side is applied to the primary winding of the transformer, and the first rectifying element is turned on to output a predetermined voltage to the output side.
  • the transformer excitation current increases.
  • the control unit operates in the second mode in which the first switching element is turned off and the second switching element is turned on to reset the excitation of the transformer.
  • the transformer voltage becomes negative
  • the first rectifying element is reverse-biased
  • the load current flows through the second rectifying element.
  • the series circuit of the second switching element and the second capacitor constitutes a so-called active clamp circuit.
  • the control unit operates in the third mode in which the first switching element and the second switching element are simultaneously turned off after the second mode to resonate the transformer and the first capacitor. Due to the resonance, the current flowing through the second rectifying element also resonates and begins to decrease once. On the other hand, when the voltage of the transformer becomes positive due to resonance, the first rectifier element becomes forward biased, current flows through the first rectifier element, and the current of the first rectifier element starts to increase once due to resonance.
  • the current flowing through the second rectifying element due to resonance in the third mode is set to be equal to or less than a predetermined threshold value.
  • the predetermined threshold may be, for example, 0A or a small value near 0A. That is, when the current flowing through the second rectifying element once starts to decrease due to resonance, the current decreases to 0 A or near 0 A. Specifically, the excitation inductance of the transformer may be reduced so that the resonance current is about 0A.
  • the control unit turns on the first switching element and shifts to the first mode when the current flowing through the second rectifying element becomes equal to or less than the threshold value.
  • the first switching element is turned on, the potential of the first rectifying element is lowered by a voltage corresponding to the voltage of the first capacitor, and a reverse voltage is applied to the second rectifying element (becomes a reverse bias).
  • a reverse voltage is applied to the second rectifier element, the current flowing through the second rectifier element is equal to or lower than the threshold value, so that the generation of reverse recovery current can be suppressed and the generation of steep back electromotive force can be prevented. Can be suppressed.
  • the power supply device of the present disclosure satisfies the formula 2 ⁇ Im ⁇ n ⁇ Imax.
  • the winding ratio between the primary winding and the secondary winding of the transformer is n: 1
  • the maximum load current is Imax
  • the exciting current of the transformer when transitioning to the third mode is Im.
  • the second rectifying element includes a load current (Il) flowing through the load, a transformer exciting current (Im), a transformer (transformer leakage inductance), and a resonance current (Ir) due to resonance of the first capacitor.
  • the load current Il becomes a constant value by, for example, relatively increasing the inductance of the output side choke coil or the like.
  • the excitation current Im does not change during the dead time (a period in which both the first switching element and the second switching element are off) because the voltage applied to the excitation inductance is almost zero.
  • the resonance current Ir becomes the same value as the excitation current Im at the resonance start timing (however, the direction of the current is opposite) and is canceled out with the excitation current Im.
  • the second rectifier element has the same magnitude as the load current. Current flows.
  • n the turns ratio of the primary winding and the secondary winding of the transformer
  • Ir the turns ratio of the primary winding and the secondary winding of the transformer
  • the current flowing through the second rectifying element based on the excitation inductance of the transformer is set to be equal to or less than the threshold value.
  • the current flowing through the second rectifying element is set to be equal to or less than the threshold value.
  • the characteristic that the amplitude of the resonance current increases when the excitation inductance of the transformer is reduced is utilized. That is, when the excitation inductance of the transformer is reduced, the amplitude can be increased so that when the current flowing through the second rectifying element starts to decrease due to resonance, the current decreases to a threshold value or less. Thereby, it is possible to suppress the generation of steep back electromotive force by adjusting the excitation inductance of the transformer without adding special parts.
  • the second rectifying element includes a synchronous rectifying element.
  • the second rectifying element includes a synchronous rectifying element.
  • a synchronous rectifying element For example, an FET can be used as the synchronous rectifier.
  • the synchronous rectification is a rectification method in which the FET is turned on / off at a necessary timing to perform a rectification operation. Since synchronous rectifying elements such as FETs have a smaller forward voltage than diodes, loss can be reduced. However, the reverse recovery current tends to increase due to the body diode of the FET.
  • FIG. 1 is an explanatory diagram illustrating a first example of a circuit configuration of the power supply device 100 according to the present embodiment.
  • the power supply apparatus 100 includes terminals A and B on the input side and terminals C and D on the output side.
  • a DC power supply (not shown) is connected to the terminals A and B on the input side, and the output side A load is connected to the terminals C and D.
  • the power supply device 100 is, for example, a step-down converter.
  • the power supply apparatus 100 includes a transformer 30, a MOSFET (Metal / Oxide / Semiconductor / Field / Effect / Transistor, hereinafter referred to as "FET") 11 as a first switching element, a capacitor 21 as a first capacitor, and a second switching element.
  • FET 12 capacitor 22 as the second capacitor
  • diode 41 as the first rectifier
  • diode 42 as the second rectifier
  • capacitor 24 inductor 61 (choke coil on the output side)
  • FET 11 and FET 12 are turned on The control part 50 etc. which control / off are provided.
  • the FET 11 and FET 12 each have a body diode.
  • the terminal A is connected to one end of the primary winding 31 of the transformer 30.
  • the other end of the primary winding 31 is connected to the drain of the FET 11.
  • the source of the FET 11 is connected to the terminal B.
  • a capacitor 21 (resonance capacitor) is connected between the drain and source of the FET 11.
  • a series circuit of the FET 12 and the capacitor 22 is connected to both ends of the primary winding 31.
  • a series circuit of the FET 12 and the capacitor 22 constitutes an active clamp circuit.
  • one end of the capacitor 22 is connected to one end of the primary winding 31, and the drain of the FET 12 is connected to the other end of the capacitor 22.
  • the source of the FET 12 is connected to the other end of the primary winding 31.
  • the cathode of the diode 41 is connected to one end of the secondary winding 32 of the transformer 30, and the anode of the diode 41 is connected to the terminal D (ground level).
  • the other end of the secondary winding 32 is connected to the cathode of the diode 42 and one end of the inductor 61.
  • the anode of the diode 42 is connected to the anode of the diode 41.
  • the anodes of the diode 41 and the diode 42 are connected to each other.
  • the present invention is not limited to this, and the cathodes of the diode 41 and the diode 42 are connected to each other. It may be configured.
  • the other end of the inductor 61 is connected to the terminal C.
  • a capacitor 24 is connected between the terminals C and D.
  • the control unit 50 outputs a gate voltage to the gates of the FETs 11 and 12.
  • FIG. 2 is an explanatory diagram illustrating an example of the operation state D1 of the power supply device 100 according to the present embodiment.
  • the control unit 50 turns on the FET 11 and turns off the FET 12.
  • the operation state D1 corresponds to the first mode.
  • the exciting current of the transformer 30 increases and the transformer 30 is excited.
  • the power supply voltage on the input side is applied to the primary winding of the transformer 30, and the voltage of the primary winding becomes positive.
  • the voltage of the secondary winding also becomes positive, the diode 41 becomes conductive, and a predetermined voltage and current are output to the output side.
  • the exciting current of the transformer 30 increases.
  • symbol Lm represents the exciting inductance of the transformer 30, and Ls represents the leakage inductance.
  • a positive voltage is defined when the upper end potential is higher than the lower ends of the primary and secondary windings.
  • FIG. 3 is an explanatory diagram showing an example of the operation state D2 of the power supply apparatus 100 of the present embodiment.
  • the control unit 50 turns off the FET 11, and the FET 12 remains off.
  • the capacitor Cs (21) is charged.
  • the capacitor 21 is also referred to as a capacitor Cs.
  • the voltage of the transformer 30 primary winding and secondary winding
  • FIG. 4 is an explanatory diagram showing an example of the operation state D3 of the power supply device 100 according to the present embodiment.
  • the FET 11 and FET 12 remain off.
  • the diode 41 becomes reverse biased and becomes non-conductive.
  • the load current flowing in the diode 41 flows through the diode 42.
  • FIG. 5 is an explanatory diagram showing an example of the operation state D4 of the power supply device 100 according to the present embodiment.
  • the operating state D4 when the capacitor Cs is charged to a predetermined voltage, the body diode of the FET 12 becomes forward biased, and the excitation current that has flowed through the capacitor Cs flows through the body diode of the FET 12.
  • the control unit 50 turns on the FET 12.
  • the FET 12 When the FET 12 is turned on, the voltage of the capacitor 22 is applied to the transformer 30 in the reverse direction (negative voltage direction), the exciting current of the transformer 30 decreases, and the state of the transformer 30 is reset.
  • the FET 12 is turned on in the operation state D4, the second mode is supported.
  • FIG. 6 is an explanatory diagram showing an example of the operating state D5 of the power supply apparatus 100 according to the present embodiment.
  • the FET 12 is on and the FET 11 is off.
  • the operation state D5 corresponds to the second mode.
  • the exciting current of the transformer 30 is reversed (becomes negative, the current direction is reversed), the energy stored in the capacitor 22 is released, and the energy is stored in the leakage inductance Ls of the transformer 30.
  • FIG. 7 is an explanatory diagram showing an example of the operation state D6 of the power supply apparatus 100 according to the present embodiment.
  • the control unit 50 turns off the FET 12, and the FET 11 remains off.
  • An operation state D6 and an operation state D7 described later correspond to the third mode, and correspond to a dead time (pause period) in which the FET 11 and the FET 12 are simultaneously turned off before the operation state D1 (the FET 11 is turned on).
  • resonance occurs due to the transformer 30 (more specifically, the sum of the leakage inductance Ls and the excitation inductance Lm) and the resonance capacitor Cs.
  • the diode 41 becomes forward biased, a current flows through the diode 41, and the operation state D7 is entered.
  • FIG. 8 is an explanatory diagram showing an example of the operating state D7 of the power supply apparatus 100 of the present embodiment.
  • the FETs 11 and 12 remain off.
  • resonance occurs due to the transformer 30 (more specifically, the leakage inductance Ls) and the capacitor Cs.
  • the diode 42 includes a load current Il (shown by a solid line in the figure) flowing through the load, an excitation current Im (shown by a broken line in the figure) of the transformer 30, and a transformer 30 (leakage inductance Ls of the transformer 30). And a resonance current Ir (indicated by a one-dot chain line in the figure) due to resonance of the capacitor Cs flows.
  • the winding ratio of the transformer 30 is assumed to be 1: 1 for convenience, but the winding ratio of the transformer 30 is not limited to 1: 1.
  • the load current Il flows through the diode 42, the inductor 61 (also referred to as the output side choke coil), and the closed loop of the load.
  • the load current Il becomes a constant value by relatively increasing the inductance of the inductor 61.
  • the exciting current Im flows through the closed loop of the transformer 30 and the diodes 42 and 41. Since the voltage applied to the exciting inductance Lm is almost zero during the dead time (period in which the FET 11 and the FET 12 are simultaneously turned off), the exciting current Im does not change in the operating state D7, and the operating state D6 is The current value Im at the time of termination is maintained.
  • the resonance current Ir is a current due to resonance of the leakage inductance Ls of the transformer 30 and the capacitor Cs, and can be expressed by Expression (1).
  • t is time
  • the resonance current Ir excitation current Im.
  • the directions of the resonance current Ir and the excitation current Im are opposite. That is, at the start timing of the operation state D7, the resonance current Ir has the same value as the excitation current Im (however, the direction of the current is reversed) and is canceled out with the excitation current Im. The same current flows.
  • the resonance current Ir temporarily decreases. That is, the resonance current Ir decreases in a direction that cancels the load current Il, and the current flowing through the diode 42 decreases toward the threshold value as the resonance proceeds.
  • the current of the diode 41 increases as the resonance proceeds.
  • the resonance frequency f can be obtained by the equation 1 / ⁇ 2 ⁇ ⁇ ⁇ ⁇ (Ls ⁇ Cs) ⁇ .
  • the current flowing through the diode 42 due to resonance is equal to or less than a predetermined threshold value.
  • the predetermined threshold may be, for example, 0A or a small value near 0A. That is, when the current flowing through the diode 42 starts to decrease once due to resonance, the current decreases to 0 A or near 0 A. Specifically, the excitation inductance Lm of the transformer 30 may be reduced so that the resonance current is about 0A.
  • the control unit 50 turns on the FET 11. Thereby, the state shifts to an operation state D8 described later.
  • the winding ratio of the transformer 30 is described as 1: 1 for convenience.
  • the winding ratio of the transformer 30 is n: 1
  • the primary side and the secondary side of the transformer 30 are secondary.
  • the voltage may be 1 / n times and the current may be n times.
  • the expression (2) is satisfied with the excitation current Im as small as possible.
  • the condition for satisfying the expression (2) with the smallest exciting current Im can be expressed by the expression (4).
  • the inductor 61 output-side choke coil
  • the ripple current is zero. That's fine.
  • FIG. 9 is an explanatory diagram showing an example of the operating state D8 of the power supply apparatus 100 according to the present embodiment.
  • the operating state D8 when the FET 11 is turned on, the voltage of the transformer 30 decreases by a voltage corresponding to the voltage of the capacitor Cs. For this reason, when the cathode potential of the diode 41 that is forward-biased decreases, the anode potential of the diode 42 decreases, and a reverse voltage is applied to the diode 42 (becomes reverse bias).
  • the current flowing through the diode 42 is equal to or less than the threshold value, so that the generation of reverse recovery current can be suppressed and the generation of steep counter electromotive force can be suppressed.
  • the current flowing through the diode 42 is set to be equal to or less than the threshold value.
  • the characteristic that the amplitude of the resonance current is increased is used. That is, when the exciting inductance Lm of the transformer 30 is reduced, the amplitude can be increased so that when the current flowing through the diode 42 starts to decrease due to resonance, the current decreases below the threshold value.
  • the exciting inductance Lm of the transformer 30 without adding special parts, it is possible to suppress the generation of steep back electromotive force.
  • the value of the excitation inductance Lm may be calculated by calculation, or may be obtained by actual measurement or simulation.
  • the timing at which the current flowing through the diode 42 becomes equal to or lower than the threshold can be obtained based on the current value of the diode 42, the resonance frequency f, and the like.
  • FIG. 10 is an explanatory diagram showing an example of the waveform of each part in the operating state of the power supply apparatus 100 of the present embodiment.
  • the horizontal axis indicates time.
  • the waveforms in FIG. 10 indicate the gate voltage of the FET 12, the current of the diode 42, the current of the diode 42, the gate voltage of the FET 11, and the voltage of the diode 42 in order from the top.
  • the FET 12 is on and the FET 11 is off.
  • the diode 42 is forward biased, and the voltage of the diode 42 is a forward voltage.
  • a load current flows through the diode 42.
  • the diode 41 is reverse-biased and no current flows.
  • the operation state D6 In the operation state D5, when the FET 12 is turned from on to off, the operation state D6 is entered. In the operation state D6, resonance occurs due to the transformer 30 (more specifically, the sum of the leakage inductance Ls and the excitation inductance Lm) and the resonance capacitor Cs.
  • the operation state D7 is set.
  • the current flowing through the diode 42 also resonates and begins to decrease once.
  • the diode 41 becomes forward biased, a current flows through the diode 41, and the current of the diode 41 starts to increase once due to resonance.
  • the FET 11 In the operation state D7, when the current flowing through the diode 42 is equal to or less than the threshold value, the FET 11 is turned on, and the operation state D8 is entered. Since the FET 11 is turned on when the current flowing through the diode 42 becomes less than the threshold value, the forward current flowing when the diode 42 is reverse-biased by turning on the FET 11 is extremely small, so that reverse recovery is achieved. It is possible to suppress the generation of current (current in the negative direction when shifting from the operating state D7 to D8 in FIG. 10). As a result, the electromotive force generated in the diode 42 (the cathode potential with respect to the anode of the diode 42) can be suppressed. As shown in FIG. 10, it can be seen that the fluctuation of the voltage of the diode 42 is relatively small when shifting from the operating state D7 to D8.
  • FIG. 11 is an explanatory diagram showing an example of the waveform of each part in the operating state of the power supply device as a comparative example.
  • the comparative example shows a case where the current flowing through the diode 42 due to resonance is not set to be equal to or less than a predetermined threshold value.
  • the operation state D5 is the same as that in FIG.
  • the operation state D6 In the operation state D5, when the FET 12 is turned from on to off, the operation state D6 is entered. In the operation state D6, resonance occurs due to the transformer 30 (more specifically, the sum of the leakage inductance Ls and the excitation inductance Lm) and the resonance capacitor Cs.
  • the operation state D7 is set.
  • the current flowing through the diode 42 gradually decreases while resonating.
  • the current flowing through the diode 41 gradually increases while resonating.
  • the amplitude of the current flowing through the diode 42 due to resonance is increased so that the current becomes a predetermined threshold value or less, and this current becomes the threshold value or less. Since the FET 11 is turned on at the timing, even if the diode 42 is reverse biased by turning on the FET 11, the reverse recovery current can be reduced, and as a result, the generation of the surge voltage can be suppressed.
  • FIG. 12 is a flowchart showing an example of the processing procedure of the control method of the power supply apparatus 100 of the present embodiment.
  • the control unit 50 turns on the FET 11, turns off the FET 12 (S 11), excites the transformer 30 (S 12), and sets the power supply apparatus 100 to the operation state D 1.
  • the control unit 50 turns off the FET 11 (S13), shifts the power supply device 100 from the operating state D1 to the operating state D2, and charges the capacitor Cs (21). Thereafter, the power supply device 100 is set to the operation states D3 and D4.
  • the control unit 50 turns on the FET 12 at the timing when the body diode of the FET 12 becomes forward biased (S14), and shifts the power supply apparatus 100 to the operation state D5.
  • the control unit 50 releases the energy stored in the capacitor 22 of the active clamp circuit and stores the energy in the leakage inductance Ls (S15).
  • the control unit 50 turns off the FET 12 (S16) and shifts the power supply device 100 to the operation state D6.
  • the control unit 50 generates resonance by the leakage inductance Ls and the resonance capacitor Cs (S17). Thereafter, the power supply device 100 is shifted to the operation state D7.
  • the current flowing through the diode 42 decreases toward 0 A due to resonance.
  • the control unit 50 determines whether or not the current flowing through the diode 42 is equal to or less than a threshold value (S18). Note that the timing at which the current flowing through the diode 42 becomes equal to or less than the threshold value is obtained in advance, and it can also be determined based on whether or not the timing is reached.
  • control unit 50 continues the process of step S18. If the current flowing through the diode 42 is less than or equal to the threshold (YES in S18), the FET 11 is turned on ( S19), the power supply device 100 is shifted to the operation state D8.
  • the control unit 50 determines whether or not to end the process (S20). When it is determined that the process is not ended (NO in S20), the process after step S12 is continued and it is determined that the process is ended (S20). YES), the process is terminated.
  • control unit 50 is configured by, for example, a CPU (processor), a RAM (memory), and the like, and a computer program that defines the procedure of each process as shown in FIG. Is loaded into a RAM (memory), and a computer program is executed by a CPU (processor), whereby a control method of the power supply device 50 can be realized on the computer.
  • a CPU processor
  • RAM memory
  • CPU processor
  • FIG. 13 is an explanatory diagram showing a second example of the circuit configuration of the power supply device 100 of the present embodiment.
  • the difference from the first example shown in FIG. 1 is that FETs 13 and 14 as synchronous rectifier elements are provided instead of the diodes 41 and 42. That is, in the second example, the drain of the FET 13 is connected to one end of the secondary winding 32 of the transformer 30, and the source of the FET 13 is connected to the terminal D (ground level). The other end of the secondary winding 32 is connected to the drain of the FET 14. The source of the FET 14 is connected to the source of the FET 13.
  • the control unit 50 outputs a gate voltage to the gates of the FET 13 and FET 14 and controls on / off of the FET 13 and FET 14.
  • the controller 50 turns on and off the FET 13 and the FET 14 at a necessary timing to perform a synchronous rectification operation.
  • the current flowing through the FET 14 is equal to or lower than the threshold value. It can be reduced and generation of steep back electromotive force can be suppressed.
  • FIG. 14 is an explanatory diagram showing a third example of the circuit configuration of the power supply device 100 of the present embodiment.
  • a current detection unit 70 that detects the load current of the power supply device 100
  • the control unit 50 includes a dead time adjustment unit 51, a calculation unit 52, and the like.
  • the control unit 50 (dead time adjustment unit 51) adjusts the dead time indicating the time from the time when the FET 12 is turned off to the time when the FET 11 is turned on according to the load current detected by the current detection 70.
  • the third example can also be applied to the second example shown in FIG. Details will be described below.
  • FIG. 15 is a time chart showing an example of the switching state of the FETs 11 and 12 of the power supply apparatus 100 of the present embodiment.
  • the time chart illustrated in FIG. 15 is apparent from the description of the first example and the second example described above, but will be described again for convenience.
  • the FET 12 on time is T12
  • the FET 11 on time is T11
  • the dead time from the FET 12 off time to the FET 11 on time is Td1
  • the dead time from the FET 11 off time to the FET 12 on time is dead.
  • Td2 T.
  • T is the switching period of the FET 12. Note that the period T is constant.
  • the on-time T11 of the FET 11 can also be made constant when the rated input voltage of the power supply device 100 does not change.
  • the noticeable dead time is the period of the operating states D6 and D7, which is Td1.
  • FIG. 16 is an explanatory diagram showing an example of the optimum value of dead time due to the load current of the power supply device 100 of the present embodiment.
  • Td1 operation states D6 and D7
  • the current flowing through the diode 42 decreases and increases and decreases repeatedly.
  • the load current is relatively large
  • the time until the current flowing through the diode 42 reaches about 0 A is relatively long.
  • the load current is relatively small
  • the time until the current flowing through the diode 42 reaches about 0 A is relatively short. That is, it can be seen that the optimum value of the dead time Td1 (that is, the time from when the FET 12 is turned off until the current flowing through the diode 42 first reaches around 0 A) changes.
  • FIG. 17 is an explanatory diagram showing a correspondence relationship between the dead time Td1 and the load current of the power supply device 100 according to the present embodiment.
  • the horizontal axis represents the load current (corresponding to the load current detected by the current detection unit 70), and the vertical axis represents the dead time Td1.
  • the dead time Td1 can be set smaller as the load current decreases, and conversely, the dead time Td1 can be set larger as the load current increases.
  • the dead time Td1 is made smaller than the lower limit value, the FET 12 and the FET 11 are simultaneously turned on due to the operation delay of the gate drive circuit, etc., and there is a possibility that an overcurrent flows. .
  • the FET 12 may be turned off at a timing at which the resonance current flowing through the diode 42 does not become around 0 A, and the dead time Td1 is preferably set to the upper limit value or less.
  • the information indicating the correspondence between the dead time Td1 and the load current as shown in FIG. 17 may be stored in a non-volatile memory (not shown) provided inside or outside the control unit 50, or A chart as shown in FIG. 17 may be realized by an arithmetic circuit, and the dead time Td1 may be calculated from the load current.
  • the current detector 70 detects the load current while the FET 11 is on.
  • the current detection unit 70 can detect a current having a value obtained by dividing the load current by the turn ratio of the transformer 30. As a result, the load current is detected from the ON time point to the OFF time point in a certain switching cycle of the FET 12, so that the OFF time point can be determined within the switching cycle and the dead time Td1 can be adjusted.
  • the current detection unit 70 detects the load current a plurality of times while the FET 11 is turned on over a plurality of switching cycles of the FET 11, and the control unit 50 detects the load current detected a plurality of times by the current detection unit 70.
  • the dead time Td1 can be adjusted according to the statistical value.
  • the statistical value can be an average value, for example.
  • FIG. 18 is an explanatory diagram showing an example of a method for adjusting the dead time Td1 of the power supply apparatus 100 according to the present embodiment.
  • the dead time adjusting unit 51 (or the control unit 50) changes the OFF time of the FET 12 to adjust the dead time Td1.
  • the dead time Td1 can be adjusted while keeping T and T11 constant.
  • the dead time adjustment unit 51 increases the dead time Td1 when the load current is large, and shortens the dead time Td1 when the load current is small.
  • the calculation unit 52 calculates the ON time T12 of the FET 12 based on the load current detected by the current detection unit 70 and information indicating the correspondence relationship between the dead time Td1 and the load current illustrated in FIG.
  • the switching element is not limited to a MOSFET, but may be a device such as an IGBT (Insulated Gate Bipolar Transistor).
  • IGBT Insulated Gate Bipolar Transistor
  • the switching element is a MOSFET as in the present embodiment, there is an equivalently incorporated body diode between the drain and source.
  • a bipolar transistor is used as the switching element, a diode may be connected in antiparallel between the collector and emitter of the transistor.
  • the current detection unit 70 is provided between one end of the primary winding 31 of the transformer 30 and the input terminal A, and the current is detected when the FET 11 is on.
  • the present invention is not limited to this.
  • a current detector 70 may be provided between one end of the inductor 61 and the output terminal C so that the current is detected when the FET 12 is on.

Abstract

According to the present invention, a control unit: activates a first mode, in which a first switching element is turned on and a second switching element is turned off, thereby exciting a transformer; activates, after the first mode, a second mode in which the first switching element is turned off and the second switching element is turned on, thereby resetting the transformer; and activates, after the second mode, a third mode in which the first switching element and the second switching element are both turned off at the same time, thereby resonating the transformer and a first capacitor, wherein the current flowing to a second rectifying element by means of the resonance in the third mode is set to be equal to or less than a predetermined threshold, and the control unit turns on the first switching element when the current flowing to the second rectifying element is equal to or less than the threshold.

Description

電源装置及び電源装置の制御方法Power supply device and control method of power supply device
　本発明は、電源装置及び電源装置の制御方法に関する。
　本出願は、２０１７年２月１３日出願の日本出願第２０１７－０２４０５５号に基づく優先権を主張し、前記日本出願に記載された全ての記載内容を援用するものである。 The present invention relates to a power supply device and a control method for the power supply device.
This application claims priority based on Japanese Patent Application No. 2017-024055 filed on Feb. 13, 2017, and incorporates all the description content described in the above Japanese application.
　直流電圧を変換するＤＣ／ＤＣコンバータが産業用機器及び車載装置に用いられている。ＤＣ／ＤＣコンバータには、アクティブクランプ回路を備える絶縁型のＤＣ／ＤＣコンバータ（アクティブクランプ型のＤＣ／ＤＣコンバータ）がある。 DC / DC converters that convert DC voltage are used in industrial equipment and in-vehicle devices. The DC / DC converter includes an insulation type DC / DC converter (active clamp type DC / DC converter) including an active clamp circuit.
　アクティブクランプ型のＤＣ／ＤＣコンバータは、トランスの１次巻線と主スイッチング素子との直列回路が直流電源に接続され、１次巻線の両端にキャパシタと補助スイッチング素子とからなるアクティブクランプ回路が接続されている。２次巻線には、順方向の整流素子と逆方向の整流素子が接続されている。主スイッチング素子と補助スイッチング素子とを交互にオン／オフすることによって、トランスの磁化エネルギー及び漏れエネルギーをアクティブクランプ回路のキャパシタを介して循環させ、電源変換効率を向上させることができる（特許文献１参照）。 In the active clamp type DC / DC converter, a series circuit of a primary winding of a transformer and a main switching element is connected to a DC power source, and an active clamp circuit including a capacitor and an auxiliary switching element is provided at both ends of the primary winding. It is connected. A rectifying element in the forward direction and a rectifying element in the reverse direction are connected to the secondary winding. By alternately turning on and off the main switching element and the auxiliary switching element, it is possible to circulate the magnetizing energy and leakage energy of the transformer through the capacitor of the active clamp circuit, thereby improving the power conversion efficiency (Patent Document 1). reference).
特開２００９－２９０９３２号公報JP 2009-290932 A
　本開示の電源装置は、トランスと、該トランスの一次巻線に直列に接続された第１のスイッチング素子と、該第１のスイッチング素子に並列に接続された第１のキャパシタと、前記一次巻線に並列に接続された第２のスイッチング素子及び第２のキャパシタの直列回路と、前記トランスの二次巻線に直列に接続された第１の整流素子と、前記二次巻線及び第１の整流素子に対して並列に接続された第２の整流素子と、前記第１のスイッチング素子及び第２のスイッチング素子のオン／オフを制御する制御部とを備える電源装置であって、前記制御部は、前記第１のスイッチング素子をオンにし、前記第２のスイッチング素子をオフにして前記トランスを励磁する第１モードで動作させ、該第１モード後に、前記第１のスイッチング素子をオフにし、前記第２のスイッチング素子をオンにして前記トランスの励磁をリセットする第２モードで動作させ、該第２モード後に、前記第１のスイッチング素子及び第２のスイッチング素子を同時にオフにして前記トランス及び第１のキャパシタを共振させる第３モードで動作させ、該第３モードにて前記共振により前記第２の整流素子に流れる電流が所定の閾値以下となるようにしてあり、前記制御部は、前記第２の整流素子に流れる電流が前記閾値以下となった場合に、前記第１のスイッチング素子をオンにして前記第１モードに移行する。 The power supply apparatus according to the present disclosure includes a transformer, a first switching element connected in series to a primary winding of the transformer, a first capacitor connected in parallel to the first switching element, and the primary winding. A series circuit of a second switching element and a second capacitor connected in parallel to the line, a first rectifier element connected in series to the secondary winding of the transformer, the secondary winding and the first A power supply device comprising: a second rectifier element connected in parallel to the rectifier element; and a control unit that controls on / off of the first switching element and the second switching element. The first switching element is turned on, the second switching element is turned off and the transformer is operated in a first mode, and after the first mode, the first switching element is turned on. And operating in a second mode in which the second switching element is turned on to reset the excitation of the transformer, and after the second mode, the first switching element and the second switching element are simultaneously turned off and the second switching element is turned off. The transformer and the first capacitor are operated in a third mode to resonate, and the current flowing through the second rectifier element due to the resonance in the third mode is less than or equal to a predetermined threshold, and the control unit When the current flowing through the second rectifying element becomes equal to or less than the threshold value, the first switching element is turned on to shift to the first mode.
　本開示の電源装置の制御方法は、トランスと、該トランスの一次巻線に直列に接続された第１のスイッチング素子と、該第１のスイッチング素子に並列に接続された第１のキャパシタと、前記一次巻線に並列に接続された第２のスイッチング素子及び第２のキャパシタの直列回路と、前記トランスの二次巻線に直列に接続された第１の整流素子と、前記二次巻線及び第１の整流素子に対して並列に接続された第２の整流素子と、前記第１のスイッチング素子及び第２のスイッチング素子のオン／オフを制御する制御部とを備える電源装置の制御方法であって、前記制御部は、前記第１のスイッチング素子をオンにし、前記第２のスイッチング素子をオフにして前記トランスを励磁する第１モードで動作させ、該第１モード後に、前記第１のスイッチング素子をオフにし、前記第２のスイッチング素子をオンにして前記トランスの励磁をリセットする第２モードで動作させ、該第２モード後に、前記第１のスイッチング素子及び第２のスイッチング素子を同時にオフにして前記トランス及び第１のキャパシタを共振させる第３モードで動作させ、該第３モードにて前記共振により前記第２の整流素子に流れる電流が所定の閾値以下となるようにしてあり、前記制御部は、前記第２の整流素子に流れる電流が前記閾値以下となった場合に、前記第１のスイッチング素子をオンにして前記第１モードに移行する。 A control method for a power supply apparatus according to the present disclosure includes a transformer, a first switching element connected in series to a primary winding of the transformer, a first capacitor connected in parallel to the first switching element, A series circuit of a second switching element and a second capacitor connected in parallel to the primary winding, a first rectifier element connected in series to the secondary winding of the transformer, and the secondary winding And a second rectifying element connected in parallel to the first rectifying element, and a control unit for controlling on / off of the first switching element and the second switching element. The control unit operates in a first mode in which the first switching element is turned on, the second switching element is turned off to excite the transformer, and after the first mode, the first switching element is operated. of The switching element is turned off, the second switching element is turned on and the transformer is operated in a second mode to reset the excitation. After the second mode, the first switching element and the second switching element are simultaneously operated. The transformer and the first capacitor are turned off to operate in a third mode, and the current flowing through the second rectifier element due to the resonance in the third mode is set to be equal to or lower than a predetermined threshold value. The control unit turns on the first switching element and shifts to the first mode when the current flowing through the second rectifying element becomes equal to or less than the threshold value.
本実施の形態の電源装置の回路構成の第１例を示す説明図である。It is explanatory drawing which shows the 1st example of the circuit structure of the power supply device of this Embodiment. 本実施の形態の電源装置の動作状態Ｄ１の一例を示す説明図である。It is explanatory drawing which shows an example of the operation state D1 of the power supply device of this Embodiment. 本実施の形態の電源装置の動作状態Ｄ２の一例を示す説明図である。It is explanatory drawing which shows an example of the operation state D2 of the power supply device of this Embodiment. 本実施の形態の電源装置の動作状態Ｄ３の一例を示す説明図である。It is explanatory drawing which shows an example of the operation state D3 of the power supply device of this Embodiment. 本実施の形態の電源装置の動作状態Ｄ４の一例を示す説明図である。It is explanatory drawing which shows an example of the operation state D4 of the power supply device of this Embodiment. 本実施の形態の電源装置の動作状態Ｄ５の一例を示す説明図である。It is explanatory drawing which shows an example of the operation state D5 of the power supply device of this Embodiment. 本実施の形態の電源装置の動作状態Ｄ６の一例を示す説明図である。It is explanatory drawing which shows an example of the operation state D6 of the power supply device of this Embodiment. 本実施の形態の電源装置の動作状態Ｄ７の一例を示す説明図である。It is explanatory drawing which shows an example of the operation state D7 of the power supply device of this Embodiment. 本実施の形態の電源装置の動作状態Ｄ８の一例を示す説明図である。It is explanatory drawing which shows an example of the operation state D8 of the power supply device of this Embodiment. 本実施の形態の電源装置の動作状態での各部の波形の一例を示す説明図である。It is explanatory drawing which shows an example of the waveform of each part in the operation state of the power supply device of this Embodiment. 比較例としての電源装置の動作状態での各部の波形の一例を示す説明図である。It is explanatory drawing which shows an example of the waveform of each part in the operation state of the power supply device as a comparative example. 本実施の形態の電源装置の制御方法の処理手順の一例を示すフローチャートである。It is a flowchart which shows an example of the process sequence of the control method of the power supply device of this Embodiment. 本実施の形態の電源装置の回路構成の第２例を示す説明図である。It is explanatory drawing which shows the 2nd example of the circuit structure of the power supply device of this Embodiment. 本実施の形態の電源装置の回路構成の第３例を示す説明図である。It is explanatory drawing which shows the 3rd example of the circuit structure of the power supply device of this Embodiment. 本実施の形態の電源装置のＦＥＴのスイッチング状態の一例を示すタイムチャートである。It is a time chart which shows an example of the switching state of FET of the power supply device of this Embodiment. 本実施の形態の電源装置の負荷電流によるデッドタイムの最適値の一例を示す説明図である。It is explanatory drawing which shows an example of the optimal value of the dead time by the load current of the power supply device of this Embodiment. 本実施の形態の電源装置のデッドタイムと負荷電流との対応関係を示す説明図である。It is explanatory drawing which shows the correspondence of the dead time and load current of the power supply device of this Embodiment. 本実施の形態の電源装置のデッドタイムの調整方法の一例を示す説明図である。It is explanatory drawing which shows an example of the adjustment method of the dead time of the power supply device of this Embodiment.
［本開示が解決しようとする課題］
　しかし、特許文献１のような従来のアクティブクランプ型のＤＣ／ＤＣコンバータでは、整流素子に順方向電流が流れている状態で主スイッチング素子がオンするので、整流素子に逆電圧がかかる。整流素子には、順方向電流に比例した電荷が蓄積されているので、逆電圧が印加されると、蓄積した電荷による逆電流が流れ、逆電流の変化によって急峻な逆起電力が生じ、整流素子の破損やノイズの原因となる。 [Problems to be solved by the present disclosure]
However, in the conventional active clamp type DC / DC converter as disclosed in Patent Document 1, since the main switching element is turned on while a forward current flows through the rectifying element, a reverse voltage is applied to the rectifying element. Since charge proportional to the forward current is stored in the rectifier element, when a reverse voltage is applied, a reverse current flows due to the stored charge, and a steep counter electromotive force is generated due to a change in the reverse current. It may cause damage to elements and noise.
　そこで、本開示は、急峻な逆起電力の発生を抑制することができる電源装置及び電源装置の制御方法を提供することを目的とする。
［本開示の効果］ Therefore, an object of the present disclosure is to provide a power supply apparatus and a control method for the power supply apparatus that can suppress the generation of steep counter electromotive force.
[Effects of the present disclosure]
　本開示によれば、急峻な逆起電力の発生を抑制することができる。 According to the present disclosure, it is possible to suppress the generation of steep back electromotive force.
［本願発明の実施形態の説明］
　本開示の電源装置は、トランスと、該トランスの一次巻線に直列に接続された第１のスイッチング素子と、該第１のスイッチング素子に並列に接続された第１のキャパシタと、前記一次巻線に並列に接続された第２のスイッチング素子及び第２のキャパシタの直列回路と、前記トランスの二次巻線に直列に接続された第１の整流素子と、前記二次巻線及び第１の整流素子に対して並列に接続された第２の整流素子と、前記第１のスイッチング素子及び第２のスイッチング素子のオン／オフを制御する制御部とを備える電源装置であって、前記制御部は、前記第１のスイッチング素子をオンにし、前記第２のスイッチング素子をオフにして前記トランスを励磁する第１モードで動作させ、該第１モード後に、前記第１のスイッチング素子をオフにし、前記第２のスイッチング素子をオンにして前記トランスの励磁をリセットする第２モードで動作させ、該第２モード後に、前記第１のスイッチング素子及び第２のスイッチング素子を同時にオフにして前記トランス及び第１のキャパシタを共振させる第３モードで動作させ、該第３モードにて前記共振により前記第２の整流素子に流れる電流が所定の閾値以下となるようにしてあり、前記制御部は、前記第２の整流素子に流れる電流が前記閾値以下となった場合に、前記第１のスイッチング素子をオンにして前記第１モードに移行する。 [Description of Embodiment of Present Invention]
The power supply apparatus according to the present disclosure includes a transformer, a first switching element connected in series to a primary winding of the transformer, a first capacitor connected in parallel to the first switching element, and the primary winding. A series circuit of a second switching element and a second capacitor connected in parallel to the line, a first rectifier element connected in series to the secondary winding of the transformer, the secondary winding and the first A power supply device comprising: a second rectifier element connected in parallel to the rectifier element; and a control unit that controls on / off of the first switching element and the second switching element. The first switching element is turned on, the second switching element is turned off and the transformer is operated in a first mode, and after the first mode, the first switching element is turned off. And operating in a second mode in which the second switching element is turned on to reset the excitation of the transformer, and after the second mode, the first switching element and the second switching element are simultaneously turned off and the second switching element is turned off. The transformer and the first capacitor are operated in a third mode to resonate, and the current flowing through the second rectifier element due to the resonance in the third mode is less than or equal to a predetermined threshold, and the control unit When the current flowing through the second rectifying element becomes equal to or less than the threshold value, the first switching element is turned on to shift to the first mode.
　本開示の電源装置の制御方法は、トランスと、該トランスの一次巻線に直列に接続された第１のスイッチング素子と、該第１のスイッチング素子に並列に接続された第１のキャパシタと、前記一次巻線に並列に接続された第２のスイッチング素子及び第２のキャパシタの直列回路と、前記トランスの二次巻線に直列に接続された第１の整流素子と、前記二次巻線及び第１の整流素子に対して並列に接続された第２の整流素子と、前記第１のスイッチング素子及び第２のスイッチング素子のオン／オフを制御する制御部とを備える電源装置の制御方法であって、前記制御部は、前記第１のスイッチング素子をオンにし、前記第２のスイッチング素子をオフにして前記トランスを励磁する第１モードで動作させ、該第１モード後に、前記第１のスイッチング素子をオフにし、前記第２のスイッチング素子をオンにして前記トランスの励磁をリセットする第２モードで動作させ、該第２モード後に、前記第１のスイッチング素子及び第２のスイッチング素子を同時にオフにして前記トランス及び第１のキャパシタを共振させる第３モードで動作させ、該第３モードにて前記共振により前記第２の整流素子に流れる電流が所定の閾値以下となるようにしてあり、前記制御部は、前記第２の整流素子に流れる電流が前記閾値以下となった場合に、前記第１のスイッチング素子をオンにして前記第１モードに移行する。 A control method for a power supply apparatus according to the present disclosure includes a transformer, a first switching element connected in series to a primary winding of the transformer, a first capacitor connected in parallel to the first switching element, A series circuit of a second switching element and a second capacitor connected in parallel to the primary winding, a first rectifier element connected in series to the secondary winding of the transformer, and the secondary winding And a second rectifying element connected in parallel to the first rectifying element, and a control unit for controlling on / off of the first switching element and the second switching element. The control unit operates in a first mode in which the first switching element is turned on, the second switching element is turned off to excite the transformer, and after the first mode, the first switching element is operated. of The switching element is turned off, the second switching element is turned on and the transformer is operated in a second mode to reset the excitation. After the second mode, the first switching element and the second switching element are simultaneously operated. The transformer and the first capacitor are turned off to operate in a third mode, and the current flowing through the second rectifier element due to the resonance in the third mode is set to be equal to or lower than a predetermined threshold value. The control unit turns on the first switching element and shifts to the first mode when the current flowing through the second rectifying element becomes equal to or less than the threshold value.
　制御部は、第１のスイッチング素子をオンにし、第２のスイッチング素子をオフにしてトランスを励磁する第１モードで動作させる。トランスの１次巻線には、入力側の電源電圧が印加され、第１の整流素子が導通して出力側へ所定の電圧が出力される。トランスの励磁電流は増加する。 The controller operates in the first mode in which the first switching element is turned on and the second switching element is turned off to excite the transformer. A power supply voltage on the input side is applied to the primary winding of the transformer, and the first rectifying element is turned on to output a predetermined voltage to the output side. The transformer excitation current increases.
　制御部は、第１モード後に、第１のスイッチング素子をオフにし、第２のスイッチング素子をオンにしてトランスの励磁をリセットする第２モードで動作させる。なお、第２モードの開始時点では、トランスの電圧が負になり、第１の整流素子は逆バイアスとなり、負荷電流は第２の整流素子を介して流れている。第２のスイッチング素子と第２のキャパシタの直列回路は、いわゆるアクティブクランプ回路を構成する。第２のスイッチング素子をオンにすることにより、第２のキャパシタの電圧がトランスに逆方向に印加され、トランスの励磁電流は減少する。また、第２のキャパシタに蓄えられたエネルギーが放出され、トランスの漏れインダクタンスにエネルギーが蓄積される。 After the first mode, the control unit operates in the second mode in which the first switching element is turned off and the second switching element is turned on to reset the excitation of the transformer. At the start of the second mode, the transformer voltage becomes negative, the first rectifying element is reverse-biased, and the load current flows through the second rectifying element. The series circuit of the second switching element and the second capacitor constitutes a so-called active clamp circuit. By turning on the second switching element, the voltage of the second capacitor is applied to the transformer in the reverse direction, and the exciting current of the transformer is reduced. Further, the energy stored in the second capacitor is released, and the energy is stored in the leakage inductance of the transformer.
　制御部は、第２モード後に、第１のスイッチング素子及び第２のスイッチング素子を同時にオフにしてトランス及び第１のキャパシタを共振させる第３モードで動作させる。共振により、第２の整流素子に流れる電流も共振し、一旦減少し始める。一方、共振によりトランスの電圧が正になると、第１の整流素子が順バイアスとなり、第１の整流素子に電流が流れ、第１の整流素子の電流は共振により一旦増加し始める。 The control unit operates in the third mode in which the first switching element and the second switching element are simultaneously turned off after the second mode to resonate the transformer and the first capacitor. Due to the resonance, the current flowing through the second rectifying element also resonates and begins to decrease once. On the other hand, when the voltage of the transformer becomes positive due to resonance, the first rectifier element becomes forward biased, current flows through the first rectifier element, and the current of the first rectifier element starts to increase once due to resonance.
　第３モードにて共振により第２の整流素子に流れる電流が所定の閾値以下となるようにしてある。所定の閾値は、例えば、０Ａでもよく、０Ａ付近の小さい値でもよい。すなわち、第２の整流素子に流れる電流が共振により一旦減少し始めたときに、その電流が０Ａ又は０Ａ付近まで減少するようにしてある。具体的には、共振電流が０Ａ程度になるようにトランスの励磁インダクタンスを小さくすればよい。 The current flowing through the second rectifying element due to resonance in the third mode is set to be equal to or less than a predetermined threshold value. The predetermined threshold may be, for example, 0A or a small value near 0A. That is, when the current flowing through the second rectifying element once starts to decrease due to resonance, the current decreases to 0 A or near 0 A. Specifically, the excitation inductance of the transformer may be reduced so that the resonance current is about 0A.
　制御部は、第２の整流素子に流れる電流が閾値以下となった場合に、第１のスイッチング素子をオンにして第１モードに移行する。第１のスイッチング素子をオンにすると、第１の整流素子の電位が第１のキャパシタの電圧に相当する電圧だけ低下し、第２の整流素子に逆電圧がかかる（逆バイアスとなる）。第２の整流素子に逆電圧がかかる時点では、第２の整流素子に流れる電流が閾値以下となっているので、逆回復電流の発生を抑制することができ、急峻な逆起電力の発生を抑制することができる。 The control unit turns on the first switching element and shifts to the first mode when the current flowing through the second rectifying element becomes equal to or less than the threshold value. When the first switching element is turned on, the potential of the first rectifying element is lowered by a voltage corresponding to the voltage of the first capacitor, and a reverse voltage is applied to the second rectifying element (becomes a reverse bias). When a reverse voltage is applied to the second rectifier element, the current flowing through the second rectifier element is equal to or lower than the threshold value, so that the generation of reverse recovery current can be suppressed and the generation of steep back electromotive force can be prevented. Can be suppressed.
　本開示の電源装置は、２×Ｉｍ×ｎ≒Ｉｍａｘという式を充足する。ここで、トランスの一次巻線と二次巻線との巻線比をｎ：１とし、最大負荷電流をＩｍａｘとし、第３モードに遷移した際のトランスの励磁電流をＩｍとする。 The power supply device of the present disclosure satisfies the formula 2 × Im × n≈Imax. Here, the winding ratio between the primary winding and the secondary winding of the transformer is n: 1, the maximum load current is Imax, and the exciting current of the transformer when transitioning to the third mode is Im.
　第３モードでは、第２の整流素子には、負荷に流れる負荷電流（Ｉｌ）、トランスの励磁電流（Ｉｍ）、トランス（トランスの漏れインダクタンス）及び第１のキャパシタの共振による共振電流（Ｉｒ）が流れる。負荷電流Ｉｌは、例えば、出力側のチョークコイル等のインダクタンスを比較的大きくすることにより、一定の値となる。励磁電流Ｉｍは、デッドタイム（第１のスイッチング素子及び第２のスイッチング素子の両方がオフの期間）において、励磁インダクタンスに印加される電圧がほぼゼロであるため、変化しない。共振電流Ｉｒは、共振の開始タイミングでは、励磁電流Ｉｍと同じ値となり（ただし、電流の向きは逆）、励磁電流Ｉｍとの間で相殺され、第２の整流素子には負荷電流と同じ大きさの電流が流れる。トランスの一次巻線と二次巻線との巻線比をｎ：１とすると、ｎ（Ｉｒ－Ｉｍ）＋Ｉｌ≒０という関係が成立する。共振電流Ｉｒが、負荷電流を最も打ち消す向きに大きくなるのは、Ｉｒ＝－Ｉｍのときであることを考慮して、最大負荷条件にて、なるべく小さいＩｍで当該関係が成立するための条件は、ｎ（－Ｉｍ－Ｉｍ）＋Ｉｍａｘ≒０となる。励磁電流Ｉｍを小さくすることで、スイッチング素子の損失を低減することができる。 In the third mode, the second rectifying element includes a load current (Il) flowing through the load, a transformer exciting current (Im), a transformer (transformer leakage inductance), and a resonance current (Ir) due to resonance of the first capacitor. Flows. The load current Il becomes a constant value by, for example, relatively increasing the inductance of the output side choke coil or the like. The excitation current Im does not change during the dead time (a period in which both the first switching element and the second switching element are off) because the voltage applied to the excitation inductance is almost zero. The resonance current Ir becomes the same value as the excitation current Im at the resonance start timing (however, the direction of the current is opposite) and is canceled out with the excitation current Im. The second rectifier element has the same magnitude as the load current. Current flows. When the turns ratio of the primary winding and the secondary winding of the transformer is n: 1, a relationship of n (Ir−Im) + Il≈0 is established. Considering that the resonance current Ir increases in the direction that most cancels the load current, when Ir = −Im, the condition for the relationship to be established with the smallest load Im is as follows. , N (−Im−Im) + Imax≈0. By reducing the excitation current Im, the loss of the switching element can be reduced.
　本開示の電源装置は、前記トランスの励磁インダクタンスに基づいて前記第２の整流素子に流れる電流を前記閾値以下とする。 In the power supply device according to the present disclosure, the current flowing through the second rectifying element based on the excitation inductance of the transformer is set to be equal to or less than the threshold value.
　トランスの励磁インダクタンスに基づいて第２の整流素子に流れる電流が閾値以下となるようにしてある。トランスの励磁インダクタンスを小さくすると、共振電流の振幅は大きくなるという特性を利用する。すなわち、トランスの励磁インダクタンスを小さくすると、第２の整流素子に流れていた電流が共振により減少し始めるときに、その電流が閾値以下まで減少するように、振幅を大きくすることができる。これにより、特段の部品を追加することなく、トランスの励磁インダクタンスを調整することにより、急峻な逆起電力の発生を抑制することができる。 Based on the exciting inductance of the transformer, the current flowing through the second rectifying element is set to be equal to or less than the threshold value. The characteristic that the amplitude of the resonance current increases when the excitation inductance of the transformer is reduced is utilized. That is, when the excitation inductance of the transformer is reduced, the amplitude can be increased so that when the current flowing through the second rectifying element starts to decrease due to resonance, the current decreases to a threshold value or less. Thereby, it is possible to suppress the generation of steep back electromotive force by adjusting the excitation inductance of the transformer without adding special parts.
　本開示の電源装置において、前記第２の整流素子は、同期整流素子を含む。 In the power supply device according to the present disclosure, the second rectifying element includes a synchronous rectifying element.
　第２の整流素子は、同期整流素子を含む。同期整流素子は、例えば、ＦＥＴを用いることができる。同期整流は、必要なタイミングでＦＥＴをオン／オフして整流動作をさせる整流方式である。ＦＥＴなどの同期整流素子は、ダイオードに比べて順方向電圧が小さいので、損失を低減することができる。しかし、ＦＥＴのボディダイオードにより逆回復電流が大きくなる傾向がある。 The second rectifying element includes a synchronous rectifying element. For example, an FET can be used as the synchronous rectifier. The synchronous rectification is a rectification method in which the FET is turned on / off at a necessary timing to perform a rectification operation. Since synchronous rectifying elements such as FETs have a smaller forward voltage than diodes, loss can be reduced. However, the reverse recovery current tends to increase due to the body diode of the FET.
　上述の構成により、第２の整流素子に逆電圧がかかる時点では、第２の整流素子に流れる電流が閾値以下となっているので、逆回復電流の発生を抑制することができ、整流動作時の損失を低減することができるとともに、急峻な逆起電力の発生を抑制することができる。 With the above-described configuration, when a reverse voltage is applied to the second rectifier element, the current flowing through the second rectifier element is equal to or lower than the threshold value. Loss and a steep counter electromotive force can be suppressed.
［本願発明の実施形態の詳細］
　以下、本発明の実施の形態を図面に基づいて説明する。図１は本実施の形態の電源装置１００の回路構成の第１例を示す説明図である。本実施の形態の電源装置１００は、入力側の端子Ａ及びＢ、出力側の端子Ｃ及びＤを備え、入力側の端子Ａ及びＢには、直流電源（不図示）が接続され、出力側の端子Ｃ及びＤには負荷が接続される。電源装置１００は、例えば、降圧変換装置である。 [Details of the embodiment of the present invention]
Hereinafter, embodiments of the present invention will be described with reference to the drawings. FIG. 1 is an explanatory diagram illustrating a first example of a circuit configuration of the power supply device 100 according to the present embodiment. The power supply apparatus 100 according to the present embodiment includes terminals A and B on the input side and terminals C and D on the output side. A DC power supply (not shown) is connected to the terminals A and B on the input side, and the output side A load is connected to the terminals C and D. The power supply device 100 is, for example, a step-down converter.
　電源装置１００は、トランス３０、第１のスイッチング素子としてのＭＯＳＦＥＴ（Metal Oxide Semiconductor Field Effect Transistor、以下、「ＦＥＴ」と称する）１１、第１のキャパシタとしてのキャパシタ２１、第２のスイッチング素子としてのＦＥＴ１２、第２のキャパシタとしてのキャパシタ２２、第１の整流素子としてのダイオード４１、第２の整流素子としてのダイオード４２、キャパシタ２４、インダクタ６１（出力側のチョークコイル）、及びＦＥＴ１１、ＦＥＴ１２のオン／オフを制御する制御部５０などを備える。ＦＥＴ１１、ＦＥＴ１２は、それぞれボディダイオードを有する。 The power supply apparatus 100 includes a transformer 30, a MOSFET (Metal / Oxide / Semiconductor / Field / Effect / Transistor, hereinafter referred to as "FET") 11 as a first switching element, a capacitor 21 as a first capacitor, and a second switching element. FET 12, capacitor 22 as the second capacitor, diode 41 as the first rectifier, diode 42 as the second rectifier, capacitor 24, inductor 61 (choke coil on the output side), and FET 11 and FET 12 are turned on The control part 50 etc. which control / off are provided. The FET 11 and FET 12 each have a body diode.
　端子Ａには、トランス３０の１次巻線３１の一端が接続されている。１次巻線３１の他端には、ＦＥＴ１１のドレインが接続されている。ＦＥＴ１１のソースは、端子Ｂに接続されている。ＦＥＴ１１のドレイン・ソース間には、キャパシタ２１（共振用のキャパシタ）が接続されている。 The terminal A is connected to one end of the primary winding 31 of the transformer 30. The other end of the primary winding 31 is connected to the drain of the FET 11. The source of the FET 11 is connected to the terminal B. A capacitor 21 (resonance capacitor) is connected between the drain and source of the FET 11.
　１次巻線３１の両端には、ＦＥＴ１２とキャパシタ２２との直列回路が接続されている。ＦＥＴ１２とキャパシタ２２との直列回路は、アクティブクランプ回路を構成する。 A series circuit of the FET 12 and the capacitor 22 is connected to both ends of the primary winding 31. A series circuit of the FET 12 and the capacitor 22 constitutes an active clamp circuit.
　図１の例では、１次巻線３１の一端にキャパシタ２２の一端が接続され、キャパシタ２２の他端にはＦＥＴ１２のドレインが接続されている。ＦＥＴ１２のソースは、１次巻線３１の他端に接続されている。 In the example of FIG. 1, one end of the capacitor 22 is connected to one end of the primary winding 31, and the drain of the FET 12 is connected to the other end of the capacitor 22. The source of the FET 12 is connected to the other end of the primary winding 31.
　トランス３０の２次巻線３２の一端にはダイオード４１のカソードが接続され、ダイオード４１のアノードは端子Ｄ（接地レベル）に接続されている。２次巻線３２の他端には、ダイオード４２のカソード及びインダクタ６１の一端が接続されている。ダイオード４２のアノードは、ダイオード４１のアノードに接続されている。なお、図１の例では、ダイオード４１、ダイオード４２それぞれのアノード同士が接続された構成となっているが、これに限定されるものではなく、ダイオード４１、ダイオード４２それぞれのカソード同士が接続された構成にしてもよい。 The cathode of the diode 41 is connected to one end of the secondary winding 32 of the transformer 30, and the anode of the diode 41 is connected to the terminal D (ground level). The other end of the secondary winding 32 is connected to the cathode of the diode 42 and one end of the inductor 61. The anode of the diode 42 is connected to the anode of the diode 41. In the example of FIG. 1, the anodes of the diode 41 and the diode 42 are connected to each other. However, the present invention is not limited to this, and the cathodes of the diode 41 and the diode 42 are connected to each other. It may be configured.
　インダクタ６１の他端は端子Ｃに接続されている。端子Ｃ及びＤ間にはキャパシタ２４が接続されている。制御部５０は、ＦＥＴ１１、ＦＥＴ１２のゲートへゲート電圧を出力する。 The other end of the inductor 61 is connected to the terminal C. A capacitor 24 is connected between the terminals C and D. The control unit 50 outputs a gate voltage to the gates of the FETs 11 and 12.
　次に、本実施の形態の電源装置１００の動作について説明する。便宜上、電源装置１００の動作状態を状態Ｄ１からＤ８に分けて説明する。 Next, the operation of the power supply device 100 of the present embodiment will be described. For convenience, the operation state of the power supply apparatus 100 will be described separately from states D1 to D8.
　図２は本実施の形態の電源装置１００の動作状態Ｄ１の一例を示す説明図である。動作状態Ｄ１では、制御部５０は、ＦＥＴ１１をオンにし、ＦＥＴ１２をオフにする。動作状態Ｄ１は第１モードに対応する。動作状態Ｄ１では、トランス３０の励磁電流が増加し、トランス３０を励磁する。トランス３０の１次巻線には、入力側の電源電圧が印加され、１次巻線の電圧は正となる。２次巻線の電圧も正となり、ダイオード４１が導通して出力側へ所定の電圧、電流が出力される。トランス３０の励磁電流は増加する。図中、符号Ｌｍはトランス３０の励磁インダクタンスを表し、Ｌｓは漏れインダクタンスを表す。なお、便宜上、図において、１次巻線及び２次巻線の下端に対して上端の電位が高い場合を正の電圧とする。 FIG. 2 is an explanatory diagram illustrating an example of the operation state D1 of the power supply device 100 according to the present embodiment. In the operation state D1, the control unit 50 turns on the FET 11 and turns off the FET 12. The operation state D1 corresponds to the first mode. In the operating state D1, the exciting current of the transformer 30 increases and the transformer 30 is excited. The power supply voltage on the input side is applied to the primary winding of the transformer 30, and the voltage of the primary winding becomes positive. The voltage of the secondary winding also becomes positive, the diode 41 becomes conductive, and a predetermined voltage and current are output to the output side. The exciting current of the transformer 30 increases. In the figure, symbol Lm represents the exciting inductance of the transformer 30, and Ls represents the leakage inductance. For convenience, in the figure, a positive voltage is defined when the upper end potential is higher than the lower ends of the primary and secondary windings.
　図３は本実施の形態の電源装置１００の動作状態Ｄ２の一例を示す説明図である。動作状態Ｄ２では、制御部５０は、ＦＥＴ１１をオフにし、ＦＥＴ１２はオフのままである。ＦＥＴ１１をオフにすることにより、キャパシタＣｓ（２１）が充電される。なお、キャパシタ２１が共振用のキャパシタであることを表すため、キャパシタ２１をキャパシタＣｓとも称する。キャパシタＣｓの充電に伴ってトランス３０（１次巻線及び２次巻線）の電圧は減少する。 FIG. 3 is an explanatory diagram showing an example of the operation state D2 of the power supply apparatus 100 of the present embodiment. In the operation state D2, the control unit 50 turns off the FET 11, and the FET 12 remains off. By turning off the FET 11, the capacitor Cs (21) is charged. In order to indicate that the capacitor 21 is a resonance capacitor, the capacitor 21 is also referred to as a capacitor Cs. As the capacitor Cs is charged, the voltage of the transformer 30 (primary winding and secondary winding) decreases.
　図４は本実施の形態の電源装置１００の動作状態Ｄ３の一例を示す説明図である。動作状態Ｄ３では、ＦＥＴ１１及びＦＥＴ１２はオフのままである。トランス３０の電圧が減少し、負になると、ダイオード４１は逆バイアスとなり、非導通となる。ダイオード４１に流れていた負荷電流はダイオード４２を介して流れるようになる。 FIG. 4 is an explanatory diagram showing an example of the operation state D3 of the power supply device 100 according to the present embodiment. In the operating state D3, the FET 11 and FET 12 remain off. When the voltage of the transformer 30 decreases and becomes negative, the diode 41 becomes reverse biased and becomes non-conductive. The load current flowing in the diode 41 flows through the diode 42.
　図５は本実施の形態の電源装置１００の動作状態Ｄ４の一例を示す説明図である。動作状態Ｄ４では、キャパシタＣｓが所定電圧まで充電されると、ＦＥＴ１２のボディダイオードが順バイアスとなり、キャパシタＣｓを流れていた励磁電流は、ＦＥＴ１２のボディダイオードを介して流れる。このとき、制御部５０は、ＦＥＴ１２をオンにする。ＦＥＴ１２がオンになると、トランス３０には、キャパシタ２２の電圧が逆方向（負の電圧の方向）に印加され、トランス３０の励磁電流は減少し、トランス３０の励磁をリセットする状態に移行する。動作状態Ｄ４において、ＦＥＴ１２がオンされると、第２モードに対応することになる。 FIG. 5 is an explanatory diagram showing an example of the operation state D4 of the power supply device 100 according to the present embodiment. In the operating state D4, when the capacitor Cs is charged to a predetermined voltage, the body diode of the FET 12 becomes forward biased, and the excitation current that has flowed through the capacitor Cs flows through the body diode of the FET 12. At this time, the control unit 50 turns on the FET 12. When the FET 12 is turned on, the voltage of the capacitor 22 is applied to the transformer 30 in the reverse direction (negative voltage direction), the exciting current of the transformer 30 decreases, and the state of the transformer 30 is reset. When the FET 12 is turned on in the operation state D4, the second mode is supported.
　図６は本実施の形態の電源装置１００の動作状態Ｄ５の一例を示す説明図である。動作状態Ｄ５では、ＦＥＴ１２がオンであり、ＦＥＴ１１がオフである。動作状態Ｄ５は第２モードに対応する。動作状態Ｄ５では、トランス３０の励磁電流が逆転し（負になる、電流方向が逆になる）、キャパシタ２２に蓄えられたエネルギーが放出され、トランス３０の漏れインダクタンスＬｓにエネルギーが蓄積される。 FIG. 6 is an explanatory diagram showing an example of the operating state D5 of the power supply apparatus 100 according to the present embodiment. In the operating state D5, the FET 12 is on and the FET 11 is off. The operation state D5 corresponds to the second mode. In the operating state D5, the exciting current of the transformer 30 is reversed (becomes negative, the current direction is reversed), the energy stored in the capacitor 22 is released, and the energy is stored in the leakage inductance Ls of the transformer 30.
　図７は本実施の形態の電源装置１００の動作状態Ｄ６の一例を示す説明図である。動作状態Ｄ６では、制御部５０は、ＦＥＴ１２をオフにし、ＦＥＴ１１はオフのままである。動作状態Ｄ６及び後述の動作状態Ｄ７は第３モードに対応し、動作状態Ｄ１（ＦＥＴ１１がオンになる）に移行する前の、ＦＥＴ１１及びＦＥＴ１２を同時にオフにするデッドタイム（休止期間）に相当する。動作状態Ｄ６では、トランス３０（より具体的には、漏れインダクタンスＬｓと励磁インダクタンスＬｍとの和）及び共振用のキャパシタＣｓによる共振が発生する。キャパシタＣｓの電荷が放電され、キャパシタＣｓの電圧が入力電圧以下になると、ダイオード４１が順バイアスとなり、ダイオード４１に電流が流れ動作状態Ｄ７に移行する。 FIG. 7 is an explanatory diagram showing an example of the operation state D6 of the power supply apparatus 100 according to the present embodiment. In the operation state D6, the control unit 50 turns off the FET 12, and the FET 11 remains off. An operation state D6 and an operation state D7 described later correspond to the third mode, and correspond to a dead time (pause period) in which the FET 11 and the FET 12 are simultaneously turned off before the operation state D1 (the FET 11 is turned on). . In the operation state D6, resonance occurs due to the transformer 30 (more specifically, the sum of the leakage inductance Ls and the excitation inductance Lm) and the resonance capacitor Cs. When the charge of the capacitor Cs is discharged and the voltage of the capacitor Cs becomes equal to or lower than the input voltage, the diode 41 becomes forward biased, a current flows through the diode 41, and the operation state D7 is entered.
　図８は本実施の形態の電源装置１００の動作状態Ｄ７の一例を示す説明図である。動作状態Ｄ７では、ＦＥＴ１１、ＦＥＴ１２はオフのままである。動作状態Ｄ７では、トランス３０（より具体的には、漏れインダクタンスＬｓ）及びキャパシタＣｓによる共振が発生する。 FIG. 8 is an explanatory diagram showing an example of the operating state D7 of the power supply apparatus 100 of the present embodiment. In the operating state D7, the FETs 11 and 12 remain off. In the operating state D7, resonance occurs due to the transformer 30 (more specifically, the leakage inductance Ls) and the capacitor Cs.
　動作状態Ｄ７では、ダイード４２には、負荷に流れる負荷電流Ｉｌ（図中、実線で示す）、トランス３０の励磁電流Ｉｍ（図中、破線で示す）、トランス３０（トランス３０の漏れインダクタンスＬｓ）及びキャパシタＣｓの共振による共振電流Ｉｒ（図中、一点鎖線で示す）が流れる。なお、図８では、便宜上、トランス３０の巻線比を１：１として説明するが、トランス３０の巻線比は１：１に限定されない。 In the operating state D7, the diode 42 includes a load current Il (shown by a solid line in the figure) flowing through the load, an excitation current Im (shown by a broken line in the figure) of the transformer 30, and a transformer 30 (leakage inductance Ls of the transformer 30). And a resonance current Ir (indicated by a one-dot chain line in the figure) due to resonance of the capacitor Cs flows. In FIG. 8, the winding ratio of the transformer 30 is assumed to be 1: 1 for convenience, but the winding ratio of the transformer 30 is not limited to 1: 1.
　負荷電流Ｉｌは、ダイオード４２、インダクタ６１（出力側のチョークコイルとも称する）、負荷の閉ループを流れる。負荷電流Ｉｌは、例えば、インダクタ６１のインダクタンスを比較的大きくすることにより、一定の値となる。 The load current Il flows through the diode 42, the inductor 61 (also referred to as the output side choke coil), and the closed loop of the load. For example, the load current Il becomes a constant value by relatively increasing the inductance of the inductor 61.
　励磁電流Ｉｍは、トランス３０、ダイオード４２、４１の閉ループを流れる。励磁電流Ｉｍは、デッドタイム（ＦＥＴ１１及びＦＥＴ１２が同時にオフの期間）において、励磁インダクタンスＬｍに印加される電圧がほぼゼロであるため、動作状態Ｄ７においては電流値が変化せず、動作状態Ｄ６が終了した時点の電流値Ｉｍが維持される。 The exciting current Im flows through the closed loop of the transformer 30 and the diodes 42 and 41. Since the voltage applied to the exciting inductance Lm is almost zero during the dead time (period in which the FET 11 and the FET 12 are simultaneously turned off), the exciting current Im does not change in the operating state D7, and the operating state D6 is The current value Im at the time of termination is maintained.
　共振電流Ｉｒは、トランス３０の漏れインダクタンスＬｓ及びキャパシタＣｓの共振による電流であり、式（１）で表すことができる。式（１）において、ｔは時間であり、漏れインダクタンスＬｓ及びキャパシタＣｓによる共振の開始時点（動作状態Ｄ７に移行した時点）をｔ＝０とする。 The resonance current Ir is a current due to resonance of the leakage inductance Ls of the transformer 30 and the capacitor Cs, and can be expressed by Expression (1). In the equation (1), t is time, and t = 0 is a starting time of resonance by the leakage inductance Ls and the capacitor Cs (a time point when the operation state is shifted to the operation state D7).
Figure JPOXMLDOC01-appb-M000001
Figure JPOXMLDOC01-appb-M000001
　式（１）から分かるように、動作状態Ｄ７に移行した時点（ｔ＝０）においては、共振電流Ｉｒ＝励磁電流Ｉｍとなる。ただし、図示しているように、共振電流Ｉｒと励磁電流Ｉｍとの電流の向きは逆である。すなわち、動作状態Ｄ７の開始タイミングでは、共振電流Ｉｒは、励磁電流Ｉｍと同じ値となり（ただし、電流の向きは逆）、励磁電流Ｉｍとの間で相殺され、ダイオード４２には負荷電流Ｉｌと同じ大きさの電流が流れる。 As can be seen from the equation (1), at the time of transition to the operating state D7 (t = 0), the resonance current Ir = excitation current Im. However, as shown, the directions of the resonance current Ir and the excitation current Im are opposite. That is, at the start timing of the operation state D7, the resonance current Ir has the same value as the excitation current Im (however, the direction of the current is reversed) and is canceled out with the excitation current Im. The same current flows.
　励磁電流Ｉｍが、十分大きいと見なせる場合、式（２）が成立する。漏れインダクタンスＬｓ及びキャパシタＣｓによる共振が進むにつれて、共振電流Ｉｒは、一旦減少する。すなわち、共振電流Ｉｒは、負荷電流Ｉｌを相殺する方向に減少し、ダイオード４２に流れる電流は、共振が進むにつれて閾値に向かって減少する。一方、ダイオード４１の電流は共振が進むにつれて増加する。 When the exciting current Im can be considered sufficiently large, the formula (2) is established. As the resonance by the leakage inductance Ls and the capacitor Cs progresses, the resonance current Ir temporarily decreases. That is, the resonance current Ir decreases in a direction that cancels the load current Il, and the current flowing through the diode 42 decreases toward the threshold value as the resonance proceeds. On the other hand, the current of the diode 41 increases as the resonance proceeds.
　なお、共振周波数ｆは、１／{２×π×√（Ｌｓ×Ｃｓ）}　という式で求めることができる。 Note that the resonance frequency f can be obtained by the equation 1 / {2 × π × √ (Ls × Cs)}.
　本実施の形態の電源装置１００では、動作状態Ｄ７において、共振によりダイオード４２に流れる電流が所定の閾値以下となるようにしてある。所定の閾値は、例えば、０Ａでもよく、０Ａ付近の小さい値でもよい。すなわち、ダイオード４２に流れる電流が共振により一旦減少し始めたときに、その電流が０Ａ又は０Ａ付近まで減少するようにしてある。具体的には、共振電流が０Ａ程度になるようにトランス３０の励磁インダクタンスＬｍを小さくすればよい。ダイオード４２に流れる電流が閾値以下となったときに、制御部５０は、ＦＥＴ１１をオンにする。これにより、後述の動作状態Ｄ８に移行する。 In the power supply device 100 of the present embodiment, in the operating state D7, the current flowing through the diode 42 due to resonance is equal to or less than a predetermined threshold value. The predetermined threshold may be, for example, 0A or a small value near 0A. That is, when the current flowing through the diode 42 starts to decrease once due to resonance, the current decreases to 0 A or near 0 A. Specifically, the excitation inductance Lm of the transformer 30 may be reduced so that the resonance current is about 0A. When the current flowing through the diode 42 becomes equal to or less than the threshold value, the control unit 50 turns on the FET 11. Thereby, the state shifts to an operation state D8 described later.
　上述の動作状態Ｄ７の説明では、便宜上、トランス３０の巻線比を１：１として説明したが、トランス３０の巻線比がｎ：１の場合には、トランス３０の１次側と２次側とで、電圧を１／ｎ倍とし、電流をｎ倍とすればよい。 In the description of the operation state D7 described above, the winding ratio of the transformer 30 is described as 1: 1 for convenience. However, when the winding ratio of the transformer 30 is n: 1, the primary side and the secondary side of the transformer 30 are secondary. On the side, the voltage may be 1 / n times and the current may be n times.
　トランス３０の巻線比がｎ：１の場合を考慮して、負荷電流Ｉｌが最大負荷電流Ｉｍａｘ（最大負荷条件）となるときに、式（２）が成立する条件を求める。式（１）において、共振電流Ｉｒが最も負荷電流Ｉｌを打ち消す向きに大きくなるので、時間ｔが、式（３）を満たすときである。そのときの共振電流Ｉｒの値は、－Ｉｍである。 In consideration of the case where the winding ratio of the transformer 30 is n: 1, a condition for satisfying Expression (2) is obtained when the load current Il becomes the maximum load current Imax (maximum load condition). In the equation (1), the resonance current Ir increases in the direction that cancels the load current Il most, so the time t satisfies the equation (3). The value of the resonance current Ir at that time is −Im.
Figure JPOXMLDOC01-appb-M000002
  
Figure JPOXMLDOC01-appb-M000002
  
　なお、励磁電流Ｉｍが大きくなると、ＦＥＴ１１、１２の導通損失が大きくなるので、励磁電流Ｉｍが、なるべく小さい値で式（２）が成立することが望ましい。トランス３０の巻線比ｎを考慮し、なるべく小さい励磁電流Ｉｍで式（２）が成立するための条件は、式（４）で表すことができる。なお、式（４）では、便宜上、インダクタ６１（出力側のチョークコイル）のインダクタが十分大きく、リップル電流がゼロとしているが、インダクタ６１のインダクタが小さい場合には、リップルを考慮した値に置き換えればよい。 In addition, since the conduction loss of the FETs 11 and 12 increases as the excitation current Im increases, it is desirable that the expression (2) is satisfied with the excitation current Im as small as possible. In consideration of the winding ratio n of the transformer 30, the condition for satisfying the expression (2) with the smallest exciting current Im can be expressed by the expression (4). In equation (4), for convenience, the inductor 61 (output-side choke coil) has a sufficiently large inductor and the ripple current is zero. That's fine.
　図９は本実施の形態の電源装置１００の動作状態Ｄ８の一例を示す説明図である。動作状態Ｄ８では、ＦＥＴ１１をオンにすると、トランス３０の電圧が、キャパシタＣｓの電圧に相当する電圧だけ低下する。このため、順バイアスされているダイオード４１のカソードの電位が下がることによって、ダイオード４２のアノードの電位が下がり、ダイオード４２に逆電圧がかかる（逆バイアスとなる）。ダイオード４２に逆電圧がかかる時点では、ダイオード４２に流れる電流が閾値以下となっているので、逆回復電流の発生を抑制することができ、急峻な逆起電力の発生を抑制することができる。 FIG. 9 is an explanatory diagram showing an example of the operating state D8 of the power supply apparatus 100 according to the present embodiment. In the operating state D8, when the FET 11 is turned on, the voltage of the transformer 30 decreases by a voltage corresponding to the voltage of the capacitor Cs. For this reason, when the cathode potential of the diode 41 that is forward-biased decreases, the anode potential of the diode 42 decreases, and a reverse voltage is applied to the diode 42 (becomes reverse bias). At the time when the reverse voltage is applied to the diode 42, the current flowing through the diode 42 is equal to or less than the threshold value, so that the generation of reverse recovery current can be suppressed and the generation of steep counter electromotive force can be suppressed.
　トランス３０の励磁インダクタンスＬｍに基づいてダイオード４２に流れる電流が閾値以下となるようにしてある。トランス３０の励磁インダクタンスＬｍを小さくすると、共振電流の振幅は大きくなるという特性を利用する。すなわち、トランス３０の励磁インダクタンスＬｍを小さくすると、ダイオード４２に流れていた電流が共振により減少し始めるときに、その電流が閾値以下まで減少するように、振幅を大きくすることができる。これにより、特段の部品を追加することなく、トランス３０の励磁インダクタンスＬｍを調整することにより、急峻な逆起電力の発生を抑制することができる。 Based on the excitation inductance Lm of the transformer 30, the current flowing through the diode 42 is set to be equal to or less than the threshold value. When the excitation inductance Lm of the transformer 30 is reduced, the characteristic that the amplitude of the resonance current is increased is used. That is, when the exciting inductance Lm of the transformer 30 is reduced, the amplitude can be increased so that when the current flowing through the diode 42 starts to decrease due to resonance, the current decreases below the threshold value. As a result, by adjusting the exciting inductance Lm of the transformer 30 without adding special parts, it is possible to suppress the generation of steep back electromotive force.
　励磁インダクタンスＬｍの値は、計算によって算出してもよく、あるいは実測又はシミュレーション等によって求めてもよい。 The value of the excitation inductance Lm may be calculated by calculation, or may be obtained by actual measurement or simulation.
　ダイオード４２に流れる電流が閾値以下となるタイミングは、例えば、ダイオード４２の電流値、共振周波数ｆなどに基づいて求めることができる。 The timing at which the current flowing through the diode 42 becomes equal to or lower than the threshold can be obtained based on the current value of the diode 42, the resonance frequency f, and the like.
　図１０は本実施の形態の電源装置１００の動作状態での各部の波形の一例を示す説明図である。図１０において、横軸は時間を示す。図１０における波形は、上から順番に、ＦＥＴ１２のゲート電圧、ダイオード４２の電流、ダイオード４２の電流、ＦＥＴ１１のゲート電圧、及びダイオード４２の電圧を示す。 FIG. 10 is an explanatory diagram showing an example of the waveform of each part in the operating state of the power supply apparatus 100 of the present embodiment. In FIG. 10, the horizontal axis indicates time. The waveforms in FIG. 10 indicate the gate voltage of the FET 12, the current of the diode 42, the current of the diode 42, the gate voltage of the FET 11, and the voltage of the diode 42 in order from the top.
　動作状態Ｄ５では、ＦＥＴ１２がオンであり、ＦＥＴ１１はオフである。ダイオード４２は順バイアスされ、ダイオード４２の電圧は順方向電圧となっている。ダイオード４２には負荷電流が流れている。ダイオード４１は逆バイアスされ、電流は流れていない。 In the operation state D5, the FET 12 is on and the FET 11 is off. The diode 42 is forward biased, and the voltage of the diode 42 is a forward voltage. A load current flows through the diode 42. The diode 41 is reverse-biased and no current flows.
　動作状態Ｄ５において、ＦＥＴ１２がオンからオフになることによって動作状態Ｄ６に移行する。動作状態Ｄ６では、トランス３０（より具体的には、漏れインダクタンスＬｓと励磁インダクタンスＬｍとの和）及び共振用のキャパシタＣｓによる共振が発生する。 In the operation state D5, when the FET 12 is turned from on to off, the operation state D6 is entered. In the operation state D6, resonance occurs due to the transformer 30 (more specifically, the sum of the leakage inductance Ls and the excitation inductance Lm) and the resonance capacitor Cs.
　動作状態Ｄ６の後に動作状態Ｄ７となる。動作状態Ｄ７では、共振により、ダイオード４２に流れる電流も共振し、一旦減少し始める。一方、共振によりトランス３０の電圧が正になると、ダイオード４１が順バイアスとなり、ダイオード４１に電流が流れ、ダイオード４１の電流は共振により一旦増加し始める。 After the operation state D6, the operation state D7 is set. In the operating state D7, due to resonance, the current flowing through the diode 42 also resonates and begins to decrease once. On the other hand, when the voltage of the transformer 30 becomes positive due to resonance, the diode 41 becomes forward biased, a current flows through the diode 41, and the current of the diode 41 starts to increase once due to resonance.
　動作状態Ｄ７において、ダイオード４２に流れる電流が閾値以下となった場合に、ＦＥＴ１１をオンにすると、動作状態Ｄ８に移行する。ダイオード４２に流れる電流が閾値以下になったときにＦＥＴ１１をオンにするので、ＦＥＴ１１をオンにすることによってダイオード４２に逆バイアスがかかったときに流れている順方向電流が極めて少ないので、逆回復電流（図１０において、動作状態Ｄ７からＤ８へ移行するときの負方向の電流）の発生を抑制することができる。これによって、ダイオード４２に発生する起電力（ダイオード４２のアノードに対するカソードの電位）を抑制することができる。図１０に示すように、動作状態Ｄ７からＤ８へ移行するときのダイオード４２の電圧の変動は比較的小さいことがわかる。 In the operation state D7, when the current flowing through the diode 42 is equal to or less than the threshold value, the FET 11 is turned on, and the operation state D8 is entered. Since the FET 11 is turned on when the current flowing through the diode 42 becomes less than the threshold value, the forward current flowing when the diode 42 is reverse-biased by turning on the FET 11 is extremely small, so that reverse recovery is achieved. It is possible to suppress the generation of current (current in the negative direction when shifting from the operating state D7 to D8 in FIG. 10). As a result, the electromotive force generated in the diode 42 (the cathode potential with respect to the anode of the diode 42) can be suppressed. As shown in FIG. 10, it can be seen that the fluctuation of the voltage of the diode 42 is relatively small when shifting from the operating state D7 to D8.
　図１１は比較例としての電源装置の動作状態での各部の波形の一例を示す説明図である。比較例は、共振によりダイオード４２に流れる電流が所定の閾値以下となるようにしていない場合を示す。動作状態Ｄ５は、図１０の場合と同様である。 FIG. 11 is an explanatory diagram showing an example of the waveform of each part in the operating state of the power supply device as a comparative example. The comparative example shows a case where the current flowing through the diode 42 due to resonance is not set to be equal to or less than a predetermined threshold value. The operation state D5 is the same as that in FIG.
　動作状態Ｄ５において、ＦＥＴ１２がオンからオフになることによって動作状態Ｄ６に移行する。動作状態Ｄ６では、トランス３０（より具体的には、漏れインダクタンスＬｓと励磁インダクタンスＬｍとの和）及び共振用のキャパシタＣｓによる共振が発生する。 In the operation state D5, when the FET 12 is turned from on to off, the operation state D6 is entered. In the operation state D6, resonance occurs due to the transformer 30 (more specifically, the sum of the leakage inductance Ls and the excitation inductance Lm) and the resonance capacitor Cs.
　動作状態Ｄ６の後に動作状態Ｄ７となる。動作状態Ｄ７では、共振により、ダイオード４２に流れる電流は共振しながら徐々に減少する。一方、ダイオード４１に流れる電流は共振しながら徐々に増加する。 After the operation state D6, the operation state D7 is set. In the operating state D7, due to resonance, the current flowing through the diode 42 gradually decreases while resonating. On the other hand, the current flowing through the diode 41 gradually increases while resonating.
　動作状態Ｄ７の終了時点（すなわち、動作状態Ｄ８の開始時点）において、ＦＥＴ１１をオンにすると、ＦＥＴ１１がオンした時点ではダイオード４２には順方向電流が流れている。このため、ダイオード４２には、順方向電流に比例する電荷が蓄積されている。このため、ダイオード４２の逆回復電流が大きくなり（図１１において、動作状態Ｄ７から動作状態Ｄ８へ移行するときの負方向の電流）、結果としてダイオード４２の逆起電力が非常に大きくなる。図１１に示すように、動作状態Ｄ７から動作状態Ｄ８へ移行するときのダイオード４２の電圧の変動は急峻となり、そのピーク電圧も非常に大きいことがわかる。 When the FET 11 is turned on at the end point of the operation state D7 (that is, the start point of the operation state D8), a forward current flows through the diode 42 when the FET 11 is turned on. For this reason, a charge proportional to the forward current is accumulated in the diode 42. For this reason, the reverse recovery current of the diode 42 becomes large (in FIG. 11, the current in the negative direction when shifting from the operation state D7 to the operation state D8). As a result, the back electromotive force of the diode 42 becomes very large. As shown in FIG. 11, it can be seen that the voltage fluctuation of the diode 42 at the time of transition from the operation state D7 to the operation state D8 is steep, and the peak voltage thereof is very large.
　一方、上述のように、本実施の形態の電源装置１００では、共振によってダイオード４２に流れる電流の振幅を大きくして、電流が所定の閾値以下になるようにし、この電流が閾値以下になったタイミングでＦＥＴ１１をオンにするので、ＦＥＴ１１をオンにすることによってダイオード４２が逆バイアスとなっても、逆回復電流を小さくすることができ、結果としてサージ電圧の発生を抑制することができる。 On the other hand, as described above, in the power supply device 100 according to the present embodiment, the amplitude of the current flowing through the diode 42 due to resonance is increased so that the current becomes a predetermined threshold value or less, and this current becomes the threshold value or less. Since the FET 11 is turned on at the timing, even if the diode 42 is reverse biased by turning on the FET 11, the reverse recovery current can be reduced, and as a result, the generation of the surge voltage can be suppressed.
　図１２は本実施の形態の電源装置１００の制御方法の処理手順の一例を示すフローチャートである。制御部５０は、ＦＥＴ１１をオンにし、ＦＥＴ１２をオフにし（Ｓ１１）、トランス３０を励磁し（Ｓ１２）、電源装置１００を動作状態Ｄ１にする。 FIG. 12 is a flowchart showing an example of the processing procedure of the control method of the power supply apparatus 100 of the present embodiment. The control unit 50 turns on the FET 11, turns off the FET 12 (S 11), excites the transformer 30 (S 12), and sets the power supply apparatus 100 to the operation state D 1.
　　制御部５０は、ＦＥＴ１１をオフにし（Ｓ１３）、電源装置１００を動作状態Ｄ１から動作状態Ｄ２に移行させ、キャパシタＣｓ（２１）を充電する。その後、電源装置１００を動作状態Ｄ３、Ｄ４にする。 The control unit 50 turns off the FET 11 (S13), shifts the power supply device 100 from the operating state D1 to the operating state D2, and charges the capacitor Cs (21). Thereafter, the power supply device 100 is set to the operation states D3 and D4.
　制御部５０は、ＦＥＴ１２のボディダイオードが順バイアスになったタイミングでＦＥＴ１２をオンにし（Ｓ１４）、電源装置１００を動作状態Ｄ５に移行させる。制御部５０は、アクティブクランプ回路のキャパシタ２２に蓄積されたエネルギーを放出し、漏れインダクタンスＬｓにエネルギーを蓄積する（Ｓ１５）。 The control unit 50 turns on the FET 12 at the timing when the body diode of the FET 12 becomes forward biased (S14), and shifts the power supply apparatus 100 to the operation state D5. The control unit 50 releases the energy stored in the capacitor 22 of the active clamp circuit and stores the energy in the leakage inductance Ls (S15).
　制御部５０は、ＦＥＴ１２をオフにし（Ｓ１６）、電源装置１００を動作状態Ｄ６に移行させる。制御部５０は、漏れインダクタンスＬｓと共振用のキャパシタＣｓとによって共振を発生させる（Ｓ１７）。その後、電源装置１００を動作状態Ｄ７に移行させる。ダイオード４２に流れる電流は、共振により０Ａに向かって減少する。 The control unit 50 turns off the FET 12 (S16) and shifts the power supply device 100 to the operation state D6. The control unit 50 generates resonance by the leakage inductance Ls and the resonance capacitor Cs (S17). Thereafter, the power supply device 100 is shifted to the operation state D7. The current flowing through the diode 42 decreases toward 0 A due to resonance.
　制御部５０は、ダイオード４２に流れる電流が閾値以下であるか否かを判定する（Ｓ１８）。なお、ダイオード４２に流れる電流が閾値以下となるタイミングを予め求めておき、そのタイミングになったか否かで判定することもできる。 The control unit 50 determines whether or not the current flowing through the diode 42 is equal to or less than a threshold value (S18). Note that the timing at which the current flowing through the diode 42 becomes equal to or less than the threshold value is obtained in advance, and it can also be determined based on whether or not the timing is reached.
　ダイオード４２に流れる電流が閾値以下でない場合（Ｓ１８でＮＯ）、制御部５０は、ステップＳ１８の処理を続け、ダイオード４２に流れる電流が閾値以下である場合（Ｓ１８でＹＥＳ）、ＦＥＴ１１をオンにし（Ｓ１９）、電源装置１００を動作状態Ｄ８に移行させる。 If the current flowing through the diode 42 is not less than or equal to the threshold (NO in S18), the control unit 50 continues the process of step S18. If the current flowing through the diode 42 is less than or equal to the threshold (YES in S18), the FET 11 is turned on ( S19), the power supply device 100 is shifted to the operation state D8.
　制御部５０は、処理を終了するか否かを判定し（Ｓ２０）、処理を終了しないと判定した場合（Ｓ２０でＮＯ）、ステップＳ１２以降の処理を続け、処理を終了すると判定した場合（Ｓ２０でＹＥＳ）、処理を終了する。 The control unit 50 determines whether or not to end the process (S20). When it is determined that the process is not ended (NO in S20), the process after step S12 is continued and it is determined that the process is ended (S20). YES), the process is terminated.
　本実施の形態の電源装置１００の制御方法は、制御部５０を、例えば、ＣＰＵ（プロセッサ）、ＲＡＭ（メモリ）などで構成し、図１２に示すような、各処理の手順を定めたコンピュータプログラムをＲＡＭ（メモリ）にロードし、コンピュータプログラムをＣＰＵ（プロセッサ）で実行することにより、コンピュータ上で電源装置５０の制御方法を実現することができる。 In the control method of the power supply apparatus 100 according to the present embodiment, the control unit 50 is configured by, for example, a CPU (processor), a RAM (memory), and the like, and a computer program that defines the procedure of each process as shown in FIG. Is loaded into a RAM (memory), and a computer program is executed by a CPU (processor), whereby a control method of the power supply device 50 can be realized on the computer.
　図１３は本実施の形態の電源装置１００の回路構成の第２例を示す説明図である。図１に示す第１例との相違点は、ダイオード４１、４２に代えて、同期整流素子としてのＦＥＴ１３、１４を備える点である。すなわち、第２例では、トランス３０の２次巻線３２の一端にはＦＥＴ１３のドレインが接続され、ＦＥＴ１３のソースは端子Ｄ（接地レベル）に接続されている。２次巻線３２の他端には、ＦＥＴ１４のドレインが接続されている。ＦＥＴ１４のソースは、ＦＥＴ１３のソースに接続されている。 FIG. 13 is an explanatory diagram showing a second example of the circuit configuration of the power supply device 100 of the present embodiment. The difference from the first example shown in FIG. 1 is that FETs 13 and 14 as synchronous rectifier elements are provided instead of the diodes 41 and 42. That is, in the second example, the drain of the FET 13 is connected to one end of the secondary winding 32 of the transformer 30, and the source of the FET 13 is connected to the terminal D (ground level). The other end of the secondary winding 32 is connected to the drain of the FET 14. The source of the FET 14 is connected to the source of the FET 13.
　制御部５０は、ＦＥＴ１３、ＦＥＴ１４のゲートにゲート電圧を出力し、ＦＥＴ１３、ＦＥＴ１４のオン／オフを制御する。制御部５０は、必要なタイミングでＦＥＴ１３、ＦＥＴ１４をオン／オフして同期整流動作をさせる。 The control unit 50 outputs a gate voltage to the gates of the FET 13 and FET 14 and controls on / off of the FET 13 and FET 14. The controller 50 turns on and off the FET 13 and the FET 14 at a necessary timing to perform a synchronous rectification operation.
　ＦＥＴなどの同期整流素子は、ダイオードに比べて順方向電圧が小さいので、損失を低減することができる。しかし、ＦＥＴのボディダイオードにより逆回復電流が大きくなる傾向がある。 Since synchronous rectifying elements such as FETs have a smaller forward voltage than diodes, loss can be reduced. However, the reverse recovery current tends to increase due to the body diode of the FET.
　しかし、本実施の形態によれば、ＦＥＴ１４に逆電圧がかかる時点では、ＦＥＴ１４に流れる電流が閾値以下となっているので、逆回復電流の発生を抑制することができ、整流動作時の損失を低減することができるとともに、急峻な逆起電力の発生を抑制することができる。 However, according to the present embodiment, when the reverse voltage is applied to the FET 14, the current flowing through the FET 14 is equal to or lower than the threshold value. It can be reduced and generation of steep back electromotive force can be suppressed.
　図１４は本実施の形態の電源装置１００の回路構成の第３例を示す説明図である。図１に示す第１例との相違点は、電源装置１００の負荷電流を検出する電流検出部７０を備え、制御部５０は、デッドタイム調整部５１、算出部５２などを備える点である。第３例では、電流検出７０で検出した負荷電流に応じて、制御部５０（デッドタイム調整部５１）は、ＦＥＴ１２のオフ時点からＦＥＴ１１のオン時点までの時間を示すデッドタイムを調整する。なお、第３例は、図１３に示す第２例においても適用することができる。以下、詳細に説明する。 FIG. 14 is an explanatory diagram showing a third example of the circuit configuration of the power supply device 100 of the present embodiment. The difference from the first example shown in FIG. 1 is that a current detection unit 70 that detects the load current of the power supply device 100 is provided, and the control unit 50 includes a dead time adjustment unit 51, a calculation unit 52, and the like. In the third example, the control unit 50 (dead time adjustment unit 51) adjusts the dead time indicating the time from the time when the FET 12 is turned off to the time when the FET 11 is turned on according to the load current detected by the current detection 70. The third example can also be applied to the second example shown in FIG. Details will be described below.
　図１５は本実施の形態の電源装置１００のＦＥＴ１１、１２のスイッチング状態の一例を示すタイムチャートである。図１５に例示するタイムチャートは、前述の第１例及び第２例の説明によって明らかであるが、便宜上、改めて記述することにする。図１５に示すように、ＦＥＴ１２のオン時間をＴ１２とし、ＦＥＴ１１のオン時間をＴ１１とし、ＦＥＴ１２のオフ時点からＦＥＴ１１のオン時点までデッドタイムをＴｄ１とし、ＦＥＴ１１のオフ時点からＦＥＴ１２のオン時点までデッドタイムをＴｄ２とすると、Ｔ１２＋Ｔ１１＋Ｔｄ１＋Ｔｄ２＝Ｔとなる。ＴはＦＥＴ１２のスイッチング周期となる。なお、周期Ｔは一定である。また、ＦＥＴ１１のオン時間Ｔ１１も電源装置１００の定格入力電圧が変わらない場合には一定とすることができる。本実施の形態において、注目すべきデッドタイムは、動作状態Ｄ６及びＤ７の期間であり、Ｔｄ１である。 FIG. 15 is a time chart showing an example of the switching state of the FETs 11 and 12 of the power supply apparatus 100 of the present embodiment. The time chart illustrated in FIG. 15 is apparent from the description of the first example and the second example described above, but will be described again for convenience. As shown in FIG. 15, the FET 12 on time is T12, the FET 11 on time is T11, the dead time from the FET 12 off time to the FET 11 on time is Td1, and the dead time from the FET 11 off time to the FET 12 on time is dead. When the time is Td2, T12 + T11 + Td1 + Td2 = T. T is the switching period of the FET 12. Note that the period T is constant. The on-time T11 of the FET 11 can also be made constant when the rated input voltage of the power supply device 100 does not change. In the present embodiment, the noticeable dead time is the period of the operating states D6 and D7, which is Td1.
　図１６は本実施の形態の電源装置１００の負荷電流によるデッドタイムの最適値の一例を示す説明図である。ＦＥＴ１２のオフ時点からＦＥＴ１１のオン時点までデッドタイムＴｄ１の期間（動作状態Ｄ６及びＤ７）では、図１１に例示したように、ダイオード４２に流れる電流は、共振により減少した後、増減を繰り返す。この場合、図１６に示すように、負荷電流が比較的大きい場合には、ダイオード４２に流れる電流が０Ａ付近になるまでの時間が比較的長くなる。一方、負荷電流が比較的小さい場合には、ダイオード４２に流れる電流が０Ａ付近になるまでの時間が比較的短くなる。すなわち、デッドタイムＴｄ１の最適値（すなわち、ＦＥＴ１２のオフ時点からダイオード４２に流れる電流が最初に０Ａ付近に到達するまでの時間）が変化することが分かる。 FIG. 16 is an explanatory diagram showing an example of the optimum value of dead time due to the load current of the power supply device 100 of the present embodiment. In the period of the dead time Td1 (operation states D6 and D7) from the time when the FET 12 is turned off to the time when the FET 11 is turned on, as illustrated in FIG. 11, the current flowing through the diode 42 decreases and increases and decreases repeatedly. In this case, as shown in FIG. 16, when the load current is relatively large, the time until the current flowing through the diode 42 reaches about 0 A is relatively long. On the other hand, when the load current is relatively small, the time until the current flowing through the diode 42 reaches about 0 A is relatively short. That is, it can be seen that the optimum value of the dead time Td1 (that is, the time from when the FET 12 is turned off until the current flowing through the diode 42 first reaches around 0 A) changes.
　図１７は本実施の形態の電源装置１００のデッドタイムＴｄ１と負荷電流との対応関係を示す説明図である。図１７において、横軸は負荷電流（電流検出部７０で検出する負荷電流に相当）を示し、縦軸はデッドタイムＴｄ１を示す。図１７に示すように、負荷電流の減少に伴ってデッドタイムＴｄ１を小さく設定することができ、逆に、負荷電流の増加に伴ってデッドタイムＴｄ１を大きく設定することができる。しかし、デッドタイムＴｄ１を下限値より小さくするとゲートドライブ回路の動作遅延等によりＦＥＴ１２とＦＥＴ１１とが同時にオン状態となり、過電流が流れるおそれがあるので、デッドタイムＴｄ１は下限値以上とすることが好ましい。また、デッドタイムＴｄ１を上限値より大きくするとダイオード４２に流れる共振電流が０Ａ付近にならないタイミングでＦＥＴ１２をオフにすることになる場合があり、デッドタイムＴｄ１は上限値以下とすることが好ましい。 FIG. 17 is an explanatory diagram showing a correspondence relationship between the dead time Td1 and the load current of the power supply device 100 according to the present embodiment. In FIG. 17, the horizontal axis represents the load current (corresponding to the load current detected by the current detection unit 70), and the vertical axis represents the dead time Td1. As shown in FIG. 17, the dead time Td1 can be set smaller as the load current decreases, and conversely, the dead time Td1 can be set larger as the load current increases. However, if the dead time Td1 is made smaller than the lower limit value, the FET 12 and the FET 11 are simultaneously turned on due to the operation delay of the gate drive circuit, etc., and there is a possibility that an overcurrent flows. . Further, if the dead time Td1 is made larger than the upper limit value, the FET 12 may be turned off at a timing at which the resonance current flowing through the diode 42 does not become around 0 A, and the dead time Td1 is preferably set to the upper limit value or less.
　なお、図１７に示すようなデッドタイムＴｄ１と負荷電流との対応関係を示す情報は、制御部５０の内部又は外部に設けられた不揮発性のメモリ（不図示）に記憶してもよく、あるいは、図１７に示すようなチャートを演算回路で実現して、負荷電流からデッドタイムＴｄ１を算出してもよい。 Note that the information indicating the correspondence between the dead time Td1 and the load current as shown in FIG. 17 may be stored in a non-volatile memory (not shown) provided inside or outside the control unit 50, or A chart as shown in FIG. 17 may be realized by an arithmetic circuit, and the dead time Td1 may be calculated from the load current.
　電流検出部７０は、ＦＥＴ１１がオンしている状態で負荷電流を検出する。電流検出部７０は、負荷電流をトランス３０の巻数比で除算した値の電流を検出することができる。これにより、ＦＥＴ１２のあるスイッチング周期でのオン時点からオフ時点までの間に負荷電流を検出するので、当該スイッチング周期内でオフ時点を決定してデッドタイムＴｄ１を調整することができる。 The current detector 70 detects the load current while the FET 11 is on. The current detection unit 70 can detect a current having a value obtained by dividing the load current by the turn ratio of the transformer 30. As a result, the load current is detected from the ON time point to the OFF time point in a certain switching cycle of the FET 12, so that the OFF time point can be determined within the switching cycle and the dead time Td1 can be adjusted.
　また、電流検出部７０は、ＦＥＴ１１の複数回のスイッチング周期に亘ってＦＥＴ１１がオンしている状態で負荷電流を複数回検出し、制御部５０は、電流検出部７０で複数回検出した負荷電流の統計値に応じてデッドタイムＴｄ１を調整することもできる。統計値は、例えば、平均値とすることができる。複数回検出した負荷電流の統計値を用いることにより、過渡的に負荷電流の増加又は減少が生じた場合でも、より正確に負荷電流の安定的な状態を検出することができるので、より最適なデッドタイムＴｄ１を求めることができ、サージ電圧の発生を抑制することができる。 Further, the current detection unit 70 detects the load current a plurality of times while the FET 11 is turned on over a plurality of switching cycles of the FET 11, and the control unit 50 detects the load current detected a plurality of times by the current detection unit 70. The dead time Td1 can be adjusted according to the statistical value. The statistical value can be an average value, for example. By using the load current statistics that are detected multiple times, even if the load current increases or decreases transiently, the stable state of the load current can be detected more accurately. The dead time Td1 can be obtained, and the generation of a surge voltage can be suppressed.
　図１８は本実施の形態の電源装置１００のデッドタイムＴｄ１の調整方法の一例を示す説明図である。図１８に示すように、デッドタイム調整部５１（制御部５０でもよい）は、ＦＥＴ１２のオフ時点を変更してデッドタイムＴｄ１を調整する。これにより、前述のＴ１２＋Ｔ１１＋Ｔｄ１＋Ｔｄ２＝Ｔという式において、Ｔ及びＴ１１を一定としつつ、デッドタイムＴｄ１を調整することができる。 FIG. 18 is an explanatory diagram showing an example of a method for adjusting the dead time Td1 of the power supply apparatus 100 according to the present embodiment. As shown in FIG. 18, the dead time adjusting unit 51 (or the control unit 50) changes the OFF time of the FET 12 to adjust the dead time Td1. Thereby, in the above-mentioned formula of T12 + T11 + Td1 + Td2 = T, the dead time Td1 can be adjusted while keeping T and T11 constant.
　また、デッドタイム調整部５１は、負荷電流が大きいときは、デッドタイムＴｄ１を長くし、負荷電流が小さいときは、デッドタイムＴｄ１を短くする。 The dead time adjustment unit 51 increases the dead time Td1 when the load current is large, and shortens the dead time Td1 when the load current is small.
　具体的には、算出部５２は、電流検出部７０で検出した負荷電流及び図１７に例示したデッドタイムＴｄ１と負荷電流の対応関係を示す情報に基づいて、ＦＥＴ１２のオン時間Ｔ１２を算出する。 Specifically, the calculation unit 52 calculates the ON time T12 of the FET 12 based on the load current detected by the current detection unit 70 and information indicating the correspondence relationship between the dead time Td1 and the load current illustrated in FIG.
　デッドタイム調整部５１は、算出部５２で算出したオン時間Ｔ１２に応じてデッドタイムＴｄ１を調整する。例えば、前述のＴ１２＋Ｔ１１＋Ｔｄ１＋Ｔｄ２＝Ｔという式において、Ｔ、Ｔ１１及びＴｄ２を一定とすると、Ｔ１２が決まれば、デッドタイムＴｄ１を求めることができる。 The dead time adjustment unit 51 adjusts the dead time Td1 according to the on-time T12 calculated by the calculation unit 52. For example, assuming that T, T11, and Td2 are constant in the above-described equation T12 + T11 + Td1 + Td2 = T, the dead time Td1 can be obtained if T12 is determined.
　本実施の形態によれば、サージ電圧の発生を抑制することができるので、スナバなどの部品を追加する必要がなく、コストの増加又は装置の大型化を避けることができる。 According to this embodiment, since the generation of surge voltage can be suppressed, it is not necessary to add parts such as a snubber, and an increase in cost or an increase in size of the apparatus can be avoided.
　スイッチング素子はＭＯＳＦＥＴに限定されるものではなく、ＩＧＢＴ（Insulated Gate Bipolar Transistor）などのデバイスであってもよい。本実施の形態のように、スイッチング素子が、ＭＯＳＦＥＴの場合には、ドレイン・ソース間には等価的に内蔵されたボディダイオードが存在する。また、スイッチング素子として、バイポーラトランジスタを用いる場合には、トランジスタのコレクタ・エミッタ間にダイオードを逆並列に接続すればよい。 The switching element is not limited to a MOSFET, but may be a device such as an IGBT (Insulated Gate Bipolar Transistor). In the case where the switching element is a MOSFET as in the present embodiment, there is an equivalently incorporated body diode between the drain and source. When a bipolar transistor is used as the switching element, a diode may be connected in antiparallel between the collector and emitter of the transistor.
　図１４の例では、トランス３０の一次巻線３１の一端と入力端Ａとの間に電流検出部７０を設け、ＦＥＴ１１がオンのときに電流を検出する構成であったが、これに限定されない。例えば、インダクタ６１の一端と出力端Ｃとの間に電流検出部７０を設け、ＦＥＴ１２がオンのときに電流を検出するようにしてもよい。 In the example of FIG. 14, the current detection unit 70 is provided between one end of the primary winding 31 of the transformer 30 and the input terminal A, and the current is detected when the FET 11 is on. However, the present invention is not limited to this. . For example, a current detector 70 may be provided between one end of the inductor 61 and the output terminal C so that the current is detected when the FET 12 is on.
　以上に開示された実施の形態及び実施例は、全ての点で例示であって制限的なものではないと考慮されるべきである。本発明の範囲は、以上の実施の形態及び実施例ではなく、特許請求の範囲によって示され、特許請求の範囲と均等の意味及び範囲内での全ての修正や変形を含むものと意図される。 It should be considered that the embodiments and examples disclosed above are illustrative and non-restrictive in every respect. The scope of the present invention is shown not by the above embodiments and examples but by the scope of claims, and is intended to include all modifications and variations within the meaning and scope equivalent to the scope of claims. .
　１１、１２、１３、１４　ＦＥＴ
　２１、２２、２４　キャパシタ
　３０　トランス
　３１　１次巻線
　３２　２次巻線
　４１、４２　ダイオード
　５０　制御部
　５１　デッドタイム調整部
　５２　算出部
　６１　インダクタ
　７０　電流検出部
  11, 12, 13, 14 FET
21, 22, 24 Capacitor 30 Transformer 31 Primary winding 32 Secondary winding 41, 42 Diode 50 Control unit 51 Dead time adjustment unit 52 Calculation unit 61 Inductor 70 Current detection unit

Claims (10)

  1. 　トランスと、該トランスの一次巻線に直列に接続された第１のスイッチング素子と、該第１のスイッチング素子に並列に接続された第１のキャパシタと、前記一次巻線に並列に接続された第２のスイッチング素子及び第２のキャパシタの直列回路と、前記トランスの二次巻線に直列に接続された第１の整流素子と、前記二次巻線及び第１の整流素子に対して並列に接続された第２の整流素子と、前記第１のスイッチング素子及び第２のスイッチング素子のオン／オフを制御する制御部とを備える電源装置であって、
    　前記制御部は、
    　前記第１のスイッチング素子をオンにし、前記第２のスイッチング素子をオフにして前記トランスを励磁する第１モードで動作させ、
    　該第１モード後に、前記第１のスイッチング素子をオフにし、前記第２のスイッチング素子をオンにして前記トランスの励磁をリセットする第２モードで動作させ、
    　該第２モード後に、前記第１のスイッチング素子及び第２のスイッチング素子を同時にオフにして前記トランス及び第１のキャパシタを共振させる第３モードで動作させ、
    　該第３モードにて前記共振により前記第２の整流素子に流れる電流が所定の閾値以下となるようにしてあり、
    　前記制御部は、
    　前記第２の整流素子に流れる電流が前記閾値以下となった場合に、前記第１のスイッチング素子をオンにして前記第１モードに移行する電源装置。     A transformer, a first switching element connected in series to the primary winding of the transformer, a first capacitor connected in parallel to the first switching element, and connected in parallel to the primary winding A series circuit of a second switching element and a second capacitor, a first rectifying element connected in series to the secondary winding of the transformer, and parallel to the secondary winding and the first rectifying element A power supply device comprising: a second rectifying element connected to the control circuit; and a control unit that controls on / off of the first switching element and the second switching element,
    The controller is
    Turning on the first switching element, turning off the second switching element, and operating in a first mode in which the transformer is excited;
    After the first mode, the first switching element is turned off, the second switching element is turned on, and the second mode is operated in the second mode to reset the excitation of the transformer,
    After the second mode, the first switching element and the second switching element are simultaneously turned off to operate in the third mode in which the transformer and the first capacitor resonate,
    In the third mode, the current flowing through the second rectifying element due to the resonance is less than or equal to a predetermined threshold value,
    The controller is
    A power supply device that turns on the first switching element and shifts to the first mode when a current flowing through the second rectifying element becomes equal to or less than the threshold value.
  2. 　２×Ｉｍ×ｎ≒Ｉｍａｘという式を充足する請求項１に記載の電源装置。ここで、前記トランスの一次巻線と二次巻線との巻線比をｎ：１とし、最大負荷電流をＩｍａｘとし、前記第３モードに遷移した際の前記トランスの励磁電流をＩｍとする。 The power supply device according to claim 1, wherein the formula 2 × Im × n≈Imax is satisfied. Here, the winding ratio between the primary winding and the secondary winding of the transformer is n: 1, the maximum load current is Imax, and the exciting current of the transformer at the time of transition to the third mode is Im. .
  3. 　前記トランスの励磁インダクタンスに基づいて前記第２の整流素子に流れる電流を前記閾値以下とする請求項１又は請求項２に記載の電源装置。 3. The power supply device according to claim 1 or 2, wherein a current flowing through the second rectifying element based on an excitation inductance of the transformer is set to be equal to or less than the threshold value.
  4. 　前記第２の整流素子は、同期整流素子を含む請求項１から請求項３のいずれか一項に記載の電源装置。 The power supply apparatus according to any one of claims 1 to 3, wherein the second rectifying element includes a synchronous rectifying element.
  5. 　前記電源装置の負荷電流を検出する電流検出部を備え、
    　前記制御部は、
    　前記電流検出で検出した負荷電流に応じて、前記第２のスイッチング素子のオフ時点から前記第１のスイッチング素子のオン時点までの時間を示すデッドタイムを調整する請求項１から請求項４のいずれか一項に記載の電源装置。     A current detection unit for detecting a load current of the power supply device;
    The controller is
    5. The dead time indicating the time from the time when the second switching element is turned off to the time when the first switching element is turned on is adjusted according to the load current detected by the current detection. The power supply device according to any one of the above.
  6. 　前記電流検出部は、
    　前記第１のスイッチング素子がオンしている状態で負荷電流を検出する請求項５に記載の電源装置。     The current detector is
    The power supply device according to claim 5, wherein a load current is detected in a state where the first switching element is on.
  7. 　前記電流検出部は、
    　前記第１のスイッチング素子の複数回のスイッチング周期に亘って前記第１のスイッチング素子がオンしている状態で負荷電流を複数回検出し、
    　前記制御部は、
    　前記電流検出部で複数回検出した負荷電流の統計値に応じて前記デッドタイムを調整する請求項５又は請求項６に記載の電源装置。     The current detector is
    Detecting a load current a plurality of times while the first switching element is on over a plurality of switching periods of the first switching element;
    The controller is
    The power supply device according to claim 5, wherein the dead time is adjusted according to a statistical value of a load current detected a plurality of times by the current detection unit.
  8. 　前記制御部は、
    　前記第２のスイッチング素子のオフ時点を変更して前記デッドタイムを調整する請求項５から請求項７のいずれか一項に記載の電源装置。     The controller is
    The power supply device according to any one of claims 5 to 7, wherein the dead time is adjusted by changing an OFF time of the second switching element.
  9. 　前記電流検出部で検出した負荷電流及び前記デッドタイムと負荷電流の対応関係を示す情報に基づいて、前記第２のスイッチング素子のオン時間を算出する算出部を備え、
    　前記制御部は、
    　前記算出部で算出したオン時間に応じて前記デッドタイムを調整する請求項５から請求項８のいずれか一項に記載の電源装置。     Based on the load current detected by the current detection unit and information indicating a correspondence relationship between the dead time and the load current, a calculation unit that calculates the on-time of the second switching element,
    The controller is
    The power supply device according to any one of claims 5 to 8, wherein the dead time is adjusted according to an on-time calculated by the calculation unit.
  10. 　トランスと、該トランスの一次巻線に直列に接続された第１のスイッチング素子と、該第１のスイッチング素子に並列に接続された第１のキャパシタと、前記一次巻線に並列に接続された第２のスイッチング素子及び第２のキャパシタの直列回路と、前記トランスの二次巻線に直列に接続された第１の整流素子と、前記二次巻線及び第１の整流素子に対して並列に接続された第２の整流素子と、前記第１のスイッチング素子及び第２のスイッチング素子のオン／オフを制御する制御部とを備える電源装置の制御方法であって、
    　前記制御部は、
    　前記第１のスイッチング素子をオンにし、前記第２のスイッチング素子をオフにして前記トランスを励磁する第１モードで動作させ、
    　該第１モード後に、前記第１のスイッチング素子をオフにし、前記第２のスイッチング素子をオンにして前記トランスの励磁をリセットする第２モードで動作させ、
    　該第２モード後に、前記第１のスイッチング素子及び第２のスイッチング素子を同時にオフにして前記トランス及び第１のキャパシタを共振させる第３モードで動作させ、
    　該第３モードにて前記共振により前記第２の整流素子に流れる電流が所定の閾値以下となるようにしてあり、
    　前記制御部は、
    　前記第２の整流素子に流れる電流が前記閾値以下となった場合に、前記第１のスイッチング素子をオンにして前記第１モードに移行する電源装置の制御方法。
     
          A transformer, a first switching element connected in series to the primary winding of the transformer, a first capacitor connected in parallel to the first switching element, and connected in parallel to the primary winding A series circuit of a second switching element and a second capacitor, a first rectifying element connected in series to the secondary winding of the transformer, and parallel to the secondary winding and the first rectifying element A control method of a power supply device comprising: a second rectifying element connected to the control circuit; and a control unit that controls on / off of the first switching element and the second switching element,
    The controller is
    Turning on the first switching element, turning off the second switching element, and operating in a first mode in which the transformer is excited;
    After the first mode, the first switching element is turned off, the second switching element is turned on, and the second mode is operated in the second mode to reset the excitation of the transformer,
    After the second mode, the first switching element and the second switching element are simultaneously turned off to operate in the third mode in which the transformer and the first capacitor resonate,
    In the third mode, the current flowing through the second rectifying element due to the resonance is less than or equal to a predetermined threshold value,
    The controller is
    A method for controlling a power supply apparatus, wherein when the current flowing through the second rectifier element becomes equal to or less than the threshold value, the first switching element is turned on to shift to the first mode.

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