WO2018072169A1 - 光调制器直流偏置的估计方法、装置以及接收机 - Google Patents

光调制器直流偏置的估计方法、装置以及接收机 Download PDF

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Publication number
WO2018072169A1
WO2018072169A1 PCT/CN2016/102684 CN2016102684W WO2018072169A1 WO 2018072169 A1 WO2018072169 A1 WO 2018072169A1 CN 2016102684 W CN2016102684 W CN 2016102684W WO 2018072169 A1 WO2018072169 A1 WO 2018072169A1
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signal
component
phase noise
receiving end
transmitting end
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PCT/CN2016/102684
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English (en)
French (fr)
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樊洋洋
窦亮
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富士通株式会社
樊洋洋
窦亮
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Priority to CN201680088738.5A priority Critical patent/CN109644045B/zh
Priority to JP2019516966A priority patent/JP6725069B2/ja
Priority to PCT/CN2016/102684 priority patent/WO2018072169A1/zh
Publication of WO2018072169A1 publication Critical patent/WO2018072169A1/zh
Priority to US16/367,384 priority patent/US10574350B2/en

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    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B10/00—Transmission systems employing electromagnetic waves other than radio-waves, e.g. infrared, visible or ultraviolet light, or employing corpuscular radiation, e.g. quantum communication
    • H04B10/07—Arrangements for monitoring or testing transmission systems; Arrangements for fault measurement of transmission systems
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B10/00—Transmission systems employing electromagnetic waves other than radio-waves, e.g. infrared, visible or ultraviolet light, or employing corpuscular radiation, e.g. quantum communication
    • H04B10/07—Arrangements for monitoring or testing transmission systems; Arrangements for fault measurement of transmission systems
    • H04B10/075—Arrangements for monitoring or testing transmission systems; Arrangements for fault measurement of transmission systems using an in-service signal
    • H04B10/079—Arrangements for monitoring or testing transmission systems; Arrangements for fault measurement of transmission systems using an in-service signal using measurements of the data signal
    • H04B10/0799—Monitoring line transmitter or line receiver equipment
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B10/00—Transmission systems employing electromagnetic waves other than radio-waves, e.g. infrared, visible or ultraviolet light, or employing corpuscular radiation, e.g. quantum communication
    • H04B10/50—Transmitters
    • H04B10/501—Structural aspects
    • H04B10/503—Laser transmitters
    • H04B10/505—Laser transmitters using external modulation
    • H04B10/5057—Laser transmitters using external modulation using a feedback signal generated by analysing the optical output
    • H04B10/50575—Laser transmitters using external modulation using a feedback signal generated by analysing the optical output to control the modulator DC bias
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B10/00—Transmission systems employing electromagnetic waves other than radio-waves, e.g. infrared, visible or ultraviolet light, or employing corpuscular radiation, e.g. quantum communication
    • H04B10/50—Transmitters
    • H04B10/516—Details of coding or modulation
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B10/00—Transmission systems employing electromagnetic waves other than radio-waves, e.g. infrared, visible or ultraviolet light, or employing corpuscular radiation, e.g. quantum communication
    • H04B10/50—Transmitters
    • H04B10/564—Power control
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B10/00—Transmission systems employing electromagnetic waves other than radio-waves, e.g. infrared, visible or ultraviolet light, or employing corpuscular radiation, e.g. quantum communication
    • H04B10/60—Receivers
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B10/00—Transmission systems employing electromagnetic waves other than radio-waves, e.g. infrared, visible or ultraviolet light, or employing corpuscular radiation, e.g. quantum communication
    • H04B10/60—Receivers
    • H04B10/61—Coherent receivers
    • H04B10/616—Details of the electronic signal processing in coherent optical receivers
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B10/00—Transmission systems employing electromagnetic waves other than radio-waves, e.g. infrared, visible or ultraviolet light, or employing corpuscular radiation, e.g. quantum communication
    • H04B10/60—Receivers
    • H04B10/66—Non-coherent receivers, e.g. using direct detection
    • H04B10/69—Electrical arrangements in the receiver
    • H04B10/697—Arrangements for reducing noise and distortion
    • H04B10/6971—Arrangements for reducing noise and distortion using equalisation

Definitions

  • the present invention relates to the field of optical communication technologies, and in particular, to a method, an apparatus, and a receiver for estimating a DC offset of an optical modulator.
  • MZM Mach-Zehnder Modulator
  • IQ-MZM in-phase quadrature Mach-Zehnder modulator
  • Modulation is performed separately from the quadrature components.
  • FIG. 1 is a schematic diagram of the structure of an optical modulator in a polarization multiplexing system, showing the case of dual polarization states (represented by h and v).
  • the structure of IQ-MZM is shown in the dotted line of Figure 1.
  • s(t) is the driving signal of MZM
  • D is the optimal DC bias voltage.
  • each MZM in the IQ-MZM typically operates independently at its respective optimal DC bias point.
  • ⁇ represents the DC offset caused by the drift.
  • the DC offset caused by drift can affect the modulation performance, causing damage to system performance.
  • modulation formats such as 16 Quadrature Amplitude Modulation (QAM), 32QAM and even higher order formats, the system is more and more sensitive to the drift of MZM DC bias.
  • the MZM is usually automatically biased at the transmitting end, for example, using a pilot and power based control scheme, and the DC bias cannot be estimated with simple and efficient structure and operation. make up.
  • Embodiments of the present invention provide a method, an apparatus, and a receiver for estimating a DC offset of an optical modulator.
  • digital signal processing DSP, Digital Signal Processing
  • DSP Digital Signal Processing
  • a method for estimating a DC offset of an optical modulator which is applied to a receiving end for converting a received optical signal into an electrical signal, the estimating method comprising:
  • an apparatus for estimating a DC bias of an optical modulator configured to convert a received optical signal into a receiving end of an electrical signal, the estimating apparatus comprising:
  • a signal processing unit that performs signal processing on the electrical signal to obtain a phase noise compensated signal
  • a signal extracting unit that extracts, according to the phase noise compensated signal, a signal component of the receiving end corresponding to the direct current component introduced at the transmitting end due to the offset drift of the transmitting end optical modulator;
  • a signal removing unit that removes a signal component of the receiving end corresponding to the DC component from the phase noise compensated signal
  • a power calculation unit that calculates a received signal power based on a signal from which a signal component of the receiving end corresponding to the DC component is removed;
  • an offset calculation unit that calculates a DC offset of the light modulator of the transmitting end based on a signal component of the receiving end corresponding to the DC component, the power of the received signal, and a driving signal power of the light modulator of the transmitting end.
  • a receiver comprising:
  • a photoelectric converter that converts the received optical signal into an electrical signal
  • a digital signal processor that performs signal processing on the electrical signal to obtain a phase noise compensated signal; and based on the phase noise compensated signal, extracts a signal introduced at the transmitting end due to offset drift of the transmitting end optical modulator a signal component of the receiving end corresponding to the DC component; removing a signal component of the receiving end corresponding to the DC component from the phase noise compensated signal; calculating based on a signal from which the signal component of the receiving end corresponding to the DC component is removed Receiving signal power; calculating a DC offset of the transmitting end optical modulator based on a receiving end signal component corresponding to the DC component, the received signal power, and a driving signal power of the transmitting end optical modulator.
  • the beneficial effects of the embodiment of the present invention are: extracting, based on the phase noise compensated signal, a signal component of the receiving end corresponding to the direct current component introduced at the transmitting end due to the offset drift of the optical modulator of the transmitting end, based on removing the receiving
  • the signal of the end signal component calculates the received signal power; and calculates the DC offset of the transmitting end optical modulator based on the receiving end signal component, the received signal power, and the driving signal power of the transmitting end optical modulator.
  • digital signal processing can be used at the receiving end of the signal to estimate the DC offset due to drift at the transmitting end, and the DC offset can be estimated and compensated with a simple and efficient structure and operation.
  • FIG. 1 is a schematic diagram of a structure of an optical modulator in a polarization multiplexing system
  • FIG. 2 is a schematic diagram of a method for estimating a DC bias of an optical modulator according to Embodiment 1 of the present invention
  • FIG. 3 is another schematic diagram of a method for estimating a DC bias of an optical modulator according to Embodiment 1 of the present invention.
  • FIG. 4 is another schematic diagram of a method for estimating a DC bias of an optical modulator according to Embodiment 1 of the present invention.
  • Figure 5 is a schematic diagram of an apparatus for estimating a DC bias of an optical modulator according to Embodiment 2 of the present invention.
  • FIG. 6 is another schematic diagram of an apparatus for estimating a DC bias of an optical modulator according to Embodiment 2 of the present invention.
  • Figure 7 is a schematic diagram of a receiver in Embodiment 3 of the present invention.
  • Figure 8 is a schematic diagram of an optical communication system according to Embodiment 3 of the present invention.
  • Embodiments of the present invention provide a method for estimating a DC offset of an optical modulator, which is applied to a receiving end that converts a received optical signal into an electrical signal.
  • 2 is a schematic diagram of a method for estimating a DC offset of an optical modulator according to an embodiment of the present invention. As shown in FIG. 2, the estimation method includes:
  • Step 201 Perform signal processing on the electrical signal to obtain a phase noise compensated signal
  • Step 202 Extract, according to the phase noise compensated signal, a signal component of the receiving end corresponding to the DC component introduced at the transmitting end due to the offset drift of the optical modulator of the transmitting end;
  • Step 203 removing a signal component of the receiving end corresponding to the DC component from the phase noise compensated signal
  • Step 204 Calculate a received signal power based on a signal from which a signal component of the receiving end corresponding to the DC component is removed;
  • Step 205 Calculate a DC offset of the optical modulator of the transmitting end based on a signal component of the receiving end corresponding to the DC component, the received signal power, and a driving signal power of the transmitting end optical modulator.
  • a transmission model as shown in equation (1) can be established for each signal in the multiplex signal (hi, hq, vi, vq) shown in FIG. 1.
  • s(t) represents the drive signal of the transmitting end light modulator, carrying the information to be transmitted;
  • ⁇ represents the DC offset of the transmitting end optical modulator, and V ⁇ represents the transmitting end optical modulator Half-wave voltage;
  • n represents the total noise on the channel through which s(t) passes;
  • r(t) represents the electrical signal at the receiving end, ie the receiver estimates the transmitted signal s(t);
  • k represents the power at the receiving end An adjustment factor between the signal and the drive signal of the transmitting end light modulator.
  • noise n has a zero mean and is not correlated with the signal.
  • the offset drift angle can be assumed It is a small amount (e.g., a small amount relative to a predetermined value), and the noise power is much smaller than the signal power.
  • the signal component of the receiving end corresponding to the DC component and the driving of the light modulator of the transmitting end can satisfy the following relationship:
  • E[s 2 (t)] represents the driving signal power
  • E[r(t)] represents the receiving end signal component corresponding to the DC component
  • the received signal power and the driving signal power of the transmitting end optical modulator can satisfy the following relationship:
  • E[s 2 (t)] represents the drive signal power
  • E[r 2 (t)] represents the received signal power
  • E[s 2 (t)] represents the drive signal power of the transmitter optical modulator
  • E[r 2 (t)] represents the received signal power
  • E[r(t)] represents the DC component.
  • the adjustment factor k can be regarded as a positive number, and thus the sign of the offset drift can be determined by the sign of the mean value of the received signal.
  • ADC Analog Digital Converter
  • various impairments in the signal can be compensated to obtain the introduction of bias drift on each path. The DC component and the power after the signal is recovered.
  • the half-wave voltage V ⁇ of the MZM and the drive signal power E[s 2 (t)] in the equation (4) can be provided by the transmitter.
  • the half-wave voltage V ⁇ of the MZM and the drive signal power E[s 2 (t)] in the equation (4) can be provided by the transmitter.
  • the signal is a single polarization state signal, ie the optical communication system is a single polarization state system.
  • FIG. 3 is another schematic diagram of a method for estimating a DC bias of an optical modulator according to an embodiment of the present invention.
  • the steps in FIG. 2 are refined by taking a single polarization modulation as an example.
  • the signal processing of the electrical signal in step 201 to obtain the phase noise compensated signal may specifically include: sampling using an analog-to-digital converter (ADC) to perform in-phase quadrature (IQ) imbalance compensation and Resampling, equalization (first equalization) and phase noise estimation (first phase noise estimation) for phase noise compensation (first phase noise compensation).
  • ADC analog-to-digital converter
  • the IQ imbalance compensation of the receiver can be completed by using the prior art; the phase noise can be extracted from the resampled and equalized signal, and the phase noise includes the phase noise introduced by the laser frequency offset and the line width, which can be related in the prior art. Any method of estimation.
  • the equalization shown in Figure 3 includes compensation for linear or non-linear impairment of fiber optic effects, and can also be accomplished using prior art techniques. The specific content of these implementations will not be described here.
  • the signal component of the receiving end corresponding to the DC component introduced at the transmitting end due to the offset drift of the transmitting end optical modulator can be extracted.
  • the phase noise compensated signal may be averaged in the time domain, and the receiving end signal component corresponding to the DC component may be extracted.
  • the present invention is not limited thereto, and any related method of extracting signal components in the prior art can be used.
  • the receiving end signal components E[r i (t)] and E[r q (t)] corresponding to the direct current component can be obtained.
  • step 204 calculating the received signal power based on the signal of the signal component of the receiving end corresponding to the DC component is removed, which may specifically include: removing the signal of the signal component of the receiving end corresponding to the DC component.
  • Equalization (second equalization) is performed, phase noise estimation (second phase noise estimation) and phase noise compensation (second phase noise compensation) are performed, and the received signal power is calculated based on the phase noise compensated signal.
  • the phase noise compensated signal can be squared in the time domain to calculate the received signal power.
  • the present invention is not limited thereto, and any related method of calculating power in the related art can be used.
  • the received signal powers E[r 2 i (t)] and E[r 2 q (t)] can be obtained.
  • the DC offsets ⁇ i and ⁇ q of the emitter optical modulator can be calculated according to equation (4).
  • the signal is a dual polarization state signal, ie the optical communication system is a dual polarization state system.
  • FIG. 4 is another schematic diagram of a method for estimating a DC bias of an optical modulator according to an embodiment of the present invention.
  • the steps in FIG. 2 are refined by taking dual polarization modulation as an example.
  • the signal processing of the electrical signal in step 201 to obtain the phase noise compensated signal may specifically include: sampling using an analog-to-digital converter (ADC) to perform in-phase quadrature (IQ) imbalance compensation and Resampling, equalization and polarization demultiplexing (first equalization and polarization demultiplexing) and phase noise estimation (first phase noise estimation) for phase noise compensation (first phase noise compensation).
  • ADC analog-to-digital converter
  • the IQ imbalance compensation of the receiver can be completed by using the prior art; the phase noise can be extracted from the resampled and equalized signal, and the phase noise includes the phase noise introduced by the laser frequency offset and the line width, which can be related in the prior art. Any method of estimation.
  • the equalization shown in Figure 4 includes compensation for linear or nonlinear impairment of the fiber effect, which can also be achieved using prior art techniques. The specific content of these implementations will not be described here.
  • the signal component of the receiving end corresponding to the DC component introduced at the transmitting end due to the offset drift of the transmitting end optical modulator can be extracted.
  • the phase noise compensated signal may be averaged in the time domain, and the receiving end signal component corresponding to the DC component may be extracted.
  • the present invention is not limited thereto, and any related method of extracting signal components in the prior art can be used.
  • step 202 offset demultiplexing is needed to obtain the signal components E[r hi (t)], E[r hq (t)], and E of the receiving end corresponding to the DC component. r vi (t)] and E[r vq (t)].
  • the signal component of the receiving end corresponding to the DC component may be multiplied by the response matrix R to perform the polarization demultiplexing; the response matrix R is a signal component of the receiving end corresponding to the DC component removed.
  • a filter for equalization and polarization demultiplexing (second equalization and polarization demultiplexing, as described later)
  • step 204 calculating the received signal power based on the signal of the receiving end signal component corresponding to the DC component is removed, which may specifically include: removing the signal of the receiving end signal component corresponding to the DC component.
  • Perform equalization and polarization demultiplexing (second equalization and polarization demultiplexing), phase noise estimation (second phase noise estimation), phase noise compensation (second phase noise compensation), and phase noise compensation
  • the signal calculates the received signal power.
  • the phase noise compensated signal can be squared in the time domain to calculate the received signal power.
  • the present invention is not limited thereto, and any related method of calculating power in the related art can be used.
  • the received signal powers E[r 2 hi (t)], E[r 2 hq (t)], E[r 2 vi (t)], and E[r 2 can be obtained. Vq (t)].
  • the DC offsets ⁇ hi , ⁇ hq , ⁇ vi , and ⁇ vq of the emitter optical modulator can be calculated according to equation (4).
  • FIGS. 3 and 4 are only illustrative of the embodiments of the present invention, but the invention is not limited thereto.
  • the order of execution between the various steps can be appropriately adjusted, and other steps can be added or some of the steps can be reduced.
  • Those skilled in the art can appropriately modify the above based on the above contents, and are not limited to the description of the above drawings.
  • the signal based on the phase noise compensation extracts the signal component of the receiving end corresponding to the DC component introduced at the transmitting end due to the offset drift of the optical modulator at the transmitting end, based on removing the signal component of the receiving end.
  • the signal calculates the received signal power; and calculates a DC offset of the transmitting end optical modulator based on the receiving end signal component, the received signal power, and the driving signal power of the transmitting end optical modulator.
  • Embodiments of the present invention provide an apparatus for estimating a DC bias of an optical modulator, configured to convert a received optical signal into a receiving end of an electrical signal.
  • an apparatus for estimating a DC bias of an optical modulator configured to convert a received optical signal into a receiving end of an electrical signal.
  • the apparatus 350 for estimating the DC offset of the optical modulator includes:
  • a signal processing unit 501 which performs signal processing on the electrical signal to obtain a phase noise compensated signal
  • a signal extracting unit 502 based on the phase noise compensated signal, extracting a signal component of the receiving end corresponding to the DC component introduced at the transmitting end due to the offset drift of the transmitting end optical modulator;
  • a signal removing unit 503 which removes, from the phase noise compensated signal, a signal component of the receiving end corresponding to the DC component;
  • a power calculation unit 504 which calculates a received signal power based on a signal from which a signal component of the receiving end corresponding to the DC component is removed;
  • the offset calculation unit 505 calculates a DC offset of the transmitter optical modulator based on the receiver signal component corresponding to the DC component, the received signal power, and the driving signal power of the transmitter optical modulator. .
  • the driving signal of the transmitting end optical modulator and the electrical signal of the receiving end satisfy the following relationship:
  • s(t) represents the drive signal of the light modulator at the transmitting end
  • ⁇ represents the DC bias of the light modulator of the transmitting end
  • V ⁇ represents the half-wave voltage of the light modulator of the transmitting end
  • n represents s (t) the total noise on the path passed
  • r(t) represents the electrical signal at the receiving end
  • k represents the adjustment factor between the electrical signal at the receiving end and the driving signal of the transmitting end light modulator.
  • the signal component of the receiving end corresponding to the DC component and the driving signal power of the transmitting end optical modulator satisfy the following relationship:
  • E[s 2 (t)] represents the driving signal power
  • E[r(t)] represents the receiving end signal component corresponding to the DC component
  • the received signal power and the driving signal power of the transmitting end optical modulator satisfy the following relationship:
  • E[s 2 (t)] represents the drive signal power
  • E[r 2 (t)] represents the received signal power
  • the offset calculation unit 505 can calculate the DC offset of the transmitter optical modulator using the following formula:
  • E[s 2 (t)] represents the drive signal power of the transmitter optical modulator
  • E[r 2 (t)] represents the received signal power
  • E[r(t)] represents the DC component.
  • the signal is a single polarization state signal
  • the signal processing unit 501 may be specifically configured to: perform sampling by using an analog-to-digital converter, perform in-phase orthogonal imbalance compensation and re-sampling, perform equalization and phase noise estimation, and perform phase noise compensation;
  • the power calculation unit 504 is specifically configured to: perform equalization on a signal of a signal component at a receiving end corresponding to the DC component, perform phase noise estimation and phase noise compensation, and calculate the received signal based on the phase noise compensated signal. power.
  • the signal is a dual polarization state signal
  • Figure 6 is another schematic diagram of an apparatus for estimating the DC bias of an optical modulator in accordance with an embodiment of the present invention, showing the situation in a dual polarization system.
  • the optical modulator DC bias estimation apparatus 600 includes a signal processing unit 501, a signal extraction unit 502, a signal removal unit 503, a power calculation unit 504, and an offset calculation unit 505, as described above.
  • the signal processing unit 501 may be specifically configured to: perform sampling by using an analog-to-digital converter, perform in-phase orthogonal imbalance compensation and resampling, perform equalization, polarization demultiplexing, and phase noise estimation to perform phase noise compensation. ;
  • the power calculation unit 504 may be specifically configured to: perform equalization and polarization demultiplexing on a signal from which a signal component of the receiving end corresponding to the DC component is removed; perform phase noise estimation and phase noise compensation; and perform signal based on phase noise compensation The received signal power is calculated.
  • the optical modulator DC bias estimation apparatus 600 may further include:
  • a matrix multiplying unit 601 which multiplies a signal component of the receiving end corresponding to the DC component by a response matrix to perform the polarization demultiplexing
  • the response matrix R is a filter for performing equalization and polarization demultiplexing on a signal from which a signal component of a receiving end corresponding to the DC component is removed. Response at zero frequency;
  • FIGS. 5 and 6 are only illustrative of the embodiments of the present invention, but the invention is not limited thereto. For example, some other components may be added as appropriate or some of them may be reduced. Those skilled in the art can appropriately modify the above based on the above contents, and are not limited to the description of the above drawings.
  • the signal based on the phase noise compensation extracts the signal component of the receiving end corresponding to the DC component introduced at the transmitting end due to the offset drift of the optical modulator at the transmitting end, based on removing the signal component of the receiving end.
  • the signal calculates the received signal power; and calculates a DC offset of the transmitting end optical modulator based on the receiving end signal component, the received signal power, and the driving signal power of the transmitting end optical modulator.
  • the embodiment of the present invention provides a receiver that can be configured with the optical modulator DC bias estimation apparatus 500 or 600 as described in Embodiment 2; the same content of the embodiment of the present invention and Embodiments 1 and 2 is no longer Narration.
  • FIG. 7 is a schematic diagram of a receiver according to an embodiment of the present invention. As shown in FIG. 7, the receiver 700 may include:
  • a photoelectric converter 701 that converts the received optical signal into an electrical signal
  • a digital signal processor 702 that performs signal processing on the electrical signal to obtain a phase noise compensated signal; and extracts a signal based on the phase noise compensation to be introduced at the transmitting end due to offset drift of the transmitting end optical modulator a signal component of the receiving end corresponding to the DC component; removing a signal component of the receiving end corresponding to the DC component from the phase noise compensated signal; and removing a signal based on a signal component of the receiving end corresponding to the DC component Calculating the received signal power; calculating a DC offset of the transmitting end optical modulator based on the receiving end signal component corresponding to the DC component, the received signal power, and the driving signal power of the transmitting end optical modulator.
  • the photoelectric converter 701 can be configured with MZM, and the digital signal processor 702 can implement the functions/operations as described above using DSP technology.
  • Embodiments of the present invention also provide an optical communication system.
  • FIG. 8 is a schematic diagram of an optical communication system according to an embodiment of the present invention.
  • a signal transmitted by a transmitter may arrive at a receiver through different devices (eg, optical fibers, optical amplifiers, dispersion compensation fibers, etc.) in a transmission link.
  • devices eg, optical fibers, optical amplifiers, dispersion compensation fibers, etc.
  • an MZM is configured in the transmitter and/or receiver, and the receiver has a digital signal processor 702 as described above.
  • the above apparatus and method of the present invention may be implemented by hardware or by hardware in combination with software.
  • the present invention relates to a computer readable program that, when executed by a logic component, enables the logic component to implement the apparatus or components described above, or to cause the logic component to implement the various methods described above Or steps.
  • the present invention also relates to a storage medium for storing the above program, such as a hard disk, a magnetic disk, an optical disk, a DVD, a flash memory, or the like.
  • the method/apparatus described in connection with the embodiments of the invention may be embodied directly in hardware, a software module executed by a processor, or a combination of both.
  • one or more of the functional block diagrams shown in FIG. 5 and/or one or more combinations of functional block diagrams may correspond to various software of a computer program flow.
  • Modules can also correspond to individual hardware modules.
  • These software modules may correspond to the respective steps shown in FIG. 2, respectively.
  • These hardware modules can be implemented, for example, by curing these software modules using a Field Programmable Gate Array (FPGA).
  • FPGA Field Programmable Gate Array
  • the software module can reside in RAM memory, flash memory, ROM memory, EPROM memory, EEPROM memory, registers, hard disk, removable disk, CD-ROM, or any other form of storage medium known in the art.
  • a storage medium can be coupled to the processor to enable the processor to read information from, and write information to, the storage medium; or the storage medium can be an integral part of the processor.
  • the processor and the storage medium can be located in an ASIC.
  • the software module can be stored in the memory of the mobile terminal or in a memory card that can be inserted into the mobile terminal.
  • the software module can be stored in the MEGA-SIM card or a large-capacity flash memory device.
  • One or more of the functional blocks described in the figures and/or one or more combinations of functional blocks may be implemented as a general purpose processor, digital signal processor (DSP) for performing the functions described herein.
  • DSP digital signal processor
  • ASIC application specific integrated circuit
  • FPGA field programmable gate array
  • One or more of the functional blocks described with respect to the figures and/or one or more combinations of functional blocks may also be implemented as a combination of computing devices, eg, a combination of a DSP and a microprocessor, multiple microprocessors One or more microprocessors in conjunction with DSP communication or any other such configuration.

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  • Physics & Mathematics (AREA)
  • Electromagnetism (AREA)
  • Engineering & Computer Science (AREA)
  • Computer Networks & Wireless Communication (AREA)
  • Signal Processing (AREA)
  • Optics & Photonics (AREA)
  • Optical Communication System (AREA)
  • Optical Modulation, Optical Deflection, Nonlinear Optics, Optical Demodulation, Optical Logic Elements (AREA)

Abstract

一种光调制器直流偏置的估计方法、装置以及接收机。所述估计方法包括：对电信号进行信号处理以获得相位噪声补偿后的信号；基于相位噪声补偿后的信号提取出由于发射端光调制器的偏置漂移而在发射端引入的直流分量所对应的接收端信号成分；从相位噪声补偿后的信号中去除所述接收端信号成分；基于去除了所述接收端信号成分的信号计算接收信号功率；基于所述接收端信号成分、所述接收信号功率以及所述发射端光调制器的驱动信号功率，计算所述发射端光调制器的直流偏置。由此，能够以简单有效的结构和操作对直流偏置进行估计和补偿。

Description

光调制器直流偏置的估计方法、装置以及接收机 技术领域
本发明涉及光通信技术领域，特别涉及一种光调制器直流偏置的估计方法、装置以及接收机。
背景技术
马赫曾德调制器(MZM，Mach-Zehnder Modulator)被广泛应用于光纤通信系统。在相干光纤通信系统中，通常用两个并行的MZM以及一个90度移相器构成一个同相正交(IQ，In-phase Quadrature)马赫曾德调制器(IQ-MZM)，对复数信号的同相和正交分量分别进行调制。
图1是偏振复用系统中光调制器结构的示意图，示出了双偏振态(用h和v表示)调制的情况。其中，IQ-MZM的结构如图1虚线框所示，如图1所示，s(t)为MZM的驱动信号，D为最优直流偏置电压。无论是用于单偏振态调制，还是用于双偏振态调制(例如如图1所示)，IQ-MZM中的每个MZM通常都独立工作在各自的最优直流偏置点上。然而，由于环境变化和器件老化等原因，MZM的直流偏置会发生漂移。如图1所示，ε表示漂移引起的直流偏置。
漂移引起的直流偏置会影响调制性能，从而给系统性能造成损伤。特别是随着调制格式的逐渐升级，例如16正交幅度调制(QAM，Quadrature Amplitude Modulation)、32QAM甚至更高阶格式，系统对MZM直流偏置的漂移越来越敏感。
应该注意，上面对技术背景的介绍只是为了方便对本发明的技术方案进行清楚、完整的说明，并方便本领域技术人员的理解而阐述的。不能仅仅因为这些方案在本发明的背景技术部分进行了阐述而认为上述技术方案为本领域技术人员所公知。
下面列出了对于理解本发明和常规技术有益的文献，通过引用将它们并入本文中，如同在本文中完全阐明了一样。
[1]U.S.patent 6539038,James Allan Wilkerson,etal.“Reference frequency quadrature phase-based control of drive level and DC bias of laser modulator”.
[2]Kenro Sekine,etal.“A Novel Bias Control Technique for MZ Modulator with Monitoring Power of Backward Light for Advanced Modulation Formats”,OSA  1-55752-830-6.
[3]Constantinos S.Petrou,etal.“Quadrature Imbalance Compensation for PDM QPSKCoherent Optical Systems”,IEEE Photon.Technol.Lett.,vol.21,no.24,pp.1876-1878,Dec.,15,2008
[4]L.Li et al.,“Wide-Range,Accurate and Simple Digital Frequency Offset Compensator for Optical Coherent Receivers”,OFC2008,OWT4.
[5]J.Li et al.,”Laser-Linewidth-Tolerant Feed-Forward Carrier Phase Estimator With Reduced Complexity for QAM”,JLT 29,pp.2358,2011.
[6]H.Louchet et al.,“Improved DSP algorithms for coherent 16-QAM transmission”,ECOC2008,Tu.1.E.6
[7]W Yan et al.,“Low Complexity Digital Perturbation Back-propagation”,ECOC2011,Tu.3.A.2.
发明内容
发明人发现：为了保证稳定的偏置，目前通常在发射端对MZM进行自动偏置控制，例如采用基于导频和功率的控制方案，不能以简单有效的结构和操作对直流偏置进行估计和补偿。
本发明实施例提供一种光调制器直流偏置的估计方法、装置以及接收机。在信号的接收端使用数字信号处理(DSP，Digital Signal Processing)对在发射端由于漂移而引起的直流偏置进行估计。
根据本发明实施例的第一个方面，提供一种光调制器直流偏置的估计方法，应用于将接收到的光信号转换成电信号的接收端，所述估计方法包括：
对所述电信号进行信号处理以获得相位噪声补偿后的信号；
基于所述相位噪声补偿后的信号提取出由于发射端光调制器的偏置漂移而在发射端引入的直流分量所对应的接收端信号成分；
从所述相位噪声补偿后的信号中去除所述直流分量所对应的接收端信号成分；
基于去除了所述直流分量所对应的接收端信号成分的信号计算接收信号功率；
基于所述直流分量所对应的接收端信号成分、所述接收信号功率以及所述发射端光调制器的驱动信号功率，计算所述发射端光调制器的直流偏置。
根据本发明实施例的第二个方面，提供一种光调制器直流偏置的估计装置，配置于将接收到的光信号转换成电信号的接收端，所述估计装置包括：
信号处理单元，其对所述电信号进行信号处理以获得相位噪声补偿后的信号；
信号提取单元，其基于所述相位噪声补偿后的信号提取出由于发射端光调制器的偏置漂移而在发射端引入的直流分量所对应的接收端信号成分；
信号去除单元，其从所述相位噪声补偿后的信号中去除所述直流分量所对应的接收端信号成分；
功率计算单元，其基于去除了所述直流分量所对应的接收端信号成分的信号计算接收信号功率；
偏置计算单元，其基于所述直流分量所对应的接收端信号成分、所述接收信号功率以及所述发射端光调制器的驱动信号功率，计算所述发射端光调制器的直流偏置。
根据本发明实施例的第三个方面，提供一种接收机，所述接收机包括：
光电转换器，其将接收到的光信号转换成电信号；
数字信号处理器，其对所述电信号进行信号处理以获得相位噪声补偿后的信号；基于所述相位噪声补偿后的信号提取出由于发射端光调制器的偏置漂移而在发射端引入的直流分量所对应的接收端信号成分；从所述相位噪声补偿后的信号中去除所述直流分量所对应的接收端信号成分；基于去除了所述直流分量所对应的接收端信号成分的信号计算接收信号功率；基于所述直流分量所对应的接收端信号成分、所述接收信号功率以及所述发射端光调制器的驱动信号功率，计算所述发射端光调制器的直流偏置。
本发明实施例的有益效果在于：基于相位噪声补偿后的信号提取出由于发射端光调制器的偏置漂移而在发射端引入的直流分量所对应的接收端信号成分，基于去除了所述接收端信号成分的信号计算接收信号功率；基于所述接收端信号成分、所述接收信号功率以及所述发射端光调制器的驱动信号功率，计算所述发射端光调制器的直流偏置。由此，可以在信号的接收端使用数字信号处理对在发射端由于漂移而引起的直流偏置进行估计，能够以简单有效的结构和操作对直流偏置进行估计和补偿。
参照后文的说明和附图，详细公开了本发明的特定实施方式，指明了本发明的原理可以被采用的方式。应该理解，本发明的实施方式在范围上并不因而受到限制。在所附权利要求的精神和条款的范围内，本发明的实施方式包括许多改变、修改和等同。
针对一种实施方式描述和/或示出的特征可以以相同或类似的方式在一个或更多个其它实施方式中使用，与其它实施方式中的特征相组合，或替代其它实施方式中的特征。
应该强调，术语“包括/包含”在本文使用时指特征、整件、步骤或组件的存在，但并不排除一个或更多个其它特征、整件、步骤或组件的存在或附加。
附图说明
在本发明实施例的一个附图或一种实施方式中描述的元素和特征可以与一个或更多个其它附图或实施方式中示出的元素和特征相结合。此外，在附图中，类似的标号表示几个附图中对应的部件，并可用于指示多于一种实施方式中使用的对应部件。
图1是偏振复用系统中光调制器结构的示意图；
图2是本发明实施例1的光调制器直流偏置的估计方法的示意图；
图3是本发明实施例1的光调制器直流偏置的估计方法的另一示意图；
图4是本发明实施例1的光调制器直流偏置的估计方法的另一示意图；
图5是本发明实施例2的光调制器直流偏置的估计装置的示意图；
图6是本发明实施例2的光调制器直流偏置的估计装置的另一示意图；
图7是本发明实施例3的接收机的示意图；
图8为本发明实施例3的光通信系统的示意图。
具体实施方式
参照附图，通过下面的说明书，本发明的前述以及其它特征将变得明显。在说明书和附图中，具体公开了本发明的特定实施方式，其表明了其中可以采用本发明的原则的部分实施方式，应了解的是，本发明不限于所描述的实施方式，相反，本发明包括落入所附权利要求的范围内的全部修改、变型以及等同物。
实施例1
本发明实施例提供一种光调制器直流偏置的估计方法，应用于将接收到的光信号转换成电信号的接收端。图2是本发明实施例的光调制器直流偏置的估计方法的示意图，如图2所示，所述估计方法包括：
步骤201，对电信号进行信号处理以获得相位噪声补偿后的信号；
步骤202，基于相位噪声补偿后的信号提取出由于发射端光调制器的偏置漂移而在发射端引入的直流分量所对应的接收端信号成分；
步骤203，从相位噪声补偿后的信号中去除该直流分量所对应的接收端信号成分；
步骤204，基于去除了所述直流分量所对应的接收端信号成分的信号计算接收信号功率；
步骤205，基于所述直流分量所对应的接收端信号成分、所述接收信号功率以及所述发射端光调制器的驱动信号功率，计算所述发射端光调制器的直流偏置。
在本实施例中，首先以图1所示的双偏振态MZM结构为例，对本发明中各个信号之间的关系进行说明。
根据MZM的sin(·)的调制特性，可以对图1所示多路信号(hi,hq,vi,vq)中的每路信号，建立如式(1)所示的传输模型。
Figure PCTCN2016102684-appb-000001
其中，s(t)表示所述发射端光调制器的驱动信号，携带着要传送的信息；ε表示所述发射端光调制器的直流偏置，Vπ表示所述发射端光调制器的半波电压；n表示s(t)所经过通道上的总噪声；r(t)表示所述接收端的电信号，即接收机对发射信号s(t)的估计；k表示所述接收端的电信号和所述发射端光调制器的驱动信号之间的调整因子。
在实际系统中，驱动信号具有零均值E[s(t)]＝0(算子E[·]表示期望或均值)，并且电压范围在半波电压以内，即|s(t)|≤Vπ，因此
Figure PCTCN2016102684-appb-000002
成立。
通常噪声n具有零均值，且与信号不相关。在这些信号和噪声特性下，可以假定偏置漂移角度
Figure PCTCN2016102684-appb-000003
是一小量(例如相对于某一预定值较小的量)，并且噪声功率远小于信号功率。
根据上述模型，直流分量所对应的接收端信号成分和所述发射端光调制器的驱动 信号功率可以满足如下关系：
Figure PCTCN2016102684-appb-000004
其中，E[s2(t)]表示所述驱动信号功率，E[r(t)]表示所述直流分量所对应的接收端信号成分。
而接收信号功率和所述发射端光调制器的驱动信号功率可以满足如下关系：
Figure PCTCN2016102684-appb-000005
其中，E[s2(t)]表示所述驱动信号功率，E[r2(t)]表示所述接收信号功率。
联立式(2)和(3)，得到偏置漂移角度或者电压大小，如式(4)所示。即，可以使用如下公式(4)计算所述发射端光调制器的直流偏置：
Figure PCTCN2016102684-appb-000006
其中，E[s2(t)]表示所述发射端光调制器的驱动信号功率，E[r2(t)]表示所述接收信号功率，E[r(t)]表示所述直流分量所对应的接收端信号成分。
值得注意的是，上述公式(1)至(4)仅是本发明实施例的实际例子，但本发明不限于此，例如可以根据实际情况对上述公式进行适当地调整或者变型，例如可以改变其中的一个或多个参数等等。
在相干光纤通信系统中，当接收端的本振激光器和发射端激光器之间的频率偏差较小时，调整因子k可认为是正数，因此偏置漂移的符号可以由接收信号均值的符号决定。根据式(4)，在获取接收机的模数转换器(ADC，Analog Digital Converter)的采样输出后，可以对信号中的各种损伤进行补偿，以获取每个通路上的由于偏置漂移引入的直流分量以及信号恢复后的功率。
在本实施例中，式(4)中MZM的半波电压Vπ以及驱动信号功率E[s2(t)]可以由发射机提供。具体如何获取可以参考相关技术，本实施例对此不再重复说明。
以下对于如何在接收端估计偏置漂移进行进一步说明。
在一个实施方式中，所述信号为单偏振态信号，即光通信系统为单偏振态系统。
图3是本发明实施例的光调制器直流偏置的估计方法的另一示意图，以单偏振态调制为例对图2中的各个步骤进行了细化。
如图3所示，步骤201中对电信号进行信号处理以获得相位噪声补偿后的信号，具体可以包括：使用模数转换器(ADC)进行采样，进行同相正交(IQ)不平衡补偿以及重采样，进行均衡(第一次均衡)以及相位噪声估计(第一次相位噪声估计)，进行相位噪声补偿(第一次相位噪声补偿)。
在本实施方式中，为了在接收端恢复出发射端每个通路上的偏置漂移引入的直流分量，需要对信号在传输过程中受到的光纤效应引起的损伤、激光器频偏和线宽、接收机IQ不平衡等引起的系统损伤进行补偿，如图3所示。
其中，可采用现有技术完成接收机的IQ不平衡补偿；可以从经过重采样和均衡的信号中提取相位噪声，相位噪声包括激光器频偏和线宽引入的相位噪声，可用现有技术中相关的任意方法进行估计。此外，图3所示的均衡包含对光纤效应的线性或者非线性损伤的补偿，也可采用现有技术实现。关于这些实现的具体内容，此处不再赘述。
如图3所示，步骤202中基于相位噪声补偿后的信号，可以提取出由于发射端光调制器的偏置漂移而在发射端引入的直流分量所对应的接收端信号成分。
例如，可以在时域上对相位噪声补偿后的信号进行平均，提取出该直流分量所对应的接收端信号成分。但本发明不限于此，可以使用现有技术中的提取信号成分的任意相关方法。
如图3所示，经过步骤202后，可以获得直流分量所对应的接收端信号成分E[ri(t)]和E[rq(t)]。
如图3所示，步骤204中基于去除了所述直流分量所对应的接收端信号成分的信号计算接收信号功率，具体可以包括：对去除了所述直流分量所对应的接收端信号成分的信号进行均衡(第二次均衡)，进行相位噪声估计(第二次相位噪声估计)以及相位噪声补偿(第二次相位噪声补偿)，以及基于相位噪声补偿后的信号计算所述接收信号功率。
例如，可以在时域上对再次相位噪声补偿后的信号进行平方，计算出该接收信号功率。但本发明不限于此，可以使用现有技术中的计算功率的任意相关方法。
如图3所示，经过步骤204后，可以获得接收信号功率E[r2 i(t)]和E[r2 q(t)]。
由此，可以根据公式(4)计算出发射端光调制器的直流偏置εi和εq。
在另一个实施方式中，所述信号为双偏振态信号，即光通信系统为双偏振态系统。
图4是本发明实施例的光调制器直流偏置的估计方法的另一示意图，以双偏振态调制为例对图2中的各个步骤进行了细化。
如图4所示，步骤201中对电信号进行信号处理以获得相位噪声补偿后的信号，具体可以包括：使用模数转换器(ADC)进行采样，进行同相正交(IQ)不平衡补偿以及重采样，进行均衡和偏振解复用(第一次均衡和偏振解复用)以及相位噪声估计(第一次相位噪声估计)，进行相位噪声补偿(第一次相位噪声补偿)。
在本实施方式中，为了在接收端恢复出发射端每个通路上的偏置漂移引入的直流分量，需要对信号在传输过程中受到的光纤效应引起的损伤、激光器频偏和线宽、接收机IQ不平衡等引起的系统损伤进行补偿，如图4所示。
其中，可采用现有技术完成接收机的IQ不平衡补偿；可以从经过重采样和均衡的信号中提取相位噪声，相位噪声包括激光器频偏和线宽引入的相位噪声，可用现有技术中相关的任意方法进行估计。此外，图4所示的均衡包含对光纤效应的线性或者非线性损伤的补偿，也可采用现有技术实现。关于这些实现的具体内容，此处不再赘述。
如图4所示，步骤202中基于相位噪声补偿后的信号，可以提取出由于发射端光调制器的偏置漂移而在发射端引入的直流分量所对应的接收端信号成分。
例如，可以在时域上对相位噪声补偿后的信号进行平均，提取出该直流分量所对应的接收端信号成分。但本发明不限于此，可以使用现有技术中的提取信号成分的任意相关方法。
如图4所示，经过步骤202后还需要进行偏置解复用，来获得直流分量所对应的接收端信号成分E[rhi(t)]、E[rhq(t)]、E[rvi(t)]和E[rvq(t)]。
其中，可以将所述直流分量所对应的接收端信号成分与响应矩阵R相乘以进行所述偏振解复用；该响应矩阵R为对去除了所述直流分量所对应的接收端信号成分的信号进行均衡与偏振解复用(第二次均衡和偏振解复用，如后所述)的滤波器
Figure PCTCN2016102684-appb-000007
在零频处的响应；即
Figure PCTCN2016102684-appb-000008
如图4所示，步骤204中基于去除了所述直流分量所对应的接收端信号成分的信号计算接收信号功率，具体可以包括：对去除了所述直流分量所对应的接收端信号成分的信号进行均衡以及偏振解复用(第二次均衡和偏振解复用)，进行相位噪声估计(第二次相位噪声估计)以及相位噪声补偿(第二次相位噪声补偿)，以及基于相位噪声补偿后的信号计算所述接收信号功率。
例如，可以在时域上对再次相位噪声补偿后的信号进行平方，计算出该接收信号功率。但本发明不限于此，可以使用现有技术中的计算功率的任意相关方法。
如图4所示，经过步骤204后，可以获得接收信号功率E[r2 hi(t)]、E[r2 hq(t)]、E[r2 vi(t)]和E[r2 vq(t)]。
由此，可以根据公式(4)计算出发射端光调制器的直流偏置εhi、εhq、εvi和εvq。
值得注意的是，附图3和4仅示意性地对本发明实施例进行了说明，但本发明不限于此。例如可以适当地调整各个步骤之间的执行顺序，此外还可以增加其他的一些步骤或者减少其中的某些步骤。本领域的技术人员可以根据上述内容进行适当地变型，而不仅限于上述附图的记载。
由上述实施例可知，基于相位噪声补偿后的信号提取出由于发射端光调制器的偏置漂移而在发射端引入的直流分量所对应的接收端信号成分，基于去除了所述接收端信号成分的信号计算接收信号功率；基于所述接收端信号成分、所述接收信号功率以及所述发射端光调制器的驱动信号功率，计算所述发射端光调制器的直流偏置。由此，可以在信号的接收端使用数字信号处理对在发射端由于漂移而引起的直流偏置进行估计，能够以简单有效的结构和操作对直流偏置进行估计和补偿。
实施例2
本发明实施例提供一种光调制器直流偏置的估计装置，配置于将接收到的光信号转换成电信号的接收端。本发明实施例与实施例1相同的内容不再赘述。
图5是本发明实施例的光调制器直流偏置的估计装置的示意图，如图5所示，光调制器直流偏置的估计装置500包括：
信号处理单元501，其对所述电信号进行信号处理以获得相位噪声补偿后的信号；
信号提取单元502，其基于所述相位噪声补偿后的信号提取出由于发射端光调制器的偏置漂移而在发射端引入的直流分量所对应的接收端信号成分；
信号去除单元503，其从所述相位噪声补偿后的信号中去除所述直流分量所对应的接收端信号成分；
功率计算单元504，其基于去除了所述直流分量所对应的接收端信号成分的信号计算接收信号功率；
偏置计算单元505，其基于所述直流分量所对应的接收端信号成分、所述接收信号功率以及所述发射端光调制器的驱动信号功率，计算所述发射端光调制器的直流偏置。
在本实施例中，发射端光调制器的驱动信号和接收端的电信号满足如下关系：
Figure PCTCN2016102684-appb-000009
其中，s(t)表示所述发射端光调制器的驱动信号；ε表示所述发射端光调制器的直流偏置，Vπ表示所述发射端光调制器的半波电压；n表示s(t)所经过通道上的总噪声；r(t)表示所述接收端的电信号；k表示所述接收端的电信号和所述发射端光调制器的驱动信号之间的调整因子。
在本实施例中，直流分量所对应的接收端信号成分和发射端光调制器的驱动信号功率满足如下关系：
Figure PCTCN2016102684-appb-000010
其中，E[s2(t)]表示所述驱动信号功率，E[r(t)]表示所述直流分量所对应的接收端信号成分。
在本实施例中，接收信号功率和发射端光调制器的驱动信号功率满足如下关系：
Figure PCTCN2016102684-appb-000011
其中，E[s2(t)]表示所述驱动信号功率，E[r2(t)]表示所述接收信号功率。
在本实施例中，偏置计算单元505可以使用如下公式计算所述发射端光调制器的直流偏置：
Figure PCTCN2016102684-appb-000012
其中，E[s2(t)]表示所述发射端光调制器的驱动信号功率，E[r2(t)]表示所述接收信号功率，E[r(t)]表示所述直流分量所对应的接收端信号成分。
在一个实施方式中，信号为单偏振态信号；
信号处理单元501具体可以用于：使用模数转换器进行采样，进行同相正交不平衡补偿以及重采样，进行均衡以及相位噪声估计，进行相位噪声补偿；
功率计算单元504具体可以用于：对去除了所述直流分量所对应的接收端信号成分的信号进行均衡，进行相位噪声估计以及相位噪声补偿，以及基于相位噪声补偿后的信号计算所述接收信号功率。
在另一个实施方式中，信号为双偏振态信号；
图6是本发明实施例的光调制器直流偏置的估计装置的另一示意图，示出了双偏振态系统中的情况。如图6所示，光调制器直流偏置的估计装置600包括：信号处理单元501，信号提取单元502，信号去除单元503，功率计算单元504和偏置计算单元505，如上所述。
在本实施方式中，信号处理单元501具体可以用于：使用模数转换器进行采样，进行同相正交不平衡补偿以及重采样，进行均衡、偏振解复用以及相位噪声估计，进行相位噪声补偿；
功率计算单元504具体可以用于：对去除了所述直流分量所对应的接收端信号成分的信号进行均衡与偏振解复用；进行相位噪声估计以及相位噪声补偿；以及基于相位噪声补偿后的信号计算所述接收信号功率。
如图6所示，光调制器直流偏置的估计装置600还可以包括：
矩阵相乘单元601，其将所述直流分量所对应的接收端信号成分与响应矩阵相乘以进行所述偏振解复用；
其中，所述响应矩阵R为对去除了所述直流分量所对应的接收端信号成分的信号进行均衡与偏振解复用的滤波器
Figure PCTCN2016102684-appb-000013
在零频处的响应；即
Figure PCTCN2016102684-appb-000014
值得注意的是，附图5和6仅示意性地对本发明实施例进行了说明，但本发明不限于此。例如可以适当地增加其他的一些部件或者减少其中的某些部件。本领域的技术人员可以根据上述内容进行适当地变型，而不仅限于上述附图的记载。
由上述实施例可知，基于相位噪声补偿后的信号提取出由于发射端光调制器的偏置漂移而在发射端引入的直流分量所对应的接收端信号成分，基于去除了所述接收端信号成分的信号计算接收信号功率；基于所述接收端信号成分、所述接收信号功率以及所述发射端光调制器的驱动信号功率，计算所述发射端光调制器的直流偏置。由此，可以在信号的接收端使用数字信号处理对在发射端由于漂移而引起的直流偏置进行估计，能够以简单有效的结构和操作对直流偏置进行估计和补偿。
实施例3
本发明实施例提供一种接收机，该接收机可以配置有如实施例2所述的光调制器直流偏置的估计装置500或600；本发明实施例与实施例1和2相同的内容不再赘述。
图7是本发明实施例的接收机的示意图，如图7所示，接收机700可以包括：
光电转换器701，其将接收到的光信号转换成电信号；
数字信号处理器702，其对所述电信号进行信号处理以获得相位噪声补偿后的信号；基于所述相位噪声补偿后的信号提取出由于发射端光调制器的偏置漂移而在发射端引入的直流分量所对应的接收端信号成分；从所述相位噪声补偿后的信号中去除所述直流分量所对应的接收端信号成分；基于去除了所述直流分量所对应的接收端信号成分的信号计算接收信号功率；基于所述直流分量所对应的接收端信号成分、所述接收信号功率以及所述发射端光调制器的驱动信号功率，计算所述发射端光调制器的直流偏置。
在本实施例中，光电转换器701可以配置有MZM，数字信号处理器702可以使用DSP技术实现如上所述的功能/操作。
本发明实施例还提供一种光通信系统。
图8为本发明实施例的光通信系统的示意图，如图8所示，发射机发射的信号可以经过传输链路中不同的器件(例如光纤、光放大器、色散补偿光纤等)到达接收机。 其中，发射机和/或接收机中配置有MZM，并且接收机具有如上所述的数字信号处理器702。
本发明以上的装置和方法可以由硬件实现，也可以由硬件结合软件实现。本发明涉及这样的计算机可读程序，当该程序被逻辑部件所执行时，能够使该逻辑部件实现上文所述的装置或构成部件，或使该逻辑部件实现上文所述的各种方法或步骤。本发明还涉及用于存储以上程序的存储介质，如硬盘、磁盘、光盘、DVD、flash存储器等。
结合本发明实施例描述的方法/装置可直接体现为硬件、由处理器执行的软件模块或二者组合。例如，图5中所示的功能框图中的一个或多个和/或功能框图的一个或多个组合(例如，信号处理单元、信号提取单元等)，既可以对应于计算机程序流程的各个软件模块，亦可以对应于各个硬件模块。这些软件模块，可以分别对应于图2所示的各个步骤。这些硬件模块例如可利用现场可编程门阵列(FPGA)将这些软件模块固化而实现。
软件模块可以位于RAM存储器、闪存、ROM存储器、EPROM存储器、EEPROM存储器、寄存器、硬盘、移动磁盘、CD-ROM或者本领域已知的任何其它形式的存储介质。可以将一种存储介质耦接至处理器，从而使处理器能够从该存储介质读取信息，且可向该存储介质写入信息；或者该存储介质可以是处理器的组成部分。处理器和存储介质可以位于ASIC中。该软件模块可以存储在移动终端的存储器中，也可以存储在可插入移动终端的存储卡中。例如，若设备(如移动终端)采用的是较大容量的MEGA-SIM卡或者大容量的闪存装置，则该软件模块可存储在该MEGA-SIM卡或者大容量的闪存装置中。
针对附图中描述的功能方框中的一个或多个和/或功能方框的一个或多个组合，可以实现为用于执行本申请所描述功能的通用处理器、数字信号处理器(DSP)、专用集成电路(ASIC)、现场可编程门阵列(FPGA)或者其它可编程逻辑器件、分立门或者晶体管逻辑器件、分立硬件组件或者其任意适当组合。针对附图描述的功能方框中的一个或多个和/或功能方框的一个或多个组合，还可以实现为计算设备的组合，例如，DSP和微处理器的组合、多个微处理器、与DSP通信结合的一个或多个微处理器或者任何其它这种配置。
以上结合具体的实施方式对本发明进行了描述，但本领域技术人员应该清楚，这些描述都是示例性的，并不是对本发明保护范围的限制。本领域技术人员可以根据本发明的精神和原理对本发明做出各种变型和修改，这些变型和修改也在本发明的范围内。

Claims (17)

  1. 一种光调制器直流偏置的估计方法，应用于将接收到的光信号转换成电信号的接收端，所述估计方法包括：
    对所述电信号进行信号处理以获得相位噪声补偿后的信号；
    基于所述相位噪声补偿后的信号提取出由于发射端光调制器的偏置漂移而在发射端引入的直流分量所对应的接收端信号成分；
    从所述相位噪声补偿后的信号中去除所述直流分量所对应的接收端信号成分；
    基于去除了所述直流分量所对应的接收端信号成分的信号计算接收信号功率；
    基于所述直流分量所对应的接收端信号成分、所述接收信号功率以及所述发射端光调制器的驱动信号功率，计算所述发射端光调制器的直流偏置。
  2. 根据权利要求1所述的估计方法，其中，所述发射端光调制器的驱动信号和所述接收端的电信号满足如下关系：
    Figure PCTCN2016102684-appb-100001
    其中，s(t)表示所述发射端光调制器的驱动信号；ε表示所述发射端光调制器的直流偏置，Vπ表示所述发射端光调制器的半波电压；n表示s(t)所经过通道上的总噪声；r(t)表示所述接收端的电信号；k表示所述接收端的电信号和所述发射端光调制器的驱动信号之间的调整因子。
  3. 根据权利要求2所述的估计方法，其中，所述直流分量所对应的接收端信号成分和所述发射端光调制器的驱动信号功率满足如下关系：
    Figure PCTCN2016102684-appb-100002
    其中，E[s2(t)]表示所述驱动信号功率，E[r(t)]表示所述直流分量所对应的接收端信号成分。
  4. 根据权利要求2所述的估计方法，其中，所述接收信号功率和所述发射端光调制器的驱动信号功率满足如下关系：
    Figure PCTCN2016102684-appb-100003
    其中，E[s2(t)]表示所述驱动信号功率，E[r2(t)]表示所述接收信号功率。
  5. 根据权利要求2所述的估计方法，其中，使用如下公式计算所述发射端光调制器的直流偏置：
    Figure PCTCN2016102684-appb-100004
    其中，E[s2(t)]表示所述发射端光调制器的驱动信号功率，E[r2(t)]表示所述接收信号功率，E[r(t)]表示所述直流分量所对应的接收端信号成分。
  6. 根据权利要求1所述的估计方法，其中，所述电信号为单偏振态信号；
    对所述电信号进行信号处理以获得相位噪声补偿后的信号包括：使用模数转换器进行采样，进行同相正交不平衡补偿以及重采样，进行均衡以及相位噪声估计，进行相位噪声补偿；
    基于去除了所述直流分量所对应的接收端信号成分的信号计算接收信号功率包括：对去除了所述直流分量所对应的接收端信号成分的信号进行均衡，进行相位噪声估计以及相位噪声补偿，以及基于相位噪声补偿后的信号计算所述接收信号功率。
  7. 根据权利要求1所述的估计方法，其中，所述电信号为双偏振态信号；
    对所述电信号进行信号处理以获得相位噪声补偿后的信号包括：使用模数转换器进行采样，进行同相正交不平衡补偿以及重采样，进行均衡、偏振解复用以及相位噪声估计，进行相位噪声补偿；
    基于去除了所述直流分量所对应的接收端信号成分的信号计算接收信号功率包括：对去除了所述直流分量所对应的接收端信号成分的信号进行均衡与偏振解复用；进行相位噪声估计以及相位噪声补偿；以及基于相位噪声补偿后的信号计算所述接收信号功率。
  8. 根据权利要求7所述的估计方法，其中，所述估计方法还包括：
    将所述直流分量所对应的接收端信号成分与响应矩阵相乘以进行所述偏振解复用；
    其中，所述响应矩阵R为对去除了所述直流分量所对应的接收端信号成分的信号 进行均衡与偏振解复用的滤波器
    Figure PCTCN2016102684-appb-100005
    在零频处的响应；即
    Figure PCTCN2016102684-appb-100006
  9. 一种光调制器直流偏置的估计装置，配置于将接收到的光信号转换成电信号的接收端，所述估计装置包括：
    信号处理单元，其对所述电信号进行信号处理以获得相位噪声补偿后的信号；
    信号提取单元，其基于所述相位噪声补偿后的信号提取出由于发射端光调制器的偏置漂移而在发射端引入的直流分量所对应的接收端信号成分；
    信号去除单元，其从所述相位噪声补偿后的信号中去除所述直流分量所对应的接收端信号成分；
    功率计算单元，其基于去除了所述直流分量所对应的接收端信号成分的信号计算接收信号功率；
    偏置计算单元，其基于所述直流分量所对应的接收端信号成分、所述接收信号功率以及所述发射端光调制器的驱动信号功率，计算所述发射端光调制器的直流偏置。
  10. 根据权利要求9所述的估计装置，其中，所述发射端光调制器的驱动信号和所述接收端的电信号满足如下关系：
    Figure PCTCN2016102684-appb-100007
    其中，s(t)表示所述发射端光调制器的驱动信号；ε表示所述发射端光调制器的直流偏置，Vπ表示所述发射端光调制器的半波电压；n表示s(t)所经过通道上的总噪声；r(t)表示所述接收端的电信号；k表示所述接收端的电信号和所述发射端光调制器的驱动信号之间的调整因子。
  11. 根据权利要求10所述的估计装置，其中，所述直流分量所对应的接收端信号成分和所述发射端光调制器的驱动信号功率满足如下关系：
    Figure PCTCN2016102684-appb-100008
    其中，E[s2(t)]表示所述驱动信号功率，E[r(t)]表示所述直流分量所对应的接收端 信号成分。
  12. 根据权利要求10所述的估计装置，其中，所述接收信号功率和所述发射端光调制器的驱动信号功率满足如下关系：
    Figure PCTCN2016102684-appb-100009
    其中，E[s2(t)]表示所述驱动信号功率，E[r2(t)]表示所述接收信号功率。
  13. 根据权利要求10所述的估计装置，其中，所述偏置计算单元使用如下公式计算所述发射端光调制器的直流偏置：
    Figure PCTCN2016102684-appb-100010
    其中，E[s2(t)]表示所述发射端光调制器的驱动信号功率，E[r2(t)]表示所述接收信号功率，E[r(t)]表示所述直流分量所对应的接收端信号成分。
  14. 根据权利要求9所述的估计装置，其中，所述电信号为单偏振态信号；
    所述信号处理单元用于：使用模数转换器进行采样，进行同相正交不平衡补偿以及重采样，进行均衡以及相位噪声估计，进行相位噪声补偿；
    所述功率计算单元用于：对去除了所述直流分量所对应的接收端信号成分的信号进行均衡，进行相位噪声估计以及相位噪声补偿，以及基于相位噪声补偿后的信号计算所述接收信号功率。
  15. 根据权利要求9所述的估计装置，其中，所述电信号为双偏振态信号；
    所述信号处理单元用于：使用模数转换器进行采样，进行同相正交不平衡补偿以及重采样，进行均衡、偏振解复用以及相位噪声估计，进行相位噪声补偿；
    所述功率计算单元用于：对去除了所述直流分量所对应的接收端信号成分的信号进行均衡与偏振解复用；进行相位噪声估计以及相位噪声补偿；以及基于相位噪声补偿后的信号计算所述接收信号功率。
  16. 根据权利要求15所述的估计装置，其中，所述估计装置还包括：
    矩阵相乘单元，其将所述直流分量所对应的接收端信号成分与响应矩阵相乘以进行所述偏振解复用；
    其中，所述响应矩阵R为对去除了所述直流分量所对应的接收端信号成分的信号 进行均衡与偏振解复用的滤波器
    Figure PCTCN2016102684-appb-100011
    在零频处的响应；即
    Figure PCTCN2016102684-appb-100012
  17. 一种接收机，所述接收机包括：
    光电转换器，其将接收到的光信号转换成电信号；
    数字信号处理器，其对所述电信号进行信号处理以获得相位噪声补偿后的信号；基于所述相位噪声补偿后的信号提取出由于发射端光调制器的偏置漂移而在发射端引入的直流分量所对应的接收端信号成分；从所述相位噪声补偿后的信号中去除所述直流分量所对应的接收端信号成分；基于去除了所述直流分量所对应的接收端信号成分的信号计算接收信号功率；基于所述直流分量所对应的接收端信号成分、所述接收信号功率以及所述发射端光调制器的驱动信号功率，计算所述发射端光调制器的直流偏置。
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