WO2018064791A1

WO2018064791A1 - Reactance shift detection and over dissipation protection in a wireless power transmitter

Info

Publication number: WO2018064791A1
Application number: PCT/CN2016/101423
Authority: WO
Inventors: Songnan Yang; Tong Zhang; Bin Xiao; Philippe R. AUZAS; Mark A. HANSEN
Original assignee: Intel Corporation
Priority date: 2016-10-03
Filing date: 2016-10-03
Publication date: 2018-04-12

Abstract

Techniques for operating a wireless power transmitter are described herein. An example of a wireless power transmitter includes a transmitter coil configured to generate a magnetic field for wirelessly charging a battery. The wireless power transmitter also includes a power amplifier to generate alternating current (AC) power to be delivered to the transmitter coil. The power amplifier includes a transistor that is pulsed to generate the AC power. The wireless power transmitter also includes an inductive load detector coupled to the power amplifier, wherein the inductive load detector is to integrate a drain voltage of the transistor over a specified integration interval to determine a reactance of the transmitter coil when an impedance of the transmitter coil is inductive.

Description

REACTANCE SHIFT DETECTION AND OVER DISSIPATION PROTECTION IN A WIRELESS POWER TRANSMITTER

Technical Field

This disclosure relates generally to techniques for wireless charging. Specifically, this disclosure relates to tuning a wireless power transmitter and protecting the wireless power transmitter from over dissipation.

Background Art

A basic wireless charging system may include a wireless power transmitter unit (PTU) and a wireless power receiving unit (PRU) . For example, a PTU may include a transmit (Tx) coil, and a PRU may include receive (Rx) coil. Magnetic resonance wireless charging may employ a magnetic coupling between the Tx coil and the Rx coil. In some cases, a PRU is implemented in a device having various size chassis. In some cases, constant current supply design may include providing a constant current source even when various PRU device chassis size changes the resonant frequency of the PTU coil.

Brief Description of the Drawings

Fig. 1 is block diagram of a power transmitting unit to provide power to a power receiving unit.

Fig. 2 is a diagram illustrating an example of an inductive load detector.

Fig. 3 is a plot of the electrical characteristics of the power amplifier plotted on a smith chart.

Fig. 4 is a plot of the drain voltage wave forms plotted over variable load reactance shift conditions.

Fig. 5 is a plot of the switching signals implemented in the power amplifier.

Fig. 6 is a plot of the rectified integral generated by the integrator of Fig. 2.

Fig. 7 is a diagram illustrating another example of an inductive load detector.

Fig. 8 is a diagram illustrating an example inductive load detector 140 coupled to differential amplifier.

The same numbers are used throughout the disclosure and the figures to reference like components and features. Numbers in the 100 series refer to features originally found in Fig. 1； numbers in the 200 series refer to features originally found in Fig. 2； and so on.

Description of the Aspects

The present disclosure relates generally to techniques for wireless charging. Specifically, the techniques described herein provide a wireless power transmitting unit (PTU) having a transmit (Tx) coil configured to generate a magnetic field. The techniques discussed herein may be implemented in a PTU that adheres to a wireless charging standard protocol, such as the specification provided by Alliance For Wireless Power (A4WP) . However, the techniques described herein may be implemented using any other wireless charging standard protocol where applicable.

When PRU devices with various chassis sizes are presented to the PTU coil, the resonant frequency of the PTU coil changes, which in turn degrades the coupling condition between PTU and PRU and can also cause over-dissipation and over-voltage conditions on the PTU’s power amplifier. For example, a smart phone may have a relatively smaller metal chassis when compared to a laptop computer. When a computing device having a relatively larger metal chassis is placed on a wireless charging device having a PTU, a reduction of PTU coil inductance may occur as an eddy current induced on the metal chassis cancels part of the magnetic field generated by PTU, causing a detuning, or reactance shift from the resonant frequency. This detuning effect significantly reduces the efficiency of the power amplifier (PA) and its capability to provide a constant current to the resonator coil. In extreme cases of high inductive load, the level of detuning may cause an over dissipation condition that can damage the power amplifier. Over dissipation occurs when the power amplifier’s main transistor experiences high levels of current, which result in high levels of heat dissipation.

To counteract these effects, the PTU includes a tuning circuit that impedance matches the output of the power source to the input of the transmit coil. The tuning circuit is responsive to a change in the impedance of the transmit coil, for example, a change in the reactance of the transmit coil. The tuning circuit can include circuitry for retuning the transmit coil and the magnetic coupling between the PTU and PRU to the same resonant frequency. For example, the tuning circuit can include a set of capacitors configured to generate a reactance shift to counter the detected reactance shift caused by the presence of PRU devices. As the reactance shifts, additional capacitors can be engaged to counter the shift. In some situations, reactance shifts may occur rapidly. Therefore, the circuitry used to detect the reactance shift and engage the tuning should be able to accurately and reliably detect reactance shifts and respond to quickly changing conditions.

According to one technique for detecting the reactance shift, a voltage sensor, a current sensor, and a phase detector are used to measure the electrical characteristics of load. These measurements are input to a microcontroller or other processor, which digitizes the input and computes the load impedance. The micro controller determines whether to adjust the tuning circuit based on the reactance component of the calculated impedance value. Since this technique relies on a processor to calculate the reactance shift, it may not be fast enough to handle fast varying reactance variation on the PTU coil when a large device such as a notebook computer or tablet PC is placed on or removed from the PTU coil.

Another technique for quickly detecting reactance shifts involves detecting changes in the peak drain voltage of the power amplifier’s main transistor. The peak drain voltage serves as a reliable indicator of reactance levels over a wide range of operating conditions. In this way, reactance shifts may be detected directly without the use of digitization and calculation by a microcontroller, and is therefore faster and simpler to implement.

However, in beaconing modes and during mode transitions when the auto-tuning function may not be operational, the transmitter coil may still experience large reactance shifts, especially when user is placing or removing devices from PTU in a fast motion. Additionally, in a highly inductive region outside of normal operation range (e.g., reactance greater than 40Ohms for a design with nominal real impedance of 30Ohms) , the peak drain voltage of the power amplifier’s main transistor may not be a reliable indicator for determining the load condition of the transmitter coil. Therefore, the auto-tuning technique, which relies on changes in the peak drain voltage of the power amplifier’s main transistor cannot be used as a protection mechanism in all instances.

The present disclosure provides an inductive load detector that is able to determine the inductive load and thereby prevent over-dissipation under high inductive loading conditions. The inductive load detector can directly measure a physical quantity that is proportional to thermal dissipation on the power amplifier’s main transistor to allow fast detection and protection against non-ideal inductive loading conditions. The inductive load detector may supplement the reactance shift detection provided by the auto tuning device and protect the PTU from thermal runaway caused by extreme inductive load conditions.

Fig. 1 is block diagram of a PTU to provide power to a PRU. A PTU 102 may be coupled to a PRU 104 via magnetic inductive coupling between

resonators

106 and 108, as indicated by the arrow 110. The resonator 106 of the PTU 102 may be referred to herein as a Tx coil 106. The resonator 108 of the PRU 104 may be referred to herein as an Rx coil 108.

The PTU 102 may include an oscillator, 112, a power amplifier 114, a Direct Current to Direct Current (DC2DC) converter 116, and a tuning circuit 118. The oscillator 112 is configured to generate a periodic oscillating electronic signal at a specified frequency. The power amplifier 114 receives direct current power from the DC2DC converter 116, and amplifies the signal received from the oscillator 112. The power amplifier 114 may be a switching power amplifier such as a Class E amplifier.

The tuning circuit 118 matches the impedance of the power amplifier 114 to the impedance of the resonator 106 to ensure efficient power transmission under normal operating conditions. The tuning circuit 118 may include any suitable arrangement of electrical components such as capacitors, inductors, and other circuit elements that can be adjusted to impedance match the resonator 106 to the power amplifier 114. The tuning circuit 118 is controlled by the tuning control circuit 120. The load, as that term is used herein, refers to the input impedance at the output of the power amplifier 114 due to the combined effects of the resonator 106 and tuning circuit 118.

Other components of the PTU may include a Bluetooth Low Energy (BLE) module 122, a controller 124, and others. The controller 124 can be configured to control various aspects of the operation of the PTU 102. For example, the controller 124 can set a frequency, and/or power level of the power radiated by the resonator 106. The controller 124 can also control communications between the PTU 102 and the PRU 104 through the BLE module 122.

The PRU 104 may be a component of a computing device 126 configured to receive power from the PTU 102 wirelessly by the inductive coupling 110. The computing device 126 may be any suitable type of computing device, including a laptop computer, an Ultrabook, a tablet computer, a phablet, a mobile phone, smart phone, smart watch, and other types of mobile battery-powered devices.

The PRU 104 can include a rectifier 128, a DC2DC converter 130, a battery charger 132, and a battery 134. The computing device 126 receives electrical power as a magnetic flux associated with the inductive coupling that passes through the resonator 108. The rectifier 128 receives an alternating current (AC) voltage from the resonator 108 and generates a rectified DC voltage (Vrect) . The DC2DC converter 130 receives the rectified voltage from the rectifier 128, converts the voltage to a suitable voltage level, and provides the output to the battery charger 132, which charges the battery 134. The battery 134 powers the various platform hardware of the computing device 126. The platform hardware includes all of the processors, working memory, data storage devices, communication buses, I/O interfaces, communication devices, display devices, and other components that make up the computing device 126.

The PRU 104 may also include a Bluetooth Low Energy (BLE) module 136 and a controller 138. The controller 138 is configured to perform a wireless handshake with the PTU 102. As discussed above, the wireless handshake broadcast may be performed through the

BLE modules

122 and 136 or other wireless data transmission component. Various types of information may be transmitted during the wireless handshake, including power budget, wireless charging capabilities, size of the computing device 126, and other information.

The tuning control circuit 120 is configured to sense a reactance shift of the resonator 106 occurring due to inductive coupling between the PTU 102 and other objects, such as the PRU 104. Upon detection of a reactance shift above a predefined threshold, the tuning control circuit 120 can adjust the tuning circuit 118 to retune the magnetic inductive coupling 110 to a resonant frequency. As described further below in relation to Fig. 3, the tuning control circuit 120 detects reactance shifts by detecting the peak voltage at the drain terminal of the power amplifier’s main transistor. The tuning control circuit 120 can include one or more voltage sensors, current sensors, comparators, and other simply circuitry for generating a tuner control signal and sending the tuner control signal to the tuning circuit. The tuning control circuit 120 may not include an Analog-to-Digital Converter (ADC) , a microcontroller, or other type of processors for performing complex mathematical computations.

The PTU 102 also includes an inductive load detector 140. The inductive load detector 140 detects high reactance conditions that are outside of the normal operating range and cannot be reliably handled by the tuning control circuit 120. To detect high reactance conditions, the inductive load detector 104 integrates the drain voltage of the power amplifier’s main transistor over a suitable interval. The integration value provides a reliable indicator of the reactance level of the load when the load is highly inductive.

The inductive load detector 140 may implement one or more inductive load thresholds used to triggering events based on the integration value. For example, if the integration value exceeds a pre-specified threshold, the inductive load detector 140 may trigger an event configured to prevent thermal overload on the transistor under high inductive loading conditions

For example, if the integration value exceeds a predetermined threshold, the inductive load detector 140 may trigger an over-dissipation event. If an over dissipation event is triggered, the inductive load detector 140 may send a signal to the power amplifier 114 to deactivate the power amplifier and prevent thermal overload. In some examples, if the integration value exceeds a predetermined threshold, the inductive load detector 140 may trigger a retuning event. If a retuning event is triggered, the inductive load detector 140 may send a signal to the tuning control circuit 120 to activate a predetermined retuning action. The predetermined retuning action may involve switching on one or more predetermined capacitors, inductors, or other circuit elements that are configured to impedance match the resonator 106 to the power amplifier 114 during high inductive loading conditions. The inductive load detector 140 is described further below in relation to Figs. 2-8.

The block diagram of Fig. 1 is not intended to indicate that the PTU 102 and the PRU 104 are to include all of the components shown in Fig. 1. Further, the PTU 102 and/or the PRU 104 may include any number of additional components not shown in Fig. 1, depending on the details of the specific implementation.

Fig. 2 is a diagram illustrating an example of an inductive load detector. As shown in Fig. 2, the inductive load detector 140 is coupled to the power amplifier 114. The power amplifier 114 includes a gate driver 202 coupled to the power amplifier’s main transistor, which is referred to herein as transistor 204. Transistor 204 may be a bipolar transistor, Field Effect Transistor (FET) , or other type of transistor. The gate driver 202 pulses the transistor 204 to create a voltage at the output 206 of the power amplifier 114. The signal generated by the gate driver 202 is shown in Fig. 2 as V2. The output 206 of the power amplifier 114 is an alternating current (AC) that oscillates at the desired frequency of the power amplifier. The power supply for the transistor 204 is provided by the voltage source 208, VDD. The duty cycle of the pulse pattern generated by the gate driver 202 determines the magnitude of the output voltage. In some examples, the power amplifier 114 is a class E power amplifier, which is configured to exhibit the electrical characteristics described in Figs. 3 and 4 below.

The inductive load detector 140 can include an integrator 210 and a comparator 212. The integrator 210 integrates the drain voltage at the drain terminal of the transistor 204 over a specified time window referred to as the integration interval. In some examples, the integrator 210 includes a transistor 214, a capacitor 216, and a rectifier 218. However, other implementations are also possible. In this example, the transistor 214 is turned off during the integration interval, which causes the capacitor 216 to be charged. When the transistor 214 is turned on, the capacitor 216 is discharged. The output of the capacitor 216 is coupled to the rectifier 218, which rectifies the output of the capacitor 216 to a DC voltage signal, Vout, that represents the inductive loading conditions presented to the power amplifier 114. The output voltage, Vout, is coupled to the comparator 212. The integration interval is controlled by a gate drive signal shown in Fig. 2 as V3. The gate drive signal V3 may be controlled to activate the transistor 214 in synchronization with the gate driver 202 with a suitable phase delay. The relationship between the gate drive signal V2 and the gate drive signal V3 is described further in relation to Fig. 5.

The inductive load detector 140 also includes circuitry for generating a specified voltage reference, Vref. In the example shown in Fig. 2, the voltage reference is generated by a voltage divider circuit 220 coupled to the voltage source 208. The comparator 212 compares the output voltage of the integrator 210 (Vout) , to the voltage reference (Vref) to generate an output signal, referred to herein as the inductive load indicator, Vind. The inductive load indicator, Vind, indicates that the load condition has exceeded the inductive load threshold set by the power divider 220. In the example shown in Fig. 2, the inductive load indicator, Vind, is routed to the tuning circuit 118 and can trigger a re-tuning event if the integration result indicates a highly inductive load, i.e., Vout greater than Vref.

As explained above, inductive load detector 140 compares two analog voltages, and does not require complex sensing circuit or a micro controller. Accordingly, the inductive load detector 140 can be implemented completely in simple logic hardware, which is inherently faster and more robust compared to conventional impedance measurement methods that require a microcontroller.

The block diagram of Fig. 2 is not intended to indicate that the PTU 102 is to include all of the components shown in Fig. 2. Further, the PTU 102 may include any number of additional components not shown in Fig. 2, depending on the details of the specific implementation.

Fig. 3 is a plot of the electrical characteristics of the power amplifier plotted on a smith chart. Each curve represents a constant peak drain voltage contour line for the power amplifier under different loading conditions of the transmit coil 106 shown in Fig 2. Each contour line represents the complex load impedances that result in the same peak drain voltage for the main transistor of the power amplifier, i.e., transistor 204. Within the region of the smith chart identified as region 302, the constant peak drain voltage contour lines are very closely aligned with the reactance axis, also known as constant jX lines, of the smith chart. Accordingly, within region 302, the reactance shift (jX) can be directly detected by measuring the drain voltage. Thus, within region 302, the peak drain voltage relative to the supply voltage, VDD, can be used as a direct indication of the reactance level exhibited by the load. The tuning control circuit 120 can detect the peak drain voltage and drive the tuning circuit 118 to switch out capacitors to compensate for the reactance shift.

Within the region identified as region 304, the constant peak drain voltage contour lines are no longer aligned with the reactance axis. Thus, within region 304, the peak drain voltage does not provide a reliable indicator of the reactance level of the load and cannot be used for auto tuning. However, as explained further below in relation to Fig. 4, the integration process implemented by the inductive load detector 140 is able to detect high reactance conditions when the load presented to the output of the power amplifier is in region 304.

Fig. 4 is a plot of the drain voltage wave forms plotted over variable load reactance shift conditions. The plot of Fig. 4 shows simulated voltage waveforms for the power amplifier 114 shown in Fig. 2. If the power amplifier 114 is operated with zero voltage switching (ZVS) and zero voltage differential switching with 50 percent duty cycle, the ideal switch mode voltage waveform (ideal Vdrain) can be expressed as:

The dashed line 402 represents the gate drive signal, V2, driving the main transistor 204. The main transistor 204 is switched on at the rising edge of the gate drive signal and switched off at the falling edge. Curves 404 to 416 represent the drain voltage on the main transistor 204 plotted over time. Curve 404 represents the drain voltage for an ideal switch mode voltage waveform for a load condition with zero reactance.

Curves

406 and 408 represent the drain voltage resulting from a capacitive load, i.e., negative reactance. As the reactance becomes more negative, the peak of the drain voltage waveform increases monotonically. Curve 410 represents the drain voltage resulting from a load with a low level of inductance, i.e., small positive reactance around j10 . Within this range of reactance levels, from curves 408 to 410, the peak drain voltage behaves monotonically and is proportional to the reactance level. Thus, within these ranges, the peak drain voltage provides a direct indication of the reactance shift (jX) , such that the ratio between peak V_drain and V_DD can be used to detect reactance shift (jX) , and can be used by the tuning control circuit 120 as measure of the reactance level to be compensated for by the tuning circuit 118.

Curves

412, 414, and 416 represent the drain voltage for a load with increasing levels of reactance. For example, curve 412 represents a reactance level of around j20, curve 414 represents a reactance level of around j30, curve 416 represents a reactance level of around j40. As these reactance levels, the peak drain voltage increases with increasing reactance. Therefore, as the load becomes highly inductive, the peak drain voltage no longer provides an unambiguous indication of the reactance. The higher the drain voltage is when the transistor switches on, the more current will be driven in the transistor 204. If the drain voltage is too high when the transistor 204 is switched on, as seen with curve 416 for example, the transistor 204 will receive an inrush of current, which could cause high levels of heat dissipation and possible damage to the transistor 204.

It is difficult to directly detect the current surge as it is happening internal to the transistor. However, as seen in Fig. 4, the drain voltage waveform exhibits a dependency to load reactance. This dependency enables the inductive load detector 140 to detect inductive loading and non-ZVS conditions. To detect inductive loading and non-ZVS conditions, the inductive load detector 140 integrates the voltage waveform over an interval starting shortly before the transistor 204 turns on. In Fig. 4, the start of the integration is represented by the line labeled t0.

As shown in Fig. 4, the integral increases with inductive loading, while the integral will be zero for capacitive loading and pure resistive loading. The resulting integral can be used as a direct indicator of the severity of power dissipation across the transistor due to a non-ZVS condition. Therefore, by controlling the interval of the integration, the drain voltage integral will be proportional to the power dissipation due to non-ZVS, and the drain voltage integral normalized by the supply voltage (VDD) will be proportional to the inductive loading. Corresponding switching signals V2 and V3 are shown in Fig. 5.

Fig. 5 is a plot of the switching signals implemented in the power amplifier. V2 represents the gate drive signal that drives the main transistor 204, and V3 represents the gate drive signal that drives the transistor 214 of the integrator 210. The main transistor 204 is switched on at the rising edge of the gate drive signal, V2, and switched off at the falling edge. The integration transistor 214 is switched on at the rising edge of the gate drive signal, V3, and switched off at the falling edge. The falling edge of the gate drive signal V3 is labeled T0 and marks the beginning of the integration interval. Signal V3 has the same frequency as signal V2 and leads the signal V2 by a predetermined phase, D. The phase D is controlled so that the integration interval starts a fraction of a period ahead of when the main transistor 204 is turned on. In some examples, the phase D may be implemented by deriving the signals V2 and V3 from a common source and conducting both signals through different delay lines. The length of the delay lines will depend on the operating frequency and the desired phase difference, D.

Fig. 6 is a plot of the rectified integral generated by the integrator 210 of Fig. 2. With reference to Fig. 2, the rectified integral, Vout, is the output voltage of the rectifier 218. As shown in Fig. 6, the integral is equal to zero for a purely resistive load (X ＝ 0) . As the reactance increases, the rectified integral, Vout, increase in a nearly linear fashion. The rectified integral, Vout, will also vary depending on the integration interval implemented. In this example, the start of the integration interval, which is dependent on the phase delay D between the gate drive signals V2 and V3, is controlled so that Vout equals zero for a purely resistive load (X ＝ 0) .

Fig. 7 is a diagram illustrating another example of an inductive load detector. The power amplifier 114 is the same as the power amplifier shown in Fig. 2. The inductive load detector 140 shown Fig. 7 is similar to the inductive load detector 140 shown in Fig. 2, and includes the integrator 210 and a comparator 700. As described in Fig. 2, the integrator 210 integrates the drain voltage at the drain terminal of the transistor 204 over a specified integration interval, and the resulting integral, Vout, is sent to the comparator 700.

The comparator 700 compares the output voltage of the integrator 210 (Vout) , to the voltage reference (Vref) to generate the output signal, Vind. However, in the example shown in Fig. 7, the voltage reference, Vref, is a fixed voltage level, which is not dependent on the power amplifier’s supply voltage, VDD. Using a fixed reference voltage enables the inductive load detector 140 to detect the power dissipation on the transistor, where the turn-on voltage of the transistor is integrated and scales with supply voltage. The inductive load detector 140 can also include an R-C circuit 704 at the output of the rectifier 218. The R-C circuit introduces a time constant on Vout to capture the commutation of the integral in representation of the thermal dissipation on the transistor.

The inductive load indicator, Vind, indicates whether the power dissipation on the main transistor 204 has exceeded the power dissipation threshold set by Vref. If Vout reaches the threshold Vref, the inductive load indicator, Vind, can be triggered to protect the transistor 204 from thermal run-away. In the example shown in Fig. 2, the inductive load indicator, Vind, is routed to the gate driver 202 and can disable the gate driver 202 if the integration result indicates a high level of power dissipation on the transistor 204.

For the sake of clarity, different aspects of an example inductive load detector 140 are described in Figs. 2 and 7. However, it will be appreciated that the inductive load detector 140 shown in Fig. 2 can include the combined features described in Figs. 2 and 7. For example, the inductive load detector 140 may include the comparator 212 of Fig. 2, which is coupled to a reference voltage that is proportional to the source voltage, Vdd, and the comparator 700 which is coupled to a fixed reference voltage.

Fig. 8 is a diagram illustrating an example inductive load detector 140 coupled to differential amplifier. The differential power amplifier 114 can be included in the PTU 102 and may be a class E differential power amplifier. The power amplifier 114 is similar to the power amplifier shown in Figs. 2 and 7, but includes two legs, a first leg with a first power transistor 802 and a second leg with a second power transistor 804. Each

power transistor

802 and 804 is configured to deliver AC power to a different side of the transmit coil 106 in a push-pull configuration.

The first leg is driven by a first gate driver 806 and the second leg is driven by a second gate driver 808. The gate driver signals generated by the

gate drivers

806 and 808 are approximately 180 degrees out of phase from one another, so that the output of the

power transistors

802 and 804 will combine at the transmit coil 106. The

gate driver

806 and 808 may be driven by a flip-flop 810, which is itself driven by a clock 812 that generates a clock signal at the desired operating frequency of the PTU 102. The non-inverted output of the flip flop 810 is coupled to the first gate driver 806, and the inverted output of the flip flip 810 is coupled to the second gate driver 808. In some examples, the clock frequency may be approximately 13.56 Megahertz.

The inductive load detector140 is the same as the inductive load detector shown in Fig. 7 and includes the comparator 700 with the fixed reference voltage 702. In this example, the integration transistor 214 is controlled by the inverted output of the flip flop and integrates the drain voltage of the first transistor 802. Thus, the inductive load detector 140 applied to first leg of the differential power amplifier uses the clock signal of the other leg to control the integration switch 214 for drain voltage waveform integration. The propagation delay of the gate driver signal can be controlled to position the integration window at a suitable time ahead of that the first transistor turns on, which provides proper timing to realize the over dissipation protection function. The output of the comparator 700 is coupled to the enable input of the first gate driver 802. If the output of the comparator 700 indicates an over dissipation condition, the output of the comparator 700 disables the gate driver 806.

For the sake of clarity, a single inductive load detector 140 is shown in Fig. 8. However, it will be appreciated that the PTU 102 can include an additional inductive load detector 140 to integrate the drain voltage waveform of the second transistor 804. The additional inductive load detector 140 would be coupled to the power amplifier 114 in a complimentary fashion, with the drain of the integration resistor 214 coupled to the drain of the second transistor 804, the gate of the integration resistor 214 coupled to the non-inverted output of the flip-flip 810, and the output of the comparator 700 coupled to the enable input of the second gate driver 808.

The block diagram of Fig. 8 is not intended to indicate that the PTU 102 is to include all of the components shown in Fig. 8. Further, the PTU 102 may include any number of additional components not shown in Fig. 8, depending on the details of the specific implementation.

Fig. 9 is a process flow diagram summarizing an example method to operate a wireless power transmitting unit. The method 900 may be performed by the PTU 102 shown in Fig. 1. The method 900 may be implemented by logic embodied in hardware, including capacitors, diodes, logic gates, flip-flops, latches, and others. Examples of hardware that may be used to implement the method 900 is shown in Figs. 2 and 7. The method may start at block 902.

At block 902, a transistor is pulsed to generate alternating current (AC) power to be delivered to a transmitter coil.

At block 904, the drain voltage of the transistor is integrated over a specified integration interval.

At block 906, the result of the integration is compared to a threshold. The threshold may be a fixed voltage level or the threshold may be proportional to the supply voltage coupled to the transistor. In the case of a fixed threshold, as shown in Fig. 7, the result of the integration provides an indication of the level of power dissipation on the transistor. In the case of a threshold proportional to the supply voltage, as shown in Fig. 2, the result of the integration provides an indication of the load reactance experienced by the power amplifier.

At block 908, a determination is made about whether the result of the integration is equal to or greater than the threshold. If the result of the integration is equal to or greater than the threshold, the process flow advances to block 910. Otherwise, the process flow returns to block 902.

At block 910, an event is trigger based on the result of the integration to prevent over-dissipation on the transistor under high inductive loading conditions. The event may be a retuning event as described in relation to Fig. 2 and/or an over-dissipation event as described in relation to Fig. 7. The method 900 may be repeated for each period of the AC output power waveform generated by the transistor.

The method 900 should not be interpreted as meaning that the blocks are necessarily performed in the order shown. Furthermore, fewer or greater actions can be included in the method 900 depending on the design considerations of a particular implementation.

Examples

Example 1 is a wireless power transmitter. The wireless power transmitter includes a transmitter coil configured to generate a magnetic field for wirelessly charging a battery, and a power amplifier to generate alternating current (AC) power to be delivered to the transmitter coil. The power amplifier includes a transistor that is pulsed to generate the AC power. The wireless power transmitter also includes an inductive load detector coupled to the power amplifier and configured to: perform an integration of a drain voltage of the transistor over a specified integration interval； compare a result of the integration to a threshold； and if the result of the integration reaches the threshold, trigger an event that prevents over-dissipation on the transistor under high inductive loading conditions.

Example 2 includes the wireless power transmitter of example 1, including or excluding optional features. In this example, the wireless power transmitter includes a tuning circuit coupled to the transmitter coil and configured to retune the transmitter coil in response to a reactance shift of the transmitter coil, wherein the event is a retuning event in which an output of the inductive load detector causes the tuning circuit to retune the transmitter coil to cancel the reactance.

Example 3 includes the wireless power transmitter of any one of examples 1 to 2, including or excluding optional features. In this example, the threshold is proportional to a source voltage coupled to the transistor, and a result of the integration is proportional to a reactance of the transmitter coil when an impedance of the transmitter coil is inductive.

Example 4 includes the wireless power transmitter of any one of examples 1 to 3, including or excluding optional features. In this example, the event is an over-dissipation protection event in which an output of the inductive load detector causes a driving signal of the transistor to be disabled.

Example 5 includes the wireless power transmitter of any one of examples 1 to 4, including or excluding optional features. In this example, the threshold is a fixed voltage value, and a result of the integration is proportional to an amount of thermal energy dissipated on the transistor.

Example 6 includes the wireless power transmitter of any one of examples 1 to 5, including or excluding optional features. In this example, the specified integration interval starts a fraction of a period before the transistor is turned on.

Example 7 includes the wireless power transmitter of any one of examples 1 to 6, including or excluding optional features. In this example, the power amplifier is a class E switch mode power amplifier.

Example 8 includes the wireless power transmitter of any one of examples 1 to 7, including or excluding optional features. In this example, the power amplifier is a differential power amplifier comprising: additional transistor to generate AC power to be delivered to the transmitter coil； and an additional inductive load detector configured to perform integration of a drain voltage of the additional transistor.

Example 9 includes the wireless power transmitter of any one of examples 1 to 8, including or excluding optional features. In this example, the inductive load detector is independent of a processor.

Example 10 includes the wireless power transmitter of any one of examples 1 to 9, including or excluding optional features. In this example, the inductive load detector is independent of computing a digital representation of an impedance of the transmitter coil.

Example 11 is a method of operating a power amplifier of a wireless power transmitter. The method includes: pulsing a transistor to generate alternating current (AC) power to be delivered to a transmitter coil； performing an integration of a drain voltage of the transistor over a specified integration interval； comparing a result of the integration to a threshold； and if the result of the integration reaches the threshold, triggering an event that prevents over-dissipation on the transistor under high inductive loading conditions.

Example 12 includes the method of example 11, including or excluding optional features. In this example, triggering the event comprises sending a signal to a tuning control circuit to retune the transmitter coil.

Example 13 includes the method of any one of examples 11 to 12, including or excluding optional features. In this example, the threshold is proportional to a source voltage coupled to the transistor, and a result of the integration is proportional to a reactance of the transmitter coil when an impedance of the transmitter coil is inductive.

Example 14 includes the method of any one of examples 11 to 13, including or excluding optional features. In this example, triggering the event comprises disabling a driving signal of the transistor.

Example 15 includes the method of any one of examples 11 to 14, including or excluding optional features. In this example, the threshold is a fixed voltage value, and a result of the integration is proportional to an amount of thermal energy dissipated on the transistor.

Example 16 includes the method of any one of examples 11 to 15, including or excluding optional features. In this example, the specified integration interval starts a fraction of a period before the transistor is turned on.

Example 17 includes the method of any one of examples 11 to 16, including or excluding optional features. In this example, the power amplifier is a class E switch mode power amplifier.

Example 18 includes the method of any one of examples 11 to 17, including or excluding optional features. In this example, the power amplifier is a differential power amplifier.

Example 19 includes the method of any one of examples 11 to 18, including or excluding optional features. In this example, performing the integration comprises performing the integration independent of a processor.

Example 20 includes the method of any one of examples 11 to 19, including or excluding optional features. In this example, performing the integration comprises performing the integration independent of computing a digital representation of an impedance of the transmitter coil.

Example 21 is a wireless power transmitter. The wireless power transmitter includes a transmitter coil configured to generate a magnetic field for wirelessly charging a battery, and a power amplifier to generate alternating current (AC) power to be delivered to the transmitter coil. The power amplifier includes a transistor and a gate driver that pulses the transistor to generate the AC power. A drain terminal of the transistor is coupled to a voltage source. The wireless power transmitter also includes an inductive load detector. The inductive load detector includes an integrator to perform an integration of a voltage waveform at the drain terminal of the transistor over a specified integration interval, and a comparator to compare a result of the integration to a threshold and generate an indicator based on the comparison. The indicator is to trigger an event that prevents over-dissipation on the transistor under high inductive loading conditions.

Example 22 includes the wireless power transmitter of example 21, including or excluding optional features. In this example, the wireless power transmitter includes a tuning circuit coupled to the transmitter coil and configured to retune the transmitter coil in response to a reactance shift of the transmitter coil, wherein the event is a retuning event in which an output of the inductive load detector causes the tuning circuit to retune the transmitter coil to cancel the reactance.

Example 23 includes the wireless power transmitter of any one of examples 21 to 22, including or excluding optional features. In this example, the wireless power transmitter includes a voltage divider coupled to the source voltage to generate the threshold, wherein the result of the integration is proportional to a reactance of the transmitter coil when an impedance of the transmitter coil is inductive.

Example 24 includes the wireless power transmitter of any one of examples 21 to 23, including or excluding optional features. In this example, the event is an over-dissipation protection event in which the indicator is configured to disable the gate driver.

Example 25 includes the wireless power transmitter of any one of examples 21 to 24, including or excluding optional features. In this example, the threshold is a fixed voltage value, and the result of the integration is proportional to an amount of thermal energy dissipated on the transistor.

Example 26 includes the wireless power transmitter of any one of examples 21 to 25, including or excluding optional features. In this example, the specified integration interval starts a fraction of a period before the transistor is turned on.

Example 27 includes the wireless power transmitter of any one of examples 21 to 26, including or excluding optional features. In this example, the power amplifier is a class E switch mode power amplifier.

Example 28 includes the wireless power transmitter of any one of examples 21 to 27, including or excluding optional features. In this example, the power amplifier is a differential power amplifier comprising: an additional transistor to generate AC power to be delivered to the transmitter coil； an additional gate driver that pulses the additional transistor to generate the AC power； and an additional inductive load detector configured to perform integration of a drain voltage of the additional transistor.

Example 29 includes the wireless power transmitter of any one of examples 21 to 28, including or excluding optional features. In this example, the inductive load detector is independent of a processor.

Example 30 includes the wireless power transmitter of any one of examples 21 to 29, including or excluding optional features. In this example, the inductive load detector is independent of computing a digital representation of an impedance of the transmitter coil.

Example 31 is an apparatus. The apparatus includes a transmitter coil configured to generate a magnetic field for wirelessly charging a battery, and a power amplifier to generate alternating current (AC) power to be delivered to the transmitter coil. The power amplifier includes a transistor that is pulsed to generate the AC power. The apparatus also includes means for preventing over-dissipation on the transistor under high inductive loading conditions. The means for preventing over-dissipation includes means for performing an integration of a drain voltage of the transistor over a specified integration interval, and means for comparing a result of the integration to a threshold and triggering a protection event if the result of the integration reaches the threshold.

Example 32 includes the apparatus of example 31, including or excluding optional features. In this example, the apparatus includes means for retuning the transmitter coil in response to a reactance shift of the transmitter coil, wherein the protection event is a retuning event in which the means for preventing causes the means for retuning to retune the transmitter coil to cancel the reactance.

Example 33 includes the apparatus of any one of examples 31 to 32, including or excluding optional features. In this example, the threshold is proportional to a source voltage coupled to the transistor, and a result of the integration is proportional to a reactance of the transmitter coil when an impedance of the transmitter coil is inductive.

Example 34 includes the apparatus of any one of examples 31 to 33, including or excluding optional features. In this example, the protection event is an over-dissipation protection event in which an output of the means for preventing causes a driving signal of the transistor to be disabled.

Example 35 includes the apparatus of any one of examples 31 to 34, including or excluding optional features. In this example, the threshold is a fixed voltage value, and a result of the integration is proportional to an amount of thermal energy dissipated on the transistor.

Example 36 includes the apparatus of any one of examples 31 to 35, including or excluding optional features. In this example, the specified integration interval starts a fraction of a period before the transistor is turned on.

Example 37 includes the apparatus of any one of examples 31 to 36, including or excluding optional features. In this example, the power amplifier is a class E switch mode power amplifier.

Example 38 includes the apparatus of any one of examples 31 to 37, including or excluding optional features. In this example, the power amplifier is a differential power amplifier comprising: an additional transistor to generate AC power to be delivered to the transmitter coil； and means for preventing over-dissipation on the additional transistor under high inductive loading conditions.

Example 39 includes the apparatus of any one of examples 31 to 38, including or excluding optional features. In this example, the means for preventing is independent of a processor.

Example 40 includes the apparatus of any one of examples 31 to 39, including or excluding optional features. In this example, the means for preventing is independent of computing a digital representation of an impedance of the transmitter coil.

Not all components, features, structures, characteristics, etc. described and illustrated herein need be included in a particular aspect or aspects. If the specification states a component, feature, structure, or characteristic “may” , “might” , “can” or “could” be included, for example, that particular component, feature, structure, or characteristic is not required to be included. If the specification or claim refers to “a” or “an” element, that does not mean there is only one of the element. If the specification or claims refer to “an additional” element, that does not preclude there being more than one of the additional element.

It is to be noted that, although some aspects have been described in reference to particular implementations, other implementations are possible according to some aspects. Additionally, the arrangement and/or order of circuit elements or other features illustrated in the drawings and/or described herein need not be arranged in the particular way illustrated and described. Many other arrangements are possible according to some aspects.

In each system shown in a figure, the elements in some cases may each have a same reference number or a different reference number to suggest that the elements represented could be different and/or similar. However, an element may be flexible enough to have different implementations and work with some or all of the systems shown or described herein. The various elements shown in the figures may be the same or different. Which one is referred to as a first element and which is called a second element is arbitrary.

It is to be understood that specifics in the aforementioned examples may be used anywhere in one or more aspects. For instance, all optional features of the computing device described above may also be implemented with respect to either of the methods or the computer-readable medium described herein. Furthermore, although flow diagrams and/or state diagrams may have been used herein to describe aspects, the techniques are not limited to those diagrams or to corresponding descriptions herein. For example, flow need not move through each illustrated box or state or in exactly the same order as illustrated and described herein.

The present techniques are not restricted to the particular details listed herein. Indeed, those skilled in the art having the benefit of this disclosure will appreciate that many other variations from the foregoing description and drawings may be made within the scope of the present techniques. Accordingly, it is the following claims including any amendments thereto that define the scope of the present techniques.

Claims

A wireless power transmitter, comprising:

a transmitter coil configured to generate a magnetic field for wirelessly charging a battery；

a power amplifier to generate alternating current (AC) power to be delivered to the transmitter coil, the power amplifier comprising a transistor that is pulsed to generate the AC power；

an inductive load detector coupled to the power amplifier and configured to:

perform an integration of a drain voltage of the transistor over a specified integration interval；

compare a result of the integration to a threshold； and

if the result of the integration reaches the threshold, trigger an event that prevents over-dissipation on the transistor under high inductive loading conditions.
The wireless power transmitter of claim 1, comprising a tuning circuit coupled to the transmitter coil and configured to retune the transmitter coil in response to a reactance shift of the transmitter coil, wherein the event is a retuning event in which an output of the inductive load detector causes the tuning circuit to retune the transmitter coil to cancel the reactance.
The wireless power transmitter of claim 1, wherein the threshold is proportional to a source voltage coupled to the transistor, and a result of the integration is proportional to a reactance of the transmitter coil when an impedance of the transmitter coil is inductive.
The wireless power transmitter of claim 1, wherein the event is an over-dissipation protection event in which an output of the inductive load detector causes a driving signal of the transistor to be disabled.
The wireless power transmitter of claim 1, wherein the threshold is a fixed voltage value, and a result of the integration is proportional to an amount of thermal energy dissipated on the transistor.
The wireless power transmitter of any one of claims 1 to 5, wherein the specified integration interval starts a fraction of a period before the transistor is turned on.
The wireless power transmitter of any one of claims 1 to 5, wherein the power amplifier is a class E switch mode power amplifier.
The wireless power transmitter of any one of claims 1 to 5, wherein the power amplifier is a differential power amplifier comprising:

additional transistor to generate AC power to be delivered to the transmitter coil； and

an additional inductive load detector configured to perform integration of a drain voltage of the additional transistor.
The wireless power transmitter of any one of claims 1 to 5, wherein the inductive load detector is independent of a processor.
The wireless power transmitter of any one of claims 1 to 5, wherein the inductive load detector is independent of computing a digital representation of an impedance of the transmitter coil.
A method of operating a power amplifier of a wireless power transmitter, comprising:

pulsing a transistor to generate alternating current (AC) power to be delivered to a transmitter coil；

performing an integration of a drain voltage of the transistor over a specified integration interval；

comparing a result of the integration to a threshold； and

if the result of the integration reaches the threshold, triggering an event that prevents over-dissipation on the transistor under high inductive loading conditions.
The method of claim 11, wherein triggering the event comprises sending a signal to a tuning control circuit to retune the transmitter coil.
The method of claim 11, wherein the threshold is proportional to a source voltage coupled to the transistor, and a result of the integration is proportional to a reactance of the transmitter coil when an impedance of the transmitter coil is inductive.
The method of claim 11, wherein triggering the event comprises disabling a driving signal of the transistor.
The method of claim 11, wherein the threshold is a fixed voltage value, and a result of the integration is proportional to an amount of thermal energy dissipated on the transistor.
The method of any one of claims 11 to 15, wherein the specified integration interval starts a fraction of a period before the transistor is turned on.
A wireless power transmitter, comprising:

a transmitter coil configured to generate a magnetic field for wirelessly charging a battery；

a power amplifier to generate alternating current (AC) power to be delivered to the transmitter coil, the power amplifier comprising a transistor and a gate driver that pulses the transistor to generate the AC power,

wherein a drain terminal of the transistor is coupled to a voltage source； and

an inductive load detector comprising:

an integrator to perform an integration of a voltage waveform at the drain terminal of the transistor over a specified integration interval； and

a comparator to compare a result of the integration to a threshold and generate an indicator based on the comparison, wherein the indicator is to trigger an event that prevents over-dissipation on the transistor under high inductive loading conditions.
The wireless power transmitter of claim 17, comprising a voltage divider coupled to the source voltage to generate the threshold, wherein the result of the integration is proportional to a reactance of the transmitter coil when an impedance of the transmitter coil is inductive.
The wireless power transmitter of claim 17, wherein the threshold is a fixed voltage value, and the result of the integration is proportional to an amount of thermal energy dissipated on the transistor.
The wireless power transmitter of any one of claims 17 to 19, wherein the specified integration interval starts a fraction of a period before the transistor is turned on.
An apparatus, comprising:

a transmitter coil configured to generate a magnetic field for wirelessly charging a battery；

a power amplifier to generate alternating current (AC) power to be delivered to the transmitter coil, the power amplifier comprising a transistor that is pulsed to generate the AC power； and

means for preventing over-dissipation on the transistor under high inductive loading conditions, the means for preventing comprising:

means for performing an integration of a drain voltage of the transistor over a specified integration interval； and

means for comparing a result of the integration to a threshold and triggering a protection event if the result of the integration reaches the threshold.
The apparatus of claim 21, comprising means for retuning the transmitter coil in response to a reactance shift of the transmitter coil, wherein the protection event is a retuning event in which the means for preventing causes the means for retuning to retune the transmitter coil to cancel the reactance.
The apparatus of claim 21, wherein the threshold is proportional to a source voltage coupled to the transistor, and a result of the integration is proportional to a reactance of the transmitter coil when an impedance of the transmitter coil is inductive.
The apparatus of claim 21, wherein the protection event is an over-dissipation protection event in which an output of the means for preventing causes a driving signal of the transistor to be disabled.
The apparatus of claim 21, wherein the threshold is a fixed voltage value, and a result of the integration is proportional to an amount of thermal energy dissipated on the transistor.