WO2018004515A1

WO2018004515A1 - Optimizing papr performance of pulse shaping filters for single carrier waveforms

Info

Publication number: WO2018004515A1
Application number: PCT/US2016/039585
Authority: WO
Inventors: Hosein Nikopour; Sarabjot SINGH
Original assignee: Intel Corporation
Priority date: 2016-06-27
Filing date: 2016-06-27
Publication date: 2018-01-04
Also published as: EP3476089A1

Abstract

A technology is described for designing a pulse shaping filter for single carrier (SC) waveforms. A pulse shaping filter can be configured to include an odd pulse shaping filter to output data with a first power metric and an even pulse shaping filter to output data with a second power metric. The first power metric and the second power metric can be derived independent of data inputted to the odd pulse shaping filter and the even pulse shaping filter. One or more filter coefficient values can be selected to apply to the odd pulse shaping filter and the even pulse shaping filter. The one or more filter coefficient values can be selected to reduce the first power metric and the second power metric. A reduction in the first power metric and the second power metric can cause the data to have a reduced peak-to-average power ratio (PAPR).

Description

OPTIMIZING PAPR PERFORMANCE OF PULSE SHAPING FILTERS FOR

SINGLE CARRIER WAVEFORMS

BACKGROUND

[0001] Wireless mobile communication technology uses various standards and protocols to transmit data between a node (e.g., a transmission station) and a wireless device (e.g., a mobile device). Some wireless devices communicate using orthogonal frequency-division multiple access (OFDMA) in a downlink (DL) transmission and single carrier frequency division multiple access (SC-FDMA) in uplink (UL). Standards and protocols that use orthogonal frequency-division multiplexing (OFDM) for signal transmission include the third generation partnership project (3 GPP) long term evolution (LTE), the Institute of Electrical and Electronics Engineers (IEEE) 802.16 standard (e.g., 802.16e, 802.16m), which is commonly known to industry groups as WiMAX (Worldwide interoperability for Microwave Access), and the IEEE 802.11 standard, which is commonly known to industry groups as WiFi.

[0002] In 3GPP radio access network (RAN) LTE systems, the node can be a

combination of Evolved Universal Terrestrial Radio Access Network (E-UTRAN) Node Bs (also commonly denoted as evolved Node Bs, enhanced Node Bs, eNodeBs, or eNBs) and Radio Network Controllers (RNCs), which communicates with the wireless device, known as a user equipment (UE). The downlink (DL) transmission can be a

communication from the node (e.g., eNodeB) to the wireless device (e.g., UE), and the uplink (UL) transmission can be a communication from the wireless device to the node.

BRIEF DESCRIPTION OF THE DRAWINGS

[0003] Features and advantages of the disclosure will be apparent from the detailed description which follows, taken in conjunction with the accompanying drawings, which together illustrate, by way of example, features of the disclosure; and, wherein:

[0004] FIG. 1 illustrates single carrier waveform modulation in a transmitter and single carrier waveform demodulation in a receiver in accordance with an example;

[0005] FIG. 2A illustrates a frequency response of a truncated root-raised cosine (RRC) pulse shaping filter in accordance with an example; [0006] FIG. 2B illustrates a frequency response of an end-to-end system in an additive white Gaussian noise (AWGN) channel in accordance with an example;

[0007] FIG. 3 illustrates a peak-to-average power ratio (PAPR) of a single carrier waveform with a pulse shaping filter in accordance with an example;

[0008] FIG. 4 illustrates a correlation between a roll-off factor and a power spreading ratio (PSR) of a single carrier waveform with a pulse shaping filter in accordance with an example;

[0009] FIG. 5 illustrates single carrier waveform modulation using an odd filter and an even filter of a root-raised cosine (RRC) pulse shaping filter in accordance with an example;

[0010] FIG. 6 illustrates power spreading ratios (PSRs) associated with an odd filter and an even filter of a pulse shaping filter in accordance with an example;

[0011] FIG. 7A illustrates an impulse response of a novel pulse shaping filter in comparison to a root-raised cosine (RRC) pulse shaping filter in accordance with an example;

[0012] FIG. 7B illustrates a frequency response of a novel pulse shaping filter in comparison to a root-raised cosine (RRC) pulse shaping filter in accordance with an example;

[0013] FIG. 8 illustrates a peak-to-average power ratio (PAPR) gain in a single carrier waveform with a novel pulse shaping filter in accordance with an example;

[0014] FIG. 9 illustrates a power spreading ratio (PSR) in a single carrier waveform with a novel pulse shaping filter in comparison to a root-raised cosine (RRC) pulse shaping filter in accordance with an example;

[0015] FIG. 10 depicts functionality of a transmit node operable to perform pulse shaping on a single carrier waveform in accordance with an example;

[0016] FIG. 11 depicts functionality of a receive node operable to receive baseband signals on a single carrier waveform in accordance with an example;

[0017] FIG. 12 depicts a flowchart of a method for designing a pulse shaping filter for single carrier (SC) waveforms in accordance with an example; [0018] FIG. 13 illustrates a diagram of a wireless device (e.g., UE) in accordance with an example; and

[0019] FIG. 14 illustrates a diagram of a wireless device (e.g., UE) in accordance with an example.

[0020] Reference will now be made to the exemplary embodiments illustrated, and specific language will be used herein to describe the same. It will nevertheless be understood that no limitation of the scope of the technology is thereby intended.

DETAILED DESCRIPTION

[0021] Before the present technology is disclosed and described, it is to be understood that this technology is not limited to the particular structures, process actions, or materials disclosed herein, but is extended to equivalents thereof as would be recognized by those ordinarily skilled in the relevant arts. It should also be understood that terminology employed herein is used for the purpose of describing particular examples only and is not intended to be limiting. The same reference numerals in different drawings represent the same element. Numbers provided in flow charts and processes are provided for clarity in illustrating actions and operations and do not necessarily indicate a particular order or sequence.

EXAMPLE EMBODIMENTS

[0022] An initial overview of technology embodiments is provided below and then specific technology embodiments are described in further detail later. This initial summary is intended to aid readers in understanding the technology more quickly but is not intended to identify key features or essential features of the technology nor is it intended to limit the scope of the claimed subject matter.

[0023] In wireless communications, there can be two types of signal waveforms - single carrier (SC) waveforms and multi-carrier (MC) waveforms. Multi carrier waveforms can be used in OFDMA, in which digital data can be transmitted over multiple narrowband carrier frequencies. Single carrier waveform in contrast transmits a digital data symbol over the entire system bandwidth which can be practically wide. [0024] In one example, single carrier waveforms can be advantageous over multicarrier waveforms in terms of peak-to-average power ratio (PAPR). Therefore, SC can be an attractive alternative to OFDMA, especially in the uplink communications where lower PAPR can benefit a user equipment (UE) in terms of transmit power efficiency. A form of SC waveform has been adopted IEEE 802.11 ad systems. In addition, low PAPR can be an important factor for millimeter-wave (mmwave) communications (which use high- frequency signals in the millimeter-wave frequency band) where lower costs of radio frequency (RF) hardware can dictate a lower dynamic range of waveform signal to limit an operation range within a linear range of an RF chain.

[0025] In one example, the PAPR can represent a dynamic range of the single carrier waveform. A high PAPR can indicate a relatively large variation for the amplitude of the single carrier waveform in the time domain. On the other hand, a low PAPR can indicate a relatively small variation for the single carrier waveform in the time domain. As previously described, a low PAPR can be advantageous for the transmitter (e.g., UE).

[0026] In one example, the PAPR of the single carrier waveform can depend on a pulse shaping filter. The pulse shaping filter can be part of the transmitter (e.g., UE or eNodeB). Pulse shaping is the process of changing the single carrier waveform of transmitted pulses, which can make a transmitted signal better suited to its purpose or the

communication channel, typically by limiting an effective bandwidth of the transmission. By filtering the transmitted data symbols, an inter-symbol interference (ISI) caused by the channel can be kept in control. Pulse shaping also can be vital for making the signal fit in its frequency band. Typically, the pulse shaping can occur after coding and modulation at the transmitter (e.g., UE or eNodeB).

[0027] In one example, the pulse shaping filter can be a root-raised cosine (RRC) filter, which can achieve no ISI by following the Nyquist criterion. In other words, the RRC filter can be used for pulse shaping in digital modulation due to its ability to minimize ISI. The RRC filter can be characterized by two values; β, a roll-off factor, and Ts, the reciprocal of a symbol-rate. The roll-off factor is a measure of an excess bandwidth of the RRC filter. The roll-off factor of the RRC filter is a parameter that governs a trade-off between ISI sensitivity and bandwidth overhead of the system. An increased roll-off factor can reduce ISI and/or timing sensitivity at the expense of increased bandwidth overhead and hence lower spectral efficiency. On the other hand, an increased roll-off factor can lower the PAPR of the single carrier waveform, however, at the expense of reduction of the spectral efficiency.

[0028] In one example, the Nyquist criterion describes conditions which, when satisfied by the communication channel, result in no ISI. When consecutive symbols (e.g., QAM) are transmitted over a channel, the impulse response (or equivalently the frequency response) of the channel can cause a transmitted symbol to be spread in the time domain. This can cause ISI because the previously transmitted symbols can affect the currently received symbol, thus reducing tolerance for noise. The Nyquist criterion relates this time-domain condition to an equivalent frequency -domain condition for the pulse shaping filter of the SC waveform.

[0029] The present technology describes a novel pulse shaping filter with relatively low PAPR without a detrimental effect to bandwidth occupation and spectral efficiency. Since the pulse shaping filter can have a direct impact on the PAPR of the single carrier waveform, the novel pulse shaping filter can be designed to with consideration of the PAPR, the ISI and the frequency response of the pulse shaping filter while not compromising the spectral efficiency of the system. An analytical framework is designed to derive filter coefficients aimed to minimize a power spreading ratio (PSR), thereby resulting in a minimized PAPR, which can be subject to pass band and stop band ripple of the frequency response of the pulse shaping filter. Depending on a given bandwidth, the novel pulse shaping filter described herein can reduce the PAPR by up to three decibels (dB) as compared to an existing root-raised cosine (RRC) pulse shaping filter.

[0030] FIG. 1 illustrates an example of single carrier waveform modulation in a transmitter (Tx) and single carrier waveform demodulation in a receiver (Rx). The transmitter can be a UE or an eNodeB, and the receiver can be a UE or an eNodeB.

Block-wise single carrier (BWSC) is a technique of single carrier modulation, which is implemented through baseband processing in the digital domain. The transmitter can receive a block of quadrature amplitude modulation (QAM) symbols. The transmitter can perform up sampling (e.g., rate 2) and then provide the symbols to a pulse shaping filter. The up sampling rate can provide additional bandwidth (i.e., above Nyquist) for expansion of the pulse shaping filter in the frequency domain. In one example, the pulse shaping filter can be a discrete root-raised cosine (RRC) filter

[0031] Signals from the transmitter can be received at a match filter of the receiver. The match filter can maximize the SNR of the output symbols at the receiver. A match filter is "matched" to the pulse shaping filter of the transmitter. The match filter can be an RRC filter. After the signal passes through the match filter, the receiver can perform down sampling (e.g., rate 2). Therefore, the transmitter can perform single carrier waveform modulation and the receiver can perform single carrier waveform demodulation in the baseband domain using discrete shaping and match filters.

[0032] In one example, to enable digital domain processing, an impulse response of the pulse shaping filter (e.g., the RRC filter as described above) can be discretized and truncated to form a baseband finite impulse response (FIR) filter. In other words, the FIR filter can be a discrete time and truncated RRC filter. The FIR filter has an impulse response (or response to any finite length input) of finite duration, because it settles to zero in finite time.

[0033] FIG. 2A illustrates an example of a frequency response of a truncated desecrate root-raised cosine (RRC) pulse shaping filter. More specifically, FIG. 2A illustrates a frequency response for an analog RRC filter with infinite length, a frequency response for a discrete time and truncated RRC filter (35 taps), and a frequency response for a discrete time and truncated RRC filter (19 taps). The discretization of the analog RRC filter involves sampling the analog RRC filter and performing a truncation to a limited time duration, which results in ripple in the frequency response or spectrum. The frequency response can depend on the length of truncation or the length of the RRC filter and other parameters, such as roll-off factor. In the example shown in FIG. 2A, the roll-off factor can be 0.1 and an up sampling rate can be two. As previously described, the roll-off factor is a measure of an excess bandwidth of analog or discrete time and truncated RRC filter.

[0034] In one example, the filter taps can indicate coefficient/delay pairs. The number of filter taps, (often designated as "N") can indicate a level of filtering that is performed. For example, an increased number of filter taps results in more complexity, but can result in greater stopband attenuation, less ripple, etc. As shown in FIG. 2A, the frequency response of the analog RRC filter can be considered ideal since the analog RRC filter does not produce side lobes of a transmit signal. On the other hand, the discrete time and truncated RRC filter can produce side lobes, although the side lobes become less pronounced as the number of taps increases. For example, the discrete time and truncated RRC filter with 35 taps can produce less pronounced side lobes as compared to the discrete time and truncated RRC filter with 19 taps. As shown, the discrete time and truncated RRC filter can produce in-band frequency fluctuation and out-of-band sidelobes.

[0035] FIG. 2B illustrates an example of a frequency response of an end-to-end system in an additive white Gaussian noise (AWGN) channel. The end-to-end system can comprise a transmitter with a pulse shaping filter and a receiver with a match filter (as shown in FIG. 1). In other words, the frequency response can represent an input to the transmitter to an output of the receiver (which includes the pulse shaping and matching). In the example shown in FIG. 2B, the roll-off factor can be 0.1 and an up sampling rate can be two. For the end-to-end system, a flat frequency response is ideal, as produced by the analog RRC filter to avoid ISI. On the other hand, the discrete time and truncated RRC filter results in some fluctuations in frequency due to the truncation and discretization. However, the fluctuations in frequency can decrease based on the number of taps. For example, the discrete time and truncated RRC filter with 35 taps can produce less frequency fluctuation as compared to the discrete time and truncated RRC filter with 19 taps. In addition, the discrete time and truncated RRC filter can impose residual ISI after the match filter.

[0036] In one example, the single carrier waveform with the pulse shaping filter (e.g., FIR filter) at the transmitter can be associated with a defined PAPR. In other words, the output signal at the transmitter can be associated with the defined PAPR. As previously explained, the PAPR of the single carrier waveform can depend on the pulse shaping filter.

[0037] FIG. 3 illustrates an exemplary statistical distribution (ccdf) of peak-to-average power ratio (PAPR) of a single carrier waveform with a pulse shaping filter. More specifically, FIG. 3 illustrates the PAPR for single carrier waveforms and the PAPR for multi carrier waveforms (such as OFDM and SC-FDMA as defined in 3GPP LTE standards) for a root-raised cosine (RRC) pulse shaping filter. The RRC pulse shaping filter can be a discrete time and truncated RRC filter (which can also be referred to as a FIR filter). As shown, the PAPR of the single carrier waveform can be less than the PAPR of the multi carrier waveforms. In addition, within SC, different PAPRs can be achieved depending on various pulse shaping filter parameters. One such parameter that can have an impact on the PAPR is the roll-off factor of the RRC pulse shaping factor. As shown in FIG. 3, different roll-off factors (e.g., 0.055, 0.1, 0.2, 0.3, 0.5, and 0.95) can produce different values of PAPR for the single carrier waveform. In one example, the PAPR can reduce when the roll-off factor increases. For example, a roll-off factor of 0.1 can result in a higher PAPR as compared to a roll-off factor of 0.8. When designing the pulse shaped filter, the goal can be to reduce the PAPR since a reduced PAPR can be advantageous for the transmitter. However, the reduction in the PAPR by increasing roll-off factor can be at the expense of a larger bandwidth, which leads to lower spectral efficiency.

[0038] In one example, one difficulty with PAPR optimization is that the PAPR depends on the input signal (or input data) to the pulse shaping filter. In other words, as the input signal changes, the PAPR at the output also changes. As a result, designing the pulse shaping filter based on the input signal can be difficult because the input signal can be random. Therefore, in this invention a framework is developed to design the pulse shaping filter independent of the input signal while statistically achieving low PAPR.

[0039] In one example, the pulse shaping filter can be optimized based on a metric referred to as a power spreading ratio (PSR). The PSR can depend on the filter taps, whereas the PAPR can depend on the filter taps and the input signal (or input data) to the pulse shaping filter. However, the PSR can indirectly represent the PAPR. The benefit of the PSR is that it is independent of the input signal to the pulse shaping filter. The PSR can be a filter-only representation of the PAPR characteristic of a single carrier waveform for a given pulse shaping filter. The PSR can be incorporated when designing the pulse shaping filter in order to achieve a desired PAPR for single carrier waveform.

[0040] In one example, an output of filter h_n for input x_n is described as y_n =

∑c h_kx_n-_k. Every sample of y_n is the linear combination of multiple input samples. Assuming the maximum tap of the pulse shaping filter (e.g., the discrete time and truncated RRC filter or the FIR filter) is at n₀, the normalized power contribution of other neighboring samples to output sample at n₀ can be characterized by a power distribution factor which is described as Power spreading ratio(PSR) = - 1

where n₀ = arg max( | h_n | ²) . Based on the equation above, a larger PSR can result in a n

larger PAPR, and vice versa.

[0041] FIG. 4 illustrates a correlation between a roll-off factor and a power spreading ratio (PSR) of a single carrier waveform with a pulse shaping filter. The pulse shaping filter can be a discrete time and truncated root-raised cosine (RRC) filter (which can also be referred to as a FIR filter). As shown, the PSR can decrease by the roll-off factor. The roll-off factor can be of the pulse shaping filter (e.g., RRC filter). As the roll-off factor increases, a power spreading ratio (PSR) can decrease. Similarly, as the roll-off factor increases, a peak-to-average power ratio (PAPR) can decrease. For example, a roll-off factor of 0.1 can result in a higher PSR as compared to a roll-off factor of 0.8, and therefore, a higher PAPR. Therefore, high PSR can be equivalent to high PAPR and vice versa. Also, low PSR can be equivalent to low PAPR and vice versa.

[0042] FIG. 5 illustrates single carrier waveform modulation using an odd filter and an even filter of a pulse shaping filter. The pulse shaping filter can be a discrete time and truncated root-raised cosine (RRC) filter (which can also be referred to as a FIR filter). As previously described with respect to FIG. 1, single carrier waveform modulation at a transmitter can involve up sampling and pulse shaping via a pulse shaping filter. As shown in FIG. 5, the pulse shaping filter at the transmitter can be split into two separate filters - an odd filter and an even filter. The odd filter can form from odd taps of the pulse shaping filter, wherein an indexing of the pulse shaping starts from 0, i.e., 0, 1, 2, 3, and so on, and the even filter can form from even taps of the pulse shaping filter. In other words, a single carrier waveform modulation process can be represented by odd and even filters. The odd and even filters can be for a sampling rate of 2. In this example, there is no direct up sampling. Rather, the input signal (also referred to as input data or discrete data) can be provided to both the odd filter and the even filter, and then samples are merged together to produce a baseband upsampled output signal. The output of the odd filter can be merged with the output of the even filter to create the merged upsampled signal. The input signal can be a block of QAM symbols. The baseband output signal can be equivalent to an up sampled signal that has passed through a pulse shaping filter (as described in FIG. 1). The transmitter block of FIG. 1 can be equivalent to FIG. 5.

[0043] FIG. 6 illustrates exemplary power spreading ratios (PSRs) associated with an odd filter and an even filter of a pulse shaping filter. The pulse shaping filter can be a discrete time and truncated root-raised cosine (RRC) filter (which can also be referred to as a FIR filter). As shown in FIG. 6, the PSR of the even filter can be higher than the PSR of the odd filter. There can be a large difference (or imbalance) between the PSRs of the odd and even filters. The PSR can depend on the roll-off factor (except for the PSR of the odd filter in which the roll-off factor has a minimal impact). The PSR of a full filter (which includes both the odd filter and even filter, which is equivalent to the original pulse shaping filter) can be in between the PSRs of the even filter and the odd filter. As shown in FIG. 6, the PSR can indicate that the even filter is a main contributor to the PAPR of the single carrier waveform when the RRC filter is used for pulse shaping.

[0044] In one configuration, a pulse shaping filter can be designed to have a substantially equal power contribution between the even and odd filters. The pulse shaping filter can also be designed to have a low PSR (and PAPR) of the even and odd filters. In addition, the pulse shaping filter can be a low pass filter with a desired stopband and transition band. The passband can be a range of frequencies that can pass through the pulse shaping filter, the stopband can be a band of frequencies through which the pulse shaping filter does not allow signals to pass, and the transition band is a range of frequencies that allow for a transition between the passband and the stopband of the pulse shaping filter. In other words, based on the desired stopband and transition band, the pulse shaping filter can be designed to have a favorable frequency response. When these design criterion are met for the pulse shaping filter, the output from the transmitter can be a low PAPR baseband signal with a desired spectrum.

[0045] In one configuration, the pulse shaping filter can be designed such that the single carrier waveform achieves desired targets in terms of frequency response or spectrum and PAPR. The pulse shaping filter can be applied to millimeter wave (mmwave) systems for example within the context of 5G cellular or WiFi systems. The pulse shaping filter can be optimized with the following example set of inputs: an FIR pulse shaping filter represented by h = [ i₀/ _.■■■ an even filter represented by h_e = [h₀h₂ ... an odd filter represented by h₀ = [h_th₃ ... h_N_₂], a number of taps, frequency response samples represented by N_FFT = 1SN, a roll-off factor represented as /?, a passband frequency represented by f_p = 0.25 (assuming the upsampling rate is 2), a stopband frequency represented by f_s = 0.25(1 + /?), a weighting between odd and even filters represented by 0 < a < 1, frequency weighting diagonal matrices W_s, W_p, and stopband and passband limits of 5_S, δ_ρ. The values provided for the frequency response samples, the roll-off factor and the stopband frequency are mere examples and are not intended to be limiting. As an example, the number of taps can be represented by N = 4L + 1, but in another example, the number of taps can be an even value. Based on the above, the following equations can be derived: D_p = l exp ("TT""")] - k = 0, ... ,

n = 0, ... , N - 1, and D_s = [exp (^)] , k = \f_sN_FFT], ... , [0.SN_FFT\, n = 0, ... , N - 1 where D_p and D_s represent a Discrete Fourier transform (DFT) basis in pass and stop bands, respectively. In other words, the above inputs can be defined. Therefore, in one example, design criterion can be derived for the pulse shaping filter that is independent of the input signal (or input data). One such example is the PSR, which can be input signal independent. Rather, the PSR can depend on the filter taps.

[0046] In one configuration, the pulse shaping filter can be optimized based on the following equation to minimize the weighted sum of PSRs of even and odd filters as expressed below:

subject to the following constraints for pass band and stop bands of the shaping filter frequency response:

|W_sD_sh| < 5_S

1 - δ_ρ≤ |W_pD_ph| < 1 + δ_ρ

[0047] In one example, the above optimization formula (subject to constraints) for the pulse shaping filter can be a non-convex optimization equation, which can be optimal but difficult to solve. Therefore, the non-convex optimization equation can be converted to a convex suboptimal optimization problem, which can be solved more easily. The suboptimal optimization problem can minimize the weighed sum of (square-root) PSR of even and odd filters, as follows: h^* = arg min \\h_even\\ + a ||h_odd ||

h

subject to the following constraints on the maximum taps of the even/odd filters and the stopband of the pulse shaping filter, i.e.,:

m_e = [N/2\ = 2L, m₀ = [N/21 = 2L + 1,

h_m<j = 1, h_me = γ

| D_sh| < 8_S

[0048] In one example, a normalization can be performed as follows:

h⁺ = hV||h^* ||

|W_sD_sh⁺ | < <Vllh^* ||

[0049] In one example, outer loops can be applied to optimize m_e, m₀, h_mo, and γ assuming h_me = 1, 8_S can be set by multiple offline trials to reach a desired stopband attenuation, and a passband ripple constraint can be relaxed. The bandwidth can be guaranteed by an indirect constraint, such as h_mo = h_me = 1.

[0050] FIG. 7 A illustrates an exemplary impulse response of a novel pulse shaping filter in comparison to a discrete root-raised cosine (RRC) pulse shaping filter. The novel pulse shaping filter can be designed based on the inputs described above. In this example, the roll-off factor can be 0.1, the up sampling rate can be 2, and the number of filter taps can be 35. The novel pulse shaping filter and the RRC pulse shaping filter can have the same number of filter taps and the same stopband attenuation.

[0051] FIG. 7B illustrates an exemplary frequency response of a novel pulse shaping filter in comparison to a discrete root-raised cosine (RRC) pulse shaping filter. The novel pulse shaping filter can be designed based on the inputs described above. In this example, the roll-off factor can be 0.1, the up sampling rate can be 2, and the number of filter taps can be 35. The novel pulse shaping filter and the RRC pulse shaping filter can have the same number of filter taps and the same stopband attenuation. In this example, the weights utilized for the filter design are a value of 1.

[0052] FIG. 8 illustrates an exemplary peak-to-average power ratio (PAPR) gain in a single carrier waveform with a novel pulse shaping filter. In this example, the novel pulse shaping filter can be one of two stopband limits (e.g., -21.75 dB or -5 dB) and with a roll- off factor of 0.1. In addition, FIG. 8 illustrates the PAPR for single carrier (SC) waveforms with a root-raised cosine (RRC) pulse shaping filter. The RRC pulse shaping filter can be a discrete time and truncated RRC filter (which can also be referred to as a FIR filter). Within SC waveform, different PAPRs can be achieved depending on various pulse shaping filter parameters. One such parameter that can have an impact on the PAPR is the roll-off factor of the RRC pulse shaping factor. As shown in FIG. 8, different roll- off factors (e.g., 0.055, 0.1, 0.2, 0.3, 0.5, and 0.95) can produce different values of PAPR for the single carrier waveform. With respect to the RRC pulse shaping factor, the PAPR can reduce when the roll-off factor increases. In addition, the PAPR gain can be higher when a stopband attenuation is reduced. For example, a -3.5 dB PAPR gain can be achieved between the novel pulse shaping filter (with a stopband limit of -21.75 dB and a roll-off factor of 0.1) and the RRC pulse shaping filter with a roll-off value of 0.1.

Therefore, in this example, even though the roll-off factor is the same as RRC, the PAPR is 3.5 dB lower with the novel pulse shaping filter, which can be beneficial to the transmitter in terms of transmit power efficiency.

[0053] FIG. 9 illustrates an example of a power spreading ratio (PSR) in a single carrier waveform with a novel pulse shaping filter in comparison to a discrete root-raised cosine (RRC) pulse shaping filter. In the RRC pulse shaping filter, there is a large imbalance between the even filter and the odd filter in terms of the PSR. In contrast, for the novel pulse shaping filter (with a stopband limit of -21.75 dB), the PSR for the even and odd filters can be relatively balanced. Similarly, for the novel pulse shaping filter (with a stopband limit of -5 dB), the PSR for the even and odd filters can be relatively balanced. In addition, the PSR for the novel pulse shaping filter can be lower as compared to the PSR for the RRC pulse shaping filter. The lower PSR can result in a lower PAPR, which can be beneficial to the transmitter in terms of transmit power efficiency.

[0054] In one configuration, the pulse shaping filter for single carrier waveforms can be designed to include both the PAPR and the filter frequency response. The PAPR can be represented by a design criterion referred to as a power distribution factor (PSR) or another relevant function of filter coefficients in a filter optimization process. The pulse shaping filter can be split into odd and even filters, and the odd and even filters for baseband processing of the single carrier waveform can be jointly designed. The pulse shaping filter can be designed using a suboptimal convex approximation. The pulse shaping filter can be restricted to a linear phase structure. In one example, the pulse shaping filter with low PAPR (at a transmit node) can be matched at a receiver node. In addition, the single carrier waveform can be a block-wise single carrier (BWSC) waveform.

[0055] Another example provides functionality 1000 of a transmit node operable to perform pulse shaping on a single carrier waveform, as shown in FIG. 10. The transmit node can comprise one or more processors and memory configured to: identify a plurality of modulated symbols to be transmitted to a receive node, as in block 1010. The transmit node can comprise one or more processors and memory configured to: distribute the plurality of modulated symbols to a pulse shaping filter, wherein even modulated symbols are distributed to an even pulse shaping filter and odd modulated symbols are distributed to an odd pulse shaping filter, as in block 1020. The transmit node can comprise one or more processors and memory configured to: determine a first power metric for an output of the odd pulse shaping filter and a second power metric for an output of the even pulse shaping filter, wherein one or more filter coefficient values are selected to reduce the first power metric and the second power metric, as in block 1030. The transmit node can comprise one or more processors and memory configured to: scale one of the output of the odd pulse shaping filter or the output of the even pulse shaping filter to provide a substantially equal power contribution of the odd pulse shaping filter and the even pulse shaping filter, as in block 1040. The transmit node can comprise one or more processors and memory configured to: merge the output of the odd pulse shaping filter and the output of the even pulse shaping filter to produce a baseband signal for transmission from the transmit node to the receive node, wherein a reduction in the first power metric and the second power metric causes the baseband signal to have a reduced peak-to-average power ratio (PAPR), as in block 1050.

[0056] Another example provides functionality 1100 of a receive node operable to receive baseband signals on a single carrier waveform, as shown in FIG. 11. The receive node can comprise one or more processors and memory configured to: identify a baseband signal received from a transmit node, as in block 1110. The receive node can comprise one or more processors and memory configured to: filter the baseband signal using a match filter at the receive node, wherein the match filter is configured to match a pulse shaping filter at the transmit node, wherein the pulse shaping filter at the transmit node includes an odd pulse shaping filter and an even pulse shaping filter, wherein an output of the odd pulse shaping filter and an output of the even pulse shaping filter are merged to produce the baseband signal, as in block 1120.

[0057] Another example provides a method for designing a pulse shaping filter for single carrier (SC) waveforms, as shown in FIG. 12. The method can be executed as instructions on a machine, where the instructions are included on at least one computer readable medium or one non-transitory machine readable storage medium. The method can include: configuring a pulse shaping filter to include an odd pulse shaping filter to output data with a first power metric and an even pulse shaping filter to output data with a second power metric, wherein the first power metric and the second power metric are derived independent of data inputted to the odd pulse shaping filter and the even pulse shaping filter, wherein the data is modulated on the SC waveform, as in block 1210. The method can include: selecting a weighting value to enable a substantially equal power contribution between the odd pulse shaping filter and the even pulse shaping filter, as in block 1220. The method can include: selecting one or more filter coefficient values to apply to the odd pulse shaping filter and the even pulse shaping filter, wherein the one or more filter coefficient values are selected to reduce the first power metric and the second power metric, wherein a reduction in the first power metric and the second power metric causes the data to have a reduced peak-to-average power ratio (PAPR), as in block 1230.

[0058] FIG. 13 provides an example illustration of a user equipment (UE) device 1300, such as a wireless device, a mobile station (MS), a mobile wireless device, a mobile communication device, a tablet, a handset, or other type of wireless device. The UE device 1300 can include one or more antennas configured to communicate with a node 1920 or transmission station, such as a base station (BS), an evolved Node B (eNB), a baseband unit (BBU), a remote radio head (RRH), a remote radio equipment (RRE), a relay station (RS), a radio equipment (RE), a remote radio unit (RRU), a central processing module (CPM), or other type of wireless wide area network (WWAN) access point. The node 1920 can include one or more processors 1922 and memory 1924. The UE device 1300 can be configured to communicate using at least one wireless communication standard including 3GPP LTE, WiMAX, High Speed Packet Access (HSPA), Bluetooth, and WiFi. The UE device 1300 can communicate using separate antennas for each wireless communication standard or shared antennas for multiple wireless communication standards. The UE device 1300 can communicate in a wireless local area network (WLAN), a wireless personal area network (WPAN), and/or a WWAN.

[0059] In some embodiments, the UE device 1300 may include application circuitry 1302, baseband circuitry 1304, Radio Frequency (RF) circuitry 1306, front-end module (FEM) circuitry 1308 and one or more antennas 1310, coupled together at least as shown.

[0060] The application circuitry 1302 may include one or more application processors. For example, the application circuitry 1302 may include circuitry such as, but not limited to, one or more single-core or multi-core processors. The processor(s) may include any combination of general-purpose processors and dedicated processors (e.g., graphics processors, application processors, etc.). The processors may be coupled with and/or may include a storage medium, and may be configured to execute instructions stored in the storage medium to enable various applications and/or operating systems to run on the system.

[0061] The baseband circuitry 1304 may include circuitry such as, but not limited to, one or more single-core or multi-core processors. The baseband circuitry 1304 may include one or more baseband processors and/or control logic to process baseband signals received from a receive signal path of the RF circuitry 1306 and to generate baseband signals for a transmit signal path of the RF circuitry 1306. Baseband processing circuity 1304 may interface with the application circuitry 1302 for generation and processing of the baseband signals and for controlling operations of the RF circuitry 1306. For example, in some embodiments, the baseband circuitry 1304 may include a second generation (2G) baseband processor 1304a, third generation (3G) baseband processor 1304b, fourth generation (4G) baseband processor 1304c, and/or other baseband processor(s) 1304d for other existing generations, generations in development or to be developed in the future (e.g., fifth generation (5G), 6G, etc.). The baseband circuitry 1304 (e.g., one or more of baseband processors 1304a-d) may handle various radio control functions that enable communication with one or more radio networks via the RF circuitry 1306. The radio control functions may include, but are not limited to, signal modulation/demodulation, encoding/decoding, radio frequency shifting, etc. In some embodiments, modulation/demodulation circuitry of the baseband circuitry 1304 may include Fast-Fourier Transform (FFT), precoding, and/or constellation

mapping/demapping functionality. In some embodiments, encoding/decoding circuitry of the baseband circuitry 1304 may include convolution, tail-biting convolution, turbo, Viterbi, and/or Low Density Parity Check (LDPC) encoder/decoder functionality.

Embodiments of modulation/demodulation and encoder/decoder functionality are not limited to these examples and may include other suitable functionality in other embodiments.

[0062] In some embodiments, the baseband circuitry 1304 may include elements of a protocol stack such as, for example, elements of an evolved universal terrestrial radio access network (EUTRAN) protocol including, for example, physical (PHY), media access control (MAC), radio link control (RLC), packet data convergence protocol (PDCP), and/or radio resource control (RRC) elements. A central processing unit (CPU) 1304e of the baseband circuitry 1304 may be configured to run elements of the protocol stack for signaling of the PHY, MAC, RLC, PDCP and/or RRC layers. In some embodiments, the baseband circuitry may include one or more audio digital signal processor(s) (DSP) 1304f. The audio DSP(s) 104f may be include elements for compression/decompression and echo cancellation and may include other suitable processing elements in other embodiments. Components of the baseband circuitry may be suitably combined in a single chip, a single chipset, or disposed on a same circuit board in some embodiments. In some embodiments, some or all of the constituent components of the baseband circuitry 1304 and the application circuitry 1302 may be implemented together such as, for example, on a system on a chip (SOC).

[0063] In some embodiments, the baseband circuitry 1304 may provide for

communication compatible with one or more radio technologies. For example, in some embodiments, the baseband circuitry 1304 may support communication with an evolved universal terrestrial radio access network (EUTRAN) and/or other wireless metropolitan area networks (WMAN), a wireless local area network (WLAN), a wireless personal area network (WPAN). Embodiments in which the baseband circuitry 1304 is configured to support radio communications of more than one wireless protocol may be referred to as multi-mode baseband circuitry.

[0064] The RF circuitry 1306 may enable communication with wireless networks using modulated electromagnetic radiation through a non-solid medium. In various embodiments, the RF circuitry 1306 may include switches, filters, amplifiers, etc. to facilitate the communication with the wireless network. RF circuitry 1306 may include a receive signal path which may include circuitry to down-convert RF signals received from the FEM circuitry 1308 and provide baseband signals to the baseband circuitry 1304. RF circuitry 1306 may also include a transmit signal path which may include circuitry to up-convert baseband signals provided by the baseband circuitry 1304 and provide RF output signals to the FEM circuitry 1308 for transmission.

[0065] In some embodiments, the RF circuitry 1306 may include a receive signal path and a transmit signal path. The receive signal path of the RF circuitry 1306 may include mixer circuitry 1306a, amplifier circuitry 1306b and filter circuitry 1306c. The transmit signal path of the RF circuitry 1306 may include filter circuitry 1306c and mixer circuitry 1306a. RF circuitry 1306 may also include synthesizer circuitry 1306d for synthesizing a frequency for use by the mixer circuitry 1306a of the receive signal path and the transmit signal path. In some embodiments, the mixer circuitry 1306a of the receive signal path may be configured to down-convert RF signals received from the FEM circuitry 1308 based on the synthesized frequency provided by synthesizer circuitry 1306d. The amplifier circuitry 1306b may be configured to amplify the down-converted signals and the filter circuitry 1306c may be a low-pass filter (LPF) or band-pass filter (BPF) configured to remove unwanted signals from the down-converted signals to generate output baseband signals. Output baseband signals may be provided to the baseband circuitry 1304 for further processing. In some embodiments, the output baseband signals may be zero-frequency baseband signals, although this is not a requirement. In some embodiments, mixer circuitry 1306a of the receive signal path may comprise passive mixers, although the scope of the embodiments is not limited in this respect.

[0066] In some embodiments, the mixer circuitry 1306a of the transmit signal path may be configured to up-convert input baseband signals based on the synthesized frequency provided by the synthesizer circuitry 1306d to generate RF output signals for the FEM circuitry 1308. The baseband signals may be provided by the baseband circuitry 1304 and may be filtered by filter circuitry 1306c. The filter circuitry 1306c may include a low-pass filter (LPF), although the scope of the embodiments is not limited in this respect.

[0067] In some embodiments, the mixer circuitry 1306a of the receive signal path and the mixer circuitry 1306a of the transmit signal path may include two or more mixers and may be arranged for quadrature down-conversion and/or up-conversion respectively. In some embodiments, the mixer circuitry 1306a of the receive signal path and the mixer circuitry 1306a of the transmit signal path may include two or more mixers and may be arranged for image rejection (e.g., Hartley image rejection). In some embodiments, the mixer circuitry 1306a of the receive signal path and the mixer circuitry 1306a may be arranged for direct down-conversion and/or direct up-conversion, respectively. In some embodiments, the mixer circuitry 1306a of the receive signal path and the mixer circuitry 1306a of the transmit signal path may be configured for super-heterodyne operation.

[0068] In some embodiments, the output baseband signals and the input baseband signals may be analog baseband signals, although the scope of the embodiments is not limited in this respect. In some alternate embodiments, the output baseband signals and the input baseband signals may be digital baseband signals. In these altemate embodiments, the RF circuitry 1306 may include analog-to-digital converter (ADC) and digital-to-analog converter (DAC) circuitry and the baseband circuitry 1304 may include a digital baseband interface to communicate with the RF circuitry 1306.

[0069] In some dual-mode embodiments, a separate radio IC circuitry may be provided for processing signals for each spectrum, although the scope of the embodiments is not limited in this respect.

[0070] In some embodiments, the synthesizer circuitry 1306d may be a fractional -N synthesizer or a fractional N/N+l synthesizer, although the scope of the embodiments is not limited in this respect as other types of frequency synthesizers may be suitable. For example, synthesizer circuitry 1306d may be a delta-sigma synthesizer, a frequency multiplier, or a synthesizer comprising a phase-locked loop with a frequency divider.

[0071] The synthesizer circuitry 1306d may be configured to synthesize an output frequency for use by the mixer circuitry 1306a of the RF circuitry 1306 based on a frequency input and a divider control input. In some embodiments, the synthesizer circuitry 1306d may be a fractional N/N+l synthesizer.

[0072] In some embodiments, frequency input may be provided by a voltage controlled oscillator (VCO), although that is not a requirement. Divider control input may be provided by either the baseband circuitry 1304 or the applications processor 1302 depending on the desired output frequency. In some embodiments, a divider control input (e.g., N) may be determined from a look-up table based on a channel indicated by the applications processor 1302.

[0073] Synthesizer circuitry 1306d of the RF circuitry 1306 may include a divider, a delay-locked loop (DLL), a multiplexer and a phase accumulator. In some embodiments, the divider may be a dual modulus divider (DMD) and the phase accumulator may be a digital phase accumulator (DPA). In some embodiments, the DMD may be configured to divide the input signal by either N or N+l (e.g., based on a carry out) to provide a fractional division ratio. In some example embodiments, the DLL may include a set of cascaded, tunable, delay elements, a phase detector, a charge pump and a D-type flip-flop. In these embodiments, the delay elements may be configured to break a VCO period up into Nd equal packets of phase, where Nd is the number of delay elements in the delay line. In this way, the DLL provides negative feedback to help ensure that the total delay through the delay line is one VCO cycle.

[0074] In some embodiments, synthesizer circuitry 1306d may be configured to generate a carrier frequency as the output frequency, while in other embodiments, the output frequency may be a multiple of the carrier frequency (e.g., twice the carrier frequency, four times the carrier frequency) and used in conjunction with quadrature generator and divider circuitry to generate multiple signals at the carrier frequency with multiple different phases with respect to each other. In some embodiments, the output frequency may be a LO frequency (fLO). In some embodiments, the RF circuitry 1306 may include an IQ/polar converter.

[0075] FEM circuitry 1308 may include a receive signal path which may include circuitry configured to operate on RF signals received from one or more antennas 1310, amplify the received signals and provide the amplified versions of the received signals to the RF circuitry 1306 for further processing. FEM circuitry 1308 may also include a transmit signal path which may include circuitry configured to amplify signals for transmission provided by the RF circuitry 1306 for transmission by one or more of the one or more antennas 1310.

[0076] In some embodiments, the FEM circuitry 1308 may include a TX/RX switch to switch between transmit mode and receive mode operation. The FEM circuitry may include a receive signal path and a transmit signal path. The receive signal path of the FEM circuitry may include a low-noise amplifier (LNA) to amplify received RF signals and provide the amplified received RF signals as an output (e.g., to the RF circuitry 1306). The transmit signal path of the FEM circuitry 1308 may include a power amplifier (PA) to amplify input RF signals (e.g., provided by RF circuitry 1306), and one or more filters to generate RF signals for subsequent transmission (e.g., by one or more of the one or more antennas 1310.

[0077] FIG. 14 provides an example illustration of the wireless device, such as a user equipment (UE), a mobile station (MS), a mobile wireless device, a mobile

communication device, a tablet, a handset, or other type of wireless device. The wireless device can include one or more antennas configured to communicate with a node, macro node, low power node (LPN), or, transmission station, such as a base station (BS), an evolved Node B (eNB), a baseband processing unit (BBU), a remote radio head (RRH), a remote radio equipment (RRE), a relay station (RS), a radio equipment (RE), or other type of wireless wide area network (WWAN) access point. The wireless device can be configured to communicate using at least one wireless communication standard such as, but not limited to, 3 GPP LTE, WiMAX, High Speed Packet Access (HSPA), Bluetooth, and WiFi. The wireless device can communicate using separate antennas for each wireless communication standard or shared antennas for multiple wireless communication standards. The wireless device can communicate in a wireless local area network

(WLAN), a wireless personal area network (WPAN), and/or a WWAN. The wireless device can also comprise a wireless modem. The wireless modem can comprise, for example, a wireless radio transceiver and baseband circuitry (e.g., a baseband processor). The wireless modem can, in one example, modulate signals that the wireless device transmits via the one or more antennas and demodulate signals that the wireless device receives via the one or more antennas.

[0078] FIG. 14 also provides an illustration of a microphone and one or more speakers that can be used for audio input and output from the wireless device. The display screen can be a liquid crystal display (LCD) screen, or other type of display screen such as an organic light emitting diode (OLED) display. The display screen can be configured as a touch screen. The touch screen can use capacitive, resistive, or another type of touch screen technology. An application processor and a graphics processor can be coupled to internal memory to provide processing and display capabilities. A non-volatile memory port can also be used to provide data input/output options to a user. The non-volatile memory port can also be used to expand the memory capabilities of the wireless device. A keyboard can be integrated with the wireless device or wirelessly connected to the wireless device to provide additional user input. A virtual keyboard can also be provided using the touch screen.

Examples

[0079] The following examples pertain to specific technology embodiments and point out specific features, elements, or actions that can be used or otherwise combined in achieving such embodiments.

[0080] Example 1 includes an apparatus of a transmit node operable to perform pulse shaping on a single carrier waveform, the apparatus comprising one or more processors and memory configured to: identify a plurality of modulated symbols to be transmitted to a receive node; distribute the plurality of modulated symbols to a pulse shaping filter, wherein even modulated symbols are distributed to an even pulse shaping filter and odd modulated symbols are distributed to an odd pulse shaping filter; determine a first power metric for an output of the odd pulse shaping filter and a second power metric for an output of the even pulse shaping filter, wherein one or more filter coefficient values are selected to reduce the first power metric and the second power metric; scale one of the output of the odd pulse shaping filter or the output of the even pulse shaping filter to provide a substantially equal power contribution of the odd pulse shaping filter and the even pulse shaping filter; and merge the output of the odd pulse shaping filter and the output of the even pulse shaping filter to produce a baseband signal for transmission from the transmit node to the receive node, wherein a reduction in the first power metric and the second power metric causes the baseband signal to have a reduced peak-to-average power ratio (PAPR).

[0081] Example 2 includes the apparatus of Example 1, further comprising a transceiver configured to transmit the baseband signal with reduced PAPR to the receive node.

[0082] Example 3 includes the apparatus of any of Examples 1 to 2, wherein the pulse shaping filter is represented by h = [ i₀/ _.■■■ the odd pulse shaping filter is represented by h₀ = [h- h₃ ... h_N_₂], and the even pulse shaping filter is represented by h_e = [h₀h₂ ... ijv-i]. [0083] Example 4 includes the apparatus of any of Examples 1 to 3, wherein the one or more filter coefficient values include a number of filter taps (N), a number of frequency response samples (N_FFT), a roll-off factor (/?), a passband frequency (f_p), a stopband frequency (f_s), the weighting value (a) that is between 0 and 1, frequency weighting diagonal matrices (W_s, W_p), and stopband and passband limits (5_S, δ_ρ).

[0084] Example 5 includes the apparatus of any of Examples 1 to 4, wherein the one or more filter coefficient values achieve a desired stopband and transition band in a frequency response produced by the pulse shaping filter.

[0085] Example 6 includes the apparatus of any of Examples 1 to 5, wherein the first power metric and the second power metric include power spreading ratios (PSRs).

[0086] Example 7 includes the apparatus of any of Examples 1 to 6, wherein the pulse shaping filter is utilized in a millimeter wave (mmwave) communication system.

[0087] Example 8 includes the apparatus of any of Examples 1 to 7, wherein: the transmit node is a user equipment (UE) or an eNodeB; and the receive node is a user equipment (UE) or an eNodeB.

[0088] Example 9 includes an apparatus of a receive node operable to receive baseband signals on a single carrier waveform, the receive node comprising one or more processors and memory configured to: identify a baseband signal received from a transmit node; and filter the baseband signal using a match filter at the receive node, wherein the match filter is configured to match a pulse shaping filter at the transmit node, wherein the pulse shaping filter at the transmit node includes an odd pulse shaping filter and an even pulse shaping filter, wherein an output of the odd pulse shaping filter and an output of the even pulse shaping filter are merged to produce the baseband signal.

[0089] Example 10 includes the apparatus of Example 9, further comprising a transceiver configured to receive the baseband signal from the transmit node.

[0090] Example 11 includes the apparatus of any of Examples 9 to 10, wherein the pulse shaping filter is configured to distribute modulated symbols to an odd pulse shaping filter and an even pulse shaping filter.

[0091] Example 12 includes the apparatus of any of Examples 9 to 11 , wherein the baseband signal is associated with a reduced power spreading ratio (PSR) due to one or more filter coefficient values applied at the pulse shaping filter, wherein the reduced PSR of the baseband signal causes a reduced peak-to-average power ratio (PAPR) of the baseband signal.

[0092] Example 13 includes the apparatus of any of Examples 9 to 12, wherein: the transmit node is a user equipment (UE) or an eNodeB; and the receive node is a user equipment (UE) or an eNodeB.

[0093] Example 14 includes a method for designing a pulse shaping filter for single carrier (SC) waveforms, the method comprising: configuring a pulse shaping filter to include an odd pulse shaping filter to output data with a first power metric and an even pulse shaping filter to output data with a second power metric, wherein the first power metric and the second power metric are derived independent of data inputted to the odd pulse shaping filter and the even pulse shaping filter, wherein the data is modulated on the SC waveform; selecting a weighting value to enable a substantially equal power contribution between the odd pulse shaping filter and the even pulse shaping filter; and selecting one or more filter coefficient values to apply to the odd pulse shaping filter and the even pulse shaping filter, wherein the one or more filter coefficient values are selected to reduce the first power metric and the second power metric, wherein a reduction in the first power metric and the second power metric causes the data to have a reduced peak-to- average power ratio (PAPR).

[0094] Example 15 includes the method of Example 14, further comprising designing the pulse shaping filters to minimize a weighted sum of the first power metric and the second power metric in accordance with the following pulse shaping optimization formula:

wherein h represents the pulse shaping filter, m_e represents an index of an even filter tap with a maximum power, m₀ represents an index of an odd filter tap with a maximum power, L is defined as N = 4L + 1 where N is a number of filter taps, h_2n represents a tap of an even filter, h_me represents a tap of an even filter with a maximum power, a represents a weighting factor, h_2n+1 represents a tap of an odd filter, and h_mo represents a tap of an odd filter with a maximum power.

[0095] Example 16 includes the method of any of Examples 14 to 15, wherein the pulse shaping optimization formula is subject to

|W_sD_sh| < 5_S and 1— δ_ρ < |W_pD_ph| < 1 + δ_ρ, wherein W_s and W_p are frequency weighting diagonal matrices, D_p represents a Discrete Fourier transform (DFT) basis in passbands, D_s represents a DFT basis in stop bands, h represents the pulse shaping filter, 5_S represents a stopband limit, and δ_ρ represents a passband limit.

[0096] Example 17 includes the method of any of Examples 14 to 16, further comprising designing the pulse shaping filters to minimize a weighted sum of the first power metric and the second power metric in accordance with the following pulse shaping sub- optimization formula: h^* = arg min \\h_even \\ + a ||h_odd ||,

h

wherein h^* represents the pulse shaping filter, _even represents taps of an even filter, a represents a weighting factor, and _odd represents taps of an odd filter.

[0097] Example 18 includes the method of any of Examples 14 to 17, wherein the pulse shaping sub-optimization formula is subject to the following constraints: m_e = [N/2 J = 21, m₀ = [TV/21 = 21 + 1, h_mo = 1, h_me = γ and |D_sh| < 8_S, wherein m_e represents an index of an even filter tap with a maximum power, m₀ represents an index of an odd filter tap with a maximum power, N represents a number of filter taps, L is defined as N = 4L + 1 where N is a number of filter taps, γ represents an optimization parameter, D_s represents a Discrete Fourier transform (DFT) basis in stop bands, h represents the pulse shaping filter, and 8_S represents a stopband limit.

[0098] Example 19 includes the method of any of Examples 14 to 18, wherein the pulse shaping filter is represented by h = [h₀h_t ... /ijv-i], the odd pulse shaping filter is represented by h₀ = [ ι^ ... h_N_₂], and the even pulse shaping filter is represented by

[0099] Example 20 includes the method of any of Examples 14 to 19, wherein the one or more filter coefficient values include a number of filter taps (N), a number of frequency response samples (N_FFT), a roll-off factor (/?), a passband frequency (f_p), a stopband frequency (f_s), the weighting value (a) that is between 0 and 1, frequency weighting diagonal matrices (W_s, W_p), and stopband and passband limits (5_S, δ_ρ). [00100] Example 21 includes the method of any of Examples 14 to 20, wherein the one or more filter coefficient values are selected to achieve a desired stopband and transition band in a frequency response produced by the pulse shaping filter.

[00101] Example 22 includes the method of any of Examples 14 to 21, wherein the first power metric and the second power metric include power spreading ratios (PSRs).

[00102] Example 23 includes the method of any of Examples 14 to 22, wherein the first power metric and the second power metric are determined based on a number of filter taps associated with the pulse shaping filter.

[00103] Example 24 includes the method of any of Examples 14 to 23, wherein a frequency response of the pulse shaping filter produces a reduced level of in-band fluctuations and out-of-band side lobes as compared to a root-raised cosine (RRC) pulse shaping filter.

[00104] Example 25 includes the method of any of Examples 14 to 24, wherein the pulse shaping filter is a linear phase structure.

[00105] Example 26 includes the method of any of Examples 14 to 25, wherein the SC waveforms are block-wise single carrier (BWSC) waveforms.

[00106] Various techniques, or certain aspects or portions thereof, may take the form of program code (i.e., instructions) embodied in tangible media, such as floppy diskettes, compact disc-read-only memory (CD-ROMs), hard drives, non-transitory computer readable storage medium, or any other machine-readable storage medium wherein, when the program code is loaded into and executed by a machine, such as a computer, the machine becomes an apparatus for practicing the various techniques. A non-transitory computer readable storage medium can be a computer readable storage medium that does not include signal. In the case of program code execution on programmable computers, the computing device may include a processor, a storage medium readable by the processor (including volatile and non-volatile memory and/or storage elements), at least one input device, and at least one output device. The volatile and non-volatile memory and/or storage elements may be a random-access memory (RAM), erasable

programmable read only memory (EPROM), flash drive, optical drive, magnetic hard drive, solid state drive, or other medium for storing electronic data. The node and wireless device may also include a transceiver module (i.e., transceiver), a counter module (i.e., counter), a processing module (i.e., processor), and/or a clock module (i.e., clock) or timer module (i.e., timer). One or more programs that may implement or utilize the various techniques described herein may use an application programming interface (API), reusable controls, and the like. Such programs may be implemented in a high level procedural or object oriented programming language to communicate with a computer system. However, the program(s) may be implemented in assembly or machine language, if desired. In any case, the language may be a compiled or interpreted language, and combined with hardware implementations.

[00107] As used herein, the term "circuitry" may refer to, be part of, or include an Application Specific Integrated Circuit (ASIC), an electronic circuit, a processor (shared, dedicated, or group), and/or memory (shared, dedicated, or group) that execute one or more software or firmware programs, a combinational logic circuit, and/or other suitable hardware components that provide the described functionality. In some embodiments, the circuitry may be implemented in, or functions associated with the circuitry may be implemented by, one or more software or firmware modules. In some embodiments, circuitry may include logic, at least partially operable in hardware.

[00108] It should be understood that many of the functional units described in this specification have been labeled as modules, in order to more particularly emphasize their implementation independence. For example, a module may be implemented as a hardware circuit comprising custom very-large-scale integration (VLSI) circuits or gate arrays, off-the-shelf semiconductors such as logic chips, transistors, or other discrete components. A module may also be implemented in programmable hardware devices such as field programmable gate arrays, programmable array logic, programmable logic devices or the like.

[00109] Modules may also be implemented in software for execution by various types of processors. An identified module of executable code may, for instance, comprise one or more physical or logical blocks of computer instructions, which may, for instance, be organized as an object, procedure, or function. Nevertheless, the executables of an identified module may not be physically located together, but may comprise disparate instructions stored in different locations which, when joined logically together, comprise the module and achieve the stated purpose for the module.

[00110] Indeed, a module of executable code may be a single instruction, or many instructions, and may even be distributed over several different code segments, among different programs, and across several memory devices. Similarly, operational data may be identified and illustrated herein within modules, and may be embodied in any suitable form and organized within any suitable type of data structure. The operational data may be collected as a single data set, or may be distributed over different locations including over different storage devices, and may exist, at least partially, merely as electronic signals on a system or network. The modules may be passive or active, including agents operable to perform desired functions.

[00111] Reference throughout this specification to "an example" or "exemplary" means that a particular feature, structure, or characteristic described in connection with the example is included in at least one embodiment of the present technology. Thus, appearances of the phrases "in an example" or the word "exemplary" in various places throughout this specification are not necessarily all referring to the same embodiment.

[00112] As used herein, a plurality of items, structural elements, compositional elements, and/or materials may be presented in a common list for convenience. However, these lists should be construed as though each member of the list is individually identified as a separate and unique member. Thus, no individual member of such list should be construed as a de facto equivalent of any other member of the same list solely based on their presentation in a common group without indications to the contrary. In addition, various embodiments and example of the present technology may be referred to herein along with alternatives for the various components thereof. It is understood that such embodiments, examples, and alternatives are not to be construed as defacto equivalents of one another, but are to be considered as separate and autonomous representations of the present technology.

[00113] Furthermore, the described features, structures, or characteristics may be combined in any suitable manner in one or more embodiments. In the following description, numerous specific details are provided, such as examples of layouts, distances, network examples, etc., to provide a thorough understanding of embodiments of the technology. One skilled in the relevant art will recognize, however, that the technology can be practiced without one or more of the specific details, or with other methods, components, layouts, etc. In other instances, well-known structures, materials, or operations are not shown or described in detail to avoid obscuring aspects of the technology.

[00114] While the forgoing examples are illustrative of the principles of the present technology in one or more particular applications, it will be apparent to those of ordinary skill in the art that numerous modifications in form, usage and details of implementation can be made without the exercise of inventive faculty, and without departing from the principles and concepts of the technology. Accordingly, it is not intended that the technology be limited, except as by the claims set forth below.

Claims

What is claimed is:

1. An apparatus of a transmit node operable to perform pulse shaping on a single carrier waveform, the apparatus comprising one or more processors and memory configured to:

identify a plurality of modulated symbols to be transmitted to a receive node;

distribute the plurality of modulated symbols to a pulse shaping filter, wherein even modulated symbols are distributed to an even pulse shaping filter and odd modulated symbols are distributed to an odd pulse shaping filter;

determine a first power metric for an output of the odd pulse shaping filter and a second power metric for an output of the even pulse shaping filter, wherein one or more filter coefficient values are selected to reduce the first power metric and the second power metric;

scale one of the output of the odd pulse shaping filter or the output of the even pulse shaping filter to provide a substantially equal power contribution of the odd pulse shaping filter and the even pulse shaping filter; and

merge the output of the odd pulse shaping filter and the output of the even pulse shaping filter to produce a baseband signal for transmission from the transmit node to the receive node, wherein a reduction in the first power metric and the second power metric causes the baseband signal to have a reduced peak- to-average power ratio (PAPR).

2. The apparatus of claim 1, further comprising a transceiver configured to transmit the baseband signal with reduced PAPR to the receive node.

3. The apparatus of claim 1, wherein the pulse shaping filter is represented by h =

[/IQ /I-L ... ijv-i], the odd pulse shaping filter is represented by h₀ = [h_th₃ ... h_N_₂], and the even pulse shaping filter is represented by h_e = [h₀h₂ ... /ijv-il

4. The apparatus of claim 1, wherein the one or more filter coefficient values include a number of filter taps (N), a number of frequency response samples (N_FFT), a roll-off factor (/?), a passband frequency (f_p), a stopband frequency (f_s), the weighting value (a) that is between 0 and 1 , frequency weighting diagonal matrices (W_s, W_p), and stopband and passband limits (5_S, δ_ρ).

The apparatus of claim 1, wherein the one or more filter coefficient values achieve a desired stopband and transition band in a frequency response produced by the pulse shaping filter.

The apparatus of claim 1, wherein the first power metric and the second power metric include power spreading ratios (PSRs).

The apparatus of claim 1, wherein the pulse shaping filter is utilized in a millimeter wave (mmwave) communication system.

The apparatus of claim 1, wherein:

the transmit node is a user equipment (UE) or an eNodeB; and

the receive node is a user equipment (UE) or an eNodeB.

An apparatus of a receive node operable to receive baseband signals on a single carrier waveform, the receive node comprising one or more processors and memory configured to:

identify a baseband signal received from a transmit node; and

filter the baseband signal using a match filter at the receive node, wherein the match filter is configured to match a pulse shaping filter at the transmit node, wherein the pulse shaping filter at the transmit node includes an odd pulse shaping filter and an even pulse shaping filter, wherein an output of the odd pulse shaping filter and an output of the even pulse shaping filter are merged to produce the baseband signal.

The apparatus of claim 9, further comprising a transceiver configured to receive the baseband signal from the transmit node. The apparatus of claim 9, wherein the pulse shaping filter is configured to distribute modulated symbols to an odd pulse shaping filter and an even pulse shaping filter.

The apparatus of claim 9, wherein the baseband signal is associated with a reduced power spreading ratio (PSR) due to one or more filter coefficient values applied at the pulse shaping filter, wherein the reduced PSR of the baseband signal causes a reduced peak-to-average power ratio (PAPR) of the baseband signal.

The apparatus of claim 9, wherein:

the transmit node is a user equipment (UE) or an eNodeB; and

the receive node is a user equipment (UE) or an eNodeB.

A method for designing a pulse shaping filter for single carrier (SC) waveforms, the method comprising:

configuring a pulse shaping filter to include an odd pulse shaping filter to output data with a first power metric and an even pulse shaping filter to output data with a second power metric, wherein the first power metric and the second power metric are derived independent of data inputted to the odd pulse shaping filter and the even pulse shaping filter, wherein the data is modulated on the SC waveform;

selecting a weighting value to enable a substantially equal power contribution between the odd pulse shaping filter and the even pulse shaping filter; and

selecting one or more filter coefficient values to apply to the odd pulse shaping filter and the even pulse shaping filter, wherein the one or more filter coefficient values are selected to reduce the first power metric and the second power metric, wherein a reduction in the first power metric and the second power metric causes the data to have a reduced peak-to-average power ratio (PAPR). The method of claim 14, further comprising designing the pulse shaping filters to minimize a weighted sum of the first power metric and the second power metric in accordance with the following pulse shaping optimization formula:

wherein h represents the pulse shaping filter, m_e represents an index of an even filter tap with a maximum power, m₀ represents an index of an odd filter tap with a maximum power, L is defined as N = 4L + 1 where N is a number of filter taps, h_2n represents a tap of an even filter, h_m represents a tap of an even filter with a maximum power, a represents a weighting factor, h_2n+1 represents a tap of an odd filter, and h_mo represents a tap of an odd filter with a maximum power.

The method of claim 15, wherein the pulse shaping optimization formula is subject to

|W_sD_sh | < S_s and 1 - δ_ρ≤ |W_p D_ph| < 1 + δ_ρ, wherein W_s and W_p are frequency weighting diagonal matrices, D_p represents a Discrete Fourier transform (DFT) basis in passbands, D_s represents a DFT basis in stop bands, h represents the pulse shaping filter, 5_S represents a stopband limit, and δ_ρ represents a passband limit.

The method of claim 14, further comprising designing the pulse shaping filters to minimize a weighted sum of the first power metric and the second power metric in accordance with the following pulse shaping sub-optimization formula:

h^* = arg min \\ _even \\ + a||h_odd ||,

h

The method of claim 17, wherein the pulse shaping sub-optimization formula is subject to the following constraints: m_e = [N/2\ = 2L, m₀ = \N/2] = 2L + 1, h_mo = 1, h_m<i = γ and |D_sh| < S_s, wherein m_e represents an index of an even filter tap with a maximum power, m₀ represents an index of an odd filter tap with a maximum power, N represents a number of filter taps, L is defined as N = 4L + 1 where N is a number of filter taps, γ represents an optimization parameter, D_s represents a Discrete Fourier transform (DFT) basis in stop bands, h represents the pulse shaping filter, and 5_S represents a stopband limit.

19. The method of claim 14, wherein the pulse shaping filter is represented by h =

[h-oh-L ... /i_jv-iL the odd pulse shaping filter is represented by h₀ = [h- h₃ ... h_N_₂], and the even pulse shaping filter is represented by h_e = [h₀h₂ ...

20. The method of claim 14, wherein the one or more filter coefficient values include a number of filter taps (N), a number of frequency response samples (N_FFT), a roll-off factor (/?), a passband frequency (f_p), a stopband frequency (f_s), the weighting value (a) that is between 0 and 1, frequency weighting diagonal matrices (W_s, W_p), and stopband and passband limits (5_S, δ_ρ).

21. The method of claim 14, wherein the one or more filter coefficient values are selected to achieve a desired stopband and transition band in a frequency response produced by the pulse shaping filter.

22. The method of claim 14, wherein the first power metric and the second power metric include power spreading ratios (PSRs).

23. The method of claim 14, wherein the first power metric and the second power metric are determined based on a number of filter taps associated with the pulse shaping filter. 24. The method of claim 14, wherein a frequency response of the pulse shaping filter produces a reduced level of in-band fluctuations and out-of-band side lobes as compared to a root-raised cosine (RRC) pulse shaping filter. The method of claim 14, wherein the pulse shaping filter is a linear phase structure.

The method of claim 14, wherein the SC waveforms are block- wise single (BWSC) waveforms.