WO2017219459A1 - 一种次谐波混频器及Ka波段高频头 - Google Patents

一种次谐波混频器及Ka波段高频头 Download PDF

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Publication number
WO2017219459A1
WO2017219459A1 PCT/CN2016/094864 CN2016094864W WO2017219459A1 WO 2017219459 A1 WO2017219459 A1 WO 2017219459A1 CN 2016094864 W CN2016094864 W CN 2016094864W WO 2017219459 A1 WO2017219459 A1 WO 2017219459A1
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frequency signal
module
filtering module
local oscillator
subharmonic
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PCT/CN2016/094864
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English (en)
French (fr)
Inventor
陈家诚
姚建可
丁庆
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深圳市华讯方舟卫星通信有限公司
华讯方舟科技有限公司
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Publication of WO2017219459A1 publication Critical patent/WO2017219459A1/zh

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    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03D—DEMODULATION OR TRANSFERENCE OF MODULATION FROM ONE CARRIER TO ANOTHER
    • H03D7/00—Transference of modulation from one carrier to another, e.g. frequency-changing
    • H03D7/16—Multiple-frequency-changing
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04N—PICTORIAL COMMUNICATION, e.g. TELEVISION
    • H04N5/00—Details of television systems
    • H04N5/44—Receiver circuitry for the reception of television signals according to analogue transmission standards
    • H04N5/50—Tuning indicators; Automatic tuning control
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04N—PICTORIAL COMMUNICATION, e.g. TELEVISION
    • H04N7/00—Television systems
    • H04N7/20—Adaptations for transmission via a GHz frequency band, e.g. via satellite

Definitions

  • the present invention belongs to the field of Ka-band communication, and particularly relates to a subharmonic mixer and a Ka-band tuner.
  • Ka-band millimeter wave is widely used in the field of satellite communication.
  • the Ka-band tuner is an important component of the millimeter-wave transceiver, and is a key component for satellite communication, and the subharmonic in the Ka-wave tuner.
  • Wave mixers are one of the important devices that affect the performance of the transmit link.
  • GaA s gallium arsenide
  • GaAs gallium arsenide
  • the present invention is implemented as follows, a subharmonic mixer including a local oscillator signal filtering module, a high frequency signal filtering module, a subharmonic mixing module, and an intermediate frequency signal filtering module, the subharmonic mixing
  • the frequency module includes an anti-parallel Schottky diode pair, the Schottky diode pair includes a diode D1 and a diode D2, wherein a positive pole of the diode D1 and a negative pole of the diode D2 are commonly connected to form the subharmonic mixing module.
  • the input terminal, the cathode of the diode D1 and the anode of the diode D2 are connected to form an input and output mountain of the subharmonic mixing module.
  • the input end of the local oscillator signal filtering module is connected to the local oscillator signal
  • the input end of the high frequency signal filtering module is connected to the high frequency signal
  • the output end of the high frequency signal filtering module is connected to the secondary harmonic
  • An input end of the wave mixing module, an input and output end of the subharmonic mixing module and an output end of the local oscillator signal filtering module The input end of the intermediate frequency signal filtering module is connected in common, and the output end of the intermediate frequency signal filtering module outputs an intermediate frequency signal;
  • the local oscillator signal filtering module filters the accessed local oscillator signal and outputs the signal to the subharmonic mixing module
  • the high frequency signal filtering module filters the accessed high frequency signal and outputs the The subharmonic mixing module, the subharmonic mixing module mixes the filtered local oscillator signal and the filtered high frequency signal to obtain an intermediate frequency signal, and outputs the intermediate frequency signal to the intermediate frequency signal filtering module, the intermediate frequency
  • the signal filtering module filters the intermediate frequency signal and outputs the signal.
  • the local oscillator signal filter module is a parallel coupled microstrip line bandpass filter.
  • the microstrip line structure of the local oscillator signal filter module is an M-order parallel coupled microstrip structure, and the M-order parallel coupled microstrip structure includes M arranged in parallel along the length direction of each microstrip line. +1 microstrip line
  • the M+1 microstrip lines are equally spaced or unequally spaced, and when the M+1 microstrip lines are unequally spaced, the spacing between adjacent microstrip lines is in the middle two The spacing between the strip lines is decremented toward the two sides as a reference; [0011] the widths of the M+1 microstrip lines are equal and the lengths of the microstrip lines are in the order of the distribution of the microstrip lines with the middle m microstrip lines The length is decremented toward both sides. When m>l ⁇ , the lengths of the intermediate m microstrip lines are equal, wherein M and m are positive integers greater than zero.
  • the local oscillator signal connected by the local oscillator signal filtering module has a frequency range of 9.03 GHz to 9.22 G.
  • the center frequency of the local oscillator signal outputted by the two-stage parallel coupled microstrip structure is ⁇ 9.125G
  • the subharmonic mixer further includes a low noise amplifier connected to an input end of the local oscillator signal filtering module, and the local oscillator signal is amplified by the low noise amplifier and then output to the local The vibration signal filtering module performs filtering.
  • the high frequency signal filtering module is a parallel coupled microstrip line band pass filter.
  • the microstrip line structure of the high frequency signal filtering module is an N-order parallel coupled microstrip structure
  • the N-th parallel coupled microstrip structure includes N+1 microstrip lines arranged in parallel along the length direction of each microstrip line; [0016] the N+1 microstrip lines are equally or unevenly spaced, when The N+1 microstrip lines are unequally spaced apart, and the spacing between adjacent microstrip lines decreases toward both sides with reference to the spacing between the two microstrip lines; [0017] the N+1 micro The width of the strip lines is equal and the length of each microstrip line is in the middle in the order of distribution of the microstrip lines The lengths of the n microstrip lines are decremented toward the two sides. When n>l ⁇ , the lengths of the middle n microstrip lines are equal, wherein N and n are positive integers greater than 0.
  • the subharmonic mixing module further includes a current limiting resistor R1 and a current limiting resistor R2, wherein the current limiting resistor R1 is connected in series with the anode of the diode D1 and the input and output of the subharmonic mixing module. Between the terminals, the current limiting resistor R2 is connected in series between the anode of the diode D2 and the input and output terminals of the subharmonic mixing module.
  • the intermediate frequency signal filtering module is a parallel coupled microstrip line band pass filter or a filter capacitor.
  • Embodiments of the present invention also provide a Ka-band tuner including the subharmonic mixer as described above.
  • the present invention has the beneficial effects of:
  • the high frequency signal and the local oscillator signal are mixed to obtain an intermediate frequency signal, so as to achieve the second harmonic of the subharmonic mixer.
  • the wave mixing function can greatly reduce the manufacturing cost of the subharmonic mixer while achieving better mixing performance.
  • FIG. 1 is a block diagram showing the basic structure of a subharmonic mixer according to Embodiment 1 of the present invention
  • FIG. 2 is a schematic diagram of a two-stage parallel coupled microstrip structure in a local oscillator signal filtering module according to Embodiment 2 of the present invention
  • FIG. 3 is a schematic diagram of a fifth-order parallel coupled microstrip structure in a high-frequency signal filtering module according to Embodiment 2 of the present invention.
  • FIG. 4 is a block diagram showing a specific structure of a subharmonic mixing module according to Embodiment 3 of the present invention.
  • the subharmonic mixer includes a local oscillator signal filtering module 10, a high frequency signal filtering module 20, a subharmonic mixing module 30, and an intermediate frequency signal filtering module 40.
  • the subharmonic mixing module 30 includes an anti-parallel Schottky diode pair, the Schottky diode pair includes a diode D1 and a diode D2, wherein the anode of the diode D1 and the cathode of the diode D2 are connected in common.
  • the input terminal of the harmonic mixing module 30, the cathode of the diode D1 and the anode of the diode D2 are connected in common to form an input and output end of the subharmonic mixing module 30.
  • the input end of the local signal filtering module 10 is connected to the local oscillator signal
  • the input end of the high frequency signal filtering module 20 is connected to the high frequency signal
  • the output end of the high frequency signal filtering module 20 is connected to the subharmonic mixing module 30.
  • the input end of the subharmonic mixing module 30 is connected to the output end of the local oscillator signal filtering module 10 and the input end of the intermediate frequency signal filtering module 40, and the output end of the intermediate frequency signal filtering module 40 outputs an intermediate frequency signal.
  • the local oscillator signal filtering module 10 filters the accessed local oscillator signal and outputs the signal to the subharmonic mixing module 30.
  • the high frequency signal filtering module 20 filters the accessed high frequency signal and outputs the same to the subharmonic.
  • the mixing module 30, the subharmonic mixing module 30 mixes the filtered local oscillator signal and the filtered high frequency signal to obtain an intermediate frequency signal and outputs the signal to the intermediate frequency signal filtering module 40, and the intermediate frequency signal filtering module 40 pairs the intermediate frequency After filtering the signal, the filtered intermediate frequency signal is obtained and output.
  • This embodiment is a further refinement of the internal device structure of the subharmonic filter in Embodiment 1.
  • the local oscillator signal filter module 10 and the high frequency signal filtering module 20 each employ a parallel coupled microstrip line band pass filter.
  • the microstrip line structure of the local oscillator signal filter module 10 is an M-order parallel coupled microstrip structure, and the M-th order parallel coupled microstrip structure includes M+1 micro-slices arranged in parallel along the length direction of each microstrip line.
  • the M+1 microstrip lines are equally or unevenly spaced, and when the M+1 microstrip lines are unequally distributed, the spacing between adjacent microstrip lines is in the middle two The spacing between the strip lines is decremented to the sides as a reference; [0039] The widths of the M+l microstrip lines are equal and the lengths of the microstrip lines are decreased toward both sides according to the distribution order of the microstrip lines, when m>l ⁇ The lengths of the middle m microstrip lines are equal, wherein both M and m are positive integers greater than zero.
  • the microstrip line structure of the high frequency signal filtering module 20 is an N-order parallel coupled microstrip structure, and the N-th parallel coupled microstrip structure includes N+1 microstrips arranged parallel to each other along the length direction of each microstrip line.
  • the N+1 microstrip lines are equally spaced or unequally spaced, and when the N+1 microstrip lines are unequally spaced, the spacing between adjacent microstrip lines is in the middle two The spacing between the strip lines is decremented toward the two sides as a reference; [0042] the widths of the N+1 microstrip lines are equal and the lengths of the microstrip lines are in the order of the microstrip lines in the middle of the n microstrip lines The length is decremented toward both sides. When n>l ⁇ , the lengths of the middle n microstrip lines are equal, wherein N and n are positive integers greater than 0.
  • the microstrip line structure of the local oscillator signal filter module 10 is a two-order parallel coupled microstrip structure, and the two-order parallel coupled microstrip structure includes mutually along the length of each microstrip line.
  • the lengths of the three microstrip lines 11 to 13 are 117.5 mil, 235 mil, and 117.5 mil in sequence (the length of the microstrip line 11 is indicated by only the dimension line in the figure), and the width D of the three microstrip lines 11 to 13 are both 16 mil (only the dimension line indicates the width of the microstrip line 12), the spacing between two adjacent microstrip lines in the three microstrip lines 11 ⁇ 13 is 5 mil (only the dimension line is used in the figure) The spacing between the microstrip lines 12 and 13 is indicated).
  • the local oscillator signal connected to the local oscillator signal filtering module 10 has a frequency range of 9.03 GHz to 9.
  • the center frequency of the local oscillator signal outputted by the two-stage parallel coupled microstrip structure is ⁇ 9.1 25 GHz, and the amplitude of the output local oscillator signal is less than -50 dB at ⁇ 9.2 GHz.
  • the size of the microstrip line of the local oscillator signal filter module 10 can be optimally fine-tuned according to design requirements on the basis of the above dimensions, and is not limited to the above dimensions.
  • the microstrip line structure of the high frequency signal filtering module 20 is a fifth-order parallel coupled microstrip structure, and the fifth-order parallel coupled microstrip structure includes a length along each microstrip line.
  • the lengths of the six microstrip lines 21 to 26 are 105 mil, 215 mil, 215 mil, 215 mil, 215 mil, and 105 mil (the length of the microstrip line 21 is indicated by a dimension line only), and the six microstrip lines are The width is 15 mil (only the dimension line indicates the width of the microstrip line 22 in the figure), and the six microstrip lines are in the middle
  • the spacing between the adjacent two microstrip lines is 12 mils, 27 mils, 30 mils, 27 mils, and 12 mils in sequence (only the dimension lines indicate the spacing between the microstrip lines 23 and 24).
  • the high frequency signal filtering module 20 has better filtering characteristics in the frequency range of the accessed high frequency signal in the range of 19.2 GHz to 20.2 GHz, and the high frequency signal passes through the five
  • the 3dB bandwidth of the parallel parallel coupled microstrip structure is ⁇ 2.87GHz, and the input reflection coefficient is less than -20dB.
  • the input power is -35 dBm
  • the frequency of the input local oscillator signal is 9.125 GHz.
  • the input power is 5dBm ⁇
  • the frequency of the local harmonic signal generated after mixing is ⁇ 18.25GHz
  • the frequency of the intermediate frequency signal of the final mixed output is 1.95GHz (the calculation formula of the intermediate frequency signal frequency is: high frequency signal)
  • the parallel coupled microstrip line bandpass filter used by the local oscillator signal filter module 10 and the high frequency signal filtering module 20 can be formed on the PCB by copper plating or copper plating and sinking silver.
  • the thickness is 17 ⁇ 34 ⁇ , which can pass the local oscillator signal in the selected frequency range and the required Ka-band high-frequency signal.
  • This embodiment is a further refinement of the subharmonic filter provided in the first embodiment or the second embodiment.
  • the subharmonic mixer further includes a low noise amplifier 50 connected to the input end of the local oscillator signal filtering module 10, and the local oscillator signal is amplified by the low noise amplifier 50. Then, it is output to the local oscillator signal filtering module 10 for filtering.
  • the subharmonic mixing module 30 further includes a current limiting resistor R1 and a current limiting resistor R2, wherein the current limiting resistor R1 is connected in series to the input and output of the negative and subharmonic mixing module 30 of the diode D1. Between the terminals, the current limiting resistor R2 is connected in series between the anode of the diode D2 and the input and output terminals of the subharmonic mixing module 30.
  • the current limiting resistor R1 and the current limiting resistor R2 are used to stabilize the current of the Schottky diode after the temperature changes to avoid the influence of temperature variation on the current drift caused by the Schottky diode.
  • the current limiting resistor R1 and the current limiting resistor R2 can be selected from a variety of types such as thermistors, adjustable resistors, and fixed-value resistors.
  • the intermediate frequency signal filtering module 40 is a parallel coupled microstrip line bandpass filter.
  • the intermediate frequency signal filtering module 40 may also be replaced by a filter capacitor.
  • the present invention further provides a Ka-band tuner applied to a transceiver in the field of Ka-band satellite communication, wherein the Ka-band tuner includes the subharmonic mixing as described above. Device.

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Abstract

一种次谐波混频器及Ka波段高频头，所述次谐波混频器，包括本振信号滤波模块（10）、高频信号滤波模块（20）、次谐波混频模块（30）及中频信号滤波模块（40），次谐波混频模块（30）包括反向并联的肖特基二极管对；次谐波混频模块（30）对本振信号滤波模块（10）滤波后的本振信号和高频信号滤波模块（20）滤波后的高频信号进行混频得到中频信号，中频信号滤波模块（40）对中频信号进行滤波后输出。本申请通过利用反向并联的肖特基二极管的偶次谐波的混频特性，对高频信号和本振信号进行混频得到中频信号，以实现次谐波混频器的二次谐波混频功能，可在获得较好混频功能的同时大大降低次谐波混频器的生产制造成本。

Description

一种次谐波混频器及 Ka波段高频头 技术领域
[0001] 本发明属于 Ka波段通信领域， 尤其涉及一种次谐波混频器及 Ka波段高频头。
背景技术
[0002] Ka波段毫米波在卫星通信领域的应用越来越广泛， Ka波段高频头作为毫米波 收发机的重要部件， 是实现卫星通信的关键器件， 而 Ka波高频头中的次谐波混 频器是影响发射链路性能的重要器件之一。
[0003] 然而， 现有的能够处理 Ka波段信号的次谐波混频器芯片通常采用砷化镓 （GaA s) 工艺制成， 此类虽然能够简化设计结构、 同吋增强电路的可靠性， 但是价格 成本较高， 不适于广泛推广使用。
技术问题
[0004] 本发明的目的在于提供一种次谐波混频器及 Ka波段高频头， 旨在解决现有的能 够处理 Ka波段信号的次谐波混频器芯片通常采用砷化镓 （GaAs) 工艺制成， 此 类虽然能够简化设计结构、 同吋增强电路的可靠性， 但是价格成本较高， 不适 于广泛推广使用的问题。
问题的解决方案
技术解决方案
[0005] 本发明是这样实现的， 一种次谐波混频器， 包括本振信号滤波模块、 高频信号 滤波模块、 次谐波混频模块及中频信号滤波模块， 所述次谐波混频模块包括反 向并联的肖特基二极管对， 所述肖特基二极管对包括二极管 D1和二极管 D2， 其 中， 二极管 D1的正极和二极管 D2的负极共接构成所述次谐波混频模块的输入端 ， 二极管 D1的负极和二极管 D2的正极共接构成所述次谐波混频模块的输入输出 山
[0006] 所述本振信号滤波模块的输入端接入本振信号， 所述高频信号滤波模块的输入 端接入高频信号、 所述高频信号滤波模块的输出端接所述次谐波混频模块的输 入端， 所述次谐波混频模块的输入输出端与所述本振信号滤波模块的输出端和 所述中频信号滤波模块的输入端共接， 所述中频信号滤波模块的输出端输出中 频信号；
[0007] 所述本振信号滤波模块对接入的本振信号进行滤波后输出给所述次谐波混频模 块， 所述高频信号滤波模块对接入的高频信号进行滤波后输出给所述次谐波混 频模块， 所述次谐波混频模块对滤波后的本振信号和滤波后的高频信号进行混 频得到中频信号并输出给所述中频信号滤波模块， 所述中频信号滤波模块对所 述中频信号进行滤波后输出。
[0008] 优选的， 所述本振信号滤波器模块为平行耦合微带线带通滤波器。
[0009] 优选的， 所述本振信号滤波器模块的微带线结构为 M阶平行耦合微带结构， 所 述 M阶平行耦合微带结构包括沿各微带线长度方向相互平行设置的 M+1条微带线
[0010] 所述 M+1条微带线等间距或不等间距分布， 当所述 M+1条微带线不等间距分布 吋， 相邻微带线之间的间距以中间两条微带线之间的间距为基准向两侧递减； [0011] 所述 M+1条微带线的宽度相等且各微带线的长度按照微带线的分布顺序以中间 m条微带线的长度为基准向两侧递减， 当 m〉l吋， 所述中间 m条微带线的长度相 等， 其中， M和 m均为大于 0的正整数。
[0012] 优选的， 所述本振信号滤波模块接入的本振信号的频率范围为 9.03GHz~9.22G
Hz, 经所述两阶平行耦合微带结构滤波后输出的本振信号的中心频率为 ~9.125G
Hz。
[0013] 优选的， 所述次谐波混频器还包括与所述本振信号滤波模块的输入端连接的低 噪声放大器， 本振信号经由所述低噪声放大器放大之后再输出给所述本振信号 滤波模块进行滤波。
[0014] 优选的， 所述高频信号滤波模块为平行耦合微带线带通滤波器。
[0015] 优选的， 所述高频信号滤波模块的微带线结构为 N阶平行耦合微带结构， 所述
N阶平行耦合微带结构包括沿各微带线长度方向相互平行设置的 N+1条微带线； [0016] 所述 N+1条微带线等间距或不等间距分布， 当所述 N+1条微带线不等间距分布 吋， 相邻微带线之间的间距以中间两条微带线之间的间距为基准向两侧递减； [0017] 所述 N+1条微带线的宽度相等且各微带线的长度按照微带线的分布顺序以中间 n条微带线的长度为基准向两侧递减， 当 n〉l吋， 所述中间 n条微带线的长度相 等， 其中， N和 n均为大于 0的正整数。
[0018] 优选的， 所述次谐波混频模块还包括限流电阻 R1和限流电阻 R2， 其中， 限流 电阻 R1串联在二极管 D1的负极和所述次谐波混频模块的输入输出端之间， 限流 电阻 R2串联在二极管 D2的正极和所述次谐波混频模块的输入输出端之间。
[0019] 优选的， 所述中频信号滤波模块为平行耦合微带线带通滤波器或滤波电容。
[0020] 本发明实施例还提供一种 Ka波段高频头， 所述 Ka波段高频头包括如前所述次 谐波混频器。
发明的有益效果
有益效果
[0021] 本发明与现有技术相比， 其有益效果在于：
[0022] 通过利用反向并联的肖特基二极管的偶次谐波的混频特性， 对高频信号和本振 信号进行混频得到中频信号， 以实现次谐波混频器的二次谐波混频功能， 可在 获得较好混频功能的同吋大大降低次谐波混频器的生产制造成本。
对附图的简要说明
附图说明
[0023] 图 1是本发明实施例一提供的次谐波混频器的基本结构框图；
[0024] 图 2是本发明实施例二提供的本振信号滤波模块中两阶平行耦合微带结构的示 意图；
[0025] 图 3是本发明实施例二提供的高频信号滤波模块中五阶平行耦合微带结构的示 意图；
[0026] 图 4是本发明实施例三提供的次谐波混频模块的具体结构框图。
本发明的实施方式
[0027] 为了使本发明的目的、 技术方案及优点更加清楚明白， 以下结合附图及实施例 ， 对本发明进行进一步详细说明。 应当理解， 此处所描述的具体实施例仅用以 解释本发明， 并不用于限定本发明。 [0028] 实施例一
[0029] 如图 1所示， 在本实施例中， 次谐波混频器包括本振信号滤波模块 10、 高频信 号滤波模块 20、 次谐波混频模块 30及中频信号滤波模块 40。
[0030] 次谐波混频模块 30包括反向并联的肖特基二极管对， 所述肖特基二极管对包括 二极管 D1和二极管 D2， 其中， 二极管 D1的正极和二极管 D2的负极共接构成次谐 波混频模块 30的输入端， 二极管 D1的负极和二极管 D2的正极共接构成次谐波混 频模块 30的输入输出端。
[0031] 本振信号滤波模块 10的输入端接入本振信号， 高频信号滤波模块 20的输入端接 入高频信号、 高频信号滤波模块 20的输出端接次谐波混频模块 30的输入端， 次 谐波混频模块 30的输入输出端与本振信号滤波模块 10的输出端和中频信号滤波 模块 40的输入端共接， 中频信号滤波模块 40的输出端输出中频信号。
[0032] 本振信号滤波模块 10对接入的本振信号进行滤波后输出给次谐波混频模块 30， 高频信号滤波模块 20对接入的高频信号进行滤波后输出给次谐波混频模块 30， 次谐波混频模块 30对滤波后的本振信号和滤波后的高频信号进行混频得到中频 信号并输出给中频信号滤波模块 40， 中频信号滤波模块 40对所述中频信号进行 滤波后得到滤波后的中频信号并输出。
[0033] 本实施例通过采用反向并联的肖特基二极管作为次谐波混频模块， 利用反向并 联的肖特基二极管的偶次谐波的混频特性， 对高频信号和本振信号进行混频得 到中频信号， 以实现次谐波混频器的二次谐波混频功能， 可在获得较好混频功 能的同吋大大降低次谐波混频器的生产制造成本。
[0034] 实施例二
[0035] 本实施例是对实施一中的次谐波滤波器内部器件结构的进一步细化。
[0036] 在本实施例中， 本振信号滤波器模块 10和高频信号滤波模块 20均采用平行耦合 微带线带通滤波器。
[0037] 本振信号滤波器模块 10的微带线结构为 M阶平行耦合微带结构， 所述 M阶平行 耦合微带结构包括沿各微带线长度方向相互平行设置的 M+1条微带线；
[0038] 所述 M+1条微带线等间距或不等间距分布， 当所述 M+1条微带线不等间距分布 吋， 相邻微带线之间的间距以中间两条微带线之间的间距为基准向两侧递减； [0039] 所述 M+l条微带线的宽度相等且各微带线的长度按照微带线的分布顺序以中间 m条微带线的长度为基准向两侧递减， 当 m〉l吋， 所述中间 m条微带线的长度相 等， 其中， M和 m均为大于 0的正整数。
[0040] 高频信号滤波模块 20的微带线结构为 N阶平行耦合微带结构， 所述 N阶平行耦 合微带结构包括沿各微带线长度方向相互平行设置的 N+1条微带线；
[0041] 所述 N+1条微带线等间距或不等间距分布， 当所述 N+1条微带线不等间距分布 吋， 相邻微带线之间的间距以中间两条微带线之间的间距为基准向两侧递减； [0042] 所述 N+1条微带线的宽度相等且各微带线的长度按照微带线的分布顺序以中间 n条微带线的长度为基准向两侧递减， 当 n〉l吋， 所述中间 n条微带线的长度相 等， 其中， N和 n均为大于 0的正整数。
[0043] 如图 2所示， 本实施中优选本振信号滤波器模块 10的微带线结构为两阶平行耦 合微带结构， 两阶平行耦合微带结构包括沿各微带线长度方向相互平行设置的 三条微带线 11 13 ;
[0044] 三条微带线 11~13的长度依次为 117.5mil、 235mil、 117.5mil (图中仅用尺寸标 注线示意出微带线 11的长度） ， 三条微带线 11~13的宽度 D均为 16mil (图中仅用 尺寸标注线示意出微带线 12的宽度） ， 三条微带线 11~13中相邻两条微带线之间 的间距均为 5mil (图中仅用尺寸标注线示意出微带线 12和 13之间的间距） 。
[0045] 在本实施例中， 本振信号滤波模块 10接入的本振信号的频率范围为 9.03GHz~9.
22GHz, 经所述两阶平行耦合微带结构滤波后输出的本振信号的中心频率为 ~9.1 25GHz, 输出的本振信号的频率在〜 9.2GHz处的反射系数小于 -50dB。
[0046] 在具体应用中， 本振信号滤波器模块 10的微带线的尺寸可在上述尺寸的基础之 上根据设计要求进行优化微调， 并不局限于上述尺寸。
[0047] 如图 3所示， 本实施中优选高频信号滤波模块 20的微带线结构为五阶平行耦合 微带结构， 所述五阶平行耦合微带结构包括沿各微带线长度方向相互平行设置 的六条微带线 21~26；
[0048] 六条微带线 21~26的长度依次为 105mil、 215mil、 215mil、 215mil、 215mil、 105 mil (图中仅用尺寸标注线示意出微带线 21的长度） ， 所述六条微带线的宽度均 为 15mil (图中仅用尺寸标注线示意出微带线 22的宽度） ， 所述六条微带线中相 邻两条微带线之间的间距依次为 12mil、 27mil、 30mil、 27mil、 12mil (图中仅用 尺寸标注线示意出微带线 23和 24之间的间距） 。
[0049] 在本实施例中， 高频信号滤波模块 20在接入的高频信号的频率范围在 19.2GHz~ 20.2GHz范围内吋具有较好的滤波特性， 所述高频信号经所述五阶平行耦合微带 结构滤波后的 3dB带宽为〜 2.87GHz, 输入反射系数小于 -20dB。
[0050] 采用本实施例所述提供的次谐波滤波器， 当肖特基二极管对输入的高频信号的 频率为 20.2GHz、 输入功率为 -35dBm， 输入的本振信号的频率为 9.125GHz、 输 入功率为 5dBm吋， 经过混频后产生的本振次谐波信号的频率为〜 18.25GHz, 最 后混频输出的中频信号的频率为 1.95GHz (中频信号频率的计算公式为： 高频信 号频率—2倍本振信号频率 =20.2GHz -9.125GHz*2=1.95GHz) 。
[0051] 在具体应用中， 本振信号滤波器模块 10和高频信号滤波模块 20所采用的平行耦 合微带线带通滤波器均可通过镀铜或者镀铜加沉银工艺， 形成在 PCB上， 厚度为 17μηι~34μιη, 可以通过选定频率范围内的本振信号和所需的 Ka波段高频信号。
[0052] 实施例三
[0053] 本实施是对实施例一或实施例二所提供的次谐波滤波器的进一步细化。
[0054] 如图 4所示， 在本实施例中， 次谐波混频器还包括与本振信号滤波模块 10的输 入端连接的低噪声放大器 50， 本振信号经由低噪声放大器 50放大之后再输出给 本振信号滤波模块 10进行滤波。
[0055] 本实施例中， 次谐波混频模块 30还包括限流电阻 R1和限流电阻 R2， 其中， 限 流电阻 R1串联在二极管 D1的负极和次谐波混频模块 30的输入输出端之间， 限流 电阻 R2串联在二极管 D2的正极和次谐波混频模块 30的输入输出端之间。
[0056] 在具体应用中， 限流电阻 R1和限流电阻 R2用于在温度变化吋， 起到稳定肖特 基二极管的电流的作用， 以避免温度变化对肖特基二极管造成的电流漂移影响
； 限流电阻 R1和限流电阻 R2可以选用热敏电阻、 可调电阻、 定值电阻等多种类 型的电阻。
[0057] 本实施例中， 中频信号滤波模块 40为平行耦合微带线带通滤波器。
[0058] 在具体应用中， 为了减小次谐波混频器的面积， 中频信号滤波模块 40也可以选 用滤波电容代替。 [0059] 在一实施例中， 本发明还提供一种应用于 Ka波段卫星通信领域的收发机上的 K a波段高频头， 所述 Ka波段高频头包括如前所述次谐波混频器。
[0060] 以上所述仅为本发明的较佳实施例而已， 并不用以限制本发明， 凡在本发明的 精神和原则之内所作的任何修改、 等同替换和改进等， 均应包含在本发明的保 护范围之内。

Claims

权利要求书
[权利要求 1] 一种次谐波混频器， 包括本振信号滤波模块、 高频信号滤波模块、 次 谐波混频模块及中频信号滤波模块， 其特征在于， 所述次谐波混频模 块包括反向并联的肖特基二极管对， 所述肖特基二极管对包括二极管
D1和二极管 D2， 其中， 二极管 D1的正极和二极管 D2的负极共接构成 所述次谐波混频模块的输入端， 二极管 D1的负极和二极管 D2的正极 共接构成所述次谐波混频模块的输入输出端；
所述本振信号滤波模块的输入端接入本振信号， 所述高频信号滤波模 块的输入端接入高频信号、 所述高频信号滤波模块的输出端接所述次 谐波混频模块的输入端， 所述次谐波混频模块的输入输出端与所述本 振信号滤波模块的输出端和所述中频信号滤波模块的输入端共接， 所 述中频信号滤波模块的输出端输出中频信号；
所述本振信号滤波模块对接入的本振信号进行滤波后输出给所述次谐 波混频模块， 所述高频信号滤波模块对接入的高频信号进行滤波后输 出给所述次谐波混频模块， 所述次谐波混频模块对滤波后的本振信号 和滤波后的高频信号进行混频得到中频信号并输出给所述中频信号滤 波模块， 所述中频信号滤波模块对所述中频信号进行滤波后输出。
[权利要求 2] 如权利要求 1所述的次谐波混频器， 其特征在于， 所述本振信号滤波 器模块为平行耦合微带线带通滤波器。
[权利要求 3] 如权利要求 2所述的次谐波混频器， 其特征在于， 所述本振信号滤波 器模块的微带线结构为 M阶平行耦合微带结构， 所述 M阶平行耦合微 带结构包括沿各微带线长度方向相互平行设置的 M+1条微带线； 所述 M+1条微带线等间距或不等间距分布， 当所述 M+1条微带线不等 间距分布吋， 相邻微带线之间的间距以中间两条微带线之间的间距为 基准向两侧递减；
所述 M+1条微带线的宽度相等且各微带线的长度按照微带线的分布顺 序以中间 m条微带线的长度为基准向两侧递减， 当 m〉l吋， 所述中间 m条微带线的长度相等， 其中， M和 m均为大于 0的正整数。 如权利要求 3所述的次谐波混频器， 其特征在于， 所述本振信号滤波 模块接入的本振信号的频率范围为 9.03GHz~9.22GHz， 经所述两阶平 行耦合微带结构滤波后输出的本振信号的中心频率为 ~9.125GHz。 如权利要求 1~4任一项所述次谐波混频器， 其特征在于， 所述次谐波 混频器还包括与所述本振信号滤波模块的输入端连接的低噪声放大器 ， 本振信号经由所述低噪声放大器放大之后再输出给所述本振信号滤 波模块进行滤波。
如权利要求 1所述的次谐波混频器， 其特征在于， 所述高频信号滤波 模块为平行耦合微带线带通滤波器。
如权利要求 6所述的次谐波混频器， 其特征在于， 所述高频信号滤波 模块的微带线结构为 N阶平行耦合微带结构， 所述 N阶平行耦合微带 结构包括沿各微带线长度方向相互平行设置的 N+1条微带线； 所述 N+1条微带线等间距或不等间距分布， 当所述 N+1条微带线不等 间距分布吋， 相邻微带线之间的间距以中间两条微带线之间的间距为 基准向两侧递减；
所述 N+1条微带线的宽度相等且各微带线的长度按照微带线的分布顺 序以中间 n条微带线的长度为基准向两侧递减， 当 n〉l吋， 所述中间 n 条微带线的长度相等， 其中， N和 n均为大于 0的正整数。
如权利要求 1所述次谐波混频器， 其特征在于， 所述次谐波混频模块 还包括限流电阻 R1和限流电阻 R2， 其中， 限流电阻 R1串联在二极管 D1的负极和所述次谐波混频模块的输入输出端之间， 限流电阻 R2串 联在二极管 D2的正极和所述次谐波混频模块的输入输出端之间。 如权利要求 1所述次谐波混频器， 其特征在于， 所述中频信号滤波模 块为平行耦合微带线带通滤波器或滤波电容。
一种 Ka波段高频头， 其特征在于， 所述 Ka波段高频头包括如权利要 求 1~9任一项所述次谐波混频器。
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Citations (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN103633943A (zh) * 2013-12-09 2014-03-12 中国电子科技集团公司第四十一研究所 一种超宽带混频器
CN104753468A (zh) * 2015-04-18 2015-07-01 中国电子科技集团公司第四十一研究所 一种毫米波偶次谐波混频器结构
CN105048967A (zh) * 2015-08-20 2015-11-11 电子科技大学 一种340GHz八次谐波混频器
CN205912021U (zh) * 2016-06-23 2017-01-25 深圳市华讯方舟卫星通信有限公司 一种次谐波混频器及Ka波段高频头

Family Cites Families (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US4749949A (en) * 1986-04-29 1988-06-07 Hewlett-Packard Company Self biasing diode microwave frequency multiplier
JP4202123B2 (ja) * 2000-07-10 2008-12-24 三菱電機株式会社 偶高調波ミクサ
CN1234205C (zh) * 2002-03-11 2005-12-28 香港城市大学 微波及毫米波四次谐波混频器
CN103209147B (zh) * 2013-03-25 2015-10-21 电子科技大学 一种多频段毫米波接收机及方法

Patent Citations (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN103633943A (zh) * 2013-12-09 2014-03-12 中国电子科技集团公司第四十一研究所 一种超宽带混频器
CN104753468A (zh) * 2015-04-18 2015-07-01 中国电子科技集团公司第四十一研究所 一种毫米波偶次谐波混频器结构
CN105048967A (zh) * 2015-08-20 2015-11-11 电子科技大学 一种340GHz八次谐波混频器
CN205912021U (zh) * 2016-06-23 2017-01-25 深圳市华讯方舟卫星通信有限公司 一种次谐波混频器及Ka波段高频头

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