WO2015180675A1 - 发光器件的恒流驱动电路及驱动方法 - Google Patents

发光器件的恒流驱动电路及驱动方法 Download PDF

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Publication number
WO2015180675A1
WO2015180675A1 PCT/CN2015/080148 CN2015080148W WO2015180675A1 WO 2015180675 A1 WO2015180675 A1 WO 2015180675A1 CN 2015080148 W CN2015080148 W CN 2015080148W WO 2015180675 A1 WO2015180675 A1 WO 2015180675A1
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Prior art keywords
switching device
current
transformer
circuit
voltage
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PCT/CN2015/080148
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English (en)
French (fr)
Inventor
张平伟
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欧普照明股份有限公司
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Priority claimed from CN201410236053.3A external-priority patent/CN105228288B/zh
Priority claimed from CN201420284271.XU external-priority patent/CN203896559U/zh
Application filed by 欧普照明股份有限公司 filed Critical 欧普照明股份有限公司
Publication of WO2015180675A1 publication Critical patent/WO2015180675A1/zh

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    • H—ELECTRICITY
    • H05—ELECTRIC TECHNIQUES NOT OTHERWISE PROVIDED FOR
    • H05B—ELECTRIC HEATING; ELECTRIC LIGHT SOURCES NOT OTHERWISE PROVIDED FOR; CIRCUIT ARRANGEMENTS FOR ELECTRIC LIGHT SOURCES, IN GENERAL
    • H05B47/00—Circuit arrangements for operating light sources in general, i.e. where the type of light source is not relevant
    • H05B47/10—Controlling the light source

Definitions

  • the present invention relates to the field of constant current driving technology of a light emitting device, and more particularly to a constant current driving circuit and a driving method.
  • the switching device can be turned off after a substantially fixed turn-off delay time due to the turn-off delay characteristic of the switching device. . Since such a turn-off delay time is uncontrollable, when the input voltage is increased, the current flowing through the light-emitting device is increased, so that the light-emitting luminance of the light-emitting device is increased; conversely, when the input voltage is decreased, the light-emitting device is caused to flow. The current is reduced, so that the luminance of the light-emitting device is also lowered. Therefore, there is a problem that the luminance of the light-emitting device fluctuates with fluctuations in the input voltage.
  • the present invention has been made in consideration of the above problems.
  • the present invention provides a constant current driving circuit of a light emitting device which adjusts a time during which the switching device is turned off by a compensation circuit such that an average value of a current flowing through the switching device is not affected by an input voltage fluctuation.
  • a constant current driving circuit for a light emitting device comprising: a transformer including a primary winding and a secondary winding; and a switching device, wherein a primary winding of the transformer is connected in series with the light emitting device a voltage input end and a first end of the switching device; a current sampling circuit having a first end connected to the second end of the switching device and a second end connected to the second end of the secondary winding of the transformer; a driving control circuit having a first end connected to the first end of the secondary winding of the transformer, a second end connected to the second end of the secondary winding of the transformer, and a third end connected to the control of the switching device And configured to control turn-on and turn-off of the switching device; and a compensation circuit having a first end coupled to the first end of the secondary winding of the transformer and a second end coupled to the current sampling circuit First end.
  • the compensation circuit when the switching device is turned on, compensates an input voltage from the voltage input terminal by using a voltage of a secondary winding of the transformer to supply a compensation current to the current sampling circuit.
  • the compensation circuit when the switching device is turned on, supplies a compensation current to the current sampling circuit by using a voltage of a secondary winding of the transformer, thereby adjusting the current sampling circuit.
  • the voltage in turn, regulates the time during which the switching device is turned off so that the average of the current flowing through the switching device is not affected by input voltage fluctuations.
  • FIG. 1 is a schematic block diagram of a constant current driving circuit of a current-type light emitting device according to an embodiment of the present invention
  • FIG. 2 is a schematic circuit diagram showing a constant current driving circuit of a current type light emitting device according to an embodiment of the present invention
  • FIG. 3 is an equivalent circuit diagram showing a constant current driving circuit of a current-type light emitting device according to an embodiment of the present invention during a first operating period;
  • FIG. 4 is an equivalent circuit diagram of a constant current driving circuit of a current-type light emitting device according to an embodiment of the present invention in a second operating period;
  • FIG. 5 is an equivalent circuit diagram showing a constant current driving circuit of a current-type light emitting device according to an embodiment of the present invention in a third operation period;
  • Fig. 6 is a schematic waveform diagram showing signals in a constant current driving circuit of a current type light emitting device according to an embodiment of the present invention.
  • a constant current driving circuit 1 of a current type light emitting device As shown in FIG. 1, a constant current driving circuit 1 of a current type light emitting device according to an embodiment of the present invention is shown, wherein the constant current driving circuit 1 includes a transformer T1, a switching device Q1, a driving control circuit 10, and compensation. Circuit 13 and current sampling circuit 14.
  • the transformer T1 includes a primary winding LA and a secondary winding LB, and the first end A1 of the primary winding LA and the first end B1 of the secondary winding LB are the same end.
  • the turns ratio of the primary winding LA and the secondary winding LB is N.
  • the first end 101 of the drive control circuit 10 is connected to the secondary winding LB of the transformer T1.
  • the first end B1 has a second end 102 connected to the second end B2 of the secondary winding LB of the transformer T1, and a third end 103 thereof connected to the control end B of the switching device Q1, and the driving control circuit 10 is configured to control the turning on and off of the switching device Q1.
  • the switching device Q1 may be a current-driven switching device, for example, a power transistor, the control terminal being a base, the first end being a collector and the second end being an emitter.
  • the drive control circuit 10 controls the switching device Q1 to be turned on by outputting a drive current at its third terminal 103, and controls the switching device Q1 to be turned off by stopping outputting a drive current at its third end 103.
  • the drive control circuit 10 may further include: a conduction control circuit 11 and a shutdown control circuit 12.
  • the first input end 111 of the conduction control circuit 11 is connected to the first end B1 of the secondary winding LB of the transformer T1, the second input end 112 is connected to the reference ground GND, and the output end 113 is connected to the shutdown control.
  • the first input end 121 of the shutdown control circuit 12 is connected to the first end B1 of the secondary winding LB of the transformer T1, and the second input end 122 is connected to the second end B2 of the secondary winding LB of the transformer T1. (ie, reference ground GND), and the third input terminal 123 is connected to the output terminal 113 of the conduction control circuit 11 and the control terminal B of the switching device Q1.
  • the conduction control circuit 11 is configured to output a driving current when the potential of the first end B1 of the secondary winding LB of the transformer T1 is higher than the potential of the second terminal B2 (ie, the ground potential of the reference ground GND), When the potential of the first terminal B1 of the secondary winding LB of the transformer T1 is lower than the potential of the second terminal B2, the drive current is no longer output.
  • the shutdown control circuit 12 is configured not to receive the drive current when a predetermined condition is satisfied, and to receive the drive current when a predetermined condition is not satisfied, such that the drive current does not flow to the switching device Q1 Control terminal B to control the switching device Q1 to be turned off.
  • the control terminal B of the switching device Q1 receives the driving current, thereby controlling the switching device Q1 to be turned on; and when the predetermined condition is not satisfied, the shutdown control circuit 12
  • the third input terminal 123 receives the driving current, and the control terminal B of the switching device Q1 no longer receives the driving current, thereby controlling the switching device Q1 to be turned off.
  • the primary winding LA of the transformer T1 is connected in series with the light emitting device between the voltage input terminal and the first end of the switching device Q1.
  • the switching device Q1 is a current type switching device Q1, and may be, for example, a power transistor. Opening When the device Q1 is a power transistor, the control terminal is the base B of the power transistor, the first end of which is the collector C of the power transistor, and the second end of which is the emitter E of the power transistor.
  • the anode of the light emitting device is connected to the voltage input end, and the cathode of the light emitting device is connected to the first end A1 of the primary winding LA of the transformer T1, the primary side The second end A2 of the winding LA is connected to the first end C of the switching device Q1.
  • the first end A1 of the primary winding LA of the transformer T1 is connected to the voltage input terminal, and the second end A2 of the primary winding LA is The anode of the light emitting device is connected, and the cathode of the light emitting device is connected to the first end C of the switching device Q1.
  • the first end 131 of the compensation circuit 13 is connected to the first end B1 of the secondary winding LB of the transformer T1, and the second end 132 thereof is connected to the first end 141 of the current sampling circuit 14.
  • the first end 141 of the current sampling circuit 14 is connected to the second end E of the switching device Q1, and the second end 142 thereof and the reference ground GND (ie, the second end B2 of the secondary winding LB of the transformer T1) connection.
  • the compensation circuit 13 is configured to provide a compensation current to the current sampling circuit 14 by using a voltage of the secondary winding LB of the transformer T1 when the switching device Q1 is turned on to compensate the voltage input terminal The current fluctuation of the light emitting device caused by the input voltage fluctuation.
  • the constant current driving circuit 1 of the light emitting device may further include: a filter circuit 15 and a freewheeling circuit 16.
  • the filter circuit 15 is connected in parallel with the light emitting device.
  • the first end of the freewheeling circuit 16 is connected to the first end C of the switching device Q1, and the second end thereof is connected to the voltage input terminal.
  • the switching device Q1 When the switching device Q1 is turned on, the potential of the first end A1 of the primary winding LA of the transformer T1 is higher than the potential of the second end A2, and the current flowing direction is: flowing from the voltage input end
  • the filter circuit 15 and the light emitting device then flow from the first end A1 of the primary winding LA of the transformer T1 to the second end A2, then flow through the current sampling circuit 14 via the turned-on switching device Q1, and finally flow into the reference ground GND .
  • the switching device Q1 when the switching device Q1 is turned off, since the current of the primary winding LA of the transformer T1 cannot be abruptly changed, the current flow direction is: the first end A1 of the primary winding LA from the transformer T1. It flows to the second terminal A2 and then continues to flow through the freewheeling circuit 16 and flows into the filter circuit 15 and the light emitting device. At this time, the voltage of the primary winding LA of the transformer T1 is reversed, that is, the potential of the first terminal A1 is lower than the potential of the second terminal A2.
  • FIG. 2 a constant current driving power of a current type light emitting device according to an embodiment of the present invention is shown. Schematic circuit diagram of the road.
  • the compensation circuit 13 includes a third resistor R3, and the current sampling circuit 14 includes a first resistor R1.
  • the filter circuit 15 is a first capacitor C1, and the freewheel circuit 16 includes a first diode D1.
  • the first end of the freewheeling circuit 16 is an anode of the first diode D1, and the freewheeling circuit 16 The second end is the cathode of the first diode D1.
  • the conduction control circuit 11 includes a second resistor R2, a second capacitor C2, and a second diode D2.
  • the second resistor R2 and the second capacitor C2 are connected in series between the first input end 111 and the output end 113 of the conduction control circuit 11, the second The anode of the diode D2 is connected to the second input terminal 112 of the conduction control circuit 11 and the cathode is connected to the output terminal 113 of the conduction control circuit 11.
  • the second capacitor C2 is charged when the potential of the first end B1 of the secondary winding LB of the transformer T1 is higher than the potential of the second terminal B2 (ie, the ground potential of the reference ground GND), and the The conduction control circuit 11 outputs a drive current from its output terminal 113.
  • the second capacitor C2 flows.
  • the current is reversed, and the current flow direction is from the anode to the cathode of the second diode D2, flows through the second capacitor C2 and the second resistor R2, and flows into the secondary winding LB of the transformer T1.
  • the first end B1, and at this time, the output end 113 of the conduction control circuit 11 no longer outputs the drive current.
  • the shutdown control circuit 12 includes a third diode D3, a third capacitor C3, and a Zener diode ZD1.
  • the anode of the third diode D3 and the first end of the third capacitor C3 are connected to the anode of the Zener diode ZD1, and the third diode D3
  • the cathode is connected to the first input end 121 of the shutdown control circuit 12, and the second end of the third capacitor C3 is connected to the second input end 122 of the shutdown control circuit 12, the cathode of the Zener diode ZD1
  • a third input 123 of the shutdown control circuit 12 is coupled.
  • the turn-off control circuit 12 is configured not to extract the turn-on control circuit 11 from its third input terminal 123 when the potential of the control terminal B of the switching device Q1 is insufficient to cause the Zener diode ZD1 to break down.
  • the output current of the output terminal 113 is output, and the output of the conduction control circuit 11 is extracted from the third input terminal 123 when the potential of the control terminal B of the switching device Q1 is sufficient to cause the Zener diode ZD1 to break down.
  • the drive current outputted from the terminal 113 is caused to be output from the output terminal 113 of the conduction control circuit 11.
  • the resulting drive current flows completely into the third input 123 of the shutdown control circuit 12 and no longer flows into the control terminal B of the switching device Q1.
  • the shutdown control circuit 12 controls the switching device Q1 to be turned off by extracting the driving current output from the conduction control circuit 11.
  • the input voltage Vin is initially applied, at this time, there is no storage voltage among all the capacitances shown in FIG. 2 and no current flows through the inductance.
  • the switching device Q1 is in an off state. Since the voltage drop across the first capacitor C1 cannot be abrupt, and the current of the primary winding LA of the transformer T1 cannot be abruptly changed, the input voltage Vin is substantially completely applied to the switching device Q1.
  • the starting circuit can be, for example, a resistor connected between the input voltage Vin and the control terminal B of the switching device Q1, which resistor can have a large resistance value.
  • the starting circuit is not shown in the drawings.
  • the form of the startup circuit is not limited, and those skilled in the art should understand that any existing or future developed startup circuit can be employed.
  • the switching device Q1 When the switching device Q1 is turned off by the linear state to the ON state, the voltage drop across the switching device Q1 V Q1 decreases, so that the voltage V of the transformer T1 primary winding on LA LA is increased, while the secondary winding of transformer T1 LB
  • the upper coupling voltage V LB also increases, which in turn increases the charging current of the second capacitor C2, and the driving current flowing to the control terminal B of the switching device Q1 also increases, so that the switching device Q1 is turned from the linear conduction state to the saturation conduction state.
  • the pass state transition causes the voltage drop V Q1 across the switching device Q1 to further decrease. It can be seen that the switching device Q1, the primary winding LA and the secondary winding LB, the second resistor R2, and the second capacitor C2 of the transformer T1 constitute a positive feedback loop, so that the switching device Q1 can be quickly saturated and turned on.
  • FIG. 1 An equivalent circuit diagram of a constant current driving circuit of the current-type light emitting device according to an embodiment of the present invention is shown in FIG.
  • Vin represents the input voltage
  • Vo represents the voltage across the first capacitor C1
  • V LA represents the voltage across the primary winding LA of the transformer T1
  • V Q1 represents the voltage across the switching device Q1
  • V R1 represents the first The voltage across resistor R1.
  • the positive direction of each voltage is shown in FIG. 3.
  • the positive directions of the respective voltages shown in FIG. 3 are respectively taken as the reference directions of the respective voltages, and will be employed in the following description. Reference direction.
  • the voltage drop V Q1 across it is substantially zero. Since the first resistor R1 is a small resistor, the voltage drop V R1 across it is only a few volts. For example, when the voltage regulation value of the Zener diode ZD1 is 3.3V and the junction voltage of the switching device Q1 is 0.7V, the voltage across the first resistor R1 before the Zener diode ZD1 is broken down is at most 2.6V. After ignoring V Q1 and V R1 , the above voltage formula (1) is simplified as:
  • the voltage V LA of the primary winding LA of the transformer T1 can be expressed by its current i LA as:
  • the Zener diode ZD1 is broken down and starts to draw current, so that the drive current IC2 output from the conduction control circuit 11 no longer flows into the control terminal B of the switching device Q1.
  • V ZD1 represents the breakdown voltage of the Zener diode ZD1
  • i R1 represents the current of the resistor R1
  • V QBE represents the voltage between the control terminal B and the second terminal E of the switching device Q1, that is, the switching device The junction voltage of Q1. Therefore, the current i R1 flowing through the first resistor R1 and the current i Q1 flowing through the switching device Q1 can be obtained:
  • V LB is the voltage of the secondary winding LB of the transformer T1
  • i R3 is the current of the third resistor R3, which can be calculated as:
  • i Q1 (V ZD1 -V QBE )/R1-((Vin-Vo)/N-(V ZD1 -V QBE ))/R3) (8)
  • FIG. 3 a schematic equivalent circuit diagram of a first stage in accordance with an embodiment of the present invention is shown.
  • the switching device Q1 Due to the delay of the current-type switching device Q1 being turned off, even if no current flows into the control terminal B of the switching device Q1, the switching device Q1 remains turned on for the delay time ts. At this time, the voltage between the first terminal C and the second terminal E of the switching device Q1 is still approximately zero, and the current i Q1 flowing through the switching device Q1 actually continues to be determined by the charging condition of the primary winding LA of the transformer T1. That is, determined by the above formula (3), therefore, the current flowing through the primary winding LA of the transformer T1 and the current i Q1 flowing through the switching device Q1 still rise, so that at the end of the delay time ts, the transformer T1 flows. The current of the primary winding LA and the current i Q1 flowing through the switching device Q1 reach their peak currents.
  • the amount of change ⁇ i Q1 of the current i Q1 flowing through the switching device Q1 can be expressed as:
  • the peak current flowing through the switching device Q1 can be expressed as:
  • FIG. 4 a schematic equivalent circuit diagram of a second stage in accordance with an embodiment of the present invention is shown.
  • the switching device Q1 is completely turned off, and the current of the primary winding LA of the transformer T1 cannot be abruptly changed, so the current of the primary winding LA of the transformer T1 flows through the first diode D1 for freewheeling, and the primary side of the transformer T1 at this time.
  • the voltage stored in the second capacitor C2 is completely released, and in this regard, the third stage of operation may also be referred to as a recovery phase.
  • the current i LA of the primary winding LA of the transformer T1 gradually decreases from its peak value to zero.
  • the voltage across the first capacitor C1 is Vo
  • the voltage across the third capacitor C3 is Vo/N.
  • the switching device Q1 goes from the start of conduction to the saturated working area via the linear working area.
  • the voltage at the control terminal B of the switching device Q1 is about to rise to V ZD1 - V C3 , causing the Zener diode ZD1 to break down.
  • the switching device Q1 can be The potential of the control terminal B is expressed as follows:
  • the switching device Q1 Due to the delay of the current-type switching device Q1 being turned off, even if no current flows into the control terminal B of the switching device Q1, the switching device Q1 remains turned on for the delay time ts. At this time, the voltage between the first terminal C and the second terminal E of the switching device Q1 is still approximately zero, and the current i Q1 flowing through the switching device Q1 actually continues to be determined by the charging condition of the primary winding LA of the transformer T1. That is, determined by the above formula (3), therefore, the current flowing through the primary winding LA of the transformer T1 and the current i Q1 flowing through the switching device Q1 still rise, so that at the end of the delay time ts, the transformer T1 flows. The current of the primary winding LA and the current i Q1 flowing through the switching device Q1 reach their peak currents.
  • the amount of change ⁇ i Q1 of the current i Q1 flowing through the switching device Q1 can be expressed as:
  • the peak current flowing through the switching device Q1 can be expressed as:
  • the peak current flowing through the primary winding LA of the transformer T1 and flowing through the switching device Q1 can be expressed as:
  • the sixth phase is exactly the same as the third phase, and will not be described here.
  • the fourth stage, the fifth stage, and the sixth stage are sequentially repeated.
  • the inductor current gradually rises; in the fifth stage, the inductor current continues to rise to the peak current during the delay time of the switching device Q1; in the sixth stage, the inductor current gradually decreases to decrease to zero. Therefore, according to an embodiment of the present invention, the primary winding LA of the transformer T1 operates in a current critical mode.
  • the current flowing through the primary winding LA of the transformer T1 is the same as the current flowing through the switching device Q1, and both reach the peak current, namely:
  • the primary winding LA of the transformer T1 operates in the current critical mode, so that the average value of the current flowing through the primary winding LA of the transformer T1 is the peak current of the primary winding LA of the transformer T1. half.
  • the average value of the current flowing through the current-type light-emitting device is half of the peak current of the primary winding LA of the transformer T1. Therefore, the average current I LED of the current-type light-emitting device can be expressed as:
  • I LED ((V ZD1 -V QBE -Vo/N)*(1/R1+1/R3) (20)
  • the current average value I LED of the current-type light-emitting device is independent of the input voltage Vin, and therefore, even if the input voltage Vin is increased or decreased, the current average value I LED of the current-type light-emitting device is kept constant, correspondingly The luminance of the ground current type light-emitting device is also kept constant.
  • the voltage V LA and the current i LA of the primary winding LA of the transformer T1 and the control of the switching device Q1 are shown when the constant current driving circuit of the current-type light-emitting device according to the embodiment of the present invention is stabilized. terminal B of the current i B. It should be understood that the first to third stages are not shown in FIG.
  • From t0 to t1 is the fourth phase, in which the current iLA of the primary winding LA of the transformer T1 rises from zero until the voltage of the control terminal B of the switching device Q1 causes the Zener diode ZD1 to break down at time t1.
  • From t1 to t2 is the fifth stage, in which the current i LA of the primary winding LA of the transformer T1 continues to rise due to the delayed turn-off characteristic of the switching device Q1, and the difference between t2 and t1 is equal to the delay time ts of the switching device Q1 .
  • the current i LA of the primary winding LA of the transformer T1 reaches a peak value.
  • the embodiment of the present invention does not It is limited to the signal amplitude and the proportional relationship shown in FIG. 6, as long as the tendency of each signal shown in FIG. 6 to change at each time point is satisfied.
  • the constant current driving circuit of the current-type light-emitting device passes the delay off time ts according to the switching device Q1, the primary winding LA of the transformer T1 and the secondary winding LB turns ratio N, and the original of the transformer T1
  • the side winding inductance LA makes the average value of the current flowing through the current type light emitting device independent of the input voltage Vin, thereby ensuring that the brightness of the current type light emitting device is constant and does not fluctuate with fluctuations in the input voltage.
  • the light emitting device is represented as an LED in the embodiment of the present invention, the present invention is not limited thereto, and the light emitting device may include a current type light emitting device.
  • the switching device is represented as a power transistor in the embodiment of the present invention, the present invention is not limited thereto, and the switching device may include a current type switching device.

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Abstract

提供了一种发光器件的恒流驱动电路及其驱动方法。所述恒流驱动电路包括：变压器，其包括原边绕组和次边绕组；开关器件，所述变压器的原边绕组与所述发光器件串联连接在电压输入端与开关器件的第一端之间；电流采样电路，其第一端与所述开关器件的第二端连接，其第二端与所述变压器的次边绕组的第二端连接；驱动控制电路，其第一端连接所述变压器的次边绕组的第一端，其第二端连接所述变压器的次边绕组的第二端，以及其第三端连接所述开关器件的控制端，并且被配置为控制所述开关器件的导通和关断；以及补偿电路，其第一端连接所述变压器的次边绕组的第一端，其第二端连接所述电流采样电路的第一端。

Description

发光器件的恒流驱动电路及驱动方法 技术领域
本发明涉及发光器件恒流驱动技术领域，并且更具体地涉及一种恒流驱动电路及驱动方法。
背景技术
目前，在发光器件的驱动电路中，由于开关器件的关断延迟特性，在所述驱动电路给出控制开关器件关断的控制信号之后，开关器件在基本固定的关断延迟时间后才能关断。由于这样的关断延迟时间不可控，在输入电压增加时，会造成流过发光器件的电流增大，从而使得发光器件的发光亮度增加；反之，在输入电压减小时，会造成流过发光器件的电流降低，从而使得发光器件的发光亮度也降低。因此，存在发光器件的发光亮度随输入电压的波动而波动的问题。
发明内容
考虑到上述问题而提出了本发明。本发明提供了一种发光器件的恒流驱动电路，其通过补偿电路调节控制开关器件关断的时间，从而使得流过所述开关器件的电流的平均值不受到输入电压波动的影响。
根据本发明实施例，提供了一种发光器件的恒流驱动电路，包括：变压器，其包括原边绕组和次边绕组；开关器件，所述变压器的原边绕组与所述发光器件串联连接在电压输入端与开关器件的第一端之间；电流采样电路，其第一端与所述开关器件的第二端连接，其第二端与所述变压器的次边绕组的第二端连接；驱动控制电路，其第一端连接所述变压器的次边绕组的第一端，其第二端连接所述变压器的次边绕组的第二端，以及其第三端连接所述开关器件的控制端，并且被配置为控制所述开关器件的导通和关断；以及补偿电路，其第一端连接所述变压器的次边绕组的第一端，其第二端连接所述电流采样电路的第一端。
根据本发明实施例，在所述开关器件导通时，所述补偿电路通过利用所述变压器的次边绕组的电压向所述电流采样电路提供补偿电流，来补偿由所述电压输入端的输入电压波动引起的所述发光器件的电流波动。
根据本发明实施例的恒流驱动电路，在所述开关器件导通时，补偿电路利用所述变压器的次边绕组的电压向所述电流采样电路提供补偿电流，从而调节所述电流采样电路的电压，继而调节控制开关器件关断的时间，从而使得流过所述开关器件的电流的平均值不受到输入电压波动的影响。
附图说明
通过结合附图对本发明的实施例进行详细描述，本发明的上述和其它目的、特征、优点将会变得更加清楚，其中：
图1示出了根据本发明实施例的电流型发光器件的恒流驱动电路的原理性框图；
图2示出了根据本发明实施例的电流型发光器件的恒流驱动电路的示意性电路图；
图3示出了根据本发明实施例的电流型发光器件的恒流驱动电路在第一工作时段的等效电路图；
图4示出了根据本发明实施例的电流型发光器件的恒流驱动电路在第二工作时段的等效电路图；
图5示出了根据本发明实施例的电流型发光器件的恒流驱动电路在第三工作时段的等效电路图；
图6示出了根据本发明实施例的电流型发光器件的恒流驱动电路中信号的示意性波形图。
具体实施方式
下面将参考附图来说明根据本发明实施例的电流型发光器件的恒流驱动电路及该恒流驱动电路的操作方法。
如图1所示，示出了根据本发明实施例的电流型发光器件的恒流驱动电路1，其中，所述恒流驱动电路1包括：变压器T1、开关器件Q1、驱动控制电路10、补偿电路13以及电流采样电路14。
所述变压器T1包括原边绕组LA和次边绕组LB，所述原边绕组LA的第一端A1与所述次边绕组LB的第一端B1为同名端。所述原边绕组LA和所述次边绕组LB的匝数比为N。
所述驱动控制电路10的第一端101连接所述变压器T1的次边绕组LB 的第一端B1，其第二端102连接所述变压器T1的次边绕组LB的第二端B2，以及其第三端103连接所述开关器件Q1的控制端B，并且所述驱动控制电路10被配置为控制所述开关器件Q1的导通和关断。
所述开关器件Q1可以为电流驱动型开关器件，例如可以为功率三极管，其控制端为基极，其第一端为集电极，其第二端为发射极。
所述驱动控制电路10通过在其第三端103输出驱动电流来控制所述开关器件Q1导通，并且通过在其第三端103停止输出驱动电流来控制所述开关器件Q1关断。
所述驱动控制电路10可以进一步包括：导通控制电路11和关断控制电路12。
所述导通控制电路11的第一输入端111连接所述变压器T1的次边绕组LB的第一端B1，其第二输入端112连接参考地GND，其输出端113连接所述关断控制电路12的第三输入端123以及所述开关器件Q1的控制端B。
所述关断控制电路12的第一输入端121连接所述变压器T1的次边绕组LB的第一端B1，其第二输入端122连接所述变压器T1的次边绕组LB的第二端B2(即参考地GND)，以及第三输入端123连接所述导通控制电路11的输出端113以及所述开关器件Q1的控制端B。
所述导通控制电路11被配置为在所述变压器T1的次边绕组LB的第一端B1的电位高于其第二端B2的电位(即参考地GND的地电位)时输出驱动电流，而在所述变压器T1的次边绕组LB的第一端B1的电位低于其第二端B2的电位时不再输出驱动电流。
所述关断控制电路12被配置为在预定条件满足时不接收所述驱动电流，而在预定条件不满足时接收所述驱动电流，从而使得所述驱动电流不再流到所述开关器件Q1的控制端B，以控制所述开关器件Q1关断。
换句话说，在预定条件满足时，所述开关器件Q1的控制端B接收所述驱动电流，从而控制所述开关器件Q1导通；以及在预定条件不满足时，所述关断控制电路12的第三输入端123接收所述驱动电流，所述开关器件Q1的控制端B不再接收所述驱动电流，从而控制所述开关器件Q1关断。
所述变压器T1的原边绕组LA与所述发光器件串联连接在电压输入端与开关器件Q1的第一端之间。
所述开关器件Q1为电流型开关器件Q1，例如可以为功率三极管。在开 关器件Q1为功率三极管的情况下，其控制端为功率三极管的基极B，其第一端为功率三极管的集电极C，其第二端为功率三极管的发射极E。
具体地，如图1所示，所述发光器件的阳极与所述电压输入端连接，所述发光器件的阴极与所述变压器T1的原边绕组LA的第一端A1连接，所述原边绕组LA的第二端A2与所述开关器件Q1的第一端C连接。
然而，不限于图1所述的连接方式，替换地，所述变压器T1的原边绕组LA的第一端A1与所述电压输入端连接，所述原边绕组LA的第二端A2与所述发光器件的阳极连接，所述发光器件的阴极与所述开关器件Q1的第一端C连接。
所述补偿电路13的第一端131连接所述变压器T1的次边绕组LB的第一端B1，其第二端132连接所述电流采样电路14的第一端141。所述电流采样电路14的第一端141与所述开关器件Q1的第二端E连接，其第二端142与参考地GND(即所述变压器T1的次边绕组LB的第二端B2)连接。
所述补偿电路13被配置为：在所述开关器件Q1导通时，利用所述变压器T1的次边绕组LB的电压向所述电流采样电路14提供补偿电流，来补偿由所述电压输入端的输入电压波动引起的所述发光器件的电流波动。
此外，如图1所示，根据本发明实施例的发光器件的恒流驱动电路1还可以包括：滤波电路15和续流电路16。
滤波电路15与所述发光器件并联连接。续流电路16的第一端连接所述开关器件Q1的第一端C，其第二端连接所述电压输入端。
在所述开关器件Q1导通时，所述变压器T1的原边绕组LA的第一端A1的电位高于第二端A2的电位，此时电流流动方向为：从电压输入端流经所述滤波电路15和所述发光器件，然后从变压器T1的原边绕组LA的第一端A1流到第二端A2，然后经由导通的开关器件Q1流过电流采样电路14，最后流入参考地GND。
另一方面，在所述开关器件Q1关断时，由于所述变压器T1的原边绕组LA的电流不能突变，因此此时电流流动方向为：从变压器T1的原边绕组LA的第一端A1流到第二端A2，然后经由续流电路16续流并流入所述滤波电路15和所述发光器件。此时，所述变压器T1的原边绕组LA的电压反向，即其第一端A1的电位低于所述第二端A2的电位。
如图2所示，示出了根据本发明实施例的电流型发光器件的恒流驱动电 路的示意性电路图。
所述补偿电路13包括第三电阻R3，所述电流采样电路14包括第一电阻R1。
所述滤波电路15为第一电容C1，所述续流电路16包括第一二极管D1，续流电路16的第一端为所述第一二极管D1的阳极，续流电路16的第二端为所述第一二极管D1的阴极。
所述导通控制电路11包括第二电阻R2、第二电容C2以及第二二极管D2。
在所述导通控制电路11中，所述第二电阻R2和所述第二电容C2串联连接在所述导通控制电路11的第一输入端111和输出端113之间，所述第二二极管D2的阳极连接所述导通控制电路11的第二输入端112并且阴极连接所述导通控制电路11的输出端113。
在所述变压器T1的次边绕组LB的第一端B1的电位高于其第二端B2的电位(即参考地GND的地电位)时，对所述第二电容C2进行充电，并且所述导通控制电路11从其输出端113输出驱动电流。
另一方面，在所述变压器T1的次边绕组LB的第一端B1的电位低于其第二端B2的电位(即参考地GND的地电位)时，流过所述第二电容C2的电流反向，此时电流流动方向为：从所述第二二极管D2的阳极到阴极，流经所述第二电容C2和第二电阻R2，流入所述变压器T1的次边绕组LB的第一端B1，并且此时所述导通控制电路11的输出端113不再输出驱动电流。
所述关断控制电路12包括第三二极管D3、第三电容C3和稳压管ZD1。
在所述关断控制电路12中，所述第三二极管D3的阳极、所述第三电容C3的第一端与所述稳压管ZD1的阳极连接，所述第三二极管D3的阴极连接所述关断控制电路12的第一输入端121，所述第三电容C3的第二端连接所述关断控制电路12的第二输入端122，所述稳压管ZD1的阴极连接所述关断控制电路12的第三输入端123。
所述关断控制电路12被配置为在所述开关器件Q1的控制端B的电位不足以使所述稳压管ZD1击穿时不从其第三输入端123抽取所述导通控制电路11的输出端113输出的驱动电流，而在所述开关器件Q1的控制端B的电位足以使所述稳压管ZD1击穿时从其第三输入端123抽取所述导通控制电路11的输出端113输出的驱动电流，使得从所述导通控制电路11的输出端113输 出的驱动电流完全流入所述关断控制电路12的第三输入端123，而不再流入所述开关器件Q1的控制端B。
具体地，在所述导通控制电路11的输出端113输出驱动电流时，并且在所述开关器件Q1的控制端B的电位不足以使所述稳压管ZD1击穿时，所述驱动电流全部流入所述开关器件Q1的控制端B；然而，在所述开关器件Q1的控制端B的电位足以使所述稳压管ZD1击穿时，所述稳压管ZD1被击穿从所述关断控制电路12的第三输入端123抽取大量电流，使得所述驱动电流全部流入所述稳压管ZD1，而不再流入所述开关器件Q1的控制端B。由此，所述关断控制电路12通过抽取所述导通控制电路11输出的驱动电流，从而控制所述开关器件Q1关断。
下面将参考图3－6来描述根据本发明实施例的电流型发光器件的恒流驱动电路的具体操作。
第一阶段(即，启动阶段)
例如，在初始施加输入电压Vin时，此时，图2中所示的所有电容中都没有存储电压并且所述电感中都没有电流流过。开关器件Q1处于断开状态，由于第一电容C1两端压降不能突变、变压器T1的原边绕组LA的电流不能突变，因此输入电压Vin基本上完全施加在开关器件Q1上。
在初始施加输入电压Vin时，利用输入电压Vin通过启动电路向开关器件Q1的控制端B提供微小的启动驱动电流，使开关器件Q1工作于线性导通状态。所述启动电路例如可以是连接在输入电压Vin和开关器件Q1的控制端B之间的电阻，该电阻可以具有较大电阻值。为了不混淆本发明技术方案，在附图中没有示出所述启动电路。在本发明实施例中不对所述启动电路的形式做任何限定，本领域技术人员应了解可以采用任何现有的或将来开发的启动电路。
在开关器件Q1由截止状态变为线性导通状态时，开关器件Q1两端的压降VQ1降低，使得变压器T1的原边绕组LA上的电压VLA增大，同时变压器T1的次边绕组LB上的耦合电压VLB也增大，继而使得第二电容C2的充电电流增大，流到开关器件Q1的控制端B的驱动电流也增大，使得开关器件Q1由线性导通状态向饱和导通状态转变，进而使得开关器件Q1的两端的压降VQ1进一步降低。由此可见，开关器件Q1、变压器T1的原边绕组LA和次边绕组LB、第二电阻R2、第二电容C2构成了正反馈回路，使得开关器件 Q1能够迅速饱和导通。
在此过程中，随着流入开关器件Q1的控制端B的驱动电流逐渐增大，开关器件Q1从线性工作区变到饱和工作区。当开关器件Q1工作于饱和工作区时，根据本发明实施例的电流型发光器件的恒流驱动电路的等效电路图如图3所示。
在此阶段，可以建立以下电压方程：
Vin＝Vo+VLA+VQ1+VR1                   (1)
在公式(1)中，Vin表示输入电压，Vo表示第一电容C1两端的电压，VLA表示变压器T1的初级绕组LA两端的电压，VQ1表示开关器件Q1两端的电压，VR1表示第一电阻R1两端的电压。在图3中示出了各个电压的正方向，在本发明实施例中，将图3所示的各个电压的正方向分别作为各个电压的参考方向，并且在下面的描述中将一直采用这样的参考方向。
在开关器件Q1饱和以后，其两端压降VQ1基本上为0，由于第一电阻R1为很小的电阻，其两端压降VR1仅仅为几伏。例如，在稳压管ZD1的稳压值为3.3V且开关器件Q1的结电压为0.7V，则在稳压管ZD1击穿前第一电阻R1两端的电压最多为2.6V。在忽略VQ1和VR1之后，以上电压公式(1)简化为：
Vin＝Vo+VLA                   (2)
此外，根据电感的电压电流关系，可以将变压器T1的原边绕组LA的电压VLA可以通过其电流iLA表示为：
VLA＝(Vin-Vo)＝LA*(d(iLA)/dt)                    (3)
在变压器T1的原边绕组LA上的电流iLA(即开关器件Q1上的电流iQ1)增大到预定值时，例如使得开关器件Q1的控制端B的电压增大到稳压管ZD1的稳压值时，稳压管ZD1被击穿并且开始抽取电流，从而使得所述导通控制电路11输出的驱动电流IC2不再流入开关器件Q1的控制端B。
此时，流入开关器件Q1的控制端B的电流为0，可以将开关器件Q1的控制端B的电位如下地表示：
VZD1＝VQBE+VR1＝VQBE+iR1*R1                     (4)
在公式(4)中，VZD1表示稳压管ZD1的击穿电压，iR1表示电阻R1的电流，VQBE表示开关器件Q1的控制端B和第二端E之间的电压，即开关器件Q1的结电压。因此，可以得到此时流过第一电阻R1的电流iR1和流过开 关器件Q1的电流iQ1：
iR1＝(VZD1-VQBE)/R1                                   (5)
iQ1＝iR1-iR3＝iR1-((VLB-iR1*R1)/R3)                     (6)
在公式(6)中，VLB为变压器T1的次边绕组LB的电压，iR3为第三电阻R3的电流，其可以计算为：
VLB＝VLA/N＝(Vin-Vo)/N                                 (7)
由以上公式(5)、(6)和(7)可以得到：
iQ1＝(VZD1-VQBE)/R1-((Vin-Vo)/N-(VZD1-VQBE))/R3)         (8)
如图3所示，示出了根据本发明实施例的第一阶段的示意性等效电路图。
第二阶段
由于电流型开关器件Q1关断的延迟性，即使没有电流流入开关器件Q1的控制端B，开关器件Q1在延迟时间ts内也保持导通。此时，开关器件Q1的第一端C和第二端E之间的电压仍近似为零，流过开关器件Q1的电流iQ1实际上继续由变压器T1的原边绕组LA的充电情况来决定，即由以上公式(3)决定，因此，流过变压器T1的原边绕组LA的电流和流过开关器件Q1的电流iQ1仍上升，从而在该延迟时间ts结束时，流过变压器T1的原边绕组LA的电流和流过开关器件Q1的电流iQ1才达到其峰值电流。
在该延迟时间ts内，流过开关器件Q1的电流iQ1的变化量ΔiQ1可以表示为：
ΔiQ1＝((Vin-Vo)/LA)*ts                           (9)
因此，流过开关器件Q1的峰值电流可以表示为：
ipeak Q1＝iQ1+ΔiQ1                                    (10)
因此，综合以上公式(8)、(9)和(10)，流过变压器T1的原边绕组LA和流过开关器件Q1的峰值电流可以表示为：
ipeak Q1＝(VZD1-VQBE)/R1-((Vin-Vo)/N-(VZD1-VQBE))/R3+((Vin-Vo)/LA)*ts
＝(VZD1-VQBE)*(1/R1+1/R3)+(Vin-Vo)*(ts/LA-1/(N*R3))          (11)
如图4所示，示出了根据本发明实施例的第二阶段的示意性等效电路图。
第三阶段
此时，开关器件Q1完全关断，变压器T1的原边绕组LA电流不能突变， 因此变压器T1的原边绕组LA的电流流经第一二极管D1进行续流，此时变压器T1的原边绕组LA上的电压反向，如图5所示。仍以图3所示的电压方向为正方向，则此时VLA＝-Vo，变压器T1的原边绕组LA开始放电。同时，变压器T1的次边绕组LB的耦合电压也反向，如图5所示。仍以图3所示的电压方向为正方向，则此时VLB＝-Vo/N。如图5所示，耦合电压VLB一方面经由第三二级管D3对第三电容C3进行充电，另一方面经过第二二极管D2对第二电容C2进行放电。
最终，一方面，第三电容C3的电压VC3具有如图5所示的极性，并且VLB+VC3+VD2＝0，忽略第二二极管D2的电压VD2，则VC3＝Vo/N，其以图5所示的方向为正方向。另一方面，第二电容C2中存储的电压被完全释放，在这点上，该第三工作阶段也可以被称为恢复阶段。
在该第三阶段结束时，变压器T1的原边绕组LA的电流iLA从其峰值逐渐降低到零。
如图5所示，示出了根据本发明实施例的第三阶段的示意性等效电路图。
第四阶段
在变压器T1的原边绕组LA的电流iLA从其峰值逐渐降低到零时，由于第一二极管D1的存在，原边绕组LA的电流iLA不会反向。
在该第四阶段开始时，第一电容C1两端电压为Vo，第三电容C3两端的电压为Vo/N。
与第一阶段类似，开关器件Q1从开始导通经由线性工作区到达饱和工作区。在第四阶段结束时，开关器件Q1的控制端B的电压即将上升到VZD1-VC3，使稳压管ZD1击穿。
第五阶段
此时，开关器件Q1的控制端B的电压上升到VZD1-VC3，使稳压管ZD1击穿，流入开关器件Q1的控制端B的电流为0，参见图4，可以将开关器件Q1的控制端B的电位如下地表示：
VZD1＝VQBE+VR1+VC3＝VQBE+iR1*R1+Vo/N                (12)
因此，可以得到此时流过第一电阻R1的电流和流过开关器件Q1的电流：
iR1＝(VZD1-VQBE-Vo/N)/R1                           (13)
iQ1＝iR1-iR3＝iR1-((VLB-iR1*R1)/R3)＝iR1-(((Vin-Vo)/N-iR1*R1)/R3)    (14)
由于电流型开关器件Q1关断的延迟性，即使没有电流流入开关器件Q1的控制端B，开关器件Q1在延迟时间ts内也保持导通。此时，开关器件Q1的第一端C和第二端E之间的电压仍近似为零，流过开关器件Q1的电流iQ1实际上继续由变压器T1的原边绕组LA的充电情况来决定，即由以上公式(3)决定，因此，流过变压器T1的原边绕组LA的电流和流过开关器件Q1的电流iQ1仍上升，从而在该延迟时间ts结束时，流过变压器T1的原边绕组LA的电流和流过开关器件Q1的电流iQ1才达到其峰值电流。
在该延迟时间ts内，流过开关器件Q1的电流iQ1的变化量ΔiQ1可以表示为：
ΔiQ1＝((Vin-Vo)/LA)*ts                         (15)
此时，流过开关器件Q1的峰值电流可以表示为：
ipeak Q1＝iQ1+ΔiQ1                                     (16)
因此，综合以上公式(15)和(16)，流过变压器T1的原边绕组LA和流过开关器件Q1的峰值电流可以表示为：
ipeak Q1＝(VZD1-VQBE-Vo/N)/R1-((Vin-Vo)/N-(VZD1-VQBE))/R3+((Vin-Vo)/LA)*ts＝(VZD1-VQBE-Vo/N)*(1/R1+1/R3)+(Vin-Vo)*(ts/LA-1/(N*R3))     (17)
第六阶段
该第六阶段与第三阶段完全相同，在此不再赘述。
此后，依次重复第四阶段、第五阶段和第六阶段。在第四阶段中，电感电流逐渐上升；在第五阶段中，在开关器件Q1的延迟时间内电感电流继续上升达到峰值电流；在第六阶段中，电感电流逐渐下降以降低到0。因此，根据本发明实施例，变压器T1的原边绕组LA工作于电流临界模式。
如上所述，在第五阶段结束时，流过变压器T1的原边绕组LA的电流与流过开关器件Q1的电流相同，并且均达到峰值电流，即：
ipeak Q1＝ipeak LA＝(VZD1-VQBE-Vo/N)/R1-((Vin-Vo)/N-(VZD1-VQBE))/R3+((Vin-Vo)/LA)*ts＝(VZD1-VQBE-Vo/N)*(1/R1+1/R3)+(Vin-Vo)*(ts/LA-1/(N*R3))      (18)
在根据本发明实施例的电流型发光器件的恒流驱动电路达到稳态时，流 过变压器T1的原边绕组LA的电流平均值等于流入发光器件的电流平均值。
如前所述，在本发明实施例中变压器T1的原边绕组LA工作在电流临界模式，因此流过变压器T1的原边绕组LA的电流平均值是变压器T1的原边绕组LA的峰值电流的一半。换句话说，流过电流型发光器件的电流平均值是变压器T1的原边绕组LA的峰值电流的一半。因此，电流型发光器件的平均电流ILED可以表示为：
ILED＝ipeak LA/2＝((VZD1-VQBE-Vo/N)*(1/R1+1/R3)+(Vin-Vo)*(ts/LA-1/(N*R3)))/2   (19)
在以上公式(19)中存在(Vin-Vo)*(ts/LA-1/(N*R3))，在R3的值为LA/(ts*N)时，以上公式(19)可以简化为：
ILED＝((VZD1-VQBE-Vo/N)*(1/R1+1/R3)                   (20)
在公式(20)中，电流型发光器件的电流平均值ILED与输入电压Vin无关，因此，即使输入电压Vin增大或减小，电流型发光器件的电流平均值ILED都保持恒定，相应地电流型发光器件的发光亮度也保持恒定。
如图6所示，示出了根据本发明实施例的电流型发光器件的恒流驱动电路达到稳定时其中变压器T1的原边绕组LA的电压VLA和电流iLA、以及开关器件Q1的控制端B的电流iB。应了解，在图6中没有示出第一阶段到第三阶段。
从t0到t1是第四阶段，其中变压器T1的原边绕组LA的电流iLA从零开始不断上升，直至开关器件Q1的控制端B的电压使得稳压管ZD1在t1时刻击穿。
在t1时刻，开关器件Q1的控制端B的电流iB迅速降低到零。
从t1到t2是第五阶段，其中由于开关器件Q1的延迟关断特性，变压器T1的原边绕组LA的电流iLA继续上升，t2与t1之间的差值等于开关器件Q1的延迟时间ts。
在t2时刻，变压器T1的原边绕组LA的电流iLA达到峰值。
从t2到t3是第六阶段，其中变压器T1的原边绕组LA的电压反向，通过第一二极管D1续流。
在t3时刻，变压器T1的原边绕组LA的电流iLA下降到零。
然后，进入下一个从第四阶段到第六阶段的循环。
在图6中为了示意可能夸大了信号的幅度和比例关系，本发明实施例不 限于图6所示的信号幅度和比例关系，只要满足图6所示的各信号按照各时间点变化的趋势即可。
根据本发明实施例的电流型发光器件的恒流驱动电路，通过根据开关器件Q1的延迟关断时间ts、变压器T1的原边绕组LA和次边绕组LB匝数比N、以及变压器T1的原边绕组电感LA，使得流过电流型发光器件的电流的平均值与输入电压Vin无关，从而保证电流型发光器件的亮度恒定，不随着输入电压的波动而波动。
尽管在本发明实施例中，将发光器件表示为LED，然而本发明不限于此，发光器件可以包括电流型发光器件。另外，尽管在本发明实施例中，将开关器件表示为功率三极管，然而本发明不限于此，开关器件可以包括电流型开关器件。
尽管这里已经参考附图描述了示例实施例，应理解上述示例实施例仅仅是示例性的，并且不意图将本发明的范围限制于此。本领域普通技术人员可以在其中进行各种改变和修改，而不偏离本发明的范围和精神。所有这些改变和修改意在被包括在所附权利要求所要求的本发明的范围之内。
本申请要求2014年5月30日提交的申请号为“201410236053.3”且发明名称为“发光器件的恒流驱动电路及驱动方法”以及2014年5月30日提交的申请号为“201420284271.X”且实用新型名称为“发光器件的恒流驱动电路”的中国优先申请的优先权，通过引用将其全部内容并入于此。

Claims (13)

  1. 一种发光器件的恒流驱动电路，包括：
    变压器，其包括原边绕组和次边绕组；
    开关器件，所述变压器的原边绕组与所述发光器件串联连接在电压输入端与开关器件的第一端之间；
    电流采样电路，其第一端与所述开关器件的第二端连接，其第二端与所述变压器的次边绕组的第二端连接；
    驱动控制电路，其第一端连接所述变压器的次边绕组的第一端，其第二端连接所述变压器的次边绕组的第二端，以及其第三端连接所述开关器件的控制端，并且被配置为控制所述开关器件的导通和关断；以及
    补偿电路，其第一端连接所述变压器的次边绕组的第一端，其第二端连接所述电流采样电路的第一端。
  2. 如权利要求1所述的恒流驱动电路，其中，在所述开关器件导通时，所述补偿电路通过利用所述变压器的次边绕组的电压向所述电流采样电路提供补偿电流，来补偿由所述电压输入端的输入电压波动引起的所述发光器件的电流波动。
  3. 如权利要求1所述的恒流驱动电路，还包括：
    滤波电路，其与所述发光器件并联连接；以及
    续流电路，其第一端连接所述开关器件的第一端，其第二端连接所述电压输入端。
  4. 如权利要求3所述的恒流驱动电路，其中，所述开关器件为电流驱动型开关器件，其控制端为基极，其第一端为集电极，其第二端为发射极；
    所述驱动控制电路通过在其第三端输出驱动电流来控制所述开关器件导通；以及
    所述驱动控制电路通过在其第三端停止输出驱动电流来控制所述开关器件关断。
  5. 如权利要求4所述的恒流驱动电路，其中，所述补偿电路包括第三电阻，所述电流采样电路包括第一电阻。
  6. 如权利要求5所述的恒流驱动电路，其中，所述驱动控制电路包括：
    导通控制电路，其第一输入端连接所述变压器的次边绕组的第一端，其 第二输入端连接所述变压器的次边绕组的第二端，其输出端连接所述开关器件的控制端；以及
    关断控制电路，其第一端连接所述变压器的次边绕组的第一端，其第二端连接所述变压器的次边绕组的第二端，以及其第三端连接所述开关器件的控制端；
    其中，所述导通控制电路的输出端输出驱动电流；在预定条件满足时，所述开关器件的控制端接收所述驱动电流，从而控制所述开关器件导通；以及在预定条件不满足时，所述关断控制电路的第三端接收所述驱动电流，所述开关器件的控制端不再接收所述驱动电流，从而控制所述开关器件关断。
  7. 如权利要求6所述的恒流驱动电路，其中，所述关断控制电路包括第三二极管、第三电容和稳压管，
    其中，所述第三二极管的阳极、所述第三电容的第一端与所述稳压管的阳极连接，所述第三二极管的阴极连接所述关断控制电路的第一输入端，所述第三电容的第二端连接所述关断控制电路的第二输入端，所述稳压管的阴极连接所述关断控制电路的第三输入端，
    其中，所述预定条件为所述稳压管未被击穿。
  8. 如权利要求7所述的恒流驱动电路，其中，所述导通控制电路包括第二电阻、第二电容以及第二晶体管，
    其中，所述第二电阻和所述第二电容串联连接在所述导通控制电路的第一输入端和输出端之间，所述第二晶体管的阳极连接所述导通控制电路的第二输入端并且阴极连接所述导通控制电路的输出端。
  9. 如权利要求8所述的恒流驱动电路，其中，所述滤波电路包括第一电容，所述续流电路包括第一二极管。
  10. 如权利要求9所述的恒流驱动电路，其中，
    所述发光器件的阳极与所述电压输入端连接，所述发光器件的阴极与所述变压器的原边绕组的第一端连接，所述原边绕组的第二端与所述开关器件的第一端和所述第一二极管的阳极连接，所述第一二极管的阴极与所述电压输入端连接；或者
    所述变压器的原边绕组的第一端与所述电压输入端连接，所述原边绕组的第二端与所述发光器件的阳极连接，所述发光器件的阴极与所述开关器件的第一端和所述第一二极管的阳极连接，所述第一二极管的阴极与所述电压 输入端连接。
  11. 一种如权利要求1所述的发光器件的恒流驱动电路的驱动方法，包括：
    在第一时段，所述驱动控制电路输出用于控制所述开关器件导通的控制信号，所述开关器件导通，所述发光器件、所述变压器的原边绕组、所述开关器件、以及所述电流采样电路形成串联连接，并且所述补偿电路向所述电流采样电路输出补偿电流，在所述开关器件的控制端的电压达到预定电压值时，所述第一时段结束；
    在第二时段，所述驱动控制电路不再输出用于控制所述开关器件导通的控制信号，所述开关器件仍保持导通，所述发光器件、所述变压器的原边绕组、所述开关器件、以及所述电流采样电路形成串联连接，经过所述开关器件的延迟时间之后，所述第二时段结束；以及
    在第三时段，所述驱动控制电路不再输出用于控制所述开关器件导通的控制信号，所述开关器件关断。
  12. 一种如权利要求4所述的发光器件的恒流驱动电路的驱动方法，包括：
    在第一时段，所述导通控制电路输出驱动电流，所述开关器件的控制端接收所述驱动电流，所述开关器件导通，所述发光器件、所述变压器的原边绕组、所述开关器件、以及所述电流采样电路形成串联连接，并且所述补偿电路向所述电流采样电路输出补偿电流，在所述开关器件的控制端的电压达到预定电压值时，所述第一时段结束；
    在第二时段，所述导通控制电路输出驱动电流，所述关断控制电路接收所述驱动电流，所述开关器件的控制端不再接收所述驱动电流，所述开关器件仍保持导通，所述发光器件、所述变压器的原边绕组、所述开关器件、以及所述电流采样电路形成串联连接，并且所述补偿电路向所述电流采样电路输出补偿电流，经过所述开关器件的关断延迟时间之后，所述第二时段结束；以及
    在第三时段，所述开关器件关断，所述导通控制电路不再输出驱动电流，所述发光器件、所述变压器的原边绕组以及所述续流电路形成串联连接，在所述变压器的原边绕组的电流逐渐下降到零时，所述第三时段结束。
  13. 一种如权利要求10所述的发光器件的恒流驱动电路的驱动方法，包 括：
    在第一时段，所述变压器的次边绕组的电压经由第二电阻给第二电容充电的充电电流作为驱动电流流动到所述开关器件的基极，所述开关器件导通，所述发光器件、所述变压器的原边绕组、所述开关器件以及所述第一电阻形成串联连接，并且所述变压器的次边绕组的电压经由第三电阻向第一电阻输出补偿电流，在所述开关器件的基极的电压足以击穿所述稳压管时，所述第一时段结束；
    在第二时段，所述变压器的次边绕组的电压经由第二电阻给第二电容充电的充电电流被击穿的所述稳压管抽取，而不再作为驱动电流流动到所述开关器件的基极，由于所述开关器件的关断延迟特性，所述开关器件仍保持导通，所述发光器件、所述变压器的原边绕组、所述开关器件、以及所述第一电阻形成串联连接，并且所述变压器的次边绕组的电压经由第三电阻向第一电阻输出补偿电流，经过所述开关器件的关断延迟时间之后，所述第二时段结束；以及
    在第三时段，所述开关器件关断，所述变压器的原边绕组的电压反向，所述发光器件、所述变压器的原边绕组以及所述续流电路形成串联连接，同时，所述变压器的次边绕组的电压也反向，所述第二电阻、所述第二电容和所述第二二极管串联连接使所述第二电容放电，所述第三电容和所述第三二极管连接给第三电容充电，在所述变压器的原边绕组的电流逐渐下降到零时，所述第三时段结束。
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CN203407049U (zh) * 2013-08-07 2014-01-22 刘文涛 发光二极管驱动电路
CN203896559U (zh) * 2014-05-30 2014-10-22 欧普照明股份有限公司 发光器件的恒流驱动电路

Cited By (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN111212504A (zh) * 2020-03-27 2020-05-29 杰华特微电子（杭州）有限公司 Bifred变换器及其控制方法及应用其的led驱动电路
CN111212504B (zh) * 2020-03-27 2023-10-27 杰华特微电子股份有限公司 Bifred变换器及其控制方法及应用其的led驱动电路

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