WO2015107088A1

WO2015107088A1 - Receiver and method for determining phase information of a received signal, and system and method for measuring information across a distance to an object

Info

Publication number: WO2015107088A1
Application number: PCT/EP2015/050606
Authority: WO
Inventors: Thorsten Giese
Original assignee: Fraunhofer-Gesellschaft zur Förderung der angewandten Forschung e.V.
Priority date: 2014-01-15
Filing date: 2015-01-14
Publication date: 2015-07-23
Also published as: DE102014200634A1

Abstract

The invention relates to a receiver for determining phase information of a received signal, comprising a sampled value generator for providing at least one pair of sampled values. The sampled value generator is designed to sample the received signal at a sampling rate which is lower than an average frequency of the received signal. The sampled value generator is designed to provide two sampled values as a pair of sampled values, said sampled values being separated by a time interval of ¼ - T_cw (4m + 1) or ¼ - T_cw (4m + 3), wherein m is a whole number greater than or equal to 1, and T_cw is an average period duration of the received signal.

Description

A receiver and method for determining phase information of a received signal and system and method for measuring information about a

Distance to an object

description

Embodiments of the present invention relate to a receiver and a method for determining phase information of a received signal. Further embodiments of the present invention relate to a system and method for measuring information about a distance to an object. Further embodiments of the present invention relate to a method for measuring the distance and the reflectivity property of an object with a CW ("Continuous Wave") LIDAR ("Light Detection and Ranging") method, for example using a real-time algorithm.

In a known CW ("Continuous Wave") - LIDAR ("Light Detection and Ranging") - method, an intensity-modulated light signal is sent to the object, which is reflected at the object to a detector. Due to the duration of the light, which the beam needs to the object and back to the detector, a phase shift between emitted, intensity-modulated light and detected signal can be seen, because

Distance = speed ^■ time

R ^* = c - t _L

(1) where R ^{* is} the distance, c the speed and t _L the time.

If the phase shift between the two signals can be determined, then it is possible to draw conclusions about the distance to the object (compare FIGS. 12, 13). In Fig. 12, an illustration of a CW method (outgoing wave 2, reflected wave 4) is shown. In Fig. 13, a construction of a known CW system is shown. If the intensity of the laser A _T (T = "Transmit") is modulated, for example, with a sine function of the period T _C w

227π1

A _T (t) = a _r - sin - t i +

cww

(2), for example, the received detector signal A _R (R = "Receive") also contains a sinusoidal function of the period T _C w

, (t) = a _R - si i - ^ - t + ^)

The phase difference Δφ between the transmitted signal A _T (t) and the received signal A _R (t) is determined by the subtraction of both phases

Δφ = _φ τ - _ψκ

(4)

From the relationship between the transit time of the signal t _L and the period T _C w and the phase difference Δφ / 2π results in the ratio t _L Αφ

l cw 2π

(5)

From this relationship, the distance R ("Range") to the object can be calculated

2 2 TT ₍₆₎ From the above equation 6, the speed of light c and the period T _C w of the intensity modulated light signal are known. The factor ! can be explained by the fact that the light travels twice the distance 2R (round trip). However, since only the distance to the object is measured, it is divided by the factor 2. To the distance To be able to determine the phase difference Δφ between the emitted and the received, intensity-modulated light wave should be determined.

FIG. 13 shows an exemplary structure of a known CW LIDAR system. As shown in Fig. 13, a laser diode (LD) having the sinusoidal function A _T (t) is driven, whereby the intensity of the laser is sinusoidally modulated. A photodiode (PD) converts the reflected light into electrical signals. Subsequent signal conditioning generates the received signal A _R (t). The functions A _T (t) and A _R (t) differ in amplitude and phase, but not in frequency.

One difficulty with a scanning CW LIDAR system is the constantly changing phase at the detector. For this, imagine a comb structure (see Fig. 14). FIG. 14 shows a scan of a comb tip (left) and a scan of a valley (right). If the light beam points to a ridge, another phase shift is applied to the detector, as if the light beam were radiating exactly between two ridge points. The faster the light beam is scanned over a crest peak and the ridge, the faster the evaluation algorithm should be able to detect and evaluate the different phase signals. FIG. 15 shows two exemplary measurements which have a different phase difference relative to one another (resulting from the distance between transmitter and object). In Fig. 15, a phase relationship between outgoing and detected signal is shown. As described above, when the intensity is modulated with a sine function, the emitted (outgoing) intensity modulated light wave is bandlimited. If a filter stage is also present in the returning path, ie from the detector to the receiver, for example to the ADC (analog-to-digital converter), a bandpass signal is output at the receiver or at the ADC - feels.

The phase information can be extracted from a bandpass signal inter alia from a quadrature downmix. This assumes an ideal BP ² signal

U _BP {t) = ü - s (27rf _M t + (p) In quadrature down-conversion, the real BP signal (bandpass signal) is multiplied by two sine waves phase-shifted by 90 °. This is shown in FIG. 16. In Fig. 16, an exemplary quadrature demodulation is shown. From the trigonometric functions sin () - sin (/>) = '· (cos (a - b) - cos (a + b))

² (8)

sin () - cos (i) = ^■ - · (sin (a - b) + sin (a + b)) (9)

It can be seen that as soon as the arguments (a and b) are identical, two signal components are created. The proportion should be removed at twice the frequency by a suitable filter. Applied to equation 7, after a TP (low-pass) filter, cos (>)

1

sin (p) (1 1)

Thus, after the TP filter, we obtain a complex value pair (Equations 10 and 11) with two different values from which the phase relationship and amplitude can be calculated.

A disadvantage of the quadrature demodulation according to FIG. 16a, however, is that the demodulation loses half the amplitude and adds a group delay through the TP filter.

DE 100 01 015 A1 describes a method for measuring the distance of objects, atmospheric particles and the like by means of LIDAR or laser radar signals. This known method is essentially based on an FFT ("Fast Fourier Transform").

US 201 1/0317147 A1 describes a known method using time-shifted PN ("pseudo-random noise") codes for CW-LIDAR, radar and sonar, this method being known as FM (Frequency Modulation) demodulation performed for a CW signal. This FM demodulation occurs, as shown in Fig. 10 of this document, prior to data acquisition. In particular, this known method requires additional resources. In Bogatscher, S. et al., "Laser Rangefinder based on MEMS mirrors for adaptive robotics", Microsystems Technology Congress 2013, 14-16 October 2013 in Aachen, another well-known LIDAR system is described 3 shows a LIDAR system and its evaluation, a laser which is modulated at the frequency f _m and, furthermore, that an ADC is sampled at 8 times the modulation frequency Subsequently, a classic demodulation is performed via two multipliers and using low-pass filters.

A general problem of the LIDAR systems described above is that they are relatively complicated and at the same time involve comparatively high computational effort or the use of additional resources.

The object of the present invention is therefore to provide a concept for determining a phase information of a received signal, which enables a simpler, more efficient and resource-saving implementation.

This object is achieved by a receiver according to claim 1, a system according to claim 8, a method according to claim 13, a method according to claim 14 and a computer program according to claim 15.

Embodiments of the present invention provide a concept for determining phase information of a received signal with a sample provider, wherein the sample provider is arranged to sample the received signal at a sampling rate less than a mean frequency of the received signal. In this case, the sample value generator is designed to provide, as a pair of samples, two samples which are separated by a time interval

~ r _ci , (4m + l)

or from ~ T _cw (4m + 3)

4, where m is an integer greater than or equal to (>) 1, and where T _C w is an average period of the received signal.

The gist of the present invention is that the above simpler, more efficient, and more resource efficient implementation can be achieved if the received signal is sampled at a sampling rate less than a mean frequency of the received signal, and if as a pair of samples two samples are provided, which have the above-mentioned time interval. As a result, a high level of implementation complexity and at the same time a comparatively high computational effort or the use of additional resources can be avoided. Thus, the simpler, more efficient and resource-saving implementation can be made possible.

In further embodiments of the present invention, the receiver includes a parameter calculator for calculating the phase information of the received signal based on the pair of samples. Thus, the phase information of the received signal can be determined for the calculation of information about the distance to an object.

Further embodiments of the present invention provide a system for measuring information about a distance to an object comprising a transmitter, a receiver and a control unit. The transmitter is designed to transmit a transmission signal. The receiver is designed to receive a received signal. Here, the receiver is designed as above to determine phase information of a received signal. The control unit is designed to control an average frequency of the transmission signal and the sampling rate as a function of a clock frequency of the system.

In further embodiments of the present invention, the receiver of the system comprises a parameter calculator for calculating the phase information of the received signal based on the pair of samples. Furthermore, the parameter calculator is designed to provide the information about the distance to the object based on a phase information of the transmitted signal and the phase information of the received signal to calculate.

Thus, according to embodiments, the phase information of the received signal or the information about the distance to the object can be determined.

Embodiments of the present invention will be explained in more detail below with reference to the accompanying figures, in which identical or equivalent elements are designated by the same reference numerals. Show it:

1 a shows a receiver for determining a phase information of a received signal according to an exemplary embodiment of the present invention; an exemplary illustration of providing at least one pair of samples with a sample value provider of the embodiment of the receiver according to FIG. 1 a; a block diagram of an embodiment of the sample value provider of the embodiment of the receiver of FIG. 1a; a block diagram of another embodiment of the sample of the embodiment of the receiver 100 of FIG. 1 a; a block diagram of another embodiment of the sample of the embodiment of the receiver of FIG. 1 a; a block diagram of an embodiment of the Wertbaararbimmers of the embodiment of the sample value provider of FIG. 4;

6a is a block diagram of an embodiment of a

Parameter calculator of the embodiment of the receiver of FIG. 1 a; Fig. 6b, c is a schematic diagram and a block diagram, respectively, for illustrating an exemplary time-shifted sample; Fig. 6d is an exemplary equivalent circuit diagram of an ideal sampler;

7a-d exemplary embodiments of a time-shifted sampling with various exemplary parameters;

8a is an illustration of an exemplary power distribution in FIG

To scan; 8b is an illustration of an exemplary scan area of a 2D

Scanners without adaptation or with adaptive scanning range adaptation;

FIG. 8c is an illustration of an exemplary measurement according to the embodiment of FIG. 4; FIG.

9 shows a system for measuring information about a distance to a

Object according to an embodiment of the present invention;

10a-c show details of the system shown in FIG. 9 according to further embodiments of the present invention;

Fig. 1 1 shows another embodiment of a time-shifted scan;

Figs. 12, 13 are illustrations of a conventional CW system;

14 is an illustration of a scan of a comb peak and a scan of a valley in a known CW system;

FIG. 15 shows a phase relationship between outgoing and detected signal in a known CW system; FIG.

Fig. 16 is a block diagram of a known quadrature demodulation;

17 is a table summarizing various names of

Description; and 18a ~ f inventive algorithms with legend for simulating an example time-shifted sampling.

Before the present invention is explained below with reference to the figures, it is pointed out that in the exemplary embodiments illustrated below, identical elements or functionally identical elements in the figures are provided with the same reference numerals. A description of elements with the same reference numerals is therefore interchangeable and / or applicable to each other in the various embodiments.

1 a shows a receiver 100 for determining phase information 125, cp _{R of} a received signal 105, A _R , according to an embodiment of the present invention. As shown in Fig. 1a, the receiver 100 comprises a sample provider 110 for providing at least one pair of samples 15 (A, B; A _1; B ₁₎ . Here, the sample provider 110 is adapted to receive the received signal 105, A _R , with a sampling rate of 1 1 1, f _Abt , which is less than an average frequency (f _C w) of the received signal 105, A _R , is.

In embodiments, the received signal 105, A _R , is a sinusoidal signal _{R R} sm {2nf _cw t + (p _R ) where _{R R} and <p _{R are} the amplitude information and the phase information, respectively, and f _C w is the average frequency of the received signal 105 , A _R , represent.

By way of example, FIG. 1 b illustrates providing at least one pair of samples 15 (A, B, A _t , Bi) with the sample provider 110 of the embodiment of the receiver 100 according to FIG. 1 a. In Fig. 1 b exemplary diagrams 10, 20, 30 and 40 are shown. Here, the exemplary diagrams 10, 20 each illustrate a temporal distance T _Abt two samples or the pair of samples 1 15 (A, B, Ai, B,), while the exemplary diagrams 30, 40 respectively the corresponding pair of samples 1 15th (A, B; Α,, Β ·,) make clear.

In the example diagram 10, a time interval T is _Abt of _Tcw {Am + l)

4, where m is equal to 1, for example, and where T _C w is the average period of the received signal 105, A _R. In the exemplary diagram 20, a time-related distance I AM of

4 illustrates, where m is an integer, for example equal to 1, and where T _C w is the mean period of the received signal 105, A _R.

Further, in the example diagram 10, the time interval l AM corresponds to one

5/4 times the average period of the received signal 105, A _R , while in the exemplary diagram 20, the time interval T _A bt a 7/4 times the average period of the received signal 105, A _R corresponds. As shown in the exemplary diagrams 10, 20, the time interval T _A bt is greater than the average period Tcw of the received signal 105, A _R. The respective time interval T _A bt in the exemplary diagrams 10, 20 corresponds, for example, to the sampling rate 1 1 1, f _A bt _> while the mean period T _C w of the received signal 105, A _R , corresponds to the mean frequency (f _C w ) of the received signal 105, A _R. Thus, the exemplary diagrams 10, 20 essentially represent sampling of the received signal 105, A _R , at a sampling rate 1 1 1, f _Abt> where the sampling rate 1 1 1, f _Abt , is less than the average frequency (f _C w) of the received signal 105, A _R. In other words, the exemplary diagrams 10, 20 essentially illustrate undersampling of the received signal 105, A _R.

In embodiments, for m, an integer greater than or equal to (>) 1 can be selected. By setting m, the time interval T _Abt can thus be correspondingly increased or the sampling rate 1 1 1, f _Abt , be reduced accordingly.

In the exemplary diagrams 10, 20 are four samples A, B; AB, wherein the samples A, B form a first pair and the samples Ai, B form a second pair of samples. Here, the time offset between the first pair A, B and the second pair Ai, B, of samples each given by the time interval T _Abt .

The exemplary diagrams 30, 40 respectively show the corresponding samples A, B; Ai, Bi as a sequence s (n) of samples. Here n denotes an index (n = 0, 1, 2, 3...) Of the respective sample A, B; Ai, B _{1 of} the sequence s (n).

FIG. 2 shows a block diagram of an exemplary embodiment of the sample value provider 110 of the exemplary embodiment of the receiver 100 according to FIG. 1 a. In the exemplary embodiment shown in FIG. 2, sample value generator 110 includes an analog-to-digital converter 12 (ADC) for sampling the received signal 105, A _R , and a value-pairing determiner 1 14 for determining the at least one pair of samples 1 15 (A, B; A ^ B ^ based on a sampling signal 1 13 sampled by the analog-to-digital converter 12 (ADC). Here, the analog-to-digital converter 1 is 12 (ADC), for example, designed to sample the received signal 105, A _R , with the sampling rate 1 1 1, f _A bt.

FIG. 3 shows a block diagram of a further exemplary embodiment of the sample value provider 110 of the exemplary embodiment of the receiver 100 according to FIG. 1 a. FIG. 3 shows a time-shifted scan with an ADC. In the embodiment shown in FIG. 3, the sample provider 110 includes an analog-to-digital converter 12 (ADC) for sampling the received signal 105, A _R , and a value pairing determiner 1 14 for determining the at least one pair of samples 15 (FIG. A, B; Α,, B,) based on a sample signal 1 13 sampled by the analog-to-digital converter 12 (ADC). Here, the analog-to-digital converter 12 (ADC) is designed to receive the received one Signal 105, A _R , with the sampling rate 1 1 1, f _Ab t to scan. For example, the value pairing determiner 1 14 is designed to decimate the sampling signal 1 13 sampled by the analog-to-digital converter 12 (ADC) by a decimation factor 4 (block 312) by a first sample A of the at least one pair of samples 1 15 (A, B, AB,). Further, the value pairing determiner 14 may be configured to delay the sampling signal 1 13 sampled by the analog-to-digital converter 12 (ADC) by the time interval T _Abt (block 322) and by a delay signal delayed by the time interval T _Abt 323 with the decimation factor 4 (block 324) to obtain a second sample B of the at least one pair of samples 1 15 (A, B; A, B, B). For example, block 322 in FIG. 3 is a delay Z '"" for delaying the time interval T _Abt , while blocks 312, 324 in FIG. 3 are, for example, decimators for decimation with the decimation factor 4. FIG. 4 shows a block diagram of a further exemplary embodiment of the sample value renderer 110 of the exemplary embodiment of the receiver 100 according to FIG. 1 a. FIG. 4 shows an expanded structure of the time-shifted sampling or a block diagram of a "LIDAR Core Algorithm". In the embodiment shown in FIG. 4, the sample provider 110 includes an analog-to-digital converter 12 (ADC) for sampling the received signal 105, A _R , and a value pair determiner 14 for determining a first pair A, B and a second one Pair A ^ B ^ of the at least one pair of samples 1 15 (A, B; A ^ B ^ based on a sample signal 13 sampled by the analog-to-digital converter 12 (ADC). Here, the analog-to-digital converter is 1 12 (ADC) is designed to sample the received signal 105, A _R , with the sampling rate 1 1 1, f _Abt . For example, the value-pairing tester 1 14 is designed to receive the signal from the analog-to-digital converter 12 (FIG. ADC) sampled sampling signal 1 13 with a decimation factor 4 to decimate (block 312) to obtain a first sample A of the first pair A, B of the at least one pair of samples 1 15 (A, B; Α-ι, B ^) Furthermore, the value-pairing tester 1 14 may be designed to be analogous to that of the analogue digita I-converter 1 12 (ADC) sampled sampling signal I 13 to delay the time interval T _Abt (block 322) and to decimate by the time interval T _Abt delayed delay signal 323 with the decimation factor 4 (block 324) to a second Sample B of the first pair A, B of the at least one pair of samples 1 15 (A, B; A ^ B,).

Further, the value pairing determiner 14 may be configured to delay the sampling signal 1 13 sampled by the analog-to-digital converter 12 (ADC) by a double of the time interval T _Abt (block 332) and by a double of the time interval T _{A t} delayed delay signal 333 with the decimation factor 4 to decimate (block 334) to a first sample A, the second pair AB, the at least one pair of samples 1 1 5 (A, B, A,, Bi) receive. Further, the value pairing determiner 14 may be configured to delay the sampling signal 1 13 sampled by the analog-to-digital converter 12 (ADC) by a factor of three times the time interval T _{A t} (block 342) and by one-third of the time Distance T _Abt decelerated delay signal 343 with the decimation factor 4 (block 344), to obtain a second sample Eh of the second pair A, B _{1 of} the at least one pair of samples 1 15 (A, B; Ai, Bi).

For example, the blocks 322, 332, 342 in FIG. 4 are a delay unit Z ^"7 ""for delaying by the time interval T _Abt , a delay unit Z for delaying by twice the time interval T _Abt and a delay unit Z ^{~ Y} ', respectively. ^M for delaying by three times the time interval T _Abt> while the blocks 31 2, 324, 334, 344 in Fig. 4 are each a decimator for decimation with the decimation factor 4.

Optionally, the value pairing determiner 14 is adapted to multiply the first sample ΑΊ of the second pair B _{1 of} the at least one pair of samples 1 15 (A, B; Α ^ Β ^ by a factor (-1) (block 335)) Furthermore, the value pairing determiner 1 14 is designed (optionally) to include the second sample value Β-ι of the second pair Α ^ of the at least one pair of sample values 15 (A, B; Ai, B) multiplied by a factor (-1) (block 345) to obtain a modified second sample Bi '.

For example, the blocks 335, 345 in FIG. 4 are multipliers multiplied by the factor (-1).

Referring to Fig. 4, an extension of the sub-scan can be realized. According to embodiments, sub-sampling may be extended to improve signal-to-noise ratio (SNR), for example. For example, Fig. 3 compares with the associated simulation in Fig. 7a. The real or imaginary part is determined, for example, in each case from one of four samples (FIG. 7a). In embodiments, the last two samples (Ai, B ^ are discarded, however, it will be appreciated that the two latter samples (A _h ) correspond to the previous samples (A, B) when multiplied by the factor -1 4, the extended structure is shown diagrammatically All other boundary conditions, such as the ratio of the sampling frequency f _Abt to the applied frequency f _cw are identical to those according to the embodiment of Fig. 3. The first two branches are taken from Fig. 3 In the third branch, the signal is delayed by two clocks (block 332) and then decimated by the factor 4 (block 334) The fourth branch is delayed by three clocks (block 342) and also decrimped by the factor 4 (block 344) In this setup we get next to the already known complex value pair (A + jB) another complex value pair (Ai + jB,) or (Α, '+ jB). In embodiments, the signal quality can be increased by the addition (eg blocks 510, 520 in FIG. 5) of both value pairs to a value pair. Further, in further embodiments, by maintaining the value pairs, one pixel may now be split into two sub-pixels.

The extension according to the embodiment of FIG. 4 allows a doubling of the pixel resolution or an improvement of the SNR over the embodiment of FIG. 3, by a double measurement within a pixel.

For this purpose, an example can be seen in Fig. 8c to what extent the dual measurement further increases the image quality (magnitude) of a noisy image. FIG. 8c shows a comparison of both methods (according to the block diagrams of FIGS. 3 and 4). Above, an ideal magnitude image is shown, bottom left, the original method (according to the embodiment of Fig. 3), bottom right, the modified method (according to the embodiment of Fig. 4), with an SNR of -10 dB.

The above-described subscan extension of FIG. 4 will also be apparent from the simulation result of FIG. 7a. For a complex value pair ("Real ValueTlmag Value"), two samples were taken from the line "Sampled Values" (first and second samples A, B) according to FIG. The third and fourth samples (A ^ B ^ are discarded as shown in FIG. 3, for example.

However, it has been recognized that the discarded sampling points (Α-ι, B,) can be multiplied by the factor -1 to take the values of the first and second samples (A, B). This embodiment is based on the embodiment of FIG. 4.

FIG. 5 shows a block diagram of an embodiment of the value-pairing estimator 1 14 of the exemplary embodiment of the sample provider 110 in FIG. 4. In the embodiment shown in FIG. 5, the value-pairing determiner 1 14 is designed to match the first sample A of the first pair A, B of FIG at least one pair of samples 1 15 (A, B; A _1; Bi) and the first sample A, of the second pair A _{1 t} B, of the at least one pair of samples 1 15 (A, B; Α,, B) to combine (block 510) to obtain a first combination value C. Furthermore, the value pairing determiner 1 14 may be designed to determine the second sample value B of the first pair A, B of the at least one pair of samples 1 1 5 (A, B; A,, Bi) and the second sample value B, of the second sample value. pair (s) of the at least one pair of samples 1 15 (A, B; Αι, Bi) (step 520) to obtain a second combination value C-.

5, the sampling signal 1 13 sampled by the analog-to-digital converter 12 (ADC) is also referred to as "ADC_Sample." Further, in Fig. 5, the first combination value C is designated "real", while the second combination value d The blocks 322, 332, 342, 312, 324, 334, 344 in Fig. 5 substantially correspond to the corresponding blocks in Fig. 4. The blocks 322, 332, 342 in Fig. 5 are respectively denoted by the symbol 501, Z ^"n . Here, the symbol 501 represents an nth-order delay element or a delay unit for delaying n times the time interval T _Abt . Further, the blocks 312, 324, 334, 344 in FIG. 5 are indicated by the symbol 502. Here, the symbol 502 represents a "downsampling" by a factor of 4 or a decimator for decimation by the decimation factor 4. Further, in Fig. 5, the blocks 510, 520 are marked with the symbol 503. Here, the symbol 503 represents a combiner ( eg adder).

The exemplary embodiments shown in FIGS. 3 to 5 essentially represent different implementations for a value-pair determination with the value-pairing tester 1 14. For example, the sample values A, B, Α Β, A, B ^ or the combination values C, d can be flexible be determined. Thus, in the embodiments according to FIGS. 3 to 5, a flexible implementation for the sample value or combination value determination can be realized.

6a shows a block diagram of an embodiment of a parameter calculator 120 of the embodiment of the receiver 100 according to FIG. 1a. For example, as shown in FIG. 6a, the receiver 100 includes a parameter calculator 120 for calculating the phase information 125, <p _R , of the received signal 105, A _R , based on the at least one pair of samples 15 (A, B; For example, the parameter calculator 120 is configured to calculate the phase information 125, <p _R , of the received signal 105, A _R , using the equation cp _R = tan ¹ (B / A), where A and B are two Samples of the pair of samples 15 (A, B, Ai, Bi) are those having the time interval T _A bt. Thus, according to exemplary embodiments, the phase information of the received signal can be determined for a measurement of information about a distance to an object.

The above-described sampling of the received signal with the sampling value generator 110 can also be referred to as a time-delayed sampling of the received signal or of the detector signal. Embodiments of the present invention have the advantage that it is possible by the time-shifted sampling of the detector signal to be able to perform a direct downmixing (folding) into the zero position (DC component) when choosing a specific sampling frequency or sampling rate (f _A bt) ,

Figuratively imagine the inventive time-shifted scan with a wheel which rotates at a speed known to us. This wheel is illuminated with a stroboscope. Since the speed of the wheel is known, it is possible to calculate a frequency for the stroboscope that gives the impression that the wheel is not rotating. We also imagine that the wheel has only one spoke and that spoke corresponds to a pointer in the unit circle (see Fig. 6b). In Fig. 6b different positions are shown in the unit circle. If the real and imaginary parts of the pointer can be determined, then the phase position of the wheel can be calculated from this. In the following, the mathematical derivation is described, how the suitable sampling frequency can be determined, with which, as it were, the wheel stands still.

Fig. 6d shows an equivalent circuit diagram of an exemplary ideal sampler. As can be seen in FIG. 6d, in an ideal ADC, a BP signal (rate of revolution of the wheel) may be multiplied by a Dirac pulse.

In the following derivation, the multipliers of FIG. 16 are replaced by, for example, two ADCs. These ADCs sample the bandpass signal U _B p (t) at the frequency f _Abt . In this case, the clock of one ADC to the other ADC is, for example, phase-shifted by 90 ° (cf., FIG. 6c). FIG. 6c shows a schematic structure of a time-shifted scan with two ADCs. The Dirac pulse is a Fourier series representation

1 "V" 1

δ (ί) = - + / - * cos (2nf _Abt 'i · t) \ · 2π

2π Ζ π

(12)

S (t) = 1 + ^ 2 · cos (2nf _Abt -it)

(13) and whose Fourier transform is f) = Σ S (f-if _Abt )

(14)

The spectrum of a Dirac pulse offset in time by φ results from the multiplication by ^e

Af) = V 6 {f-if _Aht ye - ** f *

i (15)

(16)

From the ideal bandpass signal (signal of one frequency, i.e. bandwidth

16

U _BP (t) = cos 2nf _cw t)

(17) and its Fourier transform

UBP (I) =

arise by a convolution with the spectrum of the time-shifted Dirac pulse, the spectra of the sampled signals.

For the channel A, which results from a sampling of the signal U (t), the result is

For the channel B, which results from a sampling of the signal U _B p (t), but with the sampling phase-shifted by 90 °, the result is

U _BPB (f) W - if _Ab t - fcw) + S (f - if _Abt + fcw)] ^■

(20)

To be able to mix the center frequency f _C w in the zero position, the sampling frequency f _A bt can be selected so that the center frequency f _C w corresponds to an integer multiple of the sampling frequency f _Abt , ie

then the spectra of both channels will look like this. For channel A results

UBP _a (D = \ 2 ⁵ f ^~ ( ^£ + + S (f - (i - x) f _Abt )

^£ = - ^» (22) while for channel B

U _BPB (f) = \ XW ^~ (i + x) f _Ab t) + S (f - (i - x) f _Dept )} ^{■ β '} ^^ Ψ

i ^{= ~ m} (23) results. The channels contain the spectral components for the zero position for i = ± x. This results for channel A

(24) while for channel B

U _B (f = i <5 (- 0) ^■ _e -J ^'2nx f ^ + igf + 0) · β ~ ^ ^πχ ^ Μψ

² (25). It has been found that U _A (f) and U _B (f) are then out of phase with each other by 90 ° when the condition

TT

2nxf _Abt (p = - + 2π · m | m = integer

(26) is satisfied. Furthermore, it has been recognized that this condition is satisfied by an appropriate choice of the time offset φ, namely

4 x

(27)

Among other things, a time offset of ττ / 2 is obtained by operating the ADC at 4 times the sampling frequency. For example, the input signal or the received signal 105, A _R , can be decimated in two branches by a factor of 4, with one of the two branches first being delayed by one clock (compare the exemplary embodiment of FIG. 3). In this case, the time offset φ can be selected, for example, such that φ- 'Abbot

⁴ (28) applies and

1 + 4m

= 1

(29). According to embodiments, a suitable sampling frequency f _Ab t can be calculated from the following formula. This formula is

* ^' fcw

'Abbot with m = 1, 2, 3,4,,

(1 + 4m) (30)

In the following, exemplary embodiments of a time-shifted sampling are illustrated by means of exemplary diagrams (FIGS. 7a-d), the time-shifted sampling being based on the equation just mentioned. In other words, Figures 7a-d exemplify the provision of the at least one pair of samples 1 15 (A, B; A _{1 (} B) with the sample provider 110 of the embodiment of the receiver 100 of Figure 1 a.

Examples of time-shifted sampling can be found in "Attachment A" of Figs. 18a-f.

FIG. 7 a shows an exemplary embodiment of a time-offset sampling with the exemplary parameters f _cw = 1 MHz, m = 1 and f _Abt = 4/5 MHz = 800 kHz. In FIG. 7 a, an exemplary oversampled 1 MHz cosine signal with the phase 0 can be seen in the first diagram 710. The amplitude has been set to 1 by way of example. To represent the second diagram 720, the 1 MHz cosine signal was sampled at the sampling rate f _Abt = 800 kHz, wherein in the second diagram 720 a total of four samples with the exemplary values of [1 0 -1 0] can be recognized. The third diagram 730 and the fourth diagram 740 correspond to the signal decimated by a factor of 4, the imaginary part ("imag") being delayed by one clock. In the first to fourth diagrams 710, 720, 730, 740 of FIG. 7a, the 1 MHz cosine signal substantially corresponds to the received signal 105, A _R> while the four samples correspond to the exemplary one. essentially correspond to the at least one pair of samples 15 (A, B, Ai, Bj).

In embodiments, the phase position cp

be determined (see FIG., the embodiment of Fig. 6a), where B = "Imag" (imaginary part) and A = "Real" (real part), while the amplitude ä _R of empfange- NEN signal 105, A _R, and the magnitude of the complex value pair can be determined from a _R = ^ A ² + B ² .

FIG. 7b shows an exemplary embodiment of a time-shifted sampling with the exemplary parameters f _C w = 1 MHz, m = 2 and f _Abt = 4/9 MHz = 444, 44 kHz. Here, the 1 MHz cosine signal in the first diagram 710 of FIG. 7b substantially corresponds to the received signal 105, A _R , of FIG. 1a, while the four samples having the example values of [1 0 -1 0 ] in the second to fourth diagrams 720, 730, 740 of FIG. 7b substantially correspond to the at least one pair of samples 15 (A, B, Α, B,) of FIG. 1a.

FIG. 7 c shows an exemplary embodiment of a time-shifted sampling with the exemplary parameters f _C w -1 MHz, m = 6 and f _Abt = 4/25 MHz = 160 kHz. Here, the 1 MHz cosine signal in the first diagram 710 of FIG. 7c substantially corresponds to the received signal 105, A _R , of FIG. 1 a, while the four samples having the values of [1 0 -1 0] in the second to fourth graph 720, 730, 740 of FIG. 7c substantially correspond to the at least one pair of samples 1 1 5 (A, B; A,, B-1) of FIG. 1 a.

FIG. 7 d shows an exemplary embodiment of a time-shifted sampling with the exemplary parameters f _cw = 1 MHz, m = 1 and f _Abt = 4/5 MHz = 800 kHz and with an exemplary 45 ° phase shift. In this case, the 1 MHz cosine signal, which is phase-shifted by, for example, 45 °, corresponds in the first diagram 710 of FIG. 7d substantially. Chen the received signal 105, A _R, of Figure 1 a, while the four samples with the exemplary values. [I / 2 - I / V2 - I / V2 I / V2] in the second to fourth graph 720, 730, 740 7d substantially correspond to the at least one pair of samples 1 15 (A, B; A,, of FIG. 1a.

With reference to FIGS. 7a-d, it should be noted that the time interval T _{Abt of} the respective samples is shown there. Here, the time interval T _Abt corresponds to the sampling rate f _Abt according to the above-derived equation (Equation 30). Thus, according to exemplary embodiments, the time interval T _Abt or the sampling frequency f _{Abt can be set} as a function of m (m = 1, 2, 3, 4,...).

It has therefore been recognized that the time interval T _Abt or the sampling frequency f _Abt can be set flexibly, for example, depending on an ambient condition of the receiver. Embodiments of the present invention thus allow a more flexible adjustment of the sampling rate for a sample of the received signal in the receiver.

Furthermore, it has been recognized that if the frequency position of a bandpass signal is known, by choosing a particular sampling frequency, a direct down-conversion to zero can be achieved by the sample itself.

In the following, it will be explained with reference to the exemplary embodiments described above that different received signal parameters, such as the phase information cp _R or the amplitude information a _{R of} the received signal A _R and reflectivity information of the object p can be determined according to these embodiments.

Referring to the embodiment of FIG. 3 is located at the input of the ADC 1 12, a signal 105, A _R , whose frequency f _cw (or average period T _cw ) with the modulation frequency of the laser (or with the modulation frequency of one of a transmission signal A _T ) emitted from a transmitter, the signal 105, A _{R being provided} with an unknown phase position or phase information cp _R. As shown in Fig. 3, the digital signal 1 13 is divided after the scan with the ADC 1 12 into two branches. In the first branch, the sampled signal 1 13 is decimated by the factor 4 (block 312). The second branch is first delayed by one sample clock (T _Abt ) (block 322). and then decimated by the factor 4 (block 324). After the decimation, the clock of the pixel clock (f _dk ), for example, continues to be processed (see Fig. 3).

Referring to the embodiment of Fig. 6, from the at least one pair of samples 1 15 (A, B; AB,) and from the complex value pair (A, B), where A is the real part and B is the imaginary part of the complex value pair , the phase information 125, cp _R , of the received signal 105, A _R , and the phase angle of the detector signal, respectively, are calculated using the following equation

-1

φ _κ = tan (31)

Since the phase position of the laser intensity or the phase information cp _{T of} the signal A _T emitted by a transmitter is known, the phase difference Δφ can be calculated according to exemplary embodiments. In further embodiments, from this phase difference Δφ, the distance R to the object can be further calculated (see Equation 6).

In further embodiments, the amplitude information A _{R of} the received signal 105, A _R , or the amplitude of the signal reflected on an object may be calculated as the magnitude of the complex value pair (A, B) using the following equation

Furthermore, according to embodiments, conclusions about the reflectivity property of the object can be drawn from the amplitude of the detector signal ( _{R R} ).

Given in a LIDAR system, the optical structure and the sensitivity of the photodiode. Fig. 8a shows a schematic representation of a power distribution when detecting or scanning with a LIDAR system. FIG. 8 a shows exemplary equations for the power transmitted by a transmitter (eg aircraft), P _T , for the power received by the transmitter (eg aircraft), FIG.

the power received at an object (eg ground surface), M ^■ P _T , and the power ρ · Μ · Ρ _τ reflected at the object (eg ground surface)

2π given the assumption of a Lambertian surface. Here, P _{R is} the received light power, P _{T is} the transmitted light power, M is the atmospheric transmission, A _{r is} the surface of the receiving optics, p is the reflectivity of the surface and R is the distance to the surface.

If the equation for the power received by the transmitter from FIG. 8a is changed over to p, the result is

P _R 2n - R ²

^V P _T ^' M ² · A _r

(33)

The factor P _R / P _T can be equated here with the amplitude of the outgoing signal a _T and the received signal a _R. Thus, according to embodiments, the reflectivity of the object may be calculated by the following formula

_ _R 2π · R ²

P ^{~ ~} ä ^ ^' M ² · A _r

(34)

Fig. 9 shows a system 200 for measuring information on a distance to an object 129, _Δ φ, R. As shown in Fig. 9, the system 200 includes a transmitter 10, a receiver 100 and a control unit 210. The transmitter 10 includes is designed to transmit a transmission signal 15, A _T. The receiver 100 is designed to receive a received signal 05, AR. Here, the receiver 100 of the embodiment of FIG. 9 substantially corresponds to the receiver of the embodiment of FIG. 1 a. The control unit 210 is designed to have an average frequency 1 1, f _C w '_{> of} the transmitting signal 15, A _T , and the sampling rate 1 1 1, f _Abt , as a function of a clock frequency 205, f _dk , of the system 200 to control. In embodiments, the transmission signal 15, A _T, a sinusoidal signal a _T sin (27r _{CT f} t + _φ τ) where a _T and _φ τ the amplitude information and the phase information and f _C w 'the center frequency of the transmission signal 15, A _T , represent.

Further, it has been recognized that in the system 200, the functions A _T (t) and A _R (t) for the transmit signal 15 and the received signal 105 differ only in their amplitude information and their phase information. In this case, these functions A _T (t) and A _R (t) essentially do not differ in their frequency (fcw -fcw). In other words, a light signal (eg the transmission signal 15) is intensity-modulated with a band-limited function, so we also get a band limited signal (signal 105) at the detector.

In embodiments, the control unit 210 includes a center frequency transmitter 212 and a sample rate provider 214. In this case, the center frequency relay 212 is designed, for example, to control the average frequency 1 1, f _C w '_{> of} the transmission signal 15, A _T , as a function of the clock frequency 205, f _C | _k , to provide. Further, the sample rate generator 214 may be configured to provide the sampling rate 1 1 1, f _Abt , in response to the clock frequency 205, f _dk .

In embodiments, the control unit 210 is _configured to multiply the clock frequency 205, f _cik by a factor of 4 to provide the sampling rate 1 1 1, f _Abt .

In embodiments, the system 200 includes a clock frequency provider 201 for providing the clock frequency 205, f _dk , to the control unit 210. For example, the clock frequency provider 201 is configured to set the clock frequency 205, f _dk , for the control unit 210 based on a desired resolution (FIG. N _px ) of a scan area of the system 200 and at an oscillation frequency (f _s ) of a scan mirror of the system 200.

In embodiments, the receiver 100 comprises a parameter calculator 120 for calculating the phase information 125, cp _R , of the received signal 105, A _R , Ba. based on the at least one pair of samples 1 15 (A, B; Α ,, B,). Here, the parameter calculator 120 for the system 200 of FIG. 9 substantially corresponds to the parameter calculator 120 of Fig. 6. For example, the parameter calculator is configured 120 to the information on the distance to the object 129, _Δ φ, R, based on a phase information φ _{τ of} the transmission signal 15, A _T , and the phase information 125, cp _R , of the received signal 105, A _R.

For example, a phase difference Δφ results from a difference between the phase information φ _{τ of} the transmission signal 15 and the phase information cp _{R of} the received signal 125.

Further, the reflectivity information of the object p may be calculated based on the amplitude information a _{T of} the transmission signal 15 and the amplitude information a _{R of} the received signal 105 using Equation 34.

In embodiments according to FIG. 9, the receiver 100 has knowledge of the mean frequency 1 1, f _C w '_{> of} the transmission signal 15, A _T. For example, the parameter calculator 120 is configured to calculate the distance to the object R based on the average frequency of the transmit signal 15, A _T. In this case, for example, equation 6 can be used.

Figures 10a-c show further details of the system 200 with the control unit 210 according to further embodiments of the present invention. Here, the exemplary embodiments of FIGS. 10a-c include the elements already described above with the same reference numerals. In Fig. 10a a block diagram of a scanning system is shown. In the embodiment shown in Figure 10a, the receiver 00 further comprises a detector 1010 for providing a detection signal as an input signal 1015. Further, the receiver 100 in Figure 10a includes a filter 020 for filtering the input signal 1015, and the receive signal 105 is provided. For example, the filter 1020 ("AA") is an anti-aliasing filter designed to apply low-pass filtering to the input signal 1015. Further, the parameter calculator 120 in FIG. 10a includes two blocks 1022, 1024 configured to be the same

Phase information cp _R and the amplitude information a _{R of} the received signal 105 to deliver. Further, in Fig. 10a, the control unit 210 is called "Clock Gen". The system 200 of FIG. 10a further includes a control logic 1002 configured to control various parameters of the system 200. The control logic 1002 and the control unit 210 of FIG. 0a are in turn controlled by a mirror driver 1001 so that parameters of a scan mirror of the system 200 can be adjusted. The transmitter 10 of Figure 10a further includes a block 030 ("DAC") and a block 1040 ("Laser"). Here, block 1030 corresponds to a digital-to-analog converter (DAC), while block 1040 corresponds to a laser for transmitting the transmit signal.

Referring to Figure 10a, a simplified block diagram of a scanning LIDAR system is shown. For example, the phase <p _R corresponds to the result of equation 31, while the amplitude a _R corresponds to the result from equation 32, for example. Furthermore, the detector signal 1015 is band limited by, for example, a suitable antialiasing filter (AA). FIG. 0b shows a block diagram of an overall system. In Fig. 10b, the detector 1010 is, for example, an avalanche photodiode ("Avalanche Photo Diode", APD). The avalanche photodiode 1010 is configured to provide the input signal 1015. Further, the receiver 100 of FIG. 10b includes a block 1012 ("Analog Back End"). Block 1012 represents an analog interface for receiving received signal 105. Control unit 210 is shown in FIG. 10b as "LIDAR Core Clock Gen." designated. The control unit 210 shown in FIG. 10b sets the average frequency 1 1 ("f_cw"), the sampling rate 1 1 1, f _Ab t. and another frequency 1 1 1 1 ("fs_dac") as the control frequency for block 1030 (DAC). As shown in FIG. 10b, the control unit 210 may be controlled by a block 1001 ("Mirror IP") by means of a frequency f_reso _{1 2} . The block 1001 is designed, for example, to provide the frequency f_reso ₂ for the control unit 210 in order to realize a desired resolution of the scan area of the system. Further, the transmitter 10 shown in FIG. 10b includes an analog front end block 1032 as an interface between the DAC 1030 and the laser 1040. Further, the transmitter 10 in FIG. 0b includes a mirror 1050 controlled by the block 1001 can be. Here, the mirror 1050 and the block 1001 are connected to each other via a communication link 1055. For example, the oscillation frequency (f _s ) of the mirror 1050 may be adjusted with the block 1001. The mirror 1050 is designed to deflect the transmitted signal transmitted by the laser 1040 to an object. As shown in FIG. 10b, the transmitted signal deflected onto the object is reflected towards the avalanche photodiode 1010 after reflection from the object. 10c shows an embodiment of a control unit 210 of the embodiment of the system 200 according to FIG. 10b. FIG. 10b shows a block diagram of a "LI DAR Gore Clock Generator". As shown in FIG. 10c, the control unit 210 includes the center frequency provider 212 and the sample rate provider 214. Here, the blocks 212, 214 are configured to receive the signals 1 1 (f_cw), 1 1 1 (f _abb ), and 1 1 1 1 (FIG. fs_dac) depending on the input frequency f_reso _{1 2} . Further, it is shown in Fig. 10c that the blocks 212, 214 can be adjusted by a block labeled "register" by means of a control signal ("Resolution"). Embodiments of the present invention enable a static LIDAR measurement. In a static measurement (ie, the intensity-modulated laser radiates constantly in one direction), any useful frequency fcw for the intensity modulation can be selected according to embodiments. For example, using Equation 30, a sampling rate of the ADC 1 12 (f _A bt) at which the "intensity-modulated" detector signal (or the received signal 105) is mixed to the zero position by the sampling rate f _Abt may be calculated at this frequency. According to embodiments, therefore, a new value pair or the at least one pair of sample values 15 (A, B, Ai, B †) with a frequency of f _Ab t / 4 is available. This was illustrated by the example diagrams of Figures 7a-d (see also "Attachment A" of Figures 18a-f) It was recognized that the sample frequency of the measurement can be influenced by the factor m of Equation 30. Further recognized that the resolution of the measurement result can be influenced by the frequency f _C w, since with a lower frequency the modulated wavelength becomes larger and can thus be further measured in the distance.

Embodiments of the present invention provide a concept for time-shifted sampling, wherein the advantage of time-shifted sampling in a static LIDAR measurement lies in the freely selectable repetition time of a measurement (or in the freely selectable time interval T _Abt ). This repetition time or interval can be selected from the hour range to the microsecond range. By comparison, at a typical laser rangefinder, the duty cycle times are, for example, <0.5 seconds and a maximum of 4 seconds. Furthermore, according to embodiments, the intensity frequency (fcw ') can be varied freely. This has the further advantage that the measuring range can be varied. Additionally, according to embodiments, the required tasks may be performed by a very small μ- Controllers are done because no large calculations or complex data handling are required.

Further embodiments of the present invention provide a scanning LIDAR (1D / 2D) measurement. In a scanning system, the challenge is, depending on the scanning speed (Hz, kHz, ...) and the type of scanner (raster scanner, Lissajous scanner, etc.) and the resolution of the area to be scanned (see Fig. 8b), to accurately capture the phase of each pixel. FIG. 8b shows a scan area of a 2D scanner without adaptation (left) or with adaptive scan area adaptation (right). Embodiments provide a concept that is advantageous in that it is independent of the type and scanning speed of the scanner. For this example, a grid is thrown over the area to be scanned. The distances in the grid correspond to a pixel clock (or the clock frequency f _dk ). For example, after each bar of the pixel clock, the laser will irradiate the scanner with another area of the grid. The pixel clock is now equated, for example, with f _Abt / 4, ie a quarter of the sampling rate f _Abt of ADC 1 12. If, for example, the sampling rate f _A bt of ADC 1 12 is calculated from the pixel clock and equation 30 is changed to f _C w Thus, matching the sample rate, one obtains a frequency which automatically allows down-conversion to the zero position and each pixel clock clock receives a new complex value pair for a new pixel.

The prerequisite for a time-delayed scanning according to the invention lies in a scanning system in that the oscillation frequency of the scanner (or f _s ) is known, the spatial resolution of the scanner is known, the required resolution of the scan area (pixels in x- or in x / y direction or the parameter "resolution"), the intensity modulation is a sinusoidal function, the frequency of the intensity-modulated CW signal is known (f _C w) and the detector signal is band-limited.

From the given requirements, according to embodiments, on the one hand, a variable (frequency) output signal can be calculated. In this case, the frequency fcw, for example, from equation calculated. It consists of the frequency of the pixel clock f _p dk = 2 ^■ resolution ^■ f _{Mirror (36)}

(or f _c i _k ) provided with a factor. This factor results from the transformation of Equation 30.

Since the frequency of the intensity modulation is known, a clock can be generated in accordance with this frequency, with which the scanner of the ADC 1 12 can be operated. This clock can also be adjusted in the same way as the modulation frequency during operation and corresponds, for example, to four times the pixel clock fAbt = ⁴ * fpclk (37)

Increasing the m-factor in Equation 35 increases the modulation frequency of the CW signal (which corresponds to a shorter wavelength), which in turn allows more accurate phase determination. It is also possible by a position determination of the example MEMS mirror to perform a multiple measurement of the same pixel. Thus, by averaging the measured values over the same pixel, the accuracy of the phase determination can be further increased.

This measuring method according to the invention has the advantage, for example, in scanners that no data has to be stored for further signal processing (as is the case, for example, using FFT, digital filters, etc.), but that per pixel clock in the scan (f _Abt / 4 ) exactly one complex measuring pair (A + jB) exists. From this pair of measurements, phase shift and amplitude of the sampled signal can be calculated. From these values, further distance and brightness of the pixel can be determined. In Fig. 1 1, such a scan is simulated by way of example, in which the phase position of the detector signal varies linearly from pixel to pixel. The exemplary scan of FIG. 11 is explained in more detail below. In Fig. 11, the MATLAB script "Simulation - Timed Scan LI DAR-" of Fig. 18a-f was used. In the first diagram 1110, a cosine-shaped signal with a frequency of 68 MHz is sampled at 54.4 MHz by way of example. The respective samples can be seen in the first diagram 1 1 10 ("Function Values"). In the second and third diagrams 1 120, 1 130, the sampled signal was decimated by a factor of 4 or first delayed by one clock and then decimated ("Real Value", "Imag Value"). In the fourth diagram 1 140 the phase relationship of the sampled signal ("phase [deg]"). As can be seen in FIG. 11, these samples have a spacing of 1 / fclk. Thus, each pixel can be assigned a phase value, more specifically, an individual distance can be calculated for each pixel.

The concept just described is advantageous in that it can be used independently of the scanner used (1 D / 2D scanner, raster scanner / Lissajous scanner, etc.). According to embodiments, it is possible to vary the resolution during operation and to perform a multiple measurement of a pixel.

Other embodiments provide a concept for adaptive scan resolution. According to embodiments, the resolution of the scan area can be adjusted independently of the scanning method (static or scanning measurement). If, for example, the scan area is increased, this means that the grid will become narrower. This in turn means that the Pixeiclock (Equation 36) is increased in this range, thereby increasing the frequency of the CW signal (Equation 35) and the sampling rate of the ADC 1 12 (Equation 37). For example, if there is an area to be scanned with less high resolution, then the pixel block may be decreased in this area, resulting in the frequency of the CW signal (f _C w) and the sampling rate (f _A bt) of the ADC 1 12 also be reduced. This adaptive scan resolution according to the invention can be of interest, for example, in edge detections or edge inspections. Here, for example, an edge / column is detected in a first scan. Furthermore, in a subsequent scan in the area of the edge / column, the resolution can be increased in order to better resolve the edge / column.

Embodiments of the present invention provide a real-time evaluation algorithm that determines the distance and reflectivity of the area to be scanned in a CW method. The accuracy of the measured value can be steadily increased by averaging several measurements.

Applications of the embodiments described above are static measurements (eg methods for monitoring mechanical motion) and the scanning measurement. The static measurements include, for example, a Doppler measurement, while the scanning measurement comprises a variable adaptation to a scan frequency, an adaptive pixel resolution, a possible multiple measurement of a scan frequency. xels and 3D photography (inspection applications, biometric applications, ...).

In contrast to known methods, embodiments of the present invention provide a concept that is significantly more resource efficient. Similarly, no data collection (such as in an FFT) is needed for the inventive algorithm.

Furthermore, the present invention is advantageous in that, in contrast to known methods in which an FM demodulation is carried out for a CW signal, wherein this demodulation precedes the data acquisition, no further resources are needed.

It has been shown with reference to FIG. 16 that the demodulation is achieved, for example, by multiplication. Since the equivalent circuit of a scanner (see Fig. 6d) also contains a multiplier, it is possible to perform a quadrature demodulation by selecting a particular sampling frequency (f _Ab t). It has been recognized that for this purpose the multipliers from FIG. 16 can be replaced by ADC. It was also recognized that this eliminates the subsequent TP filtering.

According to embodiments, the computer program (algorithm) according to the invention can be realized for example with the simulation program MATLAB, as shown in FIG. Although some aspects have been described in the context of a device, it will be understood that these aspects also constitute a description of the corresponding method, so that a block or a component of a device is also to be understood as a corresponding method step or as a feature of a method step. Similarly, aspects described in connection with or as a method step also represent a description of a corresponding block or detail or feature of a corresponding device. Some or all of the method steps may be performed by a hardware device (or using a hardware device). Apparatus), such as a microprocessor, a programmable computer or an electronic circuit. In some embodiments, some or more of the most important method steps may be performed by such an apparatus. A signal according to the invention, such as a signal comprising the samples 15 (A, B; Ai, B), may be stored on a digital storage medium or may be on a transmission medium such as a wireless transmission medium or a wired transmission medium, eg Internet, to be transmitted.

The signal according to the invention may be stored on a digital storage medium, or may be transmitted on a transmission medium, such as a wireless transmission medium or a wired transmission medium, such as the Internet.

Depending on particular implementation requirements, embodiments of the invention may be implemented in hardware or in software. The implementation may be performed using a digital storage medium, such as a floppy disk, a DVD, a Blu-ray Disc, a CD, a ROM, a PROM, an EPROM, an EEPROM or FLASH memory, a hard disk, or other magnetic disk or optical memory are stored on the electronically readable control signals, which can cooperate with a programmable computer system or cooperate such that the respective method is performed. Therefore, the digital storage medium can be computer readable.

Thus, some embodiments according to the invention include a data carrier having electronically readable control signals capable of interacting with a programmable computer system such that one of the methods described herein is performed.

In general, embodiments of the present invention may be implemented as a computer program product having a program code, wherein the program code is operable to perform one of the methods when the computer program product runs on a computer.

The program code can also be stored, for example, on a machine-readable carrier.

Other embodiments include the computer program for performing any of the methods described herein, wherein the computer program is stored on a machine-readable medium. In other words, an embodiment of the The method according to the invention thus comprises a computer program which has a program code for performing one of the methods described herein when the computer program runs on a computer. A further embodiment of the inventive method is thus a data carrier (or a digital storage medium or a computer-readable medium) on which the computer program is recorded for carrying out one of the methods described herein. A further embodiment of the method according to the invention is thus a data stream or a sequence of signals, which represent the computer program for performing one of the methods described herein. The data stream or the sequence of signals may be configured, for example, to be transferred via a data communication connection, for example via the Internet.

Another embodiment includes a processing device, such as a computer or a programmable logic device, that is configured or adapted to perform one of the methods described herein.

Another embodiment includes a computer on which the computer program is installed to perform one of the methods described herein.

A further embodiment according to the invention comprises a device or a system designed to transmit a computer program for carrying out at least one of the methods described herein to a receiver. The transmission can be done for example electronically or optically. The receiver may be, for example, a computer, a mobile device, a storage device or a similar device. For example, the device or system may include a file server for transmitting the computer program to the recipient.

In some embodiments, a programmable logic device (eg, a field programmable gate array, an FPGA) may be used to perform some or all of the functionality of the methods described herein. In some embodiments, a field programmable gate array may cooperate with a microprocessor to perform one of the methods described herein. In general, in some embodiments, the methods are performed by any hardware device. This can be a universally applicable Hardware such as a computer processor (CPU) or hardware specific to the process, such as an ASIC.

One aspect of the invention is therefore configuration data for a programmable logic device or field programmable gate array (FPGA) for performing the method described herein when the configuration data is loaded into the programmable logic device or field programmable gate array.

The embodiments described above are merely illustrative of the principles of the present invention. It will be understood that modifications and variations of the arrangements and details described herein will be apparent to others of ordinary skill in the art. Therefore, it is intended that the invention be limited only by the scope of the appended claims and not by the specific details presented in the description and explanation of the embodiments herein.

Embodiments of the present invention provide a method for determining phase information 125, cp _R , of a received signal 105, A _R. The method comprises providing at least one pair of samples 1 15 (A, B; A _{1 t} Βτ), sampling the received signal 105, A _R , at a sampling rate 1 1 1, f _Abt less than a mean frequency 101 , f _C w, of the received signal 105, A _R , and providing two samples as a pair of samples 1 15 (A, B; A _{1 (} Bi) representing a time interval (T _Abt ) of

^ T _cw {4m + l)

or from

~ T _cw {4m + 3)

where m is an integer greater than or equal to (>) 1, and where T _C w is an average period of the received signal 105, A _R. Further embodiments of the present invention provide a method of measuring information about a distance to an object 129 (Δφ, R) with a system 200. The method comprises transmitting a transmit signal 15, A _T , with a transmitter 10, receiving a receive signal 105, A _R , having a receiver as described above, and controlling a mean frequency 1 1, W, of the transmission signal 15, A _T , and the sampling rate 1 1 1, f _Abt , in dependence on a clock frequency 205 , f _C | _{k of} the system 200.

Claims

Patent claims

A receiver (100) for determining phase information (125) (cp _R ) of a received signal (105) (A _R ), with the following features: a sample value provider (1 10) for providing at least one pair of sample values (1 15) (A , B; Α, , B , where the sample value provider (1 10) is designed to sample the received signal (105) (AR) with a sampling rate (1 1 1 ) (f _A bt) which is smaller than a medium frequency ( 101) (f _cw ) of the received signal (105) ( _AR ), wherein the sample value provider (1 10) is designed to provide two samples as a pair of samples (1 15) (A, B; Ai, B-, which have a time interval

~- T _cw (4m + \)

4

or from

T _cw (4m + 3)

have, where m is an integer greater than or equal to 1, and where T _cw is an average period of the received signal (105) ( _AR ), the sample provider (1 10) having an analog-digital converter (1 12) (ADC ) for sampling the received signal (105) ( _AR ) and a value pair determiner (1 14) for determining the at least one pair of sample values (1 15) (A, B;

Ai , B,) based on a sampling signal (1 13) sampled by the analog-to-digital converter (1 12) (ADC); wherein the analog-digital converter (1 12) (ADC) is designed to sample the received signal (105) ( _AR ) at the sampling rate (1 1 1) (f _A t>t), the value pair determiner (1 14) is designed to decimate (312) the sampling signal (1 13) sampled by the analog-to-digital converter (1 12) (ADC) with a decimation factor of 4 by a first sample value (A) of the at least one pair of sample values (1 1 5) (A, B; Α·, , B ); wherein the value pair determiner (1 14) is designed to delay (322) the sampling signal (1 13) sampled by the analog-to-digital converter (1 12) (ADC) by the time interval (T _A bt) and by one um to decimate (324) the time interval (T _A bt) delayed delay signal (323) with the decimation factor 4 in order to obtain a second sample value (B) of the at least one pair of sample values (1 15) (A, B; Α·, , Bi ) to obtain.

2. The receiver (100) according to claim 1, wherein the sample value provider (1 10) has an analog-digital converter (1 12) (ADC) for sampling the received signal (105) ( _AR ) and a value pair determiner (1 14). for determining a first pair (A, B) and a second pair (A^ B^ of the at least one pair of samples (1 15) (A, B; A^ B^ based on one of the analog-digital converter (1 12) (ADC). decimate (312) to obtain a first sample value (A) of the first pair (A, B) of the at least one pair of samples (1 1 5) (A, B; Ai , Β _Ί ); where the value pair determiner (1 14 ) is designed to delay (322) the sampling signal (1 13) sampled by the analog-digital converter (1 12) (ADC) by the time interval (T _Abt ) and by a time interval (T _Abt ) Delayed delay signal (323) with the decimation factor 4 to decimate (324) in order to a second sample value (B) of the first pair (A, B) of the at least one pair of samples (1 15) (A, B; A, , Bi ); wherein the value pair determiner (1 14) is designed to increase the sampling signal (1 13) sampled by the analog-to-digital converter (1 12) (ADC) by twice the time chen distance (T _Abt ) to delay (332) and to decimate (334) a delay signal (333) delayed by twice the time distance (T _Abt ) with the decimation factor 4. to obtain a first sample value (Ai) of the second pair (A-1, Bi) of the at least one pair of samples (1 15) (A, B; A, , B-ι); wherein the value pair determiner (1 14) is designed to delay (342) the sampling signal (1 13) sampled by the analog-to-digital converter (1 12) (ADC) by three times the time interval (T _Abt ) and by a to decimate (344) a delay signal (343) delayed by three times the time interval (T _Abt ) with the decimation factor 4 in order to obtain a second sample value (Β· of the second pair (A Bi) of the at least one pair of sample values (1 15 ) (A, B; A^ B^ to obtain.

The receiver (100) according to claim 2, wherein the value pair determiner (1 14) is designed to determine the first sample value (A- of the second pair (Α^ B- of the at least one pair of samples (1 15) (A, B; A-i , Bi) by a factor (-1) to multiply (335) in order to obtain a modified first sample value (A); the value pair determiner (1 14) being designed to determine the second sample value (B-i) of the second pair (Αι, B,) to multiply (345) the at least one pair of samples (1 15) (A, B; u Bi) by a factor (-1) in order to obtain a modified second sample (B).

The receiver (100) according to claim 3, wherein the value pair determiner (1 14) is designed to determine the first sample value (A) of the first pair (A, B) of the at least one pair of samples (1 15) (A, B; Au Bi) and the first sample value (Ai) of the second pair A of the at least one pair of samples (1 15) (A, B; Α, , Β _Ί ) to combine (510) to obtain a first combination value (C); wherein the value pair determiner (1 14) is designed to determine the second sample value (B) of the first pair (A, B) of the at least one pair of samples (1 15) (A, B; Au and the second sample value (B <) of the second pair (Au B,) of the at least combining (520) a pair of samples (1 15) (A, B; Α·, , Β·,) to obtain a second combination value (d);

The receiver (100) according to one of claims 1 to 4, wherein the receiver (100) has a parameter calculator (120) for calculating the phase information (125) (cp _R ) of the received signal (105) (A _R ) based on the at least one Pair of samples (1 15) (A, B; A, , B-,).

The receiver (100) according to any one of claims 1 to 5, wherein the parameter calculator (120) is designed to calculate the phase information (125) (cp _R ) of the received signal (105) (A _R ) using the equation cp _R = tan (B/A), where A and B are two samples of the pair of samples (1 15) (A, B; Αι , B^ which have the time distance (T _Abt ).

A system (200) for measuring information about a distance to an object (129) (Δφ, R), with the following features: a transmitter (10) for sending a transmission signal (15) (A _T ), a receiver (100) for receiving a reception signal (105) (A _R ) according to one of claims 1 to 4; and a control unit (210) for controlling a mean frequency (1 1 ) (f _cw ') of the transmission signal (15) (A _T ) and the sampling rate (1 1 1 ) (f _Abt ) depending on a clock frequency (205) ( f _clk ) of the system (200).

The system (200) according to claim 7, wherein the control unit (210) has a center frequency provider (212) and a sampling rate provider (214), wherein the center frequency provider (212) is designed to provide the center frequency (1 1 ) ( ) of the transmission signal (15) (A _T ) depending on the clock frequency (205) (f _dk ); wherein the sampling rate provider (214) is designed to provide the sampling rate (1 1 1 ) (f _Ab t) as a function of the clock frequency (205) (f _dk ); wherein the control unit (210) is designed to multiply the clock frequency (205) (f _dk ) by a factor of 4 to provide the sampling rate (1 1 1 ) (f _Ab t).

9. The system (200) according to claim 7 or 8, wherein the system (200) has a clock frequency provisioning unit (201) for supplying the clock frequency (205) (f _dk ) for the control unit (210); wherein the clock frequency provisioning unit (201) is designed to set the clock frequency (205) (f _c i _k ) for the control unit (210) based on a desired resolution (N _px ) of a scan area of the system (200) and on an oscillation frequency (f _s ) of a scanning mirror of the system (200).

10. The system (200) according to any one of claims 7 to 9, wherein the receiver (100) has a parameter calculator (120) for calculating the phase information (125) (cp _R ) of the received signal (105) (A _R ) based on the has at least one pair of samples (1 15) (A, B; Α·, , Bi); wherein the parameter calculator (120) is designed to calculate the information about the distance to the object (129) (Δφ, R) based on phase information (φ _τ ) of the transmission signal (15) ( _AT ) and the phase information (125) ( cp _R ) of the received signal (A _R ).

1 1. The system (200) according to claim 10, wherein the receiver (100) has knowledge of the mean frequency (1 1 ) ( ) of the transmission signal (15) (A _T ); wherein the parameter calculator (120) is designed to calculate the distance to the object (R) based on the mean frequency (f _cw ') of the transmission signal (15) (A _T ).

12. A method for determining phase information (125) (cp _R ) of a received signal (105) (A _R ), with the following steps:

Providing at least one pair of samples (1 15) (A, B; Α^ B,), sampling the received signal (105) (A _R ) at a sampling rate (1 1 1 ) (f AM . which is smaller than a medium frequency (101) (f _cw ) of the received signal (105) is ( _AR ),

Providing two samples as a pair of samples (1 15) (A, B; Αι , B^, which have a time distance (T _Abt ) of

T _cw (4m + \) or from

± T _cw {4m + 3)

have, where m is an integer greater than or equal to 1, and where T _C w is an average period length of the received signal (105) ( _AR ), a sample value provider (1 10) being used which has an analog-digital Converter (1 12) (ADC), which is used to sample the received signal (105) ( _AR ), and a value pair determiner (1 14), which is used to determine the at least one pair of sample values (1 15) (A, B ; AB,) is used based on a sampling signal (1 13) sampled by the analog-to-digital converter (1 12) (ADC); wherein the analog-to-digital converter (1 12) (ADC) is used to sample the received signal (105) ( _AR ) at the sampling rate (1 1 1) (f _Abt ), wherein the value pair determiner (1 14) is used to decimate (312) the sampling signal (1 13) sampled by the analog-digital converter (1 12) (ADC) with a decimation factor of 4 to obtain a first sample value (A ) to obtain the at least one pair of samples (1 15) (A, B; A <, B-); wherein the value pair determiner ( 1 14) is used to delay (322) the sampling signal ( 1 1 3) sampled by the analog-digital converter ( 1 1 2) (ADC) by the time interval (T _{A t} ) and by to decimate (324) a delay signal (323) delayed by the time interval (T _A M) with the decimation factor 4 in order to obtain a second sample value (B) of the at least one pair of sample values (1 15) (A, B; Ai, B -ι).

1 3. A method for measuring information about a distance to an object (1 29) (Δφ, R) with a system (200), the method comprising the following steps:

Sending a transmission signal (1 5) (A _T ) with a transmitter (1 0);

Receiving a received signal (105) (A _R ) with a receiver (1 00) according to one of claims 1 to 4; and

Controlling a mean frequency (1 1 ) (f _cw ') of the transmission signal ( 1 5) (A _T ) and the sampling rate (1 1 1 ) (f _Abt ) depending on a clock frequency (205) (f _dk ) of the system ( 200).

14. Computer program with a program code for carrying out the method according to claim 1 2 or 1 3, if the computer program runs on a computer.

15. Configuration data for a programmable logic module or for a field-programmable gate array (FPGA), for carrying out the method according to claim 1 2 or 1 3, when the configuration data is loaded into the programmable logic module or the field-programmable gate array.