WO2014016955A1 - Méthode de modulation et de démodulation optique et émetteur-récepteur optique - Google Patents

Méthode de modulation et de démodulation optique et émetteur-récepteur optique Download PDF

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WO2014016955A1
WO2014016955A1 PCT/JP2012/069143 JP2012069143W WO2014016955A1 WO 2014016955 A1 WO2014016955 A1 WO 2014016955A1 JP 2012069143 W JP2012069143 W JP 2012069143W WO 2014016955 A1 WO2014016955 A1 WO 2014016955A1
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Prior art keywords
phase
polarization
optical
signal
slip state
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PCT/JP2012/069143
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English (en)
Japanese (ja)
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吉田　剛
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三菱電機株式会社
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Priority to JP2014501343A priority Critical patent/JP5653561B2/ja
Priority to PCT/JP2012/069143 priority patent/WO2014016955A1/fr
Publication of WO2014016955A1 publication Critical patent/WO2014016955A1/fr

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    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B10/00—Transmission systems employing electromagnetic waves other than radio-waves, e.g. infrared, visible or ultraviolet light, or employing corpuscular radiation, e.g. quantum communication
    • H04B10/40—Transceivers
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/03—Shaping networks in transmitter or receiver, e.g. adaptive shaping networks
    • H04L25/03006—Arrangements for removing intersymbol interference
    • H04L25/03012—Arrangements for removing intersymbol interference operating in the time domain
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/32—Carrier systems characterised by combinations of two or more of the types covered by groups H04L27/02, H04L27/10, H04L27/18 or H04L27/26
    • H04L27/34—Amplitude- and phase-modulated carrier systems, e.g. quadrature-amplitude modulated carrier systems
    • H04L27/38—Demodulator circuits; Receiver circuits

Definitions

  • the present invention relates to an optical modulation / demodulation method and an optical transceiver, and particularly to an optical modulation / demodulation method and an optical transceiver using a digital coherent method.
  • BPSK binary phase-shift keying
  • OOK on-off keying
  • QPSK Phase-Shift Keying
  • a method for high-density wavelength multiplexing, a method is used in which the number of transmission bits per symbol is doubled by polarization multiplexing in which independent signals are assigned to two orthogonal polarization components.
  • a method of increasing the signal multiplicity and increasing the number of transmission bits per symbol such as QPSK or 16-value quadrature amplitude modulation (16 quadrature Amplitude Modulation: 16QAM).
  • QPSK and 16QAM are transmitted by assigning signals to the same phase axis (In-Phase axis: I axis) and quadrature phase axis (Quadrature-Phase axis: Q axis) in the optical transmitter.
  • Non-Patent Document 1 chromatic dispersion generated in a transmission line and polarization mode dispersion (Polarization-polarization-) are achieved by linear photoelectric conversion by synchronous detection and fixed, semi-fixed, and adaptive linear equalization by digital signal processing. It is possible to achieve excellent equalization characteristics and excellent noise tolerance against linear waveform distortion caused by Mode Dispersion (PMD).
  • PMD Mode Dispersion
  • an M-th power method (see, for example, Non-Patent Document 2) or a provisional determination type algorithm (for example, see Patent Document 1) has been used.
  • a slip of an angle (2 ⁇ / M) ⁇ N may occur when the carrier phase is restored under conditions where noise and waveform distortion are large, and a large-scale continuous error may occur (M: PSK phase).
  • M PSK phase
  • differential encoding / decoding has been generally used as a method for preventing phase slip.
  • the period count (cycle count) value is increased or decreased on the condition that the transition of the estimated carrier phase between symbols exceeds a positive or negative threshold value. It is disclosed that carrier phase estimation is accurately performed based on the period count value.
  • Patent Document 3 a reference signal (pilot signal) is inserted between data signals on the transmission side in coherent optical communication.
  • the carrier phase is estimated from the reference signal.
  • the carrier phase estimated based on the reference signal is linearly interpolated to guarantee the estimation accuracy of the carrier phase.
  • Patent Document 4 discloses that in coherent optical communication, a reference signal (SYNC burst) is inserted between data signals on the transmission side, and a continuous error that may occur due to phase slip is relieved by error correction.
  • phase slip can be generated by checking whether the decoding results of the same symbol decoded in each direction are inconsistent after overlapping and collating the two-way data decoding in the time series order and the time series reverse order. Is also detected and corrected by error correction.
  • Patent Document 2 has a problem that the detection accuracy of the phase slip is insufficient and a large-scale continuous error due to the missed phase slip is unavoidable.
  • the present invention has been made in order to solve the above-described problem, an optical modulation / demodulation method capable of preventing transmission quality deterioration due to phase slip, improving transmission quality, and improving noise immunity, and The purpose is to obtain an optical transceiver.
  • an optical transmission unit generates an optical signal having a signal point arrangement that is asymmetrical in each of the X polarization and the Y polarization, and multiplexes them.
  • the receiver simultaneously estimates the carrier frequency and the carrier phase of the received optical signal based on the digital signal using the X polarization and the Y polarization, and the estimated carrier frequency
  • a step of outputting a frequency / phase compensated digital signal for each phase slip state that can be taken for each symbol based on the carrier phase, and a likelihood for each phase slip state based on the frequency / phase compensated digital signal Calculating the frequency / phase compensated digital signal based on a preset threshold value for each phase slip state, and outputting a decoding result for each phase slip state, and the phase
  • an optical transmission unit generates an optical signal having a signal point arrangement that is asymmetrical in each of the X polarization and the Y polarization, and multiplexes them.
  • the receiver simultaneously estimates the carrier frequency and the carrier phase of the received optical signal based on the digital signal using the X polarization and the Y polarization, and the estimated carrier frequency
  • a step of outputting a frequency / phase compensated digital signal for each phase slip state that can be taken for each symbol based on the carrier phase, and a likelihood for each phase slip state based on the frequency / phase compensated digital signal Calculating the frequency / phase compensated digital signal based on a preset threshold value for each phase slip state, and outputting a decoding result for each phase slip state, and the phase
  • Embodiments of an optical transceiver and an optical modulation / demodulation method according to the present invention will be described below in detail with reference to the drawings.
  • the embodiment described below is an embodiment for embodying the present invention, and is not intended to limit the present invention to its category.
  • FIG. 1 is a diagram illustrating a configuration example of an optical transceiver according to Embodiment 1 of the present invention.
  • the optical transceiver according to the first embodiment includes an optical transmission unit 100, a transmission channel 200, and an optical reception unit 300.
  • the optical transmission unit 100 includes a light source 101, a symbol mapper 102, and an optical modulation unit 103.
  • the optical receiver 300 includes a light source 301, an optical demodulator 302, a waveform equalizer 303, a frequency / phase estimator 304 (X / Y combination), a decoder 305 (X / Y simultaneously), a phase slip
  • the state estimation unit 306 and the selection unit 307 are configured.
  • the optical transmitter 100 transmits an optical signal to a counterpart optical transceiver (not shown) via the transmission channel 200.
  • the optical receiving unit 300 receives an optical signal transmitted from a partner optical transceiver (not shown) via the transmission channel 200.
  • the counterpart optical transmitter / receiver is not shown in FIG. 1, but has the same configuration as the optical transmitter / receiver shown in FIG.
  • the light source 101 generates a carrier light signal that oscillates at a predetermined wavelength set in advance, and outputs the carrier wave signal to the light modulator 103.
  • the symbol mapper 102 performs signal point arrangement based on transmission data (two series of codes A and B, see FIG. 4) input from the outside (not shown), and outputs a 4-lane binary electrical signal.
  • FIG. 2 is a signal point arrangement in the polarization phase space of the polarization-multiplexed BPSK (DP-BPSK: Dual-Polarization BPSK) method.
  • the horizontal axis represents the optical phase of X polarization, which is one polarization component when orthogonal polarization multiplexing is performed, and the vertical axis is the optical phase of the Y polarization, which is the other polarization component when orthogonal polarization multiplexing is performed.
  • Ordinary BPSK is represented by binary values of phase 0 and phase ⁇ . In FIG. 2, these are indicated by phase ⁇ / 4 and phase -3 ⁇ / 4 obtained by rotating them by ⁇ / 4.
  • each of the X polarization and the Y polarization takes a binary phase, four signal points (symbols) 0, 1, 2, and 3 can be taken in FIG.
  • the arrangement of these signal points is symmetrical (point symmetry, line symmetry) between the X polarization and the Y polarization.
  • FIG. 3 is a signal point arrangement in the polarization phase space according to Embodiment 1 of the present invention.
  • the signal point arrangement is an asymmetric arrangement between the X polarization and the Y polarization, and the signal point arrangement differs between the X polarization and the Y polarization.
  • the phase of X polarization is ⁇ 3 ⁇ / 4 (signal points 1 and 3), from the signal point arrangement of FIG. Shift the phase by + ⁇ / 2.
  • the phase of the Y polarization is the same as the signal point arrangement in FIG.
  • the arrangement of the signal points 0, 1, 2, and 3 is neither point-symmetric nor line-symmetric.
  • the signal point arrangement of both X / Y polarization is binary phase shift keying, and one side phase of one binary phase shift keying of X / Y polarization is performed. Only in this case, the other signal point of the X / Y polarization is shifted by ⁇ / 2.
  • the arrangement of the signal points is asymmetric between the X polarization and the Y polarization.
  • the symbol mapper 102 receives two series of codes A and B as transmission data from the outside (not shown), and is a 4-lane binary (+/ ⁇ ) electrical signal XI, XQ. , YI, YQ are generated and output to the light modulation unit 103.
  • the Y polarization is apparently QPSK, but the QPSK side is not gray-coded, as can be seen from FIG. 3, between 0-1 and 0-3 rather than between symbols 0-2 and 1-3. Since the polarization phase space distance is longer between 2-1 and 2-1 and 2-3, this is due to the intention of increasing the hamming distance on the far side.
  • This processing can be realized by table processing in which A and B are input and XI, XQ, YI, and YQ are output. Alternatively, it can be obtained by the following logical operation.
  • the symbol mapper 102 generates a four-lane binary signal with XI “+”, XQ “+”, YI “+”, and YQ “+”.
  • the symbols 1, 2, and 3 and the codes A and B shown in FIG. 4 are inputted, respectively, and based on them, a binary signal of 4 lanes shown in FIG. 4 is generated.
  • the optical modulation unit 103 modulates the carrier optical signal input from the light source 101 with the 4-lane binary electrical signals (XI, XQ, YI, YQ) input from the symbol mapper 102, multiplexes them, and FIG.
  • An optical signal having the signal point arrangement shown in FIG. A signal (binary electrical signal) input from the symbol mapper 102 is generally amplified by a driver (not shown) to drive the light modulator 103, but the driver is omitted in the configuration of FIG. Yes.
  • the optical modulation unit 103 divides the carrier wave input from the light source 101 into two systems, respectively performs quadrature phase (I / Q) modulation with the binary electric signal from the symbol mapper 102, and performs orthogonal polarization multiplexing.
  • the transmission unit 100 may be provided with a digital / analog conversion unit (not shown) to perform digital signal processing such as predistortion of waveform distortion at the transmission end.
  • the transmission channel 200 transmits an optical signal input from the optical modulation unit 103 in the optical transmission unit 100 and outputs the optical signal to the optical reception unit 300 of the counterpart optical transceiver.
  • the transmission channel 200 includes devices and components generally used for transmitting optical signals, such as a wavelength multiplexing / demultiplexing device, an optical amplifying device, and a transmission line optical fiber.
  • the light source 301 generates a local oscillation optical signal that oscillates at a predetermined wavelength set in advance, and outputs the local oscillation optical signal to the optical demodulation unit 302.
  • the local oscillation optical signal is oscillated at substantially the same wavelength as the carrier optical signal generated by the light source 101.
  • the optical demodulator 302 decomposes the optical signal transmitted via the transmission channel 200 and the local oscillation optical signal input from the light source 301 into orthogonal polarization components and orthogonal phase components. Next, the optical demodulator 302 performs coherent detection that mixes the decomposed optical signal and the local oscillation optical signal, performs photoelectric conversion, and converts the signal into a four-lane electrical signal. Further, these electric signals are converted into digital signals by analog / digital conversion, and quantized and sampled. The 4-lane digital signal thus obtained is output to the waveform equalization unit 303. In the analog-digital conversion, quantization is usually performed with a resolution of 6 bits or more, and sampling is performed at a sampling rate that is twice or more the baud rate.
  • the waveform equalization unit 303 receives a 4-lane digital signal from the optical demodulation unit 302 and generates waveform distortion caused by chromatic dispersion, polarization rotation, polarization mode dispersion, fiber nonlinear optical effect, or the like generated in the transmission channel 200. And a 4-lane digital signal separated by orthogonal polarization is output to the frequency / phase estimator 304.
  • the frequency / phase estimation unit 304 receives 4-lane digital signals from the waveform equalization unit 303 and combines them between orthogonal polarizations (between X / Y polarizations) to estimate and compensate for the carrier frequency / phase. . That is, the frequency / phase estimation unit 304 compensates for the center frequency difference between the carrier optical signal and the local oscillation optical signal, which is present in the 4-lane digital signal input from the waveform equalization unit 303, and the M multiplication method. Also, carrier frequency estimation and carrier phase estimation are performed based on the provisional determination method and the like, and the 4-lane digital signal after frequency / phase compensation is output to the decoding unit 305 and the phase slip state estimation unit 306. Here, the carrier frequency estimation and the carrier phase estimation are not performed independently between the orthogonal polarizations, but are performed simultaneously by combining between the orthogonal polarizations (between the X / Y polarizations).
  • the phase slip state includes (1) a state in which the phase is not slipped in both X / Y polarization, and (2) X / Y polarization.
  • There can be a total of four states: a state where both slip +90 degrees, a state where (3) both X / Y polarizations slip 180 degrees, and a state where both (4) X / Y polarizations slip -90 degrees. If the carrier frequency and phase are estimated independently for X / Y polarization, slips will occur independently for each polarization. Therefore, the slip does not fit in these four ways, but the square of 4 (4 2 16), and phase slip detection described later becomes difficult.
  • FIG. 5 shows signal point arrangements after frequency / phase compensation for four types of phase slip states obtained by combining with X / Y polarization and simultaneously performing carrier frequency / phase estimation.
  • the phase slip state State-0 is when there is no phase slip
  • the phase slip state State-1 is when the phase slip +90 degrees
  • the phase slip state State-2 is when the phase slip +180 degrees
  • the phase slip state State- 3 is a case where the phase slip is -90 degrees.
  • the signal points do not overlap between the phase slip states, and the signal point arrangement is unique for each phase slip state.
  • these signal point arrangements are signal point arrangements that cannot be taken when there is no phase slip, it is possible to detect and compensate for a phase slip state.
  • the decoding unit 305 performs decoding for all possible phase slip states for each symbol. That is, the decoding unit 305 decodes the 4-lane digital signal input from the frequency / phase estimation unit 304 based on the threshold value indicated by the dotted line in the signal point arrangement of each State in FIG. Is output to the selection unit 307.
  • FIG. 6 illustrates a configuration example of the decoding unit 305.
  • the decoding unit 305 includes four two-dimensional phase identifiers (2-D2-Phase Slicers) 500, 501, 502, and 503.
  • the 4-lane digital signals input from the frequency / phase estimation unit 304 are equally input to the two-dimensional phase identifiers 500, 501, 502, and 503.
  • the two-dimensional phase discriminator 500 performs decoding for each 4-lane digital signal corresponding to State-0, and outputs the decoding result to the selection unit 307 (see FIG. 1).
  • the two-dimensional phase identifier 501 performs decoding for each signal of the 4-lane digital signal corresponding to State-1, and outputs the decoding result to the selection unit 307.
  • the two-dimensional phase discriminator 502 performs decoding for each signal of the 4-lane digital signal corresponding to State-2, and outputs the decoding result to the selection unit 307.
  • the two-dimensional phase discriminator 503 performs decoding for each digital lane signal corresponding to State-3 and outputs the decoding result to the selection unit 307.
  • Symbols Z ′, Z ′′, and Z ′′ ′′ are decoded as symbols Z, respectively.
  • a soft value may be passed to the selection unit 307 as a soft decision to give reliability information according to the distance from the signal point center in the polarization phase space.
  • the phase slip state estimation unit 306 calculates likelihoods for all possible phase slip states for each signal (symbol), compares them, and estimates which phase slip state has the maximum likelihood. That is, the phase slip state estimation unit 306 estimates the maximum likelihood phase slip state based on the 4-lane digital signal input from the frequency / phase estimation unit 304, and outputs it to the selection unit 307.
  • FIG. 7 illustrates a configuration example of the phase slip state estimation unit 306.
  • the phase slip state estimation unit 306 includes four likelihood generation units (MetricMeGenerators) 600, 601, 602, and 603 and one maximum likelihood state estimation unit (Maximum Likelihood State Estimator) 610.
  • the 4-lane digital signals input from the frequency / phase estimation unit 304 are equally output to the likelihood generation units 600, 601, 602, and 603.
  • the likelihood generation unit 600 calculates the likelihood that the phase slip state is State-0, and outputs the likelihood information to the maximum likelihood state estimation unit 610.
  • the likelihood generation unit 601 calculates the likelihood that the phase slip state is State-1, and outputs the likelihood information to the maximum likelihood state estimation unit 610.
  • the likelihood generation unit 602 calculates the likelihood that the phase slip state is State-2, and outputs the likelihood information to the maximum likelihood state estimation unit 610.
  • the likelihood generation unit 603 calculates the likelihood that the phase slip state is State-3, and outputs the likelihood information to the maximum likelihood state estimation unit 610.
  • the maximum likelihood state estimation unit 610 is based on the likelihood information of each phase slip state State-0, State-1, State-2, and State-3 input from the likelihood generation units 600, 601, 602, and 603.
  • the most likely phase slip state is obtained from the slip states State-0, State-1, State-2, and State-3, and is output to the selection unit 307 (not shown in FIG. 7).
  • FIG. 8 illustrates a configuration example of the likelihood generation units 600, 601, 602, and 603 in the phase slip state estimation unit 306. Since the likelihood generation units 600, 601, 602, and 603 each have the same functional block, the case of the likelihood generation unit 600 that calculates the likelihood of State-0 will be described here.
  • the likelihood generation unit 600 includes four polarization phase distance calculation units 700, 701, 702, and 703, one selection unit 710, n delay units 720A, 720B,. , 721X and one adder (sum) 722.
  • the delay units 720A, 720B,..., 720X are connected in series as shown in FIG.
  • the multipliers 721A, 721B,..., 721X are connected to delay units 720A, 720B,.
  • the 4-lane digital signals input from the frequency / phase estimation unit 304 are equally output to the polarization phase distance calculation units 700, 701, 702, and 703.
  • the polarization phase space distance calculation unit 700 applies the polarization phase from the central signal point 0 to the input 4-lane digital signal based on the State-0 signal point arrangement rule in which there is no phase slip. Find the spatial distance. For example, the sum of squares of the phase distance D_ph_x (0) in the X polarization direction and the phase distance D_ph_y (0) in the Y polarization direction ((D_ph_x (0)) 2 + (D_ph_y (0)) 2 ) is calculated. The result is output to the selection unit 710.
  • the polarization phase space distance calculation unit 701 obtains the polarization phase space distance from the signal point 1 based on the signal point arrangement rule of State-0, which is in a state without phase slip, for the 4-lane digital signal. For example, the sum of squares ((D_ph_x (1)) 2 + (D_ph_y (1)) 2 ) of the phase distance D_ph_x (1) in the X polarization direction and the phase distance D_ph_y (1) in the Y polarization direction is calculated. The result is output to the selection unit 710.
  • the polarization phase space distance calculation unit 702 obtains the polarization phase space distance from the signal point 2 based on the signal point arrangement rule of State-0, which is in a state without phase slip, for the 4-lane digital signal. For example, the sum of squares ((D_ph_x (2)) 2 + (D_ph_y (2)) 2 ) of the phase distance D_ph_x (2) in the X polarization direction and the phase distance D_ph_y (2) in the Y polarization direction is calculated. The result is output to the selection unit 710.
  • the polarization phase space distance calculation unit 703 obtains the polarization phase space distance from the signal point 3 based on the signal point arrangement rule of State-0, which is in a state without phase slip, for the 4-lane digital signal. For example, the sum of squares ((D_ph_x (3)) 2 + (D_ph_y (3)) 2 ) of the phase distance D_ph_x (3) in the X polarization direction and the phase distance D_ph_y (3) in the Y polarization direction is calculated. The result is output to the selection unit 710.
  • the selection unit 710 selects the minimum polarization phase space distance from the four polarization phase space distances input from the polarization phase space distance calculation units 700, 701, 702, and 703, and outputs the selected polarization phase space distance to the delay unit 720A. To do.
  • the delay unit 720A holds the digital signal input from the selection unit 710 for one symbol time, and outputs the digital signal to the delay unit 720B and the multiplication unit 721A.
  • Multiplication unit 721A outputs the result of multiplying the digital signal input from delay unit 720A by coefficient C [0] to addition unit 722.
  • the delay unit 720B holds the digital signal input from the delay unit 720A for one symbol time, and outputs the digital signal to the delay unit 720C (not shown) and the multiplication unit 721B connected in series to the delay unit 720B.
  • the multiplier 721B outputs the result of multiplying the digital signal input from the delay unit 720B by the coefficient C [1] to the adder 722.
  • n-th delay unit 720X holds the digital signal input from the (n ⁇ 1) -th delay unit 720W (not shown) for one symbol time, and outputs the digital signal to the multiplication unit 721X. .
  • the multiplication unit 721X outputs the result of multiplying the digital signal input from the delay unit 720X by the coefficient C [n ⁇ 1] to the addition unit 722.
  • the adding unit 722 calculates the sum of the n multiplication results input from the multiplying units 721A, 721B,..., 721X, and outputs the value of the sum to the maximum likelihood state estimating unit 610 (see FIG. 7).
  • the smaller the sum value the higher the likelihood.
  • the delay units 720A, 720B,..., 720X, the multipliers 721A, 721B,..., 721X, and the adder 722 constitute a finite impulse response filter.
  • the instantaneous likelihood at the time is averaged. Therefore, delay units 720A, 720B,..., 720X, multiplication units 721A, 721B,..., 721X, and addition unit 722 constitute an averaging unit that averages the signal output from selection unit 710. is doing.
  • the filter constituting the averaging unit is not limited to a finite impulse response filter, and a forgetting infinite impulse response filter may be used. In that case, maximum likelihood sequence estimation and posterior probability maximization can be performed more complicatedly.
  • the same processing as the processing in the likelihood generation unit 600 is performed. Here, the description is omitted.
  • the maximum likelihood state estimation unit 610 includes a total sum of phase slip states State-0, State-1, State-2, and State-3 input from the likelihood generation units 600, 601, 602, and 603,
  • the phase slip state having the smallest sum value is obtained as the “maximum likelihood phase slip state” and is output to the selection unit 307 (see FIG. 1).
  • the selection unit 307 compares the four decoding results input from the decoding unit 305 based on the phase slip state estimated by the phase slip state estimation unit 306, and corresponds to the phase slip state from those decoding results.
  • the decoding result to be selected is selected as the maximum likelihood decoding result, and is output to an external device (not shown) connected to the optical transceiver.
  • the simulation of the bit error rate characteristic when this embodiment is used was performed. Simulation conditions are shown in FIG.
  • the bit rate was 128 Gb / s, and the 11-stage pseudo-random binary sequence was used as the code sequence.
  • the number of symbols was 16384, and the calculation was repeated with the noise pattern changed 100 times.
  • the transmission line was an additive white Gaussian Noise (AWGN) channel, and the noise bandwidth when defining the optical signal power to noise power ratio was 0.1 nm.
  • AWGN additive white Gaussian Noise
  • the phase noise between the carrier optical signal and the local oscillation optical signal was 0, and the frequency difference between the carrier optical signal and the local oscillation optical signal was 0. Polarization demultiplexing is ideally performed.
  • the fourth power method was used, the phase errors detected in each polarization were averaged, and equal phase compensation was performed for each polarization.
  • the averaging of the carrier phase estimation was a 17-symbol moving average, and an unwrapping process was performed so that the phase change from the carrier phase one symbol before was within ⁇ ⁇ / 4.
  • the averaging of the phase slip state estimation was performed by a 17 symbol moving average. After obtaining the code error rate, it was converted into a Q value through a complementary error function and evaluated.
  • Figure 10 shows the simulation results.
  • a DP-DE (Differential Encoded) QPSK signal obtained by applying differential encoding / decoding to the DP-QPSK method and a case where there is no phase slip (No Slip) in the DP-QPSK signal were also calculated.
  • the horizontal axis is OSNR
  • the vertical axis is Q value (Q-factor).
  • the Q value of the present embodiment in the region where the Q value is 9 dB or more, a Q value equivalent to that without slip is obtained, and good characteristics are exhibited.
  • the Q value of the present embodiment is lower than the Q value without the phase slip, but the DP-DEQPSK signal However, the Q value still shows an excellent Q value of 0.5 dB or more.
  • signal point arrangement that is asymmetrical between orthogonally polarized waves as shown in FIG. 3 is performed in optical transmitter 100, and carrier frequency / phase estimation is performed between orthogonally polarized waves in receiver 300.
  • the phase slip can be detected by carrying out simultaneously. Thereby, the phase slip state can be estimated, and the most likely slip state can be estimated.
  • phase slip compensation can be performed by selecting a decoding result based on a decoding rule corresponding to the estimated slip state.
  • phase slip state can be detected and the phase slip compensation can be performed. Therefore, the phase slip state can be estimated on a signal basis without using differential encoding / decoding and without increasing the redundancy. Also, decoding according to the estimated slip state can be performed. As a result, transmission quality deterioration due to phase slip can be avoided, noise tolerance can be improved, and transmission quality can be improved.
  • the optical modulation / demodulation method according to the present invention is useful for an optical transmission system using a digital coherent method, and is particularly suitable for an optical transmission system with a low code error rate.

Abstract

L'invention concerne un émetteur-récepteur optique doté d'une unité d'émission optique (100) et d'une unité de réception optique (300), dans lesquelles les points de signal d'un signal optique émis sont distribués asymétriquement sur la polarisation X et la polarisation Y par l'unité d'émission optique (100) et la fréquence et la phase d'une onde porteuse d'un signal optique reçu sont estimées simultanément entre la polarisation X et la polarisation Y par l'unité de réception optique (300), ce qui détecte le glissement de phase. Comme la distribution des points de signal du signal optique reçu est spécifique de l'état de glissement de phase, il est possible d'estimer l'état de glissement de phase et d'effectuer le décodage en fonction de l'état de glissement de phase estimé pour chaque point de signal.
PCT/JP2012/069143 2012-07-27 2012-07-27 Méthode de modulation et de démodulation optique et émetteur-récepteur optique WO2014016955A1 (fr)

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JP2015204572A (ja) * 2014-04-15 2015-11-16 富士通株式会社 通信システム、受信装置および半導体装置
CN111095823A (zh) * 2017-08-31 2020-05-01 三菱电机株式会社 光收发装置和光收发方法
CN112889250A (zh) * 2018-11-29 2021-06-01 日本电信电话株式会社 接收装置、接收方法和程序

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JP2015204573A (ja) * 2014-04-15 2015-11-16 富士通株式会社 通信システム、受信装置および半導体装置
JP2015204572A (ja) * 2014-04-15 2015-11-16 富士通株式会社 通信システム、受信装置および半導体装置
CN111095823A (zh) * 2017-08-31 2020-05-01 三菱电机株式会社 光收发装置和光收发方法
CN111095823B (zh) * 2017-08-31 2023-03-24 三菱电机株式会社 光收发装置和光收发方法
CN112889250A (zh) * 2018-11-29 2021-06-01 日本电信电话株式会社 接收装置、接收方法和程序

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