WO2013117049A1

WO2013117049A1 - Self-oscillatory flyback converter

Info

Publication number: WO2013117049A1
Application number: PCT/CN2012/074151
Authority: WO
Inventors: 郭国文; 尹向阳
Original assignee: 广州金升阳科技有限公司
Priority date: 2012-02-07
Filing date: 2012-04-16
Publication date: 2013-08-15
Also published as: CN103248238A

Abstract

A self-oscillatory flyback converter is disclosed. The direct-current input signals, from a power input terminal of the self-oscillatory flyback converter input, turns into direct-current output signals after passing through an input filter circuit (11), a master power circuit (14) and an output filter circuit (16) in turn. The master power circuit (14) includes a master power tube (TR1) and a master transformer (T1).The direct-current output signals controls the master power tube (TR1) with negative feedback to stable the output via a voltage-stabilizing circuit (17), an isolating optical coupler (OC1) and a drive control circuit (13). The self-oscillatory flyback converter also includes a soft-start circuit (12) with a constant current source. The soft start circuit (12) is connected between the output terminal of the input filter circuit (11) and the drive control circuit (13). The self-oscillatory flyback converter achieves the soft-start function while providing suitable drive compensation in the startup phase and steady state phase of the product, improving the capacity of resisting disturbance and stability, and expanding the design range of the input voltage of the self-oscillatory flyback converter and the variation range of the load.

Description

Self-oscillating flyback converter

The present invention relates to a self-oscillating flyback converter, and more particularly to a self-oscillating flyback converter with a constant current source soft start circuit. Background technique

The Ringing Choke Converter is favored by designers because of its low design cost and strong market competitiveness. However, its output characteristics depend largely on discrete components. The consistency of the device, when the consistency of the transformer, main power tube, control transistor, optocoupler and other components is better, the output performance is more stable. Therefore, reducing the stringent requirements of the self-oscillating flyback converter circuit (RCC) for discrete device uniformity and improving its output voltage stability have become the focus of designers. At present, designers often use offset compensation circuits (control the base of the triode to add turn-off compensation), and the stability of the output voltage is improved to some extent. However, the scheme is difficult to debug, and the consistency of the product is still poor, especially in the output. The effect is more obvious when the power is increased and the output capacitive load is increased.

A preferred circuit form of a self-oscillating flyback converter (RCC) is disclosed in the Chinese Patent Application Publication No. CN101997423A. As shown in FIG. 1, the self-oscillating flyback converter mainly includes a filtering part and a soft start part. , MOS tube, transformer, pulse frequency modulation section (PFM), auxiliary power supply, isolated optocoupler, regulated output loop section. The input power is connected to the output circuit part through the transformer, the soft start part is connected to the gate of the MOS tube, the gate of the MOS tube is also connected to the pulse frequency modulation part, the auxiliary power supply is connected to the pulse frequency modulation part, the pulse frequency modulation part and the regulated output circuit part are The indirect reference amplification section and the isolated optocoupler form a voltage negative feedback loop.

The soft start circuit of the flyback converter improves the stability of the output voltage at the start of the product. The working principle is as follows: After the input is powered on, the voltage of the soft start circuit rises, the MOS transistor turns on, and the output voltage gradually rises. Before entering the steady state, the soft start circuit continues to provide driving energy for the MOS tube, and the voltage of the soft start circuit is gradually reduced. When the system is working normally, since the charging time of the soft start circuit is much longer than the discharge time, the soft start circuit is always kept at a lower potential. (When starting up, the soft-start voltage waveform is shown in Figure 7). The disadvantages of this soft-start circuit are: At steady state, the capacitor C9 in the soft-start circuit discharges much faster than the charging speed, and its ground-to-ground voltage is close to IV (see Figure 7), so the MOS tube drive is completely dependent on the feedback winding energy. The amount of soft-start circuit is insufficient for MOS tube compensation during normal operation. The output performance of the product is easily affected by the transformer process (feedback and input coupling degree), MOS tube conduction threshold, and output external capacitive load. When the charging speed is increased, due to the wide input voltage range and large variation of the output load, over-compensation is likely to occur, resulting in shortcomings such as power-on startup current limit and product output short-circuit power consumption.

2 is a circuit schematic diagram of an embodiment disclosed in the Chinese Patent Application Publication No. CN101997423A, which includes an input filter circuit 11, a soft start circuit 12, a pulse frequency modulation circuit 13, a main power circuit 14, an auxiliary power supply circuit 15, and an output filter. The circuit 16, the isolated optocoupler OCl, and the error amplifying ADJ circuit 17. The input filter circuit 11 is composed of a filter capacitor C0, a filter capacitor C1 and a filter inductor L0 to form a π-type filter circuit. Capacitor CO is connected to the input end of the power supply, and the other end is grounded. The inductor L0 is connected to the input end of the power supply, and the other end is connected to the same end of the transformer. One end of the capacitor C1 is connected to the same name end of the transformer, and the other end is grounded. Other existing filter circuits can also be used, which can be selected in accordance with the relevant technical manual. The soft start circuit 12 includes: a voltage dividing resistor R10, a resistor R13, a resistor R14, and a starting capacitor C9. The resistor R10, the resistor R13, and the resistor R14 are connected in series, one end of the series circuit is connected to the power input end, and one end is grounded. One end of the capacitor C9 is connected to the series junction of the resistor R10 and the resistor R13, and the other end is grounded. The working principle is that when the input voltage is connected, the current charges the capacitor C9 through the resistor R10, after the time ^ -/^^ ^^-^^

The Vin capacitor voltage reaches the MOS threshold voltage to achieve the soft-start function. When the MOS transistor is turned off, the feedback winding is reversed, and the potential of the same name is negative. Therefore, the potential of the MOS transistor is also negative. At this time, the capacitor C9 is quickly discharged through the resistor R13. When the MOS transistor is turned on, the input voltage is charged to the capacitor C9 through the resistor R10, because the resistance of the resistor R10 is far greater than the resistance of the resistor R13 (when the value of R10 is decreased, the power consumption of the product output short circuit increases, and the starting current increases, Table 2 exemplifies the shortcomings caused by the decrease in the value of R10. Therefore, the discharge speed is much faster than the charging speed. After a period of time, the energy of the capacitor C9 is released. The pulse frequency modulation circuit 13 includes: a resistor R6, a resistor R9, a resistor R11, a resistor R111, a capacitor C5, a capacitor C6, a capacitor C12, an NPN transistor TR2, a PNP transistor TR3, and a positive feedback winding P3. The capacitor C6 is connected in series with the resistor R11, the resistor R111, and the resistor R6. One end of the series circuit is connected to the same name end of the positive feedback winding P3, and the other end is connected to the collector of the transistor TR2. The capacitor C5 is connected in parallel with the resistor R9. One end of the parallel circuit is connected to the base of the transistor TR2, and the other end is connected to the collector of the transistor TR3. The capacitor C12 is connected in parallel with the resistor R111. One end of the parallel circuit is connected to the emitter of the transistor TR3, and the other end is connected to the collector of the transistor TR3. Its working principle is: positive feedback winding P3, capacitor C6, resistor Rll branch pass and The main power tube TR1 is coupled to the primary and secondary sides to form a self-excited oscillation circuit, and the control switch tube is turned on and off. At the same time, the current loop R5 branch and the voltage loop optocoupler branch are regulated by the double-tube drive control circuits TR2 and TR3. The duty cycle makes the product output normal. The main power circuit 14 includes: a transformer primary winding P1, an output winding P2, a MOS transistor TR1, and an output rectifier diode D1 for realizing conversion, transmission, and input and output isolation of power. The auxiliary power supply 15 includes: a positive feedback winding P3 and a diode D3. The working principle is that when the feedback winding has the same name and the diode D3 is turned on, the optical coupling is energized. The output filter circuit 16, including the filter capacitor C3, can also be used in other existing filter circuits, which can be selected according to the relevant technical manual. The isolated optocoupler comprises: an optocoupler OC1, which mainly performs signal transmission and input/output isolation. The error amplifying ADJ circuit 17 includes: a sampling circuit and a signal comparison amplifying circuit. The working principle is as follows: When the output voltage is high, the sampling circuit collects the signal and adjusts the primary current of the optocoupler after the signal is compared with the amplifier circuit, that is, the duty cycle of the product is adjusted by the voltage loop. At the output end of the power supply, a sampling current flows through the sampling circuit, the error amplification, the isolation optocoupler, and the pulse frequency modulation PFM circuit to perform negative feedback control on the main power tube in the main power circuit; a soft connection is connected to the output end of the input filter circuit. The startup circuit is connected to the pulse frequency modulation PFM at the other end of the soft start circuit to implement a soft start function of the power supply.

The disadvantages of the above circuits are:

1) During the start-up phase of the product, with full load, especially for large capacitive loads, it cannot enter the steady state, and the output of the product is abnormal. After the above circuit is turned on, it needs to oscillate for a period of time before the product can enter the steady state; during the non-steady state, the feedback The winding energy is weak. At this time, the capacitor C9 is used to compensate the energy. However, the voltage of the capacitor C9 starts to decrease after the first period of the MOS transistor is turned on (the waveform of the capacitor C9 is shown in Figure 7, and the voltage of the MOS transistor gradually decreases after being turned on). The energy is getting weaker and weaker, so the circuit is easy to oscillate at full load, light load, especially large capacitive load, and the output voltage is low. When the output has a capacitive load, the product output voltage rises and the rise time becomes longer. Figure 12 compares the normal operation of the power converter with the model PWB4805D and power of 3W and the output voltage waveform with 1000 capacitors, where CH1 is the normal working waveform. , the rise time is about 0.5ms, CH2 is the working waveform with lOOO f capacitor, the rise time is about 3.7ms, the voltage loop is out of control when the output voltage is not formally established. If the MOS tube drive can not get enough compensation, the feedback winding Easy to stop vibration, showing poor start. The following describes the phase of the process in detail: Taking the existing PWB4805D power converter with power of 3W as an example, the starting circuit parameters take into account the key performances such as short-circuit power consumption and starting current. The circuit principle is shown in Figure 2. The soft-start circuit 12 The values of each parameter are as follows: R10=332 K Ω , R13=3.3 K Ω , R14=150 K Ω , C9=1 μ f, feedback winding drive in control circuit 13 The branch Rll=100 Ω, C6=4700PF, the main power circuit 14 part, the parameters of the transformer T1 are: Np=25, Ns=7, Nf=8. In the initial stage of MOS tube conduction, the feedback winding voltage is small, and the G-pole potential is almost zero in this stage. Therefore, the discharge time constant in the unsteady period can be approximated as T2 R13*C9 3.3ms, and the charging time constant Tl R14* C9=332ms, so the non-steady state period, after the MOS transistor is turned on, the potential of the capacitor C9 gradually decreases. Figure 7 shows (test conditions: light load Io=0.06A, high voltage 72V test) voltage waveform on capacitor C9, MOS tube guide After the pass, the capacitor voltage gradually decreases. The above parameter R10 takes 332 K Ω, taking into account the product's startup start current and output short-circuit power consumption. The table gives an example of the minimum current limit value of the power-on startup and the output short-circuit power consumption at each point when the input voltage is 18V, 48V, 72V, full load Io=0.6A.

Table I

Table 2 shows the average voltage of capacitor C9 under different load and input voltage conditions when R10 is 332 K Ω.

Table II

As can be seen from Table 1, the starting current and short-circuit power consumption of the product are small, and the maximum short-circuit power consumption is only 0.792W. It can be seen from Table 2 that when R10 takes a large value, the Vc9 voltage varies widely, and the MOS tube The compensation strength of the drive is unstable, and the compensation is the worst at light load high voltage, and the voltage is only 1.02V. When the value of R10 becomes smaller, the charging time constant T1=R10*C9 decreases, and the energy compensation of the capacitor C9 is enhanced. Since the R10 is reduced, the compensation intensity is not easy to control, especially in products with large input voltage and load variation range. Taking into account the compensation strength of each point, the disadvantages are: 1) The startup current becomes larger 2) The short-circuit power consumption increases, especially at high When the voltage is short, the power consumption of the short circuit increases significantly. In severe cases, the product is burned. Table 3 shows the output short-circuit power consumption and starting current of the product after changing R10 from 332 K Ω to 50 Ω Ω. Table 4 exemplifies the average voltage of the capacitor C9 under this condition.

Table 3

Table 4

It can be seen from Table 3: 1) The short-circuit power consumption is increased compared with the original scheme, and the high voltage is 3.96W. The long-time short-circuit will damage the product. 2) The starting current increases obviously, especially when the high-voltage starting current is nearly 1.7 times higher than the original scheme. At this point, the customer may start the current limit when using it. Table 4 shows the average voltage value of C9 under different load and voltage conditions after R10 takes 50 K Ω. The table shows that after R10 is reduced, the voltage of capacitor C9 varies greatly under various load conditions, and the maximum voltage is 12.8V. The minimum voltage is only 4.39V, and its compensation is too strong, resulting in large starting current and short circuit power consumption. On the basis of the original scheme, after changing R13 from 3.3 K Ω to 7.5 Κ Ω, the product starts to start poorly at the full load, so the discharge time constant is allowed to change little. In summary, changing the value of R10, its compensation stability for MOS tube drive is poor, and over-compensation and under-compensation are prone to occur.

2) At steady state, under high-voltage and light-load conditions, the capacitor C9 in the soft-start circuit discharges faster than the charging speed, and its ground voltage is close to IV (see Figure 7, light-load Io=0.06A, high-voltage 72V test), so MOS The tube drive depends almost entirely on the feedback winding energy. At this time, the output performance of the product depends on two points: a. The coupling performance of the feedback winding and the primary winding; b. Output load. When the feedback winding and the input coupling deteriorate, the coupled drive energy deteriorates, especially under the condition of input voltage jump, the driving energy is insufficient, which easily leads to product oscillation; When the output load becomes small (10% load or less), the feedback winding energy Greatly reduced, its drive The energy (coupling current) is also reduced, and the product is driven into intermittent oscillation, which is manifested by a low output voltage. The following is a detailed description of the working process of the product under high voltage and light load: Take the existing PWB4805D power converter with power of 3W as an example. The starting circuit parameters take into account the key performances such as short-circuit power consumption and starting current. The circuit principle is shown in Figure 2. The values of the parameters in the soft start circuit 12 are as follows: R10=332 Κ Ω , R13=3.3 K Ω , R14=150 Κ Ω , C9=1 μ f, the feedback winding drive branch Rll=100 Ω in the control circuit 13 C6=4700PF, the main power circuit 14 part, its transformer T1 parameters are: Np=25, Ns=7, Nf=8. Under steady-state conditions, when the feedback winding is reversed (corresponding to the MOS transistor is turned off), the same name terminal-to-ground voltage

V *N _f ―

V _f = - - - ~ ^ = -4.375V, due to capacitor C6 and MOS tube GS junction capacitance Ciss (using IRFR220 f ^P Ns

The junction capacitance of the MOS transistor is about 300 PF. The capacitance is small, and the resistance R11 is only 100 Ω. Therefore, when the MOS transistor is turned off, the feedback winding is reversed, and the MOS tube Ciss stores energy on the one hand through the PNP transistor TR3, and On the one hand, the driving branches R11 and C6 are released to the same name end of the feedback winding, and the Vgs potential of the MOS transistor is rapidly pulled down. The measured waveform is shown in Fig. 8 (test condition: Vin=18V, Io=0.6A). Out, the MOS tube is turned off during the Vgs«-0.64V. Therefore, the capacitor C9 in the shutdown phase is discharged to the G pole through the resistor R13; the higher the switching frequency of the product, the smaller the duty ratio, that is, the off time per unit time is long, so the high frequency condition C9 has a long discharge time and its own potential is also low. Figure 7 shows the voltage waveform on the C9 capacitor (high voltage 72V, output load 0.06A). From this figure, it can be seen that after the MOS transistor is turned on, the capacitor voltage begins to drop. After the steady state, the capacitor voltage is only IV, and its corresponding drive. The waveform is shown in Figure 9, where T« 1.07 μs, effective on-time Τοη^Ο.01 μs (drive voltage is greater than the turn-on threshold is active high, ie Vgs > 3.3V), D=0.0093, ie under this condition The MOS tube is in the off state at 99.06%, so the effective discharge time constant T2 R13*C9=3.3ms, and its charging time constant Tl=R10*C9=332ms, so T2 < <T1. When the charging time constant R10 is increased and the value of R13 is increased, the disadvantages are the same as those described under the non-steady state condition.

3) When external factors interfere with the control signal (such as static electricity), the switch tube is completely turned off in a short time. When the interference disappears, the product needs to be charged by soft start, and the MOS transistor is turned on until the output is normal (generally the output rise time is 0.5). Ms) After two periods of time, the steady state operation can be entered, and the output appears to be powered down. Figure 11 shows the output power-down waveform of the PWB4805D-3W product under high-voltage and light-load conditions, after 4KV electrostatic interference at the input end. This waveform shows that when the input terminal is static, the output of the product is easy to be powered down, and when it is serious, it interferes with the restart of the customer system. The following is a detailed description of the cause of power failure: the circuit principle is shown in Figure 2, starting part 12, R10=332 K Ω, C9 takes 1 μ f, R13=3.3 K Ω; Switch tube adopts IRFR220, check the specification whose threshold is 2V-4V, and its conduction threshold V(th)=3.3V, so the fastest when Vin=36V Start-up time t = -R ₁₀ *C ₉ *ln(l -^^) =31.9 lms, due to the current limit of R13, the actual start-up time will be slightly

Vin

It is longer than the above calculated value, and the actual measurement is about 35ms. Therefore, when the static electricity is interrupted by the static EMI, the shutdown time will inevitably become longer and the output of the product will be powered down. The scheme reduces the value of R10, which can enhance the soft start compensation capability and reduce the risk of power failure. However, due to the influence of the input voltage and load variation range, it is difficult to reduce the compensation of R10 for different input voltage points and load conditions. When the strength is too large, the short-circuit power consumption increases, the starting current increases, and the compensation is too small to solve the above problem. Summary of the invention

SUMMARY OF THE INVENTION It is an object of the present invention to provide a self-oscillating flyback converter capable of providing a suitable drive compensation for starting and steady-state phases of a product while achieving a soft start function, thereby improving product anti-jamming capability and stability, and at the same time expanding The design range of the input voltage of the self-oscillating flyback converter and the range of variation of the load.

The object of the invention is achieved by the following technical measures:

A self-oscillating flyback converter, wherein a DC input signal sequentially passes through an input filter circuit, a main power circuit and an output filter circuit, and outputs a DC signal, the main power circuit includes a main power tube and a main transformer; and the output DC signal passes through The voltage stabilizing circuit, the isolated optocoupler and the drive control circuit perform negative feedback control on the main power tube to achieve a stable output, and further include a soft start circuit with a constant current source connected to the output end of the input filter circuit and Between the drive control circuits; when the self-oscillating flyback converter is in an unsteady state after power-on, the soft start circuit is charged to the soft start circuit with a constant current source through a constant current source, and the soft start voltage rises to the MOS tube threshold After the value is turned on, the MOS tube is turned on. Since the constant current source charging speed is balanced with the soft start capacitor discharging speed, the soft starting capacitor continuously supplies driving compensation to the MOS tube to achieve normal starting; the self-oscillating flyback converter enters the steady state. The constant current source charging speed is balanced with the soft start capacitor discharging speed, and the soft start continues to provide driving compensation to the MOS tube. Steady work properly.

As an embodiment of the present invention, the soft start circuit with a constant current source includes a constant current source, a first voltage dividing resistor, a second voltage dividing resistor, and a starting capacitor; the anode of the constant current source is connected to the self-excited a power supply input end of the oscillating flyback converter, wherein the cathode of the constant current source is sequentially connected to a power reference terminal of the self-oscillating flyback converter through the first voltage dividing resistor and the second voltage dividing resistor, the starting capacitor And the first voltage dividing resistor And a series branch of the second voltage dividing resistor is connected in parallel, and a connection point of the first voltage dividing resistor and the second voltage dividing resistor is connected to a gate of the main power tube.

More preferably, the soft start circuit with a constant current source further includes a current limiting resistor, and the anode of the constant current source is connected to the power input end of the self-oscillating flyback converter through the current limiting resistor.

More preferably, the soft start circuit with a constant current source further includes a current limiting resistor connected between a cathode of the constant current source and a connection point between the first voltage dividing resistor and the starting capacitor .

As an embodiment of the present invention, the constant current source (D1A) is a single constant current source, or a constant current source in parallel, or a constant current source formed by a combination of a constant current source and a triode, a Zener, and a resistor. , or a constant current source composed of a triode and a resistor.

Compared with the prior art, the present invention has the following advantages:

One of the advantages of the present invention is to improve the starting capability of the inverter product and the capacitive load capacity, so that the product can start normally and work stably under full load and capacitive load. Table 5 lists the original scheme of the power converter of the type WRF4815P and the power of 6W compared with the capacitive load capacity after adopting the technical scheme of the present invention.

Table 5

It can be seen from Table 5 that after adopting the technical solution of the present invention, the capacity load capacity of the converter product is greatly improved.

The second advantage of the invention is: improving the output voltage stability when the product is lightly loaded (below 10% load), reducing the low energy of the feedback winding under light load (small current at light load, weak coupling energy) Hidden dangers; at the same time, it makes it possible to design a larger rated load current for the self-oscillating flyback converter.

The third advantage of the invention is that the process requirements of the transformer are reduced (the feedback and input coupling coefficient requirements are reduced), the requirement for the consistency of the switch tube threshold is reduced, and the product productivity is improved. Figure 10 shows the threshold test between different batches of MOS tubes of the type IRFR220. Test conditions: Vgs=Vds and Id=250 μ A, T=25°C. It can be seen from the figure that the MOS tube threshold turn-on voltage has a large difference between batches, and the YG batch MOS tube threshold is higher than the OM batch. Secondary MOS tube threshold, so it can The demand for the quantity also increases. The technique of the present invention provides suitable compensation for the MOS tube, so that the MOS tube can be reliably driven, and the consistency requirement for the MOS tube conduction threshold is reduced.

The fourth advantage of the invention is: improving the anti-interference ability of the product, and effectively solving the output power-down phenomenon. In the experiment, the power converter product with the model PWB4805D and power of 3W is used to supply power to the single-chip microcomputer. When the input terminal is 4KV static, the single-chip microcomputer is reset, and the power converter output is powered down. After the technical solution of the present invention is improved, the input terminal is 4KV. Static electricity, the output of the inverter is normal, and the MCU works normally.

Advantage 5 of the present invention: The adaptability of the self-oscillating flyback converter over a wide input voltage range (4:1) is improved. Due to the start of the constant current source, the starting speed and compensation intensity of the product under low voltage, nominal and high voltage are effectively controlled, so that the MOS tube drive is reasonably compensated under the conditions of full input voltage and full load range and large capacitive load. Solved short-circuit power consumption, starting current and other issues. Table 6 lists the short-circuit power consumption and startup performance of a PWB4805D, 3W power converter with a constant current source of 0.5mA. Table 6

Comparing Table 3, it can be seen that the short-circuit power consumption is greatly reduced, and the product has a long-term short-circuit protection function; the starting current of each input voltage point is reduced compared with Table 3, and the nominal reduction is 67 mA, the reduction is 37%, and the reduction is reduced. The customer starts the power supply current limit risk. DRAWINGS

The present invention will be further described in detail below with reference to the accompanying drawings and specific embodiments:

1 is a schematic block diagram of a prior art self-oscillating flyback converter;

2 is a circuit schematic diagram of a self-oscillating flyback converter in the prior art;

3 is a schematic block diagram of a self-oscillating flyback converter of the present invention;

4 is a schematic circuit diagram of a first embodiment of the present invention;

5 is a schematic circuit diagram of a portion of a soft start circuit with a constant current source according to an embodiment of the present invention; FIG. 6 is a circuit schematic diagram of three soft start circuit portions with a constant current source according to an embodiment of the present invention; FIG. 7 is a self-oscillation A voltage waveform diagram of the starting capacitor C9 in the soft start circuit of the flyback converter; Figure 8 is a waveform diagram of Vgs driving at low voltage and full load of a power converter of the type PWB4805D and power of 3W;

Figure 9 is a waveform diagram of Vgs driving at a light load and high voltage of a power converter of model PWB4805D and power of 3W;

Figure 10 is a graph showing the normal temperature threshold of a MOS tube of the type IRFR220;

Fig. 11 is a waveform diagram of the output voltage when the power converter of the model PWB4805D and the power of 3W is input with 4KV static electricity;

Figure 12 shows the waveform of the output voltage rise of the PWB4805D, 3W power converter without capacitive load and 1000 μf capacitive load;

Figure 13 is a circuit schematic diagram of a parallel form of a constant current source;

14 is a circuit schematic diagram of a composite constant current source composed of a constant current source, a resistor, a Zener tube, and a triode; FIG. 15 is a circuit schematic diagram of a constant current source formed by a PNP transistor and a resistor;

Figure 16 is a circuit schematic of a single tube drive control circuit. detailed description

The invention will be further described in detail below with reference to the drawings and specific embodiments.

As shown in FIG. 3, in the self-oscillating flyback converter of the present invention, the DC input signal sequentially passes through the input filter circuit, the main power circuit and the output filter circuit, and then outputs a DC signal, and the main power circuit includes a main power tube and a main transformer; The output DC signal is sequentially subjected to negative feedback control of the main power tube through the voltage stabilization circuit, the isolation optocoupler and the drive control circuit to achieve stable output, and further includes a soft start circuit with a constant current source, the soft start circuit being connected Between the output end of the input filter circuit and the drive control circuit; the self-oscillating flyback converter is charged to a soft start circuit with a constant current source through a constant current source when it is in an unsteady state, soft start After the voltage rises to the MOS tube threshold, the MOS transistor is turned on. Since the constant current source charging speed is balanced with the soft start capacitor discharge speed, the soft start capacitor continuously supplies driving compensation to the MOS tube to achieve normal startup; After the converter enters the steady state, the constant current source charging speed and the soft start capacitor discharge speed are balanced, and the soft start continues to the MOS tube. For drive compensation, achieve steady state normal operation.

Referring to FIG. ₄ , a first embodiment of a self-oscillating flyback converter with a constant current source soft start circuit of the present invention is shown. In the first embodiment, the pulse frequency modulation circuit 13 is used as the drive control circuit of the converter. The implementation circuit of the embodiment mainly includes the following components: an input filter circuit 11, a soft start circuit with a constant current source, and a pulse frequency. Modulation circuit 13, main power circuit 14, auxiliary power supply 15, output filter circuit 16. The voltage stabilizing circuit 17, wherein the circuit structure of the input filter circuit 11, the pulse frequency modulation circuit 13, the main power circuit 14, the auxiliary power supply 15, the output filter circuit 16, and the voltage stabilization circuit 17 is the same as that of the circuit shown in FIG. The circuit structure is the same. The difference between this embodiment and the circuit shown in FIG. 2 is that the circuit composition of the constant current source soft start circuit 12 of the circuit shown in FIG. 2 is different from that of the circuit shown in FIG. 2:

In this embodiment, the main power tube in the main power circuit 14 uses the MOS tube TR1, and the main transformer uses the transformer T1. The transformer T1 includes the primary winding PI, the output winding P2, and the positive feedback winding P3.

The input filter circuit 11 includes a filter capacitor C0, a filter capacitor C1 and a filter inductor L0, and the structure thereof is a well-known IT type filter circuit principle structure, which will not be described in detail herein.

The soft start circuit with constant current source 12 includes a constant current source D1A, a first voltage dividing resistor R13, a second voltage dividing resistor R14, and a starting capacitor C9. The anode of the constant current source D1A is connected to the power input end, and the cathode of the constant current source D1A is sequentially connected to the power reference end of the self-oscillating flyback converter through the first voltage dividing resistor R13 and the second voltage dividing resistor R14. The starting capacitor C9 is connected in parallel with the series branch of the first voltage dividing resistor R13 and the second voltage dividing resistor R14, and the connection point of the first voltage dividing resistor R13 and the second voltage dividing resistor R14 is opposite to the main power tube The gates are connected. The working principle of this circuit is described in detail below:

Start-up phase: When the input voltage is connected, the voltage is charged to the starting capacitor C9 via the constant current source, and the elapsed time _{ί =} ^ (where υ _ώ is the starting threshold of the MOS transistor TR1, C is the capacity of the starting capacitor C9, and i is the constant current c * i

After the constant current of the source D1A working area), the voltage of the starting capacitor C9 reaches the threshold voltage of the MOS transistor TR1, and the soft start function is realized. Before the output voltage is officially established, that is, during the non-steady state, the starting capacitor C9 passes the first partial voltage. The resistor R13 supplies energy to the MOS transistor TR1. On the other hand, the constant current source D1A supplies energy to the starting capacitor C9 in time, and the appropriate constant current source D1A can be selected to meet the energy balance of the starting capacitor C9 charging and discharging, due to the energy compensation of the starting capacitor C9. The positive feedback winding P3 requires only a small amount of energy to complete the self-oscillation process, thereby avoiding the occurrence of intermittent oscillations and starting the product normally.

Normal working phase: When the input voltage changes or the output load changes, the coupling energy of the positive feedback winding P3 changes, because the starting capacitor C9 is powered by the first voltage dividing resistor R13 for the driving of the MOS transistor TR1 and the constant current source D1A is activated. Capacitor C9 provides the same level of energy, MOS tube TR1 drive is effectively compensated, the coupling energy requirement for positive feedback winding P3 is greatly reduced, ensuring that the positive feedback winding P3 self-oscillation is normal, improving the output voltage of the product under different input voltages and different load conditions. Stability. Table 7 shows the power converter of model PW4805D with power of 3W. After constant current source D1A adopts constant current source of 0.5mA, the voltage of starting capacitor C9 under different input voltage and load conditions is compared with Table 2 and Table 4. After adopting the technology of the present invention, the voltage on the starting capacitor C9 is basically stable, and the variation range is only 2.4V, which avoids the MOS tube driving over-compensation and under-compensation caused by the difference in the value of the resistor R10 in the original scheme.

Table 7

The pulse frequency modulation circuit 13 includes: a resistor R6, a resistor R9, a resistor R11, a resistor R111, a capacitor C5, a capacitor C6, a capacitor C12, a NPN transistor TR2, and a PNP transistor TR3. The capacitor C6 is connected in series with the resistor R11, the resistor R111, and the resistor R6. One end of the series circuit is connected to the same name end of the positive feedback winding P3, and the other end is connected to the collector of the transistor TR2. The capacitor C5 is connected in parallel with the resistor R9. One end of the parallel current is connected to the base of the transistor TR2, and the other end is connected to the collector of the transistor TR3. Capacitor C12 is connected in parallel with resistor R111. One end of the parallel circuit is connected to the emitter of the three-pole TR3, and the other end is connected to the base of the three-pole TR3. The working principle is as follows: The positive feedback winding P3, the capacitor C6, and the resistor R11 branch are coupled with the primary and secondary sides of the main power transformer T1 to form a self-excited oscillation circuit, and the control MOS transistor TR1 is turned on and off; and the current loop resistor R5 branch The voltage loop optocoupler OC1 branch adjusts the on-duty of the switch tube by the double-tube drive control circuit of the transistor TR2 and the transistor TR3, so that the product output is normal.

The main power circuit 14, including the primary winding P1 of the transformer T1, the output winding P2, the MOS transistor TR1, the current limiting resistor R5, the absorbing capacitor C14, and the output rectifier diode Dl, realize the conversion, transmission and isolation of the input and output of the power source.

Auxiliary power supply 15, including positive feedback winding P3, diode D3, works on the principle that the feedback winding has the same name as the timing diode D3 is conducting, providing energy for the optocoupler.

The output filter circuit 16, including the capacitor C3, can also be used with other existing filter circuits, which can be selected according to the relevant technical manual.

The voltage stabilizing circuit 17 mainly includes a voltage regulator ADJ, which is connected to the main power circuit, the auxiliary power source 15 and the pulse frequency modulation circuit 13 through the optocoupler OC1, and will not be described herein.

The other two embodiments will be briefly described below, wherein only the portion of the soft start circuit with the constant current source is shown, and the connections of the other portions are the same as those of the circuit shown in FIG. 5 is a portion of a soft start circuit with a constant current source according to a second embodiment of the present invention. The difference from the first embodiment is that the constant current source D1A and the power input end are connected in series with the constant current source soft start circuit of the embodiment. A current limiting resistor R1A for limiting the maximum current, which limits the maximum current of the startup circuit while reducing the voltage division across the constant current source.

6 is a portion of a soft start circuit with a constant current source according to a third embodiment of the present invention, which differs from the circuit shown in the second embodiment only in limiting the maximum current in the constant current source soft start circuit of the embodiment. The access position of the flow resistor R1A changes, and the current limiting resistor R1A is connected between the cathode of the constant current source D1A and the connection point of the first voltage dividing resistor R13 and the starting capacitor C9.

It should be noted that the above constant current source may have multiple compositions, which may be a parallel form of a constant current source in the prior art, as shown in FIG. 13; a constant current source and a triode, a Zener, and a resistor in the prior art The constant current source, as shown in Fig. 14, has a constant current output of Io = (V _2D - V _BE1 ) / R; it can also be a constant current source composed of a triode and a resistor in the prior art, as shown in Fig. 15, which is constant The stream output Io = V _te /R ₅ . ₅ and so on.

In addition to the several implementation circuits described above, other equivalent applications that can be naturally associated with the above description and the accompanying drawings by the skilled person, for example, the drive control circuit of the converter is as shown in FIG. 16, by the transistor TR2, the resistor R6. The single-tube drive control circuit composed of the capacitor C5 and the resistor R9 replaces the pulse frequency modulation circuit and the like in the above embodiments. For those skilled in the art, the present invention may be carried out without departing from the principle of the present invention. Modifications and modifications are intended to fall within the scope of the appended claims.

Claims

Rights request

1. A self-oscillating flyback converter, wherein a DC input signal sequentially passes through an input filter circuit, a main power circuit, and an output filter circuit, and outputs a DC signal, the main power circuit includes a main power tube and a main transformer; and the output DC signal The main power tube is negatively feedback controlled by the voltage stabilizing circuit, the isolated optocoupler and the driving control circuit to realize a stable output, and is characterized in that: a soft start circuit with a constant current source is further connected, and the soft start circuit is connected to the input Between the output of the filter circuit and the drive control circuit.

The self-oscillating flyback converter according to claim 1, wherein: the soft start circuit with a constant current source comprises a constant current source (D1A), a first voltage dividing resistor (R13), and a second a voltage dividing resistor (R14) and a starting capacitor (C9); an anode of the constant current source (D1A) is connected to a power input terminal of the self-oscillating flyback converter, and a cathode of the constant current source (D1A) sequentially passes through The first voltage dividing resistor (R13) and the second voltage dividing resistor (R14) are connected to a power reference terminal of the self-oscillating flyback converter, and the starting capacitor (C9) and the first voltage dividing resistor (R13) Connected in parallel with the series branch of the second voltage dividing resistor (R14), the connection point of the first voltage dividing resistor (R13) and the second voltage dividing resistor (R14) is connected to the gate of the main power tube.

The self-oscillating flyback converter according to claim 2, wherein: the soft start circuit with a constant current source further comprises a current limiting resistor (R1A), and an anode of the constant current source (D1A) The current limiting input (R1A) is connected to the power input terminal of the self-oscillating flyback converter.

The self-oscillating flyback converter according to claim 2, wherein: the soft start circuit with a constant current source further comprises a current limiting resistor (R1A), and the current limiting resistor (R1A) is connected to Constant current source

The cathode of (D1A) is connected to the junction of the first voltage dividing resistor (R13) and the starting capacitor (C9).

The self-oscillating flyback converter according to any one of claims 1 to 4, characterized in that: the constant current source (D1A) is a single constant current source, or a constant current source is connected in parallel, or constant A constant current source composed of a combination of a current source and a triode, a Zener tube, and a resistor, or a constant current source composed of a triode and a resistor.