WO2012049205A1

WO2012049205A1 - Inductive proximity switch

Info

Publication number: WO2012049205A1
Application number: PCT/EP2011/067794
Authority: WO
Inventors: Jaromir Palata
Original assignee: Ifm Electronic Gmbh
Priority date: 2010-10-15
Filing date: 2011-10-12
Publication date: 2012-04-19

Abstract

The invention relates to an inductive proximity switch (1), comprising a transmitting coil (2), two receiving coils (3) connected in an anti-serial manner, and a compensating winding (4) coupled to one of the receiving coils in a transforming manner. The compensating winding (4) compensates the target (5) in connection with a controllable resistor (9). The resistor (9) is part of a control loop, which controls the received signal to zero. The control variable, i.e. the direct current flowing through the controllable resistor (9), is measured by a current-measuring unit (10) and is output as a signal. In a further embodiment, the compensating winding (4) and the controllable resistor (9), which comprises a resistor (Re) and a switch, are parts of a control loop designed as a relaxation oscillator, wherein said control loop compensates the damping caused by the target (5) in the centre. The duty cycle of the relaxation oscillation that develops is a measure of the damping by the target (5) and can be output as a pulse-width-modulated measured variable.

Description

INDUCTIVE PROXIMITY SWITCH

The invention relates to an inductive proximity switch according to the features of the preamble of patent claim 1.

Inductive proximity switches are used as non-contact electronic switching devices, especially in automation technology.

They contain a transmitting coil which generates an electromagnetic magnetic field which can be influenced by a metallic trigger. The influence of the magnetic field by the metallic release is evaluated and, when a threshold value is exceeded, an electronic switching stage is activated.

Switching devices of this type are manufactured and distributed in various designs, inter alia, by the Applicant.

In this case, both the control of the transmitting coil and the evaluation of the influence of the metallic trigger can be done in different ways.

In many cases, the transmitter coil is part of an affected by the metallic shutter oscillator whose amplitude and / or frequency change is evaluated.

In addition to the widespread sinusoidal control of the transmitting coil and the evaluation of frequency and or amplitude changes, the control with a short square pulse is known. The transmitter coil is in this case not part of an oscillator, but it is subjected to strong voltage or current pulses. The echo triggered by the eddy currents produced in the metallic trigger is evaluated. This evaluation can be carried out both directly on the transmitting coil and on a magnetically coupled receiving coil.

Since this is a closed resonant circuit or a closed coil circuit, only little energy is radiated. The interaction with the metallic trigger (target) is limited to the near field. It decreases approximately with the third power of the switching distance.

In order to be able to detect minor interactions with the metallic trigger, it is advantageous to compensate the signal in the uninfluenced state and to evaluate only the changes caused by the trigger.

For this purpose, preferably two receiver coils are operated in differential circuit.

The design is chosen so that one of the two coils is more affected by the target than the other.

By zeroing in the uninfluenced state, one obtains an extremely sensitive arrangement, which is also referred to as a differential transformer (LVDT = linear variable differential transformer).

The differential transformer is adjusted so that cancel the signals of the two receiving coils in the uninfluenced state each other.

The better this balance is achieved, the higher the sensor signal can be amplified, without causing overmodulation. Since the magnetic field decreases very rapidly with increasing distance, one soon comes into areas where the temperature response, in particular of the copper windings, but also of the other materials and components involved, cause effects of the order of magnitude of the expected sensor signal. Therefore, higher switching distances can only be achieved if the temperature dependence of the arrangement over the working temperature range can be compensated.

Since this balance can be disturbed during production but also by the installation situation, a subsequent adjustment of the differential transformer is desirable both at the factory and during later operation.

In DE 10 2007 014 343 A1 it is proposed to connect a receiving coil with trimmable resistors, by means of which the switching distance is set. These are controlled by an evaluation unit (microcontroller), which is also connected to a temperature sensor, so that a temperature compensation can be made.

A disadvantage here is that the temperature compensation consists of a relatively lengthy process of the measurement by the temperature sensor, the calculation of the correction values in the microcontroller, the subsequent digital output of the correction values and the digital-to-analog conversion to the triggering of the trimmable resistors. A linearization of, for example, -25 ° C to + 70 ° C appears problematic in view of this long chain.

In WO 2007/012502 A1, a control loop with a control system comprising the sensor is proposed. Controlled variable is u.a. the amplitude of the sensor circuit.

The regulation takes place on the transmitter side, i. the amplitude is kept constant against the damping disturbance.

The energy required to control the amplitude is supplied via an adjustable resistor. The control variable for this resistance is decoupled as a quantity for the instantaneous damping and thus as a measured variable.

The disadvantage here is that the adjustable resistor, preferably a transistor,

has both a temperature response and a non-linear characteristic.

The object of the invention is to further improve this process.

The temperature dependence of the circuit is to be reduced, the characteristics of the actuator linearized and the dependence on the parameter dispersion of the actuator can be eliminated.

This object is achieved according to the features of claims 1 or 2.

Claims

3 and 4 relate to methods for operating the inductive proximity switch according to the invention.

The essential idea of the invention is to measure the manipulated variable directly instead of its temperature-dependent control variable. This measurement should be done either analog or digital as a pulse width ratio.

In a first embodiment, the resistance of the high frequency actuator is measured with DC and not its control voltage.

The main advantage is that the temperature response of the controlled system can be allowed because the manipulated variable is measured directly in the form of the damping resistor on the actuator.

Since the high-frequency actuator is usually a transistor, neither its control voltage nor its control current, but the track resistance is measured with DC and output as an analog voltage.

In a second embodiment, the actuator is designed as a switch with a defined damping resistance. This switch is closed for certain time intervals and then reopened. The resulting pulse width modulated switching signal can be output directly as a measured variable. The sensor signal is not permanently zero in this case, but it fluctuates around the zero point.

The temperature response of the controlled system can also be allowed here, because the manipulated variable in the form of the duty cycle of a switchable known damping resistor is measured directly on the actuator.

This digitized signal generated directly at the sensor can practically no longer be falsified by further processing.

The invention is explained in more detail with reference to the drawing.

Show it:

1 is a schematic diagram of an inductive proximity switch according to the invention according to the first embodiment,

2 shows a schematic representation of an inductive proximity switch according to the invention according to the second embodiment,

3 shows an embodiment according to the first embodiment with a MOSFET as a damping resistor in the actuator,

Fig. 4: Embodiment of the second embodiment with XOR gates and a JFET as a switch in the actuator.

Fig. 5: Embodiment of the second embodiment with analog multiplexers in the LC oscillator, in the phase rectifier and as a switch in the actuator.

1 shows an inductive proximity switch 1 according to the first embodiment of the invention with a transmitting coil 2, two receiving coils 3 a tuning winding 4 and another, the switching flag or the inductively coupled target symbolizing winding 5. The transmitting coil and the two receiving coils form a Differential transformer (LVDT).

The transmitting coil 2 is fed by a high-frequency generator 6. The two anti-serially connected receiving coils 3 are connected to the phase-sensitive synchronous rectifier 7. The rectified signal passes through the integrator 8 and controls the variable resistor 9. The deprives the system via the tuning winding 4 as long as energy until the signal at the synchronous rectifier 7 is zero, i. E. the balance between the target 5 and the trimming coil 4 is established.

The DC current flowing through the controllable resistor 9 (actuator) located in the circuit of the tuning coil 4 is a measure of the damping. It is measured and can be output as an analogue signal in the form of current or voltage, but also digitized and possibly displayed.

Finally, it should be noted that a construction according to the invention can also be realized with two transmitting coils and one receiving coil.

2 shows an inductive proximity switch 1 according to the second embodiment of the invention with a transmitting coil 2, two antiserial receiving coils 3 a tuning winding 4 and another, the switching flag or the target symbolizing winding 5. The transmitting coil and the two receiving coils form a differential transformer (LVDT).

The transmitting coil 2 is fed by a high-frequency generator 6. The two anti-serially connected receiving coils 3 are connected to the synchronous rectifier 7. The rectified signal passes through the integrator 8 and controls the

the Schmitt trigger 11, which switches on and off the controllable resistor 9 consisting of a switch and a resistor Re.

Due to the hysteresis of the Schmitt trigger 11, the system constantly oscillates between an overcompensated and an undercompensated state.

In this way, a control loop is formed, which controls the mean value of the sensor signal to zero and thereby generates a pulse width modulated (PWM) signal.

With optimum alignment, the switch without target remains mostly open and in the heavily damped state, it remains mostly closed. The closing time or the duty cycle is thus a measure of the damping by the target and can thus be readily output as a pulse width modulated measured variable.

By changing the resistance Re, a measuring range switchover is possible.

Fig. 3 shows a detailed embodiment according to the first embodiment of the invention. The sinusoidal transmit signal is generated by the dual-channel analog multiplexer MUX1 type 74HC4053, which operates as an inverter in the LC oscillator. For the frequency f the relation holds:

f = 1 / (2Pi *

LS * (C1 * C2) / (C1 + C2)).

The transmitting coil LS is connected to the two receiving coils L1 and L2 via the transformer coupling factors M1 and M2. These are adjusted so that the voltages generated in the receiving windings L1 and L2 are equal in the unloaded state.

The center tap of the two receiving coils L1 and L2 is at half the operating voltage and high frequency to ground.

With unipolar power supply this ensures the correct operating point at the inputs of the comparator OV1.

The other ends of the two anti-serially connected receiving coils are connected via 2 damping resistors to the inputs of the comparator OV1. This works as a differential pulse shaper.

As soon as the equilibrium is disturbed by the reduction of the resistance Rx connected to the symbolically to-be-understood target winding 5, the output of OV1 supplies a square-wave signal synchronous with the oscillator.

The comparator OV2 also provides a phase synchronous with the oscillator square wave signal. Both comparators are located in a common housing, which guarantees the same temperatures and therefore the same switching times. The multiplexer MUX2 operates as a phase-sensitive rectifier. The differential integrator with the operational amplifier OV3 integrates the signal supplied by the MUX2 and thus controls the transistor T1, a small MOSFET. The transistor T1 corresponds to the controllable resistor 9 in Fig. 1. It deprives the differential transformer so much energy until the equilibrium is restored.

The measured variable is the drain-source resistance R _DS of T1 or the drain current I _DS .

The measurement is carried out with a current measuring circuit with the operational amplifier OPV4, which corresponds to the current measuring circuit 10 shown in FIG.

For DC, the gain is Vu = - RA / Ra * with Ra * = Ra + R _Cu + R _DS . (R _Cu : copper resistance of the coil, R _DS : drain-source resistance R _DS of T1)

Ri and Ci determine the integration time and provide stability of the circuit at high gains. For DC they are irrelevant.

The blocking capacitor C _HF short-circuits the HF circuit at the balancing winding 4 and separates the high-frequency components from the DC components so that no high-frequency components reach the input of the OV 4.

The DC voltage present across the blocking capacitor C _HF appears as a useful signal amplified by the above-mentioned factor at the output of OPV4.

Neglecting the copper resistance of the coil gives:

R _DS (T1) = Uref * RA / UA-Ra.

For the output voltage UA: UA = Uref * RA / (R _DS + Ra).

For the current one obtains: I _DS (T1) = (UA - Uref) / RA.

The resistor Ra is intended to prevent resonances at the resonant circuit formed by the drain-source capacitance C _{DS of} the T1 and the tuning winding 4.

Fig. 4 shows a detailed embodiment according to the second embodiment of the invention with XOR gates and a barrier FET as a switch. Here, the control loop oscillates as a relaxation oscillator between overcompensation and undercompensation. With increasing damping, the acting as a switch JFET transistor T1 remains correspondingly longer conductive. The changing duty cycle serves as an output signal.

The LC-oscillator equipped with the XOR1 exclusive-OR gate (1/4 of the 74HC86) generates a sinusoidal signal whose frequency is determined by the inductance LS and the capacitances C1 and C2 connected in series with it.

For the frequency f, the relation holds: f = 1 / (2Pi *

LS * (C1 * C2) / (C1 + C2)). The transmitting coil LS is connected to the two receiving coils L1 and L2 via the transformer coupling factors M1 and M2. They form a differential transformer and are adjusted so that in the unloaded state, the voltages generated in the receiving coils L1 and L2 are the same.

The center tap of the two receiver coils is at half the operating voltage.

High frequency, it is grounded. In the case of a unipolar power supply, the comparator OV1 thus receives the optimum bias at its two inputs.

The other ends of the two anti-serially connected receiving coils are connected to the inputs of the comparator OV1. This compares the voltages induced in the two receiver coils L1 and L2. It works as a pulse shaper and converts smallest voltage differences into a square wave signal.

The comparator OV2 also operates as a pulse shaper and generates a phase-synchronous to the oscillator square wave signal.

Since the gate XOR2 works as a phase comparator, unequal input levels give an H at the output and the same input level an L. At a phase shift of 90 ° this results in a square wave signal with twice the frequency.

The gate XOR2 is supplied with both the oscillator signal and the received signal. When these signals are phase synchronous, the output of gate XOR2 goes to L. When shifted by 180 °, it goes to H.

The receiving coils L1 and L2 are connected so that in the attenuated state, an H appears at the output of the comparator OV1.

OV3 acts as an integrator. It integrates the incoming pulses until reaching the switching threshold of the following Schmitt trigger XOR3. With weak damping this takes longer than with heavy damping.

When the switch-on threshold of the Schmitt trigger XOR3 is reached, it opens the N-channel J-FET T1 of the type U310 and thus ensures a stronger damping of the receiving coil L2 as target at L1. The resulting overcompensation generates a phase jump at the input of OV1. Since the oscillator signal and the received signal are now in phase, an L. appears at the output of XOR2. The differential integrator is discharged and the Schmitt trigger XOR3 returns to its L level when the switch-off threshold is reached. Transistor T1 then turns off and the charging process starts from again.

As already mentioned above, the control loop oscillates as a relaxation oscillator. The integrator OV3 generates a sawtooth voltage whose frequency is determined by the RC element on the integrator OV3 and the size of the hysteresis of the Schmitt trigger XOR3. The duty cycle is a measure of the damping. Provided that L1 = L2, the transistor T1 will always be conductive until the product of its duty cycle with the resistance Re equals the symbolic damping resistor Rx.

The attenuation by the target represented by the resistor Rx is compensated only on average. The control signal for the transistor T1 can be output as a pulse width modulated square wave signal. A 74HC86 contains four XOR gates, three of which are used in the sensor part. OV1 and OV2 are housed in a common housing. This guarantees equal terms of both.

FIG. 5 shows an embodiment according to the second embodiment of the invention with type 74HC4053 analog multiplexers.

The sinusoidal transmission signal is generated by the two-channel analog multiplexer MUX1 as an inverter in an LC oscillator. For the frequency f, the above formula applies.

The square wave signal present at the multiplexer output MUX1 controls the multiplexer MUX2, which acts in conjunction with the difference integrator OV2 as a phase-sensitive rectifier. The ratios at the differential transformer correspond to those in FIG. 4. The operational amplifier OV1 is preferably a comparator.

As soon as the equilibrium is disturbed by reducing the resistance Rx connected to the symbolic target winding 5, the output of the differential integrator OV2 supplies a DC voltage. This is supplied to the Schmitt trigger OV3, which switches the multiplexer MUX3 when its switching threshold is exceeded. When the output of MUX3 is connected to ground, as described above, the differential transformer is deenergized via the trimming coil 4 with the resistor Re.

If the attenuation symbolically represented here by Rx is compensated by the target, the difference integrator OV2 no longer supplies a signal, the Schmitt trigger OV3 drops back and the multiplexer MUX3 opens again.

The control does not run continuously, but it is alternately over- or under-compensated. The balanced state is only reached on average.

The control signal for the multiplexer MUX3 is also output here as a pulse width modulated square wave signal. Since the three required multiplexers are contained in a single 74HC4053, they keep the component cost low. A single integrated circuit (IC) serves as an inverter in the oscillator, as a phase comparator in the rectifier and as a switch for clocking the damping resistor.

The invention relates to an inductive proximity switch 1 with a transmitting coil 2, two antiserially connected receiving coils 3 for generating a received signal and a trimming winding 4. They advantageously have a common core.

The transmitting coil 2 is either powered by a high frequency generator 6 or is itself part of an oscillator.

The two receiving coils 3 preferably have the same inductance L1 = L2. They form with the transmitting coil 2 a differential transformer. The arrangement is such that one of the two receiver coils 3 can be influenced more strongly by the target. Furthermore, the balancing winding 4 is arranged so that it is more strongly coupled to one of the two receiving coils 3.

The balancing winding 4 forms a closed circuit with the controllable resistor 9, so that it can be damped by this.

The electrically controllable resistor is part of a control loop, with which the received signal of the two anti-serially connected receiving coils 3 can be preferably controlled to zero.

The control loop has a synchronous rectifier 7, which is preferably designed as a differential comparator.

In a first embodiment, the circuit of the tuning winding 4 has a current measuring unit 10 for measuring the manipulated variable of the control loop.

In the second embodiment of the invention, the controllable resistor 9 has a switch with a resistor connected in series therewith.

The electrically controllable resistor 9 is also part of a control loop, with which the received signal of the two anti-serially connected receiving coils 3 can be preferably controlled to zero.

The control loop has a synchronous rectifier 7, which is preferably designed as a differential comparator. Furthermore, the control loop has a Schmitt trigger 11, which controls the switch in the controllable resistor 9. This results in a relaxation oscillator, which converts the analog control signal of the integrator into a pulse width modulated signal and can control the received signal on average preferably to zero. The measured variable is the pulse width modulated signal.

The invention further comprises a method for operating an inductive proximity switch according to claim 1. The method is characterized in that the current flowing in the balancing winding 4 and in the controllable resistor 9 is measured and evaluated.

The invention further comprises a method for operating an inductive proximity switch according to claim 2. It is characterized in that

the controllable resistor 9 is periodically switched on and off via a clock signal generated by a Schmitt trigger 11. The resulting relaxation oscillation preferably controls the received signal on average preferably to zero. The pulse width ratio of the relaxation oscillation represents the attenuation by the target and is output as a pulse width modulated measurement signal (PWM).

Induktiver Näherungsschalter
Sendespule
Empfangsspulen
Abgleichwicklung
Targetwicklung, symbolisiert die induktive Kopplung zum Target
Hochfrequenzgenerator, Oszillator
Synchrongleichrichter, Differenzkomparator zur Phasenmessung
Integrator, Differenzintegrator oder Tiefpass
Steuerbarer Widerstand, Stellglied, T1 oder Rref + Schalter
Strommesseinheit
Schmitt-Trigger

List of the most important reference numbers

Inductive proximity switch
transmitting coil
receiving coils
trim winding
Targetwicklung, symbolizes the inductive coupling to the target
High frequency generator, oscillator
Synchronous rectifier, differential comparator for phase measurement
Integrator, differential integrator or low-pass filter
Controllable resistor, actuator, T1 or Rref + switch
Current measuring unit
Schmitt trigger

Claims

Inductive proximity switch (1) having a transmitting coil (2), two antiserially connected receiving coils (3) for generating a receiving signal and a balancing winding (4), which is transformer coupled to one of the receiving coils (3), wherein the balancing winding (4) with a controllable resistor (9) is connected, which is part of a control loop, with which the received signal can be controlled to zero, characterized in that the circuit of the tuning coil (4) has a current measuring unit (10) for measuring the manipulated variable.
Inductive proximity switch (1) having a transmitting coil (2), two antiserially connected receiving coils (3) for generating a receiving signal and a balancing winding (4), which is transformer coupled to one of the receiving coils (3), wherein the balancing winding (4) with a controllable resistor (9), characterized in that the controllable resistor (9) consists of a switch and a resistor and is part of a control loop, which has a Schmitt trigger (11), the analog control signal of the integrator in a pulse width modulated Signal converts with which the switch is controlled and the received signal can be controlled on average to zero.
Method for operating an inductive proximity switch according to Claim 1, characterized in that the current flowing in the balancing winding (4) and in the controllable resistor (9) is measured and evaluated.
Method for operating an inductive proximity switch according to claim 2, characterized in that the controllable resistor (9) is periodically switched on and off via a signal generated by a Schmitt trigger (11), the received signal in this way is averaged to zero and the so resulting pulse width ratio is output as a measurement signal.